LINER LT1912 36v, 2a, 500khz step-down switching regulator Datasheet

LT1912
36V, 2A, 500kHz Step-Down
Switching Regulator
FEATURES
■
■
■
■
■
■
■
■
■
■
■
DESCRIPTION
Wide Input Range: Operation From 3.6V to 36V
2A Maximum Output Current
Adjustable Switching Frequency: 200kHz to 500kHz
Low Shutdown Current: IQ < 1µA
Integrated Boost Diode
Synchronizable Between 250kHz to 500kHz
Saturating Switch Design: 0.25Ω On-Resistance
0.790V Feedback Reference Voltage
Output Voltage: 0.79V to 20V
Soft-Start Capability
Small 10-Pin Thermally Enhanced MSOP and
(3mm × 3mm) DFN Packages
The LT®1912 is an adjustable frequency (200kHz to 500kHz)
monolithic step-down switching regulator that accepts
input voltages up to 36V. A high efficiency 0.25Ω switch
is included on the die along with a boost Schottky diode
and the necessary oscillator, control and logic circuitry.
Current mode topology is used for fast transient response
and good loop stability. The LT1912 allows the use of
ceramic capacitors resulting in low output ripple while
keeping total solution size to a minimum. The low current
shutdown mode reduces input supply current to less than
1µA while a resistor and capacitor on the RUN/SS pin
provide a controlled output voltage ramp (soft-start). The
LT1912 is available in 10-pin MSOP and 3mm × 3mm DFN
packages with exposed pads for low thermal resistance.
APPLICATIONS
■
■
■
■
■
L, LT, LTC, LTM, Burst Mode, Linear Technology and the Linear logo are registered trademarks
and ThinSOT is a trademark of Linear Technology Corporation. All other trademarks are the
property of their respective owners.
Automotive Battery Regulation
Set-Top Box
Distributed Supply Regulation
Industrial Supplies
Wall Transformer Regulation
TYPICAL APPLICATION
3.3V Step-Down Converter
Efficiency
VIN
4.5V TO 36V
VIN
20k
4.7µF
BD
RUN/SS
BOOST
0.47µF
VC
LT1912
6.8µH
SW
RT
470pF
68.1k
SYNC
VOUT = 5V
90
EFFICIENCY (%)
OFF ON
100
VOUT
3.3V
2A
GND
47µF
100k
70
60
316k
FB
VOUT = 3.3V
80
VIN = 12V
L = 6.8µF
F = 500kHz
50
0
1912 TA01
0.5
1.0
1.5
LOAD CURRENT (A)
2
1912 TA01b
1912fa
1
LT1912
ABSOLUTE MAXIMUM RATINGS
(Note 1)
VIN, RUN/SS Voltage..................................................36V
BOOST Pin Voltage....................................................56V
BOOST Pin Above SW Pin..........................................30V
FB, RT, VC Voltage........................................................5V
BD, SYNC Voltage......................................................30V
Operating Junction Temperature Range (Note 2)
LT1912E............................................. –40°C to 125°C
Storage Temperature Range................... –65°C to 150°C
Lead Temperature (Soldering, 10 sec)
MSE Only........................................................... 300°C
PIN CONFIGURATION
TOP VIEW
BD
1
BOOST
2
SW
3
VIN
4
RUN/SS
5
TOP VIEW
10 RT
11
GND
BD
BOOST
SW
VIN
RUN/SS
9 VC
8 FB
7 N/C
6 SYNC
DD PACKAGE
10-LEAD (3mm × 3mm) PLASTIC DFN
JA = 45°C/W, JC = 10°C/W
EXPOSED PAD (PIN 11) IS GND, MUST BE SOLDERED TO PCB
1
2
3
4
5
11
GND
10
9
8
7
6
RT
VC
FB
N/C
SYNC
MSE PACKAGE
10-LEAD PLASTIC MSOP
JA = 45°C/W, JC = 10°C/W
EXPOSED PAD (PIN 11) IS GND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT1912EDD#PBF
LT1912EMSE#PBF
LT1912EDD#TRPBF
LT1912EMSE#TRPBF
LDJT
LTDJS
10-Lead (3mm × 3mm) Plastic DFN
10-Lead Plastic MSOP
–40°C to 125°C
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 10V, VRUN/SS = 10V, VBOOST = 15V, VBD = 3.3V unless otherwise
noted. (Note 2)
PARAMETER
CONDITIONS
TYP
MAX
l
3
3.6
V
VRUN/SS = 0.2V
VBD = 3V, Not Switching
VBD = 0, Not Switching
l
0.01
450
1.3
0.5
600
1.7
µA
µA
mA
VRUN/SS = 0.2V
VBD = 3V, Not Switching
VBD = 0, Not Switching
l
0.01
0.9
1
0.5
1.3
5
µA
mA
µA
2.7
3
Minimum Input Voltage
Quiescent Current from VIN
Quiescent Current from BD
Minimum Bias Voltage (BD Pin)
MIN
UNITS
V
1912fa
2
LT1912
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 10V, VRUN/SS = 10V VBOOST = 15V, VBD = 3.3V unless otherwise
noted. (Note 2)
PARAMETER
CONDITIONS
Feedback Voltage
l
FB Pin Bias Current (Note 3)
VFB = 0.8V, VC = 0.4V
FB Voltage Line Regulation
4V < VIN < 36V
MIN
TYP
MAX
UNITS
780
775
790
790
800
805
mV
mV
7
30
nA
0.002
0.01
%/V
l
25
Error Amp gm
Error Amp Gain
µMho
1000
VC Source Current
45
µA
VC Sink Current
45
µA
VC Pin to Switch Current Gain
3.5
A/V
VC Clamp Voltage
Switching Frequency
2
RT = 187k
Minimum Switch Off-Time
160
l
Switch Current Limit
Duty Cycle = 5%
Switch VCESAT
ISW = 2A
Boost Schottky Reverse Leakage
VSW = 10V, VBD = 0V
3.2
V
200
240
kHz
60
150
nS
3.7
4.2
500
A
mV
0.02
2
µA
1.5
2.1
V
BOOST Pin Current
ISW = 1A
22
35
mA
RUN/SS Pin Current
VRUN/SS = 2.5V
5
10
µA
2.5
V
Minimum Boost Voltage (Note 4)
l
RUN/SS Input Voltage High
RUN/SS Input Voltage Low
0.2
SYNC Low Threshold
0.5
V
V
SYNC High Threshold
SYNC Pin Bias Current
0.7
VSYNC = 0V
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LT1912E is guaranteed to meet performance specifications
from 0°C to 125°C. Specifications over the –40°C to 125°C operating
0.1
V
µA
temperature range are assured by design, characterization and correlation
with statistical process controls.
Note 3: Bias current flows out of the FB pin.
Note 4: This is the minimum voltage across the boost capacitor needed to
guarantee full saturation of the switch.
1912fa
3
LT1912
TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency
VIN = 12V
VIN = 12V
3.5
80
VIN = 34V
80
VIN = 24V
70
4.0
VIN = 7V
85
EFFICIENCY (%)
EFFICIENCY (%)
Maximum Load Current
90
VIN = 34V
75
VIN = 24V
70
65
60
60
0
55
L: NEC PLC-0745-5R6
f: 500kHz
VOUT = 5V
50
VOUT = 3.3V
50
0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0
LOAD CURRENT (A)
0
SWITCH CURRENT LIMIT(A)
LOAD CURRENT (A)
2.5
MINIMUM
2.0
VOUT = 5V
L = 4.7µH
f = 500kHz
20
15
INPUT VOLTAGE (V)
25
30
10
20
15
INPUT VOLTAGE (V)
4.0
3.5
3.0
2.5
2.0
1.5
1.0
25
30
Switch Current Limit
DUTY CYCLE = 10 %
3.5
3.0
DUTY CYCLE = 90 %
2.5
2.0
1.5
1.0
0.5
20
0
80
60
40
DUTY CYCLE (%)
100
0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
1912 G06
1912 G05
Boost Pin Current
Switch Voltage Drop
700
80
600
70
BOOST PIN CURRENT (mA)
VOLTAGE DROP (mV)
5
4.5
1912 G04
500
400
300
200
100
0
VOUT = 3.3V
L = 4.7µH
f = 500kHz
1912 G03
SWITCH CURRENT LIMIT (A)
TYPICAL
10
MINIMUM
2.0
1.0
0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0
LOAD CURRENT (A)
4.0
5
2.5
Switch Current Limit
Maximum Load Current
1.5
3.0
1912 G02
3.5
3.0
TYPICAL
1.5
L: NEC PLC-0745-5R6
f: 500kHz
1912 G01
1.0
LOAD CURRENT (A)
Efficiency
100
90
TA = 25°C, unless otherwise noted.
60
50
40
30
20
10
0
500
1500
1000
2000
SWITCH CURRENT (mA)
2500
1912 G07
0
0
500
1000
1500
2000
SWITCH CURRENT (mA)
2500
1912 G08
1912fa
4
LT1912
TYPICAL PERFORMANCE CHARACTERISTICS
TA = 25°C, unless otherwise noted.
Switching Frequency
Feedback Voltage
Frequency Foldback
1.2
1.20
SWITCHING FREQUENCY (NORMALIZED)
840
FREQUENCY (NORMALIZED)
FEEDBACK VOLTAGE (mV)
1.15
820
800
780
1.10
1.05
1.00
0.95
0.90
0.85
760
–50 –25
0
0.80
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
0
0.2
100
80
60
40
20
100 200 300 400 500 600 700 800 900
FB PIN VOLTAGE (mV)
RUN/SS Pin Current
12
3.0
2.5
2.0
1.5
1.0
0
10
8
6
4
2
0.5
25 50 75 100 125 150
TEMPERATURE (˚C)
0
1912 G11
RUN/SS PIN CURRENT (µA)
3.5
SWITCH CURRENT LIMIT (A)
120
0
2.5
2
1.5
RUN/SS PIN VOLTAGE (V)
0.5
1
3
3.5
0
0
5
20
30
15
25
10
RUN/SS PIN VOLTAGE (V)
1912 G13
1912 G12
35
1912 G14
Error Amp Output Current
Boost Diode
1.4
50
40
1.2
30
1.0
VC PIN CURRENT (µA)
BOOST DIODE Vf (V)
MINIMUM SWITCH ON TIME (ns)
4.0
0.8
0.6
0.4
20
10
0
–10
–20
–30
0.2
0
0.4
Soft-Start
Minimum Switch On-Time
0
0.6
1912 G10
140
0
–50 –25
0.8
0
25 50 75 100 125 150
TEMPERATURE (°C)
1912 G09
1.0
–40
0
0.5
1.0
1.5
BOOST DIODE CURRENT (A)
2.0
1912 G15
–50
–200
–100
100
0
FB PIN ERROR VOLTAGE (V)
200
1912 G16
1912fa
5
LT1912
TYPICAL PERFORMANCE CHARACTERISTICS
2.50
Switching Waveforms;
Continuous Operation
Switching Waveforms;
Discontinuous Operation
VC Voltages
VSW
5V/DIV
VSW
5V/DIV
2.00
CURRENT LIMIT CLAMP
VC VOLTAGE (V)
TA = 25°C unless otherwise noted.
1.50
1.00
IL
1A/DIV
VOUT
10mV/DIV
VOUT
10mV/DIV
SWITCHING THRESHOLD
0.50
0
–50 –25
IL
0.5A/DIV
0
25 50 75 100 125 150
TEMPERATURE (°C)
1912 G19
2µs/DIV
VIN = 12V; VOUT = 3.3V
ILOAD = 110mA
1912 G21
2µs/DIV
VIN = 12V; VOUT = 3.3V
ILOAD = 1A
1912 G22
1912fa
6
LT1912
PIN FUNCTIONS
BD (Pin 1): This pin connects to the anode of the boost
Schottky diode. BD also supplies current to the internal
regulator.
BOOST (Pin 2): This pin is used to provide a drive
voltage, higher than the input voltage, to the internal bipolar
NPN power switch.
SW (Pin 3): The SW pin is the output of the internal power
switch. Connect this pin to the inductor, catch diode and
boost capacitor.
VIN (Pin 4): The VIN pin supplies current to the LT1912’s
internal regulator and to the internal power switch. This
pin must be locally bypassed.
RUN/SS (Pin 5): The RUN/SS pin is used to put the
LT1912 in shutdown mode. Tie to ground to shut down
the LT1912. Tie to 2.5V or more for normal operation. If
the shutdown feature is not used, tie this pin to the VIN
pin. RUN/SS also provides a soft-start function; see the
Applications Information section.
SYNC (Pin 6): This is the external clock synchronization
input. Ground this pin when SYNC function is not used. Tie
to a clock source for synchronization. Clock edges should
have rise and fall times faster than 1µs. See synchronizing
section in Applications Information.
N/C (Pin 7): This pin should be tied to ground.
FB (Pin 8): The LT1912 regulates the FB pin to 0.790V.
Connect the feedback resistor divider tap to this pin.
VC (Pin 9): The VC pin is the output of the internal error
amplifier. The voltage on this pin controls the peak switch
current. Tie an RC network from this pin to ground to
compensate the control loop.
RT (Pin 10): Oscillator Resistor Input. Connecting a resistor to ground from this pin sets the switching frequency.
GND (Exposed Pad Pin 11): Ground. The exposed pad
must be soldered to PCB.
BLOCK DIAGRAM
VIN
4
VIN
C1
–
+
INTERNAL 0.79V REF
5
10
RUN/SS
∑
RT
OSCILLATOR
200kHz–500kHz
RT
6
SLOPE COMP
BD
SWITCH
LATCH
BOOST
1
2
R
S
C3
Q
SW
SYNC
L1
VOUT
3
D1
C2
SOFT-START
ERROR AMP
+
–
VC CLAMP
VC
9
CC
RC
GND
11
CF
FB
8
R2
R1
1912 BD
1912fa
7
LT1912
OPERATION
The LT1912 is a constant frequency, current mode stepdown regulator. An oscillator, with frequency set by RT,
enables an RS flip-flop, turning on the internal power
switch. An amplifier and comparator monitor the current
flowing between the VIN and SW pins, turning the switch
off when this current reaches a level determined by the
voltage at VC. An error amplifier measures the output
voltage through an external resistor divider tied to the FB
pin and servos the VC pin. If the error amplifier’s output
increases, more current is delivered to the output; if it
decreases, less current is delivered. An active clamp on the
VC pin provides current limit. The VC pin is also clamped to
the voltage on the RUN/SS pin; soft-start is implemented
by generating a voltage ramp at the RUN/SS pin using an
external resistor and capacitor.
An internal regulator provides power to the control circuitry.
The bias regulator normally draws power from the VIN pin,
but if the BD pin is connected to an external voltage higher
than 3V bias power will be drawn from the external source
(typically the regulated output voltage). This improves
efficiency. The RUN/SS pin is used to place the LT1912
in shutdown, disconnecting the output and reducing the
input current to less than 1µA.
The switch driver operates from either the input or from
the BOOST pin. An external capacitor and diode are used
to generate a voltage at the BOOST pin that is higher than
the input supply. This allows the driver to fully saturate the
internal bipolar NPN power switch for efficient operation.
The oscillator reduces the LT1912’s operating frequency
when the voltage at the FB pin is low. This frequency
foldback helps to control the output current during startup and overload.
1912fa
8
LT1912
APPLICATIONS INFORMATION
FB Resistor Network
The output voltage is programmed with a resistor divider
between the output and the FB pin. Choose the 1% resistors according to:
⎛ V
⎞
R1= R2 ⎜ OUT – 1⎟
⎝ 0.79V ⎠
DCMIN = fSW tON(MIN )
Reference designators refer to the Block Diagram.
Setting the Switching Frequency
The LT1912 uses a constant frequency PWM architecture
that can be programmed to switch from 200kHz to 500kHz
by using a resistor tied from the RT pin to ground. A table
showing the necessary RT value for a desired switching
frequency is in Figure 1.
SWITCHING FREQUENCY (kHz)
RT VALUE (kΩ)
200
300
400
500
187
121
88.7
68.1
Figure 1. Switching Frequency vs RT Value
Operating Frequency Trade-Offs
Selection of the operating frequency is a trade-off between
efficiency, component size, minimum dropout voltage, and
maximum input voltage. The advantage of high frequency
operation is that smaller inductor and capacitor values may
be used. The disadvantages are lower efficiency, lower
maximum input voltage, and higher dropout voltage. The
highest acceptable switching frequency (fSW(MAX)) for a
given application can be calculated as follows:
fSW (MAX ) =
switching frequency is because the LT1912 switch has
finite minimum on and off times. The switch can turn on
for a minimum of ~150ns and turn off for a minimum of
~150ns. Typical minimum on time at 25°C is 80ns. This
means that the minimum and maximum duty cycles are:
VD + VOUT
tON(MIN ) ( VD + VIN – VSW )
where VIN is the typical input voltage, VOUT is the output
voltage, VD is the catch diode drop (~0.5V) and VSW is the
internal switch drop (~0.5V at max load). This equation
shows that slower switching frequency is necessary to
safely accommodate high VIN/VOUT ratio. Also, as shown
in the next section, lower frequency allows a lower dropout
voltage. The reason input voltage range depends on the
DCMAX = 1– fSW tOFF (MIN )
where fSW is the switching frequency, the tON(MIN) is the
minimum switch on time (~150ns), and the tOFF(MIN) is
the minimum switch off time (~150ns). These equations
show that duty cycle range increases when switching
frequency is decreased.
A good choice of switching frequency should allow adequate input voltage range (see next section) and keep
the inductor and capacitor values small.
Input Voltage Range
The maximum input voltage for LT1912 applications
depends on switching frequency, the Absolute Maximum
Ratings of the VIN and BOOST pins, and the operating mode.
While the output is in start-up, short-circuit, or other
overload conditions, the switching frequency should be
chosen according to the following equation.
VIN(MIN ) =
VOUT + VD
–V +V
1– fSW tOFF (MIN ) D SW
where VIN(MAX) is the maximum operating input voltage,
VOUT is the output voltage, VD is the catch diode drop
(~0.5V), VSW is the internal switch drop (~0.5V at max
load), fSW is the switching frequency (set by RT), and
tON(MIN) is the minimum switch on time (~150ns). Note that
a higher switching frequency will depress the maximum
operating input voltage. Conversely, a lower switching
frequency will be necessary to achieve safe operation at
high input voltages.
If the output is in regulation and no short-circuit, startup, or overload events are expected, then input voltage
transients of up to 36V are acceptable regardless of the
switching frequency. In this mode, the LT1912 may enter
1912fa
9
LT1912
APPLICATIONS INFORMATION
pulse-skipping operation where some switching pulses
are skipped to maintain output regulation. In this mode
the output voltage ripple and inductor current ripple will
be higher than in normal operation.
The minimum input voltage is determined by either the
LT1912’s minimum operating voltage of ~3.6V or by its
maximum duty cycle (see equation in previous section).
The minimum input voltage due to duty cycle is:
V +V
VIN(MAX ) = OUT D – VD + VSW
fSW tON(MIN )
where VIN(MIN) is the minimum input voltage, and tOFF(MIN)
is the minimum switch off time (150ns). Note that higher
switching frequency will increase the minimum input
voltage. If a lower dropout voltage is desired, a lower
switching frequency should be used.
Inductor Selection
For a given input and output voltage, the inductor value
and switching frequency will determine the ripple current.
The ripple current ΔIL increases with higher VIN or VOUT
and decreases with higher inductance and faster switching frequency. A reasonable starting point for selecting
the ripple current is:
ΔIL = 0.4(IOUT(MAX))
where IOUT(MAX) is the maximum output load current. To
guarantee sufficient output current, peak inductor current
must be lower than the LT1912’s switch current limit (ILIM).
The peak inductor current is:
IL(PEAK) = IOUT(MAX) + ΔIL/2
where IL(PEAK) is the peak inductor current, IOUT(MAX) is
the maximum output load current, and ΔIL is the inductor
ripple current. The LT1912’s switch current limit (ILIM) is
at least 3.5A at low duty cycles and decreases linearly to
2.5A at DC = 0.8. The maximum output current is a function of the inductor ripple current:
IOUT(MAX) = ILIM – ΔIL/2
Be sure to pick an inductor ripple current that provides
sufficient maximum output current (IOUT(MAX)).
The largest inductor ripple current occurs at the highest
VIN. To guarantee that the ripple current stays below the
specified maximum, the inductor value should be chosen
according to the following equation:
⎛V +V ⎞⎛ V +V ⎞
L = ⎜ OUT D ⎟ ⎜ 1– OUT D ⎟
VIN(MAX) ⎠
⎝ fSW ΔIL ⎠ ⎝
where VD is the voltage drop of the catch diode (~0.4V),
VIN(MAX) is the maximum input voltage, VOUT is the output
voltage, fSW is the switching frequency (set by RT), and L
is in the inductor value.
The inductor’s RMS current rating must be greater than the
maximum load current and its saturation current should be
about 30% higher. For robust operation in fault conditions
(start-up or short circuit) and high input voltage (>30V),
the saturation current should be above 3.5A. To keep the
efficiency high, the series resistance (DCR) should be less
than 0.1Ω, and the core material should be intended for
high frequency applications. Table 1 lists several vendors
and suitable types.
Table 1. Inductor Vendors
VENDOR
URL
PART SERIES
TYPE
Murata
www.murata.com
LQH55D
Open
TDK
www.componenttdk.com
SLF7045
SLF10145
Shielded
Shielded
Toko
www.toko.com
D62CB
Shielded
D63CB
Shielded
D75C
Shielded
D75F
Open
CR54
Open
CDRH74
Shielded
CDRH6D38
Shielded
CR75
Open
Sumida
www.sumida.com
Of course, such a simple design guide will not always result in the optimum inductor for your application. A larger
value inductor provides a slightly higher maximum load
current and will reduce the output voltage ripple. If your
load is lower than 2A, then you can decrease the value of
the inductor and operate with higher ripple current. This
allows you to use a physically smaller inductor, or one
1912fa
10
LT1912
APPLICATIONS INFORMATION
with a lower DCR resulting in higher efficiency. There are
several graphs in the Typical Performance Characteristics
section of this data sheet that show the maximum load
current as a function of input voltage and inductor value
for several popular output voltages. Low inductance may
result in discontinuous mode operation, which is okay
but further reduces maximum load current. For details of
maximum output current and discontinuous mode operation, see Linear Technology Application Note 44. Finally,
for duty cycles greater than 50% (VOUT/VIN > 0.5), there
is a minimum inductance required to avoid subharmonic
oscillations. See AN19.
Input Capacitor
Bypass the input of the LT1912 circuit with a ceramic capacitor of X7R or X5R type. Y5V types have poor performance
over temperature and applied voltage, and should not be
used. A 4.7µF to 10µF ceramic capacitor is adequate to
bypass the LT1912 and will easily handle the ripple current.
Note that larger input capacitance is required when a lower
switching frequency is used. If the input power source has
high impedance, or there is significant inductance due to
long wires or cables, additional bulk capacitance may be
necessary. This can be provided with a lower performance
electrolytic capacitor.
Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input
capacitor is required to reduce the resulting voltage
ripple at the LT1912 and to force this very high frequency
switching current into a tight local loop, minimizing EMI.
A 4.7µF capacitor is capable of this task, but only if it is
placed close to the LT1912 and the catch diode (see the
PCB Layout section). A second precaution regarding the
ceramic input capacitor concerns the maximum input
voltage rating of the LT1912. A ceramic input capacitor
combined with trace or cable inductance forms a high
quality (under damped) tank circuit. If the LT1912 circuit
is plugged into a live supply, the input voltage can ring to
twice its nominal value, possibly exceeding the LT1912’s
voltage rating. This situation is easily avoided (see the Hot
Plugging Safely section).
Output Capacitor and Output Ripple
The output capacitor has two essential functions. Along
with the inductor, it filters the square wave generated by the
LT1912 to produce the DC output. In this role it determines
the output ripple, and low impedance at the switching
frequency is important. The second function is to store
energy in order to satisfy transient loads and stabilize the
LT1912’s control loop. Ceramic capacitors have very low
equivalent series resistance (ESR) and provide the best
ripple performance. A good starting value is:
COUT =
100
VOUT fSW
where fSW is in MHz, and COUT is the recommended output
capacitance in µF. Use X5R or X7R types. This choice will
provide low output ripple and good transient response.
Transient performance can be improved with a higher value
capacitor if the compensation network is also adjusted
to maintain the loop bandwidth. A lower value of output
capacitor can be used to save space and cost but transient
performance will suffer. See the Frequency Compensation
section to choose an appropriate compensation network.
When choosing a capacitor, look carefully through the
data sheet to find out what the actual capacitance is under
operating conditions (applied voltage and temperature).
A physically larger capacitor, or one with a higher voltage
rating, may be required. High performance tantalum or
electrolytic capacitors can be used for the output capacitor.
Low ESR is important, so choose one that is intended for
use in switching regulators. The ESR should be specified by
the supplier, and should be 0.05Ω or less. Such a capacitor will be larger than a ceramic capacitor and will have a
larger capacitance, because the capacitor must be large to
achieve low ESR. Table 2 lists several capacitor vendors.
1912fa
11
LT1912
APPLICATIONS INFORMATION
Table 2. Capacitor Vendors
VENDOR
PHONE
URL
PART SERIES
Panasonic
(714) 373-7366
www.panasonic.com
Ceramic,
Polymer,
COMMANDS
EEF Series
Tantalum
Kemet
(864) 963-6300
www.kemet.com
Ceramic,
Tantalum
Sanyo
(408) 749-9714
www.sanyovideo.com
T494, T495
Ceramic,
Polymer,
POSCAP
Tantalum
Murata
(408) 436-1300
AVX
www.murata.com
Ceramic
www.avxcorp.com
Ceramic,
Tantalum
Taiyo Yuden
(864) 963-6300
www.taiyo-yuden.com
TPS Series
Ceramic
Catch Diode
Ceramic Capacitors
The catch diode conducts current only during switch off
time. Average forward current in normal operation can
be calculated from:
A precaution regarding ceramic capacitors concerns the
maximum input voltage rating of the LT1912. A ceramic
input capacitor combined with trace or cable inductance
forms a high quality (under damped) tank circuit. If the
LT1912 circuit is plugged into a live supply, the input voltage can ring to twice its nominal value, possibly exceeding
the LT1912’s rating. This situation is easily avoided (see
the Hot Plugging Safely section).
ID(AVG) = IOUT (VIN – VOUT)/VIN
where IOUT is the output load current. The only reason to
consider a diode with a larger current rating than necessary for nominal operation is for the worst-case condition
of shorted output. The diode current will then increase
to the typical peak switch current. Peak reverse voltage
is equal to the regulator input voltage. Use a Schottky
diode with a reverse voltage rating greater than the input
voltage. Table 3 lists several Schottky diodes and their
manufacturers.
Table 3. Diode Vendors
PART NUMBER
VR
(V)
IAVE
(A)
VF AT 1A
(mV)
VF AT 2A
(mV)
On Semicnductor
MBRM120E
MBRM140
20
40
1
1
530
550
595
Diodes Inc.
B220
B230
DFLS240L
20
30
40
2
2
2
International Rectifier
10BQ030
20BQ030
30
30
1
2
500
500
500
420
470
470
Frequency Compensation
The LT1912 uses current mode control to regulate the
output. This simplifies loop compensation. In particular, the
LT1912 does not require the ESR of the output capacitor
for stability, so you are free to use ceramic capacitors to
achieve low output ripple and small circuit size. Frequency
compensation is provided by the components tied to the
VC pin, as shown in Figure 2. Generally a capacitor (CC)
and a resistor (RC) in series to ground are used. In addition, there may be lower value capacitor in parallel. This
capacitor (CF) is not part of the loop compensation but
is used to filter noise at the switching frequency, and is
required only if a phase-lead capacitor is used or if the
output capacitor has high ESR.
Loop compensation determines the stability and transient
performance. Designing the compensation network is a bit
complicated and the best values depend on the application
1912fa
12
LT1912
APPLICATIONS INFORMATION
Capacitor C3 and the internal boost Schottky diode (see
the Block Diagram) are used to generate a boost voltage that is higher than the input voltage. In most cases
a 0.22µF capacitor will work well. Figure 2 shows three
ways to arrange the boost circuit. The BOOST pin must be
more than 2.3V above the SW pin for best efficiency. For
outputs of 3V and above, the standard circuit (Figure 4a)
is best. For outputs between 2.8V and 3V, use a 1µF boost
capacitor. A 2.5V output presents a special case because it
is marginally adequate to support the boosted drive stage
while using the internal boost diode. For reliable BOOST pin
operation with 2.5V outputs use a good external Schottky
diode (such as the ON Semi MBR0540), and a 1µF boost
capacitor (see Figure 4b). For lower output voltages the
boost diode can be tied to the input (Figure 4c), or to
another supply greater than 2.8V. Tying BD to VIN reduces
The minimum operating voltage of an LT1912 application
is limited by the minimum input voltage (3.6V) and by the
maximum duty cycle as outlined in a previous section. For
proper startup, the minimum input voltage is also limited
by the boost circuit. If the input voltage is ramped slowly,
then the boost capacitor may not be fully charged. Because
LT1912
CURRENT MODE
POWER STAGE
gm = 3.5mho
SW
OUTPUT
ERROR
AMPLIFIER
R1
CPL
FB
gm =
420µmho
3Meg
VC
CF
+
BOOST and BIAS Pin Considerations
the maximum input voltage to 30V. The circuit in Figure 4a
is more efficient because the BOOST pin current and BD
pin quiescent current comes from a lower voltage source.
You must also be sure that the maximum voltage ratings
of the BOOST and BD pins are not exceeded.
–
and in particular the type of output capacitor. A practical
approach is to start with one of the circuits in this data
sheet that is similar to your application and tune the compensation network to optimize the performance. Stability
should then be checked across all operating conditions,
including load current, input voltage and temperature. The
LT1375 data sheet contains a more thorough discussion of
loop compensation and describes how to test the stability using a transient load. Figure 2 shows an equivalent
circuit for the LT1912 control loop. The error amplifier is a
transconductance amplifier with finite output impedance.
The power section, consisting of the modulator, power
switch and inductor, is modeled as a transconductance
amplifier generating an output current proportional to
the voltage at the VC pin. Note that the output capacitor
integrates this current, and that the capacitor on the VC pin
(CC) integrates the error amplifier output current, resulting
in two poles in the loop. In most cases a zero is required
and comes from either the output capacitor ESR or from
a resistor RC in series with CC. This simple model works
well as long as the value of the inductor is not too high
and the loop crossover frequency is much lower than the
switching frequency. A phase lead capacitor (CPL) across
the feedback divider may improve the transient response.
Figure 3 shows the transient response when the load current
is stepped from 500mA to 1500mA and back to 500mA.
ESR
0.8V
C1
POLYMER
OR
TANTALUM
GND
RC
C1
+
CERAMIC
R2
CC
1912 F02
Figure 2. Model for Loop Response
VOUT
100mV/DIV
IL
0.5A/DIV
VIN = 12V; FRONT PAGE APPLICATION
10µs/DIV
1912 F03
Figure 3. Transient Load Response of the LT1912 Front Page
Application as the Load Current is Stepped from 500mA to
1500mA. VOUT = 3.3V
1912fa
13
LT1912
APPLICATIONS INFORMATION
the boost capacitor is charged with the energy stored in
the inductor, the circuit will rely on some minimum load
current to get the boost circuit running properly. This
minimum load will depend on input and output voltages, and on the arrangement of the boost circuit. The
minimum load generally goes to zero once the circuit has
started. Figure 5 shows a plot of minimum load to start
and to run as a function of input voltage. In many cases
the discharged output capacitor will present a load to the
switcher, which will allow it to start. The plots show the
worst-case situation where VIN is ramping very slowly.
For lower start-up voltage, the boost diode can be tied to
VIN; however, this restricts the input range to one-half of
the absolute maximum rating of the BOOST pin.
At light loads, the inductor current becomes discontinuous and the effective duty cycle can be very high. This
reduces the minimum input voltage to approximately
300mV above VOUT. At higher load currents, the inductor
current is continuous and the duty cycle is limited by the
maximum duty cycle of the LT1912, requiring a higher
input voltage to maintain regulation.
Soft-Start
The RUN/SS pin can be used to soft-start the LT1912,
reducing the maximum input current during start-up.
The RUN/SS pin is driven through an external RC filter to
create a voltage ramp at this pin. Figure 6 shows the startup and shutdown waveforms with the soft-start circuit.
By choosing a large RC time constant, the peak start-up
current can be reduced to the current that is required to
regulate the output, with no overshoot. Choose the value
of the resistor so that it can supply 20µA when the RUN/
SS pin reaches 2.5V.
Synchronization
Synchronizing the LT1912 oscillator to an external frequency can be done by connecting a square wave (with
20% to 80% duty cycle) to the SYNC pin. The square
wave amplitude should have valleys that are below 0.3V
and peaks that are above 0.8V (up to 6V).
The LT1912 may be synchronized over a 250kHz to 500kHz
range. The RT resistor should be chosen to set the LT1912
switching frequency 20% below the lowest synchronization
VOUT
BD
BOOST
VIN
VIN
LT1912
GND
4.7µF
C3
SW
(4a) For VOUT > 2.8V
VOUT
D2
BD
BOOST
VIN
VIN
LT1912
GND
4.7µF
C3
SW
(4b) For 2.5V < VOUT < 2.8V
VOUT
BD
BOOST
VIN
4.7µF
VIN
LT1912
GND
C3
SW
1912 FO4
(4c) For VOUT < 2.5V; VIN(MAX) = 30V
Figure 4. Three Circuits For Generating The Boost Voltage
input. For example, if the synchronization signal will be
250kHz and higher, the RT should be chosen for 200kHz.
To assure reliable and safe operation the LT1912 will only
synchronize when the output voltage is near regulation. It
is therefore necessary to choose a large enough inductor
value to supply the required output current at the frequency
set by the RT resistor. See Inductor Selection section. It
is also important to note that slope compensation is set
by the RT value: When the sync frequency is much higher
than the one set by RT, the slope compensation will be
significantly reduced which may require a larger inductor
value to prevent subharmonic oscillation.
1912fa
14
LT1912
APPLICATIONS INFORMATION
6.0
INPUT VOLTAGE (V)
5.5
TO START
(WORST CASE)
5.0
4.5
15k
4.0
TO RUN
3.5
3.0
0.22µF
RUN/SS
VRUN/SS
2V/DIV
GND
VOUT = 3.3V
TA = 25°C
L = 8.2µH
f = 500kHz
2.5
2.0
10
1
VOUT
2V/DIV
100
1000
LOAD CURRENT (A)
2ms/DIV
10000
TO START
(WORST CASE)
7.0
D4
MBRS140
6.0
VIN
5.0
1912 F06
Figure 6. To Soft-Start the LT1912, Add a Resisitor
and Capacitor to the RUN/SS Pin
8.0
INPUT VOLTAGE (V)
IL
1A/DIV
RUN
VIN
BOOST
LT1912
TO RUN
RUN/SS
VOUT
SW
VC
4.0
GND FB
VOUT = 5V
TA = 25°C
L = 8.2µH
f = 500kHz
3.0
2.0
1
10
BACKUP
100
1000
LOAD CURRENT (A)
10000
1912 F07
1912 F05
Figure 5. The Minimum Input Voltage Depends on
Output Voltage, Load Current and Boost Circuit
Shorted and Reversed Input Protection
If the inductor is chosen so that it won’t saturate excessively, an LT1912 buck regulator will tolerate a shorted
output. There is another situation to consider in systems
where the output will be held high when the input to the
LT1912 is absent. This may occur in battery charging applications or in battery backup systems where a battery
or some other supply is diode OR-ed with the LT1912’s
output. If the VIN pin is allowed to float and the RUN/SS
pin is held high (either by a logic signal or because it is
tied to VIN), then the LT1912’s internal circuitry will pull
its quiescent current through its SW pin. This is fine if
your system can tolerate a few mA in this state. If you
ground the RUN/SS pin, the SW pin current will drop to
essentially zero. However, if the VIN pin is grounded while
the output is held high, then parasitic diodes inside the
LT1912 can pull large currents from the output through
Figure 7. Diode D4 Prevents a Shorted Input from
Discharging a Backup Battery Tied to the Output. It Also
Protects the Circuit from a Reversed Input. The LT1912
Runs Only When the Input is Present
the SW pin and the VIN pin. Figure 7 shows a circuit that
will run only when the input voltage is present and that
protects against a shorted or reversed input.
PCB Layout
For proper operation and minimum EMI, care must be
taken during printed circuit board layout. Figure 8 shows
the recommended component placement with trace,
ground plane and via locations. Note that large, switched
currents flow in the LT1912’s VIN and SW pins, the catch
diode (D1) and the input capacitor (C1). The loop formed
by these components should be as small as possible. These
components, along with the inductor and output capacitor,
should be placed on the same side of the circuit board,
and their connections should be made on that layer. Place
a local, unbroken ground plane below these components.
1912fa
15
LT1912
APPLICATIONS INFORMATION
The SW and BOOST nodes should be as small as possible.
Finally, keep the FB and VC nodes small so that the ground
traces will shield them from the SW and BOOST nodes.
The exposed pad on the bottom of the package must be
soldered to ground so that the pad acts as a heat sink. To
keep thermal resistance low, extend the ground plane as
much as possible, and add thermal vias under and near
the LT1912 to additional ground planes within the circuit
board and on the bottom side.
L1
C2
VOUT
CC
RRT
RC
Hot Plugging Safely
The small size, robustness and low impedance of ceramic
capacitors make them an attractive option for the input
bypass capacitor of LT1912 circuits. However, these capacitors can cause problems if the LT1912 is plugged into a
live supply (see Linear Technology Application Note 88 for
a complete discussion). The low loss ceramic capacitor,
combined with stray inductance in series with the power
source, forms an under damped tank circuit, and the
voltage at the VIN pin of the LT1912 can ring to twice the
nominal input voltage, possibly exceeding the LT1912’s
rating and damaging the part. If the input supply is poorly
controlled or the user will be plugging the LT1912 into an
energized supply, the input network should be designed
to prevent this overshoot. Figure 9 shows the waveforms
that result when an LT1912 circuit is connected to a 24V
supply through six feet of 24-gauge twisted pair. The
first plot is the response with a 4.7µF ceramic capacitor
at the input. The input voltage rings as high as 50V and
the input current peaks at 26A. A good solution is shown
in Figure 9b. A 0.7Ω resistor is added in series with the
input to eliminate the voltage overshoot (it also reduces
the peak input current). A 0.1µF capacitor improves high
frequency filtering. For high input voltages its impact on
efficiency is minor, reducing efficiency by 1.5 percent for
a 5V output at full load operating from 24V.
R2
R1
D1
C1
GND
1912 F08
VIAS TO LOCAL GROUND PLANE
VIAS TO VOUT
VIAS TO SYNC
VIAS TO RUN/SS
VIAS TO VIN
OUTLINE OF LOCAL
GROUND PLANE
Figure 8. A Good PCB Layout Ensures Proper, Low EMI Operation
High Temperature Considerations
The PCB must provide heat sinking to keep the LT1912
cool. The exposed pad on the bottom of the package must
be soldered to a ground plane. This ground should be tied
to large copper layers below with thermal vias; these layers will spread the heat dissipated by the LT1912. Place
additional vias can reduce thermal resistance further. With
these steps, the thermal resistance from die (or junction)
to ambient can be reduced to JA = 35°C/W or less. With
100 LFPM airflow, this resistance can fall by another 25%.
Further increases in airflow will lead to lower thermal resistance. Because of the large output current capability of
the LT1912, it is possible to dissipate enough heat to raise
the junction temperature beyond the absolute maximum of
1912fa
16
LT1912
APPLICATIONS INFORMATION
125°C. When operating at high ambient temperatures, the
maximum load current should be derated as the ambient
temperature approaches 125°C.
Power dissipation within the LT1912 can be estimated by
calculating the total power loss from an efficiency measurement and subtracting the catch diode loss and inductor
loss. The die temperature is calculated by multiplying the
LT1912 power dissipation by the thermal resistance from
junction to ambient.
CLOSING SWITCH
SIMULATES HOT PLUG
IIN
VIN
+
LOW
IMPEDANCE
ENERGIZED
24V SUPPLY
Application Notes 19, 35 and 44 contain more detailed
descriptions and design information for buck regulators
and other switching regulators. The LT1376 data sheet
has a more extensive discussion of output ripple, loop
compensation and stability testing. Design Note 100
shows how to generate a bipolar output supply using a
buck regulator.
VIN
20V/DIV
DANGER
RINGING VIN MAY EXCEED
ABSOLUTE MAXIMUM RATING
LT1912
4.7µF
IIN
10A/DIV
STRAY
INDUCTANCE
DUE TO 6 FEET
(2 METERS) OF
TWISTED PAIR
0.7Ω
+
Other Linear Technology Publications
0.1µF
20µs/DIV
(9a)
LT1912
VIN
20V/DIV
4.7µF
IIN
10A/DIV
(9b)
+
22µF
35V
AI.EI.
+
LT1912
20µs/DIV
VIN
20V/DIV
4.7µF
IIN
10A/DIV
(9c)
20µs/DIV
1912 F09
Figure 9. A Well Chosen Input Network Prevents Input Voltage Overshoot and
Ensures Reliable Operation when the LT1912 is Connected to a Live Supply
1912fa
17
LT1912
TYPICAL APPLICATIONS
5V Step-Down Converter
VIN
6.8V TO 36V
VIN
ON OFF
VOUT
5V
2A
BD
RUN/SS
BOOST
L
6.8µH
0.47µF
VC
4.7µF
LT1912
SW
D
RT
16.2k
SYNC
68.1k
470pF
536k
FB
GND
47µF
100k
f = 500kHz
1912 TA02
D: DIODES INC. DFLS240L
L: TAIYO YUDEN NP06DZB6R8M
3.3V Step-Down Converter
VIN
4.4V TO 36V
VIN
ON OFF
VOUT
3.3V
2A
BD
RUN/SS
BOOST
0.47µF
VC
4.7µF
LT1912
L
6.8µH
SW
D
RT
20k
68.1k
470pF
SYNC
f = 500kHz
GND
316k
FB
47µF
100k
1912 TA03
D: DIODES INC. DFLS240L
L: TAIYO YUDEN NP06DZB4R7M
1912fa
18
LT1912
TYPICAL APPLICATIONS
2.5V Step-Down Converter
VIN
4V TO 36V
VIN
ON OFF
BD
RUN/SS
D2
BOOST
1µF
VC
4.7µF
VOUT
2.5V
2A
LT1912
L
6.8µH
SW
D1
RT
20k
68.1k
330pF
SYNC
f = 500kHz
GND
215k
FB
47µF
100k
1912 TA04
D1: DIODES INC. DFLS240L
D2: MBR0540
L: TAIYO YUDEN NP06DZB4R7M
1912fa
19
LT1912
TYPICAL APPLICATIONS
12V Step-Down Converter
VIN
15V TO 36V
VIN
ON OFF
VOUT
12V
2A
BD
RUN/SS
BOOST
0.47µF
VC
10µF
L
10µH
SW
LT1912
D
RT
26.1k
SYNC
68.1kHz
330pF
715k
FB
GND
22µF
50k
f = 500kHz
1912 TA06
D: DIODES INC. DFLS240L
L: NEC/TOKIN PLC-0755-100
1.8V Step-Down Converter
VOUT
1.8V
2A
VIN
3.5V TO 27V
VIN
ON OFF
BD
RUN/SS
BOOST
0.47µF
VC
4.7µF
LT1912
L
3.3µH
SW
D
RT
18.2k
68.1k
330pF
SYNC
f = 500kHz
GND
127k
FB
47µF
100k
1912 TA08
D: DIODES INC. DFLS240L
L: TAIYO YUDEN NP06DZB3R3M
1912fa
20
LT1912
PACKAGE DESCRIPTION
DD Package
10-Lead Plastic DFN (3mm × 3mm)
(Reference LTC DWG # 05-08-1699 Rev C)
0.70 ±0.05
3.55 ±0.05
1.65 ±0.05
2.15 ±0.05 (2 SIDES)
PACKAGE
OUTLINE
0.25 ± 0.05
0.50
BSC
2.38 ±0.05
(2 SIDES)
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
3.00 ±0.10
(4 SIDES)
R = 0.125
TYP
6
0.40 ± 0.10
10
1.65 ± 0.10
(2 SIDES)
PIN 1 NOTCH
R = 0.20 OR
0.35 × 45°
CHAMFER
PIN 1
TOP MARK
(SEE NOTE 6)
0.200 REF
0.75 ±0.05
0.00 – 0.05
5
1
(DD) DFN REV C 0310
0.25 ± 0.05
0.50 BSC
2.38 ±0.10
(2 SIDES)
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-2).
CHECK THE LTC WEBSITE DATA SHEET FOR CURRENT STATUS OF VARIATION ASSIGNMENT
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE
TOP AND BOTTOM OF PACKAGE
1912fa
21
LT1912
PACKAGE DESCRIPTION
MSE Package
10-Lead Plastic MSOP, Exposed Die Pad
(Reference LTC DWG # 05-08-1664 Rev G)
BOTTOM VIEW OF
EXPOSED PAD OPTION
1.88 ± 0.102
(.074 ± .004)
5.23
(.206)
MIN
1
0.889 ± 0.127
(.035 ± .005)
0.05 REF
10
0.305 ± 0.038
(.0120 ± .0015)
TYP
RECOMMENDED SOLDER PAD LAYOUT
3.00 ± 0.102
(.118 ± .004)
(NOTE 3)
DETAIL “B”
CORNER TAIL IS PART OF
DETAIL “B” THE LEADFRAME FEATURE.
FOR REFERENCE ONLY
NO MEASUREMENT PURPOSE
10 9 8 7 6
DETAIL “A”
0° – 6° TYP
1 2 3 4 5
GAUGE PLANE
0.53 ± 0.152
(.021 ± .006)
DETAIL “A”
0.18
(.007)
0.497 ± 0.076
(.0196 ± .003)
REF
3.00 ± 0.102
(.118 ± .004)
(NOTE 4)
4.90 ± 0.152
(.193 ± .006)
0.254
(.010)
0.29
REF
1.68
(.066)
1.68 ± 0.102 3.20 – 3.45
(.066 ± .004) (.126 – .136)
0.50
(.0197)
BSC
1.88
(.074)
SEATING
PLANE
0.86
(.034)
REF
1.10
(.043)
MAX
0.17 – 0.27
(.007 – .011)
TYP
0.50
(.0197)
BSC
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
6. EXPOSED PAD DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD
SHALL NOT EXCEED 0.254mm (.010") PER SIDE.
0.1016 ± 0.0508
(.004 ± .002)
MSOP (MSE) 0910 REV G
1912fa
22
LT1912
REVISION HISTORY
REV
DATE
DESCRIPTION
PAGE NUMBER
A
11/10
Changed units to mA for Quiescent Current from VIN for VBD = 0, Not Switching in Electrical Characteristics
2
1912fa
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
23
LT1912
TYPICAL APPLICATION
1.2V Step-Down Converter
VOUT
1.2V
2A
VIN
3.6V TO 27V
VIN
BD
RUN/SS
ON OFF
BOOST
0.47µF
VC
4.7µF
LT1912
L
3.3µH
SW
D
RT
16.2k
SYNC
68.1k
330pF
GND
52.3k
FB
100k
f = 500kHz
47µF
1912 TA09
D: DIODES INC. DFLS240L
L: TAIYO YUDEN NP06DZB3R3M
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT1933
500mA (IOUT), 500kHz Step-Down Switching Regulator in
SOT-23
VIN: 3.6V to 36V, VOUT(MIN) = 1.2V, IQ = 1.6mA, ISD <1µA, ThinSOT™
Package
LT3437
60V, 400mA (IOUT), Micropower Step-Down DC/DC
Converter with Burst Mode
VIN: 3.3V to 80V, VOUT(MIN) = 1.25V, IQ = 100µA, ISD <1µA,
10-Pin 3mm × 3mm DFN and 16-Pin TSSOP Packages
LT1936
36V, 1.4A (IOUT), 500kHz High Efficiency Step-Down
DC/DC Converter
VIN: 3.6V to 36V, VOUT(MIN) = 1.2V, IQ = 1.9mA, ISD <1µA, MS8E Package
LT3493
36V, 1.2A (IOUT), 750kHz High Efficiency Step-Down
DC/DC Converter
VIN: 3.6V to 40V, VOUT(MIN) = 0.8V, IQ = 1.9mA, ISD <1µA,
6-Pin 2mm × 3mm DFN Package
LT1976/LT1977
60V, 1.2A (IOUT), 200kHz/500kHz, High Efficiency
Step-Down DC/DC Converter with Burst Mode® Operation
VIN: 3.3V to 60V, VOUT(MIN) = 1.2V, IQ = 100µA, ISD <1µA,
16-Pin TSSOP Package
LT1767
25V, 1.2A (IOUT), 1.1MHz, High Efficiency Step-Down
DC/DC Converter
VIN: 3V to 25V, VOUT(MIN) = 1.2V, IQ = 1mA, ISD <6µA, MS8E Package
LT1940
Dual 25V, 1.4A (IOUT), 1.1MHz, High Efficiency Step-Down
DC/DC Converter
VIN: 3.6V to 25V, VOUT(MIN) = 1.2V, IQ = 3.8mA, ISD <30µA,
16-Pin TSSOP Package
LT1766
60V, 1.2A (IOUT), 200kHz, High Efficiency Step-Down
DC/DC Converter
VIN: 5.5V to 60V, VOUT(MIN) = 1.2V, IQ = 2.5mA, ISD = 25µA,
16-Pin TSSOP Package
LT3434/LT3435
60V, 2.4A (IOUT), 200/500kHz, High Efficiency Step-Down
DC/DC Converter with Burst Mode Operation
VIN: 3.3V to 60V, VOUT(MIN) = 1.2V, IQ = 100µA, ISD <1µA,
16-Pin TSSOP Package
LT3480
38V, 2A (IOUT), 2.4MHz, High Efficiency Step-Down DC/DC VIN: 3.6V to 38V, VOUT(MIN) = 0.79V, IQ = 70µA, ISD <1µA,
Converter with Burst Mode Operation
10-Pin 3mm × 3mm DFN and 10-Pin MSOP Packages
LT3481
36V, 2A (IOUT), 2.8MHz, High Efficiency Step-Down DC/DC VIN: 3.6V to 34V, VOUT(MIN) = 1.26V, IQ = 50µA, ISD <1µA,
Converter with Burst Mode Operation
10-Pin 3mm × 3mm DFN and 10-Pin MSOP Packages
LT3684
36V, 2A (IOUT), 2.8MHz, High Efficiency Step-Down
DC/DC Converter
VIN: 3.6V to 34V, VOUT(MIN) = 1.26V, IQ = 1.5mA, ISD <1µA,
10-Pin 3mm × 3mm DFN and 10-Pin MSOP Packages
LT3685
38V, 2A(IOUT) 2.4MHz Step-Down DC/DC Converter with
60V Transient Protection
VIN: 3.6V to 38V, VOUT(MIN) = 0.79V, IQ = 450µA, ISD < 1µA,
3mm × 3mm DFN, MSOP-10 Packages
1912fa
24 Linear Technology Corporation
LT 1110 REV A • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
 LINEAR TECHNOLOGY CORPORATION 2007