STMicroelectronics AN3338 In recent years the variable speed motor control market has required high performance Datasheet

AN3338
Application note
SLLIMM™
small low-loss intelligent molded module
By Carmelo Parisi and Giovanni Tomasello
Introduction
In recent years the variable speed motor control market has required high performance
solutions able to satisfy the increasing energy saving requirements, compactness, reliability,
and system costs in home appliances, such as washing machines, dish washers,
refrigerators, air conditioning compressor drives, and in low power industrial applications,
such as sewing machines, pumps, tools, etc. To meet these market needs,
STMicroelectronics has developed a new family of compact, high efficiency, dual-in-line
intelligent power modules, with optional extra features, called small low-loss intelligent
molded module (SLLIMM™).
The SLLIMM product family combines optimized silicon chips, integrated in three main
inverter blocks:
• power stage
– six short-circuit rugged IGBTs
– six freewheeling diodes
• driving network
– three high voltage gate drivers
– discrete gate resistors
– three bootstrap diodes
• protection and optional features
– op amps for advanced current sensing
– comparators for fault protection against overcurrent and short-circuit
– NTC sensor for temperature control
– smart shutdown function
– dead time, interlocking function and undervoltage lockout.
Thanks to the state of art DBC mounting technology, the fully isolated SLLIMM package
(SDIP) offers extremely low thermal resistance with optimum cost-effectiveness and quality
level.
Compared to discrete-based inverters, including power devices, and driver and protection
circuits, the SLLIMM family provides a high integrated level that means simplified circuit
design, reduced component count, smaller weight, and high reliability.
The aim of this application note is to provide a detailed description of SLLIMM products,
providing guidelines to motor drive designers for an efficient, reliable, and fast design when
using the new ST SLLIMM family.
March 2015
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Contents
AN3338
Contents
1
2
3
4
2/72
Inverter design concept and SLLIMM solution . . . . . . . . . . . . . . . . . . . . 6
1.1
Product synopsis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
1.2
Product line-up and nomenclature . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
1.3
Internal circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .11
1.4
Absolute maximum ratings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Electrical characteristics and functions . . . . . . . . . . . . . . . . . . . . . . . . 17
2.1
IGBTs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
2.2
Freewheeling diodes . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
2.3
High voltage gate drivers . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
2.3.1
Logic inputs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
2.3.2
High voltage level shift . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
2.3.3
Undervoltage lockout . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
2.3.4
Dead time and interlocking function management . . . . . . . . . . . . . . . . . 22
2.3.5
Comparators for fault sensing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
2.3.6
Short-circuit protection and smart shutdown function . . . . . . . . . . . . . . 25
2.3.7
Timing chart of short-circuit protection and smart shutdown function . . 26
2.3.8
Current sensing shunt resistor selection . . . . . . . . . . . . . . . . . . . . . . . . 27
2.3.9
RC filter network selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
2.3.10
Overtemperature protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
2.3.11
Op amps for advanced current sensing . . . . . . . . . . . . . . . . . . . . . . . . . 32
2.3.12
Bootstrap circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34
2.3.13
Bootstrap capacitor selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
2.3.14
Initial bootstrap capacitor charging . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37
Package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39
3.1
DBC substrate . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39
3.2
PCB . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40
3.3
Package structure . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40
3.4
Package outline and dimensions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 43
3.5
Input and output pins description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46
Power losses and dissipation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 53
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Contents
4.1
Conduction power losses . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 53
4.2
Switching power losses . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 56
4.3
Thermal impedance overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 57
4.4
Power losses calculation example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 60
Design and mounting guidelines . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 62
5.1
Layout suggestions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 62
5.1.1
5.2
General suggestions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 62
Mounting instructions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 66
5.2.1
Heatsink mounting . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 66
5.2.2
Mounting torque . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 67
5.2.3
General handling precaution and storage notices . . . . . . . . . . . . . . . . . 68
6
References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 70
7
Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 71
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List of tables
AN3338
List of tables
Table 1.
Table 2.
Table 3.
Table 4.
Table 5.
Table 6.
Table 7.
Table 8.
Table 9.
Table 10.
Table 11.
Table 12.
Table 13.
Table 14.
Table 15.
4/72
SLLIMM line-up. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
Inverter part of STGIPL14K60 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
Control part of STGIPL14K60 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
Supply voltage and operation behavior . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
Total STGIPL14K60 system . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
Integrated pull-up/down resistor values . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
Interlocking function truth table of STGIPS10K60A . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
Interlocking function truth table of STGIPS14K60, STGIPL14K60, STGIPS20K60, and STGIPL20K6023
SDIP-25L mechanical data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 44
SDIP-38L mechanical data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46
Input and output pins of SDIP-25L package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 47
Input and output pins of SDIP-38L package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 48
RC Cauer thermal network elements by device . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 60
Mounting torque and heatsink flatness . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 67
Document revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 71
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AN3338
List of figures
List of figures
Figure 1.
Figure 2.
Figure 3.
Figure 4.
Figure 5.
Figure 6.
Figure 7.
Figure 8.
Figure 9.
Figure 10.
Figure 11.
Figure 12.
GIPL20K60
Figure 13.
Figure 14.
Figure 15.
Figure 16.
Figure 17.
Figure 18.
Figure 19.
Figure 20.
Figure 21.
Figure 22.
Figure 23.
Figure 24.
Figure 25.
Figure 26.
Figure 27.
Figure 28.
Figure 29.
Figure 30.
Figure 31.
Figure 32.
Figure 33.
Figure 34.
Figure 35.
Figure 36.
Figure 37.
Figure 38.
Figure 39.
Figure 40.
Figure 41.
Figure 42.
Figure 43.
Figure 44.
Figure 45.
Figure 46.
Figure 47.
Inverter motor drive block diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
Discrete-based inverter vs. SLLIMM solution comparison . . . . . . . . . . . . . . . . . . . . . . . . . . 7
SLLIMM block diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
SLLIMM nomenclature . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
Internal circuit of STGIPS10K60A . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
Internal circuit of STGIPS14K60 and STGIPS20K60 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
Internal circuit of STGIPL14K60 and STGIPL20K60 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Stray inductance components of output stage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
High voltage gate drive die image . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
High voltage gate driver block diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
Logic input configuration for STGIPS10K60A. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
Logic input configuration for STGIPS14K60, STGIPL14K60, STGIPS20K60, and ST20
Timing chart of undervoltage lockout function . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
Timing chart of dead time function . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
Smart shutdown equivalent circuitry . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
Timing chart of smart shutdown function . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
Examples of SC protection circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
Example of SC event . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
NTC resistance vs. temperature curve . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
Example of overtemperature protection circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
3-phase system . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33
General advanced current sense scheme and waveforms. . . . . . . . . . . . . . . . . . . . . . . . . 33
Bootstrap circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34
Bootstrap capacitor vs. switching frequency. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37
Initial bootstrap charging time . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38
DCB structure . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39
PCB structure . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40
Images and internal view of SDIP-25L package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41
Images and internal view of SDIP-38L package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 42
Outline drawing of SDIP-25L package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 43
Outline drawing of SDIP-38L package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 45
Pinout of SDIP-25L package (bottom view) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 47
Pinout of SDIP-38L package (bottom view) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 48
Typical IGBT power losses . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 53
IGBT and diode approximation of the output characteristics . . . . . . . . . . . . . . . . . . . . . . . 54
Typical switching waveforms of STGIPL14K60 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 57
Equivalent thermal circuit with heatsink single IGBT . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 58
Thermal impedance curves (STGIPS14K60 and STGIPL14K60) . . . . . . . . . . . . . . . . . . . 59
Thermal impedance RC Cauer thermal network. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 59
Maximum IC(RMS) current vs. fsw simulated curves . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 61
General suggestions 1 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 63
General suggestions 2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 64
Example 1 of a possible wrong layout . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 65
Example 2 of a possible wrong layout . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 66
Recommended silicon grease thickness and positioning . . . . . . . . . . . . . . . . . . . . . . . . . . 67
Measurement point of Cu heatsink flatness . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 68
Recommended fastening order of mounting screws . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 68
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Inverter design concept and SLLIMM solution
1
AN3338
Inverter design concept and SLLIMM solution
Motor drive applications, ranging from a few tens of watts to mega watts, are mainly based
on the inverter concept thanks to the fact that this solution can meet efficiency, reliability,
size, and cost constraints required in a number of markets.
As shown in Figure 1, an inverter for motor drive applications is basically composed of a
power stage, mainly based on IGBTs and freewheeling diodes; a driving stage, based on
high voltage gate drivers; a control unit, based on microcontrollers or DSPs; some optional
sensors for protections and feedback signals for controls.
The approach of this solution with discrete devices produces high manufacturing costs
associated with high reliability risks, bigger size and higher weight, a considerable number
of components and the significant stray inductances and dispersions in the board layout.
Figure 1. Inverter motor drive block diagram
Mains
Microcontroller
Gate driver
Bridge rectifier
Power stage
M
Sensors
Feedback
!-V
In recent years, the use of intelligent power modules has rapidly increased thanks to the
benefits of greater integration levels. The new ST SLLIMM family is able to replace more
than 30 discrete devices in a single package. Figure 2 shows a comparison between a
discrete-based inverter and the SLLIMM solution, the advantages of SLLIMM can be easily
understood and can be summarized in a significantly improved design time, reduced
manufacturing efforts, higher flexibility in a wide range of applications, and increased
reliability and quality level.
In addition, the optimized silicon chips in both control and power stages and the optimized
board layout provide maximized efficiency, reduced EMI and noise generation, higher levels
of protection, and lower propagation delay time.
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AN3338
Inverter design concept and SLLIMM solution
Figure 2. Discrete-based inverter vs. SLLIMM solution comparison
Passive components:
Diodes
Resistors
Capacitors
HV gate drivers
Reduce total
system cost
IGBTs + FWDs
Easy layout
and design
Reduced EMI
and noise
M
LIM
SL
High quality
and reliability
Advanced
protection
function
Improve
Efficiency
!-V
1.1
Product synopsis
The SLLIMM family has been designed to satisfy the requirements of a wide range of final
applications in the range of 300 W - 2.0 kW, such as:
•
washing machines
•
dish washers
•
refrigerators
•
air conditioning compressor drives
•
sewing machines
•
pumps
•
tools
•
low power industrial applications
The main features and integrated functions can be summarized as follows:
•
600 V, 10 - 30 A ratings
•
3-phase IGBT inverter bridge including:
•
–
six low-loss and short-circuit protected IGBTs
–
six low forward voltage drop and soft recovery freewheeling diodes
three control ICs for gate driving and protection including:
–
smart shutdown function
–
comparator for fault protection against overcurrent and short-circuit
–
op amps for advanced current sensing
–
three integrated bootstrap diodes
–
interlocking function
–
undervoltage lockout
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Inverter design concept and SLLIMM solution
AN3338
•
NTC thermistor for temperature monitor
•
open emitter configuration for individual phase current sensing
•
DBC fully isolated package for enhanced thermal behavior
•
isolation voltage rating of 2500 VRMS
•
several passive components for IGBT switching speed optimum setting
•
gate driver proper biasing and noise filtering.
Figure 3 shows the block diagram of SLLIMM included in the inverter solution
Figure 3. SLLIMM block diagram
Mains
Bridge rectifier
Gate driver
UVLO /
Dead time
Level
Shift
Smart
Comparator Shut
Down
Bootstrap
diode
Half bridge
Op-Amp
Gate driver
Microcontroller
UVLO /
Dead time
Level
Shift
Smart
Comparator Shut
Down
Bootstrap
diode
Half bridge
M
Op-Amp
Gate driver
UVLO /
Dead time
Level
Shift
Smart
Comparator Shut
Down
Bootstrap
diode
Half bridge
Op-Amp
NTC
temperature
monitoring
SLLIMM
Feedback
!-V
The power devices (IGBTs and freewheeling diodes), incorporated in the half bridge block,
are tailored for a motor drive application delivering the greatest overall efficiency, thanks to
the optimized trade-off between conduction and switching power losses and very low EMI
generation, as a result of reduced dV/dt and di/dt.
The IC gate drivers have been selected in order to meet two levels of functionality giving the
designers more freedom to choose: a basic version which includes the essential features for
a cost-effective solution and a fully featured version which provides advanced options for a
sophisticated control method.
The fully isolated SDIP package is available in a 25-lead version (SDIP-25L) and 38-lead
version (SDIP-38L) and offers excellent heat dissipation characteristics, thanks to the state
of the art DBC mounting technology, ensuring at the same time, very high voltage isolation
rating (2500 VRMS), compact size and high reliability.
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1.2
Inverter design concept and SLLIMM solution
Product line-up and nomenclature
Table 1. SLLIMM line-up(1)
Basic version
Fully featured version
Features
STGIPS10K60A
STGIPS14K60
STGIPL14K60
STGIPS20K60
STGIPL20K60
Voltage (V)
600
600
600
600
600
Current @ TC=25 °C
(A)
10
14
15
18
20
RthJC max. single IGBT
(°C/W)
3.8
3
2.8
2.4
2.2
Package type
SDIP-25L
SDIP-25L
SDIP-38L
SDIP-25L
SDIP-38L
Package size
(mm) X, Y, Z
44.4x22.0x5.4
44.4x22.0x5.4
49.6x24.5x5.4
44.4x22.0x5.4
49.6x24.5x5.4
DBC substrate
Yes
Yes
Yes
Yes
Yes
NTC
Yes
No
Yes
No
Yes
Integrated bootstrap
diode
Yes
Yes
Yes
Yes
Yes
SD function
No
Yes
Yes
Yes
Yes
Comparator for fault
protection
No
Yes (1 pin)
Yes (3 pins)
Yes (1 pin)
Yes (3 pins)
Smart shutdown
function
No
Yes
Yes
Yes
Yes
Op amps for advanced
current sensing
No
No
Yes
No
Yes
Interlocking function
Yes
Yes
Yes
Yes
Yes
Undervoltage lockout
Yes
Yes
Yes
Yes
Yes
Yes (3 pins)
Yes (3 pins)
Yes (3 pins)
Yes (3 pins)
Yes (3 pins)
Yes
Yes
Yes
Yes
Yes
High side IGBT input
signal
Active High
Active High
Active High
Active High
Active High
Low side IGBT input
signal
Active High
Active Low
Active Low
Active Low
Active Low
Open emitter
configuration
3.3 / 5 V input interface
compatibility
1. For additional information and the complete product portfolio, refer to www.st.com/modules.
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Inverter design concept and SLLIMM solution
AN3338
Figure 4. SLLIMM nomenclature
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1.3
Inverter design concept and SLLIMM solution
Internal circuit
Figure 5. Internal circuit of STGIPS10K60A
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Inverter design concept and SLLIMM solution
Figure 6. Internal circuit of STGIPS14K60 and STGIPS20K60
12/72
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AN3338
AN3338
Inverter design concept and SLLIMM solution
Figure 7. Internal circuit of STGIPL14K60 and STGIPL20K60
1.4
Absolute maximum ratings
The absolute maximum ratings represent the extreme capability of the device and they can
be normally used as a worst limit design condition. It is important to note that the absolute
maximum value is given according to a set of testing conditions such us temperature,
frequency, voltage, and so on. The device performances can change according to the
applied condition.
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The SLLIMM specifications are described below by using the STGIPL14K60 datasheet as
an example. Please refer to the respective product datasheets for a detailed description of
other types.
Table 2. Inverter part of STGIPL14K60
Symbol
Parameter
Value
Unit
VPN
Supply voltage applied between P-NU, NV, NW
450
V
VPN(surge)
Supply voltage (surge) applied between P-NU, NV, NW
500
V
600
V
VCES
Collector emitter voltage
(VIN(1)=0)
±IC(2)
Each IGBT continuous collector current at TC=25 °C
15
A
±IC(3)
Each IGBT pulsed collector current
30
A
PTOT
Each IGBT total dissipation at TC=25 °C
44
W
tSCW
Short-circuit withstand time, VCE=0.5· V(BR)CES, Tj=125 °C,
VCC=Vboot=15 V, VIN(1)=0÷5 V
5
µs
1. Applied between HINU, HINV, HINW; LINU, LINV, LINW and GND.
2. Calculated according to the iterative Equation 1.
3. Pulse width limited by max. junction temperature.
Equation 1
IC (TC ) =
Tjmax − TC
Rth (j−c ) ⋅ VCE(sat)(max)(@Tj max,IC(TC ))
•
VPN: supply voltage applied between P-NU, NV, NW
•
VPN(surge): supply voltage (surge) applied between P-NU, NV, NW
•
VCES: collector emitter voltage
The power stage of SLLIMM is based on IGBTs (and freewheeling diodes) having 600 V
VCES rating. Considering the SLLIMM internal stray inductances during the commutations,
which can generate up to 100 V of surge voltage, the maximum surge voltage between P-N
(VPN(surge)) allowed is 500 V. At the same time, the maximum supply voltage (in steadystate) applied between P-N (VPN) allowed is 450 V because of an additional 50 V of surge
voltage generated by the stray inductance between the SLLIMM and the DC-link capacitor.
Figure 8 shows the parasitic inductances of the output stage. It is possible to note that there
are two major components, the first is due to the internal layout of SLLIMM, while the
second is due to the layout of the board.
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AN3338
Inverter design concept and SLLIMM solution
Figure 8. Stray inductance components of output stage
The real voltage over the IGBT Due to di/dt value and parasitic
Flat VPN
can exceed the rating voltage inductance the over-voltage spike
value
can appear on the SLLIMM pins
VPN
VPN(surge)
3
High
di/dt
value
HVIC
to motor
89:
&
9EXV
1
SLLIMM
Parasitic inductance
due to the SLLIMM internal layout
Parasitic inductance
due to PCB layout
!-V
•
±IC: each IGBT continuous collector current
The allowable DC current continuously flowing at collector electrode (TC = 25 °C). The IC
parameter is calculated according to Equation 1.
•
tSCW: short-circuit withstand time
The IGBTs incorporated inside the SLLIMM are tailored for a motor control application,
therefore, short-circuit self-protection is one of the main module features.
tSCW represents the short-circuit, non-repetitive, withstand time. If the short-circuit
conditions exceed the above specifications, the lifetime of the device is drastically
shortened. It is strongly recommended that the SLLIMM should not be operated under these
conditions.
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AN3338
Table 3. Control part of STGIPL14K60
Symbol
Parameter
Value
Unit
VOUT
Output voltage applied between OUTU, OUTV, OUTW, and
GND (VCC=15 V)
Vboot -21 to Vboot +0.3
V
VCC
Low voltage power supply
-0.3 to 21
V
VCIN
Comparator input voltage
-0.3 to VCC +0.3
V
Vboot
Bootstrap voltage
-0.3 to 620
V
VIN
Logic input voltage applied between HIN, LIN and GND
-0.3 to 15
V
VSD/OD
Open drain voltage
-0.3 to 15
V
dVOUT/dt
Allowed output slew rate
50
V/ns
•
VCC: low voltage power supply
VCC represents the supply voltage of the control part. A local filtering is recommended to
enhance the SLLIMM noise immunity. Generally, the use of one electrolytic capacitor (with a
greater value but not negligible ESR) and a good quality (low ESR, low ESL) filter capacitor
(hundreds of nF), faster than the electrolytic one to provide current, is suggested. Small filter
capacitors are already connected inside the SLLIMM, directly on the involved pins (see
internal circuits Figure 5, 6, and 7).
Please refer to Table 4 in order to properly drive the SLLIMM.
Table 4. Supply voltage and operation behavior
VCC voltage (typ. value) (1)
Operating behavior
< 12 V
As the voltage is lower than the UVLO threshold the control circuit is not fully
turned on. A perfect functionality cannot be guaranteed.
12 V – 13.5 V
IGBTs can work, however conduction and switching losses increase due to
low voltage gate signal.
13.5 V – 18 V
Recommended value (see relevant datasheets).
18 V – 21 V
IGBTs can work. Switching speed is faster and saturation current higher,
increasing short-circuit broken risk and EMI issues.
> 21 V
Control circuit is destroyed. Absolute max. rating is 21 V.
1. Except for STGIPS10K60A. For further information please refer to the relevant datasheet.
Table 5. Total STGIPL14K60 system
Symbol
Parameter
Value
Unit
VISO
Isolation withstands voltage applied between each pin and
heatsink plate (AC voltage, t = 60 sec.)
2500
V
Tj
Operating junction temperature
-40 to 150
°C
TC
Module case operation temperature
-40 to 125
°C
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AN3338
2
Electrical characteristics and functions
Electrical characteristics and functions
In this section the main electrical characteristics of the power stage are discussed, together
with a detailed description of all the SLLIMM functions.
2.1
IGBTs
The SLLIMM achieves power savings in the inverter stage thanks to the use of IGBTs
manufactured with the proprietary advanced PowerMESH™ process.
These power devices, optimized for the typical motor control switching frequency, offer an
excellent trade-off between voltage drop (VCE(sat)) and switching speed (tfall), and therefore
minimize the two major sources of energy loss, conduction and switching, reducing the
environmental impact of daily-use equipment. A full analysis on the power losses of the
complete system is reported in Section 4: Power losses and dissipation.
This IGBT family is capable of surviving short-circuits lasting up to 5 microseconds, as
expected by targeted applications.
2.2
Freewheeling diodes
The Turbo 2 ultrafast high voltage diodes have been adequately selected for the SLLIMM
family and carefully tuned to achieve the best trr/VF trade-off and softness as freewheeling
diodes in order to further improve the total performance of the inverter and significantly
reduce the electromagnetic interference (EMI) in motor control applications which are quite
sensitive to this phenomena.
2.3
High voltage gate drivers
The SLLIMM is equipped with a versatile high voltage gate driver IC (HVIC), designed using
BCD offline (Bipolar, CMOS, and DMOS) technology (see Figure 9) and particularly suited
to field oriented control (FOC) motor driving applications, able to provide all the functions
and current capability necessary for high side and low side IGBT driving. This driver can be
used in all applications where high voltage shifted control is necessary and it includes a
patented internal circuitry which replaces the external bootstrap diode.
Each high voltage gate driver chip controls two IGBTs in half bridge topology, offering the
basic functions such as dead time, interlocking, integrated bootstrap diode, and also the
advanced features such as smart shutdown (patented), fault comparator, and a dedicated
high performance op amp for advanced current sensing. A schematic summary of the
features by device are listed in Table 1.
In this application note the main characteristics of a high voltage gate drive related to the
SLLIMM are discussed. For a greater understanding, please refer to the AN2738 application
note.
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Figure 9. High voltage gate drive die image
Figure 10. High voltage gate driver block diagram
Bootstrap driver
VCC
Floating structure
BOOT
+
VCC
UV
detection
from μC
HIN
Shoottrough
prevention
CSD
OUT
SD/OD
GND
to motor
U,V,W
VCC
VBias
Shutdown
latch
Smart
shut
down
LVG
driver
LVG
N
+5V
CP+
+
CIN
Comp
-
RSF
CSF
+
VREF
DT
CDF
RDF
to ADC
Dead
time
VBias
VCC
OPOUT
OP+
+
Op-amp
HVIC
OP-
-
RSHUNT
SLLIMM
!-V
18/72
to DC-link
CBOOT
HVG
S
R
LIN
from μC
from/to μC
HVG
driver
Level
shifter
Logic
+5V
RSD
P
UV
UV
DETECTION
detection
from LVG
VBOOT
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AN3338
2.3.1
Electrical characteristics and functions
Logic inputs
The high voltage gate driver IC has two logic inputs, HIN and LIN, to separately control the
high side and low side outputs, HVG and LVG. Please refer to Table 1 for the input signal
logics by device.
In order to prevent any cross conduction between high side and low side IGBT a safety time
(dead time) is introduced (see Section 2.3.4: Dead time and interlocking function
management for further details).
All the logic inputs are provided with hysteresis (~1 V) for low noise sensitivity and are
TTL/CMOS 3.3 V compatible. Thanks to this low voltage interface logic compatibility, the
SLLIMM can be used with any kind of high performance controller, such as microcontrollers,
DSPs or FPGAs.
As shown in the block diagrams of Figure 11 and Figure 12, the logic inputs have internal
pull-down (or pull-up) resistors in order to set a proper logic level in case of interruption in
the logic lines. If logic inputs are left floating, the gate driver outputs LVG and HVG are set to
low level. This simplifies the interface circuit by eliminating the six external resistors and,
therefore saving cost, board space and number of components.
Figure 11. Logic input configuration for STGIPS10K60A
Bootstrap driver
9%227
3
9&&
UV detection
UV detection
High side
level
shifting
driver
Logic
+,1
287
Shoot-trough
prevention
Low side
driver
/,1
HVIC
1
SLLIMM
!-V
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Figure 12. Logic input configuration for STGIPS14K60, STGIPL14K60, STGIPS20K60, and
STGIPL20K60
Bootstrap driver
VBOOT
P
VCC
UV detection
UV detection
High side
level
shifting
driver
Logic
HIN
OUT
+5V
Shoot-trough
prevention
Low side
driver
LIN
Shutdown
N
Smart SD
SD
CIN
+
HVIC
-
VREF
SLLIMM
!-V
The typical values of the integrated pull-up/down resistors are shown in Table 6:
Table 6. Integrated pull-up/down resistor values
Input pin
PN
Input pin logic
High side gate driving
HINU, HINV, HINW
STGIPS10K60A
Active high
500 kΩ
Low side gate driving
LINU, LINV, LINW
STGIPS10K60A
Active high
500 kΩ
High side gate driving
HINU, HINV, HINW
STGIPS14K60
STGIPL14K60
STGIPS20K60
STGIPL20K60
Active high
85 kΩ
Low side gate driving
LINU, LINV, LINW
STGIPS14K60
STGIPL14K60
STGIPS20K60
STGIPL20K60
Active low
SD / OD shutdown
STGIPS14K60
STGIPL14K60
STGIPS20K60
STGIPL20K60
Active low
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Internal pull-up
Internal pull-down
720 kΩ
125 kΩ
AN3338
2.3.2
Electrical characteristics and functions
High voltage level shift
The built-in high voltage level shift allows direct connection between the low voltage control
inputs and the high voltage power half bridge in any power application up to 600 V. It is
obtained thanks to the BCD offline technology which integrates, in the same die bipolar
devices, low and medium voltage CMOS for analog and logic circuitry and high voltage
DMOS transistors with a breakdown voltage in excess of 600 V. This key feature eliminates
the need for external optocouplers, resulting in significant savings regarding component
count and power losses. Other advantages are high-frequency operation and short input-tooutput delays.
2.3.3
Undervoltage lockout
The SLLIMM supply voltage VCC is continuously monitored by an undervoltage lockout
(UVLO) circuitry which turns off the gate driver outputs when the supply voltage goes below
the VCC_thOFF threshold specified on the datasheet and turns on the IC when the supply
voltage goes above the VCC_thON voltage. A hysteresis of about 1.5 V is provided for noise
rejection purposes. The high voltage floating supply Vboot is also provided with a similar
undervoltage lockout circuitry. When the driver is in UVLO condition, both gate driver
outputs are set to low level, setting the half bridge power stage output to high impedance.
The timing chart of undervoltage lockout, plotted in Figure 13, is based on the following
steps:
•
t1: when the VCC supply voltage raises the VCC_thON threshold, the gate driver starts to
work after the next input signal HIN/LIN is on. The circuit state becomes RESET.
•
t2: input signal HIN/LIN is on and the IGBT is turned on.
•
t3: when the VCC supply voltage goes below the VCC_thOFF threshold, the UVLO event
is detected. The IGBT is turned off in spite of input signal HIN/LIN. The state of the
circuit is now SET.
•
t4: the gate driver re-starts once the VCC supply voltage again raises the VCC_thON
threshold.
•
t5: input signal HIN/LIN is on and the IGBT is turned on again.
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Figure 13. Timing chart of undervoltage lockout function
VCC_thON
VCC
≈
≈
VCC_thOFF
IC
Circuit state
Time
SET
RESET
RESET
t1
≈ ≈
HIN/LIN
t2
t3
t4 t5
!-V
2.3.4
Dead time and interlocking function management
In order to prevent any possible cross-conduction between high side and low side IGBTs,
the SLLIMM provides both the dead time and the interlocking functions. The interlocking
function is a logic operation which sets both the outputs to low level when the inputs are
simultaneously active. The dead time function is a safety time introduced by the device
between the falling edge transition of one driver output and the rising edge of the other
output. If the rising edge set externally by the user occurs before the end of this dead time, it
is ignored and results as delayed until the end of the dead time.
Table 7. Interlocking function truth table of STGIPS10K60A
Logic input (VI)
Outputs
Condition
LIN
HIN
LVG
HVG
Interlocking
half bridge tri-state
H
H
L
L
0 “logic state”
half bridge tri-state
L
L
L
L
1 “logic state”
low side direct driving
H
L
H
L
1 “logic state”
high side direct driving
L
H
L
H
The dead time is internally set at 320 ns as the typical value of STGIPS10K60A.
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Electrical characteristics and functions
Table 8. Interlocking function truth table of STGIPS14K60, STGIPL14K60, STGIPS20K60, and
STGIPL20K60
Logic input (VI)
Outputs
Condition
Note:
SD
LIN
HIN
LVG
HVG
Shutdown enable
half bridge tri-state
L
X
X
L
L
Interlocking
half bridge tri-state
H
L
H
L
L
0 “logic state”
half bridge tri-state
H
H
L
L
L
1 “logic state”
low side direct driving
H
L
L
H
L
1 “logic state”
high side direct driving
H
H
H
L
H
X: not important.
The dead time is internally set at 600 ns as typical value. In Figure 14 the details of dead
time and interlocking function management of the STGIPS14K60, STGIPL14K60,
STGIPS20K60, and STGIPL20K60 products are described.
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Electrical characteristics and functions
AN3338
Figure 14. Timing chart of dead time function
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Comparators for fault sensing
The SLLIMM family integrates up to three comparators (with reference to the product line-up
in Table 1) intended for advanced fault protection, such as overcurrent, overtemperature or
any other type of fault measurable via a voltage signal. Each comparator has an internal
reference voltage VREF, specified in the datasheet, on its inverting input (see Figure 10),
while the non-inverting input is available on CIN pins (one per half bridge). The comparators
input can be connected to an external shunt resistor, in order to implement a simple
overcurrent or short-circuit detection function, as discussed in detail in Section 2.3.6: Shortcircuit protection and smart shutdown function. Nevertheless, in the case of three internal
comparators, they can be separately used in order to implement three independent controls.
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2.3.6
Electrical characteristics and functions
Short-circuit protection and smart shutdown function
The SLLIMM is able to monitor the output current and provide protection against overcurrent
and short-circuit conditions in a very short time (comparator triggering to high/low side driver
turn-off propagation delay tisd = 200 ns), thanks to the smart shutdown function. This feature
is based on an innovative patented circuitry which provides an intelligent fault management
operation and greatly reduces the protection intervention delay independently on the
protection time duration which can be set as desired by the device user.
As already mentioned in Section 2.3.5: Comparators for fault sensing and shown in
Figure 10, each comparator input can be connected to an external shunt resistor, RSHUNT, in
order to implement a simple overcurrent detection function. An RC filter network (RSF and
CSF) is necessary to prevent erroneous operation of the protection. The output signal of the
comparators is fed to an integrated MOSFET with the open drain available on the SD/OD
pin, shared with the SD input. When the comparator triggers, the device is set in shutdown
state and all its outputs are set to low level, leaving the half bridge in tri-state. In common
overcurrent protection architectures, usually the comparator output is connected to the SD
input and an external RC network (RSD and CSD) is connected to this SD/OD line in order to
provide a mono-stable circuit which implements a protection time when a fault condition
occurs.
Contrary to common fault detection systems, the new smart shutdown structure allows to
immediately turn off the output gate driver in the case of fault, without waiting for the
external capacitor to be discharged. This strategy minimizes the propagation delay between
the fault detection event and the actual outputs switch off. In fact, the time delay between
the fault and outputs disabling is not dependent on the RC value of the external SD circuitry
but, thanks to the new architecture, has a preferential path internally in the driver. Then the
device immediately turns off the driver outputs and latches the turn-on of the open drain
switch, until the SD signal has reached its lower threshold. After the SD signal goes below
the lower threshold, the open drain is switched off (see Figure 16).
The smart shutdown system provides the possibility to increase the value of the external RC
network across the SD pin (sized to fix the disable time generated after the fault event) as
much as desired by the user without compromising the intervention time delay of the
SLLIMM protection.
A block diagram of the smart shutdown architecture is depicted in Figure 15.
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Electrical characteristics and functions
AN3338
Figure 15. Smart shutdown equivalent circuitry
LIN
LVG
HIN
HVG
VBias
SD
Q
FSD
S
+
CP+
Comp
-
+
VREF
Q
R
SET dominant FF
HVIC
SLLIMM
Except for STGIPS10K60A
!-V
In normal operation the outputs follow the commands received from the respective input
signals.
When a fault detection event occurs, the fault signal (FSD) is set to high by the fault
detection circuit output and the FF receives a SET input signal. Consequently, the FF
outputs set the SLLIMM output signals to low level and, at the same time, turn on the open
drain MOSFET which works as active pull-down for the SD signal. Note that the gate driver
outputs stay at low level until the SD pin has experienced both a falling edge and a rising
edge, although the fault signal could be returned to low level immediately after the fault
sensing. In fact, even if the FF is reset by the falling edge of the SD input, the SD signal also
works as enable for the outputs, thanks to the two AND ports. Moreover, once the internal
open drain transistor has been activated, due to the latch, it cannot be turned off until the SD
pin voltage reaches the low logic level. Note that, since the FF is SET dominant, oscillations
of the SD pin are avoided if the fault signal remains steady at high level.
2.3.7
Timing chart of short-circuit protection and smart shutdown function
With reference to Figure 16, the short-circuit protection is based on the following steps:
26/72
•
t1: when the output current is lower than the max. allowed level, the SLLIMM is working
in normal operation.
•
t2: when the output current reaches the max. allowed level (ISC), the overcurrent/shortcircuit event is detected and the protection is activated. The voltage across the shunt
resistor, and then on the CIN pin, exceeds the VREF value, the comparator triggers,
setting the device in shutdown state and both its outputs are set to low level leading the
half bridge to tri-state. The smart shutdown switches off the IGBT gate (HVG, LVG)
through a preferential path (200 ns as typical internal delay time) and, at the same time,
it switches on the M1 internal MOSFET. The SD signal starts the discharge phase and
its value drops with a time constant τA. The time constant τA value is given by:
Doc ID 18441 Rev 4
AN3338
Electrical characteristics and functions
Equation 2
(
)
τ A = RON _ OD // RSD ⋅ CSD
•
t3: the SD signal reaches the lower threshold Vsd_L_THR and the control unit switches
off the input HIN and LIN. The smart shutdown is disabled (M1 off) and SD can rise up
with a time constant τB, given by:
Equation 3
τB = RSD ⋅ CSD
•
t4: when the SD signal reaches the upper threshold Vsd_H_THR, the system is reenabled.
Figure 16. Timing chart of smart shutdown function
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Current sensing shunt resistor selection
As previously discussed, the shunt resistors RSHUNT externally connected between the N
pin and ground (see Figure 10) are used to realize the overcurrent detection.
When the output current exceeds the short-circuit reference level (ISC), the CIN signal
overtakes the VREF value and the short-circuit protection is active. For a reliable and stable
operation the current sensing resistor should be a high quality, low tolerance non-inductive
type. In fact, stray inductance in the circuit, which includes the layout, the RC filter, and also
the shunt resistor, must be minimized in order to avoid undesired short-circuit detection.
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Electrical characteristics and functions
AN3338
For these reasons, the shunt resistor and the filtering components must be placed as close
as possible to the SLLIMM pins, for additional suggestions refer to Section 5.1: Layout
suggestions.
The value of the current sense resistor can be calculated by following different guidelines,
functions of the design specifications, or requirements. A common criterion is presented
here based on the following steps:
•
Defining of the overcurrent threshold value (IOC_th). For example, it can be fixed
considering the IGBT typical working current in the application and adding 20-30% as
overcurrent.
•
Calculation of the shunt resistor value according to the conditioning network. An
example of the conditioning network is shown in Figure 22. Further details can be
found in the user manuals listed (see References 7, References 8, and References 9).
•
Selection of the closest shunt resistor commercial value.
•
Calculation of the power rating of the shunt resistor, taking into account that this
parameter is strongly temperature dependent. Therefore, the power derating ratio of
the shunt resistor, ΔP(T)%, shown in the manufacturer's datasheet, must be considered
in the calculation as follows:
Equation 4
PSHUNT( T ) =
RSHUNT ⋅ I2RMS
Δ P(T )%
where IRMS is the IGBT RMS working current.
For a proper selection of the shunt resistor, a safety margin of at least 30% is recommended
on the calculated power rating.
28/72
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AN3338
2.3.9
Electrical characteristics and functions
RC filter network selection
Two options of shunt (1- or 3-shunt) resistor circuit can be adopted in order to implement
different control technique and short-circuit protection, as shown in Figure 17.
Figure 17. Examples of SC protection circuit
NU
NV
NW
RSF
CIN
RSHUNT
SLLIMM
NU RSHUNT_U
R
NV SHUNT_V
NW RSHUNT_W
CSF
SLLIMM
1-shunt resistor circuit
CIN
RSF
RSHUNT
CSF
3-shunt resistors circuit
!-V
A RC filter network is required to prevent undesired short-circuit operation due to the noise
on the shunt resistor.
Both solutions allow to detect the total current in all three phases of the inverter. The filter is
based on the RSF and CSF network and its time constant is given by:
Equation 5
tSF = RSF ⋅ CSF
In addition to the RC time constant, the turn-off propagation delay of the gate driver, tisd
(specified in the datasheet) and the IGBT turn-off time (in the range of tens of ns), must be
considered in the total delay time (tTotal), which is the time necessary to completely switch
off the IGBT once the short-circuit event is detected. Therefore, the tTotal is calculated as
follows:
Equation 6
t Total = tSF + tisd + toff
also considering that the IGBT short-circuit withstand time (tSC) is 5 µs, the tSF is
recommended to be set in the range of 1~2 µs.
In the case of a 3-shunt resistors circuit, a specific control technique can be implemented by
using the three shunt resistors (RSHUNT_U, RSHUNT_V and RSHUNT_W) able to monitor each
phase current.
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Electrical characteristics and functions
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An example of a short-circuit event is shown in Figure 18, where it is possible to note the
very fast protection, thanks to the smart shutdown function, against fault events. The main
steps are:
•
t1: collector current IC starts to rise. SC event is not detected yet due to the RC network
on the CIN pin
•
t2: voltage on VCIN reaches the VREF. SC event is detected and the smart shutdown
starts to turn off the SLLIMM.
•
t3: the SLLIMM is definitively turned off in less than 300 ns (including the td(off) time of
IGBT) from SC detection.
Finally, the total disable time is t3-t2 and the total SC action time is t3-t1.
Figure 18. Example of SC event
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2.3.10
Overtemperature protection
STGIPS10K60A, STGIPL14K60, and STGIPL20K60 are equipped with a negative
temperature coefficient (NTC) thermistor for an easy overtemperature protection, in the
case of slow case temperature drift or just for the temperature measurements, sending this
information to the microcontroller in real-time. Due to the thermal impedance of SLLIMM
and its own time constant, the NTC thermistor is not suited to detect rapid junction
temperature rise of the power devices directly. Therefore, it cannot be used for short-circuit
or overcurrent protection, but only for slow changes in temperature monitoring.
The resistance versus temperature characteristic of NTC thermistor, represented in
Figure 19, is non-linear and is described by the following expression:
Equation 7
R(T ) = R25
30/72
1 1 
B −

⋅ e  T 298 
Doc ID 18441 Rev 4
AN3338
Electrical characteristics and functions
where T is the temperature in Kelvin, B and R25, respectively, are a constant value in the
SLLIMM working range and the resistance value at 25 °C, both parameters are shown in the
datasheet.
Figure 19. NTC resistance vs. temperature curve
An easy circuit, using a voltage divider, for both overtemperature protection and
temperature monitoring, is shown in Figure 20:
Figure 20. Example of overtemperature protection circuit
+VDD
T1
NTC
T2
+
COT
ROT VNTC_th
SD
SLLIMM
!-V
The external comparator is used to send a shutdown signal to the SLLIMM in case of
overtemperature. The VNTC_th is a threshold voltage, fixed by design, and connected on the
non inverting input, whilst the inverting input is connected on a voltage divider based on the
NTC and ROT resistors. When voltage on the inverting input exceeds the VNTC_th value the
comparator triggers, pulling down the SD and, consequently, switching off the IGBTs.
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For a proper sizing of the voltage divider, first of all the maximum allowed temperature level
(TOT_Max) must be fixed, consequently the thermistor resistance is given by Equation 7, as
well as by Figure 19. The value of ROT resistance can be calculated by using the voltage
divider formula:
Equation 8
−
V (T) =
ROT
⋅ VDD
RNTC(T) + ROT
taking into account that, if T = TOT_Max then V-(TOT_Max) = VNTC_th.
The maximum allowed power on the thermistor should not exceed 50 mW in all the
operating range, in order to guarantee a safe working condition and avoid power
consumption affecting the temperature measurement through self-heating. Therefore,
considering (T = TOT_Max), it must be:
Equation 9
2

VDD
 ≤ 50mW

R
 NTC + ROT 

RNTC ⋅ I2 = RNTC ⋅ 
Finally, to increase the noise immunity of the NTC thermistor, it is recommended to parallel a
decoupling capacitor (COT), whose value must be between 10 to 100 nF.
2.3.11
Op amps for advanced current sensing
The SLLIMM devices, in the SDIP-38L package, integrate also three operational amplifiers
optimized for field oriented control (FOC) applications. In a typical FOC application the
currents in the three half bridges are sensed using a shunt resistor. The analog current
information is transformed into a discontinuous sense voltage signal, having the same
frequency as the PWM signal driving the bridge. The sense voltage is a bipolar analog
signal, whose sign depends on the direction of the current (see Figure 21):
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Electrical characteristics and functions
Figure 21. 3-phase system
3-phase driver
Sinusoidal Vector Control
Sensing:
Discontinuous Voltage at fPWM frequency
96
96
96
Power
stage
IPHASE
SKDVH
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AM09338v1
The sense voltage signals must be provided to an A/D converter. They are usually shifted
and amplified by dedicated op amps in order to exploit the full range of the A/D converter.
The typical scheme and principle waveforms are shown in Figure 22:
Figure 22. General advanced current sense scheme and waveforms
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Electrical characteristics and functions
AN3338
ADCs used in vector control applications have a typical full scale range (FSR) of about 3.3
V. The sense signals must be shifted and centered on FSR/2 voltage (about 1.65 V) and
amplified with a gain which provides the matching between the maximum value of the
sensed signal and the FSR of the ADC. Some typical examples of sense network sizing can
be found in the user manuals listed (see References 7, References 8, and References 9).
2.3.12
Bootstrap circuit
In the 3-phase inverter the emitters of the low side IGBTs are connected to the negative DC
bus (VDC-) as common reference ground, which allows all low side gate drivers to share the
same power supply, while, the emitter of high side IGBTs is alternately connected to the
positive (VDC+) and negative (VDC-) DC bus during the running conditions.
A bootstrap method is a simple and cheap solution to supply the high voltage section. This
function is normally accomplished by a high voltage fast recovery diode. The SLLIMM family
includes a patented integrated structure that replaces the external diode. It is realized with a
high voltage DMOS driven synchronously with the low side driver (LVG) and a diode in
series. An internal charge pump provides the DMOS driving voltage.
The operation of the bootstrap circuit is shown in Figure 23. The floating supply capacitor
CBOOT is charged, from the VCC supply, when the VOUT voltage is lower than the VCC
voltage (e.g. low side IGBT is on), through the bootstrap diode and the DMOS path with
reference to the “bootstrap charge current path”. During the high side IGBT on phase, the
bootstrap circuit provides the right gate voltage to properly drive the IGBT (see “bootstrap
discharge current path”). This circuit is iterated for all three half bridges.
Figure 23. Bootstrap circuit
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34/72
%RRWVWUDS
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Doc ID 18441 Rev 4
1
AN3338
Electrical characteristics and functions
The value of the CBOOT capacitor should be calculated according to the application
condition and must take the following into account:
2.3.13
•
voltage across CBOOT must be maintained at a value higher than the undervoltage
lockout level for the IC driver. This enables the high side IGBT to work with a correct
gate voltage (lower dissipation and better overall performances). Bear in mind that if a
voltage below the UVLO threshold is applied on the bootstrap channel, the IC disables
itself (no output) without any fault signal.
•
the voltage across CBOOT is affected by different components such as drop across the
integrated bootstrap structure, drop across the low side IGBT, and others.
•
when the high side IGBT is on, the CBOOT capacitor discharges mainly to provide the
right IGBT gate charge but other phenomena must be considered such as leakage
currents, quiescent current, etc.
Bootstrap capacitor selection
A simple method to properly size the bootstrap capacitor considers only the amount of
charge that is needed when the high voltage side of the driver is floating and IGBT gate is
driven once. This approach does not take into account either the duty cycle of the PWM, or
the fundamental frequency of the current.
During the bootstrap capacitor charging phase, the low side IGBT is on and the voltage
across CBOOT (VCBOOT) can be calculated as follows:
Equation 10
VCBOOT = VCC − VF − VRDS(on) − VCE(sat) max
where:
VCC: supply voltage of gate driver
VF: bootstrap diode forward voltage drop
VCE(sat)max: maximum emitter collector voltage drop of low side IGBT
VRDS(on): DMOS voltage drop
The dimension of the bootstrap capacitance CBOOT value is based on the minimum voltage
drop (ΔVCBOOT) to guarantee when the high side IGBT is on, and must be:
Equation 11
ΔVCBOOT = VCC − VF − VRDS(on) − VGE(min) − VCE(sat) max
under the condition:
Equation 12
VCBOOT(min) > VBS _ thON
where:
VGE(min): minimum gate emitter voltage of high side IGBT
VBS_thON: bootstrap turn-on undervoltage threshold (maximum value, see datasheet)
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Considering the factors contributing to VCBOOT decreasing, the total charge supplied by the
bootstrap capacitor (during high side on phase) is:
Equation 13
(
)
Q TOT = Q GA TE + ILKGE + IQBO + ILK + ILKDiode + ILKCap ⋅ tHon + Q LS
where:
QGATE: total IGBT gate charge
ILKGE: IGBT gate emitter leakage current
IQBO: bootstrap circuit quiescent current
ILK: bootstrap circuit leakage current
ILKDiode: bootstrap diode leakage current
ILKCap: bootstrap capacitor leakage current (relevant when using an electrolytic capacitor
but can be ignored if other types of capacitors are used)
tHon: high side on time
QLS: charge required by the internal level shifters
Finally, the minimum size of the bootstrap capacitor is:
Equation 14
CBOOT =
QTOT
ΔVCBOOT
For an easier selection of bootstrap capacitor, Figure 24 shows the behavior of CBOOT
(calculated) versus switching frequency (fsw), with different values of ΔVCBOOT,
corresponding to Equation 14 for a continuous sinusoidal modulation and for STGIPS20K60
and STGIPL20K60 (worst case) and a duty cycle δ = 50%. For all the other devices the
bootstrap capacitor can be calculated using the same curve.
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Electrical characteristics and functions
Figure 24. Bootstrap capacitor vs. switching frequency
AM09341v1
5
STGIPx20K60
δ=50%
C BOOT Calculated (µF)
4
3
ΔVCBOOT =0.1V
2
ΔVCBOOT =0.3V
ΔVCBOOT =0.5V
1
0
0
5
10
15
20
fsw (kHz)
Considering the limit cases during the PWM control and further leakages and dispersions in
the board layout, the capacitance value to use in the bootstrap circuit must be selected two
or three times higher than the CBOOT calculated in the graph of Figure 24. The bootstrap
capacitor should be with a low ESR value for a good local decoupling, therefore, in case an
electrolytic capacitor is used, a good quality (low ESR, low ESL) filter capacitor placed
directly on the SLLIMM pins is strictly recommended.
2.3.14
Initial bootstrap capacitor charging
During the startup phase, the bootstrap capacitor must be charged for a suitable time to
complete the initial charging time (tCHARGE), which is, at least, the time VCBOOT needs to
exceed the turn-on undervoltage threshold VBS_thON, as already stated in Equation 12. For
a normal operation, the voltage across the bootstrap capacitor must never drop down to the
turn-off undervoltage threshold VBS_thOFF throughout the working conditions. For the period
of startup, only the low side IGBT is switched on and, just after this phase, the PWM is run,
as shown in the following steps of Figure 25:
•
t1: the bootstrap capacitor starts to charge through the low side IGBT (LVG)
•
t2: the voltage across the bootstrap capacitor (VCBOOT) reaches its turn-on
undervoltage threshold VBS_thON.
•
t3: the bootstrap capacitor is fully charged, this enables the high side IGBT and the
CBOOT capacitor starts to discharge in order to provide the right IGBT gate charge. The
bootstrap capacitor recharges during the on state of low side IGBT (LVG).
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Figure 25. Initial bootstrap charging time
VCC
DC Bus VPN
HVG
LVG
VBS_thON
VBS_thOFF
VCBOOT
Time
t1
t2
t3
!-V
The initial charging time is given by Equation 15 and must be, for safety reasons, at least
three times longer than the calculated value.
Equation 15
tCHARGE ≥
CBOOT ⋅ RDS(on)
δ
 VCC


* ln
 ΔV

 CBOOT 
where δ is the duty cycle of the PWM signal and RDS(on) is 120 Ω typical value, as shown in
the datasheet.
A practical example can be done by considering a motor drive application where the PWM
switching frequency is 12.5 kHz, with a duty cycle of 50%, and ΔVCBOOT = 0.1 V (that
means, a gate driver supply voltage VCC = 17.6 V). From the graph in Figure 24 the
bootstrap capacitance is 1.5 µF, therefore, the CBOOT can be selected by using a value
between 3.0 and 4.5 µF. According to the commercial value the bootstrap capacitor can be
3.3 µF. From Equation 15, the initial charging time is:
Equation 16
tCHARGE ≥
3.3 ⋅ 10−6 ⋅120  17.6 
⋅ ln
 = 4ms
0.5
 0.1 
For safety reasons, the initial charging time must be at least 12 ms.
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3
Package
Package
The SLLIMM benefits from a compact package while providing high power density, the best
thermal performance, and great electrical isolation (> 2500 VRMS).
The SDIP is a dual-in-line transfer mold package, available in 25-lead version (SDIP-25L)
and 38-lead version (SDIP-38L) and based on the state of the art DBC mounting technology
for the power stage, whilst the control stage is assembled on a PCB layer. A vacuum
soldering process is used to avoid any gas inclusion (voids) during the soldering process
that could cause potential hot spots. It results in a further increase in the reliability of the
SLLIMM family due to the improved thermal and electrical conductivity.
This technology makes it possible to achieve extremely low thermal resistance values, high
stability in thermal cycling, small size with optimum cost-effectiveness, and quality level.
3.1
DBC substrate
DBC means direct bonded copper and denotes a process in which copper and a ceramic
material are directly bonded, as shown in Figure 26. Direct bonded copper substrates have
been proven for many years to be an excellent solution for electrical isolation and thermal
management of high power semiconductor modules.
Figure 26. DCB structure
The advantages of DBC substrates are, firstly, high current-carrying capability, due to thick
copper metallization, and secondly, a thermal expansion coefficient close to the silicon one
at the copper surface.
DBC has two layers of copper that are directly bonded onto an aluminum-oxide (Al2O3)
ceramic base. The DBC process yields a super-thin base and eliminates the need for thick,
heavy copper bases that were used prior to this process. Because SLLIMM with DBC bases
has fewer layers, it has much lower thermal resistance values than those based on different
materials.
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The main properties of DBC ceramic substrates
The main properties of DBC include good mechanical strength, mechanically stable shape,
good adhesion and corrosion resistance, and also offer:
3.2
•
Excellent electrical isolation
•
Very good thermal conductivity
•
The thermal expansion coefficient is close to that of silicon, so no interface layers are
required
•
Good heat spreading
•
May be structured just like printed circuit boards or “IMS substrates”
•
Environmentally clean
PCB
A PCB (printed circuit board) is used to mechanically support the gate driver ICs and to
electrically connect those using conductive pathways.
Thanks to the internal PCB it is possible to realize various electric configurations, necessary
to add advanced features, and to insert several passive components, such as resistors or
capacitors, to properly bias the gate drivers. The insertion of filter capacitors, directly across
the gate driver pins, improves the SLLIMM noise immunity and helps users to work in a
safer condition. Figure 27 shows the internal PCB detail.
Figure 27. PCB structure
3.3
Package structure
Figure 28 and Figure 29 contain images and an internal structure illustration of the SDIP25L and SDIP-38L package.
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Package
Figure 28. Images and internal view of SDIP-25L package
Top view
Bottom view
z
x
SLLIMM
y
SDIP-25L
Main dimensions
HVI C
PCB
IGBT
FWD
DBC
x = 44.4 mm
y1 = 22 mm (body only)
y2 = 25.22 mm (including leads)
z1 = 5.4 mm (body only)
z2 = 11.6 mm (including leads)
Internal view
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Figure 29. Images and internal view of SDIP-38L package
Top view
Bottom view
z
x
SLLIMM
y
SDIP-38L
Main dimensions
HVGD
PCB
IGBT
x = 49.6 mm
y1 = 24.5 mm (body only)
y2 = 29.1 mm (including leads)
z1 = 5.4 mm (body only)
z2 = 10.91 mm (including leads)
FWD
DBC
Internal view
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3.4
Package
Package outline and dimensions
Figure 30. Outline drawing of SDIP-25L package
B
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Table 9. SDIP-25L mechanical data
mm
Dim.
44/72
Min.
Typ.
Max.
A
43.90
44.40
44.90
A1
1.15
1.35
1.55
A2
1.40
1.60
1.80
A3
38.90
39.40
39.90
B
21.50
22.00
22.50
B1
11.25
11.85
12.45
B2
24.83
25.23
25.63
C
5.00
5.40
6.00
C1
6.50
7.00
7.50
C2
11.20
11.70
12.20
C3
2.90
3.00
3.10
e
2.15
2.35
2.55
e1
3.40
3.60
3.80
e2
4.50
4.70
4.90
e3
6.30
6.50
6.70
D
33.30
D1
5.55
E
11.20
E1
1.40
F
0.85
1.00
1.15
F1
0.35
0.50
0.65
R
1.55
1.75
1.95
T
0.45
0.55
0.65
V
0°
6°
Doc ID 18441 Rev 4
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Package
Figure 31. Outline drawing of SDIP-38L package
B
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Table 10. SDIP-38L mechanical data
mm
Dim.
3.5
Min.
Typ.
Max.
A
49.10
49.60
50.10
A1
1.10
1.30
1.50
A2
1.40
1.60
1.80
A3
44.10
44.60
45.10
B
24.00
24.50
25.00
B1
11.25
11.85
12.45
B2
27.10
27.60
28.10
B3
28.60
29.10
29.60
C
5.00
5.40
6.00
C1
6.50
7.00
7.50
C2
10.35
10.85
11.35
C3
2.90
3.00
3.10
e
1.10
1.30
1.50
e1
3.20
3.40
3.60
e2
5.80
6.00
6.20
e3
4.60
4.80
5.00
e4
5.60
5.80
6.00
e5
6.30
6.50
6.70
e6
4.50
4.70
4.90
D
38.10
D1
5.75
E
11.80
E1
2.15
F
0.85
1.00
1.15
F1
0.35
0.50
0.65
R
1.55
1.75
1.95
T
0.45
0.55
0.65
V
0°
6°
Input and output pins description
This paragraph defines the input and output pins of SLLIMM. For a more accurate
description and layout suggestions, please consult the relevant sections.
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Package
Figure 32. Pinout of SDIP-25L package (bottom view)
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Table 11. Input and output pins of SDIP-25L package
Name
Pin #
Description
STGIPS14K60
STGIPS10K60A
STGIPS20K60
STGIPS14K60
STGIPS10K60A
STGIPS20K60
1
OUTU
High side reference output for U phase
2
VbootU
Bootstrap voltage for U phase
3
LINU
Low side logic input for U
phase (active high)
LINU
Low side logic input for U
phase (active low)
4
HINU
High side logic input for U phase
5
VCC
Low voltage power supply
6
OUTV
High side reference output for V phase
7
VbootV
Bootstrap voltage for V phase
8
GND
Ground
9
LINV
Low side logic input for V
phase (active high)
LINV
Low side logic input for V
phase (active low)
10
HINV
High side logic input for V phase
11
OUTW
High side reference output for W phase
12
VbootW
Bootstrap voltage for W phase
13
LINW
14
Low side logic input for W
phase (active high)
LINW
HINW
Low side logic input for W
phase (active low)
High side logic input for W phase
15
T1
SD / OD
NTC thermistor terminal 1
SD logic input (active low) /
open drain (comp output)
16
T2
CIN
NTC thermistor terminal 2
Comparator input
17
NW
Negative DC input for W phase
18
W
W phase output
19
P
Positive DC input
20
NV
Negative DC input for V phase
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Table 11. Input and output pins of SDIP-25L package (continued)
Name
Pin #
Description
STGIPS14K60
STGIPS10K60A
STGIPS20K60
STGIPS14K60
STGIPS10K60A
STGIPS20K60
21
V
V phase output
22
P
Positive DC input
23
NU
Negative DC input for U phase
24
U
U phase output
25
P
Positive DC input
Figure 33. Pinout of SDIP-38L package (bottom view)
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Table 12. Input and output pins of SDIP-38L package
STGIPL14K60 STGIPL20K60
Pin #
Name
Description
1
OUTU
High side reference output for U phase
2
VbootU
Bootstrap voltage for U phase
3
LINU
Low side logic input for U phase (active low)
4
HINU
High side logic input for U phase
5
OP-U
Op amp inverting input for U phase
6
OPOUTU
Op amp output for U phase
7
OP+U
Op amp non inverting input for U phase
8
CINU
Comparator input for U phase
9
OUTV
High side reference output for V phase
10
VbootV
Bootstrap voltage for V phase
11
LINV
Low side logic input for V phase (active low)
12
HINV
High side logic input for V phase
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Package
Table 12. Input and output pins of SDIP-38L package (continued)
STGIPL14K60 STGIPL20K60
Pin #
Name
Description
13
OP-V
Op amp inverting input for V phase
14
OPOUTV
Op amp output for V phase
15
OP+V
Op amp non inverting input for V phase
16
CINV
Comparator input for V phase
17
OUTW
High side reference output for W phase
18
VbootW
Bootstrap voltage for W phase
19
LINW
Low side logic input for W phase (active low)
20
HINW
High side logic input for W phase
21
OP-W
Op amp inverting input for W phase
22
OPOUTW
Op amp output for W phase
23
OP+W
Op amp non inverting input for W phase
24
CINW
Comparator input for W phase
25
VCC
Low voltage power supply
26
SD / OD
Shutdown logic input (active low) / open drain (comparator output)
27
GND
Ground
28
T2
NTC thermistor terminal 2
29
T1
NTC thermistor terminal 1
30
NW
Negative DC input for W phase
31
W
W phase output
32
P
Positive DC input
33
NV
Negative DC input for V phase
34
V
V phase output
35
P
Positive DC input
36
NU
Negative DC input for U phase
37
U
U phase output
38
P
Positive DC input
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High-Side bias voltage pins /high-side bias voltage reference
Pins: VbootU-OUTU, VbootV-OUTV, VbootW-OUTW
•
The bootstrap section is designed to realize a simple and efficient floating power
supply, in order to provide the gate voltage signal to the high-side IGBTs.
•
The SLLIMM family integrates the bootstrap diodes. This helps customer to save cost,
board space, and number of components.
•
The advantage of the ability to bootstrap the circuit scheme is that no external power
supplies are required for the high-side IGBTs.
•
Each bootstrap capacitor is charged from the VCC supply during the on-state of the
corresponding low side IGBT.
•
To prevent malfunctions caused by noise and ripple in supply voltage, a good quality
(low ESR, low ESL) filter capacitor should be mounted close to these pins.
•
The value of bootstrap capacitors is strictly related to the application conditions. Please
consult Section 2.3.12: Bootstrap circuit.
Gate driver bias voltage
Pin: VCC
•
Control supply pin for the built-in ICs.
•
To prevent malfunctions caused by noise and ripple in the supply voltage, a good
quality (low ESR, low ESL) filter capacitor should be mounted close to this pin.
Gate drive supply ground
Pin: GND
•
Ground reference pin for the built-in ICs.
•
To avoid noise influences, the main power circuit current should not be allowed to flow
through this pin (see Section 5.1: Layout suggestions).
Signal input
Pins: HINU, HINV, HINW; LINU, LINV, LINW; LINU, LINV, LINW
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•
These pins control the operation of the built-in IGBTs.
•
The signal logic of HINU, HINV, HINW, LINU, LINV, and LINW pins is active high. The
IGBT associated with each of these pins is turned on when a sufficient logic (higher
than a specific threshold) voltage is applied to these pins.
•
The signal logic of LINU, LINV, LINW pins is active low. The IGBT associated with each
of these pins is turned on when a logic voltage (lower than a specific threshold voltage)
is applied to these pins.
•
The wiring of each input should be as short as possible to protect the SLLIMM against
noise influences.
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Package
Internal comparator non-inverting
Pins: CINU, CINV, CINW
•
The current sensing shunt resistor, connected on each phase leg, could be used by the
internal comparator (pins CINU, CINV and CINW) to detect short-circuit current.
•
The shunt resistor should be selected to meet the detection levels matched for the
specific application.
•
An RC filter (typically ~ 1 µs) should be connected to the CINU, CINV, CINW pins to
eliminate noise.
•
The connection length between the shunt resistor and CINU, CINV, CINW pins should
be minimized.
•
If a voltage signal, higher than the specified VREF (see datasheet), is applied to this pin,
the SLLIMM automatically shuts down and the SD / OD pin is pulled down (to inform
the microcontroller).
Shutdown / open drain
Pin: SD / OD
•
The SD / OD pin works as an enable/disable pin.
•
The signal logic of the SD / OD pin is active low. The SLLIMM shuts down if a voltage
lower than a specific threshold is applied to this pin, leading each half bridge in tri-state.
•
The SD / OD status is connected also to the internal comparator status (Section 2.3.6:
Short-circuit protection and smart shutdown function). When the comparator triggers,
the SD / OD pin is pulled down acting as a FAULT pin.
•
The SD / OD, when pulled down by the comparator, is open drain configured. The SD /
OD voltage should be pulled up to the 3.3 V or 5 V logic power supply through a pull-up
resistor.
Thermistor
Pins: T1, T2
•
A co-packaged NTC is available for temperature monitor purposes.
•
A simple voltage divider (as shown in Section 2.3.10: Overtemperature protection) can
be realized with an external resistor in order to realize a temperature dependent
voltage signal.
•
The NTC is not able to sense IGBT junction temperature fast variation (due to its slow
dynamic).
Integrated operational amplifier (only for STGIPL14K60 and STGIPL20K60)
Pins: OP-U, OP-V, OP-W; OPOUTU, OPOUTV, OPOUTW; OPU, OPV, OPW
•
The op amps are completely uncommitted.
•
The op amps performances are optimized for advanced control technique (FOC).
•
Thanks to the integrated op amps it is possible to realize compact and efficient board
layout, minimizing the required BOM list.
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Positive DC-link
Pin: P
•
These are three DC-link positive power supply pins of the inverter, which offer
designers more flexibility in their approach. They are internally connected to the
collectors of the high-side IGBTs.
•
To suppress the surge voltage caused by the DC-link wiring or PCB pattern inductance,
connect decoupling capacitors close to this pin and power ground (typically, high
frequency, high voltage, non-inductive capacitors of about 0.1 or 0.22 μF are used).
Negative DC-link
Pins: NU, NV, NW
•
These are the DC-link negative power supply pins (power ground) of the inverter.
•
These pins are connected to the low side IGBT emitters of each phase.
•
The power ground of the application should be separated from the logic ground of the
system and they should be reconnected at one specific point (star connection).
Inverter power output
Pins: U, V, W
•
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Inverter output pins for connecting to the inverter load (e.g. motor).
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4
Power losses and dissipation
Power losses and dissipation
The total power losses in an inverter are comprised of conduction losses, switching losses,
and off-state losses and they are essentially generated by the power devices of the inverter
stage, such as the IGBTs and the freewheeling diodes. The conduction losses (Pcond) are
the on-state losses during the conduction phase. The switching losses (Psw) are the
dynamic losses encountered during the turn-on and the turn-off. The off-state losses, due to
the blocking voltage and leakage current, can be neglected.
Finally, the total power losses are given by:
Equation 17
Ptot ≈ Pcond + Psw
Figure 34 shows a typical waveform of an inductive hard switching application such as a
motor drive, where the major sources of power losses are specified.
Figure 34. Typical IGBT power losses
VCE
IC
10% VCE
10% IC
VCE(sat)
10% IC
Esw(off)
Esw(on)
tc(on)
10% VCE
conduction
tc(off)
AM09357v1
4.1
Conduction power losses
The conduction losses are caused by IGBT and freewheeling diode forward voltage drop at
rated current. They can be calculated using a linear approximation of the forward
characteristics for both IGBT and diode, having a series connection of DC voltage source
representing the threshold voltage, VTO for IGBT, (and VFO for diode) and a collector emitter
on-state resistance, RCE, (and anode cathode on-state resistance, RAK), as shown in
Figure 35, for reference.
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Power losses and dissipation
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Figure 35. IGBT and diode approximation of the output characteristics
AM09345v1
RAK = ΔVFM / ΔIFM
ΔIFM
RCE = ΔVCE / ΔIC
ΔIC
ΔVFM
ΔVCE
VTO
VFO
Both forward characteristics are temperature dependent, and so must be considered under
a specified temperature.
The linear approximations can be translated for IGBT in the following equation:
Equation 18
v ce(ic ) = VTO + RCE ⋅ ic
and, for freewheeling diode:
Equation 19
v fm (ifm ) = VFO + RAK ⋅ ifm
The conduction losses of IGBT and diode can be derived as the time integral of the product
of conduction current and voltage across the devices, as follows:
Equation 20
Pcond_IGBT =
1 T

T 0
vce ⋅ ic (t)dt =
1 T
2
 VTO ⋅ ic (t) + R ce ⋅ ic (t) dt

T 0

Equation 21
Pcond_Diode =
1 T
1 T
v f ⋅ if (t)dt =
 V ⋅ i (t) + R AK ⋅ i2f (t)  dt

T 0
T 0  FO f


where T is the fundamental period.
The different utilization mode of SLLIMM, modulation technique, and working conditions
make the power losses very difficult to estimate, it is therefore necessary to fix some starting
points.
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Power losses and dissipation
Assuming that:
1.
the application is a variable voltage variable frequency (VVVF) inverter based on
sinusoidal PWM technique
2.
the switching frequency is high and therefore the output currents are sinusoidal
3.
the load is ideal inductive.
Under these conditions, the output inverter current is given by:
Equation 22
i = Î cos(θ- φ )
where Î is the current peak, θ stands for ωt and φ is the phase angle between output voltage
and current.
The conduction power losses can be obtained as:
Equation 23
V ⋅ Î
Pcond_IGBT = TO
2π

π
+φ
2
π
− +φ
2
R ⋅ Î 2
ξ cos(θ - φ)dθ + CE
2π

π
+φ
2
ξ cos2 (θ - φ )dθ
π
− +φ
2
Equation 24
V Î
Pcond_Diode = FO
2π

π
+φ
2
(1 − ξ) cos(θ - φ)dθ + RAK
Î2
2π
π
− +φ
2

π
+φ
2
(1− ξ) cos2(θ - φ)dθ
π
− +φ
2
where ξ is the duty cycle for this PWM technique and is given by:
Equation 25
ξ=
1+ ma ⋅ cos θ
2
and ma is the PWM amplitude modulation index.
Finally, solving Equation 23 and Equation 24, we have:
Equation 26
m a ⋅ cos φ
m a ⋅ cos φ
1
2 1
P cond_IGBT = V TO ⋅ Î  ------ + ----------------------- + R CE ⋅ Î  --- + ---------------------- 2π

8

8
3π
Equation 27
m a ⋅ cos φ
m a ⋅ cos φ
1
2 1
P cond_Diode = VFO ⋅ Î  ------ + ----------------------- + R AK ⋅ Î  --- + ---------------------- 2π

8

8
3π
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Power losses and dissipation
AN3338
and therefore, the conduction power losses of one device (IGBT and diode) are:
Equation 28
Pcond = Pcond_IGBT + Pcond_Diode
Of course, the total conduction losses per inverter are six times this value.
4.2
Switching power losses
The switching loss is the power consumption during the turn-on and turn-off transients. As
already shown in Figure 34, it is given by the pulse of power dissipated during the turn-on
(ton) and turn-off (toff). Experimentally, it can be calculated by the time integral of product of
the collector current and collector-emitter voltage for the switching period. Anyway, the
dynamic performances are strictly related to many parameters such as voltage and current,
temperature, so it is necessary to use the same assumptions of conduction power losses
(Section 4.1: Conduction power losses) to simplify the calculations.
Under these conditions, the switching energy losses are given by:
Equation 29
Eon(θ) = Êon cos(θ- φ )
Equation 30
Eoff ( θ) = Ê off cos(θ- φ)
where Êon and Êoff are the maximum values taken at Tjmax and Îc, θ stands for ωt and φ is
the phase angle between output voltage and current.
Finally, the switching power losses per device depend on the switching frequency (fsw) and
are calculated as follows:
Equation 31
Psw =
1
2π

π
+φ
2
(EIGBT + EDiode) ⋅ fswdθ =
π
- +φ
2
(EIGBT + EDiode) ⋅ fsw
π
where EIGBT and EDiode are the total switching energy for IGBT and freewheeling diode,
respectively. Also in this case, the total switching losses per inverter are six times this value.
Figure 36 shows the real turn-on and turn-off waveforms of STGIPL14K60 under the
following conditions:
•
VPN = 300 V, IC = 7 A, Tj = 25 °C with inductive load on full bridge topology, taken on
the high side IGBT.
The red plots represent instantaneous power as a result of IC (in blue) and VCE (in green)
waveforms multiplication, during the switching transitions. The areas under these plots are
the switching energies computed by graphic integration thanks to the digital oscilloscope.
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Power losses and dissipation
Figure 36. Typical switching waveforms of STGIPL14K60
ton = 264ns
Turn on
Turn off
toff = 332ns
STGIPL14K60
High side
Tj=25rC
STGIPL14K60
High side
Tj=25rC
VHIN
VHIN
IC
IC
VCE
VCE
VHIN = 5V/Div
Eon=142μJ(*)
VCE = 100V/Div
Eoff=100μJ(*)
IC = 2A/Div
Eon = ∫ (VCE · IC) dt
t = 80ns/Div
!-V
Eon and Eoff are the areas under the red plots.
4.3
Thermal impedance overview
During operation, power losses generate heat which elevates the temperature in the
semiconductor junctions contained in the SLLIMM, limiting its performance and lifetime. To
ensure safe and reliable operation, the junction temperature of power devices must be kept
below the limits defined in the datasheet, therefore, the generated heat must be conducted
away from the power chips and into the environment using an adequate cooling system.
The most common schemes are based on one heatsink designed for free conventional air
flow or, in some cases, for forced air operation. Free conventional air flow systems require
bigger heatsinks (about 50% more) than a forced air based heatsink, for a given thermal
resistance. Therefore, the choice of the cooling system becomes the starting point for the
application designer and the thermal aspect of the system is one of the key factors in
designing high efficiency and high reliability equipment. In this environment the package
and its thermal resistance play a fundamental role.
Thermal resistance quantifies the capability of a given thermal path to transfer heat in the
steady-state and it generically is given as the ratio between the temperature increase above
the reference and the relevant power flow:
Equation 32
Rth =
ΔT
ΔP
The thermal resistance specified in the datasheet is the junction-case Rth(j-c) which is
defined as the difference in temperature between junction and case reference divided by the
power dissipation per device:
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Equation 33
R th(j-c) =
Tj − Tc
PD
The SLLIMM family benefits from the state of the art DBC substrate and therefore offers a
very low Rth(j-c) value. The backside of the DBC substrate is used as the cooling interface to
the heatsink. Thermal grease or another thermal interface material between the DBC and
the heatsink is used to reduce the thermal resistance of the interface (Rth(c-h)) and, of
course, it depends of the material and its thickness.
Basically, the sum of the three thermal resistance components mentioned above gives the
thermal resistance between junction and ambient Rth(j-a), as shown in Figure 37.
Figure 37. Equivalent thermal circuit with heatsink single IGBT
SLLIMM
Junction
Rth(j-c)
Power Stage
Tj
Case
IGBT
FWD
DBC
Heatsink
Ptot
Rth(c-h)
Rth(h-a)
Tc
Th
Ambient
Tamb
!-V
As the power loss Ptot is cyclic, also the transient thermal impedance must be considered. It
is defined as the ratio between the time dependent temperature increase above the
reference, ΔT(t), and the relevant heat flow:
Equation 34
Zth(t) =
ΔT(t )
ΔP
Contrary to that already seen, regarding the thermal resistance, the thermal impedance is
typically represented by an RC equivalent circuit. For pulsed power loss, the thermal
capacitance effect delays the rise in junction temperature and therefore the advantage of
this behavior is the short-term overload capability of the SLLIMM.
For example, Figure 38 shows thermal impedance from junction to case curves of
STGIPS14K60 (in SDIP-25L package) and STGIPL14K60 (in SDIP-38L package). As per all
the other SLLIMM curves, the thermal impedance reaches saturation in about 10 seconds.
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Power losses and dissipation
Figure 38. Thermal impedance curves (STGIPS14K60 and STGIPL14K60)
STGIPL14K60
4
3
3
Zth(j-c) (°C/W)
Zth(j-c) (°C/W)
STGIPS14K60
4
2
2
1
1
0
1.E-05
AM09348v1
1.E-04
1.E-03
1.E-02
1.E-01
1.E+00
1.E+01
0
1.E-05
1.E+02
1.E-04
1.E-03
1.E-02
1.E-01
1.E+00
1.E+01
1.E+02
time (sec)
time (sec)
More generally, in the case of the device, power is time dependent too. The device
temperature can be calculated by using the convolution integral method applied to Equation
34Equation 34, as follows:
Equation 35
Δ T( t) =
t
Z
0
th (t − τ) ⋅ P( τ)dτ
An alternative method, very useful for the simulator tools, is the transient thermal
impedance model, which provides a simple method to estimate the junction temperature
rise under a transient condition.
By using the thermo-electrical analogy, the transient thermal impedance Zth(t) can be
transformed into an electrical equivalent RC network. The number of RC sections increases
the model details, therefore a ninth order model, based on the Cauer network, has been
used in order to improve the accuracy of the model, as shown the Figure 39.
Figure 39. Thermal impedance RC Cauer thermal network
7M
3WRW W
=WK W
R1
C1
R2
C2
R3
C3
R4
C4
R5
C5
R6
C6
R7
C7
R8
C8
R9
C9
7FDVH
!-V
Temperatures inside the electrical RC network represent voltages, power flows represent
currents, electrical resistances and capacitances represent thermal resistances and
capacitances respectively. The case temperature is represented with a DC voltage source
and can be interpreted as the initial junction temperature.
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Transient thermal impedance models are derived by curve fitting an equation to the
measured data. Values for the individual resistors and capacitors are the variables from that
equation and are defined device by device in Table 13.
Table 13. RC Cauer thermal network elements by device
Element
STGIPS10K60A
STGIPS14K60
STGIPL14K60
STGIPS20K60
STGIPL20K60
R1 (°C/W)
8.80E-02
1.61E-02
8.15E-03
1.00E-04
3.85E-03
R2 (°C/W)
1.54E-02
9.42E-02
1.07E-01
5.00E-03
1.50E-02
R3 (°C/W)
3.16E-01
1.20E-02
5.00E-02
7.00E-02
5.17E-03
R4 (°C/W)
3.96E-03
3.50E-01
2.00E-01
1.03E-02
4.68E-02
R5 (°C/W)
8.16E-01
5.86E-01
6.57E-01
6.00E-01
4.18E-01
R6 (°C/W)
4.32E-01
1.58E-03
1.00E-02
1.15E-01
6.71E-02
R7 (°C/W)
1.23E-02
7.50E-01
8.00E-01
1.00E-03
1.27E-01
R8 (°C/W)
4.48E-01
1.60E-02
2.00E-02
1.00E-01
6.14E-01
R9 (°C/W)
1.66E+00
1.10E+00
9.50E-01
1.50E+00
0.90E+00
C1 (W·sec/°C)
3.20E-04
9.20E-04
1.00E-03
1.80E-03
1.50E-03
C2 (W·sec/°C)
6.30E-04
9.07E-05
9.96E-05
3.09E-05
9.82E-05
C3 (W·sec/°C)
9.00E-05
1.00E-03
9.59E-05
8.94E-05
9.62E-05
C4 (W·sec/°C)
5.00E-04
4.14E-05
1.85E-05
9.29E-05
9.48E-05
C5 (W·sec/°C)
5.00E-03
1.40E-02
9.68E-03
1.20E-02
9.97E-03
C6 (W·sec/°C)
1.20E-02
3.57E-05
2.00E-02
7.04E-05
7.86E-05
C7 (W·sec/°C)
1.49E-03
3.00E-03
1.76E-03
2.93E-04
2.91E-03
C8 (W·sec/°C)
8.09E-04
5.75E-04
8.27E-04
9.43E-04
5.50E-02
C9 (W·sec/°C)
1.20E-01
1.54E-01
5.00E-01
1.00E-01
6.21E-02
4.4
Power losses calculation example
As a result of power loss calculation and thermal aspects, fully treated in the previous
sections, we are able to simulate the maximum IC(RMS) current versus switching frequency
curves for a VVVF inverter using a 3-phase continuous PWM modulation to synthesize
sinusoidal output currents.
The curves graphed in Figure 40 represent the maximum current managed by SLLIMM in
safety conditions, when the junction temperature rises to the maximum junction temperature
of 150 °C and case temperature is 100 °C, which is a typical operating condition to
guarantee the reliability of the system. These curves, functions of the motor drive typology
and control scheme, are simulated under the following conditions:
•
60/72
VPN = 300 V, ma = 0.8, cos = 0.6, Tj = 150 °C, Tc = 100 °C, fSINE = 60 Hz, max. value of
Rth(j-c), typical VCE(sat) and Etot values.
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AN3338
Power losses and dissipation
Figure 40. Maximum IC(RMS) current vs. fsw simulated curves
!-V
22
Maximum I C(RMS) current (A)
20
18
16
14
12
10
8
6
4
4
8
12
16
20
fsw (kHz)
STGIPS10K60A
STGIPS14K60
STGIPL14K60
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STGIPS20K60
STGIPL20K60
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Design and mounting guidelines
5
AN3338
Design and mounting guidelines
In this section the main layout suggestions for an optimized design and major mounting
recommendations, to appropriately handle and assemble the SLLIMM family, are
introduced.
5.1
Layout suggestions
Optimization of PCB layout for high voltage, high current and high switching frequency
applications is a critical point. PCB layout is a complex matter as it includes several aspects,
such as length and width of track and circuit areas, but also the proper routing of the traces
and the optimized reciprocal arrangement of the various system elements in the PCB area.
A good layout can help the application to properly function and achieve expected
performance. On the other hand, PCB without a careful layout can generate EMI issues
(both induced and perceived by the application), can provide overvoltage spikes due to
parasitic inductances along the PCB traces, and can produce higher power loss and even
malfunction in the control and sensing stages.
The compactness of the SLLIMM solution, which offers optimized gate driving network and
reduced parasitic elements, allows designers to focus only on some specific issues, such as
the ground issue or noise filter. Anyhow, in order to avoid all the aforementioned conditions,
the following general guidelines and suggestions must be followed in PCB layout for 3phase applications.
5.1.1
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General suggestions
•
PCB traces should be designed to be as short as possible and the area of the circuit
(power or signal) should be minimized to avoid the sensitivity of such structures to
surrounding noise.
•
Ensure a good distance between switching lines with high voltage transitions and the
signal line sensitive to electrical noise. Specifically, the tracks of each OUT phase,
bringing significant currents and high voltages, should be separated from the logic lines
and analog sensing circuit of op amps and comparators.
•
Place the RSENSE resistors as close as possible to the low side pins of the SLLIMM
(NU, NV and NW). Parasitic inductance can be minimized by connecting the ground line
(also called driver ground) of the SLLIMM directly to the cold terminal of sense
resistors. Use of a low inductance type resistor, such as an SMD resistor instead of
long-lead type resistors, can help to further decrease the parasitic inductance.
•
Avoid any ground loop. Only a single path must connect two different ground nodes.
•
Place each RC filter as close as possible to the SLLIMM pins in order to increase their
efficiency.
•
In order to prevent surge destruction, the wiring between the decoupling capacitor and
the P pin and power ground should be as short as possible. The use of a high
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Design and mounting guidelines
frequency, high voltage non-inductive capacitor about 0.1 or 0.22 µF between the P
and N pins is recommended.
•
Fixed voltage tracks, such as GND or HV lines, can be used to shield the logic and
analog lines from the electrical noise produced by the switching lines (e.g. OUTU,
OUTV and OUTW).
•
Generally it is recommended to connect each half bridge ground in a star configuration
and the three RSENSE very close to each other and to the power ground.
In Figure 41 and Figure 42 the general suggestions for all SLLIMM products are
summarized.
Figure 41. General suggestions 1
C onnect all s ignal ground
together and then connect the
s ignal ground and power
ground at only one point
W iring between NU,NV,NW and
s hunt res is tor s hould be as
s hort as pos s ible
Dec oupling
c apac itor
P
OUT U
L IN U
HIN U
VCC
NU
+
P
OUT V
-
V B OOT V
V
P
W
P ower G ND (N)
Layer 1
NW
to
VIN(UL)
VIN(UH) MCU
+15V from
power
G ND s ource
G ND
L IN V
HIN V
NV
S hunt
res is tor
Isolation dis tances between
high voltage block (bootstrap)
and low voltage block mus t be
kept
V B OOT U
U
B us
c apac itor
to motor
Boots trap capacitor s hould be
located as close as pos s ible to
the SLLIMM pins
to
VIN(VL)
VIN(VH) MCU
OUT W
V B OOT W
L IN W
HIN W
S D/OD
C IN
VIN(WL) to
VIN(WH) MCU
+3.3V
S hutdown/Fault
Layer 2
Use of low inductance type res is tor,
s uch as S MD res is tor ins tead of
long-lead type res is tor, can help to
further decreas e the paras itic
inductance
S ignal ground and power ground mus t be
connected at only one point (star connections)
avoiding long connections. P le as e ens ure a s afety
dis tances between ground tracks and noisy tracks
(high voltage or high frequency s ignals tracks)
C IN connections
mus t be as s hort
as pos s ible
C onnect C IN filter
capacitor to s ignal
ground. T his
connection s hould be
as s hort as pos s ible
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Figure 42. General suggestions 2
Use of low inductance type
res is tor, s uch as S MD one,
can help to further
decreas e the paras itic
inductance
R educe all
dis tances between
s hunt res is tors and
SLLIMM power
G ND
Isolation dis tances
between high voltage
block (bootstrap) and
low voltage block
mus t be kept
Boots trap capacitor
s hould be located
as close as
pos s ible to the
SLLIMM pins
OUT U
V B OOT U
P
L IN U
HIN U
OP -U
OP OUT U
OP +U
C IN U
U
+
NU
to motor
-
L IN V
HIN V
OP -V
OP OUT V
OP +V
C IN V
V
Dec oupling
c apac itor
S hunt
res is tors
Layer 1
P hase current U
to
VIN(VL)
VIN(VH) MCU
P hase current V
OUT W
V B OOT W
NV
L IN W
HIN W
OP -W
OP OUT W
OP +W
C IN W
VCC
S D/OD
G ND
T2
T1
P
W
P ower G ND (N)
VIN(UL) to
VIN(UH) MCU
OUT V
V B OOT V
P
B us
c apac itor
P la ce the S MD components as
close as pos s ible the op amp pins.
K eep a s afety dis tance between
noisy tracks and op amp C IN tracks
NW
VIN(WL) to
VIN(WH) MCU
P hase current W
+15V
G ND
+3.3V
S hutdown/Fault
T emp feedback
Layer 2
S ignal ground and power ground mus t be connected
at only one point (star connections), avoiding long
connections. P le as e ens ure a s afety dis tances
between ground tracks and noisy tracks (high
voltage or high frequency s ignals tracks)
P la ce an R C filter
directly acros s the
C IN (for each phase)
pin to avoid false
s hort-circuit trigger
C onnect all the s ignal
ground together and
after this, connect them
to the power ground at
only one point
P la ce an
R C filter
directly
acros s S D
pin
NTC will
provide a
temperature
feedback to
the MCU
AM09352v1
Special attention must be paid to some wrong layouts. In Figure 43 and Figure 44 some
common PCB mistakes are shown.
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Design and mounting guidelines
Figure 43. Example 1 of a possible wrong layout
W RONG !
Decoupling capacitor is too far from
SLLIMM. C onnect it as close as
pos s ible to the P pin
W RONG !
R ight-angled track turns produce a
field concentration at the inner edge.
P refer 45 angled tracks
P
W RONG !
S tub connections and vias produce reflections ,
especially on critical s ignal tracks. P refer s tar
connections and reduce number of vias
OUT U
V B OOT U
U
Dec oupling
c apac itor
P
B us
c apac itor
to motor
L IN U
HIN U
VCC
NU
OUT V
V B OOT V
V
+
-
NV
P
Layer 1
S hunt
res is tor
+15V from
power
G ND s ource
G ND
L IN V
HIN V
C IN
filter
NW
to
VIN(VL)
VIN(VH) MCU
OUT W
V B OOT W
L IN W
HIN W
S D/OD
W
P ower G ND (N)
to
VIN(UL)
VIN(UH) MCU
C IN
VIN(WL) to
VIN(WH) MCU
+3.3V
S hutdown/Fault
Layer 2
W RONG !
C IN filter is close to high voltage
s witching track (W pin). Noise will
influence comparator performances
W RONG !
C IN filter ground is not the
s ame as per SLLIMM ground.
T his may cause noise
W RONG !
Long dis tance between C IN filter and SLLIMM C IN
pin. It is important to minimize this dis tance in
order to reduce the noise impact
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Figure 44. Example 2 of a possible wrong layout
Ground path
WRONG!
Very large ground loop.
Does not used the suggested star
connection.
Long ground path could be affected
by noise (due to high voltage
switching tracks) and could affect
driver (or application) performance
Sense resistor
cold terminal
WRONG!
The cold terminal of the sense
resistor is not chosen as star centre
SLLIMM
ground
Bulk
capacitor
WRONG!
Connection between the SLLIMM
and ground is not minimized
SLLIMM
!-V
5.2
Mounting instructions
The purpose of the mounting instructions is to define some basic assembly rules in order to
limit thermal and mechanical stresses or assure the best thermal conduction and electrical
isolation of both SDIP-25L and SDIP-38L packages when mounting on a heatsink.
For further details please refer to the TN0107 technical note.
5.2.1
Heatsink mounting
The following precautions should be observed to maximize the effect of the heatsink and
minimize stresses on the device. Smooth the surface by removing burrs and protrusions; it
is essential to ensure an optimal contact between the SLLIMM and the heatsink.
Apply a uniform layer of silicon grease, from 100 µm up to 200 µm of thickness, between the
device and the heatsink to reduce the contact thermal resistance, as shown in Figure 45. Be
sure to apply the coating thinly and evenly, paying attention to not having any voids
remaining on the contact surface between the SLLIMM and the heatsink. We recommend
using high quality grease with stable performance within the operating temperature range of
the SLLIMM.
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Design and mounting guidelines
Figure 45. Recommended silicon grease thickness and positioning
SLLIMM
6LOLFRQ
JUHDVH
100~200˩m
thickness
Heatsink
!-V
5.2.2
Mounting torque
While mounting the SLLIMM to a heatsink make sure not to apply excessive force during the
assembly. Table 14 provides the specified fastening torque. Inappropriate mounting can
damage the device and over tightening the screws may cause DBC substrate or molding
compound cracks. Avoid mechanical stress due to tightening on one side only. It is
recommended to temporarily fasten both screws, then fasten them permanently to the
specified torque value using a torque wrench. Figure 47 shows the screw fastening order.
Table 14. Mounting torque and heatsink flatness
Limits
Parameter
Units
Min.
Typ.
Max.
Mounting torque (M3 screw)
0.4
0.7
1.0
Nm
Heatsink flatness
-50
150
µm
SDIP-25L package weight
13
g
SDIP-38L package weight
17
g
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Figure 46. Measurement point of Cu heatsink flatness
#U HEAT SINK SURFACE
4OP VIEW
!-V
Figure 47. Recommended fastening order of mounting screws
4EMPORARY FASTENING
0ERMANENT FASTENING
!-V
5.2.3
General handling precaution and storage notices
The incidence of thermal and/or mechanical stress to the semiconductor devices due to
improper handling may result in significant deterioration of their electrical characteristics
and/or reliability.
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Design and mounting guidelines
The SLLIMM is an ESD sensitive device, it may be damaged in the case of ESD shocks. All
equipment used to handle power modules must comply with ESD standards including
transportation, storage, and assembly.
Transportation
Be careful when handling the SLLIMM and packaging material. Ensure that the module is
not subjected to mechanical vibration or shock during transport. Do not toss or drop to
ensure the SLLIMM is correctly functioning before boarding. Wet conditions are dangerous
and moisture can also adversely affect the packaging. Hold the package avoiding touching
the leads during mounting. Put package boxes upside down, leaning them or giving them
uneven stress may cause the terminals to be deformed or the resin to be damaged.
Throwing or dropping the packaging boxes may cause the modules to be damaged. Wetting
the packaging boxes may cause the breakdown of modules when operating. Pay particular
care when transporting in wet conditions.
Storage
•
Do not force or load the external pressure to the modules while they are in storage
•
Humidity should be kept within the range of 40% to 75%, the temperature should not go
over 35 °C or below 5 °C
•
Lead solder ability is degraded by lead oxidation or corrosion. So using storage areas
where there is minimal temperature fluctuation is highly recommended
•
The presence of harmful gases or dusty conditions is not acceptable for storage.
•
Use antistatic containers
Electrical shock and thermal injury
•
Do not touch either module or heatsink when SLLIMM is operating to avoid sustaining
an electrical shock and/or a burn injury.
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References
6
AN3338
References
1.
STGIPS10K60A datasheet
2.
STGIPS14K60 datasheet
3.
STGIPL14K60 datasheet
4.
STGIPS20K60 datasheet
5.
STGIPL20K60 datasheet
6.
AN2738 application note
7.
UM0969 user manual
8.
UM0900 user manual
9.
UM1036 user manual
10. Minimum-Loss Strategy for Three-Phase PWM Rectifier, IEEE, JUNE 1999
11. TN0107 technical note.
Note:
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SLLIMM™ and PowerMESH™ are trademarks of STMicroelectronics.
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7
Revision history
Revision history
Table 15. Document revision history
Date
Revision
21-Mar-2011
1
Initial release.
12-Jul-2011
2
Modified: RthJC Table 1 on page 9, Figure 32 on page 47 and
Heatsink flatness max. value Table 14 on page 67.
17-Sep-2012
3
Updated: Figure 4 on page 10, Figure 18 on page 30, Figure 41 on
page 63, Figure 42 on page 64 and Figure 43 on page 65.
4
Updated product features in Section 1.1
Added footnote to Table 1
Updated figures: Figure 4, Figure 14, Figure 16, Figure 22,
Figure 23, Figure 30, Figure 31
Removed footnote from Table 5
Updated Table 9, Table 10
In Section 3.5, updated Positive DC-link pin details
Updated Equations 26 and 27
Updated Table 13
Removed Section 5.2.4 Packaging specifications
17-Mar-2015
Changes
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© 2015 STMicroelectronics – All rights reserved
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