LMR12007
www.ti.com
SNVS982 – SEPTEMBER 2013
LMR12007 Thin SOT23 750mA Load Step-Down DC-DC Regulator
Check for Samples: LMR12007
FEATURES
DESCRIPTION
•
•
•
•
•
The LMR12007 regulator is a monolithic, high
frequency, PWM step-down DC/DC converter in a 6pin Thin SOT package. It provides all the active
functions to provide local DC/DC conversion with fast
transient response and accurate regulation in the
smallest possible PCB area.
1
23
•
•
•
•
•
•
•
Thin SOT-6 Package
3.0V to 18V Input Voltage Range
1.25V to 16V Output Voltage Range
750mA Output Current
550kHz (LMR12007Y) and 1.6MHz (LMR12007X)
Switching Frequencies
350mΩ NMOS Switch
30nA Shutdown Current
1.25V, 2% Internal Voltage Reference
Internal Soft-Start
Current-Mode, PWM Operation
WEBENCH® Online Design Tool
Thermal Shutdown
APPLICATIONS
•
•
•
•
•
•
•
Local Point of Load Regulation
Core Power in HDDs
Set-Top Boxes
Battery Powered Devices
USB Powered Devices
DSL Modems
Notebook Computers
With a minimum of external components and online
design support through WEBENCH®, the LMR12007
is easy to use. The ability to drive 750mA loads with
an internal 350mΩ NMOS switch using state-of-theart 0.5µm BiCMOS technology results in the best
power density available. The world class control
circuitry allows for on-times as low as 13ns, thus
supporting exceptionally high frequency conversion
over the entire 3V to 18V input operating range down
to the minimum output voltage of 1.25V. Switching
frequency is internally set to 550kHz (LMR12007Y) or
1.6MHz (LMR12007X), allowing the use of extremely
small surface mount inductors and chip capacitors.
Even though the operating frequencies are very high,
efficiencies up to 90% are easy to achieve. External
shutdown is included, featuring an ultra-low stand-by
current of 30nA. The LMR12007 utilizes current-mode
control and internal compensation to provide highperformance regulation over a wide range of
operating conditions. Additional features include
internal soft-start circuitry to reduce inrush current,
pulse-by-pulse current limit, thermal shutdown, and
output over-voltage protection.
Typical Application Circuit
Efficiency vs Load Current "X"
VIN = 5V, VOUT = 3.3V
D2
VIN
BOOST
VIN
C3
C1
L1
SW
ON
VOUT
D1
EN
C2
R1
OFF
FB
GND
R2
1
2
3
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
WEBENCH is a registered trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2013, Texas Instruments Incorporated
LMR12007
SNVS982 – SEPTEMBER 2013
www.ti.com
Connection Diagram
BOOST
1
6
SW
1
6
GND
2
5
VIN
2
5
FB
3
4
EN
3
4
Figure 1. 6-Lead SOT
See Package Number DDC (R-PDSO-G6)
Figure 2. Pin 1 Indentification
PIN DESCRIPTIONS
Pin
Name
1
BOOST
Function
2
GND
3
FB
Feedback pin. Connect FB to the external resistor divider to set output voltage.
4
EN
Enable control input. Logic high enables operation. Do not allow this pin to float or be greater than VIN + 0.3V.
5
VIN
Input supply voltage. Connect a bypass capacitor to this pin.
6
SW
Output switch. Connects to the inductor, catch diode, and bootstrap capacitor.
Boost voltage that drives the internal NMOS control switch. A bootstrap capacitor is connected between the
BOOST and SW pins.
Signal and Power ground pin. Place the bottom resistor of the feedback network as close as possible to this pin
for accurate regulation.
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
Absolute Maximum Ratings (1)
VIN
-0.5V to 22V
SW Voltage
-0.5V to 22V
Boost Voltage
-0.5V to 28V
Boost to SW Voltage
-0.5V to 6.0V
FB Voltage
-0.5V to 3.0V
EN Voltage
-0.5V to (VIN + 0.3V)
Junction Temperature
150°C
ESD Susceptibility (2)
2kV
Storage Temp. Range
Soldering Information
(1)
(2)
2
-65°C to 150°C
Infrared/Convection Reflow (15sec)
220°C
Wave Soldering Lead Temp. (10sec)
260°C
If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and
specifications.
Human body model, 1.5kΩ in series with 100pF.
Submit Documentation Feedback
Copyright © 2013, Texas Instruments Incorporated
Product Folder Links: LMR12007
LMR12007
www.ti.com
SNVS982 – SEPTEMBER 2013
Operating Ratings (1)
VIN
3V to 18V
SW Voltage
-0.5V to 18V
Boost Voltage
-0.5V to 23V
Boost to SW Voltage
1.6V to 5.5V
−40°C to +125°C
Junction Temperature Range
Thermal Resistance θJA (2)
(1)
(2)
118°C/W
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is intended to be functional, but specific performance is not ensured. For specific specifications and the test conditions,
see Electrical Characteristics.
Thermal shutdown will occur if the junction temperature exceeds 165°C. The maximum power dissipation is a function of TJ(MAX) , θJA
and TA . The maximum allowable power dissipation at any ambient temperature is PD = (TJ(MAX) – TA)/θJA . All numbers apply for
packages soldered directly onto a 3” x 3” PC board with 2oz. copper on 4 layers in still air. For a 2 layer board using 1 oz. copper in still
air, θJA = 204°C/W.
Electrical Characteristics
Specifications with standard typeface are for TJ = 25°C, and those in boldface type apply over the full Operating
Temperature Range (TJ = -40°C to 125°C). VIN = 5V, VBOOST - VSW = 5V unless otherwise specified. Datasheet min/max
specification limits are ensured by design, test, or statistical analysis.
Symbol
VFB
ΔVFB/ΔVIN
IFB
UVLO
Parameter
Conditions
Feedback Voltage
Feedback Voltage Line Regulation
VIN = 3V to 18V
Feedback Input Bias Current
Sink/Source
Undervoltage Lockout
VIN Rising
Undervoltage Lockout
VIN Falling
UVLO Hysteresis
FSW
Switching Frequency
DMAX
Maximum Duty Cycle
DMIN
Minimum Duty Cycle
RDS(ON)
Typ (2)
Max (1)
Units
1.225
1.250
1.275
V
0.01
%/V
10
250
2.74
2.90
2.0
2.3
0.30
0.44
LMR12007X
1.2
1.6
1.9
LMR12007Y
0.40
0.55
0.66
LMR12007X
85
92
LMR12007Y
90
96
LMR12007X
2
LMR12007Y
1
0.62
%
VBOOST - VSW = 3V
Switch Current Limit
VBOOST - VSW = 3V
IQ
Quiescent Current
Switching
Quiescent Current (shutdown)
VEN = 0V
30
LMR12007X (50% Duty Cycle)
2.2
3.3
LMR12007Y (50% Duty Cycle)
0.9
1.6
Boost Pin Current
Shutdown Threshold Voltage
VEN Falling
Enable Threshold Voltage
VEN Rising
IEN
Enable Pin Current
Sink/Source
ISW
Switch Leakage
VEN_TH
MHz
%
Switch ON Resistance
1.0
nA
V
ICL
IBOOST
(1)
(2)
Min (1)
350
650
1.5
2.3
mΩ
A
1.5
2.5
mA
nA
0.4
1.8
mA
V
10
nA
40
nA
Specified to Texas Instruments' Average Outgoing Quality Level (AOQL).
Typicals represent the most likely parametric norm.
Submit Documentation Feedback
Copyright © 2013, Texas Instruments Incorporated
Product Folder Links: LMR12007
3
LMR12007
SNVS982 – SEPTEMBER 2013
www.ti.com
Typical Performance Characteristics
All curves taken at VIN = 5V, VBOOST - VSW = 5V, L1 = 4.7 µH ("X"), L1 = 10 µH ("Y"), and TA = 25°C, unless specified
otherwise.
4
Efficiency vs Load Current - "X" VOUT = 5V
Efficiency vs Load Current - "Y" VOUT = 5V
Figure 3.
Figure 4.
Efficiency vs Load Current - "X" VOUT = 3.3V
Efficiency vs Load Current - "Y" VOUT = 3.3V
Figure 5.
Figure 6.
Efficiency vs Load Current - "X" VOUT = 1.5V
Efficiency vs Load Current - "Y" VOUT = 1.5V
Figure 7.
Figure 8.
Submit Documentation Feedback
Copyright © 2013, Texas Instruments Incorporated
Product Folder Links: LMR12007
LMR12007
www.ti.com
SNVS982 – SEPTEMBER 2013
Typical Performance Characteristics (continued)
All curves taken at VIN = 5V, VBOOST - VSW = 5V, L1 = 4.7 µH ("X"), L1 = 10 µH ("Y"), and TA = 25°C, unless specified
otherwise.
Oscillator Frequency vs Temperature - "X"
Oscillator Frequency vs Temperature - "Y"
Figure 9.
Figure 10.
Current Limit vs Temperature VIN = 18V, VIN = 5V
VFB vs Temperature
Figure 11.
Figure 12.
RDSON vs Temperature
IQ Switching vs Temperature
Figure 13.
Figure 14.
Submit Documentation Feedback
Copyright © 2013, Texas Instruments Incorporated
Product Folder Links: LMR12007
5
LMR12007
SNVS982 – SEPTEMBER 2013
www.ti.com
Typical Performance Characteristics (continued)
All curves taken at VIN = 5V, VBOOST - VSW = 5V, L1 = 4.7 µH ("X"), L1 = 10 µH ("Y"), and TA = 25°C, unless specified
otherwise.
6
Line Regulation - "X" VOUT = 1.5V, IOUT = 500mA
Line Regulation - "Y" VOUT = 1.5V, IOUT = 500mA
Figure 15.
Figure 16.
Line Regulation - "X" VOUT = 3.3V, IOUT = 500mA
Line Regulation - "Y" VOUT = 3.3V, IOUT = 500mA
Figure 17.
Figure 18.
Submit Documentation Feedback
Copyright © 2013, Texas Instruments Incorporated
Product Folder Links: LMR12007
LMR12007
www.ti.com
SNVS982 – SEPTEMBER 2013
Block Diagram
VIN
VIN
Current-Sense Amplifier
OFF
EN
Internal
Regulator
and
Enable
Circuit
RSENSE
+
-
CIN
D2
Thermal
Shutdown
BOOST
VBOOST
Under
Voltage
Lockout
Current
Limit
Oscillator
Output
Control
Logic
Reset
Pulse
+
ISENSE
+
+
Corrective Ramp
0.3:
Switch
Driver
SW
OVP
Comparator
-
ON
Error
Signal
D
1
+
PWM
Comparator
CBOOST
VSW L
IL
VOUT
COUT
1.375V
+
-
R1
FB
Internal
Compensation
+
Error Amplifier
+
-
VREF
1.25V
R2
GND
Figure 19.
APPLICATION INFORMATION
THEORY OF OPERATION
The LMR12007 is a constant frequency PWM buck regulator IC that delivers a 750mA load current. The
regulator has a preset switching frequency of either 550kHz (LMR12007Y) or 1.6MHz (LMR12007X). These high
frequencies allow the LMR12007 to operate with small surface mount capacitors and inductors, resulting in
DC/DC converters that require a minimum amount of board space. The LMR12007 is internally compensated, so
it is simple to use, and requires few external components. The LMR12007 uses current-mode control to regulate
the output voltage.
The following operating description of the LMR12007 will refer to the Simplified Block Diagram (Figure 19) and to
the waveforms in Figure 20. The LMR12007 supplies a regulated output voltage by switching the internal NMOS
control switch at constant frequency and variable duty cycle. A switching cycle begins at the falling edge of the
reset pulse generated by the internal oscillator. When this pulse goes low, the output control logic turns on the
internal NMOS control switch. During this on-time, the SW pin voltage (VSW) swings up to approximately VIN, and
the inductor current (IL) increases with a linear slope. IL is measured by the current-sense amplifier, which
generates an output proportional to the switch current. The sense signal is summed with the regulator’s
corrective ramp and compared to the error amplifier’s output, which is proportional to the difference between the
feedback voltage and VREF. When the PWM comparator output goes high, the output switch turns off until the
next switching cycle begins. During the switch off-time, inductor current discharges through Schottky diode D1,
which forces the SW pin to swing below ground by the forward voltage (VD) of the catch diode. The regulator
loop adjusts the duty cycle (D) to maintain a constant output voltage.
Submit Documentation Feedback
Copyright © 2013, Texas Instruments Incorporated
Product Folder Links: LMR12007
7
LMR12007
SNVS982 – SEPTEMBER 2013
www.ti.com
VSW
D = TON/TSW
VIN
SW
Voltage
TOFF
TON
0
VD
t
TSW
IL
IPK
Inductor
Current
t
0
Figure 20. LMR12007 Waveforms of SW Pin Voltage and Inductor Current
BOOST FUNCTION
Capacitor CBOOST and diode D2 in Figure 21 are used to generate a voltage VBOOST. VBOOST - VSW is the gate
drive voltage to the internal NMOS control switch. To properly drive the internal NMOS switch during its on-time,
VBOOST needs to be at least 1.6V greater than VSW. Although the LMR12007 will operate with this minimum
voltage, it may not have sufficient gate drive to supply large values of output current. Therefore, it is
recommended that VBOOST be greater than 2.5V above VSW for best efficiency. VBOOST – VSW should not exceed
the maximum operating limit of 5.5V.
5.5V > VBOOST – VSW > 2.5V for best performance.
VBOOST
D2
BOOST
VIN
VIN
CIN
CBOOST
L
SW
VOUT
GND
D1
COUT
Figure 21. VOUT Charges CBOOST
When the LMR12007 starts up, internal circuitry from the BOOST pin supplies a maximum of 20mA to CBOOST.
This current charges CBOOST to a voltage sufficient to turn the switch on. The BOOST pin will continue to source
current to CBOOST until the voltage at the feedback pin is greater than 1.18V.
There are various methods to derive VBOOST:
1. From the input voltage (VIN)
2. From the output voltage (VOUT)
3. From an external distributed voltage rail (VEXT)
4. From a shunt or series zener diode
In the Simplifed Block Diagram of Figure 19, capacitor CBOOST and diode D2 supply the gate-drive current for the
NMOS switch. Capacitor CBOOST is charged via diode D2 by VIN. During a normal switching cycle, when the
internal NMOS control switch is off (TOFF) (refer to Figure 20), VBOOST equals VIN minus the forward voltage of D2
(VFD2), during which the current in the inductor (L) forward biases the Schottky diode D1 (VFD1). Therefore the
voltage stored across CBOOST is
VBOOST - VSW = VIN - VFD2 + VFD1
8
(1)
Submit Documentation Feedback
Copyright © 2013, Texas Instruments Incorporated
Product Folder Links: LMR12007
LMR12007
www.ti.com
SNVS982 – SEPTEMBER 2013
When the NMOS switch turns on (TON), the switch pin rises to
VSW = VIN – (RDSON x IL),
(2)
forcing VBOOST to rise thus reverse biasing D2. The voltage at VBOOST is then
VBOOST = 2VIN – (RDSON x IL) – VFD2 + VFD1
(3)
which is approximately
2VIN - 0.4V
(4)
for many applications. Thus the gate-drive voltage of the NMOS switch is approximately
VIN - 0.2V
(5)
An alternate method for charging CBOOST is to connect D2 to the output as shown in Figure 21. The output
voltage should be between 2.5V and 5.5V, so that proper gate voltage will be applied to the internal switch. In
this circuit, CBOOST provides a gate drive voltage that is slightly less than VOUT.
In applications where both VIN and VOUT are greater than 5.5V, or less than 3V, CBOOST cannot be charged
directly from these voltages. If VIN and VOUT are greater than 5.5V, CBOOST can be charged from VIN or VOUT
minus a zener voltage by placing a zener diode D3 in series with D2, as shown in Figure 22. When using a
series zener diode from the input, ensure that the regulation of the input supply doesn’t create a voltage that falls
outside the recommended VBOOST voltage.
(VINMAX – VD3) < 5.5V
(VINMIN – VD3) > 1.6V
D2
D3
VIN
VIN
BOOST
VBOOST
CBOOST
CIN
L
GND
VOUT
SW
D1
COUT
Figure 22. Zener Reduces Boost Voltage from VIN
An alternative method is to place the zener diode D3 in a shunt configuration as shown in Figure 23. A small
350mW to 500mW 5.1V zener in a SOT or SOD package can be used for this purpose. A small ceramic
capacitor such as a 6.3V, 0.1µF capacitor (C4) should be placed in parallel with the zener diode. When the
internal NMOS switch turns on, a pulse of current is drawn to charge the internal NMOS gate capacitance. The
0.1 µF parallel shunt capacitor ensures that the VBOOST voltage is maintained during this time.
Resistor R3 should be chosen to provide enough RMS current to the zener diode (D3) and to the BOOST pin. A
recommended choice for the zener current (IZENER) is 1 mA. The current IBOOST into the BOOST pin supplies the
gate current of the NMOS control switch and varies typically according to the following formula for the X version:
IBOOST = 0.49 x (D + 0.54) x (VZENER – VD2) mA
(6)
IBOOST can be calculated for the Y version using the following:
IBOOST = 0.20 x (D + 0.54) x (VZENER - VD2) µA
(7)
where D is the duty cycle, VZENER and VD2 are in volts, and IBOOST is in milliamps. VZENER is the voltage applied to
the anode of the boost diode (D2), and VD2 is the average forward voltage across D2. Note that this formula for
IBOOST gives typical current. For the worst case IBOOST, increase the current by 40%. In that case, the worst case
boost current will be
IBOOST-MAX = 1.4 x IBOOST
(8)
R3 will then be given by
R3 = (VIN - VZENER) / (1.4 x IBOOST + IZENER)
(9)
For example, using the X-version let VIN = 10V, VZENER = 5V, VD2 = 0.7V, IZENER = 1mA, and duty cycle D = 50%.
Then
Submit Documentation Feedback
Copyright © 2013, Texas Instruments Incorporated
Product Folder Links: LMR12007
9
LMR12007
SNVS982 – SEPTEMBER 2013
www.ti.com
IBOOST = 0.49 x (0.5 + 0.54) x (5 - 0.7) mA = 2.19mA
R3 = (10V - 5V) / (1.4 x 2.19mA + 1mA) = 1.23kΩ
(10)
(11)
VZ
C4
D2
D3
R3
VIN
BOOST
VIN
VBOOST
CBOOST
CIN
L
VOUT
SW
GND
D1
COUT
Figure 23. Boost Voltage Supplied from the Shunt Zener on VIN
ENABLE PIN / SHUTDOWN MODE
The LMR12007 has a shutdown mode that is controlled by the enable pin (EN). When a logic low voltage is
applied to EN, the part is in shutdown mode and its quiescent current drops to typically 30nA. Switch leakage
adds another 40nA from the input supply. The voltage at this pin should never exceed VIN + 0.3V.
SOFT-START
This function forces VOUT to increase at a controlled rate during start up. During soft-start, the error amplifier’s
reference voltage ramps from 0V to its nominal value of 1.25V in approximately 200µs. This forces the regulator
output to ramp up in a more linear and controlled fashion, which helps reduce inrush current.
OUTPUT OVERVOLTAGE PROTECTION
The overvoltage comparator compares the FB pin voltage to a voltage that is 10% higher than the internal
reference Vref. Once the FB pin voltage goes 10% above the internal reference, the internal NMOS control
switch is turned off, which allows the output voltage to decrease toward regulation.
UNDERVOLTAGE LOCKOUT
Undervoltage lockout (UVLO) prevents the LMR12007 from operating until the input voltage exceeds 2.74V(typ).
The UVLO threshold has approximately 440mV of hysteresis, so the part will operate until VIN drops below
2.3V(typ). Hysteresis prevents the part from turning off during power up if VIN is non-monotonic.
CURRENT LIMIT
The LMR12007 uses cycle-by-cycle current limiting to protect the output switch. During each switching cycle, a
current limit comparator detects if the output switch current exceeds 1.5A (typ), and turns off the switch until the
next switching cycle begins.
THERMAL SHUTDOWN
Thermal shutdown limits total power dissipation by turning off the output switch when the IC junction temperature
exceeds 165°C. After thermal shutdown occurs, the output switch doesn’t turn on until the junction temperature
drops to approximately 150°C.
10
Submit Documentation Feedback
Copyright © 2013, Texas Instruments Incorporated
Product Folder Links: LMR12007
LMR12007
www.ti.com
SNVS982 – SEPTEMBER 2013
Design Guide
INDUCTOR SELECTION
The Duty Cycle (D) can be approximated quickly using the ratio of output voltage (VO) to input voltage (VIN):
VO
D=
VIN
(12)
The catch diode (D1) forward voltage drop and the voltage drop across the internal NMOS must be included to
calculate a more accurate duty cycle. Calculate D by using the following formula:
VO + VD
D=
VIN + VD - VSW
(13)
VSW can be approximated by:
VSW = IO x RDS(ON)
(14)
The diode forward drop (VD) can range from 0.3V to 0.7V depending on the quality of the diode. The lower VD is,
the higher the operating efficiency of the converter.
The inductor value determines the output ripple current. Lower inductor values decrease the size of the inductor,
but increase the output ripple current. An increase in the inductor value will decrease the output ripple current.
The ratio of ripple current (ΔiL) to output current (IO) is optimized when it is set between 0.3 and 0.4 at 750mA.
The ratio r is defined as:
r=
'iL
lO
(15)
One must also ensure that the minimum current limit (1.0A) is not exceeded, so the peak current in the inductor
must be calculated. The peak current (ILPK) in the inductor is calculated by:
ILPK = IO + ΔIL/2
(16)
If r = 0.7 at an output of 750mA, the peak current in the inductor will be 1.0125A. The minimum ensured current
limit over all operating conditions is 1.0A. One can either reduce r to 0.6 resulting in a 975mA peak current, or
make the engineering judgement that 12.5mA over will be safe enough with a 1.5A typical current limit and 6
sigma limits. When the designed maximum output current is reduced, the ratio r can be increased. At a current of
0.1A, r can be made as high as 0.9. The ripple ratio can be increased at lighter loads because the net ripple is
actually quite low, and if r remains constant the inductor value can be made quite large. An equation empirically
developed for the maximum ripple ratio at any current below 2A is:
r = 0.387 x IOUT-0.3667
(17)
Note that this is just a guideline.
The LMR12007 operates at frequencies allowing the use of ceramic output capacitors without compromising
transient response. Ceramic capacitors allow higher inductor ripple without significantly increasing output ripple.
See the OUTPUT CAPACITOR section for more details on calculating output voltage ripple.
Now that the ripple current or ripple ratio is determined, the inductance is calculated by:
L=
VO + VD
IO x r x fS
x (1-D)
(18)
where fs is the switching frequency and IO is the output current. When selecting an inductor, make sure that it is
capable of supporting the peak output current without saturating. Inductor saturation will result in a sudden
reduction in inductance and prevent the regulator from operating correctly. Because of the speed of the internal
current limit, the peak current of the inductor need only be specified for the required maximum output current. For
example, if the designed maximum output current is 0.5A and the peak current is 0.7A, then the inductor should
Submit Documentation Feedback
Copyright © 2013, Texas Instruments Incorporated
Product Folder Links: LMR12007
11
LMR12007
SNVS982 – SEPTEMBER 2013
www.ti.com
be specified with a saturation current limit of >0.7A. There is no need to specify the saturation or peak current of
the inductor at the 1.5A typical switch current limit. The difference in inductor size is a factor of 5. Because of the
operating frequency of the LMR12007, ferrite based inductors are preferred to minimize core losses. This
presents little restriction since the variety of ferrite based inductors is huge. Lastly, inductors with lower series
resistance (DCR) will provide better operating efficiency. For recommended inductors see Example Circuits.
INPUT CAPACITOR
An input capacitor is necessary to ensure that VIN does not drop excessively during switching transients. The
primary specifications of the input capacitor are capacitance, voltage, RMS current rating, and ESL (Equivalent
Series Inductance). The recommended input capacitance is 10µF, although 4.7µF works well for input voltages
below 6V. The input voltage rating is specifically stated by the capacitor manufacturer. Make sure to check any
recommended deratings and also verify if there is any significant change in capacitance at the operating input
voltage and the operating temperature. The input capacitor maximum RMS input current rating (IRMS-IN) must be
greater than:
IRMS-IN = IO x
r2
D x 1-D +
12
(19)
It can be shown from the above equation that maximum RMS capacitor current occurs when D = 0.5. Always
calculate the RMS at the point where the duty cycle, D, is closest to 0.5. The ESL of an input capacitor is usually
determined by the effective cross sectional area of the current path. A large leaded capacitor will have high ESL
and a 0805 ceramic chip capacitor will have very low ESL. At the operating frequencies of the LMR12007,
certain capacitors may have an ESL so large that the resulting impedance (2πfL) will be higher than that required
to provide stable operation. As a result, surface mount capacitors are strongly recommended. Sanyo POSCAP,
Tantalum or Niobium, Panasonic SP or Cornell Dubilier ESR, and multilayer ceramic capacitors (MLCC) are all
good choices for both input and output capacitors and have very low ESL. For MLCCs it is recommended to use
X7R or X5R dielectrics. Consult capacitor manufacturer datasheet to see how rated capacitance varies over
operating conditions.
OUTPUT CAPACITOR
The output capacitor is selected based upon the desired output ripple and transient response. The initial current
of a load transient is provided mainly by the output capacitor. The output ripple of the converter is:
'VO = 'iL x (RESR +
1
)
8 x fS x CO
(20)
When using MLCCs, the ESR is typically so low that the capacitive ripple may dominate. When this occurs, the
output ripple will be approximately sinusoidal and 90° phase shifted from the switching action. Given the
availability and quality of MLCCs and the expected output voltage of designs using the LMR12007, there is really
no need to review any other capacitor technologies. Another benefit of ceramic capacitors is their ability to
bypass high frequency noise. A certain amount of switching edge noise will couple through parasitic
capacitances in the inductor to the output. A ceramic capacitor will bypass this noise while a tantalum will not.
Since the output capacitor is one of the two external components that control the stability of the regulator control
loop, most applications will require a minimum at 10 µF of output capacitance. Capacitance can be increased
significantly with little detriment to the regulator stability. Like the input capacitor, recommended multilayer
ceramic capacitors are X7R or X5R. Again, verify actual capacitance at the desired operating voltage and
temperature.
Check the RMS current rating of the capacitor. The RMS current rating of the capacitor chosen must also meet
the following condition:
IRMS-OUT = IO x
12
r
12
(21)
Submit Documentation Feedback
Copyright © 2013, Texas Instruments Incorporated
Product Folder Links: LMR12007
LMR12007
www.ti.com
SNVS982 – SEPTEMBER 2013
CATCH DIODE
The catch diode (D1) conducts during the switch off-time. A Schottky diode is recommended for its fast switching
times and low forward voltage drop. The catch diode should be chosen so that its current rating is greater than:
ID1 = IO x (1-D)
(22)
The reverse breakdown rating of the diode must be at least the maximum input voltage plus appropriate margin.
To improve efficiency choose a Schottky diode with a low forward voltage drop.
BOOST DIODE
A standard diode such as the 1N4148 type is recommended. For VBOOST circuits derived from voltages less than
3.3V, a small-signal Schottky diode is recommended for greater efficiency. A good choice is the BAT54 small
signal diode.
BOOST CAPACITOR
A ceramic 0.01µF capacitor with a voltage rating of at least 6.3V is sufficient. The X7R and X5R MLCCs provide
the best performance.
OUTPUT VOLTAGE
The output voltage is set using the following equation where R2 is connected between the FB pin and GND, and
R1 is connected between VO and the FB pin. A good value for R2 is 10kΩ.
R1 =
VO
VREF
- 1 x R2
(23)
PCB Layout Considerations
When planning layout there are a few things to consider when trying to achieve a clean, regulated output. The
most important consideration when completing the layout is the close coupling of the GND connections of the CIN
capacitor and the catch diode D1. These ground ends should be close to one another and be connected to the
GND plane with at least two through-holes. Place these components as close to the IC as possible. Next in
importance is the location of the GND connection of the COUT capacitor, which should be near the GND
connections of CIN and D1.
There should be a continuous ground plane on the bottom layer of a two-layer board except under the switching
node island.
The FB pin is a high impedance node and care should be taken to make the FB trace short to avoid noise pickup
and inaccurate regulation. The feedback resistors should be placed as close as possible to the IC, with the GND
of R2 placed as close as possible to the GND of the IC. The VOUT trace to R1 should be routed away from the
inductor and any other traces that are switching.
High AC currents flow through the VIN, SW and VOUT traces, so they should be as short and wide as possible.
However, making the traces wide increases radiated noise, so the designer must make this trade-off. Radiated
noise can be decreased by choosing a shielded inductor.
The remaining components should also be placed as close as possible to the IC. Please see Application Note
AN-1229 SNVA054 for further considerations and the LMR12007 demo board as an example of a four-layer
layout.
Submit Documentation Feedback
Copyright © 2013, Texas Instruments Incorporated
Product Folder Links: LMR12007
13
LMR12007
SNVS982 – SEPTEMBER 2013
www.ti.com
LMR12007X Circuit Examples
D2
VIN
BOOST
VIN
C3
C1
L1
R3
VOUT
SW
ON
D1
EN
C2
R1
OFF
FB
GND
R2
Figure 24. LMR12007X (1.6MHz)
VBOOST Derived from VIN
5V to 1.5V/750mA
Table 1. Bill of Materials for Figure 24
Part ID
Part Value
Part Number
Manufacturer
U1
750mA Buck Regulator
LMR12007X
Texas Instruments
C1, Input Cap
10µF, 6.3V, X5R
C3216X5ROJ106M
TDK
C2, Output Cap
10µF, 6.3V, X5R
C3216X5ROJ106M
TDK
C3, Boost Cap
0.01uF, 16V, X7R
C1005X7R1C103K
TDK
D1, Catch Diode
0.3VF Schottky 1A, 10VR
MBRM110L
ON Semi
D2, Boost Diode
1VF @ 50mA Diode
1N4148W
Diodes, Inc.
L1
4.7µH, 1.7A,
VLCF4020T- 4R7N1R2
TDK
R1
2kΩ, 1%
CRCW06032001F
Vishay
R2
10kΩ, 1%
CRCW06031002F
Vishay
R3
100kΩ, 1%
CRCW06031003F
Vishay
14
Submit Documentation Feedback
Copyright © 2013, Texas Instruments Incorporated
Product Folder Links: LMR12007
LMR12007
www.ti.com
SNVS982 – SEPTEMBER 2013
D2
VIN
BOOST
VIN
C3
C1
R3
L1
SW
ON
VOUT
D1
C2
EN
OFF
R1
FB
GND
R2
Figure 25. LMR12007X (1.6MHz)
VBOOST Derived from VOUT
12V to 3.3V/750mA
Table 2. Bill of Materials for Figure 25
Part ID
Part Value
Part Number
Manufacturer
U1
750mA Buck Regulator
LMR12007X
Texas Instruments
C1, Input Cap
10µF, 25V, X7R
C3225X7R1E106M
TDK
C2, Output Cap
22µF, 6.3V, X5R
C3216X5ROJ226M
TDK
C3, Boost Cap
0.01µF, 16V, X7R
C1005X7R1C103K
TDK
D1, Catch Diode
0.34VF Schottky 1A, 30VR
SS1P3L
Vishay
D2, Boost Diode
30V, 200 mA Schottky
BAT54
Diodes Inc.
L1
4.7µH, 1.7A,
VLCF4020T- 4R7N1R2
TDK
R1
16.5kΩ, 1%
CRCW06031652F
Vishay
R2
10.0 kΩ, 1%
CRCW06031002F
Vishay
R3
100kΩ, 1%
CRCW06031003F
Vishay
Submit Documentation Feedback
Copyright © 2013, Texas Instruments Incorporated
Product Folder Links: LMR12007
15
LMR12007
SNVS982 – SEPTEMBER 2013
www.ti.com
C4
D3
R4
D2
BOOST
VIN
VIN
C3
C1
R3
L1
VOUT
SW
ON
D1
C2
EN
OFF
R1
FB
GND
R2
Figure 26. LMR12007X (1.6MHz)
VBOOST Derived from VSHUNT
18V to 1.5V/750mA
Table 3. Bill of Materials for Figure 26
Part ID
Part Value
Part Number
Manufacturer
U1
750mA Buck Regulator
LMR12007X
Texas Instruments
C1, Input Cap
10µF, 25V, X7R
C3225X7R1E106M
TDK
C2, Output Cap
22µF, 6.3V, X5R
C3216X5ROJ226M
TDK
C3, Boost Cap
0.01µF, 16V, X7R
C1005X7R1C103K
TDK
C4, Shunt Cap
0.1µF, 6.3V, X5R
C1005X5R0J104K
TDK
D1, Catch Diode
0.4VF Schottky 1A, 30VR
SS1P3L
Vishay
D2, Boost Diode
1VF @ 50mA Diode
1N4148W
Diodes, Inc.
D3, Zener Diode
5.1V 250Mw SOT
BZX84C5V1
Vishay
L1
6.8µH, 1.6A,
SLF7032T-6R8M1R6
TDK
R1
2kΩ, 1%
CRCW06032001F
Vishay
R2
10kΩ, 1%
CRCW06031002F
Vishay
R3
100kΩ, 1%
CRCW06031003F
Vishay
R4
4.12kΩ, 1%
CRCW06034121F
Vishay
16
Submit Documentation Feedback
Copyright © 2013, Texas Instruments Incorporated
Product Folder Links: LMR12007
LMR12007
www.ti.com
SNVS982 – SEPTEMBER 2013
D3
D2
BOOST
VIN
VIN
C1
C3
L1
R3
VOUT
SW
ON
D1
EN
C2
R1
OFF
FB
GND
R2
Figure 27. LMR12007X (1.6MHz)
VBOOST Derived from Series Zener Diode (VIN)
15V to 1.5V/750mA
Table 4. Bill of Materials for Figure 27
Part ID
Part Value
Part Number
Manufacturer
U1
750mA Buck Regulator
LMR12007X
Texas Instruments
C1, Input Cap
10µF, 25V, X7R
C3225X7R1E106M
TDK
C2, Output Cap
22µF, 6.3V, X5R
C3216X5ROJ226M
TDK
C3, Boost Cap
0.01µF, 16V, X7R
C1005X7R1C103K
TDK
D1, Catch Diode
0.4VF Schottky 1A, 30VR
SS1P3L
Vishay
D2, Boost Diode
1VF @ 50mA Diode
1N4148W
Diodes, Inc.
D3, Zener Diode
11V 350Mw SOT
BZX84C11T
Diodes, Inc.
L1
6.8µH, 1.6A,
SLF7032T-6R8M1R6
TDK
R1
2kΩ, 1%
CRCW06032001F
Vishay
R2
10kΩ, 1%
CRCW06031002F
Vishay
R3
100kΩ, 1%
CRCW06031003F
Vishay
Submit Documentation Feedback
Copyright © 2013, Texas Instruments Incorporated
Product Folder Links: LMR12007
17
LMR12007
SNVS982 – SEPTEMBER 2013
www.ti.com
D3
D2
VIN
BOOST
VIN
C3
R3
C1
L1
VOUT
SW
ON
D1
EN
C2
R1
OFF
FB
GND
R2
Figure 28. LMR12007X (1.6MHz)
VBOOST Derived from Series Zener Diode (VOUT)
15V to 9V/750mA
Table 5. Bill of Materials for Figure 28
Part ID
Part Value
Part Number
Manufacturer
U1
750mA Buck Regulator
LMR12007X
Texas Instruments
C1, Input Cap
10µF, 25V, X7R
C3225X7R1E106M
TDK
C2, Output Cap
22µF, 16V, X5R
C3216X5R1C226M
TDK
C3, Boost Cap
0.01µF, 16V, X7R
C1005X7R1C103K
TDK
D1, Catch Diode
0.4VF Schottky 1A, 30VR
SS1P3L
Vishay
D2, Boost Diode
1VF @ 50mA Diode
1N4148W
Diodes, Inc.
D3, Zener Diode
4.3V 350mw SOT
BZX84C4V3
Diodes, Inc.
L1
6.8µH, 1.6A,
SLF7032T-6R8M1R6
TDK
R1
61.9kΩ, 1%
CRCW06036192F
Vishay
R2
10kΩ, 1%
CRCW06031002F
Vishay
R3
100kΩ, 1%
CRCW06031003F
Vishay
18
Submit Documentation Feedback
Copyright © 2013, Texas Instruments Incorporated
Product Folder Links: LMR12007
LMR12007
www.ti.com
SNVS982 – SEPTEMBER 2013
LMR12007Y Circuit Examples
D2
VIN
BOOST
VIN
C3
C1
L1
R3
VOUT
SW
ON
D1
EN
C2
R1
OFF
FB
GND
R2
Figure 29. LMR12007Y (550kHz)
VBOOST Derived from VIN
5V to 1.5V/750mA
Table 6. Bill of Materials for Figure 29
Part ID
Part Value
Part Number
Manufacturer
U1
750mA Buck Regulator
LMR12007Y
Texas Instruments
C1, Input Cap
10µF, 6.3V, X5R
C3216X5ROJ106M
TDK
C2, Output Cap
22µF, 6.3V, X5R
C3216X5ROJ226M
TDK
C3, Boost Cap
0.01µF, 16V, X7R
C1005X7R1C103K
TDK
D1, Catch Diode
0.3VF Schottky 1A, 10VR
MBRM110L
ON Semi
D2, Boost Diode
1VF @ 50mA Diode
1N4148W
Diodes, Inc.
L1
10µH, 1.6A,
SLF7032T-100M1R4
TDK
R1
2kΩ, 1%
CRCW06032001F
Vishay
R2
10kΩ, 1%
CRCW06031002F
Vishay
R3
100kΩ, 1%
CRCW06031003F
Vishay
Submit Documentation Feedback
Copyright © 2013, Texas Instruments Incorporated
Product Folder Links: LMR12007
19
LMR12007
SNVS982 – SEPTEMBER 2013
www.ti.com
D2
VIN
BOOST
VIN
C3
C1
R3
L1
SW
ON
VOUT
D1
C2
EN
OFF
R1
FB
GND
R2
Figure 30. LMR12007Y (550kHz)
VBOOST Derived from VOUT
12V to 3.3V/750mA
Table 7. Bill of Materials for Figure 30
Part ID
Part Value
Part Number
Manufacturer
U1
750mA Buck Regulator
LMR12007Y
Texas Instruments
C1, Input Cap
10µF, 25V, X7R
C3225X7R1E106M
TDK
C2, Output Cap
22µF, 6.3V, X5R
C3216X5ROJ226M
TDK
C3, Boost Cap
0.01µF, 16V, X7R
C1005X7R1C103K
TDK
D1, Catch Diode
0.34VF Schottky 1A, 30VR
SS1P3L
Vishay
D2, Boost Diode
30V, 200 mA Schottky
BAT54
Diodes Inc.
L1
10µH, 1.6A,
SLF7032T-100M1R4
TDK
R1
16.5kΩ, 1%
CRCW06031652F
Vishay
R2
10.0 kΩ, 1%
CRCW06031002F
Vishay
R3
100kΩ, 1%
CRCW06031003F
Vishay
20
Submit Documentation Feedback
Copyright © 2013, Texas Instruments Incorporated
Product Folder Links: LMR12007
LMR12007
www.ti.com
SNVS982 – SEPTEMBER 2013
C4
D3
R4
D2
BOOST
VIN
VIN
C3
C1
R3
L1
VOUT
SW
ON
D1
C2
EN
OFF
R1
FB
GND
R2
Figure 31. LMR12007Y (550kHz)
VBOOST Derived from VSHUNT
18V to 1.5V/750mA
Table 8. Bill of Materials for Figure 31
Part ID
Part Value
Part Number
Manufacturer
U1
750mA Buck Regulator
LMR12007Y
Texas Instruments
C1, Input Cap
10µF, 25V, X7R
C3225X7R1E106M
TDK
C2, Output Cap
22µF, 6.3V, X5R
C3216X5ROJ226M
TDK
C3, Boost Cap
0.01µF, 16V, X7R
C1005X7R1C103K
TDK
C4, Shunt Cap
0.1µF, 6.3V, X5R
C1005X5R0J104K
TDK
D1, Catch Diode
0.4VF Schottky 1A, 30VR
SS1P3L
Vishay
D2, Boost Diode
1VF @ 50mA Diode
1N4148W
Diodes, Inc.
D3, Zener Diode
5.1V 250Mw SOT
BZX84C5V1
Vishay
L1
15µH, 1.5A
SLF7045T-150M1R5
TDK
R1
2kΩ, 1%
CRCW06032001F
Vishay
R2
10kΩ, 1%
CRCW06031002F
Vishay
R3
100kΩ, 1%
CRCW06031003F
Vishay
R4
4.12kΩ, 1%
CRCW06034121F
Vishay
Submit Documentation Feedback
Copyright © 2013, Texas Instruments Incorporated
Product Folder Links: LMR12007
21
LMR12007
SNVS982 – SEPTEMBER 2013
www.ti.com
D3
D2
BOOST
VIN
VIN
C3
C1
L1
R3
VOUT
SW
ON
D1
EN
C2
R1
OFF
FB
GND
R2
Figure 32. LMR12007Y (550kHz)
VBOOST Derived from Series Zener Diode (VIN)
15V to 1.5V/750mA
Table 9. Bill of Materials for Figure 32
Part ID
Part Value
Part Number
Manufacturer
U1
750mA Buck Regulator
LMR12007Y
Texas Instruments
C1, Input Cap
10µF, 25V, X7R
C3225X7R1E106M
TDK
C2, Output Cap
22µF, 6.3V, X5R
C3216X5ROJ226M
TDK
C3, Boost Cap
0.01µF, 16V, X7R
C1005X7R1C103K
TDK
D1, Catch Diode
0.4VF Schottky 1A, 30VR
SS1P3L
Vishay
D2, Boost Diode
1VF @ 50mA Diode
1N4148W
Diodes, Inc.
D3, Zener Diode
11V 350Mw SOT
BZX84C11T
Diodes, Inc.
L1
15µH, 1.5A,
SLF7045T-150M1R5
TDK
R1
2kΩ, 1%
CRCW06032001F
Vishay
R2
10kΩ, 1%
CRCW06031002F
Vishay
R3
100kΩ, 1%
CRCW06031003F
Vishay
22
Submit Documentation Feedback
Copyright © 2013, Texas Instruments Incorporated
Product Folder Links: LMR12007
LMR12007
www.ti.com
SNVS982 – SEPTEMBER 2013
D3
D2
VIN
BOOST
VIN
C3
C1
R3
L1
VOUT
SW
ON
D1
EN
C2
R1
OFF
FB
GND
R2
Figure 33. LMR12007Y (550kHz)
VBOOST Derived from Series Zener Diode (VOUT)
15V to 9V/750mA
Table 10. Bill of Materials for Figure 33
Part ID
Part Value
Part Number
Manufacturer
U1
750mA Buck Regulator
LMR12007Y
Texas Instruments
C1, Input Cap
10µF, 25V, X7R
C3225X7R1E106M
TDK
C2, Output Cap
22µF, 16V, X5R
C3216X5R1C226M
TDK
C3, Boost Cap
0.01µF, 16V, X7R
C1005X7R1C103K
TDK
D1, Catch Diode
0.4VF Schottky 1A, 30VR
SS1P3L
Vishay
D2, Boost Diode
1VF @ 50mA Diode
1N4148W
Diodes, Inc.
D3, Zener Diode
4.3V 350mw SOT
BZX84C4V3
Diodes, Inc.
L1
22µH, 1.4A,
SLF7045T-220M1R3-1PF
TDK
R1
61.9kΩ, 1%
CRCW06036192F
Vishay
R2
10kΩ, 1%
CRCW06031002F
Vishay
R3
100kΩ, 1%
CRCW06031003F
Vishay
Submit Documentation Feedback
Copyright © 2013, Texas Instruments Incorporated
Product Folder Links: LMR12007
23
PACKAGE OPTION ADDENDUM
www.ti.com
23-Sep-2013
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
Lead/Ball Finish
(2)
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
LMR12007XMK
ACTIVE
SOT
DDC
6
1000
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
-40 to 125
SP1B
LMR12007XMKX
ACTIVE
SOT
DDC
6
3000
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
-40 to 125
SP1B
LMR12007YMK
ACTIVE
SOT
DDC
6
1000
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
-40 to 125
SP2B
LMR12007YMKX
ACTIVE
SOT
DDC
6
3000
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
-40 to 125
SP2B
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4)
There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5)
Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
Addendum-Page 1
Samples
PACKAGE OPTION ADDENDUM
www.ti.com
23-Sep-2013
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
11-Oct-2013
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
Diameter Width (mm)
(mm) W1 (mm)
B0
(mm)
K0
(mm)
P1
(mm)
W
Pin1
(mm) Quadrant
LMR12007XMK
SOT
DDC
6
1000
178.0
8.4
3.2
3.2
1.4
4.0
8.0
Q3
LMR12007XMKX
SOT
DDC
6
3000
178.0
8.4
3.2
3.2
1.4
4.0
8.0
Q3
LMR12007YMK
SOT
DDC
6
1000
178.0
8.4
3.2
3.2
1.4
4.0
8.0
Q3
LMR12007YMKX
SOT
DDC
6
3000
178.0
8.4
3.2
3.2
1.4
4.0
8.0
Q3
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
11-Oct-2013
*All dimensions are nominal
Device
Package Type
Package Drawing
Pins
SPQ
Length (mm)
Width (mm)
Height (mm)
LMR12007XMK
SOT
DDC
6
1000
210.0
185.0
35.0
LMR12007XMKX
SOT
DDC
6
3000
210.0
185.0
35.0
LMR12007YMK
SOT
DDC
6
1000
210.0
185.0
35.0
LMR12007YMKX
SOT
DDC
6
3000
210.0
185.0
35.0
Pack Materials-Page 2
IMPORTANT NOTICE
Texas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, enhancements, improvements and other
changes to its semiconductor products and services per JESD46, latest issue, and to discontinue any product or service per JESD48, latest
issue. Buyers should obtain the latest relevant information before placing orders and should verify that such information is current and
complete. All semiconductor products (also referred to herein as “components”) are sold subject to TI’s terms and conditions of sale
supplied at the time of order acknowledgment.
TI warrants performance of its components to the specifications applicable at the time of sale, in accordance with the warranty in TI’s terms
and conditions of sale of semiconductor products. Testing and other quality control techniques are used to the extent TI deems necessary
to support this warranty. Except where mandated by applicable law, testing of all parameters of each component is not necessarily
performed.
TI assumes no liability for applications assistance or the design of Buyers’ products. Buyers are responsible for their products and
applications using TI components. To minimize the risks associated with Buyers’ products and applications, Buyers should provide
adequate design and operating safeguards.
TI does not warrant or represent that any license, either express or implied, is granted under any patent right, copyright, mask work right, or
other intellectual property right relating to any combination, machine, or process in which TI components or services are used. Information
published by TI regarding third-party products or services does not constitute a license to use such products or services or a warranty or
endorsement thereof. Use of such information may require a license from a third party under the patents or other intellectual property of the
third party, or a license from TI under the patents or other intellectual property of TI.
Reproduction of significant portions of TI information in TI data books or data sheets is permissible only if reproduction is without alteration
and is accompanied by all associated warranties, conditions, limitations, and notices. TI is not responsible or liable for such altered
documentation. Information of third parties may be subject to additional restrictions.
Resale of TI components or services with statements different from or beyond the parameters stated by TI for that component or service
voids all express and any implied warranties for the associated TI component or service and is an unfair and deceptive business practice.
TI is not responsible or liable for any such statements.
Buyer acknowledges and agrees that it is solely responsible for compliance with all legal, regulatory and safety-related requirements
concerning its products, and any use of TI components in its applications, notwithstanding any applications-related information or support
that may be provided by TI. Buyer represents and agrees that it has all the necessary expertise to create and implement safeguards which
anticipate dangerous consequences of failures, monitor failures and their consequences, lessen the likelihood of failures that might cause
harm and take appropriate remedial actions. Buyer will fully indemnify TI and its representatives against any damages arising out of the use
of any TI components in safety-critical applications.
In some cases, TI components may be promoted specifically to facilitate safety-related applications. With such components, TI’s goal is to
help enable customers to design and create their own end-product solutions that meet applicable functional safety standards and
requirements. Nonetheless, such components are subject to these terms.
No TI components are authorized for use in FDA Class III (or similar life-critical medical equipment) unless authorized officers of the parties
have executed a special agreement specifically governing such use.
Only those TI components which TI has specifically designated as military grade or “enhanced plastic” are designed and intended for use in
military/aerospace applications or environments. Buyer acknowledges and agrees that any military or aerospace use of TI components
which have not been so designated is solely at the Buyer's risk, and that Buyer is solely responsible for compliance with all legal and
regulatory requirements in connection with such use.
TI has specifically designated certain components as meeting ISO/TS16949 requirements, mainly for automotive use. In any case of use of
non-designated products, TI will not be responsible for any failure to meet ISO/TS16949.
Products
Applications
Audio
www.ti.com/audio
Automotive and Transportation
www.ti.com/automotive
Amplifiers
amplifier.ti.com
Communications and Telecom
www.ti.com/communications
Data Converters
dataconverter.ti.com
Computers and Peripherals
www.ti.com/computers
DLP® Products
www.dlp.com
Consumer Electronics
www.ti.com/consumer-apps
DSP
dsp.ti.com
Energy and Lighting
www.ti.com/energy
Clocks and Timers
www.ti.com/clocks
Industrial
www.ti.com/industrial
Interface
interface.ti.com
Medical
www.ti.com/medical
Logic
logic.ti.com
Security
www.ti.com/security
Power Mgmt
power.ti.com
Space, Avionics and Defense
www.ti.com/space-avionics-defense
Microcontrollers
microcontroller.ti.com
Video and Imaging
www.ti.com/video
RFID
www.ti-rfid.com
OMAP Applications Processors
www.ti.com/omap
TI E2E Community
e2e.ti.com
Wireless Connectivity
www.ti.com/wirelessconnectivity
Mailing Address: Texas Instruments, Post Office Box 655303, Dallas, Texas 75265
Copyright © 2013, Texas Instruments Incorporated