Octal Ultrasound Analog Front End AD9674 Data Sheet FEATURES GENERAL DESCRIPTION 8 channels of LNA, VGA, AAF, ADC, and digital RF decimator Low power: 150 mW per channel, TGC mode, 40 MSPS; 62.5 mW per channel, CW mode; <30 mW in power-down Time gain compensation (TGC) channel input referred noise: 0.82 nV/√Hz, maximum gain Flexible power-down modes Fast recovery from low power standby mode: <2 μs Low noise preamplifier (LNA) Input referred noise voltage: 0.78 nV/√Hz, gain = 21.6 dB Programmable gain: 15.6 dB/17.9 dB/21.6 dB 0.1 dB compression: 1.00 V p-p/ 0.75 V p-p/0.45 V p-p Flexible active input impedance matching Variable gain amplifier (VGA) Attenuator range: 45 dB, linear in dB gain control Postamplifier gain (PGA): 21 dB/24 dB/27 dB/30 dB Antialiasing filter (AAF) Programmable second-order low-pass filter (LPF) from 8 MHz to 18 MHz or 13.5 MHz to 30 MHz and high-pass filter (HPF) Analog-to-digital converter (ADC) Signal-to-noise ratio (SNR): 75 dB, 14 bits up to 125 MSPS Configurable serial low voltage differential signaling (LVDS) Continuous wave (CW) Doppler mode harmonic rejection I/Q demodulator Individual programmable phase rotation Dynamic range per channel: >160 dBFS/√Hz Close in SNR: 156 dBc/√Hz, 1 kHz offset, −3 dBFS input Radio frequency (RF) digital HPF and decimation by 2 10 mm × 10 mm, 144-ball CSP_BGA The AD9674 is designed for low cost, low power, small size, and ease of use for medical ultrasound. It contains eight channels of a VGA with an LNA, a CW harmonic rejection I/Q demodulator with programmable phase rotation, an AAF, an ADC, a digital HPF, and RF decimation by 2. Each channel features a maximum gain of up to 52 dB, a fully differential signal path, and an active input preamplifier termination. The channel is optimized for high dynamic performance and low power in applications where a small package size is critical. The LNA has a single-ended to differential gain that is selectable through the serial port interface (SPI). Assuming a 15 MHz noise bandwidth (NBW) and a 21.6 dB LNA gain, the LNA input SNR is 94 dB. In CW Doppler mode, each LNA output drives an I/Q demodulator that has independently programmable phase rotation with 16 phase settings. Power-down of individual channels is supported to increase battery life for portable applications. Standby mode allows quick power-up for power cycling. In CW Doppler operation, the VGA, AAF, and ADC are powered down. The ADC contains several features designed to maximize flexibility and minimize system cost, such as a programmable clock, data alignment, and programmable digital test pattern generation. The digital test patterns include built in fixed patterns, built in pseudorandom patterns, and custom user defined test patterns entered via the SPI. APPLICATIONS Medical imaging/ultrasound Nondestructive Testing (NDT) Rev. A Document Feedback Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 ©2016 Analog Devices, Inc. All rights reserved. Technical Support www.analog.com AD9674* PRODUCT PAGE QUICK LINKS Last Content Update: 02/23/2017 COMPARABLE PARTS DISCUSSIONS View a parametric search of comparable parts. View all AD9674 EngineerZone Discussions. DOCUMENTATION SAMPLE AND BUY Data Sheet Visit the product page to see pricing options. • AD9674: Octal Ultrasound AFE With RF Decimator Data Sheet TECHNICAL SUPPORT REFERENCE MATERIALS Submit a technical question or find your regional support number. Press • Low Cost, Octal Ultrasound Receiver with On-Chip RF Decimator and JESD204B Serial Interface DOCUMENT FEEDBACK Submit feedback for this data sheet. DESIGN RESOURCES • AD9674 Material Declaration • PCN-PDN Information • Quality And Reliability • Symbols and Footprints This page is dynamically generated by Analog Devices, Inc., and inserted into this data sheet. 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AD9674 Data Sheet TABLE OF CONTENTS Features .............................................................................................. 1 Analog Test Signal Generation ................................................. 31 Applications ....................................................................................... 1 CW Doppler Operation ............................................................. 32 General Description ......................................................................... 1 Digital RF Decimator ..................................................................... 33 Revision History ............................................................................... 2 Vector Profile .............................................................................. 33 Functional Block Diagram .............................................................. 3 RF Decimator .............................................................................. 34 Specifications..................................................................................... 4 Digital Test Waveforms.............................................................. 34 AC Specifications.......................................................................... 4 Digital block Power Saving scheme ......................................... 35 Digital Specifications ................................................................... 7 Serial Port Interface (SPI)................................................................ 36 Switching Specifications .............................................................. 8 Hardware Interface ..................................................................... 36 ADC Timing Diagram ................................................................. 9 Memory Map .................................................................................. 38 CW Doppler Timing Diagram ................................................... 9 Reading the Memory Map Table .............................................. 38 Absolute Maximum Ratings .......................................................... 11 Reserved Locations .................................................................... 38 Thermal Impedance ................................................................... 11 Default Values ............................................................................. 38 ESD Caution ................................................................................ 11 Logic Levels ................................................................................. 38 Pin Configuration and Function Descriptions ........................... 12 Recommended Start-Up Sequence .......................................... 38 Typical Performance Characteristics ........................................... 15 Memory Map Register Descriptions ........................................ 46 TGC Mode ................................................................................... 15 Outline Dimensions ....................................................................... 47 CW Doppler Mode..................................................................... 19 Ordering Guide .......................................................................... 47 Theory of Operation ...................................................................... 20 TGC Operation ........................................................................... 20 REVISION HISTORY 1/16—Revision A: Initial Version Rev. A | Page 2 of 47 Data Sheet AD9674 FUNCTIONAL BLOCK DIAGRAM AVDD1 AVDD2 LO-A TO LO-H PDWN STBY DVDD DRVDD CWQ+ CWQ– CWI+ CWI– CWD I/Q DEMODULATOR LOSW-A TO LOSW-H LI-A TO LI-H LNA LG-A TO LG-H VGA 14-BIT ADC AAF FILTER/ DECIMATOR SERIALIZER LVDS DOUTA+ TO DOUTH+ DOUTA– TO DOUTH– AD9674 8 CHANNELS Figure 1. Rev. A | Page 3 of 47 FCO+ FCO– DCO+ DCO– 11293-001 CLK– DATA RATE MULTIPLIER CLK+ SDIO CSB SCLK GPO0 TO GPO3 SERIAL PORT INTERFACE ADDR0 TO ADDR4 TX_TRIG– NCO TX_TRIG+ VREF RBIAS REFERENCE GAIN+ GAIN– MLO– MLO+ RESET– RESET+ LO GENERATION AD9674 Data Sheet SPECIFICATIONS AC SPECIFICATIONS AVDD1 = 1.8 V, AVDD2 = 3.0 V, DVDD = 1.4 V, DRVDD = 1.8 V, 1.0 V internal ADC reference, full temperature range (0°C to 85°C), fIN = 5 MHz, local oscillator (LO) band mode, RS = 50 Ω, RFB = ∞ (unterminated), LNA gain = 21.6 dB, LNA bias = midhigh, programmable gain amplifier (PGA) gain = 27 dB, analog gain control, VGAIN = (GAIN+) − (GAIN−) = 1.6 V, AAF LPF cutoff = fSAMPLE/3 in Mode I 1/Mode II,1 AAF LPF cutoff = fSAMPLE/4.5 in Mode III1/Mode IV,1 HPF cutoff = LPF cutoff/12.00, Mode I1 = fSAMPLE = 40 MSPS, Mode II1 = fSAMPLE = 65 MSPS, Mode III1 = fSAMPLE = 80 MSPS, Mode IV1 = fSAMPLE = 125 MSPS, RF decimator bypassed, digital filter bypassed, and low power LVDS mode, unless otherwise noted. All gain setting options are listed, which can be configured via SPI registers, and all power supply currents and power dissipations are listed for the four mode settings (Mode I, Mode II, Mode III, and Mode IV).1 Table 1. Parameter 2 LNA CHARACTERISTICS Gain 0.1 dB Input Compression Point 1 dB Input Compression Point Input Common Mode (LI-x, LG-x) Output Common Mode (LO-x) Output Common Mode (LOSW-x) Input Resistance (LI-x) Input Capacitance (LI-x) Input Referred Noise Voltage Input SNR Input Referred Noise Current FULL CHANNEL (TGC) CHARACTERISTICS AAF Low-Pass Cutoff In Range AAF Bandwidth Tolerance Group Delay Variation Input Referred Noise Voltage Noise Figure Active Termination Matched Unterminated Correlated Noise Ratio Test Conditions/Comments Min Single-ended input to differential output Single-ended input to single-ended output LNA gain = 15.6 dB LNA gain = 17.9 dB LNA gain = 21.6 dB LNA gain = 15.6 dB LNA gain = 17.9 dB LNA gain = 21.6 dB Switch off Switch on Switch off Switch on RFB = 300 Ω RFB = 1350 Ω RFB = ∞ (unterminated) RS = 0 Ω LNA gain = 15.6 dB LNA gain = 17.9 dB LNA gain = 21.6 dB Noise bandwidth = 15 MHz, LNA gain = 21.6 dB −3 dB, programmable, low band mode −3 dB, programmable, high band mode f = 1 MHz to 18 MHz, VGAIN = −1.6 V to +1.6 V LNA gain = 15.6 dB LNA gain = 17.9 dB LNA gain = 21.6 dB RS = 50 Ω LNA gain = 15.6 dB, RFB = 150 Ω LNA gain = 17.9 dB, RFB = 200 Ω LNA gain = 21.6 dB, RFB = 300 Ω LNA gain = 15.6 dB LNA gain = 17.9 dB LNA gain = 21.6 dB No signal, correlated/uncorrelated Rev. A | Page 4 of 47 Typ Max Unit 15.6/17.9/21.6 3 9.6/11.9/15.63 dB dB 1.00 0.75 0.45 1.20 0.90 0.60 2.2 High-Z 1.5 High-Z 1.5 50 200 6 20 V p-p V p-p V p-p V p-p V p-p V p-p V Ω V Ω V Ω Ω kΩ pF 0.83 0.82 0.78 94 nV/√Hz nV/√Hz nV/√Hz dB 2.6 pA/√Hz 8 13.5 18 30 ±10 ±350 0.96 0.90 0.82 MHz MHz % ps nV/√Hz nV/√Hz nV/√Hz 5.6 4.8 3.8 3.2 2.9 2.6 −30 dB dB dB dB dB dB dB Data Sheet Parameter 2 Output Offset SNR Close-In SNR Second Harmonic Third Harmonic Two-Tone Intermodulation Distortion (IMD3) Channel to Channel Crosstalk GAIN ACCURACY Gain Law Conformance Error Linear Gain Error Channel to Channel Matching PGA Gain GAIN CONTROL INTERFACE Control Range Control Common Mode Input Impedance Gain Range Scale Factor Response Time CW DOPPLER MODE LO Frequency Phase Resolution Output DC Bias (Single-Ended) Output AC Current Range Transconductance (Differential) Input Referred Noise Voltage Noise Figure Dynamic Range AD9674 Test Conditions/Comments fIN = 5 MHz at −12 dBFS, VGAIN = −1.6 V fIN = 5 MHz at −1 dBFS, VGAIN = 1.6 V fIN = 3.5 MHz at −1 dBFS, VGAIN = 0 V, 1 kHz offset fIN = 5 MHz at −12 dBFS, VGAIN = −1.6 V fIN = 5 MHz at −1 dBFS, VGAIN = 1.6 V fIN = 5 MHz at −12 dBFS, VGAIN = −1.6 V fIN = 5 MHz at −1 dBFS, VGAIN = 1.6 V fRF1 = 5.015 MHz, fRF2 = 5.020 MHz, ARF1 = −1 dBFS, ARF2 = −21 dBFS, VGAIN = 1.6 V, IMD3 relative to ARF2 fIN = 5 MHz at −1 dBFS Overrange condition 4 TA = 25°C −1.6 < VGAIN < −1.28 V −1.28 V < VGAIN < +1.28 V 1.28 V < VGAIN < 1.6 V VGAIN = 0 V, normalized for ideal AAF loss −1.28 V < VGAIN < +1.28 V, 1 σ Differential GAIN+, GAIN− GAIN+, GAIN− Min −100 Max +100 −70 −62 −61 −55 −54 dBc dBc dBc dBc dBc −60 −55 dB dB 0.4 +1.3 −0.5 −1.3 +1.3 0.1 21/24/27/303 Analog Digital step size Analog 45 dB change Rev. A | Page 5 of 47 69 59 −130 Unit LSB dBFS dBFS dBc/√Hz −1.3 −1.6 0.7 fLO = fMLO/M Per channel, 4LO 5 mode Per channel, 8LO5 mode, 16LO5 mode CWI+, CWI−, CWQ+, CWQ− Per CWI+, CWI−, CWQ+, and CWQ−, each channel is enabled (2 × fLO and baseband signal) Demodulated IOUT/VIN, per CWI+, CWI−, CWQ+, and CWQ− LNA gain = 15.6 dB LNA gain = 17.9 dB LNA gain = 21.6 dB RS = 0 Ω, RFB = ∞ LNA gain = 15.6 dB LNA gain = 17.9 dB LNA gain = 21.6 dB RS = 50 Ω, RFB = ∞ LNA gain = 15.6 dB LNA gain = 17.9 dB LNA gain = 21.6 dB RS = 0 Ω, RFB = ∞ LNA gain = 15.6 dB LNA gain = 17.9 dB LNA gain = 21.6 dB Typ 0.8 10 45 14 3.5 750 1 45 22.5 AVDD2/2 ±2.2 dB dB dB dB dB dB +1.6 0.9 V V MΩ dB dB/V dB ns 10 MHz Degrees Degrees V mA ±2.5 3.3 4.3 6.6 mA/V mA/V mA/V 1.6 1.3 1.0 nV/√Hz nV/√Hz nV/√Hz 5.7 4.5 3.4 dB dB dB 164 162 160 dBFS/√Hz dBFS/√Hz dBFS/√Hz AD9674 Parameter 2 Close In SNR Two-Tone Intermodulation Distortion (IMD3) LO Harmonic Rejection Quadrature Phase Error I/Q Amplitude Imbalance Channel to Channel Matching POWER SUPPLY AVDD1 AVDD2 DVDD DRVDD IAVDD1 IAVDD2 IDVDD IDRVDD Total Power Dissipation (Including Output Drivers) Power-Down Dissipation Standby Power Dissipation ADC Resolution SNR ADC REFERENCE Output Voltage Error Load Regulation at 1.0 mA Input Resistance Data Sheet Test Conditions/Comments −3 dBFS input, fRF = 2.5 MHz, fLO = 40 MHz, 1 kHz offset, 16LO5 mode, one channel enabled −3 dBFS input, fRF = 2.5 MHz, fLO = 40 MHz, 1 kHz offset, 16LO5 mode, eight channels enabled fRF1 = 5.015 MHz, fRF2 = 5.020 MHz, fLO = 80 MHz, ARF1 = −1 dBFS, ARF2 = −21 dBFS, IMD3 relative to ARF2 Min I to Q, all phases, 1 σ I to Q, all phases, 1 σ Phase I to I, Q to Q, 1 σ Amplitude I to I, Q to Q, 1 σ Mode I/Mode II/Mode III/Mode IV1, 3 Max dBc/√Hz −58 dBc −20 dBc Degrees dB Degrees dB 1.8 3.0 1.4 1.8 144/188/224/2943 4 230 239 140 1.9 3.6 1.9 1.9 V V V V mA mA mA mA mA 47/75/57/913 mA 30/48/42/653 mA 125/170/128/1693 109/155/114/1543 1190/1385/ 1365/16003 1325/1535/ 1515/17653 mA mA mW 1215/1425/ 1385/16403 1350/1575/ 1535/18003 500 VREF = 1 V VREF = 1 V mW mW 30 fIN = 5 MHz Unit dBc/√Hz 161 0.15 0.015 0.5 0.25 1.7 2.85 1.3 1.7 TGC mode, LO band mode CW Doppler mode TGC mode, no signal, low band mode TGC mode, no signal, high band mode CW Doppler mode, eight channels enabled RF decimator enabled in Mode III1 and Mode IV,1 digital HPF enabled RF decimator enabled in Mode III1 and Mode IV,1 digital HPF disabled ANSI-644 mode Low power (IEEE 1596.3 similar) mode TGC mode, no signal, RF decimator enabled in Mode III and Mode IV, digital HPF disabled TGC mode, no signal, RF decimator enabled in Mode III1 and Mode IV, 1 digital HPF enabled CW Doppler mode, eight channels enabled Typ 156 630 mW mW 14 75 Bits dB ±50 2 7.5 mV mV kΩ The ADC speed modes depending on the encoding clock rate. For a complete set of definitions and information about how these tests were completed, see the AN-835 Application Note, Understanding High Speed ADC Testing and Evaluation. 3 The slashes mean that the four different power and current values are listed for the four different modes (Mode I, Mode II, Mode III, Mode IV). 4 The overrange condition is specified as 6 dB more than the full-scale input range. 5 The internal LO frequency, fLO, is generated from the supplied multiplier local oscillator frequency, fMLO, by dividing it up by a configurable divider value (M) that can be 4, 8, or 16; the MLO signal is named 4LO, 8LO, or 16LO, accordingly. 1 2 Rev. A | Page 6 of 47 Data Sheet AD9674 DIGITAL SPECIFICATIONS AVDD1 = 1.8 V, AVDD2 = 3.0 V, DVDD = 1.4 V, DRVDD = 1.8 V, 1.0 V internal ADC reference, full temperature range (0°C to 85°C), unless otherwise noted. Table 2. Parameter 1 INPUTS (CLK+, CLK−, TX_TRIG+, TX_TRIG−) Logic Compliance Differential Input Voltage 2 Input Voltage Range Input Common-Mode Voltage Input Resistance (Differential) Input Capacitance INPUTS (MLO±, RESET±) Logic Compliance Differential Input Voltage2 Input Voltage Range Input Common-Mode Voltage Input Resistance (Single-Ended) Input Capacitance LOGIC INPUTS (PDWN, STBY, SCLK, SDIO, ADDRx) Logic 1 Voltage Logic 0 Voltage Input Resistance Input Capacitance LOGIC INPUT (CSB) Logic 1 Voltage Logic 0 Voltage Input Resistance Input Capacitance LOGIC OUTPUT (SDIO) 3 Logic 1 Voltage (IOH = 800 µA) Logic 0 Voltage (IOL = 50 µA) DIGITAL OUTPUTS (DOUTx+, DOUTx−), ANSI-644 Logic Compliance Differential Output Voltage (VOD) Output Offset Voltage (VOS) Output Coding (Default) DIGITAL OUTPUTS (DOUTx+, DOUTx−), LOW POWER, REDUCED SIGNAL OPTION Logic Compliance Differential Output Voltage (VOD) Output Offset Voltage (VOS) Output Coding (Default) LOGIC OUTPUT (GPO0/GPO1/GPO2/GPO3) Logic 0 Voltage (IOL = 50 µA) 1 2 3 Temperature Full Full Full Full 25°C 25°C Full Full Full Full 25°C 25°C Min 0.2 GND − 0.2 Unit V p-p V V kΩ pF LVDS/LVPECL 2 × AVDD2 AVDD2 + 0.2 V p-p V V kΩ pF DRVDD + 0.3 0.3 V V kΩ pF DRVDD + 0.3 0.3 V V kΩ pF AVDD2/2 20 1.5 1.2 Full Full 25°C 25°C 1.2 30 (26 for SDIO) 2 (5 for SDIO) 26 2 Full Full Full Full Full Full Full Full Max CMOS/LVDS/LVPECL 3.6 AVDD1 + 0.2 0.9 15 4 0.250 GND − 0.2 Full Full 25°C 25°C Full Full Full Full Typ 1.79 0.05 V V 454 1.375 mV V 250 1.30 mV V 0.05 V LVDS 247 1.125 Offset binary LVDS 150 1.10 Offset binary For a complete set of definitions and information about how these tests were completed, see the AN-835 Application Note, Understanding High Speed ADC Testing and Evaluation. Specified for LVDS and LVPECL only. Specified for 13 SDIO pins sharing the same connection. Rev. A | Page 7 of 47 AD9674 Data Sheet SWITCHING SPECIFICATIONS AVDD1 = 1.8 V, AVDD2 = 3.0 V, DVDD = 1.4 V, DRVDD = 1.8 V, full temperature range (0°C to 85°C), RF decimator bypassed, and digital HPF bypassed, unless otherwise noted. Table 3. Parameter 1 CLOCK 2 Clock Rate 40 MSPS (Mode I) 65 MSPS (Mode II) 80 MSPS (Mode III) 3 125 MSPS (Mode IV) 4 Clock Pulse Width High (tEH) Clock Pulse Width Low (tEL) OUTPUT PARAMETERS2, 5 Propagation Delay (tPD) Rise Time (tR) (20% to 80%) Fall Time (tF) (20% to 80%) DCO± Period (tDCO) 6 FCO± Propagation Delay (tFCO) DCO± Propagation Delay (tCPD) 7 DCO± to Data Delay (tDATA)7 DCO± to FCO± Delay (tFRAME)7 Data to Data Skew (tDATA-MAX − tDATA-MIN) TX_TRIG± to CLK± Setup Time (tSETUP) TX_TRIG± to CLK± Hold Time (tHOLD) Wake-Up Time (Standby) Wake-Up Time (Power-Down) ADC Pipeline Latency APERTURE Aperture Uncertainty (Jitter), tA LO GENERATION MLO± Frequency 4LO Mode 8LO Mode 16LO Mode RESET± to MLO± Setup Time (tSETUP) RESET± to MLO± Hold Time (tHOLD) Temperature Min Full Full Full Full Full Full 20.5 20.5 20.5 20.5 Full Full Full Full Full Full Full Full Full 25°C 25°C 25°C 25°C Full 10.8 − 1.5 × tDCO Max Unit 40 65 80 125 MHz MHz MHz MHz ns ns 10.8 + 1.5 × tDCO 2 375 16 ns ps ps ns ns ns ps ps ps ns ns µs µs Clock cycles <1 ps rms 3.75 3.75 10.8 − 1.5 × tDCO (tSAMPLE/28) − 300 (tSAMPLE/28) − 300 10.8 300 300 tSAMPLE/7 10.8 tFCO + (tSAMPLE/28) tSAMPLE/28 tSAMPLE/28 ±225 10.8 + 1.5 × tDCO (tSAMPLE/28) + 300 (tSAMPLE/28) + 300 ±400 1 1 25°C Full Full Full Full Full Typ 4 8 16 1 1 40 80 160 tMLO/2 tMLO/2 MHz MHz MHz ns ns For a complete set of definitions and information about how these tests were completed, see the AN-835 Application Note, Understanding High Speed ADC Testing and Evaluation. The clock can be adjusted via the SPI. 3 Mode III must have the RF decimator enabled, unless DVDD runs at 1.8 V and 12-bit mode is configured. 4 Mode IV must have the RF decimator enabled. 5 Measurements were made using the device soldered to FR-4 material. 6 tSAMPLE/7 is based on the number of bits (14) divided by 2 because the interface uses DDR sampling. 7 tSAMPLE/28 is based on the number of bits (14) multiplied by 2 because the delays are based on half duty cycles. 1 2 Rev. A | Page 8 of 47 Data Sheet AD9674 ADC Timing Diagram N–1 AIN tA N tSETUP TX_TRIG+ tHOLD TX_TRIG– tEH tEL CLK– CLK+ tCPD DCO– DCO+ tFRAME tFCO FCO– FCO+ tPD tDATA DOUTx– 11293-002 MSB D12 D0 D1 D2 D3 D4 D5 D7 D6 D8 D9 D10 D11 D12 MSB N – 17 N – 17 N – 17 N – 17 N – 17 N – 17 N – 17 N – 17 N – 17 N – 17 N – 17 N – 17 N – 17 N – 17 N – 16 N – 16 DOUTx+ Figure 2. 14-Bit Data Serial Stream (Default, RF Decimator Bypassed, Digital HPF Bypassed), One Channel per Lane Mode, FCO Mode = Word CW Doppler Timing Diagram tMLO MLO– MLO+ tSETUP tHOLD 11293-003 RESET– RESET+ Figure 3. CW Doppler Mode Input MLO±, Continuous Synchronous RESET± Timing, Sampled on the Falling MLO± Edge, 4LO Mode tMLO MLO– MLO+ tHOLD tSETUP 11293-004 RESET– RESET+ Figure 4. CW Doppler Mode Input MLO±, Continuous Synchronous RESET± Timing, Sampled on the Falling MLO± Edge, 8LO Mode Rev. A | Page 9 of 47 AD9674 Data Sheet tMLO MLO– MLO+ tHOLD tSETUP 11293-105 RESET– RESET+ Figure 5. CW Doppler Mode Input MLO±, Pulse Synchronous RESET± Timing, 4LO/8LO/16LO Mode tMLO MLO– MLO+ tHOLD tSETUP 11293-106 RESET– RESET+ Figure 6. CW Doppler Mode Input MLO±, Pulse Asynchronous RESET± Timing, 4LO/8LO/16LO Mode Rev. A | Page 10 of 47 Data Sheet AD9674 ABSOLUTE MAXIMUM RATINGS Table 4. Parameter AVDD1 to GND AVDD2 to GND DVDD to GND DRVDD to GND GND to GND AVDD2 to AVDD1 AVDD1 to DRVDD AVDD2 to DRVDD Digital Outputs (DOUTx+, DOUTx−, DCO+, DCO−, FCO+, FCO−) to GND LI-x, LG-x, LO-x, LOSW-x, CWI−, CWI+, CWQ−, CWQ+, GAIN+, GAIN−, RESET+, RESET−, MLO+, MLO−, GPO0, GPO1, GPO2, GPO3 to GND CLK+, CLK−, TX_TRIG+, TX_TRIG−, VREF to GND SDIO, PDWN, STBY, SCLK, CSB, ADDRx Operating Temperature Range (Ambient) Storage Temperature Range (Ambient) Maximum Junction Temperature Lead Temperature (Soldering, 10 sec) Rating −0.3 V to +2.0 V −0.3 V to +3.9 V −0.3 V to +2.0 V −0.3 V to +2.0 V −0.3 V to +0.3 V −2.0 V to +3.9 V −2.0 V to +2.0 V −2.0 V to +3.9 V −0.3 V to DRVDD + 0.3 V Stresses at or above those listed under Absolute Maximum Ratings may cause permanent damage to the product. This is a stress rating only; functional operation of the product at these or any other conditions above those indicated in the operational section of this specification is not implied. Operation beyond the maximum operating conditions for extended periods may affect product reliability. THERMAL IMPEDANCE Table 5. Symbol θJA −0.3 V to AVDD2 + 0.3 V ΨJB −0.3 V to AVDD1 + 0.3 V ΨJT −0.3 V to DRVDD + 0.3 V 0°C to 85°C 1 −65°C to +150°C 150°C 300°C Description Junction to ambient thermal resistance, 0.0 m/sec airflow per JEDEC JESD51-2 (still air) Junction to board thermal characterization parameter, 0 m/sec airflow per JEDEC JESD51-8 (still air) Junction to top of package characterization parameter, 0 m/sec airflow per JEDEC JESD51-2 (still air) Value1 22.0 Unit °C/W 9.2 °C/W 0.12 °C/W Results are from simulations. The printed circuit board (PCB) is JEDEC multilayer. Thermal performance for actual applications requires careful inspection of the conditions in the application to determine whether they are similar to those assumed in these calculations. ESD CAUTION Rev. A | Page 11 of 47 AD9674 Data Sheet PIN CONFIGURATION AND FUNCTION DESCRIPTIONS 1 2 3 4 5 6 7 8 9 10 11 12 A LI-E LI-F LI-G LI-H VREF RBIAS GAIN+ GAIN– LI-A LI-B LI-C LI-D B LG-E LG-F LG-G LG-H GND GND CLNA GND LG-A LG-B LG-C LG-D C LO-E LO-F LO-G LO-H GND GND GND GND LO-A LO-B LO-C LO-D GND GND GND GND LOSW-A LOSW-B LOSW-C LOSW-D E GND AVDD2 AVDD2 AVDD2 GND GND GND GND AVDD2 AVDD2 AVDD2 GND F AVDD1 GND AVDD1 GND AVDD1 GND GND AVDD1 GND AVDD1 GND AVDD1 G GND AVDD1 GND DVDD GND GND GND GND AVDD1 GND DVDD GND H CLK– TX_TRIG– GND GND GND GND ADDR4 ADDR3 ADDR2 ADDR1 ADDR0 CSB J CLK+ TX_TRIG+ CWQ+ GND CWI+ AVDD2 MLO+ RESET– GPO3 GPO1 PDWN SDIO K GND GND CWQ– GND CWI– AVDD2 MLO– RESET+ GPO2 GPO0 STBY SCLK DCO+ FCO+ DOUTD+ DOUTC+ DOUTB+ DOUTA+ DRVDD DCO– FCO– DOUTD– DOUTC– DOUTB– DOUTA– M DRVDD DOUTH+ DOUTG+ DOUTF+ DOUTE+ GND DOUTH– DOUTG– DOUTF– DOUTE– Figure 7. Pin Configuration 4 2 1 3 6 5 10 8 7 9 12 11 A B C D E F G H J K L M TOP VIEW (Not to Scale) Figure 8. CSP_BGA Pin Location Rev. A | Page 12 of 47 11293-006 L GND 11293-005 D LOSW-E LOSW-F LOSW-G LOSW-H Data Sheet AD9674 Table 6. Pin Function Descriptions Pin No. B5, B6, B8, C5 to C8, D5 to D8, E1, E5 to E8, E12, F2, F4, F6, F7, F9, F11, G1, G3, G5 to G8, G10, G12, H3 to H6, J4, K1, K2, K4, M1, M12 F1, F3, F5, F8, F10, F12, G2, G9 G4, G11 E2 to E4, E9 to E11, J6, K6 B7 L1, L12 C1 D1 A1 B1 C2 D2 A2 B2 C3 D3 A3 B3 C4 D4 A4 B4 H1 J1 H2 J2 H11 H10 H9 H8 H7 M2 L2 M3 L3 M4 L4 M5 L5 M6 L6 M7 L7 M8 L8 M9 L9 M10 L10 M11 Mnemonic GND Description Ground. Tie to a quiet analog ground. AVDD1 DVDD AVDD2 CLNA DRVDD LO-E LOSW-E LI-E LG-E LO-F LOSW-F LI-F LG-F LO-G LOSW-G LI-G LG-G LO-H LOSW-H LI-H LG-H CLK− CLK+ TX_TRIG− TX_TRIG+ ADDR0 ADDR1 ADDR2 ADDR3 ADDR4 DOUTH− DOUTH+ DOUTG− DOUTG+ DOUTF− DOUTF+ DOUTE− DOUTE+ DCO− DCO+ FCO− FCO+ DOUTD− DOUTD+ DOUTC− DOUTC+ DOUTB− DOUTB+ DOUTA− 1.8 V Analog Supply. 1.4 V/1.8 V Digital Supply. 3.0 V Analog Supply. LNA External Capacitor. 1.8 V Digital Output Driver Supply. LNA Analog Inverted Output for Channel E. LNA Analog Switched Output for Channel E. LNA Analog Input for Channel E. LNA Ground for Channel E. LNA Analog Inverted Output for Channel F. LNA Analog Switched Output for Channel F. LNA Analog Input for Channel F. LNA Ground for Channel F. LNA Analog Inverted Output for Channel G. LNA Analog Switched Output for Channel G. LNA Analog Input for Channel G. LNA Ground for Channel G. LNA Analog Inverted Output for Channel H. LNA Analog Switched Output for Channel H. LNA Analog Input for Channel H. LNA Ground for Channel H. Clock Input Complement. Clock Input True. Transmit Trigger Complement. Transmit Trigger True. Chip Address Bit 0. Chip Address Bit 1. Chip Address Bit 2. Chip Address Bit 3. Chip Address Bit 4. ADC Channel H Digital Output Complement. ADC Channel H Digital Output True. ADC Channel G Digital Output Complement. ADC Channel G Digital Output True. ADC Channel F Digital Output Complement. ADC Channel F Digital Output True. ADC Channel E Digital Output Complement. ADC Channel E Digital Output True. Digital Clock Output Complement. Digital Clock Output True. Frame Clock Digital Output Complement. Frame Clock Digital Output True. ADC Channel D Digital Output Complement. ADC Channel D Digital Output True. ADC Channel C Digital Output Complement. ADC Channel C Digital Output True. ADC Channel B Digital Output Complement. ADC Channel B Digital Output True. ADC Channel A Digital Output Complement. Rev. A | Page 13 of 47 AD9674 Pin No. L11 K11 J11 K12 J12 H12 B9 A9 D9 C9 B10 A10 D10 C10 B11 A11 D11 C11 B12 A12 D12 C12 K10 J10 K9 J9 J8 K8 K7 J7 A8 A7 A6 A5 K5 J5 K3 J3 Data Sheet Mnemonic DOUTA+ STBY PDWN SCLK SDIO CSB LG-A LI-A LOSW-A LO-A LG-B LI-B LOSW-B LO-B LG-C LI-C LOSW-C LO-C LG-D LI-D LOSW-D LO-D GPO0 GPO1 GPO2 GPO3 RESET− RESET+ MLO− MLO+ GAIN− GAIN+ RBIAS VREF CWI− CWI+ CWQ− CWQ+ Description ADC Channel A Digital Output True. Standby Power-Down. Full Power-Down. Serial Clock. Serial Data Input/Output. Chip Select Bar. LNA Ground for Channel A. LNA Analog Input for Channel A. LNA Analog Switched Output for Channel A. LNA Analog Inverted Output for Channel A. LNA Ground for Channel B. LNA Analog Input for Channel B. LNA Analog Switched Output for Channel B. LNA Analog Inverted Output for Channel B. LNA Ground for Channel C. LNA Analog Input for Channel C. LNA Analog Switched Output for Channel C. LNA Analog Inverted Output for Channel C. LNA Ground for Channel D. LNA Analog Input for Channel D. LNA Analog Switched Output for Channel D. LNA Analog Inverted Output for Channel D. General-Purpose Open-Drain Output 0. General-Purpose Open-Drain Output 1. General-Purpose Open-Drain Output 2. General-Purpose Open-Drain Output 3. Synchronizing Input for LO Divide-by-M Counter Complement. Synchronizing Input for LO Divide-by-M Counter True. CW Doppler Multiple Local Oscillator Input Complement. CW Doppler Multiple Local Oscillator Input True. Gain Control Voltage Input Complement. Gain Control Voltage Input True. External Resistor to Set the Internal ADC Core Bias Current. Voltage Reference Input/Output. CW Doppler I Output Complement. CW Doppler I Output True. CW Doppler Q Output Complement. CW Doppler Q Output True. Rev. A | Page 14 of 47 Data Sheet AD9674 TYPICAL PERFORMANCE CHARACTERISTICS TGC MODE Mode I = fSAMPLE = 40 MSPS, fIN = 5 MHz, LO band mode, RS = 50 Ω, RFB = ∞ (unterminated), LNA gain = 21.6 dB, LNA bias = midhigh, PGA gain = 27 dB, VGAIN = (GAIN+) − (GAIN−) = 1.6 V, AAF LPF cutoff = fSAMPLE/3, HPF cutoff = LPF cutoff/12 (default), RF decimator bypassed, and digital HPF bypassed, unless otherwise noted. 25 2.0 PERCENTAGE OF UNITS (%) 1.5 GAIN ERROR (dB) 1.0 0°C 0.5 0 25°C –0.5 85°C –1.0 20 15 10 5 –1.5 –0.4 0 0.4 0.8 1.2 1.6 VGAIN (V) GAIN ERROR (dB) Figure 12. Gain Error Histogram, VGAIN = 1.28 V Figure 9. Gain Error vs. VGAIN 20 25 20 PERCENTAGE OF UNITS (%) 15 10 15 10 5 5 0 GAIN ERROR (dB) –1.0 –0.9 –0.8 –0.7 –0.6 –0.5 –0.4 –0.3 –0.2 –0.1 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 11293-108 –1.0 –0.9 –0.8 –0.7 –0.6 –0.5 –0.4 –0.3 –0.2 –0.1 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 0 CHANNEL-TO-CHANNEL GAIN MATCHING (dB) 11293-111 PERCENTAGE OF UNITS (%) 11293-110 –0.8 –1.0 –0.9 –0.8 –0.7 –0.6 –0.5 –0.4 –0.3 –0.2 –0.1 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 0 –1.2 11293-107 –2.0 –1.6 Figure 13. Gain Matching Histogram, VGAIN = −1.2 V Figure 10. Gain Error Histogram, VGAIN = −1.28 V 20 35 PERCENTAGE OF UNITS (%) 25 20 15 10 15 10 5 5 CHANNEL-TO-CHANNEL GAIN MATCHING (dB) Figure 14. Gain Matching Histogram, VGAIN = 1.2 V Figure 11. Gain Error Histogram, VGAIN = 0 V Rev. A | Page 15 of 47 11293-112 11293-109 GAIN ERROR (dB) –1.0 –0.9 –0.8 –0.7 –0.6 –0.5 –0.4 –0.3 –0.2 –0.1 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 0 0 –1.0 –0.9 –0.8 –0.7 –0.6 –0.5 –0.4 –0.3 –0.2 –0.1 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 PERCENTAGE OF UNITS (%) 30 AD9674 Data Sheet 1.4 70 LNA GAIN = 17.9dB 1.2 66 64 1.0 SNR (dBFS) INPUT REFERRED NOISE (nV/√Hz) 68 0.8 LNA GAIN = 15.6dB 62 60 LNA GAIN = 21.6dB 58 56 0.6 54 2 3 4 5 6 7 8 9 10 FREQUENCY (MHz) 50 10 11293-008 1 15 20 25 30 35 Figure 15. Short-Circuit, Input Referred Noise vs. Frequency 45 50 55 Figure 18. SNR vs. Channel Gain and LNA Gain, Output Amplitude (AOUT) = −1.0 dBFS 74 –132 PGA GAIN = 21dB PGA GAIN = 21dB 72 –134 70 –136 PGA GAIN = 24dB 68 SNR (dBFS) OUTPUT REFERRED NOISE (dBc/√Hz) 40 CHANNEL GAIN (dB) 11293-011 52 0.4 –138 –140 66 64 PGA GAIN = 27dB 62 60 –142 PGA GAIN = 30dB 58 –144 56 5 10 25 20 15 30 35 40 45 CHANNEL GAIN (dB) 54 –5 11293-009 0 0 5 10 15 20 25 Figure 16. Short-Circuit, Output Referred Noise vs. Channel Gain, PGA Gain = 21 dB, VGAIN = 1.6 V 0 40 45 50 55 SPEED MODE = I (40MSPS) LO BAND MODE PGA GAIN = 21dB 68 –1 66 –2 PGA GAIN = 24dB AMPLITUDE (dBFS) 64 62 PGA GAIN = 27dB 58 PGA GAIN = 30dB 56 54 –3 –4 –5 –6 –7 –8 52 –9 15 20 25 30 35 40 45 50 55 CHANNEL GAIN (dB) –10 0 5 10 15 20 INPUT FREQUENCY (MHz) Figure 20. AAF Pass-Band Response, LPF Cutoff = 1 × (1/3) × fSAMPLE, HPF = LPF Cutoff/12 Figure 17. SNR vs. Channel Gain and PGA Gain, AOUT = −1.0 dBFS Rev. A | Page 16 of 47 11293-013 LNA GAIN = 21.6dB 50 10 11293-010 SNR (dBFS) 35 Figure 19. SNR vs. Channel Gain and PGA Gain, Input Amplitude (AIN) = −45 dBm 70 60 30 CHANNEL GAIN (dB) 11293-117 LNA GAIN = 21.6dB –146 –5 Data Sheet AD9674 –30 –40 THIRD-ORDER, MIN VGAIN –50 THIRD-ORDER, MAX VGAIN –60 –70 SECOND-ORDER, MIN VGAIN –80 –90 2 3 4 5 6 7 8 9 10 11 INPUT FREQUENCY (MHz) 11293-014 SECOND-ORDER, MAX V GAIN –100 –40 –50 VGAIN = 0V –70 –80 –90 VGAIN = +1.6V –100 –110 –120 –40 –35 –30 –25 –20 –15 –10 0 –5 Figure 24. Second-Order Harmonic Distortion vs. ADC Output Level (AOUT) THIRD-ORDER HARMONIC DISTORTION (dBFS) 0 PGA GAIN = 24dB –10 –20 –30 –40 –50 –60 LNA GAIN = 17.9dB –70 LNA GAIN = 21.6dB –80 LNA GAIN = 15.6dB –90 –100 10 VGAIN = –1.2V –60 0 15 20 25 30 35 40 45 50 CHANNEL GAIN (dB) Figure 22. Second-Order Harmonic Distortion vs. Channel Gain, AOUT = −1.0 dBFS –10 –20 –30 –40 VGAIN = –1.2V –50 –60 VGAIN = 0V –70 –80 –90 VGAIN = +1.6V –100 –110 –120 –40 –35 –30 –25 –20 –15 –10 –5 0 ADC OUTPUT LEVEL (dBFS) Figure 25. Third-Order Harmonic Distortion vs. ADC Output Level (AOUT) 0 –100 PGA GAIN = 24dB –10 –110 PHASE NOISE (dBc/√Hz) –20 –30 –40 LNA GAIN = 17.9dB –50 LNA GAIN = 21.6dB –60 LNA GAIN = 15.6dB –70 –120 –130 –140 –80 –150 –90 –100 10 15 20 25 30 35 40 45 CHANNEL GAIN (dB) Figure 23. Third-Order Harmonic Distortion vs. Channel Gain, AOUT = −1.0 dBFS –160 100 11293-016 THIRD-ORDER HARMONIC DISTORTION (dBFS) –30 ADC OUTPUT LEVEL (dBFS) 11293-015 SECOND-ORDER HARMONIC DISTORTION (dBFS) Figure 21. Second-Order and Third-Order Harmonic Distortion vs. Input Frequency, AOUT = −1.0 dBFS –20 11293-123 –20 0 –10 11293-122 LNA GAIN = 21.6dB PGA GAIN = 27dB MIN VGAIN, AOUT = –12.0dBFS MAX VGAIN, AOUT = –1.0dBFS 1k 10k OFFSET FREQUENCY FROM CARRIER (Hz) 100k 11293-017 HARMONIC DISTORTION (dBFS) SECOND-ORDER HARMONIC DISTORTION (dBFS) 0 –10 Figure 26. TGC Path Phase Noise, LNA Gain = 21.6 dB, PGA Gain = 27 dB, VGAIN = 0 V Rev. A | Page 17 of 47 Data Sheet 0 8 7 6 5 4 3 2 1 0 100k –10 fIN1 = 5.0MHz fIN2 = 5.01MHz –20 FUND1 LEVEL = –1dBFS FUND2 LEVEL = –21dBFS –30 1M 0 –10 –20 –30 –40 –50 –60 –70 –80 –90 100k 10M FREQUENCY (Hz) IMD3 (dBFS) –40 100M –50 –60 –90 –100 10M FREQUENCY (Hz) 100M –120 –40 –20 –25 –20 –15 –10 –5 0 Figure 29. IMD3 vs. ADC Output Level (AOUT) 7 FUND1 LEVEL = –1dBFS FUND2 LEVEL = –21dBFS RS = 50Ω 6 NOISE FIGURE (dB) –30 –40 –50 –60 RIN = 1000Ω –70 –80 5 4 3 2 RIN = 50Ω –90 20 25 30 35 RIN = 300Ω 40 CHANNEL GAIN (dB) Figure 28. IMD3 vs. Channel Gain 45 50 1 11293-019 IMD3 (dBFS) –30 ADC OUTPUT LEVEL (dBFS) fIN1 = 2.3MHz fIN2 = 2.31MHz –100 15 –35 11293-127 1M VGAIN = 0V 0 2 4 6 8 10 12 FREQUENCY (MHz) 14 16 18 20 11293-020 0 VGAIN = +1.6V –110 Figure 27. LNA Input Impedance Magnitude and Phase, Unterminated –10 VGAIN = –1.2V –70 –80 11293-018 PHASE (Degrees) MAGNITUDE (kΩ) AD9674 Figure 30. Noise Figure vs. Frequency, RS = RIN = 100 Ω, LNA Gain = 17.9 dB, PGA Gain = 30 dB, VGAIN = 1.6 V Rev. A | Page 18 of 47 Data Sheet AD9674 CW DOPPLER MODE fIN = 5 MHz, fLO = 20 MHz, 4LO mode, RS = 50 Ω, LNA gain = 21.6 dB, LNA bias = midhigh, all CW channels enabled, phase rotation = 0°. 10 165 9 160 155 SNR (dBc/√Hz) 7 6 5 4 3 150 145 140 2 135 0 0 1000 2000 3000 4000 5000 6000 7000 8000 9000 10000 BASEBAND FREQUENCY (Hz) Figure 31. Noise Figure vs. Baseband Frequency 130 0 1000 2000 3000 4000 5000 6000 7000 8000 9000 10000 BASEBAND FREQUENCY (Hz) Figure 32. SNR vs. Baseband Frequency, −3 dBFS LNA Input Rev. A | Page 19 of 47 11293-022 1 11293-021 NOISE FIGURE (dB) 8 AD9674 Data Sheet THEORY OF OPERATION MLO– MLO+ RESET+ RESET– RFB1 LO-x RFB2 LOSW-x T/R SWITCH C S LO GENERATION CWI+ CWI– CWQ+ CWQ– LI-x ATTENUATOR –45dB TO 0dB LNA 15.6dB, 17.9dB, 21.6dB CLG TRANSDUCER GAIN INTERPOLATOR gm GAIN+ POST AMP FILTER 14-BIT ADC FILTER DEC SERIAL LVDS DOUTx+ DOUTx– 21dB, 24dB, 27dB, 30dB 11293-023 LG-x CSH GAIN– Figure 33. Simplified Block Diagram of a Single Channel Each channel of the AD9674 contains both a TGC signal path and a CW Doppler signal path. Common to both signal paths, the LNA provides four user adjustable input impedance termination options for matching different probe impedances. The CW Doppler path includes an I/Q demodulator with the programmable phase rotation needed for analog beamforming. The TGC path includes a differential X-AMP® VGA, an antialiasing filter, an ADC, and a digital HPF and RF decimator. Figure 33 shows a simplified block diagram with the external components. TGC OPERATION The system gain is distributed as listed in Table 7. Table 7. Channel Analog Gain Distribution Section LNA Attenuator VGA Amplifier Filter ADC Nominal Gain (dB) 15.6/17.9/21.6 (LNAGAIN)1 −45 to 0 (VGAATT) 21/24/27/30 (PGAGAIN)1 0 0 The slashes represent the LNA and PGA gain settings that can change using SPI registers. 1 Each LNA output is dc-coupled to a VGA input. The VGA consists of an attenuator with a −45 dB to 0 dB range followed by an amplifier with 21 dB, 24 dB, 27 dB, or 30 dB of gain. The X-AMP gain interpolation technique results in low gain error and uniform bandwidth; differential signal paths minimize distortion. The linear in dB gain (law conformance) range of the TGC path is 45 dB. The slope of the gain control interface is 14 dB/V, and the gain control range is −1.6 V to +1.6 V. Equation 1 is the expression for the differential voltage, VGAIN, at the gain control interface. Equation 2 is the expression for the VGA attenuation, VGAATT, as a function of VGAIN. VGAIN (V) = (GAIN+) − (GAIN−) (1) VGAATT (dB) = −14 (dB/V) × (1.6 − VGAIN) (2) The total channel gain can then be calculated as shown in Equation 3. Channel Gain (dB) = LNAGAIN + VGAATT + PGAGAIN (3) In its default condition, the LNA has a gain of 21.6 dB (12×), and the VGA postamplifier gain is 24 dB. If the voltage on the GAIN+ pin is 0 V and the voltage on the GAIN− pin is 1.6 V (45.1 dB attenuation), the total gain of the channel is 0.5 dB if the LNA input is unmatched. The channel gain is −5.5 dB if the LNA is matched to 50 Ω (RFB = 300 Ω). However, if the voltage on the GAIN+ pin is 1.6 V and the voltage on the GAIN− pin is 0 V (0 dB attenuation), VGAATT is 0 dB. This results in a total gain of 45.3 dB through the TGC path if the LNA input is unmatched, or in a total gain of 39.3 dB, if the LNA input is matched. Similarly, if the LNA input is unmatched and has a gain of 21.6 dB (12×), and the VGA postamplifier gain is 30 dB, the channel gain is approximately 52 dB with 0 dB VGAATT. In addition to the analog VGA attenuation described in Equation 2, the attenuation level can be digitally controlled in 3.5 dB increments. Equation 3 is still valid, and the value of VGAATT is equal to the attenuation level set in Address 0x011, Bits[7:4]. Low Noise Amplifier (LNA) Good system sensitivity relies on a proprietary ultralow noise LNA at the beginning of the signal chain, which minimizes the noise contribution in the following VGA. Active impedance control optimizes noise performance for applications that benefit from input impedance matching. The LNA input, LI-x, is capacitively coupled to the source. An on-chip bias generator establishes dc input bias voltages of approximately 2.2 V and centers the output common-mode levels at 1.5 V (AVDD2 divided by 2). A capacitor, CLG, of the same value as the input coupling capacitor, CS, is connected from LG-x to ground. The LNA supports three gain settings, 21.6 dB, 17.9 dB, or 15.6 dB, set through the SPI. Overload protection ensures quick recovery time from large input voltages. Rev. A | Page 20 of 47 Data Sheet AD9674 The LNA consists of a single-ended voltage gain amplifier with differential outputs; the negative output is externally available on two output pins (LO-x and LOSW-x) that are controlled via internal switches. This configuration allows active input impedance synthesis of three different impedance values (and an unterminated value) by connecting up to two external resistances in parallel and controlling the internal switch states via the SPI. For example, with a fixed gain of 8× (17.9 dB), an active input termination is synthesized by connecting a feedback resistor between the negative output pin, LO-x, and the positive input pin, LI-x. This well-known technique is used for interfacing multiple probe impedances to a single system. The input resistance calculation is shown in Equation 4. RIN = (RFB1 + 20 Ω) || (RFB2 + 20 Ω) + 30 Ω 1 + A 2 (4) where A/2 is the single-ended gain or the gain from the LI-x inputs to the LO-x outputs, RFB1 and RFB2 are the external feedback resistors, the 20 Ω is the internal switch on resistance, and the 30 Ω is an internal series resistance common to the two internal switches. RFB can equal to RFB1, RFB2, or (RFB1 + 20 Ω)||(RFB2 + 20 Ω) depending on the connection status of the internal switches. Because the amplifier has a gain of 8× from its input to its differential output, it is important to note that the gain, A/2, is the gain from the LI-x pin to the LO-x pin, and that it is 6 dB less than the gain of the amplifier, or 12.1 dB (4×). The input resistance is reduced by an internal bias resistor of 6 kΩ in parallel with the source resistance connected to the LI-x pin and with the LG-x pin ac grounded. Equation 5 can be used to calculate the required RFB for a desired RIN, even for higher values of RIN. RIN = (RFB1 + 20 Ω) || (RFB2 + 20 Ω) + 30 Ω || 6 k Ω 1 + A 2 RFB is the resulting impedance of the RFB1 and RFB2 combination (see Figure 33). Using Address 0x02C in the SPI memory, the AD9674 can be programmed for four impedance matching options: three active terminations and one unterminated option. Table 8 shows an example of how to select RFB1 and RFB2 for RIN = 66 Ω, 100 Ω, and 200 Ω input impedances for an LNA gain = 21.6 dB (12×). Table 8. Active Termination Example for LNA Gain = 21.6 dB, RFB1 = 650 Ω, and RFB2 = 1350 Ω Reg. 0x02C, Bits[1:0] 00 (default) 01 10 11 1 LO-x Switch On On Off Off LOSW-x Switch Off On On Off RFB (Ω) RFB1 RFB1||RFB2 RFB2 ∞ RIN (Ω) (Eq. 4) 100 66 200 ∞ N/A means not applicable. The bandwidth (BW) of the LNA is greater than 80 MHz. Ultimately, the BW of the LNA limits the accuracy of the synthesized RIN. RIN = RS up to approximately 200 Ω. The best match is between 100 kHz and 10 MHz where the lower frequency limit is determined by the size of the ac coupling capacitors and the upper limit is determined by the LNA BW. Furthermore, the input capacitance and RS limit the BW at higher frequencies. Figure 34 shows input resistance (RIN) vs. frequency for various RFB values. 1k RS = 500Ω, RFB = 2kΩ RS = 200Ω, RFB = 800Ω RS = 100Ω, RFB = 400Ω, CSH = 20pF 100 RS = 50Ω, RFB = 200Ω, CSH = 70pF 10 100k 1M 10M FREQUENCY (Hz) (5) For example, to set RIN to 200 Ω with a single-ended LNA gain of 12.1 dB (4×), the value of RFB from Equation 4 must be 950 Ω while the switch for RFB2 is open. If the more accurate equation (Equation 5) is used to calculate RIN, the value is then 194 Ω instead of 200 Ω, resulting in a gain error of less than 0.27 dB. Some factors, such as the presence of a dynamic source resistance, may influence the absolute gain accuracy more significantly. RS (Ω) 100 50 200 N/A1 100M 11293-024 Active Impedance Matching At higher frequencies, the input capacitance of the LNA must be considered. The user must determine the level of matching accuracy and adjust RFB accordingly. INPUT RESISTANCE (Ω) Low value feedback resistors and the current driving capability of the output stage allow the LNA to achieve a low input referred noise voltage of 0.78 nV/√Hz (at a gain of 21.6 dB). On-chip resistor matching results in precise single-ended gains, which are critical for accurate impedance control. The use of a fully differential topology and negative feedback minimizes distortion. Low second-order harmonic distortion is particularly important in harmonic ultrasound imaging applications. Figure 34. Input Resistance (RIN) vs. Frequency for Various RFB Values (Effects of RS and CSH Are Also Shown) For larger RIN values, parasitic capacitance starts rolling off the signal BW before the LNA can produce peaking. CSH further degrades the match; therefore, do not use CSH for values of RIN that are greater than 100 Ω (see Figure 34). Rev. A | Page 21 of 47 AD9674 Data Sheet Figure 36 shows the noise figure as it relates to RS for various values of RIN, which is helpful for design purposes. 8 7 LNA Gain (dB) 15.6 17.9 21.6 15.6 17.9 21.6 15.6 17.9 21.6 6 RFB (Ω) 150 200 300 350 450 650 750 950 1350 Minimum CSH (pF) 90 70 50 30 20 10 Not applicable Not applicable Not applicable RIN = 50Ω RIN = 75Ω RIN = 100Ω RIN = 200Ω UNTERMINATED 5 4 3 2 1 0 10 1k 100 RS (Ω) Figure 36. Noise Figure vs. RS for Various Fixed Values of RIN, Active Termination Matched Inputs, VGAIN = 1.6 V LNA Noise The short-circuit noise voltage (input referred noise) is an important limit on system performance. The short-circuit noise voltage for the LNA is 0.78 nV/√Hz at a gain of 21.6 dB, including the VGA noise at a VGA postamplifier gain of 27 dB. These measurements, which were taken without a feedback resistor, provide the basis for calculating the input noise and noise figure (NF) performance. Figure 35 and Figure 36 are simulations of noise figure vs. RS results with different input configurations and an input referred noise voltage of 2.5 nV/√Hz for the VGA. The unterminated (RFB = ∞) operation exhibits the lowest equivalent input noise and noise figure. Figure 36 shows the noise figure vs. the source resistance rising at low RS, where the LNA voltage noise is large compared with the source noise, and at high RS due to the noise contribution from RFB. The lowest NF is achieved when RS matches RIN. CLNA Connection CLNA (Ball B7) must have a 1 nF capacitor attached to AVDD2. DC Offset Correction/High-Pass Filter The AD9674 LNA architecture is designed to correct for dc offset voltages that can develop on the external CS capacitor due to leakage of the transmit/receive switch during ultrasound transmit cycles. The dc offset correction, as shown in Figure 37, provides a feedback mechanism to the LG-x input of the LNA to correct for this dc voltage. Figure 35 shows the relative noise figure performance. With an LNA gain of 21.6 dB, the input impedance is swept with RS to preserve the match at each point. The noise figures for a source impedance of 50 Ω are 7 dB, 4 dB, and 2.5 dB for the shunt termination, active termination, and unterminated configurations, respectively. The noise figures for 200 Ω are 4.5 dB, 1.7 dB, and 1 dB, respectively. 12.0 AD9674 CFB RFB1 LO-x RFB2 LOSW-x T/R SWITCH CS LI-x LG-x CSH CLG LNA 15.6dB, 17.9dB, 21.6dB TRANSDUCER gm DC OFFSET CORRECTION 10.5 Figure 37. Simplified LNA Input Configuration 7.5 SHUNT TERMINATION 6.0 4.5 3.0 ACTIVE TERMINATION UNTERMINATED 1.5 0 10 100 RS (Ω) 1k 11293-025 NOISE FIGURE (dB) 9.0 Figure 35. Noise Figure vs. RS for Shunt Termination, Active Termination Matched and Unterminated Inputs, VGAIN = 1.6 V Rev. A | Page 22 of 47 11293-035 RIN (Ω) 50 50 50 100 100 100 200 200 200 NOISE FIGURE (dB) Table 9. Active Termination External Component Values 11293-026 Table 9 lists the recommended values for RFB and CSH in terms of RIN. CFB is needed in series with RFB because the dc levels at the LO-x pin and the LI-x pin are unequal. Data Sheet AD9674 Table 10. High-Pass Filter Cutoff Frequency, fHP, for CLG = 10 nF Addr. 0x120[4:3] 00 (default) 01 10 11 gm (mS) 0.5 mS 1.0 mS 1.5 mS 2.0 mS LNAGAIN = 15.6 dB 41 kHz 83 kHz 133 kHz 167 kHz LNAGAIN = 17.9 dB 55 kHz 110 kHz 178 kHz 220 kHz LNAGAIN = 21.6 dB 83 kHz 167 kHz 267 kHz 330 kHz For other values of CLG, the high-pass filter cutoff frequency can be determined by scaling the values from Table 10 or by calculating the value based on CLG, LNAGAIN, and gm, as shown in Equation 6. f HP (C LG ) = 10 nF g 1 × LNAGAIN × m = f HP (Table 10) × 2× π C LG C LG (6) Variable Gain Amplifier (VGA) The differential X-AMP VGA provides precise input attenuation and interpolation. It has a low input referred noise of 2.5 nV/√Hz and excellent gain linearity. The VGA is driven by a fully differential input signal from the LNA. The X-AMP architecture produces a linear in dB gain law conformance and low distortion levels, deviating only ±0.5 dB or less from the ideal. The gain slope is monotonic with respect to the control voltage and is stable with variations in process, temperature, and supply. The resulting total gain range is 45 dB, allowing range loss at the endpoints. The X-AMP inputs are part of a programmable gain amplifier (PGA) that completes the VGA. The PGA in the VGA can be programmed to a gain of 21 dB, 24 dB, 27 dB, or 30 dB, allowing optimization of the channel gain for different imaging modes in the ultrasound system. The VGA bandwidth is greater than 100 MHz. The input stage is designed to ensure excellent frequency response uniformity across the gain setting. For TGC mode, the design of the input stage minimizes time delay variation across the gain range. Gain Control The analog gain control interface, GAIN±, is a differential input. VGAIN varies the gain of all VGAs through the interpolator by selecting the appropriate input stages connected to the input attenuator. The nominal VGAIN range is 14 dB/V from −1.6 V to +1.6 V, with the best gain linearity from approximately −1.44 V to +1.44 V, where the error is typically less than ±0.5 dB. For VGAIN voltages greater than +1.44 V and less than −1.44 V, the error increases. The value of GAIN± can exceed the supply voltage by 1 V without gain foldover. The gain control response time is less than 750 ns to settle within 10% of the final value for a change from minimum to maximum gain. The differential input pins, GAIN+ and GAIN−, can interface to an amplifier, as shown in Figure 38. Decouple and drive the GAIN+ and GAIN− pins to accommodate a 3.2 V full-scale input. 249Ω AD9674 GAIN+ ±0.8V DC 100Ω AT 0.8V CM GAIN– 100Ω 0.01µF 249Ω ADA4938-1/ ADA4938-2 0.01µF AVDD2 31.3kΩ ±1.6V 0.8V CM 249Ω ±0.8V DC AT 0.8V CM 249Ω 10kΩ 11293-027 The feedback acts as a high-pass filter providing dynamic correction of the dc offset. The cutoff frequency of the high-pass filter response is dependent on the value of the CLG capacitor, the gain of the LNA (LNAGAIN), and the gm of the feedback transconductance amplifier. The gm value is programmed in Address 0x120, Bits[4:3]. It is required that CS be equal to CLG for proper operation. Figure 38. Differential GAIN± Pin Configuration The analog gain control can be disabled and the attenuator can be controlled digitally using Address 0x011, Bits[7:4]. The control range is 45 dB, and the step size is 3.5 dB. VGA Noise In a typical application, a VGA compresses a wide dynamic range input signal to within the input span of an ADC. The input referred noise of the LNA limits the minimum resolvable input signal, whereas the output referred noise, which depends primarily on the VGA, limits the maximum instantaneous dynamic range that can be processed at any one particular gain control voltage. This latter limit is set in accordance with the total noise floor of the ADC. The output referred noise is a flat 40 nV/√Hz (postamplifier gain = 24 dB) over most of the gain range because it is dominated by the fixed output referred noise of the VGA. At the high end of the gain control range, the noise of the LNA and the source prevail. The input referred noise reaches its minimum value near the maximum gain control voltage, where the input referred contribution of the VGA is miniscule. At lower gains, the input referred noise and, therefore, the noise figure increase as the gain decreases. The instantaneous dynamic range of the system is not lost, however, because the input capacity increases as the input referred noise increases. The contribution of the ADC noise floor has the same dependence. The important relationship is the magnitude of the VGA output noise floor relative to that of the ADC. Gain control noise is a concern in very low noise applications. Thermal noise in the gain control interface can modulate the channel gain. The resulting noise is proportional to the output signal level and is usually evident only when a large signal is present. Take care to minimize noise impinging at the GAIN± inputs. An external RC filter can be used to remove VGAIN source noise. The filter bandwidth must be sufficient to accommodate the desired control bandwidth and attenuate unwanted switching noise from the external digital-to-analog converters used to drive the gain control. The AD9674 can bypass the GAIN± inputs and control the gain of the attenuator digitally (see the Gain Control section). This mode removes any external noise contributions when active gain control is not needed. Rev. A | Page 23 of 47 AD9674 Data Sheet Antialiasing Filter (AAF) The filter that the signal reaches prior to the ADC is used to reject dc signals and to band limit the signal for antialiasing. The antialiasing filter is a combination of a single-pole high-pass filter and a second-order low-pass filter. The high-pass filter can be configured as a ratio of the low-pass filter cutoff frequency. This is selectable using Address 0x02B, Bits[1:0]. The filter uses on-chip tuning to trim the capacitors and set the desired low-pass cutoff frequency and reduce variations. The default −3 dB low-pass filter cutoff is 1/3, 1/4.5, or 1/6 of the ADC sample clock rate. The cutoff can be scaled to 0.75, 0.8, 0.9, 1.0, 1.13, 1.25, or 1.45 times this frequency using Address 0x00F. The cutoff tolerance (±10%) is maintained from 8 MHz to 18 MHz for low band mode or 13.5 MHz to 30 MHz for high band mode. Table 11 and Table 12 calculate the valid SPI-selectable low-pass filter settings and the expected cutoff frequencies for low band mode and high band mode at the minimum and the maximum sample frequency in each speed mode. Table 11. SPI-Selectable Low-Pass Filter Cutoff Options for Low Band Mode at Example Sampling Frequencies Address 0x00F[7:3] 0 0000 LPF Cutoff Frequency (MHz) 1.45 × (1/3) × fSAMPLE 20.5 9.91 0 0001 1.25 × (1/3) × fSAMPLE 8.54 0 0010 1.13 × (1/3) × fSAMPLE 0 0011 1.0 × (1/3) × fSAMPLE 0 0100 0.9 × (1/3) × fSAMPLE 0 0101 0.8 × (1/3) × fSAMPLE 0 0110 0.75 × (1/3) × fSAMPLE 0 1000 1.45 × (1/4.5) × fSAMPLE 0 1001 1.25 × (1/4.5) × fSAMPLE 0 1010 1.13 × (1/4.5) × fSAMPLE 0 1011 1.0 × (1/4.5) × fSAMPLE 0 1100 0.9 × (1/4.5) × fSAMPLE 0 1101 0.8 × (1/4.5) × fSAMPLE 0 1110 0.75 × (1/4.5) × fSAMPLE 1 0000 1.45 × (1/6) × fSAMPLE 1 0001 1.25 × (1/6) × fSAMPLE 1 0010 1.13 × (1/6) × fSAMPLE 1 0011 1.0 × (1/6) × fSAMPLE 1 0100 0.9 × (1/6) × fSAMPLE 1 0101 0.8 × (1/6) × fSAMPLE 1 0110 0.75 × (1/6) × fSAMPLE Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range 40 Out of tunable filter range 16.67 15.00 13.33 12.00 10.67 10.00 Sampling Frequency (MHz) 65 80 Out of tunable Out of tunable filter filter range range Out of tunable Out of tunable filter filter range range Out of tunable Out of tunable filter filter range range Out of tunable Out of tunable filter filter range range Out of tunable Out of tunable filter filter range range 17.33 Out of tunable filter range 16.25 16.82 125 Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range 17.50 12.89 20.94 11.11 18.06 10.00 16.25 8.89 14.44 Out of tunable filter range Out of tunable filter range Out of tunable filter range 17.78 8.00 13.00 16.00 Out of tunable filter range Out of tunable filter range 9.67 11.56 14.22 10.83 13.33 15.71 8.33 13.54 Out of tunable filter range 16.67 Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range 12.19 15.00 10.83 13.33 9.75 12.00 8.67 10.67 Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range 16.67 8.13 10.00 15.63 Rev. A | Page 24 of 47 Data Sheet AD9674 Table 12. SPI-Selectable Low-Pass Filter Cutoff Options for High Band Mode at Example Sampling Frequencies Address 0x00F[7:3] 0 0000 LPF Cutoff Frequency (MHz) 1.45 × (1/3) × fSAMPLE 0 0001 1.25 × (1/3) × fSAMPLE 0 0010 1.13 × (1/3) × fSAMPLE 0 0011 1.0 × (1/3) × fSAMPLE 0 0100 0.9 × (1/3) × fSAMPLE 0 0101 0.8 × (1/3) × fSAMPLE 0 0110 0.75 × (1/3) × fSAMPLE 0 1000 1.45 × (1/4.5) × fSAMPLE 0 1001 1.25 × (1/4.5) × fSAMPLE 0 1010 1.13 × (1/4.5) × fSAMPLE 0 1011 1.0 × (1/4.5) × fSAMPLE 0 1100 0.9 × (1/4.5) × fSAMPLE 0 1101 0.8 × (1/4.5) × fSAMPLE 0 1110 0.75 × (1/4.5) × fSAMPLE 1 0000 1.45 × (1/6) × fSAMPLE 1 0001 1.25 × (1/6) × fSAMPLE 1 0010 1.13 × (1/6) × fSAMPLE 1 0011 1.0 × (1/6) × fSAMPLE 1 0100 0.9 × (1/6) × fSAMPLE 1 0101 0.8 × (1/6) × fSAMPLE 1 0110 0.75 × (1/6) × fSAMPLE 20.5 Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range 40 19.33 16.67 15.00 Sampling Frequency (MHz) 65 80 Out of tunable Out of tunable filter range filter range 27.08 Out of tunable filter range 24.38 30.00 Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Rev. A | Page 25 of 47 21.67 26.67 19.50 24.00 17.33 21.33 16.25 20.00 20.94 25.78 18.06 22.22 16.25 20.00 14.44 17.78 125 Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range 27.78 Out of tunable filter range Out of tunable filter range Out of tunable filter range 15.71 16.00 25.00 14.22 22.22 Out of tunable filter range 19.33 20.83 13.54 16.67 Out of tunable filter range 26.04 Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range 15.00 23.44 Out of tunable filter range Out of tunable filter range Out of tunable filter range Out of tunable filter range 20.83 18.75 16.67 15.63 AD9674 Data Sheet The back to back Schottky diodes across the secondary transformer limit clock excursions into the AD9674 to approximately 0.8 V p-p differential. These diodes help prevent large voltage swings of the clock from feeding through to other portions of the AD9674, and they preserve the fast rise and fall times of the signal, which is critical to low jitter performance. 3.3V 0.1µF OUT MINI-CIRCUITS® ADT1-1WT, 1:1Z 0.1µF XFMR 50Ω 100Ω CLK– 1 Ratio1 12 9 6 3 0.1µF High-Pass Cutoff Frequency Low-Pass Low-Pass Cutoff = 8 MHz Cutoff = 18 MHz 670 kHz 1.5 MHz 890 kHz 2.0 MHz 1.33 MHz 3.0 MHz 2.67 MHz 6.0 MHz Ratio means low-pass filter cutoff frequency/high-pass filter cutoff frequency. AAF/VGA Test Mode For debugging and testing, there is a bypass switch to view the AAF output on the GPO2 and GPO3 pins. This mode can be enabled via Address 0x109, Bit 4. The differential AAF output allows only one channel to be accessed at a time. The dc output voltage is 1.5 V (or AVDD2/2), and the maximum ac output voltage is 2 V p-p. If a low jitter clock is available, another option is to ac couple a differential positive emitter coupled logic (PECL) signal to the sample clock input pins, as shown in Figure 40. Analog Devices,Inc., offers a family of clock drivers with excellent jitter performance,including the AD9516-0, AD9516-1, AD9516-2, AD9516-3, and AD9516-5 (these five devices are represented by AD9516-x in Figure 40, Figure 41, and Figure 42), as well as the AD9524. 3.3V AD9516-x OR AD9524 VFAC3 0.1µF 0.1µF CLK+ CLK OUT 50Ω* 0.1µF 100Ω PECL DRIVER ADC 0.1µF CLK– CLK 240Ω 11293-029 240Ω *50Ω RESISTOR IS OPTIONAL. Figure 40. Differential PECL Sample Clock A third option is to ac couple a differential LVDS signal to the sample clock input pins, as shown in Figure 41. 3.3V AD9516-x OR AD9524 VFAC3 The output staging block aligns the data, corrects errors, and passes the data to the output buffers. The data is then serialized and aligned to the frame and output clocks. 0.1µF 0.1µF CLK+ CLK OUT 50Ω* 0.1µF Clock Input Considerations LVDS DRIVER 100Ω ADC 0.1µF CLK For optimum performance, clock the AD9674 sample clock inputs (CLK+ and CLK−) with a differential signal. This signal is typically ac-coupled into the CLK+ and CLK− pins via a transformer or capacitors. These pins are biased internally and require no additional bias. SCHOTTKY DIODES: HSM2812 Figure 39. Transformer-Coupled Differential Clock ADC The AD9674 uses a pipelined ADC architecture. The quantized output from each stage is combined into a 14-bit result in the digital correction logic. The pipelined architecture permits the first stage to operate on a new input sample and the remaining stages to operate on the preceding samples. Sampling occurs on the rising edge of the clock. ADC 0.1µF VFAC3 Table 13. High-Pass Filter Cutoff Options Addr. 0x02B[1:0] High-Pass Filter Cutoff 00 (default) 01 10 11 CLK+ 11293-028 Four SPI-programmable settings allow users to vary the highpass filter cutoff frequency as a function of the low-pass cutoff frequency. Two examples are shown in Table 13: an 8 MHz lowpass cutoff frequency and an 18 MHz low-pass cutoff frequency. In both cases, as the ratio decreases, the amount of rejection on the low end frequencies increases. Therefore, making the entire AAF frequency pass band narrow can reduce low frequency noise or maximize the dynamic range for harmonic processing. Figure 39 shows the preferred method for clocking the AD9674. A low jitter clock source, such as the Valpey Fisher oscillator, VFAC3BHL-50 MHz, is converted from a single-ended configuration to a differential configuration using an RF transformer. CLK– *50Ω RESISTOR IS OPTIONAL. 11293-030 Tuning is normally off to avoid changing the capacitor settings during critical times. The tuning circuit is enabled through the SPI. It is disabled automatically after 512 cycles of the ADC sample clock. Initializing the tuning of the filter must be performed after initial power-up and after reprogramming of the filter cutoff scaling or the ADC sample rate. The tuning is initiated using Address 0x02B, Bit 6. Figure 41. Differential LVDS Sample Clock In some applications, it is acceptable to drive the sample clock inputs with a single-ended CMOS signal. In such applications, drive CLK+ directly from a CMOS gate, and bypass the CLK− pin to ground with a 0.1 µF capacitor (see Figure 42). Rev. A | Page 26 of 47 Data Sheet AD9674 130 3.3V RMS CLOCK JITTER REQUIREMENT CLK 50Ω* CMOS DRIVER OPTIONAL 0.1µF 100Ω 120 110 CLK+ ADC CLK 0.1µF CLK– 11293-031 0.1µF *50Ω RESISTOR IS OPTIONAL. 16 BITS 90 14 BITS 80 12 BITS 70 10 BITS 60 Figure 42. Single-Ended 1.8 V CMOS Sample Clock 50 Clock Duty Cycle Considerations 8 BITS 40 Typical high speed ADCs use both clock edges to generate a variety of internal timing signals. As a result, these ADCs can be sensitive to the clock duty cycle. Commonly, a 5% tolerance is required on the clock duty cycle to maintain dynamic performance characteristics. The AD9674 contains a duty cycle stabilizer (DCS) that retimes the nonsampling edge, providing an internal clock signal with a nominal 50% duty cycle. This feature allows a wide range of clock input duty cycles without affecting the performance of the AD9674. When the DCS is on, noise and distortion performance are nearly flat for a wide range of duty cycles. However, some applications may require the DCS function to be off. When the DCS function is off, the dynamic range performance can be affected. The duty cycle stabilizer uses a delay-locked loop (DLL) to create the nonsampling edge. As a result, any changes to the sampling frequency require approximately eight clock cycles to allow the DLL to acquire and lock to the new rate. Clock Jitter Considerations High speed, high resolution ADCs are sensitive to the quality of the clock input. The degradation in SNR at a given input frequency (fA) due only to aperture jitter (tJ) can be calculated as follows: SNR Degradation = 20 × log 10(1/2 × π × fA × tJ) 100 (7) In Equation 7, the rms aperture jitter represents the root mean square of all jitter sources, including the clock input, analog input signal, and ADC aperture jitter (see Figure 43). Treat the clock input as an analog signal when aperture jitter may affect the dynamic range of the AD9674. Separate power supplies for clock drivers from the ADC output driver supplies to avoid modulating the clock signal with digital noise. Low jitter, crystal controlled oscillators, such as the Valpey Fisher VFAC3 series, make the best clock sources. When the clock is generated from another type of source (by gating, dividing, or other methods), retime it by the original clock during the last step. 0.125ps 0.25ps 0.5ps 1.0ps 2.0ps 30 1 10 100 ANALOG INPUT FREQUENCY (MHz) 1000 11293-033 0.1µF SNR (dB) VFAC3 OUT AD9516-x OR AD9524 Figure 43. Ideal SNR vs. Analog Input Frequency and Jitter Power Dissipation and Power-Down Mode The power dissipated by the AD9674 is proportional to its sample rate. The digital power dissipation does not vary significantly because it is determined primarily by the DRVDD supply and the bias current of the LVDS output drivers. The AD9674 features scalable LNA bias currents (see Table 25, Address 0x012). The default LNA bias current settings are midhigh. By asserting the PDWN pin high, the AD9674 is placed into power-down mode. In this state, the device dissipates at a maximum of 30 mW. During power-down, the LVDS output drivers are placed into a high impedance state. The AD9674 returns to normal operating mode when the PDWN pin is pulled low. This pin is only 1.8 V tolerant. To drive the PDWN pin from a 3.3 V logic level, insert a 1 kΩ resistor in series with this pin to limit the current. By asserting the STBY pin high, the AD9674 is placed in standby mode. In this state, the device typically dissipates 630 mW. During standby, the entire device, except the internal references, powers down. The LVDS output drivers are placed into a high impedance state. This mode is well suited for applications that require power savings because it allows the device to be powered down when not in use and then to be quickly powered up. In addition, the time to power up the device is greatly reduced. The AD9674 returns to normal operating mode when the STBY pin is pulled low. This pin is only 1.8 V tolerant. To drive the STBY pin from a 3.3 V logic level, insert a 1 kΩ resistor in series with this pin to limit the current. For more information on how jitter performance relates to ADCs, refer to the AN-501 Application Note and AN-756 Application Note. Rev. A | Page 27 of 47 AD9674 Data Sheet Other power-down options are available when using the SPI port interface. The user can individually power down each channel or place the entire device into standby mode. When fast wake-up times are required, standby mode allows the user to keep the internal PLL powered up. The wake-up time is slightly dependent on gain. To achieve a 2 µs wake-up time when the device is in standby mode, apply 0.8 V to the GAIN± pins. Power and Ground Connection Recommendations When connecting power to the AD9674, use two separate 1.8 V supplies: one for analog (AVDD1) and one for digital (DRVDD). When only one 1.8 V supply is available, route it to the AVDD1 pin first, tap it off, and isolate it with a ferrite bead or a filter choke preceded by decoupling capacitors for the DRVDD pin. The DVDD pin can be tied to the 1.8 V DRVDD supply. When this is done, route the DVDD supply first, tap it off, and isolate it with a ferrite bead or filter choke preceded by decoupling capacitors for the DRVDD pin. It is not recommended to use the same supply for AVDD1, DVDD, and DRVDD to avoid noise issues. For compatibility with the AD9674 or for lower power operation, the DVDD pin can be tied to 1.4 V. To cover both high and low frequencies, use several decoupling capacitors on all supplies. Locate these capacitors close to the point of entry at the PCB level and close to the device, with minimal trace lengths. When using the AD9674, a single PCB ground plane is sufficient. With proper decoupling and smart partitioning of the analog, digital, and clock sections of the PCB, optimum performance is easily achievable. TX_TRIG± DIGITAL POWER ANALOG POWER POWER_STOP (PROFILE SPECIFIC) POWER_START (PROFILE SPECIFIC) POWER_SETUP (SPI SET) 11293-034 In power-down mode, low power dissipation is achieved by shutting down the reference, reference buffer, phase-locked loop (PLL), and biasing networks. The decoupling capacitors on VREF are discharged when entering power-down mode and must be recharged when returning to normal operation. As a result, the wake-up time is related to the time spent in power-down mode: shorter cycles result in proportionally shorter wake-up times. To restore the device to full operation, approximately 375 µs is required when using the recommended 1 µF and 0.1 µF decoupling capacitors on the VREF pin and the 0.01 µF decoupling capacitors on the GAIN± pins. Most of this time is dependent on gain decoupling; higher value decoupling capacitors on the GAIN± pins result in longer wake-up times. Figure 44. Power Sequencing Digital Outputs and Timing The AD9674 differential outputs conform to the ANSI-644 LVDS standard on default power-up. This setting can be changed to a low power, reduced signal option similar to the IEEE 1596.3 standard via the SPI using Address 0x015, Bit 7. This LVDS standard can further reduce the overall power dissipation of the device by approximately 36 mW. The LVDS driver current is derived on chip and sets the output current at each output equal to a nominal 3.5 mA. A 100 Ω differential termination resistor placed at the LVDS receiver inputs results in a nominal 350 mV swing at the receiver. The AD9674 LVDS outputs facilitate interfacing with LVDS receivers in custom ASICs and FPGAs that have LVDS capability for superior switching performance in noisy environments. Single point to point network topologies are recommended with a 100 Ω termination resistor placed as close to the receiver as possible. No far-end receiver termination and poor differential trace routing may result in timing errors. The trace length must be no longer than 24 inches; keep the differential output traces close together and at equal lengths. Figure 45 and Figure 46 show an example of the LVDS output using the ANSI-644 standard (default) data eye and a time interval error (TIE) jitter histogram with trace lengths of less than 24 inches on standard FR-4 material. Figure 47 and Figure 48 show an example of the trace lengths exceeding 24 inches on standard FR-4 material. Notice that the TIE jitter histogram reflects the decrease of the data eye opening as the edge deviates from the ideal position. Therefore, the user must determine whether the waveforms meet the timing budget of the design when the trace lengths exceed 24 inches. Advanced Power Control For an ultrasound system, not all channels are needed during all scanning periods. The POWER_START and POWER_STOP values in the vector profile can be used to delay the channel startup and turn the channel off after a certain number of samples. These counters are relative to TX_TRIG±. The analog circuitry must power up before the digital circuitry. The analog circuitry must power up (POWER_SETUP) before POWER_START is set up in Register 0x112 (see Table 25). Rev. A | Page 28 of 47 Data Sheet AD9674 80 EYE: ALL BITS ULS: 11197/11197 70 TIE JITTER HISTOGRAM (Hits) 300 200 100 0 –100 –200 –300 60 50 40 30 20 –400 –1.5ns –1.0ns –0.5ns 0ns 0.5ns 1.0ns 1.5ns 11293-144 10 Figure 45. Data Eye for LVDS Outputs in ANSI-644 Mode with Trace Lengths of Less Than 24 Inches on Standard FR-4 60 TIE JITTER HISTOGRAM (Hits) –200ps –100ps 0ps 100ps 200ps 300ps Figure 48. TIE Jitter Histogram for LVDS Outputs in ANSI-644 Mode with Trace Lengths of Greater Than 24 Inches on Standard FR-4 Additional SPI options let the user further increase the internal current of all eight outputs to drive longer trace lengths. Even though this produces sharper rise and fall times on the data edges, increasing the internal current is less prone to bit errors and improves frequency distribution. The power dissipation of the DRVDD supply increases when this option is used. 70 50 40 In applications that require increased drive current, Address 0x015 allows the user to adjust the drivers from 2 mA to 3.72 mA. Note that this feature requires Bit 3 of Address 0x015 to be set to 1. The drive current can be adjusted for both ANSI-644 and IEEE 1596.3 (low power) mode. See Table 25 for more details. 30 20 0 –150ps –100ps –50ps 0ps 50ps 100ps 150ps 11293-044 10 Figure 46. TIE Jitter Histogram for LVDS Outputs in ANSI-644 Mode with Trace Lengths of Less Than 24 Inches on Standard FR-4 EYE: ALL BITS The format of the output data is twos complement by default. Table 14 provides an example of the output coding format. To change the output data format to twos complement, see the Memory Map section. Table 14. Digital Output Coding with RF Decimator Bypassed, Digital HPF Bypassed ULS: 11199/11199 400 EYE DIAGRAM VOLTAGE (mV) 0 –300ps 11293-045 EYE DIAGRAM VOLTAGE (mV) 400 Code 16384 8192 8191 0 300 200 100 0 –100 –200 –300 –1.5ns –1.0ns –0.5ns 0ns 0.5ns 1.0ns 1.5ns 11293-145 –400 Figure 47. Data Eye for LVDS Outputs in ANSI-644 Mode with Trace Lengths of Greater Than 24 Inches on Standard FR-4 (VIN+) − (VIN−), Input Span = 2 V p-p (V) +1.00 0.00 −0.000488 −1.00 Digital Output Mode: Twos Complement (D13 to D0) 01 1111 1111 1111 00 0000 0000 0000 11 1111 1111 1111 10 0000 0000 0000 Digital data from each channel is serialized based on the number of lanes that are enabled (see Table 25). The maximum data rate for each serial output lane is 1 Gbps. For one channel per lane with a 14-bit data stream and ADC sample clock of 70 MHz, the output data rate is 980 Mbps (14 bits × 70 MHz = 980 Mbps) with the RF decimator bypassed, and digital HPF bypassed. For higher sample rates, enabling the RF decimator is required. Two output clocks are provided to assist in capturing data from the AD9674. The digital clock outputs (DCO±) are used to clock the output data and are equal to seven times the sampling clock rate in 14-bit mode with the RF decimator bypassed and digital HPF bypassed. Rev. A | Page 29 of 47 AD9674 Data Sheet Data is clocked out of the AD9674 and must be captured on the rising and falling edges of DCO±, which support double data rate (DDR) capturing. The frame clock outputs (FCO±) signal the start of a new output byte and are equal to the sampling clock rate. A 12-, 14-, or 16-bit serial stream can also be initiated from Address 0x021, Bits[1:0]. The user can implement different serial streams and test device compatibility with lower and higher resolution systems using these modes. When using the SPI, all the data outputs can also invert from their nominal state by setting Bit 2 in the output mode register (Address 0x014). This feature is not to be confused with inverting the serial stream to an LSB first mode. In default mode, as shown in Figure 2, the MSB is represented first in the data output serial stream. However, using Address 0x000, Bit 6, this order can be inverted so that the LSB is represented first in the data output serial stream. Digital Output Test Patterns Nine digital output test pattern options can be initiated through the SPI using Address 0x0D. These options are useful when validating receiver capture and timing. See Table 16 for the output test mode bit sequencing options. Some test patterns have two serial sequential words and can be alternated in various ways depending on the test pattern chosen. Note that some patterns may not adhere to the data format select option. In addition, custom user defined test patterns can be assigned in the user pattern registers (Address 0x019 through Address 0x020). All test mode options except the pseudonoise (PN) sequence short and PN sequence long can support 8- to 14-bit word lengths to verify data capture to the receiver. The PN sequence short pattern produces a pseudorandom bit sequence that repeats itself every 29 − 1 bits, or 511 bits. A description of the PN sequence short pattern and how it is generated can be found in Section 5.1 of the ITU-T O.150 (05/96) standard. However, the PN sequence long pattern differs from the ITU-T O.150 (05/96) standard because it begins with a specific value instead of 1s (see Table 15 for the initial values). The PN sequence long pattern produces a pseudorandom bit sequence that repeats itself every 223 − 1 bits, or 8,388,607 bits. A description of the PN sequence long pattern and how it is generated can be found in Section 5.6 of the ITU-T O.150 (05/96) standard. The PN sequence long pattern differs from the standard, however, because the starting value of the pattern is a specific value rather than a value of only 1s and the AD9674 inverts the bit stream (see Table 15 for the initial values). The output sample size depends on the selected bit length. Table 15. PN Sequence Initial Values Sequence PN Sequence Short PN Sequence Long Initial Value 0x092 0x003 First Three Output Samples (MSB First, 16-Bit) 0x496F, 0xC9A9, 0x980C 0xFF5C, 0x0029, 0xB80A See the Memory Map section for information on how to change these additional digital output timing features through the SPI. SDIO Pin The SDIO pin is required to operate the SPI. The pin has an internal 30 kΩ pull-down resistor that pulls this pin low and is only 1.8 V tolerant. If applications require that this pin be driven from a 3.3 V logic level, insert a 1 kΩ resistor in series with this pin to limit the current. SCLK Pin The SCLK pin is required to operate the SPI. The pin has an internal 30 kΩ pull-down resistor that pulls this pin low and is only 1.8 V tolerant. To drive the SCLK pin from a 3.3 V logic level, insert a 1 kΩ resistor in series with this pin to limit the current. CSB Pin The CSB pin is required to operate the SPI. The pin has an internal 70 kΩ pull-up resistor that pulls this pin high and is only 1.8 V tolerant. To drive the CSB pin from a 3.3 V logic level, insert a 1 kΩ resistor in series with this pin to limit the current. RBIAS Pin To set the internal core bias current of the ADC, place a resistor nominally equal to 10.0 kΩ to ground at the RBIAS pin. Using a resistor other than the recommended 10.0 kΩ resistor for RBIAS degrades the performance of the device. Therefore, it is imperative that at least a 1% tolerance on this resistor be used to achieve consistent performance. VREF Pin A stable and accurate 0.5 V voltage reference is built into the AD9674. This voltage reference is gained up internally by a factor of 2, setting VREF to 1.0 V, which results in a full-scale differential input span of 2.0 V p-p for the ADC. VREF is set internally by default, but the VREF pin can be driven externally with a 1.0 V reference to achieve more accuracy. However, the AD9674 does not support ADC full-scale ranges less than 2.0 V p-p. When applying the decoupling capacitors to the VREF pin, use ceramic, low equivalent series resistance (ESR) capacitors. Ensure that these capacitors are close to the reference pin and on the same layer of the PCB as the AD9674. The VREF pin must have both a 0.1 µF capacitor and a 1 µF capacitor that are connected in parallel to the analog ground. These capacitor values are recommended for the ADC to properly settle and acquire the next valid sample. Rev. A | Page 30 of 47 Data Sheet AD9674 Table 16. Flexible Output Test Modes Output Test Mode Bit Sequence 0000 0001 0010 0011 0100 0101 0110 0111 1000 1111 Pattern Name Off (default) Midscale short Positive full-scale short Negative full-scale short Checkerboard PN sequence long PN sequence short One-word/zero-word toggle User input Ramp output Digital Output Word 1 Not applicable 10 0000 0000 0000 11 1111 1111 1111 00 0000 0000 0000 10 1010 1010 1010 Not applicable Not applicable 11 1111 1111 1111 Address 0x019 and Address 0x01A 00 0000 0000 0000 General-Purpose Output Pins The general-purpose output pins, GPO0, GPO1, GPO2 and GPO3, can be used in a system to provide programmable inputs to other chips in the system. The value of each pin is set via Address 0x00E to either Logic 0 or Logic 1 (see Table 25). Chip Address Pins The chip address pins can be used to address individual AD9674 chips among multiple chips in a system. The chip address mode is enabled using Address 0x115, Bit 5 (see Table 25). If the value written to Bits[4:0] matches the value on the chip address bit pins (ADDR4 to ADDR0]), the device is selected and any subsequent SPI writes or reads to addresses indicated as chip registers are written only to that device. If chip address mode is disabled, all addresses can be written to regardless of the value on the address pins. ANALOG TEST SIGNAL GENERATION The AD9674 can generate analog test signals that can be switched to the input of the LNA of each channel to be used for channel gain calibration. The test signal amplitude at the LNA output is dependent on LNA gain, as shown in Table 17. The test signal amplitude at the input to the ADC can be calculated given the LNA gain, attenuator control voltage, and the PGA gain. Table 18 and Table 19 give example calculations. Table 18. Test Signal Fundamental Amplitude at ADC Input, VGAIN = 0 V, PGA Gain = 21 dB Address 0x116, Bits[3:2], Analog Test Tones 00 (default) 01 10 LNA Gain 15.6 dB 80 mV p-p 160 mV p-p 320 mV p-p LNA Gain 17.9 dB 98 mV p-p 196 mV p-p 391 mV p-p LNA Gain 15.6 dB −29 dBFS −23 dBFS −17 dBFS LNA Gain 17.9 dB −28 dBFS −22 dBFS −16 dBFS LNA Gain 21.6 dB −26 dBFS −20 dBFS −14 dBFS Table 19. Test Signal Fundamental Amplitude at ADC Input, VGAIN = 0 V, PGA Gain = 30 dB Address 0x116, Bits [3:2], Analog Test Tones 00 (default) 01 10 Table 17. Test Signal Fundamental Amplitude at LNA Output Address 0x116, Bits[3:2], Analog Test Tones 00 (default) 01 10 Digital Output Word 2 Not applicable Same Same Same 01 0101 0101 0101 Not applicable Not applicable 00 0000 0000 0000 Address 0x01B and Address 0x01C 00 0000 0000 0001 Subject to Resolution Select Not applicable Yes Yes Yes No Yes Yes No No Yes LNA Gain 21.6 dB 119 mV p-p 238 mV p-p 476 mV p-p Rev. A | Page 31 of 47 LNA Gain 15.6 dB −20 dBFS −14 dBFS −8 dBFS LNA Gain 17.9 dB −19 dBFS −13 dBFS −7 dBFS LNA Gain 21.6 dB −17 dBFS −11 dBFS −5 dBFS AD9674 Data Sheet CW DOPPLER OPERATION Each channel of the AD9674 includes an I/Q demodulator. Each demodulator has an individual programmable phase shifter. The I/Q demodulator is ideal for phased array beamforming applications in medical ultrasound. Each channel can be programmed for 16 phase settings/360° (or 22.5°/step), selectable via the SPI port. The device has a RESET± input that is used to synchronize the LO dividers of each channel. If multiple AD9674 devices are used, a common reset across the array ensures a synchronized phase for all channels. Internal to the AD9674, the individual Channel I and Channel Q outputs are current summed. If multiple AD9674 devices are used, the I and Q outputs from each AD9674 can be current summed and converted to a voltage using an external transimpedance amplifier. Quadrature Generation The internal 0° and 90° LO phases are digitally generated by a divide by M logic circuit, where M is 4, 8, or 16. The internal divider is selected via Address 0x02E, Bits[2:1] (see Table 25). The divider is dc-coupled and inherently broadband; the maximum LO frequency is limited only by its switching speed. The duty cycle of the quadrature LO signals must be as close to 50% as possible for the 4LO and 8LO modes. The 16LO mode does not require a 50% duty cycle. Furthermore, the divider is implemented so the multiple LO signal reclocks the final flip flops that generate the internal LO signals and, therefore, minimizes noise introduced by the divide circuitry. For optimum performance, the MLO± input is driven differentially, as on the AD9670 evaluation board. The common-mode voltage on each pin is approximately 1.2 V with the nominal 3 V supply. It is important to ensure that the MLO± source has very low phase noise (jitter), a fast slew rate, and an adequate input level to obtain optimum performance of the CW signal chain. Beamforming applications require a precise channel-to-channel phase relationship for coherence among multiple channels. The RESET± input is provided to synchronize the LO divider circuits in different AD9674 devices when they are used in arrays. The RESET± input is a synchronous edge triggered input that resets the dividers to a known state after power is applied to multiple AD9674 devices. The RESET± signal can be either a continuous signal or a single pulse, and can be either synchronized with the MLO± clock edge (recommended) or it can be asynchronous. If a continuous signal is used for the RESET±, it must be at the LO rate. For a synchronous RESET±, the device can be configured to sample the RESET± signal with either the falling or rising edge of the MLO± clock, which makes it easier to align the RESET± signal with the opposite MLO± clock edge. Register 0x02E is used to configure the RESET± signal behavior. Synchronize the RESET± input to the MLO± input. Accurate channel to channel phase matching can be achieved via a common clock on the RESET± input when using more than one AD9674 device. I/Q Demodulator and Phase Shifter The I/Q demodulators consist of double balanced, harmonic rejection, passive mixers. The RF input signals are converted into currents by transconductance stages that have a maximum differential input signal capability matching the full-scale LNA output. These currents are then presented to the mixers, which convert them to baseband (RF − LO) and 2× RF (RF + LO). The signals are phase shifted according to the codes that are programmed into the SPI latch (see Table 20). The phase shift function is an integral part of the overall circuit. The phase shift listed in Table 20 is defined as being between the baseband I or Q channel outputs. As an example, for a common signal applied to a pair of RF inputs to an AD9674, the baseband outputs are in phase for matching phase codes. However, if the phase code for Channel 1 is 0000 and the phase code for Channel 2 is 0001, Channel 2 leads Channel 1 by 22.5°. Table 20. Phase Select Code for Channel to Channel Phase Shift Φ Shift 0° 22.5° 45° 67.5° 90° 112.5° 135° 157.5° 180° 202.5° 225° 247.5° 270° 292.5° 315° 337.5° Rev. A | Page 32 of 47 I/Q Demodulator Phase (Address 0x02D, Bits[3:0]) 0000 0001 (not valid in 4LO mode) 0010 0011 (not valid in 4LO mode) 0100 0101 (not valid in 4LO mode) 0110 0111 (not valid in 4LO mode) 1000 1001 (not valid in 4LO mode) 1010 1011 (not valid in 4LO mode) 1100 1101 (not valid in 4LO mode) 1110 1111 (not valid in 4LO mode) Data Sheet AD9674 DIGITAL RF DECIMATOR The AD9674 contains digital processing capability. Each channel has two stages of processing available: RF decimator and HPF. For test purposes, the input to the decimator can be a test waveform. Normally, the input to the decimator is the output of the ADC. The output of the decimator and filter is sent to the serializer for output formatting. The maximum data rate of the serializer is 1000 MSPS. Therefore, if the sample rate of the ADC is greater than 65 MSPS, the RF decimator (fixed rate of 2) must be enabled. The ADC resolution is 14 bits. Saturation of the ADC is determined after the dc offset calibration to ensure maximum dynamic range. VECTOR PROFILE To minimize the time needed to reconfigure device settings while operating, the device supports configuration profiles. Up to 32 profiles can be stored in the device. A profile is selected by a 5-bit index. A profile consists of a 64-bit vector, as described in Table 21. Each parameter is concatenated to form the 64-bit profile vector. The profile memory starts at Address 0xF00 and ends at Address 0xFFF. The memory can be written in either stream mode or address selected data mode. However, the memory must be read using stream mode. When writing or reading in stream mode while the SPI configuration is set to MSB first mode (default setting for Register 0x000), the write/read address must refer to the last register address, not the first one. For example, when writing or reading the first profile that spans the address space between 0xF00 and 0xF07, and the SPI port is configured as MSB first, the referenced address must be 0xF07 to allow reading from or writing to the 64-bit profile in MSB mode. For more information about stream mode, see the AN-877 Application Note, Interfacing to High Speed ADCs via SPI. A buffer in the device stores the current profile data. When the profile index is written in Register 0x10C, the selected profile is read from memory and stored in the current profile buffer. The profile memory is read/written in the SPI clock domain. After the SPI writes the profile index value, it takes four SPI clock cycles to read the profile from RAM and store it in the current profile buffer. If the SPI is in LSB mode, these additional SPI clock cycles are provided when the profile index register is written. If the SPI is in MSB mode, an additional byte needs to be read or written to update the profile buffer. Updating the profile memory does not affect the data in the profile buffer. The profile index register must be written to cause a refresh of the current profile data, even if the profile index register is written with the same value. RF DECIMATOR MULTIBAND AAF DECIMATE BY 2 ADC OUTPUT OR TEST WAVEFORM HIGH-PASS FILTER FRAMER SERIALIZER 11293-038 DC OFFSET CALIBRATION Figure 49. Simplified Block Diagram of a Single Channel of RF Decimator Table 21. Profile Definition Field Reserved HPF bypass Bits 32 1 POWER_START 15 Reserved POWER_STOP 1 15 Description Reserved Digital HPF bypass 0 = disable (filter enabled) 1 = enable (filter bypassed) ADC clock cycles counted from TX_TRIG± when the active channels are powered up 0x0000 = 0 clock cycles 0x0001 = 1 clock cycle … 0x7FFF = 32,767 clock cycles Reserved ADC clock cycles counted from TX_TRIG± when the active channels are powered down 0x0000 = 0 clock cycles 0x0001 = 1 clock cycle … 0x7FFF = Continuous run mode Rev. A | Page 33 of 47 AD9674 Data Sheet RF DECIMATOR High-Pass Filter The input to the RF decimator is either the ADC output data or a test waveform, as described in the Digital Test Waveforms section. The test waveforms are enabled per channel using Address 0x11A (see Table 25). A second-order Butterworth, high-pass, infinite impulse response (IIR) filter can be applied after the RF decimator. The IIR filter has a settling time of 2.5 µs and a cutoff frequency of 700 kHz for an encode clock of 50 MHz. Therefore, if the ADC clock is 50 MHz, the first 125 samples (2.5 µs/0.02 µs) must be ignored. The filter can be bypassed or enabled in the vector profile if the filter is enabled using Address 0x113, Bit 5. If the filter is bypassed by setting Address 0x113, Bit 5, to 1, the filter cannot be enabled from the vector profile. DC Offset Calibration DC offset can be reduced through a manual system calibration process. The dc offset of every channel in the system is measured, followed by setting a calibration value in Address 0x110 and Address 0x111. Note that these registers are both chip and local addresses, meaning the registers are accessed using the chip address and device index. The dc offset calibration can be bypassed using Address 0x10F, Bits[2:0]. Multiband AAF and Decimate by 2 The multiband filter is a finite impulse response (FIR) filter. It is programmable with low or high band filtering. The filter requires 11 input samples to populate the filter. The decimation rate is fixed at 2×. Therefore, the decimation frequency is fDEC = fSAMPLE/2. Figure 50 and Figure 51 show the frequency response of the filter, depending on this mode. Figure 50 shows the attenuation amplitude over the Nyquist frequency range. Figure 51 shows the pass band response as nearly flat. DIGITAL TEST WAVEFORMS Digital test waveforms can be used in the digital processing block instead of the ADC output. To enable digital test waveforms, use Address 0x11B. Each channel can be individually enabled in Address 0x11A. Waveform Generator For testing and debugging, a programmable waveform generator can be used in place of ADC data. The waveform generator can vary offset, amplitude, and frequency. The generator uses the ADC sample frequency, fSAMPLE, and ADC full-scale amplitude, AFULL-SCALE, as references. The values are set in Address 0x117, Address 0x118, and Address 0x119 (see Table 25). x = C + A × sin(2 × π × N) 10 0 AMPLITUDE (dBFS) LOW BAND FILTER HIGH BAND FILTER f SAMPLE × n , see Address 0x117 64 (9) A= AFULL−SCALE , see Address 0x118 2x (10) C = AFULL-SCALE × a × 2−(13 − b), see Address 0x119 –30 (11) Channel ID and Ramp Generator –40 –60 0 2 4 6 8 10 12 14 16 18 20 FREQUENCY (MHz) 11293-039 –50 In Channel ID test mode, the output is a concatenated value. Bits[6:0] are a ramp. Bit 7 is reserved as 0. Bits[10:8] are the channel ID such that Channel A is coded as 000 and Channel B is 001. Bits[15:11] compose the chip address. Figure 50. AAF Frequency Response (Frequency Scale Assumes fADC = 2 × fDEC = 40 MHz) 2 1 0 –1 LOW BAND FILTER HIGH BAND FILTER –2 –3 –4 –5 –6 –7 –8 0 2 4 6 8 10 12 14 16 18 FREQUENCY (MHz) 20 11293-040 AMPLITUDE (dBFS) N= –10 –20 (8) Figure 51. AAF Frequency Response Zoomed In (Frequency Scale Assumes fADC = 2 × fDEC = 40 MHz) Rev. A | Page 34 of 47 Data Sheet AD9674 DIGITAL BLOCK POWER SAVING SCHEME To put the digital block back into the idle state (while the rest of the chip is still running) and save power, raise the TX_TRIG signal high or write to the profile index (Register 0x10C, Bits[0:4]). The digital block will also switch to the idle state if the power stop expires when using the advanced power control feature. Figure 52 illustrates the digital block power saving scheme. RUN CHIP DIGITAL DECIMATOR/FILTER IDLE TX_TRIG IS HIGH, PROFILE INDEX WRITE, OR POWER STOP EXPIRES Rev. A | Page 35 of 47 NEGATIVE EDGE TX_TRIG OR SOFTWARE TX_TRIG DIGITAL DECIMATOR/FILTER RUNNING Figure 52. Digital Block Power Saving Scheme 11293-252 To reduce power consumption in the digital block after the ADC, the RF decimator and filter start in an idle state after running the chip (Register 0x008, Bits[2:0] = 000). The digital block only switches to a running state when the negative edge of the TX_TRIG signal pulse is detected, or with a software TX_TRIG signal write (Register 0x10C, Bit 5 = 1). CHIP IN POWER-DOWN, STANDBY, OR CW MODE Data Sheet AD9674 SERIAL PORT INTERFACE (SPI) Table 22. Serial Port Pins Pin SCLK SDIO CSB Function Serial clock. Serial shift clock input. SCLK is used to synchronize serial interface reads and writes. Serial data input/output. Dual-purpose pin that typically serves as an input or an output, depending on the instruction sent and the relative position in the timing frame. Chip select bar (active low). This control gates the read and write cycles. The falling edge of CSB, in conjunction with the rising edge of SCLK, determines the start of the framing sequence. During the instruction phase, a 16-bit instruction is transmitted, followed by one or more data bytes, which is determined by the W0 and W1 bit fields. An example of the serial timing and definitions are shown in Figure 54 and Table 23. During normal operation, CSB signals to the AD9674 that SPI commands must be received and processed. When CSB is brought low, the device processes SCLK and SDIO to execute instructions. Normally, CSB remains low until the communication cycle is complete. However, if connected to a slow device, CSB can be brought high between bytes, allowing older microcontrollers enough time to transfer data into shift registers. CSB can be stalled when transferring one, two, or three bytes of data. When W0 and W1 are set to 11, the device enters streaming mode and continues to process data, either reading or writing, until CSB is taken high to end the communication cycle. This mode allows complete memory transfers without the need for additional instructions. Regardless of the mode, if CSB is taken high in the middle of a byte transfer, the SPI state machine is reset, and the device waits for a new instruction. The SPI port can be configured to operate in different manners. CSB can also be tied low to enable 2-wire mode. When CSB is tied low, SCLK and SDIO are the only pins required for communication. In addition to word length, the instruction phase determines whether the serial frame is a read or write operation, allowing the serial port to be used both to program the chip and to read the contents of the on-chip memory. If the instruction is a readback operation, performing a readback causes the SDIO pin to change direction from an input to an output at the appropriate point in the serial frame. Data can be sent in MSB first mode or LSB first mode. MSB first mode is the default at power-up and can be changed by adjusting the configuration register (Address 0x00). For more information about this and other features, see the AN-877 Application Note, Interfacing to High Speed ADCs via SPI. HARDWARE INTERFACE The pins described in Table 22 constitute the physical interface between the programming device and the serial port of the AD9674. The SCLK and CSB pins function as inputs when using the SPI. The SDIO pin is bidirectional, functioning as an input during write phases and as an output during readback. If multiple SDIO pins share a common connection, ensure that proper VOH levels are met. Figure 53 shows the number of SDIO pins that can be connected together and the resulting VOH levels, assuming the same load for each AD9674. 1.800 1.795 1.790 1.785 1.780 1.775 1.770 1.765 1.760 1.755 1.750 1.745 1.740 1.735 1.730 1.725 1.720 1.715 0 10 20 30 40 50 60 70 80 90 NUMBER OF SDIO PINS CONNECTED TOGETHER 100 11293-041 The SCLK, SDIO, and CSB pins define the SPI (see Table 22). The SCLK (serial clock) pin synchronizes the read and write data presented to the device. The SDIO pin is a dual-purpose pin that allows data to be sent to and read from the internal memory map registers of the device. The CSB pin is an active low control that enables or disables the read and write cycles. Although the device is synchronized during power-up, caution must be exercised when using 2-wire mode to ensure that the serial port remains synchronized with the CSB line. When operating in 2-wire mode, it is recommended that a 1-, 2-, or 3-byte transfer be used exclusively. Without an active CSB line, streaming mode can be entered but not exited. VOH (V) The AD9674 SPI allows the user to configure the signal chain for specific functions or operations through the structured register space provided inside the chip. The SPI offers the user added flexibility and customization, depending on the application. Addresses are accessed via the serial port and can be written to or read from via the port. Memory is organized into bytes that can be further divided into fields, as documented in the Memory Map section. For detailed operational information, see the AN-877 Application Note, Interfacing to High Speed ADCs via SPI. Figure 53. SDIO Pin Loading This interface is flexible enough to be controlled either by serial programmable read-only memories (PROMs) or by PIC microcontrollers, which provide the user with an alternative to a full SPI controller for programming the device (see the AN-812 Application Note, Microcontroller-Based Serial Port Interface (SPI®) Boot Circuit). Rev. A | Page 36 of 47 Data Sheet AD9674 tDS tS tHIGH tCLK tH tDH tLOW CSB DON’T CARE SDIO DON’T CARE DON’T CARE R/W W1 W0 A12 A11 A10 A9 A8 A7 D5 D4 D3 D2 D1 D0 DON’T CARE 11293-042 SCLK Figure 54. Serial Timing Details Table 23. Serial Timing Definitions Parameter tDS tDH tCLK tS tH tHIGH tLOW tEN_SDIO Timing (ns min) 12.5 5 40 5 2 16 16 15 tDIS_SDIO 15 Description Setup time between the data and the rising edge of SCLK Hold time between the data and the rising edge of SCLK Period of the clock Setup time between CSB and SCLK Hold time between CSB and SCLK Minimum period that SCLK must be in a logic high state Minimum period that SCLK must be in a logic low state Minimum time for the SDIO pin to switch from an input to an output relative to the SCLK falling edge (not shown in Figure 54) Minimum time for the SDIO pin to switch from an output to an input relative to the SCLK rising edge (not shown in Figure 54) Rev. A | Page 37 of 47 AD9674 Data Sheet MEMORY MAP READING THE MEMORY MAP TABLE RESERVED LOCATIONS Each row in the memory map register table has eight bit locations. The memory map is roughly divided into two sections: the chip configuration register map (Address 0x000 to Address 0x1A1) and the profile register map (Address 0xF00 to Address 0xFFF). Registers that are designated as local registers use the device index in Address 0x004 and Address 0x005 to determine to which channels of a device the command is applied. Registers that are designated as chip registers use the chip address mode in Address 0x115 to determine whether the device is to be updated by writing to the chip register. Do not write to undefined memory locations except when writing the default values suggested in this data sheet. Addresses that have values marked as 0 must be considered reserved and have a 0 written into their registers during power-up. The address hex column of Table 25 indicates the register address. The default value is shown in the default value column. The Bit 7 (MSB) column is the start of the default hexadecimal value given. For example, Address 0x009, the global clock register, has a default value of 0x01, meaning that Bit 7 = 0, Bit 6 = 0, Bit 5 = 0, Bit 4 = 0, Bit 3 = 0, Bit 2 = 0, Bit 1 = 0, and Bit 0 = 1, or 0000 0001 in binary. This setting is the default for the duty cycle stabilizer in the on condition. “Bit is set” is synonymous with “bit is set to Logic 1” or “writing Logic 1 for the bit.” Similarly, “bit is cleared” is synonymous with “bit is set to Logic 0” or “writing Logic 0 for the bit.” For more information about the SPI memory map and other functions, see the AN-877 Application Note, Interfacing to High Speed ADCs via SPI. DEFAULT VALUES After a reset, critical registers are automatically loaded with default values. These values are indicated in Table 25, where an X refers to an undefined feature. LOGIC LEVELS RECOMMENDED START-UP SEQUENCE To save system power during programming, the AD9674 powers up in power-down mode. To start the device up and initialize the data interface, the SPI commands listed in Table 24 are recommended. At a minimum, the profile memory for an index of 0 must be written (Address 0xF00 to Address 0xF03). If additional profiles and coefficient memory are required, these can be written after Profile Memory 0. Rev. A | Page 38 of 47 Data Sheet AD9674 Table 24. SPI Write Start-Up Sequence Example Address 0x000 0x002 0x0FF 0x004 0x005 0x113 0x011 0xF00 0xF01 0xF02 0xF03 0x10C 1 0x014 0x008 0x021 0x199 0x19B 0x188 0x18B 0x18C 0x182 0x10C 3 0x00F 0x02B Value 0x3C 0x0X (default) 0x01 0x0F 0x3F 0x00 0x06 (default) 0xFF 0x7F 0x00 0x80 0x00 (default) 0x00 0x00 0x05 0x80 0x50 0x01 0x27 0x72 0x82 0x20 0x18 (default) 0x40 Description Initiates SPI reset Sets speed mode to 40 MHz Enables speed mode change (required after Register 0x002 writes) Sets local registers to all channels Sets local registers to all channels Bypasses RF decimator; enable digital HPF Sets LNA gain = 21.6 dB, VGA gain = external, and PGA gain = 24 dB Continuous run mode enabled; do not power down channels (POWER_STOP LSB) Continuous run mode enabled; do not power down channels (POWER_STOP MSB) Powers up all channels 0 clock cycles after TX_TRIG± (POWER_STOP LSB) Digital high-pass bypassed (POWER_STOP MSB) Sets the profile index (required after profile memory writes) Sets output data format TGC run mode 2 14 bits, 8 lanes, frame clock output (FCO) covers entire frame Enables automatic clocks per sample calculation Serial format Enables start code Sets start code MSB Sets start code LSB Autoconfigures PLL Sets SPI TX_TRIG and profile index Sets low-pass filter cutoff frequency and bandwidth mode Sets analog LPF and HPF to defaults, tune filters 4 Setting the profile index requires an additional SPI write in SPI MSB mode before the chip runs to complete the current profile buffer update. Running the chip from full power-down mode requires 375 µs wake-up time, as listed in Table 3. Software TX_TRIG switches the demodulator/decimator digital block to a running state. The software TX_TRIG signal may not be needed if a hardware TX_TRIG signal is used to run the digital block. 4 Tuning the filters requires 512 ADC clock cycles. 1 2 3 Rev. A | Page 39 of 47 Data Sheet AD9674 Table 25. Memory Map Registers Addr. (Hex) Register Name Bit 7 (MSB) Chip Configuration Registers 0x000 CHIP_PORT_ 0 CONFIG Bit 6 Bit 5 Bit 4 Bit 3 Bit 2 Bit 1 LSB first: 0 = off (default), 1 = on SPI reset: 0 = off (default), 1 = on 1 1 SPI reset: 0 = off (default), 1 = on LSB first: 0 0 = off (default), 1 = on f0x001 CHIP_ID Bit 0 (LSB) Chip ID, Bits[7:0] (AD9674 = 0xA8), default 0x002 CHIP_GRADE X X 0x0FF DEVICE_UPDATE X Default Value Comments 0x18 0xA8 Mirror nibbles so LSB first or MSB Mode I is set correctly regardless of shift mode. SPI reset reverts all registers (including LVDS registers), except Register 0x000, to their default values, and Register 0x000, Bit 2 and Bit 5 bits automatically clear. Default is unique chip ID, different for each device; read only register. Speed mode used to differentiate ADC speed power modes (must update Register 0x0FF after for the speed mode changes to take effect). X X X 0x0X X Speed mode, Bits[5:4] X (identify device variants of chip ID): 00 = Mode I (40 MSPS) (default), 01 = Mode II (65 MSPS), 10 = Mode III (80 MSPS), 11 = Mode III (125 MSPS) X X X X X X 0x00 0x004 DEVICE_INDEX_2 X X X X X Data Channel E: 0 = off, 1 = on (default) Data Channel A: 0 = off, 1 = on (default) 0x3F Bits are set to determine which on-chip channel receives the next write command. LNA input impedance: 0 = 6 kΩ (default), 1 = 3 kΩ Clock Channel FCO±: 0 = off, 1 = on (default) 0 Data Channel F: 0 = off, 1 = on (default) Data Channel B: 0 = off, 1 = on (default) 0x008 GLOBAL_MODES X Clock Channel DCO±: 0 = off, 1 = on (default) X Data Channel G: 0 = off, 1 = on (default) Data Channel C: 0 = off, 1 = on (default) 0x0F 0x005 DEVICE_INDEX_1 X Data Channel H: 0 = off, 1 = on (default) Data Channel D: 0 = off, 1 = on (default) 0 Determines generic modes of chip operation (global). 0x009 GLOBAL_CLOCK X X X X X 0x00A PLL_STATUS PLL lock status: 0 = not locked (default), 1 = locked X X X X Internal power-down mode: 0x01 000 = chip run (TGC mode), 001 = full power-down (default), 010 = standby, 011 = reset all LVDS registers, 100 = CW Doppler mode (TGC is powered down) X X DCS: 0x01 0 = off, 1 = on (default) X X X 0x00 Rev. A | Page 40 of 47 A write to Register 0x0FF (value does not matter) resets all default register values (analog and ADC registers only; not JESD204B ones and not Register 0x00 or Register 0x02, Bits[5:4]) if Register 0x02 has been previously written since the last reset/load of defaults. Bits are set to determine which on-chip channel receives the next write command. Turns the internal DCS on and off (global). Monitor PLL lock status (read only, global). Data Sheet Addr. (Hex) Register Name 0x00D TEST_IO 0x00E GPO AD9674 Bit 7 (MSB) Bit 6 User test mode: 0 = X continuous, repeat user patterns (1, 2, 3, 4, 1, 2, 3, 4, …) (default), 1 = single clock cycle user patterns, then zeros (1, 2, 3, 4, 0, 0, …) Bit 5 Reset PN long gen 0 = on, PN long running (default), 1 = off, PN long held in reset Bit 4 Reset PN short gen: 0 = on, PN short running (default), 1 = off, PN short held in reset X X X 0x00F FLEX_CHANNEL _INPUT 0x010 FLEX_OFFSET 0x011 FLEX_GAIN X 0x012 BIAS_CURRENT X 0x013 RESERVED_13 0x014 OUTPUT_MODE 0 X X Filter cutoff frequency control: 00000 = 1.45 × (1/3) × fSAMPLE, 00001 = 1.25 × (1/3) × fSAMPLE, 00010 = 1.13 × (1/3) × fSAMPLE, 00011 = 1.0 × (1/3) × fSAMPLE (default), 00100 = 0.9 × (1/3) × fSAMPLE, 00101 = 0.8 × (1/3) × fSAMPLE, 00110 = 0.75 × (1/3) × fSAMPLE, 00111 = reserved, 01000 = 1.45 × (1/4.5) × fSAMPLE, 01001 = 1.25 × (1/4.5) × fSAMPLE, 01010 = 1.13 × (1/4.5) × fSAMPLE, 01011 = 1.0 × (1/4.5) × fSAMPLE, 01100 = 0.9 × (1/4.5) × fSAMPLE, 01101 = 0.8 × (1/4.5) × fSAMPLE, 01110 = 0.75 × (1/4.5) × fSAMPLE, 01111 = reserved, 10000 = 1.45 × (1/6) × fSAMPLE, 10001 = 1.25 × (1/6) × fSAMPLE, 10010 = 1.13 × (1/6) × fSAMPLE, 10011 = 1.0 × (1/6) × fSAMPLE, 10100 = 0.9 × (1/6) × fSAMPLE, 10101 = 0.8 × (1/6) × fSAMPLE, 10110 = 0.75 × (1/6) × fSAMPLE, 1 0111 = reserved X 1 0 Digital VGA gain control: 0000 = GAIN± pins enabled (default), 0001 = 0.0 dB (maximum gain, GAIN± pins disabled), 0010 = −3.5 dB, 0011 = −7.0 dB, …., 1110 = 45 dB 1111 = 45 dB X X X 0x015 OUTPUT_ADJUST LVDS output standard: 0 = ANSI-644 (default), 1 = IEEE 1596.3 (low power) Bit 3 Bit 2 Bit 1 Bit 0 (LSB) Output test mode: 0000 = off (default), 0001 = midscale short, 0010 = positive full-scale short, 0011 = negative full-scale short, 0100 = checkerboard output, 0101 = PN sequence long, 0110 = PN sequence short, 0111 = one-word/zero-word toggle, 1000 = user input, 1001:1110 = reserved, 1111 = ramp output (see Table 16) GPO3 GPO2 GPO1 GPO0 output: output: output: output: 0 = low 0 = low 0 = low 0 = low (default); (default); (default): (default); 1 = high 1 = high 1 = high 1 = high Band X mode: 0 = low (default, 8 MHz to 18 MHz), 1 = high (13.5 MHz to 30 MHz) 0 0 PGA gain: 00 = 21 dB, 01 = 24 dB (default), 10 = 27 dB, 11 = 30 dB 1 0 X 0 X 0 0 Output data X enable: 0 = enable (default), 1 = disable 1 1 0 LVDS drive strength enable: 0 = disable (default), 1 = enable Rev. A | Page 41 of 47 PGA bias: 0 = 100% (default), 1 = 60% 0 Default Value Comments 0x00 When register is set, the test data is placed on the output pins in place of normal data (local). 0x00 Values placed on GPOx pins (global). X 0x18 Antialiasing filter cutoff (global). 0 LNA gain: 00 = 15.6 dB, 01 = 17.9 dB, 10 = 21.6 dB (default) 0x20 0x06 Reserved. LNA and PGA gain adjustment (global). 0x09 LNA bias current adjustment (global). 0x00 0x01 Reserved. Data output modes (local). 0x61 Data output levels (global). LNA bias: 00 = high, 01 = midhigh (default), 10 = midlow, 11 = low 0 0 0 Output Output data format: data 00 = offset binary, invert: 01 = twos complement 0 = disable (default), (default), 10 = gray code, 1 = enable 11 = reserved LVDS drive current: 000 = 3.72 mA, 001 = 3.5 mA (default), 010 = 3.30 mA, 011 = 2.96 mA, 100 = 2.82 mA, 101 = 2.57 mA, 110 = 2.27 mA, 111 = 2.0 mA (reduced range) AD9674 Addr. (Hex) Register Name 0x016 FLEX_OUTPUT_ PHASE Data Sheet Bit 4 DCO signal invert: 0 = disable (default), 1 = enable Bit 3 X Bit 2 X Bit 1 Bit 0 (LSB) DCO signal phase adjust with respect to DOUT: 00 = +90° (default), 01 = 0°, 10 = 0°, 11 = −90° DCO signal clock delay: 00000: 100 ps (default), 00001 = 200 ps, 00010 = 300 ps, …, 11101 = 3.0 ns, 11110 = 3.1 ns, 11111 = 3.2 ns 1 0 0 B2 B1 B0 Default Value Comments 0x00 DCO signal inversion and coarse phase adjustment (global). Bit 7 (MSB) X Bit 6 X Bit 5 0 0x017 FLEX_OUTPUT_ DELAY DCO signal delay enable: 0 = disable (default), 1 = enable X X 0x018 RESERVED_018 0x019 USER_ PATT1_LSB 0x01A USER_ PATT1_MSB 0x01B USER_ PATT2_LSB 0x01C USER_ PATT2_MSB 0x01D USER_ PATT3_LSB 0x01E USER_ PATT3_MSB 0x01F USER_ PATT4_LSB 0x020 USER_ PATT4_MSB 0x021 FLEX_ SERIAL_CTRL X B7 X B6 X B5 X B4 X B3 B15 B14 B13 B12 B11 B10 B9 B8 0x00 B7 B6 B5 B4 B3 B2 B1 B0 0x00 B15 B14 B13 B12 B11 B10 B9 B8 0x00 B7 B6 B5 B4 B3 B2 B1 B0 0x00 B15 B14 B13 B12 B11 B10 B9 B8 0x00 B7 B6 B5 B4 B3 B2 B1 B0 0x00 B15 B14 B13 B12 B11 B10 B9 B8 0x00 0 FCO signal invert: 0 = not inverted (default), 1 = inverted Lane low rate: 0 = normal (default), 1 = low sample frequency (<32 MHz) X Output word length: 00 = 12 bits (default), 01 = 14 bits, 10 = 16 bits, 11 = reserved 0x00 0x022 SERIAL_ CH_STAT X X Lane mode: 00 = 1-channel/lane (8 lanes) (default), 01 = 2-channel/lane (4 lanes), 10 = 4-channel/lane (2 lanes), 11 = 8-channel/lane (1 lane) X X X X 0x02B FLEX_FILTER X 0x02C LNA_TERM X Enables X automatic low-pass tuning: 1 = on (self clearing) X X X Bypass X analog HPF: 0 = off (default), 1 = on X X Rev. A | Page 42 of 47 X X 0x00 DCO signal delay (global). 0x04 0x00 Reserved (global). User-Defined Pattern 1, LSB (global). User-Defined Pattern 1, MSB (global). User-Defined Pattern 2, LSB (global). User-Defined Pattern 2, MSB (global). User-Defined Pattern 3, LSB (global). User-Defined Pattern 3, MSB (global). User-Defined Pattern 4, LSB (global). User-Defined Pattern 4, MSB (global). LVDS control (global). Channel 0x00 power-down: 1 = on, 0 = off (default) Analog high-pass filter 0x00 cutoff: 00 = fLP/12 (default), 01 = fLP/9, 10 = fLP/6, 11 = fLP/3 LO-x, LOSW-x 0x00 connection: 00 = RFB1 (default), 01 = (RFB1||RFB2), 10 = RFB2, 11 = ∞ Used to power down individual channels (local). Filter cutoff (global); (fLP = low-pass filter cutoff frequency in MSPS). LNA active termination/ input impedance (global). Data Sheet Addr. (Hex) Register Name 0x02D CW_ENABLE_ PHASE 0x02E CW_LO_MODE 0x02F CW_OUTPUT 0x102 0x103 0x104 0x105 0x106 0x107 RESERVED_102 RESERVED_103 RESERVED_104 RESERVED_105 RESERVED_106 RESERVED_107 AD9674 Bit 7 (MSB) X Bit 6 X Bit 5 X Bit 4 Bit 3 Bit 2 Bit 1 Bit 0 (LSB) CW Doppler I/Q demodulator phase: channel 0000 = 0° (default), enable: 0001 = 22.5° (not valid for 4LO mode), 0 = off 0010 = 45°, (default), 0011 = 67.5° (not valid for 4LO mode), 0100 = 90°, 1 = on 0101 = 112.5° (not valid for 4LO mode), 0110 = 135°, 0111 = 157.5° (not valid for 4LO mode), 1000 = 180°, 1001 = 202.5° (not valid for 4LO mode), 1010 = 225°, 1011 = 247.5° (not valid for 4LO mode), 1100 = 270°, 1101 = 292.5° (not valid for 4LO mode), 1110 = 315°, 1111 = 337.5° (not valid for 4LO mode) Partially enable RESET± with SynchroRESET± MLO± and LO mode LVDS during CW MLO± clock nous signal RESET± 00X = 4LO, third to fifth odd 0: LVDS link edge: RESET± polarity: buffer enable harmonic rejection (default) disabled during CW 0 = synchro- sampling 0 = active (in all modes nous MLO± high 010 = 8LO, third to fifth odd (default), except clock edge: (default), harmonic rejection 1: LVDS link partially (default), CW mode): 011 = 8LO, third to 13th odd enabled during CW, 1 = asynchro- 0 = falling 1 = active 0 = power (default), PLL, FCO, and DCO down harmonic rejection nous low are enabled, while (default), 100 = 16LO, third to fifth odd 1 = rising LVDS data drivers harmonic rejection 1 = enable are disabled 101 = 16LO, third to 13th odd (switching activity harmonic rejection can degrade CW 11X = reserved performance) CW output dc bias 0 0 0 0 0 0 0 voltage: 0 = bypass, 1 = enable (default) 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 1 1 1 1 1 1 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 9 0 0 0 0 0 0 X X 0x108 RESERVED_108 0x109 VGA_TEST 0 X 0 X 0 X 0 VGA/AAF test mode enable: 0 = off (default), 1 = on 0x10C PROFILE_INDEX X X 0x10D RESERVED_10D 0x10E RESERVED_10E 0x10F DIG_OFFSET_CAL 1 1 0 1 1 0 Manual TX_TRIG: 0 = off, use pin (default), 1 = on, auto generate TX_TRIG (self clears) 1 1 1 1 0 0 0 X 0 0 0 VGA/AAF output test mode: 000 = Channel A (default), 001 = Channel B, 010 = Channel C, 011 = Channel D, 100 = Channel E, 101 = Channel F, 110 = Channel G, 111 = Channel H Profile index, Bits[4:0] 1 1 1 1 1 1 1 1 Digital offset Digital offset calibration: calibration 000 = disable correction, reset status: correction value (default), 0 = not 001 = average 210 samples, complete 010 = average 211 samples, (default), …, 1 = complete 111 = average 216 samples Rev. A | Page 43 of 47 Default Value Comments 0x00 Phase of demodulators (local, chip). 0x00 CW mode functions (global). 0x80 CW dc voltage output control (global). 0x00 0x00 0x3F 0x00 0x00 Read only 0x00 0x00 Reserved. Reserved. Reserved. Reserved. Reserved. Reserved. 0x00 Index for profile memory; selects active profile (global). 0xFF 0xFF 0x00 Reserved. Reserved. Controls digital offset calibration enable and number of samples used (global). Reserved. VGA/AAF test mode, enables AAF output to GPO2 and GPO3 pins (global). AD9674 Addr. (Hex) Register Name 0x110 DIG_OFFSET_ CORR1 0x111 DIG_OFFSET_ CORR2 Data Sheet Bit 7 (MSB) D7 D15 Bit 6 D6 Bit 5 D5 Bit 3 D3 Bit 2 D2 Bit 1 D1 Bit 0 (LSB) D0 D14 D13 D12 D11 D10 D9 D8 Digital offset calibration (read back if auto calibration enabled with Register 0x10F. Otherwise, force correction value.) Offset correction = [D15:D0] × AFULL-SCALE/216, 0111 1111 1111 1111 (215 − 1) = +1/2 × AFULL-SCALE − 1/216 × AFULL-SCALE, 0111 1111 1111 1110 (215 − 2) = +1/2 × AFULL-SCALE − 2/216 × AFULL-SCALE, …, 0000 0000 0000 0001 (+1) = 1/216 × AFULL-SCALE, 0000 0000 0000 0000 = no correction (default), 1111 1111 1111 1111 (−1) = −1/216 × AFULL-SCALE, …, 1000 0000 0000 0000 (−215) = −1/2 AFULL-SCALE X X Power-up setup time (POWER_SETUP): 0 0000 = 0, 0 0001 = 1 × 40/fSAMPLE, 0 0010 = 2 × 40/fSAMPLE (default), 0 0011 = 3 × 40/fSAMPLE, …, 1 1111 = 31 × 40/fSAMPLE X Digital 0 Decimator and X X high-pass filter enable: filter: 00 = RF 2× decimator 0 = enable bypassed (default), (default), 01 = RF 2× decimator enabled and low 1 = bypass band filter, 1X = RF 2× decimator enabled and high band filter, 0x112 POWER_ MASK_CONFIG X 0x113 DIG_CONFIG X 0x115 CHIP_ADDR_EN X X 0x116 ANALOG_ TEST_TONE X X Chip address mode: 0 = disable (default), 1 = enable X 0x117 DIG_SINE_ TEST_FREQ X X X 0x118 DIG_SINE_ TEST_AMP X X X 0x119 DIG_SINE_ TEST_OFFSET 0x11A TEST_MODE_ CH_ENABLE Bit 4 D4 Channel H enable: 0 = off (default), 1 = on Chip address qualifier: 0 0000 (default), if read, returns state of Pin ADDR4 to Pin ADDR0 X X Analog test signal amplitude (see Table 17 to Table 19) Analog test signal frequency: 00 = fSAMPLE/4 (default), 01 = fSAMPLE/8, 10 = fSAMPLE/16, 11 = fSAMPLE/32 Digital test tone frequency: 0 0000 = 1 × fSAMPLE/64, 0 0001 = 2 × fSAMPLE/64, …, 1 1111 = 32 × fSAMPLE/64 Digital test tone amplitude: 0000 = AFULL-SCALE (default), 0001 = AFULL-SCALE/2, 0010 = AFULL-SCALE/22, …, 1111 = AFULL-SCALE/215 Offset exponent (b): 000 = 0 (default), 001 = 1, …, 111 = 7 Offset multiplier (a): 0 1111 = 15, 0 1110 = 14, …, 0 0000 = 0 (default), 1 1111 = −1, …, 1 0000 = −16 Offset = AFULL-SCALE × a × 2− (13 − b), offset range is ~0.5 dB, maximum positive offset = 15 × 2− (13 − 7) = 0.25 × AFULL-SCALE, maximum negative offset = –16 × 2− (13 − 7) ≈ −0.25 × AFULL-SCALE Channel G Channel F Channel E Channel D Channel C enable: enable: enable: enable: enable: 0 = off 0 = off 0 = off 0 = off 0 = off (default), (default), (default), (default), (default), 1 = on 1 = on 1 = on 1 = on 1 = on Rev. A | Page 44 of 47 Channel B enable: 0 = off (default), 1 = on Channel A enable: 0 = off (default), 1 = on Default Value Comments 0x00 Offset correction LSB (local, chip). 0x00 Offset correction MSB (local, chip). 0x02 Power setup time used to set the power-up time (global). 0x00 Enables stages of the digital processing (global). 0x00 Chip address mode enables the addressing of specific devices if the value of Bits[4:0] qualifier equals the state on the ADDR4 to ADDR0 pins (global). Analog test tone amplitude and frequency (global). 0x00 0x00 Digital sine test tone frequency (global). 0x00 Digital sine test tone amplitude (global). 0x00 Digital sine test tone offset (global). 0x00 Enables channels for test mode (global). Data Sheet AD9674 Addr. (Hex) Register Name 0x11B TEST_MODE_ CONFIG Bit 7 (MSB) X Bit 6 X Bit 5 X Bit 4 X 0x11C 0x11D 0x11E 0x11F 0x120 0 0 0 0 0 0 0 0 0 CW I/Q output swap: 0 = disable (default), 1 = enable 0 0 0 0 LNA offset cancellation: 0 = enable (default), 1 = disable 0 0 0 0 0 0 0 0 LNA offset cancellation transconductance: 00 = 0.5 mS (default), 01 = 1.0 mS, 10 = 1.5 mS, 11 = 2.0 mS 0 0 0 0 0 0 0 0 0 0 0 0 Bit 1 Bit 0 (LSB) Test mode selection: 000 = disable test modes (default), 001 = enable digital sine test mode, 010 = reserved 011 = enable channel ID test mode (16-bit data = digital ramp (7 bits) + reserved bit (0) + Channel ID (3 bits) + chip address (5 bits), 100 = enable analog test tone, 101 = reserved, 110 = reserved, 111 = reserved 0 0 0 0 0 0 0 0 0 0 0 0 CW analog test tone 0 override for Register 0x116 < Bits[1:0] > 00 = disable override (default) 01 = set analog test tone frequency to fLO 1X = set analog test tone frequency to dc 1 1 1 0 0 0 0 1 0 0 0 0 0 0 0 0 1 1 0 0 0 0 0 0 0 0 1 0 0 1 0 1 0 0 1 0 1 0 0 0 0 B14 B6 0 0 0 0 0 0 0 0 0 0 B13 B5 0 0 0 0 0 0 0 0 0 0 B12 B4 1 0 1 0 1 0 1 0 0 0 B11 B3 0 0 1 0 1 0 1 0 0 0 B10 B2 0 0 0 0 1 0 0 0 0 0 B9 B1 0 0 0 0 0 0 0 0 RESERVED_11C RESERVED_11D RESERVED_11E RESERVED_11F CW_TEST_TONE 0x180 RESERVED_180 0x181 RESERVED_181 0x182 PLL_STARTUP 0x183 0x184 0x186 0x187 0x188 RESERVED_183 RESERVED_184 RESERVED_186 RESERVED_187 START_CODE_EN 1 0 PLL auto configure: 0 = disable (default), 1 = enable 0 0 1 0 0 0x189 0x18A 0x18B 0x18C 0x190 0x191 0x192 0x193 0x194 0x195 0x196 0x197 RESERVED_189 RESERVED_18A START_CODE_MSB START_CODE_LSB RESERVED_190 RESERVED_191 RESERVED_192 RESERVED_193 RESERVED_194 RESERVED_195 RESERVED_196 RESERVED_197 0 0 B15 B7 0 0 0 0 0 0 0 0 Bit 3 X Rev. A | Page 45 of 47 Bit 2 1 0 0 0 Start code identifier: 0 = disable, 1 = enable (default) 0 0 B8 B0 0 0 0 0 0 0 0 0 Default Value Comments 0x00 Enables digital test modes (global). 0x00 0x00 0x00 0x00 0x00 Reserved. Reserved. Reserved. Reserved. Sets the frequency of the analog test tone to fLO in CW Doppler mode; enables I/Q output swap; LNA offset cancellation control (global). 0x87 0x00 0x02 Reserved. Reserved. PLL control (global). 0x07 0x00 0xAE 0x20 0x01 Reserved. Reserved. Reserved. Reserved. Enables start code identifier (global). 0x00 0x00 0x27 0x72 0x10 0x00 0x18 0x00 0x1C 0x00 0x18 0x00 Reserved. Reserved. Start code MSB (global). Start code LSB (global). Reserved. Reserved. Reserved. Reserved. Reserved. Reserved. Reserved. Reserved. AD9674 Data Sheet Addr. (Hex) Register Name Bit 7 (MSB) 0x198 CLOCK_DOUBLING 0 Bit 6 0 Bit 5 0 Bit 4 0 Bit 3 0x199 SAMPLE_CLOCK_ Enables clocks per sample auto COUNTER calculation: 0 = off (default), 1 = on 0x19A DATA_OUTPUT_ X INVERT 0 0 0 0 X X X X 0x19B SERIAL_FORMAT X Enables FCO for start code sample: 0 = disable, 1 = enable (default) Enables FCO continuously: 0 = only during burst, 1 = continuous (default) 0x19C RESERVED_19C 0x19D RESERVED_19D 0x19E RESERVED_19E 0x19F RESERVED_19F 0x1A0 RESERVED_1A0 0x1A1 RESERVED_1A1 Profile Memory Registers 0xF00 Profile memory to 0xFFF 0 0 0 0 0 0 0 0 0 0 0 0 Enables FCO for extra sample at end of burst: 0 = disable, 1 = enable (default) 0 0 0 0 0 0 1 0 1 0 0 0 Bit 2 Bit 1 Bit 0 (LSB) DCO frequency doubling/divider: 0000 = 1 (default), 0001 = 2, 0010 = 4, 0011 = 8, 0100 = 16, 0101 = 32, 0110 = 64, 0111 = 128, 1000 = 1/256, 1001 = 1/128, 1010 = 1/64, 1011 = 1/32, 1100 = 1/16, 1101 = 1/8, 1110 = 1/4, 1111 = 1/2 0 0 0 X 0 0 0 0 0 0 32 × 64 bits For more information about the SPI memory map and other functions, consult the AN-877 Application Note, Interfacing to High Speed ADCs via SPI. Transfer (Register 0x0FF) All registers except Register 0x002 update as soon as they are written. Writing to Register 0x0FF (the value written is don’t care) initializes and updates the speed mode (Address 0x002) and resets all other registers to their default values (analog and ADC registers only, and not JESD204B registers, Register 0x000 or Register 0x002). 0x00 Inverts data 0x00 output: 0= noninverted (default), 1 = inverted FCO signal rotate: 0x70 0000 = FCO signal aligned with DOUT signal, 0001 = FCO 1 bit before DOUT, 0010 = FCO 2 bits before DOUT, …, 1101 = FCO 3 bits after DOUT, 1110 = FCO 2 bits after DOUT, 1111 = FCO 1 bit after DOUT 0 0 0 0 0 0 MEMORY MAP REGISTER DESCRIPTIONS Default Value Comments 0x00 DCO frequency control (global). X 0 0 0 0 0 0 0 0 0 0 0 0 Enables FCO function (global). Inverts DOUT signal outputs (global). FCO signal controls (global). 0x10 0x00 0x10 0x00 0x00 0x00 Reserved. Reserved. Reserved. Reserved. Reserved. Reserved. 0x00 Vector profile memory (global). Set the speed mode in Register 0x002 and write to Register 0x0FF at the beginning of the setup of the SPI writes after the device is powered up to avoid rewriting other registers after Register 0x0FF is written. Profile Index and Manual TX_TRIG (Register 0x10C) The vector profile is selected using the profile index in Register 0x10C, Bits[4:0]. The manual TX_TRIG control in Bit 5 generates a TX_TRIG signal internal to the device. This signal is asynchronous to the ADC sample clock. Therefore, it cannot be used to align the data output or initiate advanced power mode across multiple devices in the system. The external pin driven TX_TRIG± control is recommended for systems that require synchronization of these features across multiple AD9674 devices. Rev. A | Page 46 of 47 Data Sheet AD9674 OUTLINE DIMENSIONS A1 BALL CORNER 10.10 10.00 SQ 9.90 A1 BALL CORNER 12 11 10 9 8 7 6 5 4 3 2 1 A B C D 8.80 BSC SQ E F G H 0.80 J K L M TOP VIEW 0.60 REF BOTTOM VIEW DETAIL A *1.40 MAX DETAIL A 0.65 MIN 0.25 MIN 0.50 COPLANARITY 0.45 0.12 0.40 BALL DIAMETER *COMPLIANT WITH JEDEC STANDARDS MO-275-EEAB-1 WITH THE EXCEPTION OF PACKAGE HEIGHT. 03-28-2013-B SEATING PLANE Figure 55. 144-Ball Chip Scale Package, Ball Grid Array [CSP_BGA] (BC-144-1) Dimensions shown in millimeters ORDERING GUIDE Model 1 AD9674KBCZ AD9670EBZ 1 Temperature Range 0°C to 85°C Package Description 144-Ball Chip Scale Package, Ball Grid Array [CSP_BGA] Evaluation Board Z = RoHS Compliant Part. ©2016 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D11293-0-1/16(A) Rev. A | Page 47 of 47 Package Option BC-144-1