LINER LT1223M 100mhz current feedback amplifier Datasheet

LT1223
100MHz Current
Feedback Amplifier
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DESCRIPTIO
FEATURES
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100MHz Bandwidth at AV = 1
1000V/µs Slew Rate
Wide Supply Range: ±5V to ±15V
1mV Input Offset Voltage
1µA Input Bias Current
5MΩ Input Resistance
75ns Settling Time to 0.1%
50mA Output Current
6mA Quiescent Current
The LT1223 is a 100MHz current feedback amplifier with
very good DC characteristics. The LT1223’s high slew
rate, 1000V/µs, wide supply range, ±15V, and large output
drive, ±50mA, make it ideal for driving analog signals over
double- terminated cables. The current feedback amplifier
has high gain bandwidth at high gains, unlike conventional
op amps.
The LT1223 comes in the industry standard pinout and
can upgrade the performance of many older products.
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APPLICATI
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The LT1223 is manufactured on Linear Technology’s
proprietary complementary bipolar process.
Video Amplifiers
Buffers
IF and RF Amplification
Cable Drivers
8-, 10-, 12-Bit Data Acquisition Systems
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TYPICAL APPLICATI
Video Cable Driver
Voltage Gain vs Frequency
60
LT1223
40
–
RF
1k
75Ω
CABLE
VOUT
RG
1k
100MHz GAIN
BANDWIDTH
50
75Ω
VOLTAGE GAIN (dB)
+
V IN
30
20
10
0
+
RG = 10
–
RG = 33
RG
1k
RG = 110
RG = 470
RG = ∞
75Ω
–10
–20
100k
R
AV = 1 + F
RG
AT AMPLIFIER OUTPUT.
6dB LESS AT VOUT .
1M
10M
100M
1G
FREQUENCY (Hz)
LT1223 • TPC01
LT1223 • TA02
1
LT1223
W
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W W
W
AXI U
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ABSOLUTE
PACKAGE/ORDER I FOR ATIO
RATI GS
Supply Voltage ...................................................... ±18V
Differential Input Voltage ......................................... ±5V
Input Voltage ............................ Equal to Supply Voltage
Output Short Circuit Duration (Note 1) ......... Continuous
Operating Temperature Range
LT1223M ........................................ –55°C to 125°C
LT1223C ................................................ 0°C to 70°C
Storage Temperature Range ................. –65°C to 150°C
Junction Temperature Plastic Package ........... 150°C
Junction Temperature Ceramic Package ........ 175°C
Lead Temperature (Soldering, 10 sec.)................. 300°C
ORDER PART
NUMBER
TOP VIEW
NULL
1
8
SHUTDOWN
–IN
2
7
V+
+IN
3
6
OUT
V–
4
5
NULL
LT1223MJ8
LT1223CJ8
LT1223CN8
LT1223CS8
J8 PACKAGE
N8 PACKAGE
8-LEAD CERAMIC DIP 8-LEAD PLASTIC DIP
S8 PART MARKING
S8 PACKAGE
8-LEAD PLASTIC SOIC
LT1223 • POI01
1223
TJ MAX = 175°C, θJA = 100°C/W(J8)
TJ MAX = 150°C, θJA = 100°C/W(N8)
TJ MAX = 150°C, θJA = 150°C/W(S8)
ELECTRICAL CHARACTERISTICS VS = ± 15V, TA = 25°C, unless otherwise noted.
LT1223M/C
TYP
MAX
VCM = 0V
±1
±3
mV
Noninverting Input Current
VCM = 0V
±1
±3
µA
Inverting Input Current
VCM = 0V
±1
±3
µA
en
Input Noise Voltage Density
f = 1kHz, RF = 1k, RG = 10Ω
3.3
nV/√Hz
in
Input Noise Current Density
f = 1kHz, RF = 1k, RG = 10Ω
2.2
pA/√Hz
RIN
Input Resistance
VIN = ±10V
1
10
MΩ
CIN
Input Capacitance
1.5
pF
±10
±12
V
SYMBOL
PARAMETER
CONDITIONS
VOS
Input Offset Voltage
IIN+
IIN–
Input Voltage Range
CMRR
PSRR
MIN
UNITS
Common-Mode Rejection Ratio
VCM = ±10V
Inverting Input Current Common-Mode Rejection
VCM = ±10V
Power Supply Rejection Ratio
VS = ±4.5V to ±18V
Noninverting Input Current Power Supply Rejection
VS = ±4.5V to ±18V
12
100
nA/V
Inverting Input Current Power Supply Rejection
VS = ±4.5V to ±18V
60
500
nA/V
56
63
30
68
dB
100
80
nA/V
dB
AV
Large Signal Voltage Gain
RLOAD = 400Ω, VOUT = ±10V
70
89
dB
ROL
Transresistance, ∆VOUT/∆IIN–
RLOAD = 400Ω, VOUT = ±10V
1.5
5
MΩ
VOUT
Maximum Output Voltage Swing
RLOAD = 200Ω
±10
±12
IOUT
Maximum Output Current
RLOAD = 200Ω
50
60
mA
SR
Slew Rate
RF = 1.5k, RG = 1.5k, (Note 2)
800
1300
V/µs
BW
Bandwidth
RF = 1k, RG = 1k, VOUT = 100mV
100
MHz
tr
Rise Time
RF = 1.5k, RG = 1.5k, VOUT = 1V
6.0
ns
tPD
Propagation Delay
RF = 1.5k, RG = 1.5k, VOUT = 1V
6.0
ns
Overshoot
RF = 1.5k, RG = 1.5k, VOUT = 1V
5
%
Settling Time, 0.1%
RF = 1k, RG = 1k, VOUT = 10V
75
ns
Differential Gain
RF = 1k, RG = 1k, RL = 150Ω
0.02
%
Differential Phase
RF = 1k, RG = 1k, RL = 150Ω
0.12
Deg
ROUT
Open-Loop Output Resistance
VOUT = 0, IOUT = 0
IS
Supply Current
VIN = 0V
6
10
mA
Supply Current, Shutdown
Pin 8 Current = 200µA
2
4
mA
ts
2
V
Ω
35
LT1223
ELECTRICAL CHARACTERISTICS VS = ± 15V, VCM = 0V, 0°C ≤ TA ≤ 70°C, unless otherwise noted.
MIN
LT1223C
TYP
MAX
●
±1
±3
mV
●
±1
±3
µA
VCM = 0V
●
±1
±3
µA
VIN = ±10V
●
1
10
MΩ
●
±10
±12
V
56
SYMBOL
PARAMETER
CONDITIONS
VOS
Input Offset Voltage
VCM = 0V
IIN+
Noninverting Input Current
VCM = 0V
IIN–
Inverting Input Current
RIN
Input Resistance
Input Voltage Range
CMRR
PSRR
UNITS
Common-Mode Rejection Ratio
VCM = ±10V
●
Inverting Input Current Common-Mode Rejection
VCM = ±10V
●
Power Supply Rejection Ratio
VS = ±4.5V to ±18V
●
Noninverting Input Current Power Supply Rejection
VS = ±4.5V to ±18V
●
12
100
nA/V
Inverting Input Current Power Supply Rejection
VS = ±4.5V to ±18V
●
60
500
nA/ V
63
30
68
dB
100
80
nA/V
dB
AV
Large-Signal Voltage Gain
RLOAD = 400Ω, VOUT = ±10V
●
70
89
dB
ROL
Transresistance, ∆VOUT/∆IIN–
RLOAD = 400Ω, VOUT = ±10V
●
1.5
5
MΩ
VOUT
Maximum Output Voltage Swing
RLOAD = 200Ω
●
±10
±12
IOUT
Maximum Output Current
RLOAD = 200Ω
●
50
60
IS
Supply Current
VIN = 0V
●
6
10
mA
Supply Current, Shutdown
Pin 8 Current = 200µA
●
2
4
mA
V
mA
ELECTRICAL CHARACTERISTICS VS = ± 15V, VCM = 0V, – 55°C ≤ TA ≤ 125°C, unless otherwise noted.
LT1223M
TYP
MAX
●
±1
±5
mV
VCM = 0V
●
±1
±5
µA
VCM = 0V
●
±1
±10
µA
VIN = ±10V
●
1
10
●
±10
±12
V
●
56
63
dB
SYMBOL
PARAMETER
CONDITIONS
VOS
Input Offset Voltage
VCM = 0V
IIN+
Noninverting Input Current
IIN–
Inverting Input Current
RIN
Input Resistance
CMRR
Common-Mode Rejection Ratio
VCM = ±10V
Inverting Input Current Common-Mode Rejection
VCM = ±10V
●
PSRR
Power Supply Rejection Ratio
VS = ±4.5V to ±15V
●
Noninverting Input Current Power Supply Rejection
VS = ±4.5V to ±15V
●
Input Voltage Range
MIN
30
68
UNITS
MΩ
100
80
nA/V
dB
12
200
nA/V
60
500
nA/V
Inverting Input Current Power Supply Rejection
VS = ±4.5V to ±15V
●
AV
Large-Signal Voltage Gain
RLOAD = 400Ω, VOUT = ±10V
●
70
ROL
Transresistance, ∆VOUT/∆IIN–
RLOAD = 400Ω, VOUT = ±10V
●
VOUT
Maximum Output Voltage Swing
RLOAD = 200Ω
●
IOUT
Maximum Output Current
RLOAD = 200Ω
●
IS
Supply Current
VIN = 0V
●
6
10
mA
Supply Current, Shutdown
Pin 8 Current = 200µA
●
2
4
mA
89
dB
1.5
5
MΩ
±7
±12
V
35
60
mA
The ● denotes the specifications which apply over the full operating
temperature range.
Note 1: A heat sink may be required.
Note 2: Noninverting operation, VOUT = ±10V, measured at ±5V.
3
LT1223
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TYPICAL PERFOR A CE CHARACTERISTICS
Supply Current vs Supply Voltage,
VIN = 0 (Operating)
Supply Current vs Supply Voltage
(Shutdown)
4
100
OUTPUT SHORT CIRCUIT CURRENT (mA)
10
PIN 8 = 0V
125°C
SUPPLY CURRENT (mA)
SUPPLY CURRENT (mA)
8
25°C
6
–55°C
4
3
25°C
125°C
2
–55°C
1
2
0
0
2
4
6
8
2
0
10 12 14 16 18 20
4
6
8
5
–1
4
V S = 15V
+4
+3
VS = –5V
0
100
125
–1
–3
–6
–8
–10
–5
0
5
15
OUTPUT VOLTAGE SWING (V)
VOS (mV)
15
25°C
–55°C
–10
–15
–5
0
5
10
15
COMMON MODE VOLTAGE (V)
LT1223 • TPC08
–5
0
5
10
15
LT1223 • TPC07
Output Voltage Swing vs
Supply Voltage
20
VS = ±15V
10
15
125°C
25°C, –55°C
5
0
–5
–10
25°C, –55°C
125°C
–15
–10
–10
COMMON MODE VOLTAGE (V)
20
0
–20
–15
10
–10
–15
Output Voltage Swing vs
Load Resistor
V S = ±15V
125
25°C
LT1223 • TPC06
20
100
–55°C
–2
–4
VOS vs Common-Mode Voltage
125°C
75
0
COMMON MODE VOLTAGE (V)
5
50
V S = ±15V
2
–2
LT1223 • TPC05
10
25
4
125°C
TEMPERATURE (°C)
–5
0
125°C
25°C
1
–5
–15
15
–25
6
V–
–50
75
10
8
–55°C
–4
50
20
–IB vs Common-Mode Voltage
VS = ±15V
+1
25
30
LT1223 • TPC04
–l B (µA)
–4
0
40
CASE TEMPERATURE (°C)
3
VS = 5V
–25
50
10
2
VS = –15V
60
+IB vs Common-Mode Voltage
V+
+lB (µA)
COMMON MODE RANGE (V)
Input Common-Mode Limit vs
Temperature
+2
70
LT1223 • TPC03
LT1223 • TPC02
–3
80
SUPPLY VOLTAGE (±V)
SUPPLY VOLTAGE (±V)
–2
90
0
–50
10 12 14 16 18 20
OUTPUT VOLTAGE SWING (V)
0
4
Output Short Circuit-Current vs
Temperature
–20
100
125°C
10
–55°C
25°C
5
0
25°C
–5
–10
125°C
–55°C
–15
–20
1000
10000
0
2
4
6
8
10 12 14 16 18 20
SUPPLY VOLTAGE (±V)
LOAD RESISTOR (Ω)
LT1223 • TPC09
LT1223 • TPC10
LT1223
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TYPICAL PERFOR A CE CHARACTERISTICS
100
70
60
50
40
30
80
70
40
10
1
2
RF = 1.5k
30
10
0
RF = 1k
50
20
RF = 2k
5
10
1k
100
3
80
70
125°C
60
1000
25°C
6
125°C
5
4
–55°C
3
1
0
100
10000
1000
LT1223 • TPC16
Output Impedance vs Frequency
80
MAGNITUDE OF OUTPUT IMPEDANCE (Ω)
100
VS = ±15V
RF = 1k
60
POSITIVE
40
NEGATIVE
20
+i n
10k
0
10k
100k
1M
10M
100M
FREQUENCY (Hz)
FREQUENCY (Hz)
LT1223 • TPC17
10000
LOAD RESISTOR (Ω)
Power Supply Rejection vs
Frequency
POWER SUPPLY REJECTION (dB)
SPOT NOISE (nV/√Hz OR pA/√Hz)
7
2
VS = ±15V
VO = ± 10V
50
8
LT1223 • TPC15
1000
60
VS = ± 15V
VO = ± 10V
9
–55°C
Spot Noise Voltage and Current vs
Frequency
1
50
Transimpedance vs Load Resistor
LOAD RESISTOR (Ω)
en
40
10
LT1223 • TPC14
–i n
30
LT1223 • TPC13
90
FEEDBACK RESISTOR (kΩ)
1k
20
VOLTAGE GAIN (V/V)
25°C
40
100
10
100
10
0
TRANSIMPEDANCE (MΩ)
OPEN LOOP VOLTAGE GAIN (dB)
CAPACITIVE LOAD (pF)
A V = 2; RF = RG
R L = 100; VS = ± 15V
PEAKING < 5dB
10
300
Open-Loop Voltage Gain vs
Load Resistor
100
10
2dB PEAKING
400
LT1223 • TPC12
10k
100
500
SUPPLY VOLTAGE (± V)
Maximum Capacitive Load vs
Feedback Resistor
2
0dB PEAKING
600
15
LT1223 • TPC11
1
700
100
0
FEEDBACK RESISTOR (k Ω)
0
800
200
0
3
VS = ±15V
R L = 100
900
RF = 750
60
20
0
RF = RG
AV = 2
RL = 100 Ω
TA = 25°C
90
–3dB BANDWIDTH (MHz)
80
Minimum Feedback Resistor vs
Voltage Gain
1000
100
A V = 2; RF = RG
R L = 100 Ω ; VS = ±15V
NO CAPACITIVE LOAD
90
–3dB BANDWIDTH (MHz)
–3dB Bandwidth vs
Supply Voltage
FEEDBACK RESISTOR (Ω)
–3dB Bandwidth vs
Feedback Resistor
VS = ±15V
10
1
RF = RG = 3k
RF = RG = 1k
0.1
0.01
10k
100k
1M
10M
100M
FREQUENCY (Hz)
LT1223 • TPC18
LT1223 • TPC19
5
LT1223
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TYPICAL PERFOR A CE CHARACTERISTICS
Voltage Gain and Phase vs
Frequency
Total Harmonic Distortion vs
Frequency
225
VOLTAGE GAIN (dB)
135
GAIN
5
RL = 100Ω
90
RL ≥ 1k
0
45
PHASE
–5
–10
RL = 100Ω
0
RL ≥ 1k
–45
PHASE SHIFT (DEGREES)
10
0.1
180
–15
–90
–20
–135
–25
–180
–225
–30
1M
10M
100M
–20
VS = ±15V
VO = 7VRMS
RL = 400 Ω
RF = RG =1k
0.01
THD
100
10
FREQUENCY (Hz)
1k
10k
1
TO 10mV
6
0
–2
–4
Inverting Amplifier Settling
Time vs Output Step
4
6
TO 1mV
2
0
–2
–4
TO 1mV
–6
TO 10mV
4
0
–4
–8
–10
60
80
100
0
LT1223 • TPC23
2
0
20
40
60
80
100
SETTLING TIME (ns)
LT1223 • TPC24
LT1223 • TPC25
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APPLICATI
1
SETTLING TIME (µs)
TO 1mV
TO 10mV
–6
–8
40
TO 1mV
–2
–10
SETTLING TIME (ns)
TO 10mV
2
–10
20
A V = –1
RF = 1k
VS = ± 15V
RL = 1k
8
–8
0
100
LT1223 • TPC22
OUTPUT STEP (V)
2
–6
10
FREQUENCY (MHz)
10
A V = +1
R F = 1k
VS = ± 15V
RL = 1k
8
OUTPUT STEP (V)
OUTPUT STEP (V)
–70
100k
10
4
–50
Noninverting Amplifier Settling
Time to 1mV vs Output Step
10
6
3RD
LT1223 • TPC21
Noninverting Amplifier Settling
Time to 10mV vs Output Step
A V = +1
RF = 1k
VS = ± 15V
RL = 1k
2ND
–40
FREQUENCY (Hz)
LT1223 • TPC20
8
= ± 15V
= 2VP-P
= 100
= 1k
= 10dB
–60
0.001
1G
VS
VO
RL
RF
AV
–30
DISTORTION (dBc)
VS = ±15V
RF = RG = 1k
15
TOTAL HARMONIC DISTORTION (%)
20
2nd and 3rd Harmonic
Distortion vs Frequency
S I FOR ATIO
Current Feedback Basics
The small-signal bandwidth of the LT1223, like all current
feedback amplifiers, isn’t a straight inverse function of the
closed-loop gain. This is because the feedback resistors
determine the amount of current driving the amplifier’s
internal compensation capacitor. In fact, the amplifier’s
feedback resistor (RF) from output to inverting input
works with internal junction capacitances of the LT1223 to
set the closed-loop bandwidth.
Even though the gain set resistor (RG) from inverting input
to ground works with RF to set the voltage gain just like it
6
does in a voltage feedback op amp, the closed-loop
bandwidth does not change. This is because the equivalent gain bandwidth product of the current feedback amplifier is set by the Thevenin equivalent resistance at the
inverting input and the internal compensation capacitor.
By keeping RF constant and changing the gain with RG, the
Thevenin resistance changes by the same amount as the
change in gain. As a result, the net closed-loop bandwidth
of the LT1223 remains the same for various closed-loop
gains.
LT1223
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APPLICATI
S I FOR ATIO
The curve on the first page shows the LT1223 voltage gain
versus frequency while driving 100Ω, for five gain settings
from 1 to 100. The feedback resistor is a constant 1k and
the gain resistor is varied from infinity to 10Ω. Shown for
comparison is a plot of the fixed 100MHz gain bandwidth
limitation that a voltage feedback amplifier would have. It
is obvious that for gains greater than one, the LT1223
provides 3 to 20 times more bandwidth. It is also evident
that second order effects reduce the bandwidth somewhat
at the higher gain settings.
Feedback Resistor Selection
Because the feedback resistor determines the compensation of the LT1223, bandwidth and transient response can
be optimized for almost every application. To increase the
bandwidth when using higher gains, the feedback resistor
(and gain resistor) can be reduced from the nominal 1k
value. The Minimum Feedback Resistor versus Voltage
Gain curve shows the values to use for ±15V supplies.
Larger feedback resistors can also be used to slow down
the LT1223 as shown in the –3dB Bandwidth versus
Feedback Resistor curve.
Capacitive Loads
The LT1223 can be isolated from capacitive loads with a
small resistor (10Ω to 20Ω) or it can drive the capacitive
load directly if the feedback resistor is increased. Both
techniques lower the amplifier’s bandwidth about the
same amount. The advantage of resistive isolation is that
the bandwidth is only reduced when the capacitive load is
present. The disadvantage of resistor isolation is that
resistive loading causes gain errors. Because the DC
accuracy is not degraded with resistive loading, the desired way of driving capacitive loads, such as flash converters, is to increase the feedback resistor. The Maximum
Capacitive Load versus Feedback Resistor curve shows
the value of feedback resistor and capacitive load that
gives 5dB of peaking. For less peaking, use a larger
feedback resistor.
Power Supplies
The LT1223 may be operated with single or split supplies
as low as ±4V (8V total) to as high as ±18V (36V total). It
is not necessary to use equal value split supplies, however, the offset voltage will degrade about 350µV per volt
of mismatch. The internal compensation capacitor decreases with increasing supply voltage. The –3dB Bandwidth versus Supply Voltage curve shows how this affects
the bandwidth for various feedback resistors. Generally,
the bandwidth at ±5V supplies is about half the value it is
at ±15V supplies for a given feedback resistor.
The LT1223 is very stable even with minimal supply
bypassing, however, the transient response will suffer if
the supply rings. It is recommended for good slew rate and
settling time that 4.7µF tantalum capacitors be placed
within 0.5 inches of the supply pins.
Input Range
The noninverting input of the LT1223 looks like a 10M
resistor in parallel with a 3pF capacitor until the common
mode range is exceeded. The input impedance drops
somewhat and the input current rises to about 10µA when
the input comes too close to the supplies. Eventually,
when the input exceeds the supply by one diode drop, the
base collector junction of the input transistor forward
biases and the input current rises dramatically. The input
current should be limited to 10mA when exceeding the
supplies. The amplifier will recover quickly when the input
is returned to its normal common mode range unless the
input was over 500mV beyond the supplies, then it will
take an extra 100ns.
Offset Adjust
Output offset voltage is equal to the input offset voltage
times the gain plus the inverting input bias current times
the feedback resistor. For low gain applications (3 or less)
a 10kΩ pot connected to pins 1 and 5 with wiper to V+ will
trim the inverting input current (±10µA) to null the output;
it does not change the offset voltage very much. If the
LT1223 is used in a high gain application, where input
offset voltage is the dominate error, it can be nulled by
pulling approximately 100µA from pin 1 or 5. The easy way
to do this is to use a 10kΩ pot between pin 1 and 5 with a
150k resistor from the wiper to ground for 15V supply
applications. Use a 47k resistor when operating on a 5V
supply.
7
LT1223
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APPLICATI
S I FOR ATIO
Shutdown
Output Slew Rate of 500V/µs
Pin 8 activates a shutdown control function. Pulling more
than 200µA from pin 8 drops the supply current to less than
3mA, and puts the output into a high impedance state. The
easy way to force shutdown is to ground pin 8, using an
open collector (drain) logic stage. An internal resistor limits
current, allowing direct interfacing with no additional parts.
When pin 8 is open, the LT1223 operates normally.
Slew Rate
The slew rate of a current feedback amplifier is not independent of the amplifier gain configuration the way it is in
a traditional op amp. This is because the input stage and
the output stage both have slew rate limitations. Inverting
amplifiers do not slew the input and are therefore limited
only by the output stage. High gain, noninverting amplifiers are similar. The input stage slew rate of the LT1223 is
about 350V/µs before it becomes nonlinear and is enhanced by the normally reverse-biased emitters on the
input transistors. The output slew rate depends on the size
of the feedback resistors. The peak output slew rate is
about 2000V/µs with a 1k feedback resistor and drops
proportionally for larger values. At an output slew rate of
1000V/µs or more, the transistors in the “mirror circuits”
will begin to saturate due to the large feedback currents.
This causes the output to have slew induced overshoot and
is somewhat unusual looking; it is in no way harmful or
dangerous to the device. The photos show the LT1223 in
a noninverting gain of three (RF = 1k, RG = 500Ω) with a
20V peak-to-peak output slewing at 500V/µs, 1000V/µs
and 2000V/µs.
Settling Time
The Inverting Amplifier Settling Time versus Output Step
curve shows that the LT1223 will settle to within 1mV of
final value in less than 100ns for all output changes of 10V
or less. When operated as an inverting amplifier there is
less than 500µV of thermal settling in the amplifier.
However, when operating the LT1223 as a noninverting
amplifier, there is an additional thermal settling component that is about 200µV for every volt of input common
mode change. So a noninverting gain of one amplifier will
8
Output Slew Rate of 1000V/µs
Output Slew Rate at 2000V/µs Shows Aberrations (See Text)
LT1223
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APPLICATI
S I FOR ATIO
have about 2.5mV thermal tail on a 10V step. Unfortunately, reducing the input signal and increasing the gain
always results in a thermal tail of about the same amount
for a given output step. For this reason we show separate
graphs of 10mV and 1mV non-inverting amplifier settling
times. Just as the bandwidth of the LT1223 is fairly
constant for various closed-loop gains, the settling time
remains constant as well.
Adjustable Gain Amplifier
To make a variable gain amplifier with the LT1223, vary the
value of RG. The implementation of RG can be a pot, a light
controlled resistor, a FET, or any other low capacitance
variable resistor. The value of RF should not be varied to
change the gain. If RF is changed, then the bandwidth will
be reduced at maximum gain and the circuit will oscillate
when RF is very small.
Accurate Bandwidth Limiting The LT1223
It is very common to limit the bandwidth of an op amp by
putting a small capacitor in parallel with RF. DO NOT PUT
A SMALL CAPACITOR FROM THE INVERTING INPUT OF
A CURRENT FEEDBACK AMPLIFIER TO ANYWHERE ELSE,
ESPECIALLY NOT TO THE OUTPUT. The capacitor on the
inverting input will cause peaking or oscillations. If you
need to limit the bandwidth of a current feedback amplifier,
use a resistor and capacitor at the noninverting input (R1
& C1). This technique will also cancel (to a degree) the
peaking caused by stray capacitance at the inverting input.
Unfortunately, this will not limit the output noise the way
it does for the op amp.
V IN
R1
+
LT1223
C1
VOUT
–
V IN
+
RF
VOUT
LT1223
–
RG
R1 = 300Ω
C1 = 100pF
BW = 5MHz
RF
LT1223 • TA05
RG
LT1223 • TA03
Current Feedback Amplifier Integrator
Adjustable Bandwidth Amplifier
Because the resistance at the inverting input determines
the bandwidth of the LT1223, an adjustable bandwidth
circuit can be made easily. The gain is set as before with
RF and RG; the bandwidth is maximum when the variable
resistor is at a minimum.
V IN
+
LT1223
+
VOUT
LT1223
–
VOUT
= 1
sCI RI
VIN
5k
RG
Since we remember that the inverting input wants to see
a resistor, we can add one to the standard integrator
circuit. This generates a new summing node where we can
apply capacitive feedback. The LT1223 integrator has
excellent large signal capability and accurate phase shift at
high frequencies.
RF
VIN
LT1223 • TA04
RI
VOUT
–
RF
1k
CI
LT1223 • TA06
9
LT1223
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APPLICATI
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Summing Amplifier (DC Accurate)
The summing amplifier is easily made by adding additional
inputs to the basic inverting amplifier configuration. The
LT1223 has no IOS spec because there is no correlation
between the two input bias currents. Therefore, we will not
improve the DC accuracy of the inverting amplifier by
putting in the extra resistor in the noninverting input.
+
VOUT
LT1223
R 1
G
V I1
–
R 2
G
R
V I2
•
•
•
R n
G
F
VOUT = –R F
( RV
I1 + VI2 + VIn
R Gn
G1 R G2
VIn
)
inverting input (A1) senses the shield and the non-inverting input (A2) senses the center conductor. Since this
amplifier does not load the cable (take care to minimize
stray capacitance) and it rejects common mode hum and
noise, several amplifiers can sense the signal with only
one termination at the end of the cable. The design
equations are simple. Just select the gain you need (it
should be two or more) and the value of the feedback
resistor (typically 1k) and calculate RG1 and RG2. The gain
can be tweaked with RG2 and the CMRR with RG1 if needed.
The bandwidth of the noninverting input signal is not
reduced by the presence of the other amplifier, however,
the inverting input signal bandwidth is reduced since it
passes two amplifiers. The CMRR is good at high frequencies because the bandwidth of the amplifiers are about the
same even though they do not necessarily operate at the
same gain.
LT1223 • TA07
RG1
1k
Difference Amplifier
RF1
1k
The LT1223 difference amplifier delivers excellent
performance if the source impedance is very low. This is
because the common mode input resistance is only equal
to RF + RG.
RG2
1k
–
–
A1
LT1223
RG
(RF –50)
100
OPTIONAL TRIM
FOR CMRR
A2
LT1223
+
VIN –
V1
RF2
1k
VOUT = G (VIN+ – VIN–)
R
RF1 = RF2; RG1 = (G – 1) RF2; RG2 = F2
G–1
TRIM GAIN (G) WITH RG2; TRIM CMRR WITH RG1
VOUT
+
VIN +
LT1223 • TA09
+
LT1223
RG
V2
VOUT =
VOUT
–
RF
(V1 – V2 )
RG
RF
LT1223 • TA08
Video Instrumentation Amplifier
This instrumentation amplifier uses two LT1223s to increase the input resistance to well over 1M. This makes an
excellent “loop through” or cable sensing amplifier if the
10
Cable Driver
The cable driver circuit is shown on the front page. When
driving a cable it is important to properly terminate both
ends if even modest high frequency performance is
required. The additional advantage of this is that it isolates
the capacitive load of the cable from the amplifier so it can
operate at maximum bandwidth.
LT1223
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TYPICAL APPLICATI
150mA Output Current Video Amp
V+
V+
V IN
+
LT1223
IN
LT1010
20 Ω
BIAS
OUT
75Ω
75Ω
75Ω
75Ω
75Ω
75Ω
75Ω
75Ω
75Ω
75Ω
–
V–
V–
2k
2k
R f = 2k TO STABILIZE CIRCUIT
DIFFERENTIAL GAIN = 1%
DIFFERENTIAL PHASE = 1°
LT1223 • TA10
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7
15k
1
5
BIAS
10k
8
3
6
2
BIAS
4
LT1223 • TA01
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
11
LT1223
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PACKAGE DESCRIPTIO
Dimensions in inches (millimeters) unless otherwise noted.
J8 Package
8-Lead Ceramic DIP
0.005
(0.127)
MIN
0.200
(5.080)
MAX
0.290 – 0.320
(7.366 – 8.128)
0.015 – 0.060
(0.381 – 1.524)
0.008 – 0.018
(0.203 – 0.460)
0.405
(10.287)
MAX
8
6
7
5
0.025
(0.635)
RAD TYP
0.220 – 0.310
(5.588 – 7.874)
0° – 15°
1
0.038 – 0.068
(0.965 – 1.727)
0.385 ± 0.025
(9.779 ± 0.635)
0.125
3.175
0.100 ± 0.010 MIN
(2.540 ± 0.254)
0.014 – 0.026
(0.360 – 0.660)
2
3
4
0.055
(1.397)
MAX
J8 0392
N8 Package
8-Lead Plastic DIP
0.300 – 0.320
(7.620 – 8.128)
0.045 – 0.065
(1.143 – 1.651)
0.130 ± 0.005
(3.302 ± 0.127)
8
7
6
5
0.065
(1.651)
TYP
0.009 – 0.015
(0.229 – 0.381)
(
0.400
(10.160)
MAX
+0.025
0.325 –0.015
+0.635
8.255
–0.381
0.250 ± 0.010
(6.350 ± 0.254)
0.045 ± 0.015
(1.143 ± 0.381)
)
0.100 ± 0.010
(2.540 ± 0.254)
0.125
(3.175)
MIN
0.020
(0.508)
MIN
1
2
3
4
0.018 ± 0.003
(0.457 ± 0.076)
N8 0392
S8 Package
8-Lead Plastic SOIC
0.189 – 0.197
(4.801 – 5.004)
0.010 – 0.020
× 45°
(0.254 – 0.508)
0°– 8° TYP
0.014 – 0.019
(0.355 – 0.483)
0.050
(1.270)
BSC
6
5
0.228 – 0.244
(5.791 – 6.197)
0.150 – 0.157
(3.810 – 3.988)
1
12
7
0.004 – 0.010
(0.101 – 0.254)
0.008 – 0.010
(0.203 – 0.254)
0.016 – 0.050
0.406 – 1.270
8
0.053 – 0.069
(1.346 – 1.752)
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7487
(408) 432-1900 ● FAX: (408) 434-0507 ● TELEX: 499-3977
2
3
4
SO8 0392
LT/GP 1092 5K REV A
 LINEAR TECHNOLOGY CORPORATION 1992