AD AD8551 Zero-drift, single-supply, rail-to-rail input/output operational amplifier Datasheet

Zero-Drift, Single-Supply, Rail-to-Rail
Input/Output Operational Amplifiers
AD8551/AD8552/AD8554
APPLICATIONS
Temperature sensors
Pressure sensors
Precision current sensing
Strain gage amplifiers
Medical instrumentation
Thermocouple amplifiers
PIN CONFIGURATIONS
8
NC
V+
OUT A
NC
AD8551
4
5
01101-001
1
NC
–IN A
+IN A
V–
NC = NO CONNECT
Figure 1. 8-Lead MSOP (RM Suffix)
8 NC
NC 1
–IN A 2
+IN A 3
AD8551
V– 4
7 V+
6 OUT A
5 NC
NC = NO CONNECT
01101-002
Low offset voltage: 1 μV
Input offset drift: 0.005 μV/°C
Rail-to-rail input and output swing
5 V/2.7 V single-supply operation
High gain, CMRR, PSRR: 130 dB
Ultralow input bias current: 20 pA
Low supply current: 700 μA/op amp
Overload recovery time: 50 μs
No external capacitors required
Figure 2. 8-Lead SOIC (R Suffix)
OUT A
–IN A
+IN A
V–
1
8
AD8552
4
5
V+
OUT B
–IN B
+IN B
01101-003
FEATURES
Figure 3. 8-Lead TSSOP (RU Suffix)
GENERAL DESCRIPTION
With an offset voltage of only 1 μV and drift of 0.005 μV/°C, the
AD855x are perfectly suited for applications in which error
sources cannot be tolerated. Temperature, position and pressure
sensors, medical equipment, and strain gage amplifiers benefit
greatly from nearly zero drift over their operating temperature
range. The rail-to-rail input and output swings provided by the
AD855x family make both high-side and low-side sensing easy.
The AD855x family is specified for the extended industrial/auto
motive temperature range (−40°C to +125°C). The AD8551
single amplifier is available in 8-lead MSOP and 8-lead narrow
SOIC packages. The AD8552 dual amplifier is available in 8-lead
narrow SOIC and 8-lead TSSOP surface-mount packages. The
AD8554 quad is available in 14-lead narrow SOIC and 14-lead
TSSOP packages.
+IN A 3
8 V+
AD8552
V– 4
7 OUT B
6 –IN B
5 +IN B
01101-004
–IN A 2
Figure 4. 8-Lead SOIC (R Suffix)
OUT A
–IN A
+IN A
V+
+IN B
–IN B
OUT B
1
14
AD8554
7
8
OUT D
–IN D
+IN D
V–
+IN C
–IN C
OUT C
01101-005
The AD855x family provides the benefits previously found only
in expensive auto-zeroing or chopper-stabilized amplifiers.
Using Analog Devices, Inc. topology, these new zero-drift
amplifiers combine low cost with high accuracy. No external
capacitors are required.
OUT A 1
Figure 5. 14-Lead TSSOP (RU Suffix)
OUT A 1
14 OUT D
–IN A 2
13 –IN D
+IN A 3
12 +IN D
V+ 4
AD8554
11 V–
+IN B 5
10 +IN C
–IN B 6
9 –IN C
OUT B 7
8 OUT C
01101-006
This family of amplifiers has ultralow offset, drift, and bias
current. The AD8551, AD8552, and AD8554 are single, dual,
and quad amplifiers featuring rail-to-rail input and output swings.
All are guaranteed to operate from 2.7 V to 5 V with a single supply.
Figure 6. 14-Lead SOIC (R Suffix)
Rev. C
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
Trademarks and registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.461.3113 ©1999–2007 Analog Devices, Inc. All rights reserved.
AD8551/AD8552/AD8554
TABLE OF CONTENTS
Features .............................................................................................. 1
1/f Noise Characteristics ........................................................... 16
Applications....................................................................................... 1
Intermodulation Distortion ...................................................... 17
General Description ......................................................................... 1
Broadband and External Resistor Noise Considerations...... 18
Pin Configurations ........................................................................... 1
Output Overdrive Recovery...................................................... 18
Revision History ............................................................................... 2
Input Overvoltage Protection ................................................... 18
Specifications..................................................................................... 3
Output Phase Reversal............................................................... 19
Electrical Characteristics............................................................. 3
Capacitive Load Drive ............................................................... 19
Absolute Maximum Ratings............................................................ 5
Power-Up Behavior .................................................................... 19
Thermal Characteristics .............................................................. 5
Applications..................................................................................... 20
ESD Caution.................................................................................. 5
5 V Precision Strain Gage Circuit ............................................ 20
Typical Performance Characteristics ............................................. 6
3 V Instrumentation Amplifier ................................................ 20
Functional Description .................................................................. 14
High Accuracy Thermocouple Amplifier............................... 21
Amplifier Architecture .............................................................. 14
Precision Current Meter............................................................ 21
Basic Auto-Zero Amplifier Theory.......................................... 14
Precision Voltage Comparator.................................................. 21
High Gain, CMRR, PSRR.......................................................... 16
Outline Dimensions ....................................................................... 22
Maximizing Performance Through Proper Layout ............... 16
Ordering Guide .......................................................................... 23
REVISION HISTORY
3/07—Rev. B to Rev. C
Changes to Specifications Section.................................................. 3
2/07—Rev. A to Rev. B
Updated Format..................................................................Universal
Changes to Figure 54...................................................................... 16
Deleted Spice Model Section......................................................... 19
Deleted Figure 63, Renumbered Sequentially ............................ 19
Changes to Ordering Guide .......................................................... 24
11/02—Rev. 0 to Rev. A
Edits to Figure 60............................................................................ 16
Updated Outline Dimensions ....................................................... 20
Rev. C | Page 2 of 24
AD8551/AD8552/AD8554
SPECIFICATIONS
ELECTRICAL CHARACTERISTICS
VS = 5 V, VCM = 2.5 V, VO = 2.5 V, TA = 25°C, unless otherwise noted.
Table 1.
Parameter
INPUT CHARACTERISTICS
Offset Voltage
Symbol
Conditions
Min
VOS
Typ
Max
Unit
1
5
10
50
1.5
300
4
70
200
150
400
5
μV
μV
pA
nA
pA
nA
pA
pA
pA
pA
V
dB
dB
dB
dB
μV/°C
−40°C ≤ TA ≤ +125°C
Input Bias Current
AD8551/AD8554
AD8552
AD8552
Input Offset Current
AD8551/AD8554
AD8552
AD8552
Input Voltage Range
Common-Mode Rejection Ratio
Large Signal Voltage Gain1
Offset Voltage Drift
OUTPUT CHARACTERISTICS
Output Voltage High
IB
10
1.0
160
2.5
20
150
30
150
−40°C ≤ TA ≤ +125°C
−40°C ≤ TA ≤ +85°C
−40°C ≤ TA ≤ +125°C
IOS
−40°C ≤ TA ≤ +125°C
−40°C ≤ TA ≤ +85°C
−40°C ≤ TA ≤ +125°C
CMRR
AVO
ΔVOS/ΔT
VOH
Output Voltage Low
VOL
Output Short-Circuit Limit Current
ISC
VCM = 0 V to +5 V
−40°C ≤ TA ≤ +125°C
RL = 10 kΩ, VO = 0.3 V to 4.7 V
−40°C ≤ TA ≤ +125°C
−40°C ≤ TA ≤ +125°C
RL = 100 kΩ to GND
RL = 100 kΩ to GND @ −40°C to +125°C
RL = 10 kΩ to GND
RL = 10 kΩ to GND @ −40°C to +125°C
RL = 100 kΩ to V+
RL = 100 kΩ to V+ @ −40°C to +125°C
RL = 10 kΩ to V+
RL = 10 kΩ to V+ @ −40°C to +125°C
0
120
115
125
120
4.99
4.99
4.95
4.95
±25
−40°C to +125°C
Output Current
IO
−40°C to +125°C
POWER SUPPLY
Power Supply Rejection Ratio
Supply Current/Amplifier
DYNAMIC PERFORMANCE
Slew Rate
Overload Recovery Time
Gain Bandwidth Product
NOISE PERFORMANCE
Voltage Noise
Voltage Noise Density
Current Noise Density
1
PSRR
ISY
SR
VS = 2.7 V to 5.5 V
−40°C ≤ TA ≤ +125°C
VO = 0 V
−40°C ≤ TA ≤ +125°C
RL = 10 kΩ
GBP
en p-p
en p-p
en
in
0 Hz to 10 Hz
0 Hz to 1 Hz
f = 1 kHz
f = 10 Hz
Gain testing is dependent upon test bandwidth.
Rev. C | Page 3 of 24
120
115
140
130
145
135
0.005
4.998
4.997
4.98
4.975
1
2
10
15
±50
±40
±30
±15
130
130
850
1000
0.4
0.05
1.5
1.0
0.32
42
2
0.04
10
10
30
30
975
1075
0.3
V
V
V
V
mV
mV
mV
mV
mA
mA
mA
mA
dB
dB
μA
μA
V/μs
ms
MHz
μV p-p
μV p-p
nV/√Hz
fA/√Hz
AD8551/AD8552/AD8554
VS = 2.7 V, VCM = 1.35 V, VO = 1.35 V, TA = 25°C, unless otherwise noted.
Table 2.
Parameter
INPUT CHARACTERISTICS
Offset Voltage
Symbol
Conditions
Min
VOS
Typ
Max
Unit
1
5
10
50
1.5
300
4
50
200
150
400
2.7
μV
μV
pA
nA
pA
nA
pA
pA
pA
pA
V
dB
dB
dB
dB
μV/°C
−40°C ≤ TA ≤ +125°C
Input Bias Current
AD8551/AD8554
AD8552
AD8552
Input Offset Current
AD8551/AD8554
AD8552
AD8552
Input Voltage Range
Common-Mode Rejection Ratio
Large Signal Voltage Gain 1
Offset Voltage Drift
OUTPUT CHARACTERISTICS
Output Voltage High
IB
10
1.0
160
2.5
10
150
30
150
−40°C ≤ TA ≤ +125°C
−40°C ≤ TA ≤ +85°C
−40°C ≤ TA ≤ +125°C
IOS
−40°C ≤ TA ≤ +125°C
−40°C ≤ TA ≤ +85°C
−40°C ≤ TA ≤ +125°C
CMRR
AVO
ΔVOS/ΔT
VOH
Output Voltage Low
VOL
Short-Circuit Limit
ISC
VCM = 0 V to 2.7 V
−40°C ≤ TA ≤ +125°C
RL = 10 kΩ, VO = 0.3 V to 2.4 V
−40°C ≤ TA ≤ +125°C
−40°C ≤ TA ≤ +125°C
RL = 100 kΩ to GND
RL = 100 kΩ to GND @ −40°C to +125°C
RL = 10 kΩ to GND
RL = 10 kΩ to GND @ −40°C to +125°C
RL = 100 kΩ to V+
RL = 100 kΩ to V+ @ −40°C to +125°C
RL = 10 kΩ to V+
RL = 10 kΩ to V+ @ −40°C to +125°C
0
115
110
110
105
2.685
2.685
2.67
2.67
±10
−40°C to +125°C
Output Current
IO
−40°C to +125°C
POWER SUPPLY
Power Supply Rejection Ratio
Supply Current/Amplifier
DYNAMIC PERFORMANCE
Slew Rate
Overload Recovery Time
Gain Bandwidth Product
NOISE PERFORMANCE
Voltage Noise
Voltage Noise Density
Current Noise Density
1
PSRR
ISY
SR
VS = 2.7 V to 5.5 V
−40°C ≤ TA ≤ +125°C
VO = 0 V
−40°C ≤ TA ≤ +125°C
2.697
2.696
2.68
2.675
1
2
10
15
±15
±10
±10
±5
130
130
750
950
0.04
10
10
20
20
900
1000
V
V
V
V
mV
mV
mV
mV
mA
mA
mA
mA
dB
dB
μA
μA
RL = 10 kΩ
0.5
0.05
1
V/μs
ms
MHz
0 Hz to 10 Hz
f = 1 kHz
f = 10 Hz
1.6
75
2
μV p-p
nV/√Hz
fA/√Hz
GBP
en p-p
en
in
120
115
130
130
140
130
0.005
Gain testing is dependent upon test bandwidth.
Rev. C | Page 4 of 24
AD8551/AD8552/AD8554
ABSOLUTE MAXIMUM RATINGS
THERMAL CHARACTERISTICS
Table 3.
Parameter
Supply Voltage
Input Voltage
Differential Input Voltage1
ESD (Human Body Model)
Output Short-Circuit Duration to GND
Storage Temperature Range
Operating Temperature Range
Junction Temperature Range
Lead Temperature Range (Soldering, 60 sec)
1
Rating
6V
GND to VS + 0.3 V
±5.0 V
2000 V
Indefinite
−65°C to +150°C
−40°C to +125°C
−65°C to +150°C
300°C
Table 4.
Package Type
8-Lead MSOP (RM)
8-Lead TSSOP (RU)
8-Lead SOIC (R)
14-Lead TSSOP (RU)
14-Lead SOIC (R)
ESD CAUTION
Differential input voltage is limited to ±5.0 V or the supply voltage,
whichever is less.
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
Rev. C | Page 5 of 24
θJA
190
240
158
180
120
θJC
44
43
43
36
36
Unit
°C/W
°C/W
°C/W
°C/W
°C/W
AD8551/AD8552/AD8554
TYPICAL PERFORMANCE CHARACTERISTICS
180
180
VSY = 2.7V
VCM = 1.35V
TA = 25°C
VSY = 5V
VCM = 2.5V
TA = 25°C
160
140
NUMBER OF AMPLIFIERS
120
100
80
60
40
20
140
120
100
80
60
40
–1.5
–0.5
0.5
1.5
OFFSET VOLTAGE (µV)
0
–2.5
01101-007
0
–2.5
2.5
–1.5
Figure 7. Input Offset Voltage Distribution at 2.7 V
50
20
10
+25°C
0
–40°C
8
6
4
2
–20
0
1
2
3
4
INPUT COMMON-MODE VOLTAGE (V)
0
01101-008
–30
VSY = 5V
VCM = 2.5V
TA = –40°C TO +125°C
10
+85°C
NUMBER OF AMPLIFIERS
INPUT BIAS CURRENT (pA)
12
30
–10
2.5
Figure 10. Input Offset Voltage Distribution at 5 V
VSY = 5V
TA = –40°C, +25°C, +85°C
40
0.5
–0.5
1.5
OFFSET VOLTAGE (µV)
01101-010
20
5
Figure 8. Input Bias Current vs. Common-Mode Voltage
1500
1
2
3
4
INPUT OFFSET DRIFT (nV/°C)
5
6
Figure 11. Input Offset Voltage Drift Distribution at 5 V
10k
VSY = 5V
TA = 125°C
1000
0
01101-011
NUMBER OF AMPLIFIERS
160
VSY = 5V
TA = 25°C
OUTPUT VOLTAGE (mV)
0
–500
–1000
100
SOURCE
10
SINK
1
–2000
0
1
2
3
4
INPUT COMMON-MODE VOLTAGE (V)
5
Figure 9. Input Bias Current vs. Common-Mode Voltage
0.1
0.0001
0.001
0.01
0.1
1
LOAD CURRENT (mA)
10
100
Figure 12. Output Voltage to Supply Rail vs. Load Current at 5 V
Rev. C | Page 6 of 24
01101-012
–1500
01101-009
INPUT BIAS CURRENT (pA)
1k
500
AD8551/AD8552/AD8554
800
10k
TA = +25°C
100
SOURCE
10
SINK
1
0.001
0.01
0.1
1
LOAD CURRENT (mA)
10
100
500
400
300
200
100
0
01101-013
0.1
0.0001
600
1
2
3
4
SUPPLY VOLTAGE (V)
5
6
Figure 16. Supply Current per Amplifier vs. Supply Voltage
Figure 13. Output Voltage to Supply Rail vs. Load Current at 2.7 V
60
0
VCM = 2.5V
VSY = 5V
50
40
–250
OPEN-LOOP GAIN (dB)
INPUT BIAS CURRENT (pA)
0
–500
–750
VSY = 2.7V
CL = 0pF
RL = ∞
0
30
45
20
90
10
135
0
180
–10
225
–20
270
PHASE SHIFT (Degrees)
OUTPUT VOLTAGE (mV)
1k
700
01101-016
SUPPLY CURRENT PER AMPLIFIER (µA)
VSY = 2.7V
TA = 25°C
–25
0
25
50
75
TEMPERATURE (°C)
100
125
150
–40
10k
60
1.0
50
5V
OPEN-LOOP GAIN (dB)
40
2.7V
0.6
0.4
0.2
10M
100M
VSY = 5V
CL = 0pF
RL = ∞
0
30
45
20
90
10
135
0
180
–10
225
–20
270
–50
–25
0
25
50
75
TEMPERATURE (°C)
100
125
150
–40
10k
100k
1M
FREQUENCY (Hz)
10M
100M
Figure 18. Open-Loop Gain and Phase Shift vs. Frequency at 5 V
Figure 15. Supply Current vs. Temperature
Rev. C | Page 7 of 24
01101-018
–30
0
–75
01101-015
SUPPLY CURRENT (mA)
0.8
1M
FREQUENCY (Hz)
Figure 17. Open-Loop Gain and Phase Shift vs. Frequency at 2.7 V
Figure 14. Input Bias Current vs. Temperature
VCM = 2.5V
VSY = 5V
100k
PHASE SHIFT (Degrees)
–50
01101-014
–1000
–75
01101-017
–30
AD8551/AD8552/AD8554
60
300
VSY = 2.7V
CL = 0pF
RL = 2kΩ
AV = –100
20
10
0
240
OUTPUT IMPEDANCE (Ω)
30
AV = –10
AV = +1
–10
180
150
120
–30
30
1k
10k
100k
FREQUENCY (Hz)
1M
10M
30
20
10
0
1k
10k
100k
FREQUENCY (Hz)
1M
10M
VSY = 2.7V
CL = 300pF
RL = 2kΩ
AV = 1
VSY = 5V
CL = 0pF
RL = 2kΩ
AV = –100
AV = 1
Figure 22. Output Impedance vs. Frequency at 5 V
60
40
AV = 10
0
100
Figure 19. Closed-Loop Gain vs. Frequency at 2.7 V
50
AV = 100
90
60
–40
100
CLOSED-LOOP GAIN (dB)
210
–20
01101-019
CLOSED-LOOP GAIN (dB)
40
VSY = 5V
270
AV = –10
AV = +1
–10
–30
2µs
1k
10k
100k
FREQUENCY (Hz)
1M
10M
Figure 20. Closed-Loop Gain vs. Frequency at 5 V
Figure 23. Large Signal Transient Response at 2.7 V
300
270
VSY = 5V
CL = 300pF
RL = 2kΩ
AV = 1
VSY = 2.7V
240
210
180
150
120
90
AV = 100
60
0
100
AV = 1
1k
10k
100k
FREQUENCY (Hz)
1M
5µs
10M
1V
Figure 21. Output Impedance vs. Frequency at 2.7 V
Figure 24. Large Signal Transient Response at 5 V
Rev. C | Page 8 of 24
01101-024
AV = 10
30
01101-021
OUTPUT IMPEDANCE (Ω)
500mV
01101-020
–40
100
01101-023
–20
01101-022
50
AD8551/AD8552/AD8554
45
VSY = ±1.35V
CL = 50pF
VSY = ±2.5V
RL = 2kΩ
TA = 25°C
40
35
30
25
20
+OS
15
–OS
10
5
0
10
Figure 25. Small Signal Transient Response at 2.7 V
100
1k
CAPACITANCE (pF)
10k
Figure 28. Small Signal Overshoot vs. Load Capacitance at 5 V
VSY = ±2.5V
CL = 50pF
0V
RL = ∞
AV = 1
VIN
VSY = ±2.5V
VIN = –200mV p-p
(RET TO GND)
CL = 0pF
RL = 10kΩ
AV = –100
VOUT
0V
20µs
1V
01101-029
01101-026
50mV
5µs
BOTTOM SCALE: 1V/DIV
TOP SCALE: 200mV/DIV
Figure 26. Small Signal Transient Response at 5 V
SMALL SIGNAL OVERSHOOT (%)
45
40
Figure 29. Positive Overvoltage Recovery
VSY = ±1.35V
RL = 2kΩ
TA = 25°C
VIN
0V
35
30
25
0V
+OS
20
15
–OS
VOUT
10
5
20µs
100
1k
CAPACITANCE (pF)
10k
01101-027
0
10
VSY = ±2.5V
VIN = 200mV p-p
(RET TO GND)
CL = 0pF
RL = 10kΩ
AV = –100
1V
BOTTOM SCALE: 1V/DIV
TOP SCALE: 200mV/DIV
Figure 27. Small Signal Overshoot vs. Load Capacitance at 2.7 V
Figure 30. Negative Overvoltage Recovery
Rev. C | Page 9 of 24
01101-030
50
01101-028
50mV
5µs
01101-025
SMALL SIGNAL OVERSHOOT (%)
RL = ∞
AV = 1
AD8551/AD8552/AD8554
140
VS = ±2.5V
RL = 2kΩ
AV = –100
VIN = 60mV p-p
VSY = ±1.35V
120
PSRR (dB)
100
80
60
+PSRR
40
0
100
Figure 31. No Phase Reversal
10k
100k
FREQUENCY (Hz)
1M
10M
Figure 34. PSRR vs. Frequency at ±1.35 V
140
VSY = 2.7V
120
120
100
100
80
80
PSRR (dB)
CMRR (dB)
140
1k
01101-034
20
01101-031
1V
200µs
–PSRR
60
VSY = ±2.5V
+PSRR
60
40
40
–PSRR
1k
10k
100k
FREQUENCY (Hz)
1M
10M
0
100
01101-032
Figure 32. CMRR vs. Frequency at 2.7 V
140
3.0
120
1M
10M
VSY = ±1.35V
RL = 2kΩ
AV = 1
THD+N < 1%
TA = 25°C
2.5
OUTPUT SWING (V p-p)
100
80
60
40
2.0
1.5
1.0
0.5
20
1k
10k
100k
FREQUENCY (Hz)
1M
10M
01101-033
CMRR (dB)
10k
100k
FREQUENCY (Hz)
Figure 35. PSRR vs. Frequency at ±2.5 V
VSY = 5V
0
100
1k
Figure 33. CMRR vs. Frequency at 5 V
0
100
1k
10k
FREQUENCY (Hz)
100k
1M
Figure 36. Maximum Output Swing vs. Frequency at 2.7 V
Rev. C | Page 10 of 24
01101-036
0
100
01101-035
20
20
AD8551/AD8552/AD8554
5.5
VSY = ±2.5V
RL = 2kΩ
AV = 1
THD+N < 1%
TA = 25°C
5.0
4.0
156
en (nV/√Hz)
3.5
3.0
2.5
2.0
130
104
78
52
1.5
1.0
26
1k
10k
FREQUENCY (Hz)
100k
1M
0
01101-037
0
100
Figure 37. Maximum Output Swing vs. Frequency at 5 V
0.5
1.0
1.5
FREQUENCY (kHz)
2.0
2.5
01101-040
0.5
Figure 40. Voltage Noise Density at 2.7 V from 0 Hz to 2.5 kHz
VSY = ±1.35V
AV = 10000
VSY = 2.7V
RS = 0Ω
112
en (nV/√Hz)
96
0V
80
64
48
32
2mV
0
Figure 38. 0.1 Hz to 10 Hz Noise at 2.7 V
5
10
15
FREQUENCY (kHz)
20
25
01101-041
16
01101-038
1s
Figure 41. Voltage Noise Density at 2.7 V from 0 Hz to 25 kHz
VSY = ±2.5V
AV = 10000
VSY = 5V
RS = 0Ω
91
en (nV/√Hz)
78
65
52
39
26
2mV
13
0
Figure 39. 0.1 Hz to 10 Hz Noise at 5 V
0.5
1.0
1.5
FREQUENCY (kHz)
2.0
2.5
Figure 42. Voltage Noise Density at 5 V from 0 Hz to 2.5 kHz
Rev. C | Page 11 of 24
01101-042
1s
01101-039
OUTPUT SWING (V p-p)
4.5
VSY = 2.7V
RS = 0Ω
182
AD8551/AD8552/AD8554
150
VSY = 5V
RS = 0Ω
VSY = 2.7V TO 5.5V
80
64
48
32
0
5
10
15
FREQUENCY (kHz)
20
25
01101-043
16
145
140
135
130
125
–75
50
VSY = 5V
RS = 0Ω
0
25
50
75
TEMPERATURE (°C)
100
125
150
VSY = 2.7V
SHORT-CIRCUIT CURRENT (mA)
40
144
120
96
72
48
24
30
ISC–
20
10
0
ISC+
–10
–20
–30
–40
0
5
FREQUENCY (Hz)
10
01101-044
en (nV/√Hz)
–25
Figure 45. Power Supply Rejection vs. Temperature
Figure 43. Voltage Noise Density at 5 V from 0 Hz to 25 kHz
168
–50
–50
–75
–50
–25
0
25
50
75
TEMPERATURE (°C)
100
125
Figure 46. Output Short-Circuit Current vs. Temperature
Figure 44. Voltage Noise Density at 5 V from 0 Hz to 10 Hz
Rev. C | Page 12 of 24
150
01101-046
en (nV/√Hz)
96
01101-045
POWER SUPPLY REJECTION (dB)
112
AD8551/AD8552/AD8554
OUTPUT VOLTAGE TO SUPPLY RAIL (mV)
80
SHORT-CIRCUIT CURRENT (mA)
250
VSY = 5.0V
ISC–
60
40
20
0
–20
ISC+
–40
–60
–100
–75
–50
–25
0
25
50
75
TEMPERATURE (°C)
100
125
150
01101-047
–80
VSY = 2.7V
200
175
150
125
RL = 1kΩ
100
75
50
RL = 100kΩ
25
0
–75
RL = 10kΩ
–50
–25
0
25
50
75
TEMPERATURE (°C)
100
125
150
01101-048
OUTPUT VOLTAGE TO SUPPLY RAIL (mV)
225
VSY = 5.0V
200
175
150
RL = 1kΩ
125
100
75
50
RL = 100kΩ
25
0
–75
RL = 10kΩ
–50
–25
0
25
50
75
TEMPERATURE (°C)
100
125
150
Figure 49. Output Voltage to Supply Rail vs. Temperature
Figure 47. Output Short-Circuit Current vs. Temperature
250
225
Figure 48. Output Voltage to Supply Rail vs. Temperature
Rev. C | Page 13 of 24
01101-049
100
AD8551/AD8552/AD8554
FUNCTIONAL DESCRIPTION
The AD855x family of amplifiers are high precision, rail-to-rail
operational amplifiers that can be run from a single-supply
voltage. Their typical offset voltage of less than 1 μV allows
these amplifiers to be easily configured for high gains without
risk of excessive output voltage errors. The extremely small
temperature drift of 5 nV/°C ensures a minimum of offset
voltage error over its entire temperature range of −40°C to
+125°C, making the AD855x amplifiers ideal for a variety of
sensitive measurement applications in harsh operating
environments, such as underhood and braking/suspension
systems in automobiles.
The AD855x family are CMOS amplifiers and achieve their
high degree of precision through auto-zero stabilization. This
autocorrection topology allows the AD855x to maintain its low
offset voltage over a wide temperature range and over its
operating lifetime.
AMPLIFIER ARCHITECTURE
Each AD855x op amp consists of two amplifiers, a main amplifier and a secondary amplifier, used to correct the offset voltage
of the main amplifier. Both consist of a rail-to-rail input stage,
allowing the input common-mode voltage range to reach both
supply rails. The input stage consists of an NMOS differential
pair operating concurrently with a parallel PMOS differential
pair. The outputs from the differential input stages are combined
in another gain stage whose output is used to drive a rail-to-rail
output stage.
The wide voltage swing of the amplifier is achieved by using two
output transistors in a common-source configuration. The
output voltage range is limited by the drain-to-source resistance
of these transistors. As the amplifier is required to source or
sink more output current, the rDS of these transistors increases,
raising the voltage drop across these transistors. Simply put, the
output voltage does not swing as close to the rail under heavy
output current conditions as it does with light output current.
This is a characteristic of all rail-to-rail output amplifiers.
Figure 12 and Figure 13 show how close the output voltage can
get to the rails with a given output current. The output of the
AD855x is short-circuit protected to approximately 50 mA of
current.
The AD855x amplifiers have exceptional gain, yielding greater
than 120 dB of open-loop gain with a load of 2 kΩ. Because the
output transistors are configured in a common-source
configuration, the gain of the output stage, and thus the openloop gain of the amplifier, is dependent on the load resistance.
Open-loop gain decreases with smaller load resistances. This is
another characteristic of rail-to-rail output amplifiers.
BASIC AUTO-ZERO AMPLIFIER THEORY
Autocorrection amplifiers are not a new technology. Various IC
implementations have been available for more than 15 years with
some improvements made over time. The AD855x design offers
a number of significant performance improvements over previous
versions while attaining a very substantial reduction in device
cost. This section offers a simplified explanation of how the
AD855x is able to offer extremely low offset voltages and high
open-loop gains.
As noted in the Amplifier Architecture section, each AD855x
op amp contains two internal amplifiers. One is used as the
primary amplifier, the other as an autocorrection, or nulling,
amplifier. Each amplifier has an associated input offset voltage
that can be modeled as a dc voltage source in series with the
noninverting input. In Figure 50 and Figure 51 these are labeled
as VOSX, where x denotes the amplifier associated with the offset:
A for the nulling amplifier and B for the primary amplifier. The
open-loop gain for the +IN and −IN inputs of each amplifier is
given as AX. Both amplifiers also have a third voltage input with
an associated open-loop gain of BX.
There are two modes of operation determined by the action of
two sets of switches in the amplifier: an auto-zero phase and an
amplification phase.
Auto-Zero Phase
In this phase, all φA switches are closed and all φB switches are
opened. Here, the nulling amplifier is taken out of the gain loop
by shorting its two inputs together. Of course, there is a degree
of offset voltage, shown as VOSA, inherent in the nulling amplifier
which maintains a potential difference between the +IN and
−IN inputs. The nulling amplifier feedback loop is closed through
φB2 and VOSA appears at the output of the nulling amp and on
CM1, an internal capacitor in the AD855x. Mathematically, this
is expressed in the time domain as
VOA[t] = AAVOSA[t] − BAVOA[t]
B
(1)
which can be expressed as
VOA [t ] =
AAVOSA [t ]
1 + BA
(2)
This demonstrates that the offset voltage of the nulling amplifier
times a gain factor appears at the output of the nulling amplifier
and, thus, on the CM1 capacitor.
Rev. C | Page 14 of 24
AD8551/AD8552/AD8554
VIN+
BB
ФB
VOA
VOSA
+
ФA
VOA [t ] = AAVIN [t ] +
VOUT
AB
VIN–
ФB
AA
CM2
⎛
⎞
V
VOA [t ] = AA ⎜⎜ VIN [t ] + OSA ⎟⎟
1 + BA ⎠
⎝
CM1
01101-050
ФA
(6)
or
VNB
–BA
AA (1 + BA )VOSA − AA BAVOSA
1 + BA
VNA
Figure 50. Auto-Zero Phase of the AD855x
Amplification Phase
When the φB switches close and the φA switches open for the
amplification phase, this offset voltage remains on CM1 and,
essentially, corrects any error from the nulling amplifier. The
voltage across CM1 is designated as VNA. Furthermore, VIN is
designated as the potential difference between the two inputs to
the primary amplifier, or VIN = (VIN+ − VIN−). Thus, the nulling
amplifier can be expressed as
VOA [t ] = A A (V IN [t ] − VOSA [t ]) − B AVNA [t ]
(3)
(7)
From these equations, the auto-zeroing action becomes evident.
Note the VOS term is reduced by a 1 + BA factor. This shows how
the nulling amplifier has greatly reduced its own offset voltage
error even before correcting the primary amplifier. This results
in the primary amplifier output voltage becoming the voltage at
the output of the AD855x amplifier. It is equal to
VOUT [t ] = AB (VIN [t ] + VOSB ) + BBVNB
(8)
In the amplification phase, VOA = VNB, so this can be rewritten as
⎡ ⎛
V
VOUT [t ] = A B VIN [t ] + A B VOSB + B B ⎢ A A ⎜⎜ VIN [t ] + OSB
1 + BA
⎢⎣ ⎝
⎞⎤
⎟⎥
⎟
⎠⎥⎦
(9)
Combining terms,
VIN+
VOUT
AB
VIN–
VOUT [t ] = VIN [t ](AB + AB BB ) +
BB
ФB
VOA
VOSA
+
ФA
ФB
CM2
AA BAVOSA
+ ABVOSA
1 + BA
(10)
The AD855x architecture is optimized in such a way that
AA
VNB
AA = AB and BA = BB and BA >> 1
B
–BA
ФA
CM1
B
B
B
Also, the gain product of AABB is much greater than AB. These
allow Equation 10 to be simplified to
VNA
01101-051
B
VOUT [t ] ≈ VIN [t ]AA BA + AA (VOSA + VOSB )
Figure 51. Output Phase of the Amplifier
Because φA is now open and there is no place for CM1 to
discharge, the voltage (VNA), at the present time (t), is equal to
the voltage at the output of the nulling amp (VOA) at the time
when φA was closed. If the period of the autocorrection switching
frequency is labeled tS, then the amplifier switches between
phases every 0.5 × tS. Therefore, in the amplification phase
1
VNA [t ] = VNA ⎡⎢t − t S ⎤⎥
⎣ 2 ⎦
(4)
Substituting Equation 4 and Equation 2 into Equation 3 yields
1
AA B AVOSA ⎡⎢t − t S ⎤⎥
2 ⎦
⎣
VOA [t ] = AAVIN [t ] + AAVOSA [t ] −
1 + BA
B
(11)
Most obvious is the gain product of both the primary and
nulling amplifiers. This AABA term is what gives the AD855x its
extremely high open-loop gain. To understand how VOSA and
VOSB relate to the overall effective input offset voltage of the
complete amplifier, establish the generic amplifier equation of
B
B
VOUT = k × (VIN + VOS , EFF )
(12)
where k is the open-loop gain of an amplifier and VOS, EFF is its
effective offset voltage.
Putting Equation 12 into the form of Equation 11 gives
VOUT [t ] ≈ VIN [t ]AA BA + VOS , EFF AA BA
(13)
(5)
Thus, it is evident that
For the sake of simplification, assume that the autocorrection
frequency is much faster than any potential change in VOSA or
VOSB. This is a valid assumption because changes in offset
voltage are a function of temperature variation or long-term
wear time, both of which are much slower than the auto-zero
clock frequency of the AD855x. This effectively renders VOS
time invariant; therefore, Equation 5 can be rearranged and
rewritten as
VOS , EFF ≈
VOSA + VOSB
BA
(14)
B
The offset voltages of both the primary and nulling amplifiers
are reduced by the Gain Factor BA. This takes a typical input
offset voltage from several millivolts down to an effective input
offset voltage of submicrovolts. This autocorrection scheme is
the outstanding feature of the AD855x series that continues to
Rev. C | Page 15 of 24
AD8551/AD8552/AD8554
HIGH GAIN, CMRR, PSRR
Common-mode and power supply rejection are indications
of the amount of offset voltage an amplifier has as a result of a
change in its input common-mode or power supply voltages. As
shown in the previous section, the autocorrection architecture
of the AD855x allows it to quite effectively minimize offset voltages. The technique also corrects for offset errors caused by
common-mode voltage swings and power supply variations.
This results in superb CMRR and PSRR figures in excess of
130 dB. Because the autocorrection occurs continuously, these
figures can be maintained across the entire temperature range
of the device, from −40°C to +125°C.
MAXIMIZING PERFORMANCE THROUGH
PROPER LAYOUT
To achieve the maximum performance of the extremely high
input impedance and low offset voltage of the AD855x, care is
needed in laying out the circuit board. The PC board surface
must remain clean and free of moisture to avoid leakage currents between adjacent traces. Surface coating of the circuit
board reduces surface moisture and provides a humidity barrier,
reducing parasitic resistance on the board. The use of guard
rings around the amplifier inputs further reduces leakage currents. Figure 52 shows proper guard ring configuration, and
Figure 53 shows the top view of a surface-mount layout. The
guard ring does not need to be a specific width, but it should
form a continuous loop around both inputs. By setting the
guard ring voltage equal to the voltage at the noninverting
input, parasitic capacitance is minimized as well. For further
reduction of leakage currents, components can be mounted to
the PC board using Teflon standoff insulators.
Other potential sources of offset error are thermoelectric
voltages on the circuit board. This voltage, also called Seebeck
voltage, occurs at the junction of two dissimilar metals and is
proportional to the temperature of the junction. The most
common metallic junctions on a circuit board are solder-toboard trace and solder-to-component lead. Figure 54 shows a
cross-section of the thermal voltage error sources. If the
temperature of the PC board at one end of the component (TA1)
is different from the temperature at the other end (TA2), the
resulting Seebeck voltages are not equal, resulting in a thermal
voltage error.
This thermocouple error can be reduced by using dummy
components to match the thermoelectric error source. Placing
the dummy component as close as possible to its partner ensures
both Seebeck voltages are equal, thus canceling the thermocouple error. Maintaining a constant ambient temperature on
the circuit board further reduces this error. The use of a ground
plane helps distribute heat throughout the board and reduces
EMI noise pickup.
COMPONENT
LEAD
VSC1 +
SURFACE-MOUNT
COMPONENT
VTS1 +
+
PC BOARD
TA1
TA2
IF TA1 ≠ TA2, THEN
VTS1 + VSC1 ≠ VTS2 + VSC2
COPPER
TRACE
Figure 54. Mismatch in Seebeck Voltages Causes
Thermoelectric Voltage Error
RF
R1
VOUT
VIN
AD8551/
AD8552/
AD8554
VIN
AD8552
AV = 1 + (RF/R1)
VOUT
NOTES
1. RS SHOULD BE PLACED IN CLOSE PROXIMITY AND
ALIGNMENT TO R1 TO BALANCE SEEBECK VOLTAGES.
AD8552
Figure 55. Using Dummy Components to Cancel
Thermoelectric Voltage Errors
VIN
VOUT
01101-052
AD8552
Figure 52. Guard Ring Layout and Connections to Reduce
PC Board Leakage Currents
R1
V+
R2
AD8552
VIN1
R2
R1
VREF
VREF
GUARD
RING
V–
01101-053
VIN2
GUARD
RING
01101-055
VIN
SOLDER
+ VTS2
RS = R1
VOUT
VSC2
1/f NOISE CHARACTERISTICS
Another advantage of auto-zero amplifiers is their ability to
cancel flicker noise. Flicker noise, also known as 1/f noise, is
noise inherent in the physics of semiconductor devices, and it
increases 3 dB for every octave decrease in frequency. The 1/f
corner frequency of an amplifier is the frequency at which the
flicker noise is equal to the broadband noise of the amplifier.
At lower frequencies, flicker noise dominates, causing higher
degrees of error for sub-Hertz frequencies or dc precision
applications.
Figure 53. Top View of AD8552 SOIC Layout with Guard Rings
Rev. C | Page 16 of 24
01101-054
earn the reputation of being among the most precise amplifiers
available on the market.
AD8551/AD8552/AD8554
0
INTERMODULATION DISTORTION
The 4 kHz auto-zero clock frequency appears at the output with
less than 2 μV of amplitude. Harmonics are also present, but at
reduced levels from the fundamental auto-zero clock frequency.
The amplitude of the clock frequency feedthrough is proportional
to the closed-loop gain of the amplifier. Like other autocorrection
amplifiers, at higher gains there is more clock frequency
feedthrough. Figure 57 shows the spectral output with the
amplifier configured for a gain of 60 dB.
VSY = 5V
AV = 0dB
–80
–100
1
2
3
4
5
6
FREQUENCY (kHz)
7
8
9
10
01101-057
0
Figure 57. Spectral Analysis of AD855x Output with +60 dB Gain
When an input signal is applied, the output contains some
degree of intermodulation distortion (IMD). This is another
characteristic feature of all autocorrection amplifiers. IMD
appears as sum and difference frequencies between the input
signal and the 4 kHz clock frequency (and its harmonics) and is
at a level similar to, or less than, the clock feedthrough at the
output. The IMD is also proportional to the closed-loop gain of
the amplifier. Figure 58 shows the spectral output of an AD8552
configured as a high gain stage (+60 dB) with a 1 mV input
signal applied. The relative levels of all IMD products and
harmonic distortion add up to produce an output error of
−60 dB relative to the input signal. At unity gain, these add
up to only −120 dB relative to the input signal.
0
VSY = 5V
AV = 60dB
OUTPUT SIGNAL
1V rms @ 200Hz
–40
–20
–80
–100
0
1
2
3
4
5
6
FREQUENCY (kHz)
7
8
9
10
–60
–80
IMD < 100µV rms
–100
01101-056
–120
–40
–120
Figure 56. Spectral Analysis of AD8552 Output in Unity Gain Configuration
0
1
2
3
4
5
6
FREQUENCY (kHz)
7
8
9
10
01101-058
–60
–140
–60
–140
OUTPUT SIGNAL (dB)
OUTPUT SIGNAL (dB)
–20
–40
–120
The AD855x can be used as a conventional op amp for gain/
bandwidth combinations up to 1.5 MHz. The auto-zero correction
frequency of the device is fixed at 4 kHz. Although a trace
amount of this frequency feeds through to the output, the
amplifier can be used at much higher frequencies. Figure 56
shows the spectral output of the AD8552 with the amplifier
configured for unity gain and the input grounded.
0
VSY = 5V
AV = 60dB
–20
OUTPUT SIGNAL (dB)
Because the AD855x amplifiers are self-correcting op amps, they
do not have increasing flicker noise at lower frequencies. In
essence, low frequency noise is treated as a slowly varying offset
error and is greatly reduced as a result of autocorrection. The
correction becomes more effective as the noise frequency
approaches dc, offsetting the tendency of the noise to increase
exponentially as frequency decreases. This allows the AD855x
to have lower noise near dc than standard low noise amplifiers
that are susceptible to 1/f noise.
Figure 58. Spectral Analysis of AD8552 in High Gain with a 1 mV Input Signal
For most low frequency applications, the small amount of autozero clock frequency feedthrough does not affect the precision
of the measurement system. If it is desired, the clock frequency
feedthrough can be reduced through the use of a feedback
capacitor around the amplifier. However, this reduces the
bandwidth of the amplifier. Figure 59 and Figure 60 show a
configuration for reducing the clock feedthrough and the
corresponding spectral analysis at the output. The −3 dB
bandwidth of this configuration is 480 Hz.
Rev. C | Page 17 of 24
AD8551/AD8552/AD8554
3.3nF
Because the input current noise of the AD855x is very small,
it does not become a dominant term unless RS is greater than
4 GΩ, which is an impractical value of source resistance.
100kΩ
100Ω
The total noise (en, TOTAL) is expressed in volts per square root
Hertz, and the equivalent rms noise over a certain bandwidth
can be found as
01101-059
VIN = 1mV rms
@ 200Hz
Figure 59. Reducing Autocorrection Clock Noise Using a Feedback Capacitor
en = en,TOTAL × BW
(16)
0
VSY = 5V
AV = 60dB
where BW is the bandwidth of interest in Hertz.
OUTPUT SIGNAL
–20
OUTPUT OVERDRIVE RECOVERY
The AD855x amplifiers have an excellent overdrive recovery of
only 200 μs from either supply rail. This characteristic is particularly difficult for autocorrection amplifiers because the
nulling amplifier requires a nontrivial amount of time to error
correct the main amplifier back to a valid output. Figure 29 and
Figure 30 show the positive and negative overdrive recovery
times for the AD855x.
–40
–60
–80
–120
0
2
1
3
4
5
6
FREQUENCY (kHz)
7
8
9
10
01101-060
–100
Figure 60. Spectral Analysis Using a Feedback Capacitor
BROADBAND AND EXTERNAL RESISTOR NOISE
CONSIDERATIONS
The total broadband noise output from any amplifier is primarily
a function of three types of noise: input voltage noise from the
amplifier, input current noise from the amplifier, and Johnson
noise from the external resistors used around the amplifier.
Input voltage noise, or en, is strictly a function of the amplifier
used. The Johnson noise from a resistor is a function of the resistance and the temperature. Input current noise, or in, creates
an equivalent voltage noise proportional to the resistors used
around the amplifier. These noise sources are not correlated
with each other and their combined noise sums in a rootsquared-sum fashion. The full equation is given as
[
en _ TOTAL = en2 + 4kTrS + (in RS )2
]
1
2
(15)
Where:
en = the input voltage noise density of the amplifier.
in = the input current noise of the amplifier.
RS = source resistance connected to the noninverting terminal.
k = Boltzmann’s constant (1.38 × 10−23 J/K).
T = ambient temperature in Kelvin (K = 273.15 + °C).
The input voltage noise density (en) of the AD855x is 42 nV/√Hz,
and the input noise, in, is 2 fA/√Hz. The en, TOTAL is dominated by
the input voltage noise, provided the source resistance is less
than 106 kΩ. With source resistance greater than 106 kΩ, the
overall noise of the system is dominated by the Johnson noise of
the resistor itself.
The output overdrive recovery for an autocorrection amplifier is
defined as the time it takes for the output to correct to its final
voltage from an overload state. It is measured by placing the
amplifier in a high gain configuration with an input signal that
forces the output voltage to the supply rail. The input voltage is
then stepped down to the linear region of the amplifier, usually
to halfway between the supplies. The time from the input signal
stepdown to the output settling to within 100 μV of its final
value is the overdrive recovery time.
INPUT OVERVOLTAGE PROTECTION
Although the AD855x is a rail-to-rail input amplifier, exercise
care to ensure that the potential difference between the inputs
does not exceed 5 V. Under normal operating conditions, the
amplifier corrects its output to ensure the two inputs are at the
same voltage. However, if the device is configured as a comparator,
or is under some unusual operating condition, the input voltages
may be forced to different potentials. This can cause excessive
current to flow through internal diodes in the AD855x used to
protect the input stage against overvoltage.
If either input exceeds either supply rail by more than 0.3 V,
large amounts of current begin to flow through the ESD protection diodes in the amplifier. These diodes connect between
the inputs and each supply rail to protect the input transistors
against an electrostatic discharge event and are normally
reverse-biased. However, if the input voltage exceeds the supply
voltage, these ESD diodes become forward-biased. Without
current limiting, excessive amounts of current can flow through
these diodes, causing permanent damage to the device. If inputs
are subjected to overvoltage, appropriate series resistors should
be inserted to limit the diode current to less than 2 mA maximum.
Rev. C | Page 18 of 24
AD8551/AD8552/AD8554
OUTPUT PHASE REVERSAL
Output phase reversal occurs in some amplifiers when the input
common-mode voltage range is exceeded. As common-mode
voltage moves outside of the common-mode range, the outputs
of these amplifiers suddenly jump in the opposite direction to
the supply rail. This is the result of the differential input pair
shutting down and causing a radical shifting of internal
voltages, resulting in the erratic output behavior.
The AD855x amplifiers have been carefully designed to prevent
any output phase reversal, provided both inputs are maintained
within the supply voltages. If there is the potential of one or
both inputs exceeding either supply voltage, place a resistor in
series with the input to limit the current to less than 2 mA to
ensure the output does not reverse its phase.
CAPACITIVE LOAD DRIVE
The AD855x family has excellent capacitive load driving
capabilities and can safely drive up to 10 nF from a single 5 V
supply. Although the device is stable, capacitive loading limits
the bandwidth of the amplifier. Capacitive loads also increase
the amount of overshoot and ringing at the output. An R-C
snubber network, shown in Figure 61, can be used to
compensate the amplifier against capacitive load ringing and
overshoot.
The optimum value for the resistor and capacitor is a function
of the load capacitance and is best determined empirically because
actual CLOAD (CL) includes stray capacitances and may differ
substantially from the nominal capacitive load. Table 5 shows
some snubber network values that can be used as starting points.
Table 5. Snubber Network Values for Driving Capacitive Loads
CLOAD
1 nF
4.7 nF
10 nF
RX
200 Ω
60 Ω
20 Ω
CX
1 nF
0.47 μF
10 μF
POWER-UP BEHAVIOR
At power-up, the AD855x settles to a valid output within 5 μs.
Figure 63 shows an oscilloscope photo of the output of the
amplifier with the power supply voltage, and Figure 64 shows
the test circuit. With the amplifier configured for unity gain, the
device takes approximately 5 μs to settle to its final output
voltage. This turn-on response time is much faster than most
other autocorrection amplifiers, which can take hundreds of
microseconds or longer for their output to settle.
VOUT
5V
0V
RX
60Ω
CX
0.47µF
VOUT
CL
4.7nF
V+
0V
Although the snubber does not recover the loss of amplifier
bandwidth from the load capacitance, it does allow the
amplifier to drive larger values of capacitance while maintaining
a minimum of overshoot and ringing. Figure 62 shows the
output of an AD855x driving a 1 nF capacitor with and without
a snubber network.
1V
5µs
Figure 61. Snubber Network Configuration for Driving Capacitive Loads
01101-063
AD8551/
AD8552/
AD8554
01101-061
VIN
200mV p-p
BOTTOM TRACE = 2V/DIV
TOP TRACE = 1V/DIV
Figure 63. AD855x Output Behavior on Power-Up
100kΩ
VSY = 0V TO 5V
WITH
SNUBBER
AD8551/
AD8552/
AD8554
01101-064
VOUT
100kΩ
10µs
Figure 64. AD855x Test Circuit for Turn-On Time
VSY = 5V
CLOAD = 4.7nF
100mV
01101-062
WITHOUT
SNUBBER
Figure 62. Overshoot and Ringing are Substantially Reduced
Using a Snubber Network
Rev. C | Page 19 of 24
AD8551/AD8552/AD8554
R2
APPLICATIONS
V2
The extremely low offset voltage of the AD8552 makes it an
ideal amplifier for any application requiring accuracy with high
gains, such as a weigh scale or strain gage. Figure 65 shows a
configuration for a single-supply, precision, strain gage
measurement system.
V1
A REF192 provides a 2.5 V precision reference voltage for A2.
The A2 amplifier boosts this voltage to provide a 4.0 V reference for the top of the strain gage resistor bridge. Q1 provides
the current drive for the 350 Ω bridge network. A1 is used to
amplify the output of the bridge with the full-scale output
voltage equal to
2 × (R1 + R2 )
RB
B
Using the values given in Figure 65, the output voltage linearly
varies from 0 V with no strain to 4.0 V under full strain.
1kΩ
6
A2
R3
=
R2
R1
, THEN VOUT =
R2
R1
× (V1 – V2)
In an ideal difference amplifier, the ratio of the resistors are set
exactly equal to
AV =
R2 R4
=
R1 R3
(19)
Which sets the output voltage of the system to
VOUT = AV (V1 − V2)
(20)
Due to finite component tolerance, the ratio between the four
resistors is not exactly equal, and any mismatch results in a
reduction of common-mode rejection from the system.
Referring to Figure 66, the exact common-mode rejection ratio
can be expressed as
CMRR =
3
R1R4 + 2R2 R4 + R2 R3
2R1R4 − 2R2 R3
(21)
In the three-op amp, instrumentation amplifier configuration
shown in Figure 67, the output difference amplifier is set to
unity gain with all four resistors equal in value. If the tolerance
of the resistors used in the circuit is given as δ, the worst-case
CMRR of the instrumentation amplifier is
20kΩ
40mV
FULL-SCALE
R2
100Ω
A1
AD8552-A
R3
17.4kΩ
VOUT
0V TO 4.0V
CMRRMIN =
R4
100Ω
NOTES
1. USE 0.1% TOLERANCE RESISTORS.
The high common-mode rejection, high open-loop gain, and
operation down to 3 V of supply voltage makes the AD855x an
excellent choice of op amp for discrete single-supply instrumentation amplifiers. The common-mode rejection ratio of the
AD855x is greater than 120 dB, but the CMRR of the system is
also a function of the external resistor tolerances. The gain of
the difference amplifier shown in Figure 66 is given as
⎞
⎟⎟
⎠
R
R
R
R
R
RG
3 V INSTRUMENTATION AMPLIFIER
⎞
⎛R
⎟⎟ − V 2⎜⎜ 2
⎠
⎝ R1
(22)
AD8554-A
V2
Figure 65. A 5 V Precision Strain Gage Amplifier
⎞⎛ R1
⎟⎜1 +
⎟⎜ R
2
⎠⎝
1
2δ
(18)
V1
VOUT
AD8554-B
VOUT = 1 +
R
AD8554-C
RTRIM
2R
(V1 – V2)
RG
01101-067
12.0kΩ
⎛ R4
VOUT = V 1⎜⎜
⎝ R3 + R 4
R4
R4
4
AD8552-B
R1
17.4kΩ
350Ω
LOAD
CELL
REF192
01101-065
4.0V
2.5V
R3
AD8551/
AD8552/
AD8554
Figure 66. Using the AD855x as a Difference Amplifier
2
5V
Q1
2N2222
OR
EQUIVALENT
VOUT
IF
(17)
where RB is the resistance of the load cell.
R1
01101-066
5 V PRECISION STRAIN GAGE CIRCUIT
Figure 67. A Discrete Instrumentation Amplifier Configuration
Consequently, using 1% tolerance resistors results in a worstcase system CMRR of 0.02, or 34 dB. Therefore, either high
precision resistors or an additional trimming resistor, as shown
in Figure 67, should be used to achieve high common-mode
rejection. The value of this trimming resistor should be equal
to the value of R multiplied by its tolerance. For example, using
10 kΩ resistors with 1% tolerance requires a series trimming
resistor equal to 100 Ω.
Rev. C | Page 20 of 24
AD8551/AD8552/AD8554
⎛R
Monitor Output = R2 × ⎜⎜ SENSE
⎝ R1
HIGH ACCURACY THERMOCOUPLE AMPLIFIER
Figure 68 shows a K-type thermocouple amplifier configuration
with cold junction compensation. Even from a 5 V supply, the
AD8551 can provide enough accuracy to achieve a resolution of
better than 0.02°C from 0°C to 500°C. D1 is used as a temperature measuring device to correct the cold junction error from
the thermocouple and should be placed as close as possible to
the two terminating junctions. With the thermocouple measuring
tip immersed in a 0°C ice bath, R6 should be adjusted until the
output is at 0 V.
4
R5
40.2kΩ
IL
V+
3V
R8
124kΩ
5V
D1
R1
100Ω
10µF
+
3
0.1µF
R2
2.74kΩ
R7
453Ω
R6
200Ω
R4
5.62kΩ
RSENSE
0.1Ω
3V
1N4148
K-TYPE
THERMOCOUPLE
40.7µV/°C
For the component values shown in Figure 70, the output
transfer function decreases from V+ at −2.5 V/A.
5.000V
R1
10.7kΩ
R3
53.6Ω
(24)
2
0.1µF
8
1/2
AD8552
1
4
2
3
–
8
AD8551
+
4
M1
Si9433
1
0V TO 5.00V
(0°C TO 500°C)
S
MONITOR
OUTPUT
G
D
R2
2.49kΩ
01101-069
REF02EZ 6
Figure 70 shows the low-side monitor equivalent. In this circuit,
the input common-mode voltage to the AD8552 is at or near
ground. Again, a 0.1 Ω resistor provides a voltage drop proportional to the return current. The output voltage is given as
Figure 69. A High-Side Load Current Monitor
Figure 68. A Precision K-Type Thermocouple Amplifier with
Cold Junction Compensation
V+
PRECISION CURRENT METER
R2
2.49kΩ
Because of its low input bias current and superb offset voltage at
single supply voltages, the AD855x is an excellent amplifier for
precision current monitoring. Its rail-to-rail input allows the
amplifier to be used as either a high-side or low-side current
monitor. Using both amplifiers in the AD8552 provides a simple
method to monitor both current supply and return paths for
load or fault detection.
Figure 69 shows a high-side current monitor configuration. In
this configuration, the input common-mode voltage of the
amplifier is at or near the positive supply voltage. The rail-torail input of the amplifier provides a precise measurement even
with the input common-mode voltage at the supply voltage. The
CMOS input structure does not draw any input bias current,
ensuring a minimum of measurement error.
The 0.1 Ω resistor creates a voltage drop to the noninverting
input of the AD855x. The output of the amplifier is corrected
until this voltage appears at the inverting input. This creates a
current through R1, which in turn flows through R2. The
monitor output is given by
VOUT
Q1
V+
R1
100Ω
RSENSE
0.1Ω
1/2 AD8552
RETURN TO
GROUND
01101-070
2
Using the components shown in Figure 69, the monitor output
transfer function is 2.5 V/A.
01101-068
0.1µF
(23)
⎛R
⎞
VOUT = (V + ) − ⎜⎜ 2 × RSENSE × I L ⎟⎟
⎝ R1
⎠
Using the values shown in Figure 68, the output voltage tracks
temperature at 10 mV/°C. For a wider range of temperature
measurement, R9 can be decreased to 62 kΩ. This creates a
5 mV/°C change at the output, allowing measurements of up
to 1000°C.
12V
⎞
⎟ × IL
⎟
⎠
Figure 70. A Low-Side Load Current Monitor
PRECISION VOLTAGE COMPARATOR
The AD855x can be operated open-loop and used as a precision
comparator. The AD855x has less than 50 μV of offset voltage
when run in this configuration. The slight increase of offset
voltage stems from the fact that the autocorrection architecture
operates with lowest offset in a closed-loop configuration, that
is, one with negative feedback. With 50 mV of overdrive, the
device has a propagation delay of 15 μs on the rising edge and
8 μs on the falling edge. Ensure the maximum differential
voltage of the device is not exceeded. For more information,
refer to the Input Overvoltage Protection section.
Rev. C | Page 21 of 24
AD8551/AD8552/AD8554
OUTLINE DIMENSIONS
5.00 (0.1968)
4.80 (0.1890)
3.20
3.00
2.80
8
1
1
5
4
6.20 (0.2440)
5.80 (0.2284)
4
1.27 (0.0500)
BSC
PIN 1
0.65 BSC
0.38
0.22
0.80
0.60
0.40
8°
0°
0.23
0.08
0.51 (0.0201)
0.31 (0.0122)
COPLANARITY
0.10
SEATING
PLANE
1.10 MAX
COPLANARITY
0.10
1.75 (0.0688)
1.35 (0.0532)
0.25 (0.0098)
0.10 (0.0040)
0.95
0.85
0.75
0.15
0.00
8
4.00 (0.1574)
3.80 (0.1497)
5.15
4.90
4.65
5
0.50 (0.0196)
0.25 (0.0099)
45°
8°
0°
0.25 (0.0098)
0.17 (0.0067)
1.27 (0.0500)
0.40 (0.0157)
COMPLIANT TO JEDEC STANDARDS MS-012-A A
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN.
SEATING
PLANE
060506-A
3.20
3.00
2.80
COMPLIANT TO JEDEC STANDARDS MO-187-AA
Figure 73. 8-Lead Standard Small Outline Package [SOIC_N]
Narrow Body (R-8)
Dimensions shown in millimeters and (inches)
Figure 71. 8-Lead Mini Small Outline Package [MSOP]
(RM-8)
Dimensions shown in millimeters
5.10
5.00
4.90
3.10
3.00
2.90
8
5
14
4.50
4.40
4.30
1
8
4.50
4.40
4.30
6.40 BSC
4
6.40
BSC
1
PIN 1
7
PIN 1
0.65 BSC
0.15
0.05
1.20
MAX
COPLANARITY
0.10
0.30
0.19
0.65
BSC
1.05
1.00
0.80
SEATING 0.20
PLANE
0.09
8°
0°
1.20
MAX
0.15
0.05
0.75
0.60
0.45
0.30
0.19
0.20
0.09
SEATING
COPLANARITY
PLANE
0.10
8°
0°
COMPLIANT TO JEDEC STANDARDS MO-153-AA
COMPLIANT TO JEDEC STANDARDS MO-153-AB-1
Figure 72. 8-Lead Thin Shrink Small Outline Package [TSSOP]
(RU-8)
Dimensions shown in millimeters
Figure 74. 14-Lead Thin Shrink Small Outline Package [TSSOP]
(RU-14)
Dimensions shown in millimeters
8.75 (0.3445)
8.55 (0.3366)
8
14
1
7
1.27 (0.0500)
BSC
0.25 (0.0098)
0.10 (0.0039)
COPLANARITY
0.10
0.51 (0.0201)
0.31 (0.0122)
6.20 (0.2441)
5.80 (0.2283)
0.50 (0.0197)
0.25 (0.0098)
1.75 (0.0689)
1.35 (0.0531)
SEATING
PLANE
45°
8°
0°
0.25 (0.0098)
0.17 (0.0067)
1.27 (0.0500)
0.40 (0.0157)
COMPLIANT TO JEDEC STANDARDS MS-012-AB
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN.
Figure 75. 14-Lead Standard Small Outline Package [SOIC_N]
Narrow Body (R-14)
Dimensions shown in millimeters and (inches)
Rev. C | Page 22 of 24
060606-A
4.00 (0.1575)
3.80 (0.1496)
0.75
0.60
0.45
AD8551/AD8552/AD8554
ORDERING GUIDE
Model
AD8551AR
AD8551AR-REEL
AD8551AR-REEL7
AD8551ARZ 1
AD8551ARZ-REEL1
AD8551ARZ-REEL71
AD8551ARM-R2
AD8551ARM-REEL
AD8551ARMZ-R21
AD8551ARMZ-REEL1
AD8552AR
AD8552AR-REEL
AD8552AR-REEL7
AD8552ARZ1
AD8552ARZ-REEL1
AD8552ARZ-REEL71
AD8552ARU
AD8552ARU-REEL
AD8552ARUZ1
AD8552ARUZ-REEL1
AD8554AR
AD8554AR-REEL
AD8554AR-REEL7
AD8554ARZ1
AD8554ARZ-REEL1
AD8554ARZ-REEL71
AD8554ARU
AD8554ARU-REEL
AD8554ARUZ1
AD8554ARUZ-REEL1
1
Temperature Range
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
Package Description
8-Lead SOIC_N
8-Lead SOIC_N
8-Lead SOIC_N
8-Lead SOIC_N
8-Lead SOIC_N
8-Lead SOIC_N
8-Lead MSOP
8-Lead MSOP
8-Lead MSOP
8-Lead MSOP
8-Lead SOIC_N
8-Lead SOIC_N
8-Lead SOIC_N
8-Lead SOIC_N
8-Lead SOIC_N
8-Lead SOIC_N
8-Lead TSSOP
8-Lead TSSOP
8-Lead TSSOP
8-Lead TSSOP
14-Lead SOIC_N
14-Lead SOIC_N
14-Lead SOIC_N
14-Lead SOIC_N
14-Lead SOIC_N
14-Lead SOIC_N
14-Lead TSSOP
14-Lead TSSOP
14-Lead TSSOP
14-Lead TSSOP
Z = RoHS Compliant Part, # denotes RoHS compliant part may be top or bottom marked.
Rev. C | Page 23 of 24
Package Option
R-8
R-8
R-8
R-8
R-8
R-8
RM-8
RM-8
RM-8
RM-8
R-8
R-8
R-8
R-8
R-8
R-8
RU-8
RU-8
RU-8
RU-8
R-14
R-14
R-14
R-14
R-14
R-14
RU-14
RU-14
RU-14
RU-14
Branding
AHA
AHA
AHA#
AHA#
AD8551/AD8552/AD8554
NOTES
©1999–2007 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
C01101-0-3/07(C)
Rev. C | Page 24 of 24