TI1 LMR23610AQDDARQ1 Simple switcher 36 v, 1 a synchronous step-down converter Datasheet

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LMR23610-Q1
SNVSAR4 – DECEMBER 2016
LMR23610-Q1 SIMPLE SWITCHER® 36 V, 1 A Synchronous Step-Down Converter
1 Features
3 Description
•
•
The LMR23610-Q1 SIMPLE SWITCHER® is an easy
to use 36 V, 1 A synchronous step down regulator.
With a wide input range from 4 V to 36 V, it is
suitable for various applications from industrial to
automotive for power conditioning from unregulated
sources. Peak current mode control is employed to
achieve simple control loop compensation and cycleby-cycle current limiting. A quiescent current of 75 µA
makes it suitable for battery powered systems.
Internal loop compensation means that the user is
free from the tedious task of loop compensation
design. This also minimizes the external components.
An extended family is available in 1.5 A (LMR23615Q1), 2.5 A (LMR23625-Q1) and 3 A (LMR23630-Q1)
load current options in pin-to-pin compatible
packages which allows simple, optimum PCB layout.
A precision enable input allows simplification of
regulator control and system power sequencing.
Protection features include cycle-by-cycle current
limit, hiccup mode short circuit protection and thermal
shutdown due to excessive power dissipation.
1
•
•
•
•
•
•
•
•
•
•
•
•
•
Qualified for Automotive Applications
AEC-Q100 Qualified With the Following Results:
- Device Temperature Grade 1: -40 °C to 125 °C
Ambient Operating Temperature Range
- Device HBM ESD Classification Level H1C
- Device CDM ESD Classification Level C4A
4 V to 36 V Input Range
1 A Continuous Output Current
Integrated Synchronous Rectification
Current Mode Control
Minimum Switch-On Time: 60 ns
Internal Compensation for Ease of Use
400 kHz Switching Frequency With PFM Mode
Frequency Synchronization to External Clock
75 µA Quiescent Current at No Load
Soft-Start into a Pre-Biased Load
High Duty Cycle Operation Supported
Output Short-Circuit Protection with Hiccup Mode
8-Pin HSOIC with PowerPAD™ Package Options
2 Applications
•
•
•
•
Device Information(1)
PART NUMBER
PACKAGE
BODY SIZE (NOM)
LMR23610AQDDARQ1
HSOIC (8)
4.9 mm x 3.9 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
Automotive Battery Regulation
Industrial Power Supplies
Telecom and Datacom Systems
General Purpose Wide Vin Regulation
space
Simplified Schematic
Efficiency vs Load, VIN = 12 V
VIN up to 36 V
100
CIN
VIN
90
BOOT
CBOOT
AGND
L
VOUT
SW
RFBT
COUT
VCC
FB
CVCC
RFBB
Efficiency (%)
EN/SYNC
80
70
60
PGND
50
VOUT = 5 V
VOUT = 3.3 V
40
0.0001
0.001
0.01
IOUT (A)
0.1
1
D000
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
LMR23610-Q1
SNVSAR4 – DECEMBER 2016
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
4
6.1
6.2
6.3
6.4
6.5
6.6
6.7
6.8
4
4
4
5
5
6
7
8
Absolute Maximum Ratings ......................................
ESD Ratings..............................................................
Recommended Operating Conditions ......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Timing Characteristics...............................................
Switching Characteristics ..........................................
Typical Characteristics ..............................................
7.4 Device Functional Modes........................................ 17
8
Application and Implementation ........................ 18
8.1 Application Information............................................ 18
8.2 Typical Applications ................................................ 18
9 Power Supply Recommendations...................... 24
10 Layout................................................................... 24
10.1
10.2
10.3
10.4
10.5
Layout Guidelines .................................................
Compact Layout for EMI Reduction ......................
Ground Plane and Thermal Considerations..........
Feedback Resistors ..............................................
Layout Example ....................................................
24
24
25
25
26
11 Device and Documentation Support ................. 27
11.1
11.2
11.3
11.4
11.5
Detailed Description ............................................ 10
7.1 Overview ................................................................. 10
7.2 Functional Block Diagram ....................................... 10
7.3 Feature Description................................................. 11
Receiving Notification of Documentation Updates
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
27
27
27
27
27
12 Mechanical, Packaging, and Orderable
Information ........................................................... 28
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
2
DATE
REVISION
NOTES
December 2016
*
Initial release.
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5 Pin Configuration and Functions
DDA Package
8-Pin HSOIC
Top View
SW
1
BOOT
2
VCC
FB
8
PGND
7
VIN
3
6
AGND
4
5
EN/SYNC
Thermal Pad
(9)
Pin Functions
PIN
NAME
NO.
I/O
(1)
DESCRIPTION
SW
1
P
Switching output of the regulator. Internally connected to both power MOSFETs. Connect to
power inductor.
BOOT
2
P
Boot-strap capacitor connection for high-side driver. Connect a high quality 100 nF capacitor
from BOOT to SW.
VCC
3
P
Internal bias supply output for bypassing. Connect a 2.2 μF/ 16 V or higher capacitance
bypass capacitor from this pin to AGND. Do not connect external loading to this pin. Never
short this pin to ground during operation.
FB
4
A
Feedback input to regulator, connect the feedback resistor divider tap to this pin.
EN/SYNC
5
A
Enable input to regulator. High = On, Low = Off. Can be connected to VIN. Do not float.
Adjust the input under voltage lockout with two resistors. The internal oscillator can be
synchronized to an external clock by coupling a positive pulse into this pin through a small
coupling capacitor. See Enable/Sync for detail.
AGND
6
G
Analog ground pin. Ground reference for internal references and logic. Connect to system
ground.
VIN
7
P
Input supply voltage.
PGND
8
G
Power ground pin, connected internally to the low side power FET. Connect to system
ground, PAD, AGND, ground pins of CIN and COUT. Path to CIN must be as short as possible.
PAD
9
G
Low impedance connection to AGND. Connect to PGND on PCB. Major heat dissipation
path of the die. Must be used for heat sinking to ground plane on PCB.
(1)
A = Analog, P = Power, G = Ground
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6 Specifications
6.1 Absolute Maximum Ratings
Over the recommended operating junction temperature range of -40 °C to 125 °C (unless otherwise noted)
PARAMETER
(1)
MIN
MAX
VIN to PGND
-0.3
42
EN/SYNC to AGND
-5.5
VIN + 0.3
FB to AGND
-0.3
4.5
AGND to PGND
-0.3
0.3
SW to PGND
-1
VIN + 0.3
SW to PGND less than 10 ns transients
-5
42
BOOT to SW
-0.3
5.5
VCC to AGND
-0.3
TJ
Junction temperature
-40
150
°C
Tstg
Storage temperature
-65
150
°C
Input Voltages
Output Voltages
(1)
(2)
4.5
UNIT
V
V
(2)
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
In shutdown mode, the VCC to AGND maximum value is 5.25 V.
6.2 ESD Ratings
VALUE
V(ESD)
(1)
Electrostatic discharge
(1)
±2500
Charged-device model (CDM)
±1000
Human-body model (HBM)
UNIT
V
AEC Q100-002 indicates that HBM stressing shall be in accordance with the ANSI/ESDA/JEDEC JS-001 specification.
6.3 Recommended Operating Conditions
Over the recommended operating junction temperature range of -40 °C to 125 °C (unless otherwise noted)
VIN
Input Voltage
EN/SYNC
FB
MIN
MAX
4
36
-5
36
-0.3
1.2
(1)
UNIT
V
Output Voltage
VOUT
1
28
Output Current
IOUT
0
1
A
Temperature
Operating junction temperature, TJ
-40
125
°C
(1)
4
V
Operating Ratings indicate conditions for which the device is intended to be functional, but do not guarantee specific performance limits.
For guaranteed specifications, see Electrical Characteristics.
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6.4 Thermal Information
THERMAL METRIC
(1) (2)
DDA (8 PINS)
RθJA
Junction-to-ambient thermal resistance
ψJT
Junction-to-top characterization parameter
5.9
ψJB
Junction-to-board characterization parameter
23.4
RθJC(top)
Junction-to-case (top) thermal resistance
45.8
RθJC(bot)
Junction-to-case (bottom) thermal resistance
3.6
RθJB
Junction-to-board thermal resistance
23.4
(1)
(2)
UNIT
42.0
°C/W
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report, SPRA953.
Power rating at a specific ambient temperature TA should be determined with a maximum junction temperature (TJ) of 125 °C, which is
illustrated in Recommended Operating Conditions section.
6.5 Electrical Characteristics
Limits apply over the recommended operating junction temperature (TJ) range of -40 °C to +125 °C, unless otherwise stated.
Minimum and Maximum limits are specified through test, design or statistical correlation. Typical values represent the most
likely parametric norm at TJ = 25 °C, and are provided for reference purposes only.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
Rising threshold
36
V
3.3
3.7
3.9
Falling threshold
2.9
3.3
3.5
4.0
POWER SUPPLY (VIN PIN)
VIN
Operation input voltage
VIN_UVLO
Under voltage lockout thresholds
4
ISHDN
Shutdown supply current
VEN = 0 V, VIN = 12 V, TJ = -40 °C to 125
°C
2.0
IQ
Operating quiescent current (nonswitching)
VIN =12 V, VFB = 1.1 V, TJ = -40 °C to
125 °C, PFM mode
75
V
μA
μA
ENABLE (EN PIN)
VEN_H
Enable rising threshold Voltage
VEN_HYS
Enable hysteresis voltage
VWAKE
Wake-up threshold
IEN
1.4
1.55
1.7
0.4
V
0.4
VIN = 4 V to 36 V, VEN= 2 V
Input leakage current at EN pin
V
V
10
VIN = 4 V to 36 V, VEN= 36 V
100
nA
1
μA
VOLTAGE REFERENCE (FB PIN)
VREF
Reference voltage
ILKG_FB
Input leakage current at FB pin
VIN = 4 V to 36 V, TJ = 25 °C
0.985
1.0
1.015
VIN = 4 V to 36 V, TJ = -40 °C to 125 °C
0.980
1.0
1.020
VFB= 1 V
V
10
nA
4.1
V
INTERNAL LDO (VCC PIN)
VCC
Internal LDO output voltage
VCC_UVLO
VCC under voltage lockout thresholds
Rising threshold
2.8
3.2
3.6
Falling threshold
2.4
2.8
3.2
V
CURRENT LIMIT
IHS_LIMIT
Peak inductor current limit
1.4
2.0
2.6
A
ILS_LIMIT
Valley inductor current limit
1.0
1.5
2.1
A
IL_ZC
Zero cross current limit
-0.04
A
INTEGRATED MOSFETS
RDS_ON_HS
High-side MOSFET ON-resistance
VIN = 12 V, IOUT = 1 A
185
mΩ
RDS_ON_LS
Low-side MOSFET ON-resistance
VIN = 12 V, IOUT = 1 A
105
mΩ
THERMAL SHUTDOWN
TSHDN
Thermal shutdown threshold
THYS
Hysteresis
162
170
178
15
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°C
5
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6.6 Timing Characteristics
Over the recommended operating junction temperature range of -40 °C to 125 °C (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
HICCUP MODE
NOC (1)
Number of cycles that LS current limit is
tripped to enter Hiccup mode
64
TOC
Hiccup retry delay time
5
Cycles
ms
SOFT START
TSS
(1)
6
Internal soft-start time
The time of internal reference to increase
from 0 V to 1.0 V
1
2
3
ms
Guaranteed by design.
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6.7 Switching Characteristics
Over the recommended operating junction temperature range of -40 °C to 125 °C (unless otherwise noted)
PARAMETER
MIN
TYP
MAX
UNIT
340
SW (SW PIN)
fSW
Default switching frequency
400
460
kHz
TON_MIN
Minimum turn-on time
60
90
ns
TOFF_MIN (1)
Minimum turn-off time
100
ns
SYNC (EN/SYNC PIN)
fSYNC
SYNC frequency range
200
2200
kHz
VSYNC
Amplitude of SYNC clock AC signal (measured at SYNC pin)
2.8
5.5
V
TSYNC_MIN
Minimum sync clock ON and OFF time
(1)
100
ns
Guaranteed by design.
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6.8 Typical Characteristics
100
100
90
90
80
80
70
70
Efficiency (%)
Efficiency (%)
Unless otherwise specified the following conditions apply: VIN = 12 V, fSW = 400 kHz, L = 22 µH, COUT = 47 µF × 2, TA = 25°C.
60
50
40
60
50
40
30
30
VIN = 8 V
VIN = 12 V
VIN = 24 V
VIN = 36 V
20
10
0
1E-5
0.0001
0.001
0.01
IOUT (A)
fSW = 400 kHz
VIN = 8 V
VIN = 12 V
VIN = 24 V
VIN = 36 V
20
10
0.1
0
1E-5
1
0.0001
fSW = 400 kHz
VOUT = 5 V
100
100
90
90
80
80
70
70
Efficiency (%)
Efficiency (%)
0.1 0.2 0.5 1
D002
VOUT = 3.3 V
Figure 2. Efficiency vs. Load Current
Figure 1. Efficiency vs. Load Current
60
50
40
30
60
50
40
30
VIN = 8 V
VIN = 12 V
VIN = 24 V
VIN = 36 V
20
10
0
1E-5
0.0001
0.001
0.01
IOUT (A)
fSW = 1000 kHz
(Sync)
VIN = 8 V
VIN = 12 V
VIN = 24 V
VIN = 36 V
20
10
0.1
0
1E-5
1
0.0001
0.001
0.01
IOUT (A)
D003
VOUT = 5 V
fSW = 1000 kHz
(Sync)
Figure 3. Efficiency vs. Load Current
0.1
1
D004
VOUT = 3.3 V
Figure 4. Efficiency vs. Load Current
5.02
5.09
IOUT = 0.3 A
IOUT = 0.5 A
IOUT = 0.8 A
IOUT = 1.0 A
VIN = 8 V
VIN = 12 V
VIN = 24 V
VIN = 36 V
5.08
5.07
5.015
VOUT (V)
5.06
VOUT (V)
0.001
0.01
IOUT (A)
D001
5.05
5.04
5.01
5.03
5.02
5.01
5.005
5
0
0.2
0.4
0.6
IOUT (A)
0.8
1
0
5
D005
15
20
VIN (V)
25
30
35
40
D006
VOUT = 5 V
VOUT = 5 V
Figure 6. Line Regulation
Figure 5. Load Regulation
8
10
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Typical Characteristics (continued)
Unless otherwise specified the following conditions apply: VIN = 12 V, fSW = 400 kHz, L = 22 µH, COUT = 47 µF × 2, TA = 25°C.
3.6
5.5
5
VOUT (V)
VOUT (V)
3.3
4.5
4
3
3.5
IOUT = 0.2 A
IOUT = 0.5 A
IOUT = 1.0 A
IOUT = 0.2 A
IOUT = 0.5 A
IOUT = 1.0 A
2.7
3.3
3
4
4.5
5
VIN (V)
5.5
6
3.4
3.5
D007
VOUT = 5 V
3.7
3.8
Figure 7. Dropout Curve
D008
Figure 8. Dropout Curve
3.67
VIN UVLO Rising Threshold (V)
75
70
65
60
-50
0
50
Temperature (°C)
VIN = 12 V
100
3.66
3.65
3.64
3.63
3.62
3.61
-50
150
0
D008
50
Temperature (°C)
100
150
D009
VFB = 1.1 V
Figure 9. IQ vs. Junction Temperature
Figure 10. VIN UVLO Rising Threshold vs. Junction
Temperature
0.425
2.4
LS Limit
HS Limit
Current Limit (A)
2.2
VIN UVLO Hysteresis (V)
3.9
VOUT = 3.3 V
80
IQ (µA)
3.6
VIN (V)
0.42
0.415
2
1.8
1.6
1.4
0.41
-50
0
50
Temperature (°C)
100
150
1.2
-50
D010
0
50
Temperature (qC)
100
150
D009
VIN = 12 V
Figure 11. VIN UVLO Hysteresis vs. Junction Temperature
Figure 12. HS & LS Current Limit vs. Junction Temperature
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7 Detailed Description
7.1 Overview
The LMR23610-Q1 SIMPLE SWITCHER® regulator is an easy to use synchronous step-down DC-DC converter
operating from 4 V to 36 V supply voltage. It is capable of delivering up to 1 A DC load current with good thermal
performance in a small solution size. An extended family is available in multiple current options from 1 A to 3 A in
pin-to-pin compatible packages.
The LMR23610-Q1 employs fixed frequency peak current mode control. The device enters PFM mode at light
load to achieve high efficiency. The device is internally compensated, which reduces design time, and requires
few external components. The LMR23610-Q1 is capable of synchronization to an external clock within the range
of 200 kHz to 2.2 MHz.
Additional features such as precision enable and internal soft-start provide a flexible and easy to use solution for
a wide range of applications. Protection features include thermal shutdown, VIN and VCC under-voltage lockout,
cycle-by-cycle current limit, and hiccup mode short-circuit protection.
The family requires very few external components and has a pin-out designed for simple, optimum PCB layout.
7.2 Functional Block Diagram
EN/SYNC
SYNC Signal
SYNC
Detector
VCC
VCC
Enable
LDO
VIN
Precision
Enable
Internal
SS
BOOT
HS I Sense
EA
REF
Rc
TSD
UVLO
Cc
PWM CONTROL LOGIC
PFM
Detector
OV/UV
Detector
SW
FB
Slope
Comp
Freq
Foldback
Zero
Cross
HICCUP
Detector
SYNC Signal
Oscillator
LS I Sense
FB
PGND
AGND
10
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7.3 Feature Description
7.3.1 Fixed Frequency Peak Current Mode Control
The following operating description of the LMR23610-Q1 will refer to the Functional Block Diagram and to the
waveforms in Figure 13. LMR23610-Q1 is a step-down synchronous buck regulator with integrated high-side
(HS) and low-side (LS) switches (synchronous rectifier). The LMR23610-Q1 supplies a regulated output voltage
by turning on the HS and LS NMOS switches with controlled duty cycle. During high-side switch ON time, the
SW pin voltage swings up to approximately VIN, and the inductor current iL increase with linear slope (VIN – VOUT)
/ L. When the HS switch is turned off by the control logic, the LS switch is turned on after an anti-shoot-through
dead time. Inductor current discharges through the LS switch with a slope of –VOUT / L. The control parameter of
a buck converter is defined as Duty Cycle D = tON / TSW, where tON is the high-side switch ON time and TSW is
the switching period. The regulator control loop maintains a constant output voltage by adjusting the duty cycle
D. In an ideal buck converter, where losses are ignored, D is proportional to the output voltage and inversely
proportional to the input voltage: D = VOUT / VIN.
VSW
SW Voltage
D = tON/ TSW
VIN
tON
tOFF
t
0
-VD
Inductor Current
iL
TSW
ILPK
IOUT
'iL
t
0
Figure 13. SW Node and Inductor Current Waveforms in
Continuous Conduction Mode (CCM)
The LMR23610-Q1 employs fixed frequency peak current mode control. A voltage feedback loop is used to get
accurate DC voltage regulation by adjusting the peak current command based on voltage offset. The peak
inductor current is sensed from the high-side switch and compared to the peak current threshold to control the
ON time of the high-side switch. The voltage feedback loop is internally compensated, which allows for fewer
external components, makes it easy to design, and provides stable operation with almost any combination of
output capacitors. The regulator operates with fixed switching frequency at normal load condition. At light load
condition, the LMR23610-Q1 will operate in PFM mode to maintain high efficiency.
7.3.2 Adjustable Output Voltage
A precision 1.0 V reference voltage is used to maintain a tightly regulated output voltage over the entire
operating temperature range. The output voltage is set by a resistor divider from output voltage to the FB pin. It
is recommended to use 1% tolerance resistors with a low temperature coefficient for the FB divider. Select the
low-side resistor RFBB for the desired divider current and use Equation 1 to calculate high-side RFBT. RFBT in the
range from 10 kΩ to 100 kΩ is recommended for most applications. A lower RFBT value can be used if static
loading is desired to reduce VOUT offset in PFM operation. Lower RFBT will reduce efficiency at very light load.
Less static current goes through a larger RFBT and might be more desirable when light load efficiency is critical.
But RFBT larger than 1 MΩ is not recommended because it makes the feedback path more susceptible to noise.
Larger RFBT value requires more carefully designed feedback path on the PCB. The tolerance and temperature
variation of the resistor dividers affect the output voltage regulation.
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Feature Description (continued)
VOUT
RFBT
FB
RFBB
Figure 14. Output Voltage Setting
RFBT
VOUT VREF
u RFBB
VREF
(1)
7.3.3 Enable/Sync
The voltage on the EN pin controls the ON or OFF operation of LMR23610-Q1. A voltage less than 1 V (typ) will
shut-down the device while a voltage higher than 1.6 V (typ) is required to start the regulator. The EN pin is an
input and can not be left open or floating. The simplest way to enable the operation of the LMR23610-Q1 is to
connect the EN to VIN. This allows self-start-up of the LMR23610-Q1 when VIN is within the operation range.
Many applications will benefit from the employment of an enable divider RENT and RENB (Figure 15) to establish a
precision system UVLO level for the converter. System UVLO can be used for supplies operating from utility
power as well as battery power. It can be used for sequencing, ensuring reliable operation, or supply protection,
such as a battery discharge level. An external logic signal can also be used to drive EN input for system
sequencing and protection.
VIN
RENT
EN/SYNC
RENB
Figure 15. System UVLO by Enable Divider
The EN pin also can be used to synchronize the internal oscillator to an external clock. The internal oscillator can
be synchronized by AC coupling a positive edge into the EN pin. The AC coupled peak-to-peak voltage at the EN
pin must exceed the SYNC amplitude threshold of 2.8 V (typ) to trip the internal synchronization pulse detector,
and the minimum SYNC clock ON and OFF time must be longer than 100ns (typ). A 3.3 V or a higher amplitude
pulse signal coupled through a 1 nF capacitor CSYNC is a good starting point. Keeping RENT // RENB (RENT parallel
with RENB) in the 100 kΩ range is a good choice. RENT is required for this synchronization circuit, but RENB can be
left unmounted if system UVLO is not needed. LMR23610-Q1 switching action can be synchronized to an
external clock from 200 kHz to 2.2 MHz. Figure 17 and Figure 18 show the device synchronized to an external
system clock.
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Feature Description (continued)
VIN
CSYNC
RENT
EN/SYNC
RENB
Clock
Source
Figure 16. Synchronize to external clock
Figure 17. Synchronizing in PWM Mode
Figure 18. Synchronizing in PFM Mode
7.3.4 VCC, UVLO
The LMR23610-Q1 integrates an internal LDO to generate VCC for control circuitry and MOSFET drivers. The
nominal voltage for VCC is 4.1 V. The VCC pin is the output of an LDO and must be properly bypassed. A high
quality ceramic capacitor with a value of 2.2 µF to 10 µF, 16 V or higher rated voltage should be placed as close
as possible to VCC and grounded to the exposed PAD and ground pins. The VCC output pin should not be
loaded, or shorted to ground during operation. Shorting VCC to ground during operation may cause damage to
the LMR23610-Q1.
VCC under voltage lockout (UVLO) prevents the LMR23610-Q1 from operating until the VCC voltage exceeds 3.2
V (typ). The VCC UVLO threshold has 400 mV (typ) of hysteresis to prevent undesired shutdown due to
temporary VIN drops.
7.3.5 Minimum ON-time, Minimum OFF-time and Frequency Foldback at Drop-out Conditions
Minimum ON-time, TON_MIN, is the smallest duration of time that the HS switch can be on. TON_MIN is typically 60
ns in the LMR23610-Q1. Minimum OFF-time, TOFF_MIN, is the smallest duration that the HS switch can be off.
TOFF_MIN is typically 100 ns in the LMR23610-Q1. In CCM operation, TON_MIN and TOFF_MIN limit the voltage
conversion range given a selected switching frequency.
The minimum duty cycle allowed is:
DMIN = TON_MIN x fSW
(2)
And the maximum duty cycle allowed is:
DMAX = 1 - TOFF_MIN x fSW
(3)
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Feature Description (continued)
Given fixed TON_MIN and TOFF_MIN, the higher the switching frequency the narrower the range of the allowed duty
cycle. In the LMR23610-Q1, a frequency foldback scheme is employed to extend the maximum duty cycle when
TOFF_MIN is reached. The switching frequency will decrease once longer duty cycle is needed under low VIN
conditions. Wide range of frequency foldback allows the LMR23610-Q1 output voltage stay in regulation with a
much lower supply voltage VIN. This leads to a lower effective drop-out voltage.
Given an output voltage, the choice of the switching frequency affects the allowed input voltage range, solution
size and efficiency. The maximum operation supply voltage can be found by:
VOUT
VIN _ MAX
fSW u TON _ MIN
(4)
At lower supply voltage, the switching frequency will decrease once TOFF_MIN is tripped. The minimum VIN without
frequency foldback can be approximated by:
VOUT
VIN _ MIN
1 fSW u TOFF _ MIN
(5)
Taking considerations of power losses in the system with heavy load operation, VIN_MAX is higher than the result
calculated in Equation 4. With frequency foldback, VIN_MIN is lowered by decreased fSW.
450
400
Frequency (kHz)
350
300
250
200
150
IOUT = 0.4 A
IOUT = 0.6 A
IOUT = 0.8 A
IOUT = 1.0 A
100
50
0
5
5.1
5.2
5.3
5.4
VIN (V)
5.5
5.6
5.7
5.8
D010
Figure 19. Frequency Foldback at Dropout (VOUT = 5 V, fSW = 400 kHz)
7.3.6 Internal Compensation and CFF
The LMR23610-Q1 is internally compensated as shown in Functional Block Diagram. The internal compensation
is designed such that the loop response is stable over the entire operating frequency and output voltage range.
Depending on the output voltage, the compensation loop phase margin can be low with all ceramic capacitors.
An external feed-forward capacitor CFF is recommended to be placed in parallel with the top resistor divider RFBT
for optimum transient performance.
VOUT
RFBT
CFF
FB
RFBB
Figure 20. Feedforward Capacitor for Loop Compensation
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Feature Description (continued)
The feed-forward capacitor CFF in parallel with RFBT places an additional zero before the cross over frequency of
the control loop to boost phase margin. The zero frequency can be found by
1
fZ _ CFF
2S u CFF u RFBT
(6)
An additional pole is also introduced with CFF at the frequency of
1
fP _ CFF
2S u CFF u RFBT //RFBB
(7)
The zero fZ_CFF adds phase boost at the crossover frequency and improves transient response. The pole fP-CFF
helps maintaining proper gain margin at frequency beyond the crossover. Table 1 lists the combination of COUT,
CFF and RFBT for typical applications, designs with similar COUT but RFBT other than recommended value, please
adjust CFF such that (CFF × RFBT) is unchanged and adjust RFBB such that (RFBT / RFBB) is unchanged.
Designs with different combinations of output capacitors need different CFF. Different types of capacitors have
different Equivalent Series Resistance (ESR). Ceramic capacitors have the smallest ESR and need the most
CFF. Electrolytic capacitors have much larger ESR and the ESR zero frequency
1
fZ _ESR
2S u COUT u ESR
(8)
would be low enough to boost the phase up around the crossover frequency. Designs using mostly electrolytic
capacitors at the output may not need any CFF.
The CFF creates a time constant with RFBT that couples in the attenuate output voltage ripple to the FB node. If
the CFF value is too large, it can couple too much ripple to the FB and affect VOUT regulation. Therefore, CFF
should be calculated based on output capacitors used in the system. At cold temperatures, the value of CFF
might change based on the tolerance of the chosen component. This may reduce its impedance and ease noise
coupling on the FB node. To avoid this, more capacitance can be added to the output or the value of CFF can be
reduced.
7.3.7 Bootstrap Voltage (BOOT)
The LMR23610-Q1 provides an integrated bootstrap voltage regulator. A small capacitor between the BOOT and
SW pins provides the gate drive voltage for the high-side MOSFET. The BOOT capacitor is refreshed when the
high-side MOSFET is off and the low-side switch conducts. The recommended value of the BOOT capacitor is
0.1 μF. A ceramic capacitor with an X7R or X5R grade dielectric with a voltage rating of 16V or higher is
recommended for stable performance over temperature and voltage.
7.3.8 Over Current and Short Circuit Protection
The LMR23610-Q1 is protected from over-current conditions by cycle-by-cycle current limit on both the peak and
valley of the inductor current. Hiccup mode will be activated if a fault condition persists to prevent over-heating.
High-side MOSFET over-current protection is implemented by the nature of the Peak Current Mode control. The
HS switch current is sensed when the HS is turned on after a set blanking time. The HS switch current is
compared to the output of the Error Amplifier (EA) minus slope compensation every switching cycle. Please refer
to Functional Block Diagram for more details. The peak current of HS switch is limited by a clamped maximum
peak current threshold IHS_LIMIT which is constant. So the peak current limit of the high-side switch is not affected
by the slope compensation and remains constant over the full duty cycle range.
The current going through LS MOSFET is also sensed and monitored. When the LS switch turns on, the inductor
current begins to ramp down. The LS switch will not be turned OFF at the end of a switching cycle if its current is
above the LS current limit ILS_LIMIT. The LS switch will be kept ON so that inductor current keeps ramping down,
until the inductor current ramps below the LS current limit ILS_LIMIT. Then the LS switch will be turned OFF and
the HS switch will be turned on after a dead time. This is somewhat different than the more typical peak current
limit, and results in Equation 9 for the maximum load current.
VIN VOUT
V
IOUT _ MAX ILS _ LIMIT
u OUT
2 u fSW u L
VIN
(9)
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Feature Description (continued)
If the current of the LS switch is higher than the LS current limit for 64 consecutive cycles, hiccup current
protection mode will be activated. In hiccup mode, the regulator will be shut down and kept off for 5 ms typically
before the LMR23610-Q1 tries to start again. If over-current or short-circuit fault condition still exist, hiccup will
repeat until the fault condition is removed. Hiccup mode reduces power dissipation under severe over-current
conditions, prevents over-heating and potential damage to the device.
7.3.9 Thermal Shutdown
The LMR23610-Q1 provides an internal thermal shutdown to protect the device when the junction temperature
exceeds 170 °C (typ). The device is turned off when thermal shutdown activates. Once the die temperature falls
below 155 °C (typ), the device reinitiates the power up sequence controlled by the internal soft-start circuitry.
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7.4 Device Functional Modes
7.4.1 Shutdown Mode
The EN pin provides electrical ON and OFF control for the LMR23610-Q1. When VEN is below 1 V (typ), the
device is in shutdown mode. The LMR23610-Q1 also employs VIN and VCC under voltage lock out protection. If
VIN or VCC voltage is below their respective UVLO level, the regulator will be turned off.
7.4.2 Active Mode
The LMR23610-Q1 is in Active Mode when VEN is above the precision enable threshold, VIN and VCC are above
their respective UVLO level. The simplest way to enable the LMR23610-Q1 is to connect the EN pin to VIN pin.
This allows self startup when the input voltage is in the operating range: 4 V to 36 V. Please refer to VCC, UVLO
and Enable/Sync for details on setting these operating levels.
In Active Mode, depending on the load current, the LMR23610-Q1 will be in one of three modes:
1. Continuous conduction mode (CCM) with fixed switching frequency when load current is above half of the
peak-to-peak inductor current ripple.
2. Discontinuous conduction mode (DCM) with fixed switching frequency when load current is lower than half of
the peak-to-peak inductor current ripple in CCM operation.
3. Pulse frequency modulation mode (PFM) when switching frequency is decreased at very light load.
7.4.3 CCM Mode
CCM operation is employed in the LMR23610-Q1 when the load current is higher than half of the peak-to-peak
inductor current. In CCM operation, the frequency of operation is fixed, output voltage ripple will be at a minimum
in this mode and the maximum output current of 1 A can be supplied by the LMR23610-Q1.
7.4.4 Light Load Operation
When the load current is lower than half of the peak-to-peak inductor current in CCM, the LMR23610-Q1 will
operate in Discontinuous Conduction Mode (DCM), also known as Diode Emulation Mode (DEM). In DCM, the
LS switch is turned off when the inductor current drops to IL_ZC (-40 mA typ). Both switching losses and
conduction losses are reduced in DCM, compared to forced PWM operation at light load.
At even lighter current loads, Pulse Frequency Modulation (PFM) is activated to maintain high efficiency
operation. When either the minimum HS switch ON time (tON_MIN ) or the minimum peak inductor current IPEAK_MIN
(300 mA typ) is reached, the switching frequency will decrease to maintain regulation. In PFM, switching
frequency is decreased by the control loop when load current reduces to maintain output voltage regulation.
Switching loss is further reduced in PFM operation due to less frequent switching actions. The external clock
synchronizing will not be valid when LMR23610-Q1 enters into PFM mode.
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The LMR23610-Q1 is a step down DC-to-DC regulator. It is typically used to convert a higher DC voltage to a
lower DC voltage with a maximum output current of 1 A. The following design procedure can be used to select
components for the LMR23610-Q1. Alternately, the WEBENCH® software may be used to generate complete
designs. When generating a design, the WEBENCH® software utilizes iterative design procedure and accesses
comprehensive databases of components. Please go to ti.com for more details.
8.2 Typical Applications
The LMR23610-Q1 only requires a few external components to convert from a wide voltage range supply to a
fixed output voltage. Figure 21 shows a basic schematic.
VIN 12 V
CBOOT
0.1 F
BOOT
VIN
L
22 H
CIN
10 F
VOUT
5 V/1 A
SW
EN/
SYNC
PAD
CFF
75 pF
RFBT
88.7 kŸ
FB
CVCC
2.2 F
RFBB
22.1 kŸ
VCC
PGND
COUT
68 F
AGND
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Figure 21. Application Circuit
The external components have to fulfill the needs of the application, but also the stability criteria of the device's
control loop. Table 1 can be used to simplify the output filter component selection.
Table 1. L, COUT and CFF Typical Values
fSW (kHz)
VOUT (V)
L (µH)
COUT (µF)
CFF (pF)
RFBT (kΩ)
400
3.3
15
82
100
51
400
5
22
68
75
88.7
400
12
47
33
See note (5)
243
400
24
47
22
See note (5)
510
1.
2.
3.
4.
Inductance value is calculated based on VIN = 36 V.
All the COUT values are after derating. Add more when using ceramic capacitors.
RFBT = 0 Ω for VOUT = 1 V. RFBB = 22.1 kΩ for all other VOUT setting.
For designs with RFBT other than recommended value, please adjust CFF such that (CFF × RFBT) is
unchanged and adjust RFBB such that (RFBT / RFBB) is unchanged.
5. High ESR COUT will give enough phase boost and CFF not needed.
8.2.1 Design Requirements
Detailed design procedure is described based on a design example. For this design example, use the
parameters listed in Table 2 as the input parameters.
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Table 2. Design Example Parameters
Input Voltage, VIN
12 V typical, range from 8 V to 28 V
Output Voltage, VOUT
5V
Maximum Output Current IO_MAX
1A
Transient Response 0.1 A to 1 A
5%
Output Voltage Ripple
50 mV
Input Voltage Ripple
400 mV
Switching Frequency fSW
400 kHz
8.2.2 Detailed Design Procedure
8.2.2.1
Output Voltage Set-Point
The output voltage of LMR23610-Q1 is externally adjustable using a resistor divider network. The divider network
is comprised of top feedback resistor RFBT and bottom feedback resistor RFBB. Equation 10 is used to determine
the output voltage:
VOUT VREF
u RFBB
RFBT
VREF
(10)
Choose the value of RFBB to be 22.1 kΩ. With the desired output voltage set to 5 V and the VREF = 1.0 V, the
RFBB value can then be calculated using Equation 10. The formula yields to a value 88.7 kΩ.
8.2.2.2
Switching Frequency
The default switching frequency of the LMR23610-Q1 is 400 kHz. For other switching frequency, the device must
be synchronized to an external clock, please refer to Enable/Sync for more details.
8.2.2.3
Inductor Selection
The most critical parameters for the inductor are the inductance, saturation current and the rated current. The
inductance is based on the desired peak-to-peak ripple current ΔiL. Since the ripple current increases with the
input voltage, the maximum input voltage is always used to calculate the minimum inductance LMIN. Use
Equation 12 to calculate the minimum value of the output inductor. KIND is a coefficient that represents the
amount of inductor ripple current relative to the maximum output current of the device. A reasonable value of
KIND should be 30% to 50%. During an instantaneous short or over current operation event, the RMS and peak
inductor current can be high. The inductor current rating should be higher than the current limit of the device.
'iL
LMIN
VOUT u VIN _ MAX
VOUT
VIN _ MAX u L u fSW
VIN _ MAX
VOUT
IOUT u KIND
(11)
VOUT
u
VIN _ MAX u fSW
(12)
In general, it is preferable to choose lower inductance in switching power supplies, because it usually
corresponds to faster transient response, smaller DCR, and reduced size for more compact designs. But too low
of an inductance can generate too large of an inductor current ripple such that over current protection at the full
load could be falsely triggered. It also generates more conduction loss and inductor core loss. Larger inductor
current ripple also implies larger output voltage ripple with same output capacitors. With peak current mode
control, it is not recommended to have too small of an inductor current ripple. A larger peak current ripple
improves the comparator signal to noise ratio.
For this design example, choose KIND = 0.5, the minimum inductor value is calculated to be 20.5 µH. Choose the
nearest standard 22 μH ferrite inductor with a capability of 2 A RMS current and 2.5 A saturation current.
8.2.2.4
Output Capacitor Selection
The output capacitor(s), COUT, should be chosen with care since it directly affects the steady state output voltage
ripple, loop stability and the voltage over/undershoot during load current transients.
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The output ripple is essentially composed of two parts. One is caused by the inductor current ripple going
through the Equivalent Series Resistance (ESR) of the output capacitors:
'VOUT_ESR 'iL u ESR KIND u IOUT u ESR
(13)
The other is caused by the inductor current ripple charging and discharging the output capacitors:
KIND u IOUT
'iL
'VOUT _ C
8 u fSW u COUT
8 u fSW u COUT
(14)
The two components in the voltage ripple are not in phase, so the actual peak-to-peak ripple is smaller than the
sum of two peaks.
Output capacitance is usually limited by transient performance specifications if the system requires tight voltage
regulation with presence of large current steps and fast slew rate. When a fast large load increase happens,
output capacitors provide the required charge before the inductor current can slew up to the appropriate level.
The regulator’s control loop usually needs six or more clock cycles to respond to the output voltage droop. The
output capacitance must be large enough to supply the current difference for six clock cycles to maintain the
output voltage within the specified range. Equation 15 shows the minimum output capacitance needed for
specified output undershoot. When a sudden large load decrease happens, the output capacitors absorb energy
stored in the inductor. which results in an output voltage overshoot. Equation 16 calculates the minimum
capacitance required to keep the voltage overshoot within a specified range.
6 u IOH IOL
COUT !
fSW u VUS
(15)
COUT !
2
2
IOH
IOL
VOUT u VOS
2
2
VOUT
uL
(16)
where
• KIND = Ripple ratio of the inductor ripple current (ΔiL / IOUT)
• IOL = Low level output current during load transient
• IOH = High level output current during load transient
• VUS = Target output voltage undershoot
• VOS = Target output voltage overshoot
For this design example, the target output ripple is 50 mV. Presuppose ΔVOUT_ESR = ΔVOUT_C = 50 mV, and
chose KIND = 0.5. Equation 13 yields ESR no larger than 100 mΩ and Equation 14 yields COUT no smaller than
3.1 μF. For the target over/undershoot range of this design, VUS = VOS = 5% × VOUT = 250 mV. The COUT can be
calculated to be no smaller than 54 μF and 8.5 μF by Equation 15 and Equation 16 respectively. Consider of
derating, one 82 μF, 16 V ceramic capacitor with 5 mΩ ESR is used.
8.2.2.5
Feed-Forward Capacitor
The LMR23610-Q1 is internally compensated. Depending on the VOUT and frequency fSW, if the output capacitor
COUT is dominated by low ESR (ceramic types) capacitors, it could result in low phase margin. To improve the
phase boost an external feedforward capacitor CFF can be added in parallel with RFBT. CFF is chosen such that
phase margin is boosted at the crossover frequency without CFF. A simple estimation for the crossover frequency
(fX) without CFF is shown in Equation 17, assuming COUT has very small ESR, and COUT value is after derating.
8.32
fX
VOUT u COUT
(17)
The following equation for CFF was tested:
1
CFF
2S u fX u RFBT
(18)
For designs with higher ESR, CFF is not needed when COUT has very high ESR and CFF calculated from
Equation 18 should be reduced with medium ESR. Table 1 can be used as a quick starting point.
For the application in this design example, a 75 pF, 50 V, COG capacitor is selected.
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Input Capacitor Selection
The LMR23610-Q1 device requires high frequency input decoupling capacitor(s) and a bulk input capacitor,
depending on the application. The typical recommended value for the high frequency decoupling capacitor is 4.7
μF to 10 μF. A high-quality ceramic capacitor type X5R or X7R with sufficiency voltage rating is recommended.
To compensate the derating of ceramic capacitors, a voltage rating of twice the maximum input voltage is
recommended. Additionally, some bulk capacitance can be required, especially if the LMR23610-Q1 circuit is not
located within approximately 5 cm from the input voltage source. This capacitor is used to provide damping to the
voltage spike due to the lead inductance of the cable or the trace. For this design, two 4.7 μF, 50 V, X7R ceramic
capacitors are used. A 0.1 μF for high-frequency filtering and place it as close as possible to the device pins.
8.2.2.7
Bootstrap Capacitor Selection
Every LMR23610-Q1 design requires a bootstrap capacitor (CBOOT). The recommended capacitor is 0.1 μF and
rated 16 V or higher. The bootstrap capacitor is located between the SW pin and the BOOT pin. The bootstrap
capacitor must be a high-quality ceramic type with an X7R or X5R grade dielectric for temperature stability.
8.2.2.8 VCC Capacitor Selection
The VCC pin is the output of an internal LDO for LMR23610-Q1. To insure stability of the device, place a
minimum of 2.2 μF, 16 V, X7R capacitor from this pin to ground.
8.2.2.9 Under Voltage Lockout Set-Point
The system undervoltage lockout (UVLO) is adjusted using the external voltage divider network of RENT and
RENB. The UVLO has two thresholds, one for power up when the input voltage is rising and one for power down
or brown outs when the input voltage is falling. The following equation can be used to determine the VIN UVLO
level.
R
RENB
VIN _ RISING VENH u ENT
RENB
(19)
The EN rising threshold (VENH) for LMR23610-Q1 is set to be 1.55 V (typ). Choose the value of RENB to be 287
kΩ to minimize input current from the supply. If the desired VIN UVLO level is at 6.0 V, then the value of RENT can
be calculated using the equation below:
§ VIN _ RISING
·
RENT ¨¨
1¸¸ u RENB
© VENH
¹
(20)
The above equation yields a value of 820 kΩ. The resulting falling UVLO threshold, equals 4.4 V, can be
calculated by below equation, where EN hysteresis (VEN_HYS) is 0.4 V (typ).
R
RENB
VIN _ FALLING
VENH VEN _ HYS u ENT
RENB
(21)
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8.2.3 Application Curves
Unless otherwise specified the following conditions apply: VIN = 12 V, fSW = 400 kHz, L = 22 µH, COUT = 47 µF × 2, TA = 25
°C.
VOUT = 5 V
IOUT = 1 A
fSW = 400 kHz
VOUT = 5 V
Figure 22. CCM Mode
VIN = 12 V
VOUT = 5 V
VOUT = 5 V
IOUT = 1 A
VIN = 12 V
VOUT = 5 V
IOUT = 1 A
Figure 25. Start Up by EN
IOUT = 0.1 A to 1 A,
100 mA / μs
VOUT = 7 V to 36
V, 2 V / μs
Figure 26. Load Transient
22
fSW = 400 kHz
Figure 23. PFM Mode
Figure 24. Start Up by VIN
VIN = 12 V
IOUT = 0 mA
VOUT = 5 V
IOUT = 1 A
Figure 27. Line Transient
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Unless otherwise specified the following conditions apply: VIN = 12 V, fSW = 400 kHz, L = 22 µH, COUT = 47 µF × 2, TA = 25
°C.
VOUT = 5 V
IOUT = 1 A to short
VOUT = 5 V
Figure 28. Short Protection
IOUT = short to 1 A
Figure 29. Short Recovery
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9 Power Supply Recommendations
The LMR23610-Q1 is designed to operate from an input voltage supply range between 4 V and 36 V. This input
supply should be able to withstand the maximum input current and maintain a stable voltage. The resistance of
the input supply rail should be low enough that an input current transient does not cause a high enough drop at
the LMR23610-Q1 supply voltage that can cause a false UVLO fault triggering and system reset. If the input
supply is located more than a few inches from the LMR23610-Q1, additional bulk capacitance may be required in
addition to the ceramic input capacitors. The amount of bulk capacitance is not critical, but a 47 μF or 100 μF
electrolytic capacitor is a typical choice.
10 Layout
10.1 Layout Guidelines
Layout is a critical portion of good power supply design. The following guidelines will help users design a PCB
with the best power conversion performance, thermal performance, and minimized generation of unwanted EMI.
1. The input bypass capacitor CIN must be placed as close as possible to the VIN and PGND pins. Grounding
for both the input and output capacitors should consist of localized top side planes that connect to the PGND
pin and PAD.
2. Place bypass capacitors for VCC close to the VCC pin and ground the bypass capacitor to device ground.
3. Minimize trace length to the FB pin net. Both feedback resistors, RFBT and RFBB should be located close to
the FB pin. Place CFF directly in parallel with RFBT. If VOUT accuracy at the load is important, make sure VOUT
sense is made at the load. Route VOUT sense path away from noisy nodes and preferably through a layer on
the other side of a shielded layer.
4. Use ground plane in one of the middle layers as noise shielding and heat dissipation path.
5. Have a single point ground connection to the plane. The ground connections for the feedback and enable
components should be routed to the ground plane. This prevents any switched or load currents from flowing
in the analog ground traces. If not properly handled, poor grounding can result in degraded load regulation or
erratic output voltage ripple behavior.
6. Make VIN, VOUT and ground bus connections as wide as possible. This reduces any voltage drops on the
input or output paths of the converter and maximizes efficiency.
7. Provide adequate device heat-sinking. Use an array of heat-sinking vias to connect the exposed pad to the
ground plane on the bottom PCB layer. If the PCB has multiple copper layers, these thermal vias can also be
connected to inner layer heat-spreading ground planes. Ensure enough copper area is used for heat-sinking
to keep the junction temperature below 125 °C.
10.2 Compact Layout for EMI Reduction
Radiated EMI is generated by the high di/dt components in pulsing currents in switching converters. The larger
area covered by the path of a pulsing current, the more EMI is generated. High frequency ceramic bypass
capacitors at the input side provide primary path for the high di/dt components of the pulsing current. Placing
ceramic bypass capacitor(s) as close as possible to the VIN and PGND pins is the key to EMI reduction.
The SW pin connecting to the inductor should be as short as possible, and just wide enough to carry the load
current without excessive heating. Short, thick traces or copper pours (shapes) should be used for high current
conduction path to minimize parasitic resistance. The output capacitors should be placed close to the VOUT end
of the inductor and closely grounded to PGND pin and exposed PAD.
The bypass capacitors on VCC should be placed as close as possible to the pin and closely grounded to PGND
and the exposed PAD.
24
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10.3 Ground Plane and Thermal Considerations
It is recommended to use one of the middle layers as a solid ground plane. Ground plane provides shielding for
sensitive circuits and traces. It also provides a quiet reference potential for the control circuitry. The AGND and
PGND pins should be connected to the ground plane using vias right next to the bypass capacitors. PGND pin is
connected to the source of the internal LS switch. They should be connected directly to the grounds of the input
and output capacitors. The PGND net contains noise at switching frequency and may bounce due to load
variations. PGND trace, as well as VIN and SW traces, should be constrained to one side of the ground plane.
The other side of the ground plane contains much less noise and should be used for sensitive routes.
It is recommended to provide adequate device heat sinking by utilizing the PAD of the IC as the primary thermal
path. Use a minimum 4 by 2 array of 12 mil thermal vias to connect the PAD to the system ground plane heat
sink. The vias should be evenly distributed under the PAD. Use as much copper as possible, for system ground
plane, on the top and bottom layers for the best heat dissipation. Use a four-layer board with the copper
thickness for the four layers, starting from the top of, 2 oz / 1 oz / 1 oz / 2 oz. Four layer boards with enough
copper thickness provides low current conduction impedance, proper shielding and lower thermal resistance.
The thermal characteristics of the LMR23610-Q1 are specified using the parameter θJA, which characterize the
junction temperature of silicon to the ambient temperature in a specific system. Although the value of θJA is
dependent on many variables, it still can be used to approximate the operating junction temperature of the
device. To obtain an estimate of the device junction temperature, one may use the following relationship:
TJ = PD x θJA + TA
(22)
where
TJ = Junction temperature in °C
PD = VIN x IIN x (1 - Efficiency) - 1.1 x IOUT2 x DCR in Watt
DCR = Inductor DC parasitic resistance in Ω
θJA = Junction to ambient thermal resistance of the device in °C/W
TA = Ambient temperature in °C
The maximum operating junction temperature of the LMR23610-Q1 is 125 °C. θJA is highly related to PCB size
and layout, as well as environmental factors such as heat sinking and air flow.
10.4 Feedback Resistors
To reduce noise sensitivity of the output voltage feedback path, it is important to place the resistor divider and
CFF close to the FB pin, rather than close to the load. The FB pin is the input to the error amplifier, so it is a high
impedance node and very sensitive to noise. Placing the resistor divider and CFF closer to the FB pin reduces the
trace length of FB signal and reduces noise coupling. The output node is a low impedance node, so the trace
from VOUT to the resistor divider can be long if short path is not available.
If voltage accuracy at the load is important, make sure voltage sense is made at the load. Doing so will correct
for voltage drops along the traces and provide the best output accuracy. The voltage sense trace from the load to
the feedback resistor divider should be routed away from the SW node path and the inductor to avoid
contaminating the feedback signal with switch noise, while also minimizing the trace length. This is most
important when high value resistors are used to set the output voltage. It is recommended to route the voltage
sense trace and place the resistor divider on a different layer than the inductor and SW node path, such that
there is a ground plane in between the feedback trace and inductor/SW node polygon. This provides further
shielding for the voltage feedback path from EMI noises.
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10.5 Layout Example
Output Bypass
Capacitor
Output Inductor
SW
Input Bypass
Capacitor
PGND
BOOT Capacitor
BOOT
VCC
Capacitor
VIN
VCC
AGND
FB
EN/
SYNC
UVLO Adjust Resistor
Output Voltage Set
Resistor
Thermal VIA
VIA (Connect to GND Plane)
Figure 30. Layout
26
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11 Device and Documentation Support
11.1 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. In the upper
right corner, click on Alert me to register and receive a weekly digest of any product information that has
changed. For change details, review the revision history included in any revised document.
11.2 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
11.3 Trademarks
PowerPAD, E2E are trademarks of Texas Instruments.
SIMPLE SWITCHER is a registered trademark of Texas Instruments.
11.4 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.5 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
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12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
28
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PACKAGE OPTION ADDENDUM
www.ti.com
21-Dec-2016
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
Lead/Ball Finish
MSL Peak Temp
(2)
(6)
(3)
Op Temp (°C)
Device Marking
(4/5)
LMR23610AQDDAQ1
ACTIVE SO PowerPAD
DDA
8
75
Green (RoHS
& no Sb/Br)
CU NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
F10AQ
LMR23610AQDDARQ1
ACTIVE SO PowerPAD
DDA
8
2500
Green (RoHS
& no Sb/Br)
CU NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
F10AQ
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3)
MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4)
There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5)
Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6)
Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish
value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
Addendum-Page 1
Samples
PACKAGE OPTION ADDENDUM
www.ti.com
21-Dec-2016
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
OTHER QUALIFIED VERSIONS OF LMR23610-Q1 :
• Catalog: LMR23610
NOTE: Qualified Version Definitions:
• Catalog - TI's standard catalog product
Addendum-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
21-Dec-2016
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
LMR23610AQDDARQ1
Package Package Pins
Type Drawing
SO
Power
PAD
DDA
8
SPQ
Reel
Reel
A0
Diameter Width (mm)
(mm) W1 (mm)
2500
330.0
12.8
Pack Materials-Page 1
6.4
B0
(mm)
K0
(mm)
P1
(mm)
5.2
2.1
8.0
W
Pin1
(mm) Quadrant
12.0
Q1
PACKAGE MATERIALS INFORMATION
www.ti.com
21-Dec-2016
*All dimensions are nominal
Device
Package Type
Package Drawing
Pins
SPQ
Length (mm)
Width (mm)
Height (mm)
LMR23610AQDDARQ1
SO PowerPAD
DDA
8
2500
366.0
364.0
50.0
Pack Materials-Page 2
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