AD AD8224HBCPZ-RL Precision, dual-channel, jfet input, rail-to-rail instrumentation amplifier Datasheet

Precision, Dual-Channel, JFET Input,
Rail-to-Rail Instrumentation Amplifier
AD8224
APPLICATIONS
+VS
OUT1
OUT2
–VS
FUNCTIONAL BLOCK DIAGRAM
16
15
14
13
AD8224
–IN2
1
12
RG1
2
11
RG2
RG1
3
10
RG2
+IN1
4
9
+IN2
7
8
06286-001
6
–VS
5
REF2
–IN1
+VS
Two channels in a small 4 mm × 4 mm LFCSP
Custom LFCSP package with hidden paddle
Permits routing and vias underneath package
Allows full bias current performance
Low input currents
10 pA maximum input bias current (B Grade)
0.6 pA maximum input offset current (B Grade)
High CMRR
100 dB CMRR (minimum), G = 10 (B Grade)
90 dB CMRR (minimum) to 10 kHz, G = 10 (B Grade)
Excellent ac specifications and low power
1.5 MHz bandwidth (G = 1)
14 nV/√Hz input noise (1 kHz)
Slew rate: 2 V/μs
750 μA quiescent current per amplifier
Versatility
Rail-to-rail output
Input voltage range to below negative supply rail
4 kV ESD protection
4.5 V to 36 V single supply
±2.25 V to ±18 V dual supply
Gain set with single resistor (G = 1 to 1000)
REF1
FEATURES
Figure 1.
Table 1. In Amps and Difference Amplifiers by Category
High
Perform
AD8220 1
AD8221
AD8222
1
Low
Cost
AD85531
AD6231
High
Voltage
AD628
AD629
Mil
Grade
AD620
AD621
AD524
AD526
AD624
Low
Power
AD6271
Digital
Gain
AD82311
AD8250
AD8251
AD85551
AD85561
AD85571
Rail-to-rail output.
Medical instrumentation
Precision data acquisition
Transducer interfaces
Differential drives for high resolution input ADCs
Remote sensors
GENERAL DESCRIPTION
The AD8224 is the first single-supply, JFET input instrumentation
amplifier available in the space-saving 16-lead, 4 mm × 4 mm
LFCSP. It requires the same board area as a typical single
instrumentation amplifier yet doubles the channel density
and offers a lower cost per channel without compromising
performance.
Designed to meet the needs of high performance, portable
instrumentation, the AD8224 has a minimum common-mode
rejection ratio (CMRR) of 86 dB at dc and a minimum CMRR
of 80 dB at 10 kHz for G = 1. Maximum input bias current is
10 pA and typically remains below 300 pA over the entire
industrial temperature range. Despite the JFET inputs, the
AD8224 typically has a noise corner of only 10 Hz.
With the proliferation of mixed-signal processing, the number
of power supplies required in each system has grown. Designed
to alleviate this problem, the AD8224 can operate on a ±18 V
dual supply, as well as on a single +5 V supply. The device’s railto-rail output stage maximizes dynamic range on the low
voltage supplies common in portable applications. Its ability to
run on a single 5 V supply eliminates the need for higher
voltage, dual supplies. The AD8224 draws 750 μA of quiescent
current per amplifier, making it ideal for battery powered
devices.
In addition, the AD8224 can be configured as a single-channel,
differential output, instrumentation amplifier. Differential
outputs provide high noise immunity, which can be useful when
the output signal must travel through a noisy environment, such
as with remote sensors. The configuration can also be used to
drive differential input ADCs. For a single-channel version, use
the AD8220.
Rev. B
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
Trademarks and registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.461.3113 ©2007–2010 Analog Devices, Inc. All rights reserved.
AD8224
TABLE OF CONTENTS
Features .............................................................................................. 1
Layout .......................................................................................... 21
Applications ....................................................................................... 1
Solder Wash................................................................................. 22
Functional Block Diagram .............................................................. 1
Input Bias Current Return Path ............................................... 22
General Description ......................................................................... 1
Input Protection ......................................................................... 22
Revision History ............................................................................... 2
RF Interference ........................................................................... 23
Specifications..................................................................................... 3
Common-Mode Input Voltage Range ..................................... 23
Absolute Maximum Ratings............................................................ 9
Applications Information .............................................................. 24
Thermal Resistance ...................................................................... 9
Driving an ADC ......................................................................... 24
ESD Caution .................................................................................. 9
Differential Output .................................................................... 24
Pin Configuration and Function Descriptions ........................... 10
Driving a Differential Input ADC ............................................ 25
Typical Performance Characteristics ........................................... 11
Driving Cabling .......................................................................... 25
Theory of Operation ...................................................................... 20
Outline Dimensions ....................................................................... 26
Gain Selection ............................................................................. 20
Ordering Guide .......................................................................... 27
Reference Terminal .................................................................... 21
REVISION HISTORY
5/10—Rev. A to Rev. B
Changes to Features Section............................................................ 1
Added Table 10 ................................................................................. 9
Changes to Figure 3 and Table 11 ................................................. 10
Added Hidden Paddle Package Section and Exposed Paddle
Package Section and Figure 58...................................................... 21
Updated Outline Dimensions ....................................................... 26
Changes to Ordering Guide .......................................................... 27
4/07—Rev. 0 to Rev. A
Changes to Features, General Description, and Figure 1 ............ 1
Changes to Table 2 ............................................................................ 3
Changes to Table 3 and Table 4 ....................................................... 5
Changes to Table 5 ............................................................................ 6
Changes to Table 6 and Table 7 .......................................................8
Changes to Figure 2 ...........................................................................9
Changes to Figure 3 ........................................................................ 10
Inserted Figure 4, Figure 5, and Figure 6; Renumbered
Sequentially ..................................................................................... 11
Changes to Figure 7 ........................................................................ 11
Changes to Figure 20 and Figure 21............................................. 13
Changes to Figure 28...................................................................... 15
Changes to Theory of Operation and Figure 55 ........................ 20
Changes to Ordering Guide .......................................................... 26
1/07—Revision 0: Initial Version
Rev. B | Page 2 of 28
AD8224
SPECIFICATIONS
VS+ = +15 V, VS− = −15 V, VREF = 0 V, TA = 25°C, G = 1, RL = 2 kΩ 1 , unless otherwise noted. Table 2 displays the specifications for an
individual instrumentation amplifier configured for a single-ended output or dual instrumentation amplifiers configured for differential
outputs as shown in Figure 63.
Table 2. Individual Amplifier in Single-Ended Configuration or Dual Amplifiers in Differential Output Configuration 2 , VS = ±15 V
Parameter
COMMON-MODE REJECTION RATIO (CMRR)
Test Conditions
CMRR DC to 60 Hz with
1 kΩ Source Imbalance
G=1
G = 10
G = 100
G = 1000
CMRR at 10 kHz
G=1
G = 10
G = 100
G = 1000
NOISE
VCM = ±10 V
Voltage Noise, 1 kHz
Input Voltage Noise, eni
Output Voltage Noise, eno
RTI, 0.1 Hz to 10 Hz
G=1
Min
VOLTAGE OFFSET
Input Offset, VOSI
Average TC
Output Offset, VOSO
Average TC
Offset RTI vs. Supply (PSR)
G=1
G = 10
G = 100
G = 1000
INPUT CURRENT (PER CHANNEL)
Input Bias Current
Over Temperature 3
Input Offset Current
Over Temperature3
REFERENCE INPUT
RIN
IIN
Voltage Range
Gain to Output
Min
B Grade
Typ
Max
Unit
78
94
94
94
86
100
100
100
dB
dB
dB
dB
74
84
84
84
80
90
90
90
dB
dB
dB
dB
VCM = ±10 V
RTI noise =
√(eni2 + (eno/G)2)
VIN+, VIN− = 0 V
VIN+, VIN− = 0 V
14
90
G = 1000
Current Noise
A Grade
Typ
Max
f = 1 kHz
14
90
17
100
nV/√Hz
nV/√Hz
5
5
μV p-p
0.8
0.8
μV p-p
1
1
fA/√Hz
RTI VOS =
(VOSI) + (VOSO/G)
300
10
1200
10
T = −40°C to +85°C
T = −40°C to +85°C
VS = ±5 V to ±15 V
86
96
96
96
175
5
800
5
86
100
100
100
dB
dB
dB
dB
25
T = −40°C to +85°C
300
T = −40°C to +85°C
5
10
300
2
0.6
5
40
VIN+, VIN− = 0 V
−VS
1±
0.0001
Rev. B | Page 3 of 28
40
70
+VS
μV
μV/°C
μV
μV/°C
70
+VS
−VS
1±
0.0001
pA
pA
pA
pA
kΩ
μA
V
V/V
AD8224
Parameter
GAIN
Gain Range
Gain Error
G=1
G = 10
G = 100
G = 1000
Gain Nonlinearity
G=1
G = 10
G = 100
G = 1000
G=1
G = 10
G = 100
G=1000
Gain vs. Temperature
G=1
G > 10
INPUT
Impedance (Pin to Ground) 4
Input Operating Voltage Range 5
Over Temperature
OUTPUT
Output Swing
Over Temperature
Output Swing
Over Temperature
Short-Circuit Current
POWER SUPPLY (PER AMPLIFIER)
Operating Range
Quiescent Current
Over Temperature
TEMPERATURE RANGE
For Specified Performance
Operational 7
Test Conditions
G = 1 + (49.4 kΩ/RG)
Min
A Grade
Typ
Max
1
1000
Min
B Grade
Typ
Max
1
Unit
1000
V/V
0.04
0.2
0.2
0.2
%
%
%
%
VOUT = ±10 V
0.06
0.3
0.3
0.3
VOUT = −10 V to +10 V
RL = 10 kΩ
RL = 10 kΩ
RL = 10 kΩ
RL = 10 kΩ
RL = 2 kΩ
RL = 2 kΩ
RL = 2 kΩ
RL = 2 kΩ
8
5
15
100
15
12
35
180
15
10
25
150
20
20
50
250
8
5
15
100
15
12
35
180
15
10
25
150
20
20
50
250
ppm
ppm
ppm
ppm
ppm
ppm
ppm
ppm
3
10
−50
2
5
−50
ppm/°C
ppm/°C
104||5
104||5
VS = ±2.25 V to ±18 V
for dual supplies
T = −40°C to +85°C
−VS − 0.1
+VS − 2
−VS − 0.1
+VS − 2
GΩ||pF
V
−VS − 0.1
+VS − 2.1
−VS − 0.1
+VS − 2.1
V
RL = 2 kΩ
T = −40°C to +85°C
RL = 10 kΩ
T = −40°C to +85°C
−14.25
−14.3
−14.7
−14.6
+14.25
+14.1
+14.7
+14.6
−14.25
−14.3
−14.7
−14.6
+14.25
+14.1
+14.7
+14.6
V
V
V
V
mA
±18
800
900
V
μA
μA
+85
+125
°C
°C
15
±2.25 6
750
850
T = −40°C to +85°C
−40
−40
1
15
±18
800
900
±2.256
+85
+125
−40
−40
750
850
When the output sinks more than 4 mA, use a 47 pF capacitor in parallel with the load to prevent ringing. Otherwise, use a larger load, such as 10 kΩ.
Refers to the differential configuration shown in Figure 63.
Refer to Figure 14 and Figure 15 for the relationship between input current and temperature.
4
Differential and common-mode input impedance can be calculated from the pin impedance: ZDIFF = 2(ZPIN); ZCM = ZPIN/2.
5
The AD8224 can operate up to a diode drop below the negative supply; however, the bias current increases sharply. The input voltage range reflects the maximum
allowable voltage where the input bias current is within the specification.
6
At this supply voltage, ensure that the input common-mode voltage is within the input voltage range specification.
7
The AD8224 is characterized from −40°C to +125°C. See the Typical Performance Characteristics section for expected operation in this temperature range.
2
3
Rev. B | Page 4 of 28
AD8224
VS+ = +15 V, VS− = −15 V, VREF = 0 V, TA = 25°C, G = 1, RL = 2 kΩ 1 , unless otherwise noted. Table 3 displays the specifications for the
dynamic performance of each individual instrumentation amplifier.
Table 3. Dynamic Performance of Each Individual Amplifier—Single-Ended Output Configuration, VS = ±15 V
Parameter
DYNAMIC RESPONSE
Small Signal Bandwidth −3 dB
G=1
G = 10
G = 100
G =1000
Settling Time 0.01%
G=1
G = 10
G = 100
G =1000
Settling Time 0.001%
G=1
G = 10
G = 100
G =1000
Slew Rate
G = 1 to 100
1
Conditions
Min
A Grade
Typ
Max
Min
B Grade
Typ
Max
Unit
1500
800
120
14
1500
800
120
14
kHz
kHz
kHz
kHz
5
4.3
8.1
58
5
4.3
8.1
58
μs
μs
μs
μs
6
4.6
9.6
74
6
4.6
9.6
74
μs
μs
μs
μs
2
V/μs
ΔVO = ±10 V step
ΔVO = ±10 V step
2
When the output sinks more than 4 mA, use a 47 pF capacitor in parallel with the load to prevent ringing. Otherwise, use a larger load, such as 10 kΩ.
VS+ = +15 V, VS− = −15 V, VREF = 0 V, TA = 25°C, G = 1, RL = 2 kΩ 1 , unless otherwise noted. Table 4 displays the specifications for the
dynamic performance of both amplifiers when used in the differential output configuration shown in Figure 63.
Table 4. Dynamic Performance of Both Amplifiers—Differential Output Configuration 2 , VS = ±15 V
Parameter
DYNAMIC RESPONSE
Small Signal Bandwidth −3 dB
G=1
G = 10
G = 100
G =1000
Settling Time 0.01%
G=1
G = 10
G = 100
G =1000
Settling Time 0.001%
G=1
G = 10
G = 100
G =1000
Slew Rate
G = 1 to 100
Conditions
Min
A Grade
Typ
Max
Min
B Grade
Typ
Max
Unit
1500
800
120
14
1500
800
120
14
kHz
kHz
kHz
kHz
5
4.3
8.1
58
5
4.3
8.1
58
μs
μs
μs
μs
6
4.6
9.6
74
6
4.6
9.6
74
μs
μs
μs
μs
2
V/μs
ΔVO = ±10 V step
ΔVO = ±10 V step
2
1
When the output sinks more than 4 mA, use a 47 pF capacitor in parallel with the load to prevent ringing. Otherwise, use a larger load, such as 10 kΩ.
2
Refers to the differential configuration shown in Figure 63.
Rev. B | Page 5 of 28
AD8224
VS + = 5 V, VS− = 0 V, VREF = 2.5 V, TA = 25°C, G = 1, RL = 2 kΩ 1 , unless otherwise noted. Table 5 displays the specifications for an
individual instrumentation amplifier configured for a single-ended output or dual instrumentation amplifiers configured for differential
outputs as shown in Figure 63.
Table 5. Individual Amplifier in Single-Ended Configuration or Dual Amplifiers in Differential Output Configuration 2 , VS =+5 V
Parameter
COMMON-MODE REJECTION RATIO (CMRR)
CMRR DC to 60 Hz with
1 kΩ Source Imbalance
G=1
G = 10
G = 100
G = 1000
CMRR at 10 kHz
G=1
G = 10
G = 100
G = 1000
NOISE
Voltage Noise, 1 kHz
Input Voltage Noise, eni
Output Voltage Noise, eno
RTI, 0.1 Hz to 10 Hz
G=1
G = 1000
Current Noise
VOLTAGE OFFSET
Input Offset, VOSI
Average TC
Output Offset, VOSO
Average TC
Offset RTI vs. Supply (PSR)
G=1
G = 10
G = 100
G = 1000
INPUT CURRENT (PER CHANNEL)
Input Bias Current
Over Temperature 3
Input Offset Current
Over Temperature3
REFERENCE INPUT
RIN
IIN
Voltage Range
Gain to Output
Test Conditions
Min
A Grade
Typ
Max
Min
B Grade
Typ
Max
Unit
VCM = 0 to 2.5 V
78
94
94
94
86
100
100
100
dB
dB
dB
dB
74
84
84
84
80
90
90
90
dB
dB
dB
dB
RTI noise = √(eni2 + (eno/G)2)
VS = ±2.5 V
VIN+, VIN− = 0 V, VREF = 0 V
VIN+, VIN− = 0 V, VREF = 0 V
f = 1 kHz
RTI VOS = (VOSI) + (VOSO/G)
14
90
14
90
5
0.8
1
5
0.8
1
300
10
1200
10
T = −40°C to +85°C
T = −40°C to +85°C
86
96
96
96
250
5
800
5
T = −40°C to +85°C
5
10
300
2
0.6
5
40
VIN+, VIN− = 0 V
40
70
+VS
−VS
1±
0.0001
μV
μV/°C
μV
μV/°C
dB
dB
dB
dB
25
300
nV/√Hz
nV/√Hz
μV p-p
μV p-p
fA/√Hz
86
100
100
100
T = −40°C to +85°C
Rev. B | Page 6 of 28
17
100
70
+VS
−VS
1±
0.0001
pA
pA
pA
pA
kΩ
μA
V
V/V
AD8224
Parameter
GAIN
Gain Range
Gain Error
G=1
G = 10
G = 100
G = 1000
Nonlinearity
G=1
G = 10
G = 100
G = 1000
G=1
G = 10
G = 100
G = 1000
Gain vs. Temperature
G=1
G > 10
INPUT
Impedance (Pin to Ground) 4
Input Voltage Range 5
Over Temperature
OUTPUT
Output Swing
Over Temperature
Output Swing
Over Temperature
Short-Circuit Current
POWER SUPPLY (PER AMPLIFIER)
Operating Range
Quiescent Current
Over Temperature
TEMPERATURE RANGE
For Specified Performance
Operational 6
Test Conditions
G = 1 + (49.4 kΩ/RG)
Min
A Grade
Typ
Max
1
VOUT = 0.3 V to 2.9 V
VOUT = 0.3 V to 3.8 V
VOUT = 0.3 V to 3.8 V
VOUT = 0.3 V to 3.8 V
VOUT = 0.3 V to 2.9 V for G = 1
VOUT = 0.3 V to 3.8 V for G > 1
RL = 10 kΩ
RL = 10 kΩ
RL = 10 kΩ
RL = 10 kΩ
RL = 2 kΩ
RL = 2 kΩ
RL = 2 kΩ
RL = 2 kΩ
1000
Min
B Grade
Typ
Max
1
0.06
0.3
0.3
0.3
Unit
1000
V/V
0.04
0.2
0.2
0.2
%
%
%
%
35
35
50
90
35
35
50
175
50
50
75
115
50
50
75
200
35
35
50
90
35
35
50
175
50
50
75
115
50
50
75
200
ppm
ppm
ppm
ppm
ppm
ppm
ppm
ppm
3
10
−50
2
5
−50
ppm/°C
ppm/°C
GΩ||pF
V
V
104||6
104||6
T = −40°C to +85°C
−0.1
−0.1
+VS − 2
+VS − 2.1
−0.1
−0.1
+VS − 2
+VS − 2.1
RL = 2 kΩ
T = −40°C to +85°C
RL = 10 kΩ
T = −40°C to +85°C
0.25
0.3
0.15
0.2
4.75
4.70
4.85
4.80
0.25
0.3
0.15
0.2
4.75
4.70
4.85
4.80
V
V
V
V
mA
36
800
900
V
μA
μA
+85
+125
°C
°C
15
4.5
750
850
T = −40°C to +85°C
−40
−40
1
15
36
800
900
4.5
+85
+125
−40
−40
750
850
When the output sinks more than 4 mA, use a 47 pF capacitor in parallel with the load to prevent ringing. Otherwise, use a larger load, such as 10 kΩ.
Refers to the differential configuration shown in Figure 63.
3
Refer to Figure 14 and Figure 15 for the relationship between input current and temperature.
4
Differential and common-mode impedance can be calculated from the pin impedance: ZDIFF = 2(ZPIN); ZCM = ZPIN/2.
5
The AD8224 can operate up to a diode drop below the negative supply, but the bias current increases sharply. The input voltage range reflects the maximum
allowable voltage where the input bias current is within the specification.
6
The AD8224 is characterized from −40°C to +125°C. See the Typical Performance Characteristics section for expected operation in that temperature range.
2
Rev. B | Page 7 of 28
AD8224
VS + = 5 V, VS− = 0 V, VREF = 2.5 V, TA = 25°C, G = 1, RL = 2 kΩ 1 , unless otherwise noted. Table 6 displays the specifications for the
dynamic performance of each individual instrumentation amplifier.
Table 6. Dynamic Performance of Each Individual Amplifier—Single-Ended Output Configuration, VS = +5 V
Parameter
DYNAMIC RESPONSE
Small Signal Bandwidth −3 dB
G=1
G = 10
G = 100
G =1000
Settling Time 0.01%
G=1
G = 10
G = 100
G =1000
Settling Time 0.001%
G=1
G = 10
G = 100
G =1000
Slew Rate
G = 1 to 100
1
Conditions
Min
A Grade
Typ
Max
Min
B Grade
Typ
Max
Unit
1500
800
120
14
1500
800
120
14
kHz
kHz
kHz
kHz
ΔVO = 3 V step
ΔVO = 4 V step
ΔVO = 4 V step
ΔVO = 4 V step
2.5
2.5
7.5
60
2.5
2.5
7.5
60
μs
μs
μs
μs
ΔVO = 3 V step
ΔVO = 4 V step
ΔVO = 4 V step
ΔVO = 4 V step
3.5
3.5
8.5
75
3.5
3.5
8.5
75
μs
μs
μs
μs
2
2
V/μs
When the output sinks more than 4 mA, use a 47 pF capacitor in parallel with the load to prevent ringing. Otherwise, use a larger load, such as 10 kΩ.
VS + = 5 V, VS− = 0 V, VREF = 2.5 V, TA = 25°C, G = 1, RL = 2 kΩ 1 unless otherwise noted. Table 7 displays the specifications for the
dynamic performance of both amplifiers when used in the differential output configuration shown in Figure 63.
Table 7. Dynamic Performance of Both Amplifiers—Differential Output Configuration 2 , VS = +5 V
Parameter
DYNAMIC RESPONSE
Small Signal Bandwidth −3 dB
G=1
G = 10
G = 100
G =1000
Settling Time 0.01%
G=1
G = 10
G = 100
G =1000
Settling Time 0.001%
G=1
G = 10
G = 100
G =1000
Slew Rate
G = 1 to 100
Conditions
Min
A Grade
Typ
Max
Min
B Grade
Typ
Max
Unit
1500
800
120
14
1500
800
120
14
kHz
kHz
kHz
kHz
ΔVO = 3 V step
ΔVO = 4 V step
ΔVO = 4 V step
ΔVO = 4 V step
2.5
2.5
7.5
60
2.5
2.5
7.5
60
μs
μs
μs
μs
ΔVO = 3 V step
ΔVO = 4 V step
ΔVO = 4 V step
ΔVO = 4 V step
3.5
3.5
8.5
75
3.5
3.5
8.5
75
μs
μs
μs
μs
2
1
2
When the output sinks more than 4 mA, use a 47 pF capacitor in parallel with the load to prevent ringing. Otherwise, use a larger load, such as 10 kΩ.
2
Refers to the differential configuration shown in Figure 63.
Rev. B | Page 8 of 28
V/μs
AD8224
ABSOLUTE MAXIMUM RATINGS
THERMAL RESISTANCE
Table 8.
Rating
±18 V
See Figure 2
Indefinite 1
±VS
±VS
−65°C to +130°C
−40°C to +125°C
300°C
130°C
130°C
4 kV
1 kV
0.4 kV
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
2
Assumes the load is referenced to midsupply.
Temperature for specified performance is −40°C to +85°C. For performance
to 125°C, see the Typical Performance Characteristics section.
Exposed Paddle Package
CP-16-13: LFCSP Soldered to Board
CP-16-13: LFCSP Not Soldered to Board
θJA
48
86
Unit
°C/W
°C/W
θJA
86
Unit
°C/W
Table 10.
Hidden Paddle Package
CP-16-19: LFCSP
The θJA values in Table 9 and Table 10 assume a 4-layer JEDEC
standard board. If the thermal pad is soldered to the board, it is
also assumed it is connected to a plane. θJC at the exposed pad is
4.4°C/W.
Maximum Power Dissipation
The maximum safe power dissipation for the AD8224 is limited
by the associated rise in junction temperature (TJ) on the die. At
approximately 130°C, which is the glass transition temperature,
the plastic changes its properties. Even temporarily exceeding
this temperature limit may change the stresses that the package
exerts on the die, permanently shifting the parametric performance
of the amplifiers. Exceeding a temperature of 130°C for an
extended period can result in a loss of functionality. Figure 2
shows the maximum safe power dissipation in the package vs.
the ambient temperature for the LFCSP on a 4-layer JEDEC
standard board.
4.0
MAXIMUM POWER DISSIPATION (W)
1
Table 9.
3.5
3.0
θJA = 48°C/W WHEN THERMAL PAD
IS SOLDERED TO BOARD
2.5
2.0
1.5
1.0
0.5
θJA = 86°C/W WHEN THERMAL PAD
IS NOT SOLDERED TO BOARD
0
–60
–40
–20
0
20
40
60
80
100
120
140
AMBIENT TEMPERATURE (°C)
Figure 2. Maximum Power Dissipation vs. Ambient Temperature
ESD CAUTION
Rev. B | Page 9 of 28
06286-002
Parameter
Supply Voltage
Power Dissipation
Output Short-Circuit Current
Input Voltage (Common Mode)
Differential Input Voltage
Storage Temperature Range
Operating Temperature Range 2
Lead Temperature (Soldering, 10 sec)
Junction Temperature
Package Glass Transition Temperature
ESD (Human Body Model)
ESD (Charge Device Model)
ESD (Machine Model)
AD8224
16 +VS
15 OUT1
14 OUT2
13 –VS
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
PIN 1
INDICATOR
12 –IN2
11 RG2
10 RG2
9 +IN2
–VS 8
TOP VIEW
+VS 5
+IN1 4
AD8224
REF1 6
REF2 7
RG1 2
RG1 3
NOTES
1. THE AD8224 COMES IN TWO PACKAGE TYPES—EACH IS A 16 LEAD
4mm × 4mm LFCSP. ONE PACKAGE HAS AN EXPOSED THERMAL PAD,
WHICH IS CONNECTED TO +VS. THE OTHER PACKAGE TYPE DOES NOT
EXPOSE THE THERMAL PAD. SEE THE PACKAGE CONSIDERATIONS
SECTION FOR MORE INFORMATION.
06286-003
–IN1 1
Figure 3. Pin Configuration
Table 11. Pin Function Descriptions
Pin Number
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
16
Mnemonic
−IN1
RG1
RG1
+IN1
+VS
REF1
REF2
−VS
+IN2
RG2
RG2
−IN2
−VS
OUT2
OUT1
+VS
Description
Negative Input Instrumentation Amplifier (In-Amp) 1
Gain Resistor In-Amp 1
Gain Resistor In-Amp 1
Positive Input In-Amp 1
Positive Supply
Reference Adjust In-Amp 1
Reference Adjust In-Amp 2
Negative Supply
Positive Input In-Amp 2
Gain Resistor In-Amp 2
Gain Resistor In-Amp 2
Negative Input In-Amp 2
Negative Supply
Output In-Amp 2
Output In-Amp 1
Positive Supply
Rev. B | Page 10 of 28
AD8224
TYPICAL PERFORMANCE CHARACTERISTICS
25°C, VS = ±15 V, RL =10 kΩ, unless otherwise noted.
1000
400
VOLTAGE NOISE RTI (nV/ Hz)
300
250
200
150
100
GAIN = 100 BANDWIDTH ROLL-OFF
100
GAIN = 1
GAIN = 10
GAIN = 100/GAIN = 1000
10
GAIN = 1000 BANDWIDTH ROLL-OFF
50
–40
–20
0
20
40
1
06286-070
0
CMRR (µV/V)
1
10
100
1k
100k
10k
FREQUENCY (Hz)
06286-009
NUMBER OF UNITS
350
Figure 7. Voltage Spectral Density vs. Frequency
Figure 4. Typical Distribution of CMRR (G = 1)
XX
400
300
250
XXX (X)
NUMBER OF UNITS
350
200
150
100
50
0
100
200
VOSI (µV)
1s/DIV
XX
XX
XX
06286-010
–100
XX
06286-011
5µV/DIV
–200
06286-071
0
XXX (X)
Figure 8. 0.1 Hz to 10 Hz RTI Voltage Noise (G = 1)
Figure 5. Typical Distribution of Input Offset Voltage
XX
XXX (X)
300
200
100
0
–1200
1µV/DIV
–900
–600
–300
0
300
600
900
1200
VOSO (µV)
1s/DIV
XX
XX
06286-072
NUMBER OF UNITS
400
XXX (X)
Figure 6. Typical Distribution of Output Offset Voltage
Figure 9. 0.1 Hz to 10 Hz RTI Voltage Noise (G = 1000)
Rev. B | Page 11 of 28
AD8224
0.3
4.5
INPUT BIAS CURRENT (pA)
3.5
3.0
2.5
2.0
1.5
1.0
INPUT
OFFSET
CURRENT ±5
7
5
–15.1V
3
0.1
INPUT BIAS
CURRENT ±15
–0.1
INPUT BIAS
CURRENT ±5
–0.3
–5.1V
1
INPUT OFFSET CURRENT (pA)
4.0
DELTA VOSI (µV)
INPUT OFFSET
CURRENT ±15
9
0.5
10
TIME (s)
100
1000
–1
–16
–12
–8
–4
0
4
8
–0.5
16
12
06286-068
1
06286-012
0
0.1
COMMON-MODE VOLTAGE (V)
Figure 13. Input Bias Current and Input Offset Current vs. Common-Mode Voltage
Figure 10. Change in Input Offset Voltage vs. Warmup Time
150
INPUT BIAS CURRENT (A)
BANDWIDTH
LIMITED
GAIN = 100
110
GAIN = 10
90
GAIN = 1
70
50
IBIAS
100p
10p
IOS
1p
0.1p
30
1
10
100
1k
10k
100k
1M
FREQUENCY (Hz)
–50
06286-013
10
1n
–25
0
25
50
75
100
125
150
TEMPERATURE (°C)
06286-016
PSRR (dB)
10n
GAIN = 1000
130
Figure 14. Input Bias Current and Offset Current vs. Temperature,
VS = ±15 V, VREF = 0 V
Figure 11. Positive PSRR vs. Frequency, RTI
150
10n
130
1n
90
CURRENT (A)
GAIN = 1000
GAIN = 1
70
IBIAS
100p
10p
GAIN = 10
IOS
1p
50
GAIN = 100
10
1
10
100
1k
10k
100k
FREQUENCY (Hz)
1M
–50
–25
0
25
50
75
100
125
150
TEMPERATURE (°C)
Figure 15. Input Bias Current and Offset Current vs. Temperature,
VS = 5 V, VREF = 2.5 V
Figure 12. Negative PSRR vs. Frequency, RTI
Rev. B | Page 12 of 28
06286-017
0.1p
30
06286-014
PSRR (dB)
110
AD8224
70
160
60
140 GAIN = 1000
40
100
GAIN = 100
30
GAIN = 10
GAIN (dB)
CMRR (dB)
BANDWIDTH
LIMITED
GAIN = 100
120
GAIN = 1000
50
GAIN = 1
20
GAIN = 10
10
0
80
GAIN = 1
–10
–20
60
100
1000
10000
100000
FREQUENCY (Hz)
–40
100
06286-018
10
1k
10k
100k
06286-021
–30
40
10M
1M
FREQUENCY (Hz)
Figure 19. Gain vs. Frequency
Figure 16. CMRR vs. Frequency
160
GAIN = 100
GAIN = 10
XXX
CMRR (dB)
120
100 GAIN = 1
BANDWIDTH
LIMITED
80
NONLINEARITY (10ppm/DIV)
140 GAIN = 1000
RLOAD = 2kΩ
RLOAD = 10kΩ
06286-022
60
1
10
100
1000
10000
100000
FREQUENCY (Hz)
06286-019
VS = ±15V
40
–10
–8
–6
–4
–2
0
2
4
6
8
10
OUTPUT VOLTAGE (V)
Figure 17. CMRR vs. Frequency, 1 kΩ Source Imbalance
Figure 20. Gain Nonlinearity, G = 1
7
XXX
4
3
2
RLOAD = 2kΩ
RLOAD = 10kΩ
0
–50
06286-023
1
VS = ±15V
–30
–10
10
30
50
70
90
110
TEMPERATURE (°C)
130
06286-020
CMRR (µV/V)
5
NONLINEARITY (10ppm/DIV)
6
Figure 18. Change in CMRR vs. Temperature, G = 1
–10
–8
–6
–4
–2
0
2
4
OUTPUT VOLTAGE (V)
Figure 21. Gain Nonlinearity, G = 10
Rev. B | Page 13 of 28
6
8
10
AD8224
RLOAD = 10kΩ
VS = ±15V
–10
–8
–6
–4
–2
0
2
4
6
8
2
+0.1V, +1.7V
+4.9V, +1.7V
+5V SINGLE SUPPLY,
VREF = +2.5V
1
+0.1V, +0.5V
+4.9V, +0.5V
0
–0.3V
–1
–1
10
+3V
3
0
1
2
3
4
5
6
OUTPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
06286-027
INPUT COMMON-MODE VOLTAGE (V)
RLOAD = 2kΩ
06286-024
XXX
NONLINEARITY (20ppm/DIV)
4
Figure 25. Input Common-Mode Voltage Range vs. Output Voltage,
G = 1, VS = 5 V, VREF = 2.5 V
Figure 22. Gain Nonlinearity, G = 100
06286-025
RLOAD = 10kΩ
VS = ±15V
–8
–6
–4
–2
0
2
4
6
8
6
0
+3V
–14.9V, +5.4V
–5.3V
–14.8V, –9V
–12
–15.3V
–12
–8
–4
INPUT COMMON-MODE VOLTAGE (V)
–14.8V, +5.5V
+14.9V, +5.5V
+3V
–4.8V, +0.6V
+4.95V, +0.6V
±5V SUPPLIES
+4.95V, –3.3V
–14.8V, –8.3V
+14.9V, –8.3V
–5.3V
–12
–15.3V
–12
–8
–4
0
4
8
12
16
OUTPUT VOLTAGE (V)
06286-026
INPUT COMMON-MODE VOLTAGE (V)
±15V SUPPLIES
–4.8V, –3.3V
–18
–16
0
4
8
12
16
4
+13V
–6
+14.9V, –9V
Figure 26. Input Common-Mode Voltage Range vs. Output Voltage,
G = 100, VREF = 0 V
18
0
+4.9V, –4.1V
–6
OUTPUT VOLTAGE (V)
Figure 23. Gain Nonlinearity, G = 1000
6
+4.9V, +0.5V
±5V SUPPLIES
OUTPUT VOLTAGE (V)
12
+14.9V, +5.4V
–4.9V, +0.4V
–4.9V, –4.1V
–18
–16
10
±15V SUPPLIES
Figure 24. Input Common-Mode Voltage Range vs. Output Voltage,
G = 1, VREF = 0 V
+3V
3
2
+0.1V, +1.7V
+4.9V, +1.7V
+5V SINGLE SUPPLY,
VREF = +2.5V
1
0
+0.1V, –0.5V
–1
–1
0
1
+4.9V, –0.5V
–0.3V
2
3
4
5
6
OUTPUT VOLTAGE (V)
Figure 27. Input Common-Mode Voltage Range vs. Output Voltage,
G = 100, VS = 5 V, VREF = 2.5 V
Rev. B | Page 14 of 28
06286-029
–10
+13V
12
06286-028
INPUT COMMON-MODE VOLTAGE (V)
RLOAD = 2kΩ
NONLINEARITY (100ppm/DIV)
XXX
18
AD8224
15
+125°C
–40°C
–2
+25°C
+85°C
NOTES
1. THE AD8224 CAN OPERATE UP TO A VBE BELOW
THE NEGATIVE SUPPLY, BUT THE BIAS CURRENT
WILL INCREASE SHARPLY.
+1
–40°C
+25°C
+85°C
+125°C
4
6
8
10
12
14
16
18
SUPPLY VOLTAGE (V)
Figure 28. Input Voltage Limit vs. Supply Voltage, G = 1, VREF =0 V
+125°C
–5
+85°C
+25°C
–40°C
10k
1k
Figure 31. Output Voltage Swing vs. Load Resistance, VS = ±15 V, VREF = 0 V
–40°C
–3
OUTPUT VOLTAGE SWING (V)
+85°C
+25°C
+125°C
–4
+4
+3
+2
+125°C
+85°C
+25°C
+25°C
4
+85°C
+125°C
3
2
+125°C
1
+25°C
–40°C
+85°C
2
4
6
8
10
12
14
16
18
DUAL SUPPLY VOLTAGE (±V)
0
100
10k
1k
RLOAD (Ω)
06286-034
–40°C
06286-031
Figure 32. Output Voltage Swing vs. Load Resistance, VS = 5 V, VREF = 2.5 V
Figure 29. Output Voltage Swing vs. Dual Supply Voltage,
RLOAD = 2 kΩ, G = 10, VREF = 0 V
VS+
VS+
+125°C
–0.4
+0.4
+125°C
–40°C
–1
+85°C
+85°C
+25°C
+25°C
OUTPUT VOLTAGE SWING (V)
REFERRED TO SUPPLY VOLTAGES
–0.2
–40°C
–40°C
+125°C
–2
+85°C
+25°C
–3
–4
+4
+3
+2
+125°C
+85°C
+25°C
+1
+0.2
4
6
8
10
12
14
16
DUAL SUPPLY VOLTAGE (±V)
Figure 30. Output Voltage Swing vs. Dual Supply Voltage,
RLOAD = 10 kΩ, G = 10, VREF = 0 V
18
VS –
06286-032
2
–40°C
0
2
4
6
8
IOUT (mA)
10
12
14
16
06286-035
OUTPUT VOLTAGE SWING (V)
REFERRED TO SUPPLY VOLTAGES
–40°C
–2
+1
OUTPUT VOLTAGE SWING (V)
REFERRED TO SUPPLY VOLTAGES
0
5
–1
VS –
+125°C
RLOAD (Ω)
VS+
VS –
+85°C
5
–15
100
06286-030
2
+25°C
–10
VS –
–1
–40°C
10
OUTPUT VOLTAGE SWING (V)
INPUT VOLTAGE LIMIT (V)
–1
06286-033
VS+
Figure 33. Output Voltage Swing vs. Output Current, VS = ±15 V, VREF = 0 V
Rev. B | Page 15 of 28
AD8224
+85°C
+25°C
+125°C
–2
+2
+125°C
+1
+85°C
+25°C
–40°C
VS –
0
2
4
6
8
10
12
14
16
IOUT (mA)
Figure 34. Output Voltage Swing vs. Output Current, VS = 5 V, VREF = 2.5 V
30
25
GAIN = 10, 100, 1000
GAIN = 1
20
15
10
5
0
100
1k
10k
100k
10M
1M
FREQUENCY (Hz)
06286-039
–1
OUTPUT VOLTAGE SWING (V p-p)
35
06286-036
OUTPUT VOLTAGE SWING (V)
REFERRED TO SUPPLY VOLTAGES
VS+
Figure 37. Output Voltage Swing vs. Large Signal Frequency Response
XX
XX
NO LOAD
47pF
100pF
XXX (X)
XXX (X)
5V/DIV
0.002%/DIV
XX
XX
XX
XXX (X)
XX
XXX (X)
06286-040
20µs/DIV
5µs/DIV
06286-037
20mV/DIV
XX
XX
5µs TO 0.01%
6µs TO 0.001%
Figure 38. Large Signal Pulse Response and Settle Time, G = 1,
RLOAD = 10 kΩ, VS = ±15 V, VREF = 0 V
Figure 35. Small Signal Pulse Response for Various Capacitive Loads,
VS = ±15 V, VREF = 0 V
XX
XX
47pF
100pF
NO LOAD
XXX (X)
XXX (X)
5V/DIV
0.002%/DIV
XX
XXX (X)
Figure 36. Small Signal Pulse Response for Various Capacitive Loads,
VS = 5 V, VREF = 2.5 V
Rev. B | Page 16 of 28
XX
XX
XX
XXX (X)
Figure 39. Large Signal Pulse Response and Settle Time, G = 10,
RLOAD = 10 kΩ, VS = ±15 V, VREF = 0 V
06286-041
20µs/DIV
5µs/DIV
06286-038
20mV/DIV
XX
XX
4.3μs TO 0.01%
4.6μs TO 0.001%
AD8224
XX
XXX
XXX (X)
5V/DIV
0.002%/DIV
8.1μs TO 0.01%
9.6μs TO 0.001%
20mV/DIV
4µs/DIV
06286-042
XX
XXX (X)
XXX
06286-045
20µs/DIV
XX
XX
Figure 43. Small Signal Pulse Response,
G = 10, RLOAD = 2 kΩ, CLOAD = 100 pF, VS = ±15 V, VREF = 0 V
Figure 40. Large Signal Pulse Response and Settle Time,
G = 100, RLOAD = 10 kΩ, VS = ±15 V, VREF = 0 V
XX
XXX
XXX (X)
5V/DIV
0.002%/DIV
58μs TO 0.01%
74μs TO 0.001%
20mV/DIV
4µs/DIV
XXX
Figure 41. Large Signal Pulse Response and Settle Time, G = 1000,
RLOAD = 10 kΩ, VS = ±15 V, VREF = 0 V
06286-046
XX
XXX (X)
06286-043
200µs/DIV
XX
XX
XXX
XXX
Figure 44. Small Signal Pulse Response,
G = 100, RLOAD = 2 kΩ, CLOAD = 100 pF, VS = ±15 V, VREF = 0 V
20mV/DIV
Figure 42. Small Signal Pulse Response,
G = 1, RLOAD = 2 kΩ, CLOAD = 100 pF, VS = ±15 V, VREF = 0 V
40µs/DIV
XXX
Figure 45. Small Signal Pulse Response,
G = 1000, RLOAD = 2 kΩ, CLOAD = 100 pF, VS = ±15 V, VREF = 0 V
Rev. B | Page 17 of 28
06286-047
4µs/DIV
XXX
06286-044
20mV/DIV
XXX
XXX
AD8224
20mV/DIV
40µs/DIV
XXX
Figure 46. Small Signal Pulse Response,
G = 1, RLOAD = 2 kΩ, CLOAD = 100 pF, VS = 5 V, VREF = 2.5 V
06286-051
4µs/DIV
XXX
06286-048
20mV/DIV
Figure 49. Small Signal Pulse Response, G = 1000, RLOAD = 2 kΩ,
CLOAD = 100 pF, VS = 5 V, VREF = 2.5 V
4µs/DIV
XXX
Figure 47. Small Signal Pulse Response,
G = 10, RLOAD = 2 kΩ, CLOAD = 100 pF, VS = 5 V, VREF = 2.5 V
SETTLED TO 0.001%
SETTLED TO 0.01%
5
0
06286-049
20mV/DIV
10
0
5
10
20
15
OUTPUT VOLTAGE STEP SIZE (V)
06286-052
XXX
SETTLING TIME (µs)
15
Figure 50. Settling Time vs. Output Voltage Step Size, (G = 1) ±15 V, VREF = 0 V
4µs/DIV
XXX
Figure 48. Small Signal Pulse Response,
G = 100, RLOAD = 2 kΩ, CLOAD = 100 pF, VS = 5 V, VREF = 2.5 V
10
SETTLED TO 0.01%
1
06286-050
20mV/DIV
SETTLED TO 0.001%
1
10
100
1000
GAIN (V/V)
Figure 51. Settling Time vs. Gain for a 10 V Step, VS = ±15 V, VREF = 0 V
Rev. B | Page 18 of 28
06286-053
XXX
SETTLING TIME (µs)
100
AD8224
180
SOURCE
VOUT = 20V p-p
160
VDIFF_OUT
VCM_OUT
80
THERMAL CROSSTALK
VARIES WITH LOAD
140
120
70
GAIN = 1
100
LIMITED BY
MEASUREMENT
SYSTEM
60
50
40
30
80
20
60
1
10
100
1k
10k
100k
1M
FREQUENCY (Hz)
60
GAIN = 1000
40
GAIN = 100
20
GAIN = 10
0
GAIN = 1
10k
100k
1M
10M
FREQUENCY (Hz)
06286-055
–20
1k
1
10
100
1k
10k
100k
FREQUENCY (Hz)
Figure 54. Differential Output Configuration:
Common-Mode Output (CMROUT) vs. Frequency
Figure 52. Channel Separation vs. Frequency,
RLOAD = 2 kΩ, Source Channel at G = 1
–40
100
0
Figure 53. Differential Output Configuration: Gain vs. Frequency
Rev. B | Page 19 of 28
1M
06286-056
10
06286-069
40
GAIN (dB)
CMROUT = 20 log
90
CMROUT (dB)
CHANNEL SEPARATION (dB)
100
SOURCE V OUT
SMALLER TO
AVOID SLEW
RATE LIMIT
GAIN = 1000
AD8224
THEORY OF OPERATION
+VS
+VS
NODE A
R1
24.7kΩ
+VS
RG
+VS
NODE B
–VS
20kΩ
R2
24.7kΩ
–VS
NODE F
+VS
20kΩ
OUTPUT
20kΩ
+VS
+VS
NODE C
J1 Q1
+IN
–VS
A3
VPINCH
NODE E
NODE D
C1
C2
A1
A2
Q2
–IN
J2
VPINCH
+VS
–VS
REF
20kΩ
–VS
–VS
VB
I
06286-057
I
–VS
Figure 55. Simplified Schematic
The AD8224 is a JFET input, monolithic instrumentation amplifier
based on the classic three op amp topology (see Figure 55). Input
Transistor J1 and Input Transistor J2 are biased at a fixed current so
that any input signal forces the output voltages of A1 and A2 to
change accordingly. The input signal creates a current through RG
that flows in R1 and R2 such that the outputs of A1 and A2 provide
the correct, gained signal. Topologically, J1, A1, and R1 and J2, A2,
and R2 can be viewed as precision current feedback amplifiers with
a gain bandwidth of 1.5 MHz. The common-mode voltage and
amplified differential signal from A1 and A2 are applied to a
difference amplifier that rejects the common-mode voltage but
amplifies the differential signal. The difference amplifier employs
20 kΩ laser trimmed resistors that result in an in-amp with a gain
error of less than 0.04%. New trim techniques were developed to
ensure that the CMRR exceeds 86 dB (G = 1).
Using JFET transistors, the AD8224 offers an extremely high
input impedance, extremely low bias currents of 10 pA maximum,
low offset current of 0.6 pA maximum, and no input bias
current noise. In addition, input offset is less than 175 μV
and drift is less than 5 μV/°C. Ease of use and robustness were
considered. A common problem for instrumentation amplifiers
is that at high gains, when the input is overdriven, an excessive
milliampere input bias current can result, and the output can
undergo phase reversal.
Overdriving the input at high gains refers to when the input
signal is within the supply voltages but the amplifier cannot
output the gained signal. For example, at a gain of 100, driving
the amplifier with 10 V on ±15 V constitutes overdriving the
inputs because the amplifier cannot output 100 V.
The AD8224 has none of these problems; its input bias current
is limited to less than 10 μA, and the output does not phase
reverse under overdrive fault conditions.
The AD8224 has extremely low load induced nonlinearity. All
amplifiers that comprise the AD8224 have rail-to-rail output
capability for enhanced dynamic range. The input of the AD8224
can amplify signals with wide common-mode voltages even
slightly lower than the negative supply rail. The AD8224 operates
over a wide supply voltage range. It can operate from either a
single +4.5 V to +36 V supply or a dual ±2.25 V to ±18 V. The
transfer function of the AD8224 is
G =1+
49.4 kΩ
RG
Users can easily and accurately set the gain using a single,
standard resistor. Because the input amplifiers employ a current
feedback architecture, the AD8224 gain bandwidth product
increases with gain, resulting in a system that does not experience
as much bandwidth loss as voltage feedback architectures at
higher gains.
GAIN SELECTION
Placing a resistor across the RG terminals sets the gain of the
AD8224. This is calculated by referring to Table 12 or by using
the following gain equation
Rev. B | Page 20 of 28
RG =
49.4 kΩ
G −1
AD8224
LAYOUT
Table 12. Gains Achieved Using 1% Resistors
The AD8224 is a high precision device. To ensure optimum
performance at the PCB level, care must be taken in the design
of the board layout. The AD8224 pinout is arranged in
a logical manner to aid in this task.
Calculated Gain
1.990
4.984
9.998
19.93
50.40
100.0
199.4
495.0
991.0
Package Considerations
The AD8224 is available in two version s of the 16-lead, 4 mm ×
4 mm LFCSP package: with or without an exposed paddle. Blindly
copying the footprint from another 4 mm × 4 mm LFCSP part
is not recommended because it may not have the same thermal
pad size and leads. Refer to the Outline Dimensions section to
verify that the PCB symbol has the correct dimensions.
The AD8224 defaults to G = 1 when no gain resistor is used.
The tolerance and gain drift of the RG resistor should be added
to the AD8224 specifications to determine the total gain
accuracy of the system. When the gain resistor is not used,
gain error and gain drift are kept to a minimum.
Hidden Paddle Package
REFERENCE TERMINAL
The output voltage of the AD8224 is developed with respect to
the potential on the reference terminal. This is useful when the
output signal needs to be offset to a precise midsupply level. For
example, a voltage source can be tied to the REF1 pin or the
REF2 pin to level-shift the output so that the AD8224 can drive
a single-supply ADC. Pin REFx is protected with ESD diodes
and should not exceed either +VS or −VS by more than 0.5 V.
For best performance, source impedance to the REF terminal
should be kept below 1 Ω. As shown in Figure 55, the reference
terminal, REF, is at one end of a 20 kΩ resistor. Additional
impedance at the REF terminal adds to this 20 kΩ resistor and
results in amplification of the signal connected to the positive
input. The amplification from the additional RREF can be
computed by
The AD8224 is available in an LFCSP package with a hidden
paddle. It is the preferred package for the AD8224. Unlike
chip scale packages where the pad limits routing capability,
this package allows routes and vias directly underneath the
chip, so that the full space savings of the small LFCSP can be
realized. Although the package has no metal in the center of
the part, the manufacturing process does leave a very small
section of exposed metal at each of the package corners, shown
in Figure 57 as well as Figure 68 in the Outline Dimensions
section. This metal is connected to +VS through the part.
Because of a possibility of a short, vias should not be placed
underneath these exposed metal tabs.
HIDDEN
PADDLE
EXPOSED LEAD
FRAME TABS
BOTTOM VIEW
2 (20 kΩ + RREF )
NOTES
1. EXPOSED LEAD FRAME TABS AT THE FOUR CORNERS
OF THE PACKAGE ARE INTERNALLY CONNECTED TO
+VS. REFER TO THE OUTLINE DIMENSIONS PAGE, FOR
FURTHER INFORMATION ON PACKAGE AVAILABILITY.
40 kΩ + RREF
Figure 57. Hidden Paddle Package: Bottom View
Only the positive signal path is amplified; the negative path is
unaffected. This uneven amplification degrades the CMRR of
the amplifier.
INCORRECT
CORRECT
AD8224
Exposed Paddle Package
The AD8224 4 mm × 4 mm LFCSP is also available with an
exposed thermal paddle package version. This pad is connected
internally to +VS. The pad can either be left unconnected or
connected to the positive supply rail. Space between the leads
and thermal pad should be kept as wide as possible for the best
bias current performance. To maintain the AD8224 ultralow
bias current performance, the thermal pad area can be reduced
to extend the gap between the leads and the pad.
CORRECT
AD8224
VREF
AD8224
VREF
+
+
AD8224
–
–
Figure 56. Driving the Reference Pin
06286-058
VREF
OP2177
06286-101
1% Standard Table Value of RG (Ω)
49.9 k
12.4 k
5.49 k
2.61 k
1.00 k
499
249
100
49.9
To preserve maximum pin compatibility with other dual
instrumentation amplifiers, such as the AD8222, leave the pad
unconnected. This can be done by not soldering the paddle at
all or by soldering the part to a landing that is a not connected
to any other net. For high vibration applications, a landing is
recommended.
Rev. B | Page 21 of 28
AD8224
Because the AD8224 dissipates little power, heat dissipation is
rarely an issue. If improved heat dissipation is desired (for example,
when driving heavy loads), connect the exposed pad to the
positive supply rail. For the best heat dissipation performance,
the positive supply rail should be a plane in the board. See
the Thermal Resistance section for more information.
0.1µF
Common-Mode Rejection over Frequency
The AD8224 has a higher CMRR over frequency than typical
in-amps, which gives it greater immunity to disturbances, such
as line noise and its associated harmonics. A well-implemented
layout is required to maintain this high performance. Input
source impedances should be matched closely. Source resistance
should be placed close to the inputs so that it interacts with as
little parasitic capacitance as possible.
16
15
14
13
AD8224
1
12
2
11
3
10
4
9
RG
RG
Parasitics at the RGx pins can also affect CMRR over frequency.
The PCB should be laid out so that the parasitic capacitances at
each pin match. Traces from the gain setting resistor to the RGx
pins should be kept short to minimize parasitic inductance.
5
6
7
8
0.1µF
Reference
06286-059
Errors introduced at the reference terminal feed directly to
the output. Take care to tie the REFx pins to the appropriate
local ground.
Figure 58. Example Layout
Power Supplies
A stable dc voltage should be used to power the instrumentation
amplifier. Noise on the supply pins can adversely affect
performance.
The AD8224 has two positive supply pins (Pin 5 and Pin 16)
and two negative supply pins (Pin 8 and Pin 13). While the part
functions with only one pin from each supply pair connected,
both pins should be connected for specified performance and
optimum reliability.
The AD8224 should be decoupled with 0.1 μF bypass capacitors,
one for each supply. Place the positive supply decoupling
capacitor near Pin 16, and the negative supply decoupling
capacitor near Pin 8. Each supply should also be decoupled with
a 10 μF tantalum capacitor. The tantalum capacitor can be
placed further away from the AD8224 and can generally be
shared by other precision integrated circuits. Figure 58 shows an
example layout.
SOLDER WASH
The solder process can leave flux and other contaminants on
the board. When these contaminants are between the AD8224
leads and thermal pad, they can create leakage paths that are
larger than the AD8224 bias currents. A thorough washing
process removes these contaminants and restores the device’s
excellent bias current performance.
INPUT BIAS CURRENT RETURN PATH
The input bias current of the AD8224 must have a return path
to common. When the source, such as a transformer, cannot
provide a return current path, one should be created, as shown
in Figure 59.
INPUT PROTECTION
All terminals of the AD8224 are protected against ESD. ESD
protection is guaranteed to 4 kV (human body model). In
addition, the input structure allows for dc overload conditions
a diode drop above the positive supply and a diode drop below
the negative supply. Voltages beyond a diode drop of the
supplies cause the ESD diodes to conduct and enable current to
flow through the diode. Therefore, an external resistor should
be used in series with each of the inputs to limit current for
voltages above +Vs. In either scenario, the AD8224 safely
handles a continuous 6 mA current at room temperature.
Rev. B | Page 22 of 28
AD8224
For applications where the AD8224 encounters extreme
overload voltages, as in cardiac defibrillators, external series
resistors and low leakage diode clamps, such as BAV199Ls,
FJH1100s, or SP720s, should be used.
INCORRECT
The relationship between external, matched series resistors and the
internal gate capacitance is expressed as
FilterFreq DIFF =
CORRECT
+VS
FilterFreqCM =
+VS
AD8224
REF
–VS
1
2πRCG
To eliminate high frequency common-mode signals while using
smaller source resistors, a low-pass RC network can be placed at
the input of the instrumentation amplifier (see Figure 61). The
filter limits the input signal bandwidth according to the
following relationship:
AD8224
REF
–VS
TRANSFORMER
FilterFreq DIFF =
TRANSFORMER
+VS
+VS
C
FilterFreqCM =
C
C
R
1
fHIGH-PASS = 2πRC
AD8224
AD8224
C
REF
REF
R
–VS
06286-060
–VS
CAPACITIVELY COUPLED
CAPACITIVELY COUPLED
Figure 59. Creating an IBIAS Path
1
2πRCG
1
2πR(2 CD + CC + CG )
1
2πR(CC + CG )
Mismatched CC capacitors result in mismatched low-pass filters.
The imbalance causes the AD8224 to treat what would have
been a common-mode signal as a differential signal. To reduce
the effect of mismatched external CC capacitors, select a value of
CD greater than 10 times CC. This sets the differential filter
frequency lower than the common-mode frequency.
+15V
RF INTERFERENCE
0.1µF
CC
+IN
4.02kΩ
CD
R
REF
–IN
CC
+
VOUT
AD8224
10nF
4.02kΩ
10µF
+
1nF
R
+15V
0.1µF
10µF
1nF
10µF
0.1µF
+
–15V
06286-062
RF rectification is often a problem in applications where there are
large RF signals. The problem appears as a small dc offset voltage.
The AD8224 by its nature has a 5 pF gate capacitance (CG) at its
inputs. Matched series resistors form a natural low-pass filter that
reduces rectification at high frequency (see Figure 60).
Figure 61. RFI Suppression
+IN
COMMON-MODE INPUT VOLTAGE RANGE
CG
AD8224
–VS
R
–IN
VOUT
CG
–VS
0.1µF
REF
10µF
+
–15V
06286-061
R
The 3-op amp architecture of the AD8224 applies gain and then
removes the common-mode voltage. Therefore, internal nodes
in the AD8224 experience a combination of both the gained
signal and the common-mode signal. This combined signal can
be limited by the voltage supplies even when the individual input
and output signals are not. Figure 24 through Figure 27 show the
allowable common-mode input voltage ranges for various
output voltages, supply voltages, and gains.
Figure 60. RFI Filtering Without External Capacitors
Rev. B | Page 23 of 28
AD8224
APPLICATIONS INFORMATION
+IN
An instrumentation amplifier is often used in front of an ADC
to provide CMRR and additional conditioning such as a voltage
level shift and gain (see Figure 62). In this example, a 2.7 nF
capacitor and a 500 Ω resistor create an antialiasing filter for the
AD7685. The 2.7 nF capacitor also serves to store and deliver
the necessary charge to the switched capacitor input
of the ADC. The 500 Ω series resistor reduces the burden
of the 2.7 nF load from the amplifier. However, large source
impedance in front of the ADC can degrade the total harmonic
distortion (THD).
RG
–IN
+
AD8224
+OUT
20kΩ
–
–
AD8224
33pF
+
+IN2
REF2
–OUT
06286-064
DRIVING AN ADC
Figure 63. Differential Circuit Schematic
Setting the Common-Mode Voltage
For applications where THD performance is critical, the series
resistor needs to be small. At worst, a small series resistor can
load the AD8224, potentially causing the output to overshoot
or ring. In such cases, a buffer amplifier, such as the AD8615
should be used after the AD8224 to drive the ADC.
The output common-mode voltage is set by the average of +IN2
and REF2. The transfer function is
VCM_OUT = (V+OUT + V−OUT)/2 = (V+IN2 + VREF2)/2
+IN2 and REF2 have different properties that allow the
reference voltage to be easily set for a wide variety of applications.
+IN2 has high impedance but cannot swing to the positive
supply rail. REF2 must be driven with a low impedance but
can go 300 mV beyond the supply rails.
+5V
+
0.1µF
ADR435
+5V
4.7µF
+IN
±50mV
500Ω
AD8224
1.07kΩ
REF
2.7nF
AD7685
–IN
06286-063
+2.5V
Figure 62. Driving an ADC in a Low Frequency Application
2-Channel Differential Output Using a Dual Op Amp
DIFFERENTIAL OUTPUT
The differential configuration of the AD8224 has the same
excellent dc precision specifications as the single-ended output
configuration and is recommended for applications in the
frequency range of dc to 1 MHz.
The circuit configuration, outlined in Table 4 and Table 7, refers
to the configuration shown in Figure 63 only. The circuit includes
an RC filter that maintains the stability of the loop.
The transfer function for the differential output is
A common application sets the common-mode output voltage
to the midscale of a differential ADC. In this case, the ADC
reference voltage is sent to the +IN2 terminal, and ground is
connected to the REF2 terminal. This produces a commonmode output voltage of half the ADC reference voltage.
Another differential output topology is shown in Figure 64.
Instead of a second in-amp, ½ of a dual OP2177 op amp creates
the inverted output. Because the OP2177 comes in an MSOP,
this configuration allows the creation of a dual-channel,
precision differential output in-amp with little board area.
Errors from the op amp are common to both outputs and are,
thus, common mode. Errors from mismatched resistors also
create a common-mode dc offset. Because these errors are
common mode, they are likely to be rejected by the next
device in the signal chain.
VDIFF_OUT = V+OUT − V−OUT = (V+IN − V−IN) × G
+IN
AD8224
where:
G =1+
+OUT
–IN
49.4 kΩ
REF
RG
4.99kΩ
4.99kΩ
VREF
+
–
OP2177
–OUT
Figure 64. Differential Output Using Op Amp
Rev. B | Page 24 of 28
06286-065
10µF
AD8224
+12V
10µF
+
0.1µF
+5V
1kΩ
+IN
100pF
NPO
5%
0.1µF
+OUT
1000pF
AD8224
–IN
+IN2
100pF
NPO
5%
–OUT
REF2
IN–
2.7nF
2.7nF
+
AD7688
GND
REF
10µF
X5R
+12V
+5V REF
10µF
VDD
IN+
806Ω
(DIFF OUT)
1kΩ
806Ω
0.1µF
0.1µF
VIN
–12V
+5V REF
VOUT
0.1µF
ADR435
06286-066
GND
Figure 65. Driving a Differential ADC
The AD8224 can be configured in differential output mode
to drive a differential ADC. Figure 65 illustrates several of the
concepts.
First Antialiasing Filter
The 1 kΩ resistor, 1000 pF capacitor, and 100 pF capacitors in
front of the in-amp form a 76 kHz filter. This is the first of two
antialiasing filters in the circuit and helps to reduce the noise of
the system. The 100 pF capacitors protect against commonmode RFI signals. Note that they are 5% COG/NPO types.
These capacitors match well over time and temperature,
which keeps the CMRR of the system high over frequency.
Second Antialiasing Filter
An 806 Ω resistor and a 2.7 nF capacitor are located between
each AD8224 output and ADC input. These components
create a 73 kHz low-pass filter for another stage of antialiasing
protection.
These four elements also isolate the ADC from loading the
AD8224. The 806 Ω resistor shields the AD8224 from the
switched capacitor input of the ADC, which looks like a timevarying load. The 2.7 nF capacitor provides a charge to the
switched capacitor front end of the ADC. If the application
requires a lower frequency antialiasing filter, increase the value
of the capacitor rather than the resistor.
However, other converters have less robust inputs and may need
the added protection.
Reference
The ADR435 supplies a reference voltage to both the ADC and
the AD8224. Because REF2 on the AD8224 is grounded, the
common-mode output voltage is precisely half the reference
voltage, exactly where it needs to be for the ADC.
DRIVING CABLING
All cables have a certain capacitance per unit length, which
varies widely with cable type. The capacitive load from the cable
may cause peaking in the AD8224 output response. To reduce
peaking, use a resistor between the AD8224 and the cable.
Because cable capacitance and desired output response vary
widely, this resistor is best determined empirically. A good
starting point is 50 Ω.
The AD8224 operates at a low enough frequency that
transmission line effects are rarely an issue; therefore, the
resistor need not match the characteristic impedance of
the cable.
The 806 Ω resistors can also protect an ADC from overvoltages.
Because the AD8224 runs on wider supply voltages than a
typical ADC, there is a possibility of overdriving the ADC. This
is not an issue with a PulSAR® converter, such as the AD7688.
Its input can handle a 130 mA overdrive, which is much higher
than the short-circuit limit of the AD8224.
AD8224
(DIFF OUT)
AD8224
(SINGLE OUT)
06286-067
DRIVING A DIFFERENTIAL INPUT ADC
Figure 66. Driving a Cable
Rev. B | Page 25 of 28
AD8224
OUTLINE DIMENSIONS
4.00
BSC SQ
0.50
0.40
0.30
0.60 MAX
12 13
PIN 1
INDICATOR
3.75
BSC SQ
1
16
EXPOSED
PAD
0.65
BSC
TOP VIEW
4
2.65
2.50 SQ
2.35
5
8
9
PIN 1
INDICATOR
0.25 MIN
1.95 BCS
0.80 MAX
0.65 TYP
BOTTOM VIEW
FOR PROPER CONNECTION OF
THE EXPOSED PAD, REFER TO
THE PIN CONFIGURATION AND
FUNCTION DESCRIPTIONS
SECTION OF THIS DATA SHEET.
0.05 MAX
0.02 NOM
COPLANARITY
0.20 REF
0.08
031006-A
12° MAX
1.00
0.85
0.80
SEATING
0.30
PLANE
0.23
0.18
COMPLIANT TO JEDEC STANDARDS MO-220-VGGC.
Figure 67. 16-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
4 mm × 4 mm Body, Very Thin Quad
(CP-16-13)
Dimensions are shown in millimeters
0.60 MAX
4.00
BSC SQ
0.60 MAX
PIN 1
INDICATOR
3.75
BCS SQ
0.65
BSC
13
12
TOP VIEW
SEATING
PLANE
12° MAX
8
5
4
BOTTOM VIEW
0.80 MAX
0.65 TYP
0.35
0.30
0.25
0.05 MAX
0.02 NOM
COPLANARITY
0.08
0.20 REF
COMPLIANT TO JEDEC STANDARDS MO-263-VBBC
Figure 68. 16-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
4 mm × 4 mm Body, Very Thin Quad, with Hidden Paddle
CP-16-19
Dimensions shown in millimeters
Rev. B | Page 26 of 28
062309-B
1.00
0.85
0.80
1
1.95 REF
SQ
9
0.75
0.60
0.50
16
AD8224
ORDERING GUIDE
Model 1
AD8224ACPZ-R7
AD8224ACPZ-RL
AD8224ACPZ-WP
AD8224BCPZ-R7
AD8224BCPZ-RL
AD8224BCPZ-WP
AD8224HACPZ-R7
AD8224HACPZ-RL
AD8224HACPZ-WP
AD8224HBCPZ-R7
AD8224HBCPZ-RL
AD8224HBCPZ-WP
AD8224-EVALZ
1
Temperature Range
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
Product Description
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
Evaluation Board
Z = RoHS Compliant Part.
Rev. B | Page 27 of 28
Package Option
CP-16-13
CP-16-13
CP-16-13
CP-16-13
CP-16-13
CP-16-13
CP-16-19
CP-16-19
CP-16-19
CP-16-19
CP-16-19
CP-16-19
AD8224
NOTES
©2007–2010 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D06286-0-5/10(B)
Rev. B | Page 28 of 28