LINER LTC3707 High effi ciency, 2-phase synchronous step-down switching regulator Datasheet

LTC3707
High Efficiency, 2-Phase
Synchronous Step-Down
Switching Regulator
DESCRIPTION
FEATURES
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The LTC®3707 is a high performance dual step-down
switching regulator controller that drives N-channel
synchronous power MOSFET stages. A constant frequency
current mode architecture allows adjustment of the
frequency up to 300kHz. Power loss and noise due to the
ESR of the input capacitors are minimized by operating
the two controller output stages out of phase.
180° Phased Dual Controllers Reduce Required
Input Capacitance and Power Supply Induced Noise
OPTI-LOOP® Compensation Minimizes COUT
±1.5% Output Voltage Accuracy over Temperature
Dual N-Channel MOSFET Synchronous Drive
Power Good Output Voltage Monitor
DC Programmed Fixed Frequency 150kHz to 300kHz
Wide VIN Range: 4.5V to 28V Operation
Very Low Dropout Operation: 99% Duty Cycle
Adjustable Soft-Start Current Ramping
Foldback Output Current Limiting
Latched Short-Circuit Shutdown with Defeat Option
Output Overvoltage Protection
Remote Output Voltage Sense
Low Shutdown IQ: 20μA
5V and 3.3V Standby Regulators
Selectable Constant Frequency, Burst Mode®
Operation or PWM Operation
Small 28-Lead Narrow SSOP Package
OPTI-LOOP compensation allows the transient response to
be optimized over a wide range of output capacitance and
ESR values. The precision 0.8V reference and power good
output indicator are compatible with future microprocessor
generations, and a wide 3.5V to 30V input supply range
encompasses all battery chemistries.
A RUN/SS pin for each controller provides both soft-start
and optional timed, short-circuit shutdown. Current
foldback limits MOSFET dissipation during short-circuit
conditions when overcurrent latchoff is disabled. Output
overvoltage protection circuitry latches on the bottom
MOSFET until VOUT returns to normal. The FCB mode
pin can select among Burst Mode operation, constant
frequency mode and continuous inductor current mode
or regulate a secondary winding.
APPLICATIONS
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Notebook and Palmtop Computers, PDAs
Battery Chargers
Portable Instruments
Battery-Operated Digital Devices
DC Power Distribution Systems
TYPICAL APPLICATION
L, LT, LTC, LTM, Burst Mode, and OPTI-LOOP are registered trademarks of Linear Technology
Corporation. All other trademarks are the property of their respective owners. Protected by U.S.
Patents including 5481178, 5705919, 5929620, 6144194, 6177787, 6304066, 6580258.
+
4.7μF
D3
VIN
M1
L1
6.3μH
CB1, 0.1μF
BOOST1
BG1
BOOST2
VOUT1
5V
5A
+
COUT1
47μF
6V
SP
R1
20k
1%
CC1
220pF
RC1
15k
L2
6.3μH
CB2, 0.1μF
M4
BG2
D2
PGND
SENSE1+
SENSE2+
SENSE1–
SENSE2–
VOSENSE1
VOSENSE2
1000pF
R2
105k
1%
VIN
5.2V TO 28V
CIN
22μF
50V
CERAMIC
SW2
LTC3707
SGND
RSENSE1
0.01Ω
M3
TG2
SW1
M2
D1
D4
INTVCC
TG1
1μF
CERAMIC
RSENSE2
0.01Ω
1000pF
ITH1
RUN/SS1
CSS1
0.1μF
ITH2
RUN/SS2
CSS2
0.1μF
CC2
220pF
RC2
15k
R3
20k
1%
M1, M2, M3, M4: FDS6680A
Figure 1. High Efficiency Dual 5V/3.3V Step-Down Converter
R4
63.4k
1%
VOUT2
3.3V
5A
COUT
56μF
6V
SP
+
3707 F01
3707fb
1
LTC3707
ABSOLUTE MAXIMUM RATINGS
PIN CONFIGURATION
(Note 1)
TOP VIEW
Input Supply Voltage (VIN) .........................30V to – 0.3V
Top Side Driver Voltages
(BOOST1, BOOST2) ................................... 36V to –0.3V
Switch Voltage (SW1, SW2) ......................... 30V to –5V
INTVCC, EXTVCC, RUN/SS1, RUN/SS2, (BOOST1-SW1),
(BOOST2-SW2), PGOOD.............................. 7V to –0.3V
SENSE1+, SENSE2+, SENSE1–,
SENSE2– Voltages .........................(1.1)INTVCC to –0.3V
FREQSET, STBYMD, FCB Voltage ......... INTVCC to –0.3V
ITH1, ITH2, VOSENSE1, VOSENSE2 Voltages ... 2.7V to –0.3V
Peak Output Current <10μs (TG1, TG2, BG1, BG2) .....3A
INTVCC Peak Output Current ................................. 40mA
Operating Temperature Range (Note 2).... –40°C to 85°C
Junction Temperature (Note 3) ............................. 125°C
Storage Temperature Range................... –65°C to 150°C
Lead Temperature (Soldering, 10 sec) .................. 300°C
RUN/SS1
1
28 PGOOD
SENSE1+
2
27 TG1
SENSE1–
3
26 SW1
VOSENSE1
4
25 BOOST1
FREQSET
5
24 VIN
STBYMD
6
23 BG1
FCB
7
22 EXTVCC
ITH1
8
21 INTVCC
SGND
9
20 PGND
3.3VOUT 10
19 BG2
ITH2 11
18 BOOST2
VOSENSE2 12
17 SW2
SENSE2– 13
16 TG2
SENSE2+ 14
15 RUN/SS2
GN PACKAGE
28-LEAD PLASTIC SSOP
TJMAX = 125°C, θJA = 95°C/W
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3707EGN#PBF
LTC3707EGN#TRPBF
3707EGN
28-Lead Plastic SSOP
–40°C to 85°C
LTC3707IGN#PBF
LTC3707IGN#TRPBF
3707IGN
28-Lead Plastic SSOP
–40°C to 85°C
LEAD BASED FINISH
TAPE AND REEL
PART MARKING
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3707EGN
LTC3707EGN#TR
3707EGN
28-Lead Plastic SSOP
–40°C to 85°C
LTC3707IGN
LTC3707IGN#TR
3707IGN
28-Lead Plastic SSOP
–40°C to 85°C
Consult LTC Marketing for parts specified with wider operating temperature ranges.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, VRUN/SS1, 2 = 5V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
VOSENSE1, 2
Regulated Feedback Voltage
(Note 4); ITH1, 2 Voltage = 1.2V
IOSENSE1, 2
Feedback Current
(Note 4)
VREFLNREG
Reference Voltage Line Regulation
VIN = 3.6V to 30V (Note 4)
VLOADREG
Output Voltage Load Regulation
(Note 4)
Measured in Servo Loop; ΔITH Voltage = 1.2V to 0.7V
Measured in Servo Loop; ΔITH Voltage = 1.2V to 2.0V
gm1,2
Transconductance Amplifier gm
ITH1, 2 = 1.2V; Sink/Source 5μA; (Note 4)
l
l
l
MIN
TYP
0.788
0.800
0.812
V
–5
–50
nA
0.002
0.02
%/V
0.1
–0.1
0.5
–0.5
%
%
1.3
MAX UNITS
mmho
3707fb
2
LTC3707
which apply over the full operating
ELECTRICAL
CHARACTERISTICS The l denotes the specifi=cations
temperature range, otherwise specifications are at T = 25°C. V = 15V, V
5V unless otherwise noted.
A
IN
RUN/SS1, 2
SYMBOL
PARAMETER
CONDITIONS
gmGBW1, 2
Transconductance Amplifier GBW
ITH1, 2 = 1.2V; (Note 4)
IQ
Input DC Supply Current
Normal Mode
Standby
Shutdown
(Note 5)
EXTVCC Tied to VOUT1 = 5V
VRUN/SS1, 2 = 0V, VSTBYMD > 2V
VRUN.SS1, 2 = 0V, VSTBYMD = Open
MIN
TYP
MAX UNITS
3
l
MHz
350
125
20
35
μA
μA
μA
0.76
0.800
0.84
V
–0.30
–0.18
–0.1
μA
4.3
4.8
V
3.5
4
V
0.84
0.86
0.88
V
–60
VFCB
Forced Continuous Threshold
IFCB
Forced Continuous Pin Current
VFCB = 0.85V
VBINHIBIT
Burst Inhibit (Constant Frequency)
Threshold
Measured at FCB pin
UVLO
Undervoltage Lockout
VIN Ramping Down
l
VOVL
Feedback Overvoltage Lockout
Measured at VOSENSE1, 2
l
ISENSE
Sense Pins Total Source Current
(Each Channel); VSENSE1–, 2– = VSENSE1+, 2+ = 0V
–90
VSTBYMD MS
Master Shutdown Threshold
VSTBYMD Ramping Down
0.4
VSTBYMD KA
Keep-Alive Power On-Threshold
VSTBYMD Ramping Up, RUNSS1, 2 = 0V
DFMAX
Maximum Duty Factor
In Dropout
98
99.4
%
IRUN/SS1, 2
Soft-Start Charge Current
VRUN/SS1, 2 = 1.9V
0.5
1.2
μA
VRUN/SS1, 2 ON RUN/SS Pin ON Threshold
VRUN/SS1, VRUN/SS2, Rising
1.0
1.5
2.0
V
VRUN/SS1, 2 LT
RUN/SS Pin Latchoff Arming
Threshold
VRUN/SS1, VRUN/SS2, Rising from 3V
4.1
4.75
V
ISCL1, 2
RUN/SS Discharge Current
Soft Short Condition VOSENSE1, 2 = 0.5V;
VRUN/SS1, 2 = 4.5V
2
4
μA
ISDLHO
Shutdown Latch Disable Current
VOSENSE1, 2 = 0.5V
1.6
5
μA
VSENSE(MAX)
Maximum Current Sense Threshold
VOSENSE1, 2 = 0.7V, VOSENSE1–, 2– = 5V
75
75
85
88
mV
mV
TG1, 2 tr
TG1, 2 tf
TG Transition Time:
Rise Time
Fall Time
(Note 6)
CLOAD = 3300pF
CLOAD = 3300pF
60
60
110
110
ns
ns
BG1, 2 tr
BG1, 2 tf
BG Transition Time:
Rise Time
Fall Time
(Note 6)
CLOAD = 3300pF
CLOAD = 3300pF
50
50
110
100
ns
ns
Top Gate Off to Bottom Gate On Delay
Synchronous Switch-On Delay Time
CLOAD = 3300pF Each Driver
80
ns
BG/TG t2D
Bottom Gate Off to Top Gate On Delay
Top Switch-On Delay Time
CLOAD = 3300pF Each Driver
80
ns
tON(MIN)
Minimum On-Time
Tested with a Square Wave (Note 7)
180
ns
TG/BG t1D
0.6
1.5
0.5
l
65
62
μA
V
2
V
INTVCC Linear Regulator
VINTVCC
Internal VCC Voltage
6V < VIN < 30V, VEXTCC = 4V
4.8
5.0
5.2
V
VLDO INT
INTVCC Load Regulation
ICC = 0 to 20mA, VEXTVCC = 4V
0.2
2.0
%
VLDO EXT
EXTVCC Voltage Drop
ICC = 20mA, VEXTVCC = 5V
100
200
mV
VEXTVCC
EXTVCC Switchover Voltage
ICC = 20mA, EXTVCC Ramping Positive
VLDOHYS
EXTVCC Hysteresis
l
4.5
4.7
V
0.2
V
Oscillator
fOSC
Oscillator Frequency
VFREQSET = Open (Note 8)
190
220
250
kHz
fLOW
Lowest Frequency
VFREQSET = 0V
120
140
160
kHz
3707fb
3
LTC3707
which apply over the full operating
ELECTRICAL
CHARACTERISTICS The l denotes the specifi=cations
5V unless otherwise noted.
temperature range, otherwise specifications are at T = 25°C. V = 15V, V
A
IN
RUN/SS1, 2
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX UNITS
fHIGH
Highest Frequency
VFREQSET = 2.4V
280
310
360
kHz
IFREQSET
FREQSET Input Current
VFREQSET = 0V
–2
–1
μA
3.35
3.45
V
3.3V Linear Regulator
l
V3.3OUT
3.3V Regulator Output Voltage
No Load
3.20
V3.3IL
3.3V Regulator Load Regulation
I3.3 = 0 to 10mA
0.5
2
%
V3.3VL
3.3V Regulator Line Regulation
6V < VIN < 30V
0.05
0.2
%
VPGL
PGOOD Voltage Low
IPGOOD = 2mA
0.1
0.3
V
IPGOOD
PGOOD Leakage Current
VPGOOD = 5V
1
μA
VPG
PGOOD Trip Level, Either Controller
VOSENSE Respect to Set Output Voltage
VOSENSE Ramping Negative
VOSENSE Ramping Positive
–9.5
9.5
%
%
PGOOD Output
–6
6
–7.5
7.5
Note 4: The LTC3707 is tested in a feedback loop that servos VITH1, 2 to a
specified voltage and measures the resultant VOSENSE1, 2.
Note 5: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. See Applications Information.
Note 6: Rise and fall times are measured using 10% and 90% levels. Delay
times are measured using 50% levels.
Note 7: The IC minimum on-time is tested under an ideal condition
without external power FETs. It can be different when the IC is working in
an actual circuit. See Minimum On-Time Considerations in the Application
Information section.
Note 8: VFREQSET pin internally tied to a 1.19V reference through a
large resistance.
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC3707E is guaranteed to meet performance specifications
from 0°C to 85°C. Specifications over the – 40°C to 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls. The LTC3707I is guaranteed to meet
performance specifications over the full – 40°C to 85°C operating
temperature range.
Note 3: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formula:
LTC3707EGN = TJ = TA + (PD • 85°C/W)
TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency vs Output Current
(Figure 13)
Efficiency vs Output Current
and Mode (Figure 13)
100
100
Burst Mode
90 OPERATION
50
40
30
CONSTANT
FREQUENCY
(BURST DISABLE)
20
VIN = 20V
70
60
VIN = 15V
VOUT = 5V
10
0
0.001
80
90
VIN = 10V
VIN = 15V
EFFICIENCY (%)
FORCED
CONTINUOUS
MODE (PWM)
EFFICIENCY (%)
EFFICIENCY (%)
60
100
VIN = 7V
90
80
70
VIN = 15V
VOUT = 5V
Efficiency vs Input Voltage
(Figure 13)
0.1
0.01
1
OUTPUT CURRENT (A)
10
3707 G01
80
70
60
VOUT = 5V
IOUT = 3A
50
0.001
0.1
0.01
1
OUTPUT CURRENT (A)
50
10
3707 G02
5
15
INPUT VOLTAGE (V)
25
30
3707 G03
3707fb
4
LTC3707
TYPICAL PERFORMANCE CHARACTERISTICS
Supply Current vs Input Voltage
and Mode (Figure 13)
1000
200
600
BOTH
CONTROLLERS ON
400
200
5.05
INTVCC AND EXTVCC SWITCH VOLTAGE (V)
EXTVCC VOLTAGE DROP (mV)
800
SUPPLY CURRENT (μA)
INTVCC and EXTVCC Switch
Voltage vs Temperature
EXTVCC Voltage Drop
150
100
50
STANDBY
SHUTDOWN
0
0
5
20
15
10
INPUT VOLTAGE (V)
25
0
30
10
0
20
CURRENT (mA)
30
4.95
4.90
4.85
4.80
4.70
–50 –25
40
100
125
3707 G06
75
80
ILOAD = 1mA
70
5.0
60
4.8
4.7
50
VSENSE (mV)
4.9
VSENSE (mV)
INTVCC VOLTAGE (V)
50
25
75
0
TEMPERATURE (°C)
Maximum Current Sense
Threshold vs Percent of Nominal
Output Voltage (Foldback)
Maximum Current Sense
Threshold vs Duty Factor
Internal 5V LDO Line Reg
25
4.6
50
40
30
20
4.5
10
0
4.4
0
5
20
15
10
INPUT VOLTAGE (V)
0
30
25
20
40
60
DUTY FACTOR (%)
80
Maximum Current Sense
Threshold vs VRUN/SS (Soft-Start)
80
0
100
50
100
0
25
75
PERCENT ON NOMINAL OUTPUT VOLTAGE (%)
3707 G08
3707 G07
3707 G09
Current Sense Threshold
vs ITH Voltage
Maximum Current Sense Threshold
vs Sense Common Mode Voltage
80
VSENSE(CM) = 1.6V
90
80
70
76
60
40
VSENSE (mV)
60
VSENSE (mV)
VSENSE (mV)
EXTVCC SWITCHOVER THRESHOLD
4.75
3707 G05
3707 G04
5.1
INTVCC VOLTAGE
5.00
72
68
50
40
30
20
10
20
0
64
–10
–20
0
0
1
2
3
4
VRUN/SS (V)
5
6
3707 G10
60
0
1
3
4
2
COMMON MODE VOLTAGE (V)
5
3707 G11
–30
0
0.5
1
1.5
VITH (V)
2
2.5
3707 G12
3707fb
5
LTC3707
TYPICAL PERFORMANCE CHARACTERISTICS
Load Regulation
VITH VS VRUN/SS
2.5
FCB = 0V
VIN = 15V
FIGURE 1
SENSE Pins Total Source Current
100
VOSENSE = 0.7V
2.0
–0.2
50
ISENSE (μA)
–0.1
VITH (V)
NORMALIZED VOUT (%)
0.0
1.5
1.0
–0.3
0
–50
0.5
–0.4
0
1
0
3
2
LOAD CURRENT (A)
4
5
0
2
1
3
4
5
–100
6
2
0
VRUN/SS (V)
4
3707 G14
3707 G13
Maximum Current Sense
Threshold vs Temperature
3707 G15
Dropout Voltage vs Output Current
(Figure 13)
80
4
6
VSENSE COMMON MODE VOLTAGE (V)
RUN/SS Current vs Temperature
1.8
VOUT = 5V
1.6
76
74
3
RUN/SS CURRENT (μA)
DROPOUT VOLTAGE (V)
VSENSE (mV)
78
2
RSENSE = 0.015Ω
1
1.4
1.2
1.0
0.8
0.6
0.4
72
RSENSE = 0.010Ω
0.2
70
–50
0
–25
50
25
0
75
TEMPERATURE (°C)
100
0
125
0.5
1.0 1.5 2.0 2.5 3.0
OUTPUT CURRENT (A)
3.5
4.0
0
–50
0
25
50
75
TEMPERATURE (°C)
3707 G18
3707 G17
Soft-Start Up (Figure 13)
100
125
3707 G25
Load Step (Figure 13)
Load Step (Figure 13)
VOUT
5V/DIV
–25
VOUT
200mV/DIV
VOUT
200mV/DIV
IOUT
2A/DIV
IOUT
2A/DIV
VRUN/SS
5V/DIV
IOUT
2A/DIV
VIN = 15V
VOUT = 5V
5ms/DIV
3707 G19
20μs/DIV
VIN = 15V
VOUT = 5V
LOAD STEP = 0A TO 3A
Burst Mode OPERATION
3707 G20
20μs/DIV
VIN = 15V
VOUT = 5V
LOAD STEP = 0A TO 3A
CONTINUOUS OPERATION
3707 G21
3707fb
6
LTC3707
TYPICAL PERFORMANCE CHARACTERISTICS
Input Source/Capacitor
Instantaneous Current (Figure 13)
IIN
2A/DIV
VIN
200mV/DIV
Constant Frequency (Burst Inhibit)
Operation (Figure 13)
Burst Mode Operation (Figure 13)
VOUT
20mV/DIV
VOUT
20mV/DIV
IOUT
0.5A/DIV
IOUT
0.5A/DIV
VSW1
10V/DIV
VSW2
10V/DIV
3707 G22
1μs/DIV
VIN = 15V
VOUT = 5V
IOUT5 = IOUT3.3 = 2A
VIN = 15V
VOUT = 5V
VFCB = OPEN
IOUT = 20mA
10
350
29
27
50
25
0
75
TEMPERATURE (°C)
100
125
300
8
FREQUENCY (kHz)
EXTVCC SWITCH RESISTANCE (Ω)
31
6
4
VFREQSET = OPEN
200
VFREQSET = 0V
150
100
50
0
–50
50
25
0
75
TEMPERATURE (°C)
–25
100
0
–50 –25
125
50
25
75
0
TEMPERATURE (°C)
Undervoltage Lockout
vs Temperature
100
125
3707 G28
3707 G27
3707 G26
Shutdown Latch Thresholds
vs Temperature
3.50
4.5
3.45
3.40
3.35
3.30
3.25
3.20
–50 –25
250
2
SHUTDOWN LATCH THRESHOLDS (V)
–25
3707 G24
VFREQSET = 5V
33
25
–50
2μs/DIV
Oscillator Frequency
vs Temperature
VOUT = 5V
UNDERVOLTAGE LOCKOUT (V)
CURRENT SENSE INPUT CURRENT (μA)
VIN = 15V
VOUT = 5V
VFCB = 5V
IOUT = 20mA
EXTVCC Switch Resistance
vs Temperature
Current Sense Pin Input Current
vs Temperature
35
3707 G23
10μs/DIV
50
25
75
0
TEMPERATURE (°C)
100
125
3707 G29
LATCH ARMING
4.0
3.5
3.0
LATCHOFF
THRESHOLD
2.5
2.0
1.5
1.0
0.5
0
–50
–25
0
25
50
75
TEMPERATURE (°C)
100
125
3707 G30
3707fb
7
LTC3707
PIN FUNCTIONS
RUN/SS1, RUN/SS2 (Pins 1, 15): Combination of soft-start,
run control inputs and short-circuit detection timers. A
capacitor to ground at each of these pins sets the ramp
time to full output current. Forcing either of these pins
back below 1.0V causes the IC to shut down the circuitry
required for that particular controller. Latchoff overcurrent
protection is also invoked via this pin as described in the
Applications Information section.
3.3VOUT (Pin 10): Output of a linear regulator capable
of supplying 10mA DC with peak currents as high as
50mA.
SENSE1+, SENSE2+ (Pins 2, 14): The (+) Input to the
Differential Current Comparators. The ITH pin voltage and
controlled offsets between the SENSE– and SENSE+ pins in
conjunction with RSENSE set the current trip threshold.
INTVCC (Pin 21): Output of the Internal 5V Linear Low
Dropout Regulator and the EXTVCC Switch. The driver and
control circuits are powered from this voltage source. Must
be decoupled to power ground with a minimum of 4.7μF
tantalum or other low ESR capacitor. The INTVCC regulator
standby function is determined by the STBYMD pin.
SENSE1–, SENSE2– (Pins 3, 13): The (–) Input to the
Differential Current Comparators.
VOSENSE1, VOSENSE2 (Pins 4, 12): Receives the remotelysensed feedback voltage for each controller from an external
resistive divider across the output.
FREQSET (Pin 5): Frequency Control Input to the Oscillator.
This pin can be left open, tied to ground, tied to INTVCC
or driven by an external voltage source. This pin can also
be used with an external phase detector to build a true
phase-locked loop.
STBYMD (Pin 6): Control pin that determines which circuitry remains active when the controllers are shut down
and/or provides a common control point to shut down both
controllers. See the Operation section for details.
FCB (Pin 7): Forced Continuous Control Input. This input
acts on both controllers and is normally used to regulate a
secondary winding. Pulling this pin below 0.8V will force
continuous synchronous operation on both controllers.
Do not leave this pin floating.
PGND (Pin 20): Driver Power Ground. Connects to the
sources of bottom (synchronous) N-channel MOSFETs,
anodes of the Schottky rectifiers and the (–) terminal(s)
of CIN.
EXTVCC (Pin 22): External Power Input to an Internal Switch
Connected to INTVCC. This switch closes and supplies VCC
power, bypassing the internal low dropout regulator, whenever EXTVCC is higher than 4.7V. See EXTVCC connection
in Applications section. Do not exceed 7V on this pin.
BG1, BG2 (Pins 23, 19): High Current Gate Drives for Bottom (Synchronous) N-Channel MOSFETs. Voltage swing
at these pins is from ground to INTVCC.
VIN (Pin 24): Main Supply Pin. A bypass capacitor should
be tied between this pin and the signal ground pin.
BOOST1, BOOST2 (Pins 25, 18): Bootstrapped Supplies
to the Top Side Floating Drivers. Capacitors are connected
between the boost and switch pins and Schottky diodes are
tied between the boost and INTVCC pins. Voltage swing at
the boost pins is from INTVCC to (VIN + INTVCC).
SW1, SW2 (Pins 26, 17): Switch Node Connections to
Inductors. Voltage swing at these pins is from a Schottky
diode (external) voltage drop below ground to VIN.
ITH1, ITH2 (Pins 8, 11): Error Amplifier Output and Switching
Regulator Compensation Point. Each associated channels’
current comparator trip point increases with this control
voltage.
TG1, TG2 (Pins 27, 16): High Current Gate Drives for
Top N-Channel MOSFETs. These are the outputs of floating drivers with a voltage swing equal to INTVCC – 0.5V
superimposed on the switch node voltage SW.
SGND (Pin 9): Small Signal Ground common to both
controllers, must be routed separately from high current
grounds to the common (–) terminals of the COUT
capacitors.
PGOOD (Pin 28): Open-Drain Logic Output. PGOOD is
pulled to ground when the voltage on either VOSENSE pin
is not within ±7.5% of its set point.
3707fb
8
LTC3707
FUNCTIONAL DIAGRAM
VIN
INTVCC
DUPLICATE FOR SECOND
CONTROLLER CHANNEL
1.19V
BOOST
1M
DROP
OUT
DET
CLK1
OSCILLATOR
CLK2
PGOOD
–
0.86V
S
Q
R
Q
TG
TOP
BOT
SWITCH
LOGIC
INTVCC
BOT
VFB
4.5V
–
SHDN
+
I1
+
–
FCB
–
++
+
VREF
SLOPE
COMP
–
–
INTVCC
I2
–
+
30k SENSE
+
45k
45k
2.4V
–
EA
+
EXTVCC
5V
+
–
OV
5V
LDO
REG
+
SGND
INTERNAL
SUPPLY
6V
VFB
VOSENSE
0.80V
R1
0.86V
ITH
SHDN
RST
4(VFB)
R2
+
–
1.2μA
INTVCC
CSEC
–
30k SENSE
VIN
4.8V
DSEC
3mV
0.86V
4(VFB)
0.8V
RSENSE
BINH
+
–
3.3VOUT
VOUT
–
FCB
R5
COUT
PGND
B
3V
0.18μA
R6
0.74V
BG
+
+
+
CIN
SW
TOP ON
0.55V
+
D1
FCB
+
–
CB
+
FREQSET
VSEC
DB
RUN
SOFTSTART
CC
CC2
RC
RUN/SS
STBYMD
CSS
3707 FD/F02
Figure 2
3707fb
9
LTC3707
OPERATION
(Refer to Functional Diagram)
Main Control Loop
The LTC3707 uses a constant frequency, current mode
step-down architecture with the two controller channels
operating 180 degrees out of phase. During normal
operation, each top MOSFET is turned on when the clock
for that channel sets the RS latch, and turned off when
the main current comparator, I1, resets the RS latch.
The peak inductor current at which I1 resets the RS
latch is controlled by the voltage on the ITH pin, which
is the output of each error amplifier EA. The VOSENSE pin
receives the voltage feedback signal, which is compared
to the internal reference voltage by the EA. When the load
current increases, it causes a slight decrease in VOSENSE
relative to the 0.8V reference, which in turn causes the
ITH voltage to increase until the average inductor current
matches the new load current. After the top MOSFET has
turned off, the bottom MOSFET is turned on until either the
inductor current starts to reverse, as indicated by current
comparator I2, or the beginning of the next cycle.
The top MOSFET drivers are biased from floating bootstrap
capacitor CB, which normally is recharged during each off
cycle through an external diode when the top MOSFET
turns off. As VIN decreases to a voltage close to VOUT,
the loop may enter dropout and attempt to turn on the
top MOSFET continuously. The dropout detector detects
this and forces the top MOSFET off for about 500ns every
tenth cycle to allow CB to recharge.
The main control loop is shut down by pulling the RUN/
SS pin low. Releasing RUN/SS allows an internal 1.2μA
current source to charge soft-start capacitor CSS. When
CSS reaches 1.5V, the main control loop is enabled with the
ITH voltage clamped at approximately 30% of its maximum
value. As CSS continues to charge, the ITH pin voltage is
gradually released allowing normal, full-current operation.
When both RUN/SS1 and RUN/SS2 are low, all LTC3707
controller functions are shut down, and the STBYMD pin
determines if the standby 5V and 3.3V regulators are
kept alive.
Low Current Operation
The FCB pin is a multifunction pin providing two
functions: 1) an analog input to provide regulation for a
secondary winding by temporarily forcing continuous
PWM operation on both controllers and 2) a logic input
to select between two modes of low current operation.
When the FCB pin voltage is below 0.800V, the controller
forces continuous PWM current mode operation. In
this mode, the top and bottom MOSFETs are alternately
turned on to maintain the output voltage independent
of direction of inductor current. When the FCB pin is
below VINTVCC – 2V but greater than 0.80V, the controller
enters Burst Mode operation. Burst Mode operation sets
a minimum output current level before inhibiting the top
switch and turns off the synchronous MOSFET(s) when
the inductor current goes negative. This combination of
requirements will, at low currents, force the ITH pin below
a voltage threshold that will temporarily inhibit turn-on of
both output MOSFETs until the output voltage drops. There
is 60mV of hysteresis in the burst comparator B tied to
the ITH pin. This hysteresis produces output signals to the
MOSFETs that turn them on for several cycles, followed
by a variable “sleep” interval depending upon the load
current. The resultant output voltage ripple is held to a
very small value by having the hysteretic comparator
after the error amplifier gain block.
Constant Frequency Operation
When the FCB pin is tied to INTVCC, Burst Mode operation
is disabled and the forced minimum output current
requirement is removed. This provides constant frequency,
discontinuous (preventing reverse inductor current)
current operation over the widest possible output current
range. This constant frequency operation is not as efficient
as Burst Mode operation, but does provide a lower noise,
constant frequency operating mode down to approximately
1% of designed maximum output current. Voltage should
not be applied to the FCB pin prior to the application of
voltage to the VIN pin.
Continuous Current (PWM) Operation
Tying the FCB pin to ground will force continuous current
operation. This is the least efficient operating mode,
but may be desirable in certain applications. The output
can source or sink current in this mode. When sinking
current while in forced continuous operation, current will
3707fb
10
LTC3707
OPERATION
(Refer to Functional Diagram)
be forced back into the main power supply potentially
boosting the input supply to dangerous voltage levels—
BEWARE!
Frequency Setting
The FREQSET pin provides frequency adjustment of the
internal oscillator from approximately 140kHz to 310kHz.
This input is nominally biased through an internal resistor
to the 1.19V reference, setting the oscillator frequency to
approximately 220kHz. This pin can be driven from an external AC or DC signal source to control the instantaneous
frequency of the oscillator. Voltage should not be applied
to the FREQSET pin prior to the application of voltage to
the VIN pin.
INTVCC/EXTVCC Power
Power for the top and bottom MOSFET drivers and most
other internal circuitry is derived from the INTVCC pin. When
the EXTVCC pin is left open, an internal 5V low dropout
linear regulator supplies INTVCC power. If EXTVCC is taken
above 4.7V, the 5V regulator is turned off and an internal
switch is turned on connecting EXTVCC to INTVCC. This
allows the INTVCC power to be derived from a high efficiency
external source such as the output of the regulator itself
or a secondary winding, as described in the Applications
Information.
Standby Mode Pin
The STBYMD pin is a three-state input that controls common circuitry within the IC as follows: When the STBYMD
pin is held at ground, both controller RUN/SS pins are pulled
to ground providing a single control pin to shut down both
controllers. When the pin is left open, the internal RUN/SS
currents are enabled to charge the RUN/SS capacitor(s),
allowing the turn-on of either controller and activating
necessary common internal biasing. When the STBYMD
pin is taken above 2V, both internal linear regulators are
turned on independent of the state on the RUN/SS pins
of the two switching regulator controllers, providing an
output power source for “wake-up” circuitry. Decouple the
pin with a small capacitor (0.01μF) to ground if the pin is
not connected to a DC potential.
Output Overvoltage Protection
An overvoltage comparator, 0V, guards against transient
overshoots (>7.5%) as well as other more serious conditions that may overvoltage the output. In this case, the top
MOSFET is turned off and the bottom MOSFET is turned
on until the overvoltage condition is cleared.
Power Good (PGOOD) Pin
The PGOOD pin is connected to an open drain of an internal MOSFET. The MOSFET turns on and pulls the pin
low when both the outputs are not within ± 7.5% of their
nominal output levels as determined by their resistive
feedback dividers. When both outputs meet the ± 7.5%
requirement, the MOSFET is turned off within 10μs and
the pin is allowed to be pulled up by an external resistor
to a source of up to 7V.
Foldback Current, Short-Circuit Detection
and Short-Circuit Latchoff
The RUN/SS capacitors are used initially to limit the inrush
current of each switching regulator. After the controller
has been started and been given adequate time to charge
up the output capacitors and provide full load current, the
RUN/SS capacitor is used in a short-circuit time-out circuit.
If the output voltage falls to less than 70% of its nominal
output voltage, the RUN/SS capacitor begins discharging
on the assumption that the output is in an overcurrent
and/or short-circuit condition. If the condition lasts for
a long enough period as determined by the size of the
RUN/SS capacitor, the controllers will be shut down until
the RUN/SS pin(s) voltage(s) are recycled. This built-in
latchoff can be overridden by providing a >5μA pull-up at
a compliance of 4.2V to the RUN/SS pin(s). This current
shortens the soft start period but also prevents net discharge of the RUN/SS capacitor(s) during an overcurrent
and/or short-circuit condition. Foldback current limiting
is also activated when the output voltage falls below
70% of its nominal level whether or not the short-circuit
latchoff circuit is enabled. Even if a short is present and
the short-circuit latchoff is not enabled, a safe, low output
current is provided due to internal current foldback and
actual power dissipated is low due to the efficient nature
of the current mode switching regulator.
3707fb
11
LTC3707
OPERATION
(Refer to Functional Diagram)
THEORY AND BENEFITS OF 2-PHASE OPERATION
The LTC1628 and the LTC3707 are the first dual high
efficiency DC/DC controllers to bring the considerable
benefits of 2-phase operation to portable applications.
Notebook computers, PDAs, handheld terminals and automotive electronics will all benefit from the lower input
filtering requirement, reduced electromagnetic interference
(EMI) and increased efficiency associated with 2-phase
operation.
Why the need for 2-phase operation? Up until the LTC1628
was introduced, constant-frequency dual switching regulators operated both channels in phase (i.e., single-phase
operation). This means that both switches turned on at
the same time, causing current pulses of up to twice the
amplitude of those for one regulator to be drawn from the
input capacitor and battery. These large amplitude current
pulses increased the total RMS current flowing from the
input capacitor, requiring the use of more expensive input
capacitors and increasing both EMI and losses in the input
capacitor and battery.
With 2-phase operation, the two channels of the dualswitching regulator are operated 180 degrees out of phase.
This effectively interleaves the current pulses drawn by the
switches, greatly reducing the overlap time where they add
together. The result is a significant reduction in total RMS
input current, which in turn allows less expensive input
capacitors to be used, reduces shielding requirements for
EMI and improves real world operating efficiency.
Figure 3 compares the input waveforms for a representative single-phase dual switching regulator to the LTC3707
2-phase dual switching regulator. An actual measurement of
the RMS input current under these conditions shows that 2phase operation dropped the input current from 2.53ARMS
to 1.55ARMS. While this is an impressive reduction in itself,
remember that the power losses are proportional to IRMS2,
meaning that the actual power wasted is reduced by a factor of 2.66. The reduced input ripple voltage also means
less power is lost in the input power path, which could
include batteries, switches, trace/connector resistances
and protection circuitry. Improvements in both conducted
and radiated EMI also directly accrue as a result of the
reduced RMS input current and voltage.
Of course, the improvement afforded by 2-phase operation is a function of the dual switching regulator’s relative
duty cycles which, in turn, are dependent upon the input
voltage VIN (Duty Cycle = VOUT/VIN). Figure 4 shows how
the RMS input current varies for single-phase and 2-phase
operation for 3.3V and 5V regulators over a wide input
voltage range.
5V SWITCH
20V/DIV
3.3V SWITCH
20V/DIV
INPUT CURRENT
5A/DIV
INPUT VOLTAGE
500mV/DIV
IIN(MEAS) = 2.53ARMS
(a)
3707 F03a
IIN(MEAS) = 1.55ARMS
3707 F03b
(b)
Figure 3. Input Waveforms Comparing Single-Phase (a) and 2-Phase (b) Operation for Dual
Switching Regulators Converting 12V to 5V and 3.3V at 3A Each. The Reduced Input Ripple
with the LTC1628 2-Phase Regulator Allows Less Expensive Input Capacitors, Reduces
Shielding Requirements for EMI and Improves Efficiency
3707fb
12
LTC3707
(Refer to Functional Diagram)
It can readily be seen that the advantages of 2-phase operation are not just limited to a narrow operating range, but
in fact extend over a wide region. A good rule of thumb
for most applications is that 2-phase operation will reduce
the input capacitor requirement to that for just one channel
operating at maximum current and 50% duty cycle.
A final question: If 2-phase operation offers such an
advantage over single-phase operation for dual switching
regulators, why hasn’t it been done before? The answer
is that, while simple in concept, it is hard to implement.
Constant-frequency current mode switching regulators
require an oscillator derived “slope compensation”
signal to allow stable operation of each regulator at over
50% duty cycle. This signal is relatively easy to derive in
single-phase dual switching regulators, but required the
development of a new and proprietary technique to allow
2-phase operation. In addition, isolation between the two
channels becomes more critical with 2-phase operation
because switch transitions in one channel could potentially
disrupt the operation of the other channel.
The LTC1628 and the LTC3707 are proof that these hurdles
have been surmounted. The new device offers unique advantages for the ever-expanding number of high efficiency
power supplies required in portable electronics.
3.0
SINGLE PHASE
DUAL CONTROLLER
2.5
INPUT RMS CURRENT (A)
OPERATION
2.0
1.5
2-PHASE
DUAL CONTROLLER
1.0
0.5
0
VO1 = 5V/3A
VO2 = 3.3V/3A
0
10
20
30
INPUT VOLTAGE (V)
40
3707 F04
Figure 4. RMS Input Current Comparison
APPLICATIONS INFORMATION
Figure 1 on the first page is a basic LTC3707 application
circuit. External component selection is driven by the
load requirement, and begins with the selection of RSENSE
and the inductor value. Next, the power MOSFETs and
D1 are selected. Finally, CIN and COUT are selected. The
circuit shown in Figure 1 can be configured for operation
up to an input voltage of 28V (limited by the external
MOSFETs).
RSENSE =
50mV
IMAX
RSENSE Selection For Output Current
Because of possible PCB noise in the current sensing loop,
the AC current sensing ripple of ΔVSENSE = ΔI • RSENSE
also needs to be checked in the design to get good
signal-to-noise ratio. In general, for a reasonable good
PCB layout, a 15mV ΔVSENSE voltage is recommended
as a conservative number to start with.
RSENSE is chosen based on the required output current. The
LTC3707 current comparator has a maximum threshold
of 75mV/RSENSE and an input common mode range of
SGND to 1.1(INTVCC). The current comparator threshold
sets the peak of the inductor current, yielding a maximum
average output current IMAX equal to the peak value less
half the peak-to-peak ripple current, ΔIL.
When using the controller in very low dropout conditions,
the maximum output current level will be reduced due to
the internal compensation required to meet stability criterion for buck regulators operating at greater than 50%
duty factor. A curve is provided to estimate this reducton
in peak output current level depending upon the operating
duty factor.
Allowing a margin for variations in the LTC3707 and external
component values yields:
3707fb
13
LTC3707
APPLICATIONS INFORMATION
Selection of Operating Frequency
The LTC3707 uses a constant frequency architecture with
the frequency determined by an internal oscillator capacitor.
This internal capacitor is charged by a fixed current plus
an additional current that is proportional to the voltage
applied to the FREQSET pin.
A graph for the voltage applied to the FREQSET pin vs
frequency is given in Figure 5. As the operating frequency
is increased the gate charge losses will be higher, reducing
efficiency (see Efficiency Considerations). The maximum
switching frequency is approximately 310kHz.
FREQSET PIN VOLTAGE (V)
2.5
2.0
The inductor value also has secondary effects. The transition to Burst Mode operation begins when the average
inductor current required results in a peak current below
25% of the current limit determined by RSENSE. Lower
inductor values (higher ΔIL) will cause this to occur at
lower load currents, which can cause a dip in efficiency in
the upper range of low current operation. In Burst Mode
operation, lower inductance values will cause the burst
frequency to decrease.
1.5
Inductor Core Selection
1.0
0.5
0
120
170
220
270
OPERATING FREQUENCY (kHz)
320
3707 F05
Figure 5. FREQSET Pin Voltage vs Frequency
Inductor Value Calculation
The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the use
of smaller inductor and capacitor values. So why would
anyone ever choose to operate at lower frequencies with
larger components? The answer is efficiency. A higher
frequency generally results in lower efficiency because
of MOSFET gate charge losses. In addition to this basic
trade-off, the effect of inductor value on ripple current and
low current operation must also be considered.
The inductor value has a direct effect on ripple current.
The inductor ripple current ΔIL decreases with higher
inductance or frequency and increases with higher VIN:
ΔIL =
Accepting larger values of ΔIL allows the use of low
inductances, but results in higher output voltage ripple
and greater core losses. A reasonable starting point for
setting ripple current is ΔI = 30% • IOUT(MAX) or higher for
good load transient response and sufficient ripple current
signal in the current loop. Remember, the maximum ΔIL
occurs at the maximum input voltage.
⎛ V ⎞
1
VOUT ⎜ 1– OUT ⎟
(f)(L)
VIN ⎠
⎝
Once the value for L is known, the type of inductor must
be selected. High efficiency converters generally cannot
afford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite, molypermalloy,
or Kool Mμ® cores. Actual core loss is independent of core
size for a fixed inductor value, but it is very dependent
on inductance selected. As inductance increases, core
losses go down. Unfortunately, increased inductance
requires more turns of wire and therefore copper losses
will increase.
Ferrite designs have very low core loss and are preferred
at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite
core material saturates “hard,” which means that inductance collapses abruptly when the peak design current is
exceeded. This results in an abrupt increase in inductor
ripple current and consequent output voltage ripple. Do
not allow the core to saturate!
Molypermalloy (from Magnetics, Inc.) is a very good, low
loss core material for toroids, but it is more expensive
than ferrite. A reasonable compromise from the same
manufacturer is Kool Mμ. Toroids are very space efficient,
especially when you can use several layers of wire. Because
they generally lack a bobbin, mounting is more difficult.
3707fb
14
LTC3707
APPLICATIONS INFORMATION
However, designs for surface mount are available that do
not increase the height significantly.
Power MOSFET and D1 Selection
Two external power MOSFETs must be selected for each
controller with the LTC3707: One N-channel MOSFET for
the top (main) switch, and one N-channel MOSFET for the
bottom (synchronous) switch.
The peak-to-peak drive levels are set by the INTVCC
voltage. This voltage is typically 5V during start-up
(see EXTVCC Pin Connection). Consequently, logic-level
threshold MOSFETs must be used in most applications.
The only exception is if low input voltage is expected
(VIN < 5V); then, sub-logic level threshold MOSFETs
(VGS(TH) < 3V) should be used. Pay close attention to the
BVDSS specification for the MOSFETs as well; most of the
logic level MOSFETs are limited to 30V or less.
Selection criteria for the power MOSFETs include the “ON”
resistance RDS(ON), reverse transfer capacitance CRSS, input
voltage and maximum output current. When the LTC3707
is operating in continuous mode the duty cycles for the
top and bottom MOSFETs are given by:
Main Switch Duty Cycle =
VOUT
VIN
Synchronous Switch Duty Cycle =
VIN – VOUT
VIN
The MOSFET power dissipations at maximum output
current are given by:
PMAIN =
VOUT
2
IMAX ) (1+ δ )RDS(ON) +
(
VIN
k ( VIN ) (IMAX ) (CRSS ) ( f )
2
PSYNC =
VIN – VOUT
2
IMAX ) (1+ δ )RDS(ON)
(
VIN
where δ is the temperature dependency of RDS(ON) and k
is a constant inversely related to the gate drive current.
Both MOSFETs have I2R losses while the topside N-channel
equation includes an additional term for transition losses,
which are highest at high input voltages. For VIN < 20V
the high current efficiency generally improves with larger
MOSFETs, while for VIN > 20V the transition losses rapidly
increase to the point that the use of a higher RDS(ON) device
with lower CRSS actually provides higher efficiency. The
synchronous MOSFET losses are greatest at high input
voltage when the top switch duty factor is low or during
a short-circuit when the synchronous switch is on close
to 100% of the period.
The term (1+δ) is generally given for a MOSFET in the
form of a normalized RDS(ON) vs Temperature curve, but
δ = 0.005/°C can be used as an approximation for low
voltage MOSFETs. CRSS is usually specified in the MOSFET
characteristics. The constant k = 1.7 can be used to estimate the contributions of the two terms in the main switch
dissipation equation.
The Schottky diode D1 shown in Figure 1 conducts during the dead-time between the conduction of the two
power MOSFETs. This prevents the body diode of the
bottom MOSFET from turning on, storing charge during
the dead-time and requiring a reverse recovery period
that could cost as much as 3% in efficiency at high VIN.
A 1A to 3A Schottky is generally a good compromise for
both regions of operation due to the relatively small average current. Larger diodes result in additional transition
losses due to their larger junction capacitance. Schottky
diodes should be placed in parallel with the synchronous
MOSFETs when operating in pulse-skip mode or in Burst
Mode operation.
CIN and COUT Selection
The selection of CIN is simplified by the multiphase architecture and its impact on the worst-case RMS current
drawn through the input network (battery/fuse/capacitor).
It can be shown that the worst case RMS current occurs
when only one controller is operating. The controller with
the highest (VOUT)(IOUT) product needs to be used in the
formula below to determine the maximum RMS current
requirement. Increasing the output current, drawn from
the other out-of-phase controller, will actually decrease the
input RMS ripple current from this maximum value (see
Figure 4). The out-of-phase technique typically reduces
the input capacitor’s RMS ripple current by a factor of
3707fb
15
LTC3707
APPLICATIONS INFORMATION
30% to 70% when compared to a single phase power
supply solution.
The type of input capacitor, value and ESR rating have
efficiency effects that need to be considered in the selection process. The capacitance value chosen should be
sufficient to store adequate charge to keep high peak
battery currents down. 20μF to 40μF is usually sufficient
for a 25W output supply operating at 200kHz. The ESR of
the capacitor is important for capacitor power dissipation
as well as overall battery efficiency. All of the power (RMS
ripple current • ESR) not only heats up the capacitor but
wastes power from the battery.
Medium voltage (20V to 35V) ceramic, tantalum, OS-CON
and switcher-rated electrolytic capacitors can be used
as input capacitors, but each has drawbacks: ceramic
voltage coefficients are very high and may have audible
piezoelectric effects; tantalums need to be surge-rated;
OS-CONs suffer from higher inductance, larger case size
and limited surface-mount applicability; electrolytics’
higher ESR and dryout possibility require several to be
used. Multiphase systems allow the lowest amount of
capacitance overall. As little as one 22μF or two to three
10μF ceramic capacitors are an ideal choice in a 20W to
50W power supply due to their extremely low ESR. Even
though the capacitance at 20V is substantially below their
rating at zero-bias, very low ESR loss makes ceramics
an ideal candidate for highest efficiency battery operated
systems. Also consider parallel ceramic and high quality
electrolytic capacitors as an effective means of achieving
ESR and bulk capacitance goals.
In continuous mode, the source current of the top N-channel
MOSFET is a square wave of duty cycle VOUT/VIN. To prevent
large voltage transients, a low ESR input capacitor sized for
the maximum RMS current of one channel must be used.
The maximum RMS capacitor current is given by:
CIN RequiredIRMS ≈IMAX
⎡⎣ VOUT ( VIN − VOUT ) ⎤⎦
VIN
1/2
This formula has a maximum at V IN = 2V OUT, where
IRMS = IOUT/2. This simple worst case condition is commonly used for design because even significant deviations
do not offer much relief. Note that capacitor manufacturer’s
ripple current ratings are often based on only 2000 hours
of life. This makes it advisable to further derate the capacitor, or to choose a capacitor rated at a higher temperature
than required. Several capacitors may also be paralleled
to meet size or height requirements in the design. Always
consult the manufacturer if there is any question.
The benefit of the LTC3707 multiphase can be calculated by
using the equation above for the higher power controller
and then calculating the loss that would have resulted if
both controller channels switch on at the same time. The
total RMS power lost is lower when both controllers are
operating due to the interleaving of current pulses through
the input capacitor’s ESR. This is why the input capacitor’s
requirement calculated above for the worst-case controller
is adequate for the dual controller design. Remember that
input protection fuse resistance, battery resistance and PC
board trace resistance losses are also reduced due to the
reduced peak currents in a multiphase system. The overall
benefit of a multiphase design will only be fully realized
when the source impedance of the power supply/battery
is included in the efficiency testing. The drains of the
two top MOSFETS should be placed within 1cm of each
other and share a common CIN(s). Separating the drains
and CIN may produce undesirable voltage and current
resonances at VIN.
The selection of COUT is driven by the required effective
series resistance (ESR). Typically once the ESR requirement is satisfied the capacitance is adequate for filtering.
The output ripple (ΔVOUT) is determined by:
⎛
1 ⎞
ΔVOUT ≈ ΔIL ⎜ ESR +
8fCOUT ⎟⎠
⎝
Where f = operating frequency, COUT = output capacitance,
and ΔIL= ripple current in the inductor. The output ripple is
highest at maximum input voltage since ΔIL increases with
input voltage. With ΔIL = 0.3IOUT(MAX) the output ripple will
typically be less than 50mV at max VIN assuming:
COUT Recommended ESR < 2 RSENSE
and COUT > 1/(8fRSENSE)
The first condition relates to the ripple current into the ESR
of the output capacitance while the second term guarantees
3707fb
16
LTC3707
APPLICATIONS INFORMATION
that the output capacitance does not significantly discharge
during the operating frequency period due to ripple current.
The choice of using smaller output capacitance increases
the ripple voltage due to the discharging term but can be
compensated for by using capacitors of very low ESR to
maintain the ripple voltage at or below 50mV. The ITH pin
OPTI-LOOP compensation components can be optimized
to provide stable, high performance transient response
regardless of the output capacitors selected.
Manufacturers such as Nichicon, United Chemicon and
Sanyo can be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest (ESR)(size)
product of any aluminum electrolytic at a somewhat
higher price. An additional ceramic capacitor in parallel
with OS-CON capacitors is recommended to reduce the
inductance effects.
In surface mount applications multiple capacitors may
need to be used in parallel to meet the ESR, RMS current
handling and load step requirements of the application.
Aluminum electrolytic, dry tantalum and special polymer
capacitors are available in surface mount packages. Special
polymer surface mount capacitors offer very low ESR but
have lower storage capacity per unit volume than other
capacitor types. These capacitors offer a very cost-effective
output capacitor solution and are an ideal choice when
combined with a controller having high loop bandwidth.
Tantalum capacitors offer the highest capacitance density
and are often used as output capacitors for switching
regulators having controlled soft-start. Several excellent
surge-tested choices are the AVX TPS, AVX TPSV or
the KEMET T510 series of surface mount tantalums,
available in case heights ranging from 2mm to 4mm.
Aluminum electrolytic capacitors can be used in cost-driven
applications providing that consideration is given to ripple
current ratings, temperature and long term reliability. A
typical application will require several to many aluminum
electrolytic capacitors in parallel. A combination of the
above mentioned capacitors will often result in maximizing
performance and minimizing overall cost. Other capacitor
types include Nichicon PL series, NEC Neocap, Pansonic
SP and Sprague 595D series. Consult manufacturers for
other specific recommendations.
INTVCC Regulator
An internal P-channel low dropout regulator produces 5V
at the INTVCC pin from the VIN supply pin. INTVCC powers the drivers and internal circuitry within the LTC3707.
The INTVCC pin regulator can supply a peak current of
40mA and must be bypassed to ground with a minimum
of 4.7μF tantalum, 10μF special polymer, or low ESR type
electrolytic capacitor. A 1μF ceramic capacitor placed directly adjacent to the INTVCC and PGND IC pins is highly
recommended. Good bypassing is necessary to supply
the high transient currents required by the MOSFET gate
drivers and to prevent interaction between channels.
Higher input voltage applications in which large MOSFETs
are being driven at high frequencies may cause the maximum junction temperature rating for the LTC3707 to be
exceeded. The system supply current is normally dominated
by the gate charge current. Additional external loading of
the INTVCC and 3.3V linear regulators also needs to be
taken into account for the power dissipation calculations.
The total INTVCC current can be supplied by either the 5V
internal linear regulator or by the EXTVCC input pin. When
the voltage applied to the EXTVCC pin is less than 4.7V, all
of the INTVCC current is supplied by the internal 5V linear
regulator. Power dissipation for the IC in this case is highest: (VIN)(IINTVCC), and overall efficiency is lowered. The
gate charge current is dependent on operating frequency
as discussed in the Efficiency Considerations section.
The junction temperature can be estimated by using the
equations given in Note 2 of the Electrical Characteristics.
For example, the LTC3707 VIN current is limited to less
than 24mA from a 24V supply when not using the EXTVCC
pin as follows:
TJ = 70°C + (24mA)(24V)(95°C/W) = 125°C
Use of the EXTVCC input pin reduces the junction temperature to:
TJ = 70°C + (24mA)(5V)(95°C/W) = 81°C
Dissipation should be calculated to also include any added
current drawn from the internal 3.3V linear regulator.
To prevent maximum junction temperature from being
exceeded, the input supply current must be checked
operating in continuous mode at maximum VIN.
3707fb
17
LTC3707
APPLICATIONS INFORMATION
EXTVCC Connection
The LTC3707 contains an internal P-channel MOSFET
switch connected between the EXTVCC and INTVCC pins.
When the voltage applied to EXTVCC rises above 4.7V,
the internal regulator is turned off and the switch closes,
connecting the EXTVCC pin to the INTVCC pin thereby supplying internal power. The switch remains closed as long
as the voltage applied to EXTVCC remains above 4.5V. This
allows the MOSFET driver and control power to be derived
from the output during normal operation (4.7V < VOUT <
7V) and from the internal regulator when the output is
out of regulation (start-up, short-circuit). If more current
is required through the EXTVCC switch than is specified,
an external Schottky diode can be added between the
EXTVCC and INTVCC pins. Do not apply greater than 7V to
the EXTVCC pin and ensure that EXTVCC < VIN.
Significant efficiency gains can be realized by powering
INTVCC from the output, since the VIN current resulting
from the driver and control currents will be scaled by a
factor of (Duty Cycle)/(Efficiency). For 5V regulators this
supply means connecting the EXTVCC pin directly to VOUT.
However, for 3.3V and other lower voltage regulators,
additional circuitry is required to derive INTVCC power
from the output.
The following list summarizes the four possible connections for EXTVCC:
1. EXTVCC Left Open (or Grounded). This will cause INTVCC
to be powered from the internal 5V regulator resulting in an
efficiency penalty of up to 10% at high input voltages.
2. EXTVCC Connected directly to VOUT. This is the normal
connection for a 5V regulator and provides the highest
efficiency.
3. EXTVCC Connected to an External supply. If an external
supply is available in the 5V to 7V range, it may be used to
power EXTVCC providing it is compatible with the MOSFET
gate drive requirements.
4. EXTVCC Connected to an Output-Derived Boost Network.
For 3.3V and other low voltage regulators, efficiency gains
can still be realized by connecting EXTVCC to an outputderived voltage that has been boosted to greater than 4.7V.
This can be done with either the inductive boost winding
as shown in Figure 6a or the capacitive charge pump
shown in Figure 6b. The charge pump has the advantage
of simple magnetics.
VIN
OPTIONAL EXTVCC
CONNECTION
5V < VSEC < 7V
+
CIN
CIN
VSEC
VIN
1μF
BAT85
FCB
BG1
T1
1:N
R6
VOUT
EXTVCC
L1
SW
+
R5
COUT
+
COUT
BG1
N-CH
SGND
BAT85
RSENSE
VOUT
SW
BAT85
VN2222LL
TG1
RSENSE
EXTVCC
0.22μF
N-CH
LTC3707
TG1
1μF
+
VIN
+
N-CH
LTC3707
+
VIN
N-CH
PGND
PGND
3707 F06a
Figure 6a. Secondary Output Loop & EXTVCC Connection
3707 F06b
Figure 6b. Capacitive Charge Pump for EXTVCC
3707fb
18
LTC3707
APPLICATIONS INFORMATION
Topside MOSFET Driver Supply (CB, DB)
External bootstrap capacitors CB connected to the BOOST
pins supply the gate drive voltages for the topside MOSFETs.
Capacitor CB in the functional diagram is charged though
external diode DB from INTVCC when the SW pin is low.
When one of the topside MOSFETs is to be turned on,
the driver places the CB voltage across the gate-source
of the desired MOSFET. This enhances the MOSFET and
turns on the topside switch. The switch node voltage, SW,
rises to VIN and the BOOST pin follows. With the topside
MOSFET on, the boost voltage is above the input supply:
VBOOST = VIN + VINTVCC. The value of the boost capacitor
CB needs to be 100 times that of the total input capacitance
of the topside MOSFET(s). The reverse breakdown of the
external Schottky diode must be greater than VIN(MAX).
When adjusting the gate drive level, the final arbiter is the
total input current for the regulator. If a change is made
and the input current decreases, then the efficiency has
improved. If there is no change in input current, then there
is no change in efficiency.
Output Voltage
The LTC3707 output voltages are each set by an external
feedback resistive divider carefully placed across the output
capacitor. The resultant feedback signal is compared with
the internal precision 0.800V voltage reference by the error
amplifier. The output voltage is given by the equation:
⎛ R2 ⎞
VOUT = 0.8V ⎜ 1+ ⎟
⎝ R1⎠
SENSE+/SENSE– Pins
The common mode input range of the current comparator
sense pins is from 0V to (1.1)INTVCC. Continuous linear
operation is guaranteed throughout this range allowing
output voltage setting from 0.8V to 7.7V, depending upon
the voltage applied to EXTVCC. A differential NPN input
stage is biased with internal resistors from an internal 2.4V
source as shown in the Functional Diagram. This requires
that current either be sourced or sunk from the SENSE
pins depending on the output voltage. If the output voltage
is below 2.4V current will flow out of both SENSE pins to
the main output. The output can be easily preloaded by
the VOUT resistive divider to compensate for the current
comparator’s negative input bias current. The maximum
current flowing out of each pair of SENSE pins is:
ISENSE+ + ISENSE– = (2.4V – VOUT)/24k
Since VOSENSE is servoed to the 0.8V reference voltage,
we can choose R1 in Figure 2 to have a maximum value
to absorb this current.
⎛
⎞
0.8V
R1(MAX) = 24k ⎜
⎝ 2.4V – VOUT ⎟⎠
for VOUT < 2.4V
Regulating an output voltage of 1.8V, the maximum value
of R1 should be 32K. Note that for an output voltage above
2.4V, R1 has no maximum value necessary to absorb the
sense currents; however, R1 is still bounded by the VOSENSE
feedback current.
Soft-Start/Run Function
The RUN/SS1 and RUN/SS2 pins are multipurpose pins
that provide a soft-start function and a means to shut down
the LTC3707. Soft-start reduces the input power source’s
surge currents by gradually increasing the controller’s
current limit (proportional to VITH). This pin can also be
used for power supply sequencing.
An internal 1.2μA current source charges up the CSS capacitor. When the voltage on RUN/SS1 (RUN/SS2) reaches
1.5V, the particular controller is permitted to start operating.
As the voltage on RUN/SS increases from 1.5V to 3.0V,
the internal current limit is increased from 25mV/RSENSE
to 75mV/RSENSE. The output current limit ramps up slowly,
taking an additional 1.25s/μF to reach full current. The
output current thus ramps up slowly, reducing the starting surge current required from the input power supply.
If RUN/SS has been pulled all the way to ground there is
a delay before starting of approximately:
tDELAY =
1.5V
C = (1.25s / µF ) CSS
1.2µA SS
t IRAMP =
3V − 1.5V
C = (1.25s / µF ) CSS
1.2µA SS
3707fb
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LTC3707
APPLICATIONS INFORMATION
By pulling both RUN/SS pins below 1V and/or pulling
the STBYMD pin below 0.2V, the LTC3707 is put into
low current shutdown (IQ = 20μA). The RUN/SS pins
can be driven directly from logic as shown in Figure 7.
Diode D1 in Figure 7 reduces the start delay but allows
CSS to ramp up slowly providing the soft-start function.
Each RUN/SS pin has an internal 6V zener clamp (See
Functional Diagram).
VIN
3.3V OR 5V
INTVCC
RUN/SS
RSS*
D1
RSS*
RUN/SS
CSS
CSS
*OPTIONAL TO DEFEAT OVERCURRENT LATCHOFF
(a)
3707 F07
(b)
Figure 7. RUN/SS Pin Interfacing
Fault Conditions: Overcurrent Latchoff
The RUN/SS pins also provide the ability to latch off the
controller(s) when an overcurrent condition is detected.
The RUN/SS capacitor, CSS, is used initially to turn on
and limit the inrush current. After the controller has been
started and been given adequate time to charge up the
output capacitor and provide full load current, the RUN/SS
capacitor is used for a short-circuit timer. If the regulator’s
output voltage falls to less than 70% of its nominal value
after CSS reaches 4.2V, CSS begins discharging on the assumption that the output is in an overcurrent condition. If
the condition lasts for a long enough period as determined
by the size of the CSS and the specified discharge current,
the controller will be shut down until the RUN/SS pin voltage is recycled. If the overload occurs during start-up, the
time can be approximated by:
tLO1 ≈ [CSS (4.1 – 1.5 + 4.1 – 3.5)]/(1.2μA)
= 2.7 • 106 (CSS)
If the overload occurs after start-up the voltage on CSS will
begin discharging from the zener clamp voltage:
tLO2 ≈ [CSS (6 – 3.5)]/(1.2μA) = 2.1 • 106 (CSS)
This built-in overcurrent latchoff can be overridden by
providing a pull-up resistor to the RUN/SS pin as shown
in Figure 7. This resistance shortens the soft-start period
and prevents the discharge of the RUN/SS capacitor during
an over current condition. Tying this pull-up resistor to
VIN as in Figure 7a, defeats overcurrent latchoff. Diodeconnecting this pull-up resistor to INTVCC , as in Figure
7b, eliminates any extra supply current during controller
shutdown while eliminating the INTVCC loading from
preventing controller start-up.
Why should you defeat overcurrent latchoff? During the
prototyping stage of a design, there may be a problem
with noise pickup or poor layout causing the protection
circuit to latch off. Defeating this feature will easily allow
troubleshooting of the circuit and PC layout. The internal
short-circuit and foldback current limiting still remains
active, thereby protecting the power supply system from
failure. After the design is complete, a decision can be
made whether to enable the latchoff feature.
The value of the soft-start capacitor CSS may need to be
scaled with output voltage, output capacitance and load
current characteristics. The minimum soft-start capacitance is given by:
CSS > (COUT )(VOUT) (10 –4) (RSENSE)
The minimum recommended soft-start capacitor of
CSS = 0.1μF will be sufficient for most applications.
Fault Conditions: Current Limit and Current Foldback
The LTC3707 current comparator has a maximum sense
voltage of 75mV resulting in a maximum MOSFET current of 75mV/RSENSE. The maximum value of current
limit generally occurs with the largest VIN at the highest
ambient temperature, conditions that cause the highest
power dissipation in the top MOSFET.
The LTC3707 includes current foldback to help further
limit load current when the output is shorted to ground.
The foldback circuit is active even when the overload
shutdown latch described above is overridden. If the
output falls below 70% of its nominal output level, then
the maximum sense voltage is progressively lowered from
75mV to 25mV. Under short-circuit conditions with very
low duty cycles, the LTC3707 will begin cycle skipping in
order to limit the short-circuit current. In this situation
the bottom MOSFET will be dissipating most of the power
3707fb
20
LTC3707
APPLICATIONS INFORMATION
but less than in normal operation. The short-circuit ripple
current is determined by the minimum on-time tON(MIN)
of the LTC3707 (less than 200ns), the input voltage and
inductor value:
ΔIL(SC) = tON(MIN) (VIN/L)
The resulting short-circuit current is:
ISC =
25mV 1
+ ΔI
RSENSE 2 L(SC)
Fault Conditions: Overvoltage Protection (Crowbar)
The overvoltage crowbar is designed to blow a system
input fuse when the output voltage of the regulator rises
much higher than nominal levels. The crowbar causes huge
currents to flow, that blow the fuse to protect against a
shorted top MOSFET if the short occurs while the controller is operating.
A comparator monitors the output for overvoltage conditions. The comparator (0V) detects overvoltage faults
greater than 7.5% above the nominal output voltage. When
this condition is sensed, the top MOSFET is turned off and
the bottom MOSFET is turned on until the overvoltage
condition is cleared. The output of this comparator is
only latched by the overvoltage condition itself and will
therefore allow a switching regulator system having a poor
PC layout to function while the design is being debugged.
The bottom MOSFET remains on continuously for as long
as the 0V condition persists; if VOUT returns to a safe level,
normal operation automatically resumes. A shorted top
MOSFET will result in a high current condition which will
open the system fuse. The switching regulator will regulate
properly with a leaky top MOSFET by altering the duty
cycle to accommodate the leakage.
The Standby Mode (STBYMD) Pin Function
The Standby Mode (STBYMD) pin provides several choices
for start-up and standby operational modes. If the pin is
pulled to ground, the RUN/SS pins for both controllers
are internally pulled to ground, preventing start-up and
thereby providing a single control pin for turning off both
controllers at once. If the pin is left open or decoupled with
a capacitor to ground, the RUN/SS pins are each internally
provided with a starting current enabling external control
for turning on each controller independently. If the pin is
provided with a current of >3μA at a voltage greater than
2V, both internal linear regulators (INTVCC and 3.3V) will
be on even when both controllers are shut down. In this
mode, the onboard 3.3V and 5V linear regulators can
provide power to keep-alive functions such as a keyboard
controller. This pin can also be used as a latching “on”
and/or latching “off” power switch if so designed.
Frequency of Operation
The LTC3707 has an internal voltage controlled oscillator.
The frequency of this oscillator can be varied over a 2 to 1
range. The pin is internally self-biased at 1.19V, resulting
in a free-running frequency of approximately 220kHz. The
FREQSET pin can be grounded to lower this frequency to
approximately 140kHz or tied to the INTVCC pin to yield
approximately 310kHz. The FREQSET pin may be driven
with a voltage from 0 to INTVCC to fix or modulate the
oscillator frequency as shown in Figure 5.
Minimum On-Time Considerations
Minimum on-time tON(MIN) is the smallest time duration
that the LTC3707 is capable of turning on the top MOSFET.
It is determined by internal timing delays and the gate
charge required to turn on the top MOSFET. Low duty
cycle applications may approach this minimum on-time
limit and care should be taken to ensure that
t ON(MIN) <
VOUT
VIN (f)
If the duty cycle falls below what can be accommodated
by the minimum on-time, the LTC3707 will begin to skip
cycles. The output voltage will continue to be regulated,
but the ripple voltage and current will increase.
The typical tested minimum on-time of the LTC3707 is
180ns under an ideal condition without switching noise.
However, the minimum on-time can be affected by PCB
switching noise in the voltage and current loops. With
reasonably good PCB layout, minimum 30% inductor
current ripple and about 15mV sensing ripple voltage,
300ns minimum on-time is a conservative number to
start with.
3707fb
21
LTC3707
APPLICATIONS INFORMATION
FCB Pin Operation
The FCB pin can be used to regulate a secondary winding
or as a logic level input. Continuous operation is forced
when the FCB pin drops below 0.8V. During continuous
mode, current flows continuously in the transformer primary. The secondary winding(s) draw current only when
the bottom, synchronous switch is on. When primary
load currents are low and/or the VIN/VOUT ratio is low,
the synchronous switch may not be on for a sufficient
amount of time to transfer power from the output capacitor
to the secondary load. Forced continuous operation will
support secondary windings providing there is sufficient
synchronous switch duty factor. Thus, the FCB input pin
removes the requirement that power must be drawn from
the inductor primary in order to extract power from the
auxiliary windings. With the loop in continuous mode, the
auxiliary outputs may nominally be loaded without regard
to the primary output load.
The secondary output voltage VSEC is normally set as shown
in Figure 6a by the turns ratio N of the transformer:
VSEC ≅ (N + 1) VOUT
However, if the controller goes into Burst Mode operation
and halts switching due to a light primary load current, then
VSEC will droop. An external resistive divider from VSEC to
the FCB pin sets a minimum voltage VSEC(MIN):
⎛ R6 ⎞
VSEC(MIN) ≈ 0.8V ⎜ 1+ ⎟
⎝ R5 ⎠
If VSEC drops below this level, the FCB voltage forces
temporary continuous switching operation until VSEC is
again above its minimum.
In order to prevent erratic operation if no external connections are made to the FCB pin, the FCB pin has a 0.18μA
internal current source pulling the pin high. Include this
current when choosing resistor values R5 and R6.
The following table summarizes the possible states available on the FCB pin:
Table 1
FCB Pin
Condition
0V to 0.75V
Forced Continuous (Current Reversal
Allowed—Burst Inhibited)
0.85V < VFCB < VINTVCC – 2V
Minimum Peak Current Induces
Burst Mode Operation
No Current Reversal Allowed
Feedback Resistors
Regulating a Secondary Winding
= VINTVCC
Burst Mode Operation Disabled
Constant Frequency Mode Enabled
No Current Reversal Allowed
No Minimum Peak Current
Voltage Positioning
Voltage positioning can be used to minimize peak-to-peak
output voltage excursions under worst-case transient
loading conditions. The open-loop DC gain of the control
loop is reduced depending upon the maximum load step
specifications. Voltage positioning can easily be added to
the LTC3707 by loading the ITH pin with a resistive divider
having a Thevenin equivalent voltage source equal to the
midpoint operating voltage of the error amplifier, or 1.2V
(see Figure 8).
The resistive load reduces the DC loop gain while maintaining the linear control range of the error amplifier.
The maximum output voltage deviation can theoretically
be reduced to half or alternatively the amount of output
capacitance can be reduced for a particular application.
A complete explanation is included in Design Solutions
10. (See www.linear.com)
INTVCC
RT2
ITH
RT1
RC
LTC3707
CC
3707 F08
Figure 8. Active Voltage Positioning Applied to the LTC3707
3707fb
22
LTC3707
APPLICATIONS INFORMATION
Efficiency Considerations
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Percent efficiency can
be expressed as:
%Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage of input power.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most
of the losses in LTC3707 circuits: 1) LTC3707 VIN current (including loading on the 3.3V internal regulator),
2) INTVCC regulator current, 3) I2R losses, 4) Topside
MOSFET transition losses.
1. The VIN current has two components: the first is the DC
supply current given in the Electrical Characteristics table,
which excludes MOSFET driver and control currents; the
second is the current drawn from the 3.3V linear regulator
output. VIN current typically results in a small (<0.1%)
loss.
2. INTVCC current is the sum of the MOSFET driver and
control currents. The MOSFET driver current results from
switching the gate capacitance of the power MOSFETs.
Each time a MOSFET gate is switched from low to high
to low again, a packet of charge dQ moves from INTVCC
to ground. The resulting dQ/dt is a current out of INTVCC
that is typically much larger than the control circuit current. In continuous mode, IGATECHG =f(QT+QB), where QT
and QB are the gate charges of the topside and bottom
side MOSFETs.
Supplying INTVCC power through the EXTVCC switch input
from an output-derived source will scale the VIN current
required for the driver and control circuits by a factor of
(Duty Cycle)/(Efficiency). For example, in a 20V to 5V application, 10mA of INTVCC current results in approximately
2.5mA of VIN current. This reduces the mid-current loss
from 10% or more (if the driver was powered directly from
VIN) to only a few percent.
3. I2R losses are predicted from the DC resistances of
the fuse (if used), MOSFET, inductor, current sense resistor, and input and output capacitor ESR. In continuous
mode the average output current flows through L and
RSENSE, but is “chopped” between the topside MOSFET
and the synchronous MOSFET. If the two MOSFETs have
approximately the same RDS(ON), then the resistance of
one MOSFET can simply be summed with the resistances
of L, RSENSE and ESR to obtain I2R losses. For example, if
each RDS(ON) = 30mΩ, RL = 50mΩ, RSENSE = 10mΩ and
RESR = 40mΩ (sum of both input and output capacitance
losses), then the total resistance is 130mΩ. This results
in losses ranging from 3% to 13% as the output current
increases from 1A to 5A for a 5V output, or a 4% to 20%
loss for a 3.3V output. Efficiency varies as the inverse
square of VOUT for the same external components and
output power level. The combined effects of increasingly
lower output voltages and higher currents required by
high performance digital systems is not doubling but
quadrupling the importance of loss terms in the switching
regulator system!
4. Transition losses apply only to the topside MOSFET(s),
and become significant only when operating at high input
voltages (typically 15V or greater). Transition losses can
be estimated from:
Transition Loss = (1.7) VIN2 IO(MAX) CRSS f
Other “hidden” losses such as copper trace and internal
battery resistances can account for an additional 5% to
10% efficiency degradation in portable systems. It is
very important to include these “system” level losses
during the design phase. The internal battery and fuse
resistance losses can be minimized by making sure that
CIN has adequate charge storage and very low ESR at the
switching frequency. A 25W supply will typically require
a minimum of 20μF to 40μF of capacitance having a
maximum of 20mΩ to 50mΩ of ESR. The LTC3707 2phase architecture typically halves this input capacitance
requirement over competing solutions. Other losses
including Schottky conduction losses during dead-time
and inductor core losses generally account for less than
2% total additional loss.
3707fb
23
LTC3707
APPLICATIONS INFORMATION
Checking Transient Response
The regulator loop response can be checked by looking at
the load current transient response. Switching regulators
take several cycles to respond to a step in DC (resistive)
load current. When a load step occurs, VOUT shifts by
an amount equal to ΔILOAD (ESR), where ESR is the effective series resistance of COUT. ΔILOAD also begins to
charge or discharge COUT generating the feedback error
signal that forces the regulator to adapt to the current
change and return VOUT to its steady-state value. During
this recovery time VOUT can be monitored for excessive
overshoot or ringing, which would indicate a stability
problem. OPTI-LOOP compensation allows the transient
response to be optimized over a wide range of output
capacitance and ESR values. The availability of the ITH pin
not only allows optimization of control loop behavior but
also provides a DC coupled and AC filtered closed loop
response test point. The DC step, rise time and settling
at this test point truly reflects the closed loop response.
Assuming a predominantly second order system, phase
margin and/or damping factor can be estimated using the
percentage of overshoot seen at this pin. The bandwidth
can also be estimated by examining the rise time at the
pin. The ITH external components shown in the Figure 1
circuit will provide an adequate starting point for most
applications.
The ITH series RC-CC filter sets the dominant pole-zero
loop compensation. The values can be modified slightly
(from 0.5 to 2 times their suggested values) to optimize
transient response once the final PC layout is done and
the particular output capacitor type and value have been
determined. The output capacitors need to be selected
because the various types and values determine the loop
gain and phase. An output current pulse of 20% to 80%
of full-load current having a rise time of 1μs to 10μs will
produce output voltage and ITH pin waveforms that will
give a sense of the overall loop stability without breaking the feedback loop. Placing a power MOSFET directly
across the output capacitor and driving the gate with an
appropriate signal generator is a practical way to produce
a realistic load step condition. The initial output voltage
step resulting from the step change in output current may
not be within the bandwidth of the feedback loop, so this
signal cannot be used to determine phase margin. This
is why it is better to look at the ITH pin signal which is in
the feedback loop and is the filtered and compensated
control loop response. The gain of the loop will be increased by increasing RC and the bandwidth of the loop
will be increased by decreasing CC. If RC is increased by
the same factor that CC is decreased, the zero frequency
will be kept the same, thereby keeping the phase shift the
same in the most critical frequency range of the feedback
loop. The output voltage settling behavior is related to the
stability of the closed-loop system and will demonstrate
the actual overall supply performance.
A second, more severe transient is caused by switching
in loads with large (>1μF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel
with COUT, causing a rapid drop in VOUT. No regulator can
alter its delivery of current quickly enough to prevent this
sudden step change in output voltage if the load switch
resistance is low and it is driven quickly. If the ratio of
CLOAD to COUT is greater than1:50, the switch rise time
should be controlled so that the load rise time is limited
to approximately 25 • CLOAD. Thus a 10μF capacitor would
require a 250μs rise time, limiting the charging current
to about 200mA.
3707fb
24
LTC3707
APPLICATIONS INFORMATION
Automotive Considerations: Plugging into the
Cigarette Lighter
As battery-powered devices go mobile, there is a natural
interest in plugging into the cigarette lighter in order to
conserve or even recharge battery packs during operation. But before you connect, be advised: you are plugging into the supply from hell. The main power line in an
automobile is the source of a number of nasty potential
transients, including load-dump, reverse-battery, and
double-battery.
Load-dump is the result of a loose battery cable. When the
cable breaks connection, the field collapse in the alternator can cause a positive spike as high as 60V which takes
several hundred milliseconds to decay. Reverse-battery is
50A IPK RATING
12V
just what it says, while double-battery is a consequence of
tow-truck operators finding that a 24V jump start cranks
cold engines faster than 12V.
The network shown in Figure 9 is the most straight forward approach to protect a DC/DC converter from the
ravages of an automotive power line. The series diode
prevents current from flowing during reverse-battery,
while the transient suppressor clamps the input voltage
during load-dump. Note that the transient suppressor
should not conduct during double-battery operation, but
must still clamp the input voltage below breakdown of the
converter. Although the LTC3707 has a maximum input
voltage of 30V, most applications will be limited to 28V
by the MOSFET BVDSS.
VIN
LTC3707
TRANSIENT VOLTAGE
SUPPRESSOR
GENERAL INSTRUMENT
1.5KA24A
3707 F09
Figure 9. Automotive Application Protection
3707fb
25
LTC3707
APPLICATIONS INFORMATION
Design Example
As a design example for one channel, assume VIN =
12V(nominal), VIN = 22V(max), VOUT = 2V, IMAX = 5A,
and f = 300kHz.
The inductance value is chosen first based on a 30% ripple
current assumption. The highest value of ripple current
occurs at the maximum input voltage. Tie the FREQSET
pin to the INTVCC pin for 300kHz operation. The minimum
inductance for 30% ripple current is:
⎛ V ⎞
V
ΔIL = OUT ⎜ 1– OUT ⎟
(f)(L) ⎝
VIN ⎠
A 4.7μH inductor will produce 25% ripple current and a
3.3μH will result in 36%. The peak inductor current will be
the maximum DC value plus one half the ripple current, or
5.92A, for the 3.3μH value. Increasing the ripple current
will also help ensure that the minimum on-time of 200ns
is not violated. The minimum on-time occurs at maximum
VIN:
tON(MIN) =
VOUT
VIN(MAX)f
=
2V
= 303ns
22V(300kHz)
The RSENSE resistor value can be calculated by using the
maximum current sense voltage specification with some
accommodation for tolerances:
RSENSE ≤
60mV
≈ 0.01Ω
5.92A
Since the output voltage is below 2.4V the output resistive
divider will need to be sized to not only set the output
voltage but also to absorb the SENSE pins specified input
current.
Choosing 1% resistors; R1 = 25.5k and R2 = 32.4k yields
an output voltage of 1.816V.
The power dissipation on the top side MOSFET can be
easily estimated. Choosing a Siliconix Si4412DY results
in; RDS(ON) = 0.042Ω, CRSS = 100pF. At maximum input
voltage with T(estimated) = 50°C:
PMAIN =
2V
2
5) [1+(0.005)(50°C – 25°C)]
(
22V
(0.042Ω) + 1.7 (22V )2 (5A )(100pF )(300kHz )
= 230mW
A short-circuit to ground will result in a folded back current
of:
ISC =
25mV 1 ⎛ 200ns(22V) ⎞
+
= 3.2A
0.01Ω 2 ⎜⎝ 3.3µH ⎟⎠
with a typical value of RDS(ON) and δ = (0.005/°C)(20) =
0.1. The resulting power dissipated in the bottom MOSFET
is:
22V – 2V
2
PSYNC =
3.2A ) (1.1) ( 0.042Ω )
(
22V
= 430mW
which is less than under full-load conditions.
CIN is chosen for an RMS current rating of at least 3A at
temperature assuming only this channel is on. COUT is
chosen with an ESR of 0.02Ω for low output ripple. The
output ripple in continuous mode will be highest at the
maximum input voltage. The output voltage ripple due to
ESR is approximately:
VORIPPLE = RESR(ΔIL) = 0.02Ω(1.67A) = 33mVP–P
⎛
⎞
0.8V
R1(MAX) = 24k ⎜
⎝ 2.4V – VOUT ⎟⎠
0.8V ⎞
⎛
= 24K ⎜
= 32k
⎝ 2.4V – 1.8V ⎟⎠
3707fb
26
LTC3707
APPLICATIONS INFORMATION
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC3707. These items are also illustrated graphically in
the layout diagram of Figure 10. The Figure 11 illustrates
the current waveforms present in the various branches
of the 2-phase synchronous regulators operating in the
continuous mode. Check the following in your layout:
1. Are the top N-channel MOSFETs M1 and M3 located
within 1cm of each other with a common drain connection
at CIN? Do not attempt to split the input decoupling for
the two channels as it can cause a large resonant loop.
2. Are the signal and power grounds kept separate? The
combined LTC3707 signal ground pin and the ground return
of CINTVCC must return to the combined COUT (–) terminals.
The path formed by the top N-channel MOSFET, Schottky
diode and the CIN capacitor should have short leads and
PC trace lengths. The output capacitor (–) terminals should
be connected as close as possible to the (–) terminals
of the input capacitor by placing the capacitors next to
each other and away from the Schottky loop described
above.
3. Do the LTC3707 VOSENSE pins resistive dividers connect
to the (+) terminals of COUT? The resistive divider must
be connected between the (+) terminal of COUT and signal
ground and a small VOSENSE decoupling capacitor should
be as close as possible to the LTC3707 SGND pin. The R2
and R4 connections should not be along the high current
input feeds from the input capacitor(s).
RPU
3
R2
4
R1
5
6
8
3.3V
10
11
12
R3
R4
13
14
TG1
SENSE1–
SW1
VOSENSE1
BOOST1
FREQSET
VIN
STBYMD
BG1
FCB
EXTVCC
LTC3707
ITH1
SGND
3.3VOUT
ITH2
INTVCC
PGND
BG2
BOOST2
VOSENSE2
SW2
SENSE2–
TG2
SENSE2+
RUN/SS2
L1
27
RSENSE
VOUT1
26
25
CB1
M1
M2
D1
24
23
COUT1
RIN
22
CIN
CVIN
21
20
CINTVCC
GND
+
9
SENSE1+
PGOOD
+
7
INTVCC
PGOOD
VPULL-UP
(<7V)
+
2
RUN/SS1
28
+
1
VIN
COUT2
19
18
17
D2
CB2
M3
M4
RSENSE
16
15
VOUT2
L2
3707 F10
Figure 10. LTC3707 Recommended Printed Circuit Layout Diagram
3707fb
27
LTC3707
APPLICATIONS INFORMATION
SW1
L1
D1
RSENSE1
COUT1
VOUT1
+
RL1
VIN
RIN
CIN
+
SW2
BOLD LINES INDICATE
HIGH, SWITCHING
CURRENT LINES.
KEEP LINES TO A
MINIMUM LENGTH.
D2
L2
RSENSE2
COUT2
VOUT2
+
RL2
3707 F11
Figure 11. Branch Current Waveforms
4. Are the SENSE – and SENSE + leads routed together with
minimum PC trace spacing? The filter capacitor between
SENSE+ and SENSE– should be as close as possible to
the IC. Ensure accurate current sensing with Kelvin connections at the SENSE resistor.
from sensitive small-signal nodes, especially from the
opposites channel’s voltage and current sensing feedback
pins. All of these nodes have very large and fast moving
signals and therefore should be kept on the “output side”
of the LTC3707 and occupy minimum PC trace area.
5. Is the INTVCC decoupling capacitor connected close to
the IC, between the INTVCC and the power ground pins?
This capacitor carries the MOSFET drivers current peaks.
An additional 1μF ceramic capacitor placed immediately
next to the INTVCC and PGND pins can help improve noise
performance substantially.
7. Use a modified “star ground” technique: a low impedance, large copper area central grounding point on the same
side of the PC board as the input and output capacitors with
tie-ins for the bottom of the INTVCC decoupling capacitor,
the bottom of the voltage feedback resistive divider and
the SGND pin of the IC.
6. Keep the switching nodes (SW1, SW2), top gate nodes
(TG1, TG2), and boost nodes (BOOST1, BOOST2) away
3707fb
28
LTC3707
APPLICATIONS INFORMATION
PC Board Layout Debugging
Start with one controller on at a time. It is helpful to use
a DC-50MHz current probe to monitor the current in the
inductor while testing the circuit. Monitor the output
switching node (SW pin) to synchronize the oscilloscope
to the internal oscillator and probe the actual output voltage
as well. Check for proper performance over the operating
voltage and current range expected in the application. The
frequency of operation should be maintained over the input
voltage range down to dropout and until the output load
drops below the low current operation threshold—typically
10% to 20% of the maximum designed current level in
Burst Mode operation.
The duty cycle percentage should be maintained from cycle
to cycle in a well-designed, low noise PCB implementation.
Variation in the duty cycle at a subharmonic rate can suggest noise pickup at the current or voltage sensing inputs
or inadequate loop compensation. Overcompensation of
the loop can be used to tame a poor PC layout if regulator
bandwidth optimization is not required. Only after each
controller is checked for their individual performance
should both controllers be turned on at the same time.
A particularly difficult region of operation is when one
controller channel is nearing its current comparator trip
point when the other channel is turning on its top MOSFET.
This occurs around 50% duty cycle on either channel due
to the phasing of the internal clocks and may cause minor
duty cycle jitter.
Short-circuit testing can be performed to verify proper
overcurrent latchoff, or 5μA can be provided to the RUN/SS
pin(s) by resistors from VIN to prevent the short-circuit
latchoff from occurring.
Reduce VIN from its nominal level to verify operation
of the regulator in dropout. Check the operation of the
undervoltage lockout circuit by further lowering VIN while
monitoring the outputs to verify operation.
Investigate whether any problems exist only at higher output currents or only at higher input voltages. If problems
coincide with high input voltages and low output currents,
look for capacitive coupling between the BOOST, SW, TG,
and possibly BG connections and the sensitive voltage
and current pins. The capacitor placed across the current
sensing pins needs to be placed immediately adjacent to
the pins of the IC. This capacitor helps to minimize the
effects of differential noise injection due to high frequency
capacitive coupling. If problems are encountered with
high current output loading at lower input voltages, look
for inductive coupling between CIN, Schottky and the top
MOSFET components to the sensitive current and voltage
sensing traces. In addition, investigate common ground
path voltage pickup between these components and the
SGND pin of the IC.
An embarrassing problem, which can be missed in an
otherwise properly working switching regulator, results
when the current sensing leads are hooked up backwards.
The output voltage under this improper hookup will still
be maintained but the advantages of current mode control
will not be realized. Compensation of the voltage loop will
be much more sensitive to component selection. This
behavior can be investigated by temporarily shorting out
the current sensing resistor—don’t worry, the regulator
will still maintain control of the output voltage.
3707fb
29
LTC3707
TYPICAL APPLICATION
59k
1M
100k
0.1μF
2
180pF
1000pF
3
105k, 1%
4
20k
1%
5
INTVCC
6
33pF
7
15k
1000pF 9
33pF
3.3V
10
11
15k
1000pF 12
20k
1%
13
63.4k
1%
180pF
1000pF
14
SENSE1+
TG1
SENSE1–
SW1
VOSENSE1
BOOST1
FREQSET
VIN
STBYMD
BG1
FCB
EXTVCC
LTC3707
ITH1
SGND
3.3VOUT
ITH2
INTVCC
PGND
BG2
BOOST2
VOSENSE2
SW2
SENSE2–
TG2
SENSE2+
RUN/SS2
+
T1, 1:1.8
10μH
33μF
25V
0.015Ω
27
VOUT1
5V
3A; 4A PEAK
26
8
0.1μF
25
M1
M2
5
D1
MBRM
140T3
LT1121
3
ON/OFF
2
1
220k
24
23
10Ω
22
CMDSH-3TR
22μF
50V
150μF, 6.3V
PANASONIC SP
VOUT3
12V
120mA
+
100k
0.1μF
GND
21
20
19
1μF
10V
4.7μF
180μF, 4V
PANASONIC SP
CMDSH-3TR
18
17
1μF
25V
+
8
PGOOD
MBRS1100T3
VPULL-UP
(<7V)
PGOOD
+
0.01μF
RUN/SS1
28
+
1
0.1μF
M3
M4
D2
MBRM
140T3
VOUT2
3.3V
5A; 6A PEAK
16
15
L1
6.3μH
VIN
7V TO
28V
0.01Ω
0.1μF
1628 F12
VIN: 7V TO 28V
VOUT: 5V, 3A/3.3V, 6A/12V, 120mA
SWITCHING FREQUENCY = 300kHz
MI, M2, M3, M4: NDS8410A
L1: SUMIDA CEP123-6R3MC
T1: 10mH 1:1.8 — DALE LPE6562-A262 GAPPED E-CORE OR BH ELECTRONICS #501-0657 GAPPED TOROID
Figure 12. LTC3707 High Efficiency Low Noise 5V/3A, 3.3V/5A, 12/120mA Regulator
3707fb
30
LTC3707
PACKAGE DESCRIPTION
GN Package
28-Lead Plastic SSOP (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1641)
.386 – .393*
(9.804 – 9.982)
.045 ±.005
28 27 26 25 24 23 22 21 20 19 18 17 1615
.254 MIN
.033
(0.838)
REF
.150 – .165
.229 – .244
(5.817 – 6.198)
.0165 ±.0015
.150 – .157**
(3.810 – 3.988)
.0250 BSC
1
RECOMMENDED SOLDER PAD LAYOUT
.015 ± .004
× 45°
(0.38 ± 0.10)
.0075 – .0098
(0.19 – 0.25)
2 3
4
5 6
7
8
9 10 11 12 13 14
.0532 – .0688
(1.35 – 1.75)
.004 – .0098
(0.102 – 0.249)
0° – 8° TYP
.016 – .050
(0.406 – 1.270)
NOTE:
1. CONTROLLING DIMENSION: INCHES
INCHES
2. DIMENSIONS ARE IN
(MILLIMETERS)
.008 – .012
(0.203 – 0.305)
TYP
.0250
(0.635)
BSC
GN28 (SSOP) 0204
3. DRAWING NOT TO SCALE
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
3707fb
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
31
LTC3707
TYPICAL APPLICATION
0.1μF
27pF
2
1000pF
105k
1%
3
4
20k
1%
INTVCC
5
6
0.01μF
7
8
220pF
9
3.3V
10
11
15k
220pF
12
20k
1%
27pF
13
63.4k
1%
1000pF
14
SENSE1+
TG1
SENSE1–
SW1
VOSENSE1
BOOST1
FREQSET
VIN
STBYMD
BG1
FCB
LTC3707
ITH1
SGND
EXTVCC
INTVCC
PGND
3.3VOUT
ITH2
BG2
BOOST2
VOSENSE2
SW2
SENSE2–
TG2
SENSE2+
RUN/SS2
L1
8μH
PGOOD
27
0.015Ω
VOUT1
5V
3A; 4A PEAK
26
0.1μF
25
M1A
M1B
24
23
47μF 6.3V
10Ω
22
22μF
50V
CMDSH-3TR
0.1μF
GND
21
20
1μF
10V
19
+
15k
33pF
PGOOD
+
33pF
RUN/SS1
VPULL-UP
(<7V)
28
+
1
4.7μF
56μF, 4V
CMDSH-3TR
VIN
5.2V TO
28V
18
17
0.1μF
M2A
M2B
16
L2
8μH
15
0.015Ω
VOUT2
3.3V
3A; 4A PEAK
0.1μF
3707 F13
VIN: 5.2V TO 28V
VOUT: 5V, 4A/3.3V, 4A
SWITCHING FREQUENCY = 300kHz
MI, M2: FDS6982S
L1, L2: 8mH SUMIDA CEP1238R0MC
OUTPUT CAPACITORS: PANASONIC SP SERIES
Figure 13. LTC3707 5V/4A, 3.3V/4A Regulator
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC1159
High Efficiency Synchronous Step-Down Switching Regulator
100% DC, Logic Level MOSFETs, VIN < 40V
LTC1438/LTC1439
Dual High Efficiency Low Noise Synchronous Step-Down Switching Regulators POR, Auxiliary Regulator
LTC1438-ADJ
Dual Synchronous Controller with Auxiliary Regulator
LTC1538-AUX
Dual High Efficiency Low Noise Synchronous Step-Down Switching Regulator Auxiliary Regulator, 5V Standby
LTC1539
Dual High Efficiency Low Noise Synchronous Step-Down Switching Regulator 5V Standby, POR, Low-Battery, Aux Regulator
LTC1530
High Power Step-Down Syncrhonous DC/DC Controller in SO-8
High Efficiency 5V to 3.3V Conversion at Up to 15A
LTC1625/LTC1775
No RSENSE™ Current Mode Synchronous Step-Down Controller
97% Efficiency, No Sense Resistor, 16-Pin SSOP
POR, External Feedback Divider
LTC1628/LTC1628-PG Dual High Efficiency 2-Phase Synchronous Step-Down Switching Regulator
Constant Frequency, 5V and 3.3V LDOs, VIN ≤ 36V
LTC1629
Expandable from 2-Phase to 12-Phase, Uses All
Surface Mount Components, No Heat Sink
20A to 200A PolyPhase™ Synchronous Controller
LTC1702
No RSENSE 2-Phase Dual Synchronous Step-Down Controller
550kHz, No Sense Resistor
LTC1703
No RSENSE 2-Phase Dual Synchronous Step-Down Controller with 5-Bit
Mobile VID Control
Mobile Pentium III Processors, 550kHz, VIN ≤ 7V
LT1709
High Efficiency, 2-Phase Synchronous Step-Down Switching Regulator with
5-Bit VID
1.3V ≤ VOUT ≤ 3.5V, Current Mode Ensures Accurate
Current Sharing, 3.5V ≤ VIN ≤ 36V
LTC1735
High Efficiency Synchronous Step-Down Switching Regulator
Output Fault Protection, 16-Pin SSOP
LTC1736
High Efficiency Synchronous Controller with 5-Bit Mobile VID Control
Output Fault Protection, 24-Pin SSOP, 3.5V ≤ VIN ≤ 36V
LTC1929
2-Phase Synchronous Controller
Up to 42A, Uses All Surface Mount Components,
No Heat Sink, 3.5V ≤ VIN ≤ 36V
Adaptive Power, No RSENSE and PolyPhase are trademarks of Linear Technology Corporation. Pentium is a registered trademark of Intel Corporation.
3707fb
32 Linear Technology Corporation
LT 0508 REV B • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
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© LINEAR TECHNOLOGY CORPORATION 2001
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