ON CS5165AGDW16 5−bit synchronous cpu buck controller Datasheet

CS5165A
5−Bit Synchronous CPU
Buck Controller
The CS5165A synchronous 5−bit NFET buck controller is
optimized to manage the power of the next generation Pentium II
processors. It’s V2t control architecture delivers the fastest transient
response (100 ns), and best overall voltage regulation in the industry
today. It’s feature rich design gives end users the maximum flexibility
to implement the best price/performance solutions for their end
products.
The CS5165A has been carefully crafted to maximize performance and
protect the processor during operation. It has a 5−bit DAC on board that
holds a ±1.0% tolerance over temperature. Its on board programmable
Soft−Start insures a control startup, and the FET nonoverlap circuitry
ensures that both FETs do not conduct simultaneously.
The on board oscillator can be programmed up to 1.0 MHz to give
the designer maximum flexibility in choosing external components
and setting systems costs.
The CS5165A protects the processor during potentially catastrophic
events like overvoltage (OVP) and short circuit. The OVP feature is
part of the V2 architecture and does not require any additional
components. During short circuit, the controller pulses the MOSFETs
in a “hiccup” mode (3.0% duty cycle) until the fault is removed. With
this method, the MOSFETs do not overheat or self destruct.
The CS5165A is designed for use in both single processor desktop and
multiprocessor workstation and server applications. The CS5165A’s
current sharing capability allows the designer to build multiple parallel
and redundant power solutions for multiprocessor systems.
The CS5165A contains other control and protection features such as
Power Good, ENABLE, and adaptive voltage positioning. It is
available in a 16 lead SOIC wide body package.
Features
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V2 Control Topology
Dual N−Channel Design
100 ns Controller Transient Response
Excess of 1.0 MHz Operation
5−Bit DAC with 1.0% Tolerance
Power Good Output With Internal Delay
Enable Input Provides Micropower Shutdown Mode
5.0 V and 12 V Operation
Adaptive Voltage Positioning
Remote Sense Capability
Current Sharing Capability
VCC Monitor
Hiccup Mode Short Circuit Protection
Overvoltage Protection (OVP)
Programmable Soft−Start
150 ns PWM Blanking
65 ns FET Nonoverlap Time
40 ns Gate Rise and Fall Times (3.3 nF Load)
Pb−Free Packages are Available*
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MARKING
DIAGRAM
16
16
SO−16WB
DW SUFFIX
CASE 751G
1
CS5165A
AWLYYWWG
1
CS5165A
A
WL
YY
WW
G
= Device Code
= Assembly Location
= Wafer Lot
= Year
= Work Week
= Pb−Free Package
PIN CONNECTIONS
1
16
VFB
COMP
LGND
PWRGD
GATE(L)
PGND
GATE(H)
VCC
VID0
VID1
VID2
VID3
SS
VID4
COFF
ENABLE
ORDERING INFORMATION
Package
Shipping †
CS5165AGDW16
SOIC−16
47 Units/Rail
CS5165AGDW16G
SOIC−16
(Pb−Free)
47 Units/Rail
CS5165AGDWR16
SOIC−16 1000/Tape & Reel
Device
CS5165AGDWR16G SOIC−16 1000/Tape & Reel
(Pb−Free)
†For information on tape and reel specifications,
including part orientation and tape sizes, please
refer to our Tape and Reel Packaging Specifications
Brochure, BRD8011/D.
*For additional information on our Pb−Free strategy and soldering details, please
download the ON Semiconductor Soldering and Mounting Techniques
Reference Manual, SOLDERRM/D.
© Semiconductor Components Industries, LLC, 2005
October, 2005 − Rev. 1
1
Publication Order Number:
CS5165A/D
CS5165A
12 V
5.0 V
1200 mF/10 V × 3
1.0 mF
IRL3103
SS
0.1 mF
0.1 mF
VCC
COMP
PCB trace
6.0 mW
VCC
GATE(H)
COFF
330 pF
1.2 mH
1200 mF/10 V × 5
GATE(L)
VSS
VID4
IRL3103
PWRGD
PGND
VID3
ENABLE
LGND
VID2
CS5165A
VID1
VID0
3.3 k
VID1
VFB
1000 pF
VID0
Pentium II
System
VID2
PWRGD
VID3
ENABLE
VID4
Figure 1. Application Diagram, 5.0 V to 2.8 V @ 14.2 A for 300 MHz Pentium II
MAXIMUM RATINGS
Rating
Operating Junction Temperature, TJ
Lead Temperature Soldering:
Reflow: (SMD styles only) (Note 1)
Storage Temperature Range, TS
Value
Unit
0 to 150
°C
230 peak
°C
−65 to +150
°C
ESD Susceptibility (Human Body Model)
2.0
kV
Maximum ratings are those values beyond which device damage can occur. Maximum ratings applied to the device are individual stress limit
values (not normal operating conditions) and are not valid simultaneously. If these limits are exceeded, device functional operation is not implied,
damage may occur and reliability may be affected.
1. 60 second maximum above 183°C.
MAXIMUM RATINGS
Pin Name
IC Power Input
Soft−Start Capacitor
Compensation Capacitor
Pin Symbol
VMAX
VMIN
ISOURCE
ISINK
VCC
16 V
−0.3 V
N/A
1.5 A peak, 200 mA DC
SS
6.0 V
−0.3 V
200 mA
10 mA
COMP
6.0 V
−0.3 V
10 mA
1.0 mA
VFB
6.0 V
−0.3 V
10 mA
10 mA
COFF
6.0 V
−0.3 V
1.0 mA
50 mA
Voltage ID DAC Inputs
VID0−VID4
6.0 V
−0.3 V
1.0 mA
10 mA
High−Side FET Driver
GATE(H)
16 V
−0.3 V
1.5 A peak, 200 mA DC
1.5 A peak, 200 mA DC
Voltage Feedback Input
Off−Time Capacitor
Low−Side FET Driver
GATE(L)
16 V
−0.3 V
1.5 A peak, 200 mA DC
1.5 A peak, 200 mA DC
Enable Input
ENABLE
6.0 V
−0.3 V
100 mA
1.0 mA
Power Good Output
PWRGD
6.0 V
−0.3 V
10 mA
30 mA
Power Ground
PGND
0V
0V
1.5 A peak, 200 mA DC
N/A
Logic Ground
LGND
0V
0V
100 mA
N/A
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CS5165A
ELECTRICAL CHARACTERISTICS (0°C < TA < +70°C; 0°C < TJ < +125°C; 8.0 V < VCC < 14 V; 2.8 DAC Code:
(VID4 = VID2 = VID1 = VID0 = 1; VID3 = 0); CGATE(H) and CGATE(L) = 3.3 nF; COFF = 330 pF; CSS = 0.1 mF, unless otherwise specified.)
Characteristic
Test Conditions
Min
Typ
Max
Unit
VCC Supply Current
Operating
1.0 V < VFB < VDAC (max on−time)
No Loads on GATE(H) and GATE(L)
−
12
20
mA
Sleep Mode
ENABLE = 0 V
−
300
600
mA
VCC Monitor
Start Threshold
GATE(H) switching
3.75
3.95
4.15
V
Stop Threshold
GATE(H) not switching
3.65
3.87
4.05
V
Hysteresis
Start−Stop
−
80
−
mV
Error Amplifier
VFB Bias Current
VFB = 0 V
−
0.1
1.0
mA
COMP Source Current
COMP = 1.2 V to 3.6 V; VFB = 2.7 V
15
30
60
mA
COMP CLAMP Voltage
VFB = 2.7 V, Adjust COMP voltage for Comp current = 50 mA
0.85
1.0
1.15
V
COMP Clamp Current
COMP = 0 V
0.4
1.0
1.6
mA
COMP Sink Current
VCOMP = 1.2 V; VFB = 3.0 V; VSS > 2.5 V
180
400
800
mA
Open Loop Gain
(Note 2)
50
60
−
dB
Unity Gain Bandwidth
(Note 2)
0.5
2.0
−
MHz
PSRR @ 1.0 kHz
(Note 2)
60
85
−
dB
High Voltage at 100 mA
Measure VCC − GATE
−
1.2
2.0
V
Low Voltage at 100 mA
Measure GATE
−
1.0
1.5
V
Rise Time
1.6 V < GATE < (VCC − 2.5 V)
−
40
80
ns
Fall Time
(VCC − 2.5 V) > GATE > 1.6 V
−
40
80
ns
GATE(H) to GATE(L) Delay
GATE(H) < 2.0 V; GATE(L) > 2.0 V
30
65
100
ns
GATE(L) to GATE(H) Delay
GATE(L) < 2.0 V; GATE(H) > 2.0 V
30
65
100
ns
GATE pulldown
Resistor to PGND, (Note 2)
20
50
115
kW
SS Charge Time
VFB = 0 V
1.6
3.3
5.0
ms
SS Pulse Period
VFB = 0 V
25
100
200
ms
SS Duty Cycle
(Charge Time/Period) × 100
1.0
3.3
6.0
%
SS COMP Clamp Voltage
VFB = 2.7 V; VSS = 0 V
0.50
0.95
1.10
V
VFB Low Comparator
Increase VFB till no SS pulsing and normal Off−time
0.9
1.0
1.1
V
GATE(H) and GATE(L)
Fault Protection
PWM Comparator
Transient Response
VFB = 1.2 to 5.0 V. 500 ns after GATE(H)
(after Blanking time) to GATE(H) = (VCC −1.0 V) to 1.0 V
−
130
180
ns
Minimum Pulse Width
(Blanking Time)
Drive VFB. 1.2 to 5.0 V upon GATE(H) rising edge
(> VCC − 1.0 V), measure GATE(H) pulse width
50
150
250
ns
Normal Off−Time
VFB = 2.7 V
1.0
1.6
2.3
ms
Extended Off−Time
VSS = VFB = 0 V
5.0
8.0
12.0
ms
Time−Out Time
VFB = 2.7 V, Measure GATE(H) Pulse Width
10
30
50
ms
Fault Duty Cycle
VFB = 0V
30
50
70
%
ENABLE Threshold
GATE(H) Switching
0.8
1.15
1.30
V
Shutdown delay (Note 3)
ENABLE−to−GATE(H) < 2.0 V
−
3.0
−
ms
COFF
Time−Out Timer
Enable Input
2. Guaranteed by design, not 100% tested in production.
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CS5165A
ELECTRICAL CHARACTERISTICS (0°C < TA < +70°C; 0°C < TJ < +125°C; 8.0 V < VCC < 14 V; 2.8 DAC Code:
(VID4 = VID2 = VID1 = VID0 = 1; VID3 = 0); CGATE(H) and CGATE(L) = 3.3 nF; COFF = 330 pF; CSS = 0.1 mF, unless otherwise specified.)
Characteristic
Test Conditions
Min
Typ
Max
Unit
Enable Input
Pullup Current
ENABLE = 0 V
3.0
7.0
15
mA
Pullup Voltage
No load on ENABLE pin
1.30
1.8
3.0
V
Input Resistance
ENABLE = 5.0 V, R = (5.0 V − VPULLUP)/IENABLE
10
20
50
kW
Low to High Delay
VFB = (0.8 × VDAC) to VDAC
30
65
110
ms
High to Low Delay
VFB = VDAC to (0.8 × VDAC)
30
75
120
ms
Output Low Voltage
VFB = 2.4 V, IPWRGD = 500 mA
−
0.2
0.3
V
Sink Current Limit
VFB = 2.4 V, PWRGD = 1.0 V
0.5
4.0
15.0
mA
Power Good Output
3. Guaranteed by design, not 100% tested in production.
ELECTRICAL CHARACTERISTICS (0°C < TA < +70°C; 0°C < TJ < +125°C; 8.0 V < VCC < 14 V; 2.8 DAC Code:
(VID4 = VID2 = VID1 = VID0 = 1; VID3 = 0); CGATE(H) and CGATE(L) = 3.3 nF; COFF = 330 pF; CSS = 0.1 mF, unless otherwise specified.)
Characteristic
Test Conditions
Min
Typ
Max
Unit
−1.0
−
+1.0
%
Voltage Identification DAC
Measure VFB = COMP (COFF = 0 V)
25°C ≤ TJ ≤ 125°C; VCC = 12 V
Accuracy (all codes except 11111)
VID4
VID3
VID2
VID1
VID0
1
0
0
0
0
−
3.505
3.540
3.575
V
1
0
0
0
1
−
3.406
3.440
3.474
V
1
0
0
1
0
−
3.307
3.340
3.373
V
1
0
0
1
1
−
3.208
3.240
3.272
V
1
0
1
0
0
−
3.109
3.140
3.171
V
1
0
1
0
1
−
3.010
3.040
3.070
V
1
0
1
1
0
−
2.911
2.940
2.969
V
1
0
1
1
1
−
2.812
2.840
2.868
V
1
1
0
0
0
−
2.713
2.740
2.767
V
1
1
0
0
1
−
2.614
2.640
2.666
V
1
1
0
1
0
−
2.515
2.540
2.565
V
1
1
0
1
1
−
2.416
2.440
2.464
V
1
1
1
0
0
−
2.317
2.340
2.363
V
1
1
1
0
1
−
2.218
2.240
2.262
V
1
1
1
1
0
−
2.119
2.140
2.161
V
0
0
0
0
0
−
2.069
2.090
2.111
V
0
0
0
0
1
−
2.020
2.040
2.060
V
0
0
0
1
0
−
1.970
1.990
2.010
V
0
0
0
1
1
−
1.921
1.940
1.959
V
0
0
1
0
0
−
1.871
1.890
1.909
V
0
0
1
0
1
−
1.822
1.840
1.858
V
0
0
1
1
0
−
1.772
1.790
1.808
V
0
0
1
1
1
−
1.723
1.740
1.757
V
0
1
0
0
0
−
1.673
1.690
1.707
V
0
1
0
0
1
−
1.624
1.640
1.656
V
0
1
0
1
0
−
1.574
1.590
1.606
V
0
1
0
1
1
−
1.525
1.540
1.555
V
0
1
1
0
0
−
1.475
1.490
1.505
V
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CS5165A
ELECTRICAL CHARACTERISTICS (0°C < TA < +70°C; 0°C < TJ < +125°C; 8.0 V < VCC < 14 V; 2.8 DAC Code:
(VID4 = VID2 = VID1 = VID0 = 1; VID3 = 0); CGATE(H) and CGATE(L) = 3.3 nF; COFF = 330 pF; CSS = 0.1 mF, unless otherwise specified.)
Characteristic
Test Conditions
Min
Typ
Max
Unit
Voltage Identification DAC
0
1
1
0
1
−
1.426
1.440
1.455
V
0
1
1
1
0
−
1.376
1.390
1.405
V
0
1
1
1
1
−
1.327
1.340
1.353
V
1
1
1
1
1
−
1.223
1.247
1.273
V
Input Threshold
VID4, VID3, VID2, VID1, VID0
1.000
1.250
2.400
V
Input Pullup Resistance
VID4, VID3, VID2, VID1, VID0
25
50
100
kW
4.85
5.00
5.15
V
Input Pullup Voltage
Threshold Accuracy
Min
Lower Threshold
Typ
Max
Min
Upper Threshold
Typ
Max
Unit
DAC CODE
% of Nominal DAC Output
−12
−8.5
−5.0
5.0
8.5
12
%
VID4
VID3
VID2
VID1
VID0
1
0
0
0
0
3.115
3.239
3.363
3.717
3.841
3.965
V
1
0
0
0
1
3.027
3.148
3.268
3.612
3.732
3.853
V
1
0
0
1
0
2.939
3.056
3.173
3.507
3.624
3.741
V
1
0
0
1
1
2.851
2.965
3.078
3.402
3.515
3.629
V
1
0
1
0
0
2.763
2.873
2.983
3.297
3.407
3.517
V
1
0
1
0
1
2.675
2.782
2.888
3.192
3.298
3.405
V
1
0
1
1
0
2.587
2.690
2.793
3.087
3.190
3.293
V
1
0
1
1
1
2.499
2.599
2.698
2.982
3.081
3.181
V
1
1
0
0
0
2.411
2.507
2.603
2.877
2.973
3.069
V
1
1
0
0
1
2.323
2.416
2.508
2.772
2.864
2.957
V
1
1
0
1
0
2.235
2.324
2.413
2.667
2.756
2.845
V
1
1
0
1
1
2.147
2.233
2.318
2.562
2.647
2.733
V
1
1
1
0
0
2.059
2.141
2.223
2.457
2.539
2.621
V
1
1
1
0
1
1.971
2.050
2.128
2.352
2.430
2.509
V
1
1
1
1
0
1.883
1.958
2.033
2.250
2.322
2.397
V
0
0
0
0
0
1.839
1.912
1.986
2.195
2.268
2.341
V
0
0
0
0
1
1.795
1.867
1.938
2.142
2.213
2.285
V
0
0
0
1
0
1.751
1.821
1.810
2.090
2.159
2.229
V
0
0
0
1
1
1.707
1.775
1.843
2.037
2.105
2.173
V
0
0
1
0
0
1.663
1.729
1.796
1.985
2.051
2.117
V
0
0
1
0
1
1.619
1.684
1.748
1.932
1.996
2.061
V
0
0
1
1
0
1.575
1.638
1.701
1.880
1.942
2.005
V
0
0
1
1
1
1.531
1.592
1.653
1.827
1.888
1.949
V
0
1
0
0
0
1.487
1.546
1.606
1.775
1.834
1.893
V
0
1
0
0
1
1.443
1.501
1.558
1.722
1.779
1.837
V
0
1
0
1
0
1.399
1.455
1.511
1.670
1.725
1.781
V
0
1
0
1
1
1.355
1.409
1.463
1.617
1.671
1.724
V
0
1
1
0
0
1.311
1.363
1.416
1.565
1.617
1.669
V
0
1
1
0
1
1.267
1.318
1.368
1.512
1.562
1.613
V
0
1
1
1
0
1.223
1.272
1.321
1.460
1.508
1.557
V
0
1
1
1
1
1.179
1.226
1.273
1.407
1.454
1.501
V
1
1
1
1
1
1.097
1.141
1.185
1.309
1.353
1.397
V
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CS5165A
PACKAGE PIN DESCRIPTION
PACKAGE PIN #
SOIC−16
PIN SYMBOL
FUNCTION
1, 2, 3, 4, 6
VID0−VID4
Voltage ID DAC input pins. These pins are internally pulled up to 5.0 V if left open. VID4 selects
the DAC range. When VID4 is high (logic one), the Error Amp reference range is 2.14 V to 3.45 V
with 100 mV increments. When VID4 is low (logic zero), the Error Amp reference voltage 1.34 V
to 2.09 V with 50 mV increments.
5
SS
7
COFF
8
ENABLE
Soft−Start Pin. A capacitor from this pin to LGND sets the Soft−Start and fault timing.
Off−Time Capacitor Pin. A capacitor from this pin to LGND sets both the normal and extended off
time.
Output Enable Input. This pin is internally pulled up to 1.8 V. A logic Low (< 0.8) on this pin
disables operation and places the CS5165A into a low current sleep mode.
9
VCC
10
GATE(H)
11
PGND
12
GATE(L)
Low Side Synchronous FET driver pin.
13
PWRGD
Power Good Output. Open collector output drives low when VFB is out of regulation. Active when
ENABLE input is low.
14
LGND
Reference ground. All control circuits are referenced to this pin.
15
COMP
Error Amp output. PWM Comparator reference input. A capacitor to LGND provides Error Amp
compensation.
16
VFB
VCC
Input Power Supply Pin.
High Side Switch FET driver pin.
High current ground for the GATE(H) and GATE(L) pins.
Error Amp, PWM Comparator, and Low VFB Comparator feedback input.
−
7.0 mA
VCC Monitor
−
VCC
+
20 k
ENABLE
3.95 V
3.87V
Circuit Bias
+
1.25 V
VGATE(H)
Enable
Comparator
5.0 V
−
SS Low
Comparator
R
Q
S
Q
+
60 mA
0.7 V
SS
+
2.0 mA
FAULT
FAULT
PGND
FAULT
Latch
SS High
Comparator
VCC
−
VCC1
VGATE(L)
2.5 V
COMP
VID0
Error Amplifier
PGND
+
VID1
5 BIT
DAC
VID2
−
VID3
VID4
−8.5%
−
PWM COMP
Blanking
PWM
Comparator
+8.5%
+
−
+
−
GATE(H) = ON
Maximum
On−Time
Timeout
Q
S
Q
Normal
Off−Time
Timeout
+
Extended
Off−Time
Timeout
PWRGD
R
PWM
Latch
Off−Time
Timeout
GATE(H) = OFF
COFF
One Shot
R
S
65 ms
Delay
VFB
Time−Out
Timer
(30 ms)
−
+
1.0 V
VFB Low
Comparator
LGND
Figure 2. Block Diagram
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Edge Triggered
COFF
Q
CS5165A
200
200
180
180
160
160
140
140
Risetime (ns)
Risetime (ns)
TYPICAL PERFORMANCE CHARACTERISTICS
120
100
80
60
0
0
2000 4000
80
VCC = 12 V
40
TA = 25°C
20
100
60
VCC = 12 V
40
120
TA = 25°C
20
0
6000 8000 10000 12000 14000 16000
0
2000 4000
Load Capacitance (pF)
Figure 3. GATE(L) Risetime vs. Load Capacitance
Figure 4. GATE(H) Risetime vs. Load Capacitance
0.04
DAC Output Voltage Deviation (%)
200
180
160
Falltime (ns)
140
120
100
80
60
40
VCC = 12 V
20
TA = 25°C
0
0
2000 4000
0.02
0
−0.02
−0.04
−0.06
−0.08
−0.1
6000 8000 10000 12000 14000 16000
0
20
40
Load Capacitance (pF)
60
80
100
120
Junction Temperature (°C)
Figure 5. GATE(H) & GATE(L) Falltime vs. Load
Capacitance
Figure 6. DAC Output Voltage vs. Temperature,
DAC Code = 10111, VCC = 12 V
0.04
0.05
0.02
0
0
Output Error (%)
Output Error (%)
6000 8000 10000 12000 14000 16000
Load Capacitance (pF)
−0.02
−0.04
−0.05
−0.10
−0.15
DAC Output Voltage Setting (V)
DAC Output Voltage Setting (V)
Figure 7. Percent Output Error vs. DAC Voltage
Setting, VCC = 12 V, TA = 255C, VID4 = 0
Figure 8. Percent Output Error vs. DAC Output
Voltage Setting VCC = 12 V, TA = 255C, VID4 = 1
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3.54
3.44
3.34
3.24
3.14
3.04
2.94
2.84
2.74
2.64
2.54
2.44
2.34
2.14
2.09
2.04
1.99
1.89
1.94
1.84
1.79
1.69
1.74
1.64
1.59
1.54
1.49
−0.25
1.44
−0.10
1.39
−0.20
1.34
−0.08
2.24
−0.06
CS5165A
APPLICATIONS INFORMATION
THEORY OF OPERATION
frequency. Enhanced noise immunity improves remote sensing
of the output voltage, since the noise associated with long
feedback traces can be effectively filtered.
Line and load regulation are drastically improved because
there are two independent voltage loops. A voltage mode
controller relies on a change in the error signal to compensate
for a deviation in either line or load voltage. This change in the
error signal causes the output voltage to change corresponding
to the gain of the error amplifier, which is normally specified
as line and load regulation. A current mode controller
maintains fixed error signal under deviation in the line voltage,
since the slope of the ramp signal changes, but still relies on a
change in the error signal for a deviation in load. The V2
method of control maintains a fixed error signal for both line
and load variation, since the ramp signal is affected by both line
and load.
V2 Control Method
The V2 method of control uses a ramp signal that is
generated by the ESR of the output capacitors. This ramp is
proportional to the AC current through the main inductor and
is offset by the value of the DC output voltage. This control
scheme inherently compensates for variation in either line or
load conditions, since the ramp signal is generated from the
output voltage itself. This control scheme differs from
traditional techniques such as voltage mode, which generates
an artificial ramp, and current mode, which generates a ramp
from inductor current.
PWM
Comparator
+
GATE(H)
C
GATE(L)
−
Ramp
Signal
Error
Signal
To maximize transient response, the CS5165A uses a
constant off time method to control the rate of output pulses.
During normal operation, the off time of the high side switch
is terminated after a fixed period, set by the COFF capacitor.
To maintain regulation, the V2 control loop varies switch on
time. The PWM comparator monitors the output voltage
ramp, and terminates the switch on time.
Constant off time provides a number of advantages. Switch
duty cycle can be adjusted from 0 to 100% on a pulse by pulse
basis when responding to transient conditions. Both 0% and
100% duty cycle operation can be maintained for extended
periods of time in response to load or line transients. PWM
slope compensation to avoid sub−harmonic oscillations at
high duty cycles is avoided.
Switch on time is limited by an internal 30 ms (typical)
timer, minimizing stress to the power components.
Output
Voltage
Feedback
Error
Amplifier
COMP
Constant Off Time
−
E
+
Reference
Voltage
Figure 9. V2 Control Diagram
The V2 control method is illustrated in Figure 9. The output
voltage is used to generate both the error signal and the ramp
signal. Since the ramp signal is simply the output voltage, it is
affected by any change in the output regardless of the origin of
that change. The ramp signal also contains the DC portion of
the output voltage, which allows the control circuit to drive the
main switch to 0% or 100% duty cycle as required.
A change in line voltage changes the current ramp in the
inductor, affecting the ramp signal, which causes the V2 control
scheme to compensate the duty cycle. Since the change in
inductor current modifies the ramp signal, as in current mode
control, the V2 control scheme has the same advantages in line
transient response.
A change in load current will have an affect on the output
voltage, altering the ramp signal. A load step immediately
changes the state of the comparator output, which controls the
main switch. Load transient response is determined only by the
comparator response time and the transition speed of the main
switch. The reaction time to an output load step has no relation
to the crossover frequency of the error signal loop, as in
traditional control methods.
The error signal loop can have a low crossover frequency,
since transient response is handled by the ramp signal loop.
The main purpose of this ‘slow’ feedback loop is to provide
DC accuracy. Noise immunity is significantly improved, since
the error amplifier bandwidth can be rolled off at a low
Programmable Output
The CS5165A is designed to provide two methods for
programming the output voltage of the power supply. A 5−bit
on board digital to analog converter (DAC) is used to program
the output voltage within two different ranges. The first range
is 2.14 V to 3.54 V in 100 mV steps, the second is 1.34 V to
2.09 V in 50 mV steps, depending on the digital input code.
If all five bits are left open, the CS5165A enters adjust mode.
In adjust mode, the designer can choose any output voltage by
using resistor divider feedback to the VFB pin, as in traditional
controllers. The CS5165A is specifically designed to meet or
exceed Intel’s Pentium II specifications.
Startup
Until the voltage on the VCC supply pin exceeds the 3.95 V
monitor threshold, the Soft−Start and GATE pins are held low.
The FAULT latch is reset (no Fault condition). The output of
the error amplifier (COMP) is pulled up to 1.0 V by the
comparator clamp. When the VCC pin exceeds the monitor
threshold, the GATE(H) output is activated, and the Soft−Start
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CS5165A
capacitor begins charging. The GATE(H) output will remain
on, enabling the NFET switch, until terminated by either the
PWM comparator, or the maximum on time timer.
If the maximum on time is exceeded before the regulator
output voltage achieves the 1.0 V level, the pulse is terminated.
The GATE(H) pin drives low, and the GATE(L) pin drives high
for the duration of the extended off time. This time is set by the
time out timer and is approximately equal to the maximum on
time, resulting in a 50% duty cycle. The GATE(L) pin will then
drive low, the GATE(H) pin will drive high, and the cycle
repeats.
When regulator output voltage achieves the 1.0 V level
present at the COMP pin, regulation has been achieved and
normal off time will ensue. The PWM comparator terminates
the switch on time, with off time set by the COFF capacitor. The
V2 control loop will adjust switch duty cycle as required to
ensure the regulator output voltage tracks the output of the
error amplifier.
The Soft−Start and COMP capacitors will charge to their
final levels, providing a controlled turn on of the regulator
output. Regulator turn on time is determined by the COMP
capacitor charging to its final value. Its voltage is limited by the
Soft−Start COMP clamp and the voltage on the Soft−Start pin.
Trace 1− Soft−Start Pin (2.0 V/div.)
Trace 2− COMP PIn (error amplifier output) (1.0 V/div.)
Trace 4− Regulator Output Voltage (1.0 V/div.)
Figure 11. Demonstration Board Startup Waveforms
Power Supply Sequencing
The CS5165A offers inherent protection from undefined
startup conditions, regardless of the 12 V and 5.0 V supply
power up sequencing. The turn on slew rates of the 12 V and
5.0 V power supplies can be varied over wide ranges without
affecting the output voltage or causing detrimental effects to
the buck regulator.
M 10.0 ms
Trace 1− Regulator Output Voltage (1.0 V/div.)
Trace 2− Inductor Switching Node (5.0 V/div.)
Figure 12. Demonstration Board Enable Startup
Waveforms
Normal Operation
During normal operation, switch off time is constant and
set by the COFF capacitor. Switch on time is adjusted by the
V2 control loop to maintain regulation. This results in changes
in regulator switching frequency, duty cycle, and output
ripple in response to changes in load and line. Output voltage
ripple will be determined by inductor ripple current working
and the ESR of the output capacitors (see Figures 13 and 14).
M 250 ms
Trace 1− Regulator Output Voltage (1.0 V/div.)
Trace 2− Inductor Switching Node (2.0 V/div.)
Trace 3− 12 V Input (VCC) (5.0 V/div.)
Trace 4− 5.0 V Input (1.0 V/div.)
Figure 10. Demonstration Board Startup in
Response to Increasing 12 V and 5.0 V Input
Voltages. Extended Off Time is Followed by Normal
Off Time Operation when Output Voltage Achieves
Regulation to the Error Amplifier Output.
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CS5165A
and carries the output current. With no load, there is no DC
drop across this resistor, producing an output voltage tracking
the Error amps, including the +40 mV offset. When the full
load current is delivered, an 80 mV drop is developed across
this resistor. This results in output voltage being offset
−40 mV low.
The result of Adaptive Voltage Positioning is that
additional margin is provided for a load transient before
reaching the output voltage specification limits. When load
current suddenly increases from its minimum level, the
output capacitor is pre−positioned +40 mV. Conversely, when
load current suddenly decreases from its maximum level, the
output capacitor is pre−positioned −40 mV (see Figures 15,
16, and 17). For best Transient Response, a combination of a
number of high frequency and bulk output capacitors are
usually used.
If the Maximum On−Time is exceeded while responding to
a sudden increase in Load current, a normal off−time occurs
to prevent saturation of the output inductor.
Trace 1− GATE(H) (10 V/div.)
Trace 2− Inductor Switching Node (5.0 V/div.)
Trace 3− Output Inductor Ripple Current (2.0 A/div.)
Trace 4− VOUT ripple (20 mV/div.)
Figure 13. Normal Operation Showing Output Inductor
Ripple Current and Output Voltage Ripple, 0.5 A Load,
VOUT = +2.84 V (DAC = 10111)
Trace 1− GATE(H) (10 V/div.)
Trace 2− Inductor Switching Node (5.0 V/div.)
Trace 3− Output Inductor Ripple Current (2.0 A/div.)
Trace 4− VOUT ripple (20 mV/div.)
Trace 3− Load Current (5.0 A/10 mV/div.)
Trace 4− VOUT (100 mV/div.)
Figure 14. Normal Operation Showing Output Inductor
Ripple Current and Output Voltage Ripple,
ILOAD = 14 A, VOUT = +2.84 V (DAC = 10111)
Figure 15. Output Voltage Transient Response to
a 14 A Load Pulse, VOUT = +2.84 V (DAC = 10111)
Transient Response
The CS5165A V2 control loop’s 100 ns reaction time
provides unprecedented transient response to changes in
input voltage or output current. Pulse by pulse adjustment of
duty cycle is provided to quickly ramp the inductor current to
the required level. Since the inductor current cannot be
changed instantaneously, regulation is maintained by the
output capacitor(s) during the time required to slew the
inductor current.
Overall load transient response is further improved through
a feature called “Adaptive Voltage Positioning”. This
technique pre−positions the output capacitors voltage to
reduce total output voltage excursions during changes in load.
Holding tolerance to 1.0% allows the error amplifiers
reference voltage to be targeted +40 mV high without
compromising DC accuracy. A “Droop Resistor”,
implemented through a PC board trace, connects the Error
Amps feedback pin (VFB) to the output capacitors and load
Trace 1− GATE(H) (10 V/div.)
Trace 2− Inductor Switching Node (5.0 V/div.)
Trace 3− Load Current (5.0 A/div)
Trace 4− VOUT (100 mV/div.)
Figure 16. Output Voltage Transient Response to a
14 A Load Step, VOUT = +2.84 V (DAC = 10111)
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CS5165A
M 25.0 ms
Trace 4− 5.0 V Supply Voltage (2.0 V/div.)
Trace 1− GATE(H) (10 V/div.)
Trace 3− Soft−Start Timing Capacitor (1.0 V/div.)
Trace 2− Inductor Switching Node (5.0 V/div.)
Trace 3− Load Current (5.0 A/div)
Trace 4− VOUT (100 mV/div.)
Trace 2− Inductor Switching Node (2.0 V/div.)
Figure 18. Demonstration Board Hiccup Mode Short
Circuit Protection. Gate Pulses are Delivered While
the Soft−Start Capacitor Charges, and Cease During
Discharge
Figure 17. Output Voltage Transient Response to a 14
A Load Turn−Off, VOUT = +2.84 V (DAC = 10111)
PROTECTION AND MONITORING FEATURES
Short Circuit Protection
A lossless hiccup mode short circuit protection feature is
provided, requiring only the Soft−Start capacitor to implement.
If a short circuit condition occurs the VFB low comparator sets
the FAULT latch. This causes the top FET to shut off,
disconnecting the regulator from it’s input voltage. The
Soft−Start capacitor is then slowly discharged by a 2.0 mA
current source until it reaches it’s lower 0.7 V threshold. The
regulator will then attempt to restart normally, operating in it’s
extended off time mode with a 50% duty cycle, while the
Soft−Start capacitor is charged with a 60 mA charge current.
If the short circuit condition persists, the regulator output
will not achieve the 1.0 V low VFB comparator threshold
before the Soft−Start capacitor is charged to it’s upper 2.5 V
threshold. If this happens the cycle will repeat itself until the
short is removed. The Soft−Start charge/discharge current
ratio sets the duty cycle for the pulses (2.0 mA/60 mA = 3.3%),
while actual duty cycle is half that due to the extended off time
mode (1.65%).
This protection feature results in less stress to the regulator
components, input power supply, and PC board traces than
occurs with constant current limit protection (see Figures 18
and 19).
If the short circuit condition is removed, output voltage will
rise above the 1.0 V level, preventing the FAULT latch from
being set, allowing normal operation to resume.
M 50.0 ms
Trace 4− 5.0 V from PC Power Supply (2.0 V/div.)
Trace 2− Inductor Switching Node (2.0 V/div.)
Figure 19. Demonstration Board Startup with
Regulator Output Shorted To Ground
Overvoltage Protection
Overvoltage protection (OVP) is provided as result of the
normal operation of the V2 control topology and requires no
additional external components. The control loop responds to
an overvoltage condition within 100 ns, causing the top
MOSFET to shut off, disconnecting the regulator from it’s
input voltage. The bottom MOSFET is then activated, resulting
in a “crowbar” action to clamp the output voltage and prevent
damage to the load (see Figures 20 and 21 ). The regulator will
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CS5165A
VCORE
remain in this state until the overvoltage condition ceases or the
input voltage is pulled low. The bottom FET and board trace
must be properly designed to implement the OVP function.If
a dedicated OVP output is required, it can be implemented
using the circuit in Figure 22. In this figure the OVP signal will
go high (overvoltage condition), if the output voltage (VCORE)
exceeds 20% of the voltage set by the particular DAC code and
provided that PWRGD is low. It is also required that the
overvoltage condition be present for at least the PWRGD delay
time for the OVP signal to be activated. The resistor values
shown in Figure 22 are for VDAC = +2.8 V (DAC = 10111).
The VOVP (overvoltage trip−point) can be set using the
following equation:
ǒ
VOVP + VBEQ3 1 ) R2
R1
15 k
R1
Q3
2N3906
+5.0 V
56 k
R2
5.0 k
OVP
20 k
+5.0 V
CS5165A
10 k
PWRGD
Ǔ
10 k
Q2
2N3904
10 K
Q1
2N3906
Figure 22. Circuit To Implement A Dedicated OVP
Output Using The CS5165A
Output Enable Circuit
The Enable pin (pin 8) is used to enable or disable the
regulator output voltage, and is consistent with TTL DC
specifications. It is internally pulled−up. If pulled low (below
0.8 V), the output voltage is disabled. At the same time the
Power Good and Soft−Start pins are pulled low, so that when
normal operation resumes power−up of the CS5165A goes
through the Soft−Start sequence. Upon pulling the Enable pin
low, the internal IC bias is completely shut off, resulting in
total shutdown of the Controller IC.
Power Good Circuit
M 10.0 ms
The Power Good pin (pin 13) is an open−collector signal
consistent with TTL DC specifications. It is externally
pulled−up, and is pulled low (below 0.3 V) when the regulator
output voltage typically exceeds ± 8.5% of the nominal output
voltage. Maximum output voltage deviation before Power
Good is pulled low is ± 12%.
Trace 4− 5.0 V from PC Power Supply (5.0 V/div.)
Trace 1− Regulator Output Voltage (1.0 V/div.)
Trace 2− Inductor Switching Node 5.0 V/div.)
Figure 20. OVP Response to an Input−to−Output
Short Circuit by Immediately Providing 0% Duty
Cycle, Crow−Barring the Input Voltage to Ground
Trace 2− PWRGD (2.0 V/div.)
M 5.00 ms
Trace 4− VOUT (1.0 V/div.)
Trace 4− 5.0 V from PC Power Supply (2.0 V/div.)
Figure 23. PWRGD Signal Becomes Logic High as
VOUT Enters −8.5% of Lower PWRGD Threshold,
VOUT = +2.84 V (DAC = 10111)
Trace 1− Regulator Output Voltage (1.0 V/div.)
Figure 21. OVP Response to an Input−to−Output Short
Circuit by Pulling the Input Voltage to Ground
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CS5165A
Selecting External Components
The CS5165A buck regulator can be used with a wide range
of external power components to optimize the cost and
performance of a particular design. The following information
can be used as general guidelines to assist in their selection.
NFET Power Transistors
Both logic level and standard FETs can be used. The
reference designs derive gate drive from the 12 V supply
which is generally available in most computer systems and
utilize logic level FETs. A charge pump may be easily
implemented to support 5.0 V only systems. Multiple FET’s
may be paralleled to reduce losses and improve efficiency and
thermal management.
Voltage applied to the FET gates depends on the application
circuit used. Both upper and lower gate driver outputs are
specified to drive to within 1.5 V of ground when in the low
state and to within 2.0 V of their respective bias supplies when
in the high state. In practice, the FET gates will be driven rail
to rail due to overshoot caused by the capacitive load they
present to the controller IC. For the typical application where
VCC = 12 V and 5.0 V is used as the source for the regulator
output current, the following gate drive is provided:
Trace 2− PWRGD (2.0 V/div.)
Trace 4− VFB (1.0 V/div.)
Figure 24. Power Good Response to an Out of
Regulation Condition
Figure 24 shows the relationship between the regulated
output voltage VFB and the Power Good signal. To prevent
Power Good from interrupting the CPU unnecessarily, the
CS5165A has a built−in delay to prevent noise at the VFB pin
from toggling Power Good. The internal time delay is designed
to take about 75 ms for Power Good to go low and 65 ms for it
to recover. This allows the Power Good signal to be completely
insensitive to out of regulation conditions that are present for
a duration less than the built in delay (see Figure 25).
It is therefore required that the output voltage attains an out
of regulation or in regulation level for at least the built−in delay
time duration before the Power Good signal can change state.
VGS(TOP) + 12 V * 5.0 V + 7.0 V
VGS(BOTTOM) + 12 V
(see Figure 26)
Trace 3− GATE(H) (10 V/div.)
Trace 1− GATE(H) − 5.0 VIN
Trace 4− GATE(L) (10 V/div.)
Trace 2− Inductor Switching Node (5.0 V/div.)
Figure 26. Gate Drive Waveforms Depicting
Rail to Rail Swing
Trace 2− PWRGD (2.0 V/div.)
Trace 4− VFB (1.0 V/div.)
Figure 25. Power Good is Insensitive to Out of
Regulation Conditions that are Present for a
Duration Less Than the Built In Delay
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CS5165A
Schottky Diode for Synchronous FET
For synchronous operation, a Schottky diode may be placed
in parallel with the synchronous FET to conduct the inductor
current upon turn off of the switching FET to improve
efficiency. The CS5165A reference circuit does not use this
device due to it’s excellent design. Instead, the body diode of
the synchronous FET is utilized to reduce cost and conducts the
inductor current. For a design operating at 200 kHz or so, the
low non−overlap time combined with Schottky forward
recovery time may make the benefits of this device not worth
the additional expense. The power dissipation in the
synchronous MOSFET due to body diode conduction can be
estimated by the following equation:
@ 2.2 V
Power + VBD
Trace 1 = GATE(H) (5.0 V/div.)
Figure 27. Normal Operation Showing the Guaranteed
Non−Overlap Time Between the High and Low−Side
MOSFET Gate Drives, ILOAD = 14 A
Power + 1.6 V
RDSON
RDSON
ƪ
duty cycle
ƫ
Off Time Capacitor (COFF)
The COFF timing capacitor sets the regulator off time:
4848.5
R + R20[1 ) a20(T * 20)]
The preceding equations for duty cycle can also be used
to calculate the regulator switching frequency and select the
COFF timing capacitor:
COFF +
Perioid
where:
R20 = resistance at 20°C
(1 * duty cycle)
4848.5
a + 0.00393
°C
T = operating temperature
R = desired droop resistor value
For temperature T = 50°C, the % R change = 12%
where:
Period +
200 kHz + 0.45 W
2. Mismatch due to L/W. The variation in L/W is
governed by variations due to the PCB manufacturing
process that affect the geometry and the power
dissipation capability of the droop resistor. The error
due to L/W mismatch is typically 1.0%.
3. Thermal Considerations. Due to I2 × R power losses
the surface temperature of the droop resistor will
increase causing the resistance to increase. Also, the
ambient temperature variation will contribute to the
increase of the resistance, according to the formula:
RDSON OF SYNCH FET)
TOFF + COFF
100 ns
1.35 * 1.15 + 16%
1.25
(1 * duty cycle)
VIN)(ILOAD RDSON OF SYNCH FET)
* (ILOAD RDSON OF SWITCH FET)
14.2 A
Adaptive voltage positioning is used to help keep the output
voltage within specification during load transients. To
implement adaptive voltage positioning a “Droop Resistor”
must be connected between the output inductor and output
capacitors and load. This resistor carries the full load current
and should be chosen so that both DC and AC tolerance limits
are met. An embedded PC trace resistor has the distinct
advantage of near zero cost implementation. However, this
droop resistor can vary due to three reasons: 1) the sheet
resistivity variation causes the thickness of the PCB layer to
vary. 2) the mismatch of L/W, and 3) temperature variation.
1. Sheet Resistivity for one ounce copper, the thickness
variation typically 1.15 mil to 1.35 mil. Therefore the
error due to sheet resistivity is:
Duty Cycle =
VOUT ) (ILOAD
switching frequency
“Droop” Resistor for Adaptive Voltage Positioning
Synchronous MOSFET:
Power + ILOAD2
conduction time
This is only 1.1% of the 40 W being delivered to the load.
The CS5165A provides adaptive control of the external
NFET conduction times by guaranteeing a typical 65 ns
non−overlap between the upper and lower MOSFET gate drive
pulses. This feature eliminates the potentially catastrophic
effect of “shoot−through current”, a condition during which
both FETs conduct causing them to overheat, self−destruct, and
possibly inflict irreversible damage to the processor.
The most important aspect of FET performance is RDSON,
which effects regulator efficiency and FET thermal
management requirements.
The power dissipated by the MOSFETs may be estimated as
follows:
Switching MOSFET:
Power + ILOAD2
ILOAD
Where VBD = the forward drop of the MOSFET body
diode. For the CS5165A demonstration board:
Trace 2 = GATE(L) (5.0 V/div.)
1
switching frequency
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CS5165A
Droop Resistor Tolerance
Tolerance due to sheet resistivity variation
Tolerance due to L/W error
Tolerance due to temperature variation
Total tolerance for droop resistor
For most PCBs the copper thickness, t, is 35 mm (1.37
mils) for one ounce copper. ρ = 717.86 mW−mil
For a Pentium II load of 14.2 A the resistance needed to
create a 56 mV drop at full load is:
16%
1.0%
12%
29%
In order to determine the droop resistor value the nominal
voltage drop across it at full load has to be calculated. This
voltage drop has to be such that the output voltage full load
is above the minimum DC tolerance spec.
Response Droop + 56 mV + 56 mV + 3.9 mW
14.2 A
IOUT
The resistivity of the copper will drift with the
temperature according to the following guidelines:
[VDAC(MIN) * VDC(MIN)]
VDROOP(TYP) +
1 ) RDROOP(TOLERANCE)
DR + 12% @ TA + ) 50°C
DR + 34% @ TA + ) 100°C
Example: for a 300 MHz PentiumII, the DC accuracy spec
is 2.74 < VCC(CORE) < 2.9 V, and the AC accuracy spec is
2.67 V < VCC(CORE) < 2.9 3V. The CS5165A DAC output
voltage is +2.812 V < VDAC < +2.868 V. In order not to
exceed the DC accuracy spec, the voltage drop developed
across the resistor must be calculated as follows:
Droop Resistor Width Calculations
The droop resistor must have the ability to handle the load
current and therefore requires a minimum width which is
calculated as follows (assume one ounce copper thickness):
I
W + LOAD
0.05
[V
* VDC PENTIUMII(MIN)]
VDROOP(TYP) + DAC(MIN)
1 ) RDROOP(TOLERANCE)
where:
W = minimum width (in mils) required for proper power
dissipation, and ILOAD Load Current Amps.
The Pentium®II maximum load current is 14.2 A.
Therefore:
+ 2.812 V * 2.74 V + 56 mV
1.3
With the CS5165A DAC accuracy being 1.0%, the internal
error amplifier’s reference voltage is trimmed so that the
output voltage will be 40 mV high at no load. With no load,
there is no DC drop across the resistor, producing an output
voltage tracking the error amplifier output voltage, including
the offset. When the full load current is delivered, a drop of
−56 mV is developed across the resistor. Therefore, the
regulator output is pre−positioned at 40 mV above the
nominal output voltage before a load turn−on. The total
voltage drop due to a load step is DV−40 mV and the
deviation from the nominal output voltage is 40 mV smaller
than it would be if there was no droop resistor. Similarly at full
load the regulator output is pre−positioned at 16 mV below
the nominal voltage before a load turn−off. The total voltage
increase due to a load turn−off is DV−16 mV and the
deviation from the nominal output voltage is 16 mV smaller
than it would be if there was no droop resistor. This is because
the output capacitors are pre−charged to value that is either
40 mV above the nominal output voltage before a load
turn−on or, 16 mV below the nominal output voltage before
a load turn−off (see Figure 15).
Obviously, the larger the voltage drop across the droop
resistor ( the larger the resistance), the worse the DC and
load regulation, but the better the AC transient response.
W + 14.2 A + 284 mils + 0.7213 cm
0.05
Droop Resistor Length Calculation
RDROOP W t
ò
0.0039
284 1.37 + 2113 mil + 5.36 cm
+
717.86
L+
Output Inductor
The inductor should be selected based on its inductance,
current capability, and DC resistance. Increasing the
inductor value will decrease output voltage ripple, but
degrade transient response.
Inductor Ripple Current
Ripple Current +
Example: VIN = +5.0 V, VOUT = +2.8 V, ILOAD = 14.2 A,
L = 1.2 mH, Freq = 200 kHz
Ripple Current +
VRIPPLE + Inductor Ripple Current
The basic equation for laying an embedded resistor is:
L or R + ò
A
Output Capacitor ESR
Example:
VIN = +5.0 V, VOUT = +2.8 V, ILOAD = 14.2 A, L = 1.2 mH,
Switching Frequency = 200 kHz
Output Ripple Voltage = 5.1 A × Output Capacitor ESR
(from manufacturer’s specs)
ESR of Output Capacitors to limit Output Voltage Spikes
L
(W
[(5.0 V * 2.8 V) 2.8 V]
+ 5.1 A
[200 kHz 1.2 mH 5.0 V]
Output Ripple Voltage
Design Rules for Using a Droop Resistor
RAR + ò
[(VIN * VOUT) VOUT]
(Switching Frequency L VIN)
t)
where:
A = W × t = cross−sectional area
ρ = the copper resistivity (mW − mil)
L = length (mils)
W = width (mils)
t = thickness (mils)
ESR +
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15
DVOUT
DIOUT
CS5165A
THERMAL MANAGEMENT
This applies for current spikes that are faster than
regulator response time. Printed Circuit Board resistance
will add to the ESR of the output capacitors.
In order to limit spikes to 100 mV for a 14.2 A Load Step,
ESR = 0.1/14.2 = 0.007 W
Thermal Considerations for Power
MOSFETs and Diodes
In order to maintain good reliability, the junction
temperature of the semiconductor components should be
kept to a maximum of 150°C or lower. The thermal
impedance (junction to ambient) required to meet this
requirement can be calculated as follows:
Inductor Peak Current
Peak Current + Maximum Load Current )
ǒRipple 2CurrentǓ
Example: VIN = +5.0 V, VOUT = +2.8 V, ILOAD = 14.2 A,
L = 1.2 mH, Freq = 200 kHz
Thermal Impedance +
TJUNCTION(MAX) * TAMBIENT
Power
A key consideration is that the inductor must be able to
deliver the Peak Current at the switching frequency without
saturating.
A heatsink may be added to TO−220 components to reduce
their thermal impedance. A number of PC board layout
techniques such as thermal vias and additional copper foil
area can be used to improve the power handling capability of
surface mount components.
Response Time to Load Increase
EMI Management
(limited by Inductor value unless Maximum On−Time is
exceeded)
As a consequence of large currents being turned on and off
at high frequency, switching regulators generate noise as a
consequence of their normal operation. When designing for
compliance with EMI/EMC regulations, additional
components may be added to reduce noise emissions. These
components are not required for regulator operation and
experimental results may allow them to be eliminated. The
input filter inductor may not be required because bulk filter and
bypass capacitors, as well as other loads located on the board
will tend to reduce regulator di/dt effects on the circuit board
and input power supply. Placement of the power component to
minimize routing distance will also help to reduce emissions.
Peak Current + 14.2 A ) (5.1ń2) + 16.75 A
Response Time +
L DIOUT
(VIN * VOUT)
Example: VIN = +5.0 V, VOUT = +2.8 V, L = 1.2 mH, 14.2 A
change in Load Current
Response Time +
1.2 mH 14.2 A
+ 7.7 ms
(5.0 V * 2.8 V)
Response Time to Load Decrease
(limited by Inductor value)
Response Time +
L
Change in IOUT
VOUT
Layout Guidelines
When laying out the CPU buck regulator on a printed
circuit board, the following checklist should be used to ensure
proper operation of the CS5165A.
Example: VOUT = +2.8 V, 14.2 A change in Load Current,
L = 1.2 mH
Response Time +
1.2 mH 14.2 A
+ 6.1 ms
2.8 V
Input and Output Capacitors
These components must be selected and placed carefully to
yield optimal results. Capacitors should be chosen to provide
acceptable ripple on the input supply lines and regulator
output voltage. Key specifications for input capacitors are
their ripple rating, while ESR is important for output
capacitors. For best transient response, a combination of low
value/high frequency and bulk capacitors placed close to the
load will be required.
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16
CS5165A
1. Rapid changes in voltage across parasitic capacitors
and abrupt changes in current in parasitic inductors
are major concerns for a good layout.
2. Keep high currents out of logic grounds.
3. Avoid ground loops as they pick up noise. Use star or
single point grounding. The source of the lower
(synchronous FET) is an ideal point where the input
and output GND planes can be connected.
4. For double−sided PCBs a single large ground plane is
not recommended, since there is little control of
where currents flow and the large surface area can act
as an antenna.
5. Even though double sided PCBs are usually sufficient
for a good layout, four−layer PCBs are the optimum
approach to reducing susceptibility to noise. Use the
two internal layers as the +5.0 V and GND planes, and
the top and bottom layers for the vias.
6. Keep the inductor switching node small by placing
the output inductor, switching and synchronous FETs
close together.
7. The FET gate traces to the IC must be as short,
straight, and wide as possible. Ideally, the IC has to be
placed right next to the FETs.
8. Use fewer, but larger output capacitors, keep the
capacitors clustered, and use multiple layer traces
with heavy copper to keep the parasitic resistance
low.
9. Place the switching FET as close to the +5.0 V input
capacitors as possible.
10. Place the output capacitors as close to the load as
possible.
11. Place the VFB filter resistor in series with theVFB pin
(pin 16) right at the pin.
12. Place the VFB filter capacitor right at the VFB pin
(pin 16).
13. The “Droop” Resistor (embedded PCB trace) has to
be wide enough to carry the full load current.
14. Place the VCC bypass capacitor as close as possible to
the VCC pin.
5.0 V
MBRS120
MBRS120
1.0 mF
1200 mF/10 V × 3
MBRS120
1.0 mF
VCC
VID0
VGATE(H)
VID1 CS5165A
VID2
VGATE(L)
VID3
VID4
330 pF
Droop Resistor
(Embedded PCB trace)
6.0 mW
1.2 mH
+
1200 mF/10 V × 5
Si9410DY
COFF
SS
ENABLE
PWRGD
3.3 k
VFB
LGND
0.1 mF
VCC
VSS
PGND
COMP
0.1 mF
Si4410DY
ENABLE
PWRGD
PENTIUM II
SYSTEM
1000 pF
VID4
VID3
VID2
VID1
VID0
Figure 28. Additional Application Diagram, +5.0 V to +2.8 V @ 14.2 A for 300 MHz Pentium II
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CS5165A
PACKAGE DIMENSIONS
SOIC−16WB
DW SUFFIX
CASE 751G−03
ISSUE C
A
D
9
1
8
h X 45 _
E
0.25
H
8X
M
B
M
16
NOTES:
1. DIMENSIONS ARE IN MILLIMETERS.
2. INTERPRET DIMENSIONS AND TOLERANCES
PER ASME Y14.5M, 1994.
3. DIMENSIONS D AND E DO NOT INLCUDE
MOLD PROTRUSION.
4. MAXIMUM MOLD PROTRUSION 0.15 PER SIDE.
5. DIMENSION B DOES NOT INCLUDE DAMBAR
PROTRUSION. ALLOWABLE DAMBAR
PROTRUSION SHALL BE 0.13 TOTAL IN
EXCESS OF THE B DIMENSION AT MAXIMUM
MATERIAL CONDITION.
q
16X
M
14X
e
B
B
T A
S
B
S
L
A
0.25
MILLIMETERS
DIM MIN
MAX
A
2.35
2.65
A1 0.10
0.25
B
0.35
0.49
C
0.23
0.32
D 10.15 10.45
E
7.40
7.60
e
1.27 BSC
H 10.05 10.55
h
0.25
0.75
L
0.50
0.90
q
0_
7_
A1
SEATING
PLANE
T
C
PACKAGE THERMAL DATA
Parameter
SOIC−16WB
Unit
RqJC
Typical
23
°C/W
RqJA
Typical
105
°C/W
V2 is a trademark of Switch Power, Inc.
Pentium is a registered trademark of Intel Corporation.
ON Semiconductor and
are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice
to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does SCILLC assume any liability
arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages.
“Typical” parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All
operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. SCILLC does not convey any license under its patent rights
nor the rights of others. SCILLC products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other applications
intended to support or sustain life, or for any other application in which the failure of the SCILLC product could create a situation where personal injury or death may occur. Should
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CS5165A/D