Renesas EL7563CM Monolithic 4 amp dc/dc step-down regulator Datasheet

NOT RECOMMENDED FOR NEW DESIGNS
SEE EL7554
DATASHEET
EL7563
FN7296
Rev 2.00
May 13, 2005
Monolithic 4 Amp DC/DC Step-Down Regulator
The EL7563 is an integrated, full-featured synchronous stepdown regulator with output voltage adjustable from 1.0V to
2.5V. It is capable of delivering 4A continuous current at up
to 95% efficiency. The EL7563 operates at a constant
frequency pulse width modulation (PWM) mode, making
external synchronization possible. Patented on-chip
resistorless current sensing enables current mode control,
which provides cycle-by-cycle current limiting, over-current
protection, and excellent step load response. The EL7563
features power tracking, which makes the start-up
sequencing of multiple converters possible. A junction
temperature indicator conveniently monitors the silicon die
temperature, saving the designer time on the tedious
thermal characterization. The minimal external components
and full functionality make this EL7563 ideal for desktop and
portable applications.
Features
The EL7563 is offered in 20-pin SO and 28-pin HTSSOP
packages.
• Oscillator synchronization possible
Pinout
• Over-voltage protection
• Integrated synchronous MOSFETs and current mode
controller
• 4A continuous output current
• Up to 95% efficiency
• Internal patented current sense
• Cycle-by-cycle current limit
• 3V to 3.6V input voltage
• Adjustable output voltage 1V to 2.5V
• Precision reference
• ±0.5% load and line regulation
• Adjustable switching frequency to 1MHz
• Internal soft-start
EL7563
(20-PIN SO)
TOP VIEW
• Junction temperature indicator
• Over-temperature protection
• Under-voltage lockout
C5
0.1µF
1 VREF
EN 20
2 SGND
FB 19
3 COSC
PG 18
• Multiple supply start-up tracking
• Power-good indicator
C4
R4
C3
22
0.22µF
C2
4 VDD
VDRV 17
5 VTJ
VHI 16
2.2nF
VIN
3.3V
D2
390pF
C1
330µF
6 PGND
LX 15
7 PGND
LX 14
8 VIN
PGND 13
9 STP
PGND 12
10 STN
PGND 11
D1
C6
0.22µF
D3
C8
0.22µF
C9
0.1µF
C10
2.2nF
C7
330µF
• 28-pin HTSSOP package
• Pb-Free available (RoHS compliant)
Applications
VOUT
L1
4.7µH
• 20-pin SO (0.300”) package
D4
2.5V
4A
R2
1.58k
R1
1k
• DSP, CPU core, and I/O supplies
• Logic/Bus supplies
• Portable equipment
• DC/DC converter modules
• GTL + Bus power supply
Typical Application Diagrams continued on page 3
Manufactured Under U.S. Patent No. 5,7323,974
FN7296 Rev 2.00
May 13, 2005
Page 1 of 16
EL7563
Ordering Information
PACKAGE
TAPE &
REEL
PKG. DWG. #
EL7563CM
20-Pin SO (0.300”)
-
MDP0027
EL7563CMZ
(See Note)
20-Pin SO (0.300”)
(Pb-free)
-
MDP0027
EL7563CMZ-T13
(See Note)
20-Pin SO (0.300”)
(Pb-free)
13”
MDP0027
EL7563CRE
28-Pin HTSSOP
-
MDP0048
EL7563CRE-T7
28-Pin HTSSOP
7”
MDP0048
EL7563CRE-T13
28-Pin HTSSOP
13”
MDP0048
EL7563CREZ
(See Note)
28-Pin HTSSOP
(Pb-free)
-
MDP0048
EL7563CREZ-T7
(See Note)
28-Pin HTSSOP
(Pb-free)
7”
MDP0048
EL7563CREZ-T13
(See Note)
28-Pin HTSSOP
(Pb-free)
13”
MDP0048
PART NUMBER
NOTE: Intersil Pb-free products employ special Pb-free material sets;
molding compounds/die attach materials and 100% matte tin plate
termination finish, which are RoHS compliant and compatible with
both SnPb and Pb-free soldering operations. Intersil Pb-free products
are MSL classified at Pb-free peak reflow temperatures that meet or
exceed the Pb-free requirements of IPC/JEDEC J STD-020.
FN7296 Rev 2.00
May 13, 2005
Page 2 of 16
EL7563
Absolute Maximum Ratings (TA = 25°C)
Supply Voltage between VIN or VDD and GND . . . . . . . . . . . . +4.5V
VLX Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . VIN +0.3V
Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . GND -0.3V, VDD +0.3V
VHI Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . GND -0.3V, VLX +6V
Storage Temperature . . . . . . . . . . . . . . . . . . . . . . . .-65°C to +150°C
Operating Ambient Temperature . . . . . . . . . . . . . . . .-40°C to +85°C
Operating Junction Temperature . . . . . . . . . . . . . . . . . . . . . . +135°C
CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the
device at these or any other conditions above those indicated in the operational sections of this specification is not implied.
IMPORTANT NOTE: All parameters having Min/Max specifications are guaranteed. Typical values are for information purposes only. Unless otherwise noted, all tests
are at the specified temperature and are pulsed tests, therefore: TJ = TC = TA
DC Electrical Specifications
PARAMETER
VDD = VIN = 3.3V, TA = TJ = 25°C, COSC = 390pF, unless otherwise specified.
DESCRIPTION
VREF
Reference Accuracy
VREFTC
Reference Temperature Coefficient
VREFLOAD
Reference Load Regulation
VRAMP
Oscillator Ramp Amplitude
IOSC_CHG
Oscillator Charge Current
IOSC_DIS
CONDITIONS
MIN
TYP
MAX
1.24
1.26
1.28
50
0 < IREF < 50µA
UNIT
V
ppm/°C
-1
%
1.15
V
0.1V < VOSC < 1.25V
200
µA
Oscillator Discharge Current
0.1V < VOSC < 1.25V
8
mA
IVDD+VDRV
VDD+VDRV Supply Current
VEN = 2.7V, FOSC = 120kHz
IVDD_OFF
VDD Standby Current
EN = 0
2
5.5
6.5
mA
1
1.5
mA
VDD_OFF
VDD for Shutdown
2.4
2.65
V
VDD_ON
VDD for Startup
2.6
2.95
V
TOT
Over Temperature Threshold
135
°C
THYS
Over Temperature Hysteresis
20
°C
ILEAK
Internal FET Leakage Current
ILMAX
Peak Current Limit
RDSON
FET On Resistance
RDSONTC
RDSON Tempco
ISTP
Auxiliary Supply Tracking Positive Input
Pull Down Current
VSTP = VIN/2
ISTN
Auxiliary Supply Tracking Negative Input
Pull Up Current
VSTN = VIN/2
VPGP
Positive Power Good Threshold
With respect to target output voltage
VPGN
Negative Power Good Threshold
EN = 0, LX = 3.3V (low FET), LX = 0V
(high FET)
10
5
Wafer level test only
A
30
-4
µA
60
m
0.2
m/°C
2.5
µA
2.5
4
µA
6
16
%
With respect to target output voltage
-16
-6
%
2.7
VPG_HI
Power Good Drive High
IPG = 1mA
VPG_LO
Power Good Drive Low
IPG = -1mA
VOVP
Over Voltage Protection
VFB
Output Initial Accuracy
ILOAD = 0A
VFB_LINE
Output Line Regulation
VIN = 3.3V, VIN = 10%, ILOAD = 0A
0.5
%
VFB_LOAD
Output Load Regulation
0.5A < ILOAD < 4A
0.5
%
VFB_TC
Output Temperature Stability
-40°C < TA < 85°C, ILOAD = 2A
±1
%
IFB
Feedback Input Pull Up Current
VFB = 0V
100
VEN_HI
EN Input High Level
VEN_LO
EN Input Low Level
IEN
Enable Pull Up Current
FN7296 Rev 2.00
May 13, 2005
V
0.5
10
0.977
0.992
%
1.007
200
2.7
1
VEN = 0
-4
V
V
nA
V
V
-2.5
µA
Page 3 of 16
EL7563
Closed-Loop AC Electrical Specifications
PARAMETER
VS = VIN = 3.3V, TA = TJ = 25°C, COSC = 390pF, unless otherwise specified.
DESCRIPTION
CONDITIONS
MIN
TYP
MAX
UNIT
310
365
420
kHz
FOSC
Oscillator Initial Accuracy
tSYNC
Minimum Oscillator Sync Width
25
ns
MSS
Soft Start Slope
0.5
V/ms
tBRM
FET Break Before Make Delay
15
ns
tLEB
High Side FET Minimum On Time
150
ns
DMAX
Maximum Duty Cycle
95
%
Typical Application Diagrams
(Continued)
C5
1 VREF
EN 28
2 SGND
FB 27
3 COSC
PG 26
0.1µF
C4
22
C3
0.22µF
4 VDD
C2
5 VTJ
C1
330µF
D4
VDRV 25
VHI 24
2.2nF
VIN
3.3V
D2
390pF
R4
D1
6 PGND
LX 23
C6
0.22µF
7 PGND
LX 22
L1
8 PGND
LX 21
4.7µH
9 PGND
LX 20
10 VIN
LX 19
11 VIN
LX 18
12 NC
NC 17
13 STP
PGND 16
14 STN
PGND 15
D3
C8
0.22µF
C9
0.1µF
C10
2.2nF
C7
330µF
VOUT
2.5V
4A
R2
1.58k
R1
1k
28-PIN HTSSOP
FN7296 Rev 2.00
May 13, 2005
Page 4 of 16
EL7563
Pin Descriptions
20-PIN SO
(0.300”)
28-PIN
HTSSOP
PIN NAME
1
1
VREF
Bandgap reference bypass capacitor; typically 0.1µF to SGND
2
2
SGND
Control circuit negative supply or signal ground
3
3
COSC
Oscillator timing capacitor (see performance curves)
4
4
VDD
Control circuit positive supply; normally connected to VIN through an RC filter
5
5
VTJ
Junction temperature monitor; connected with 2.2nF to 3.3nF to SGND
6, 7
6, 7, 8, 9
PGND
8
10, 11
VIN
Power supply input of the regulator; connected to the drain of the high-side NMOS power FET
9
13
STP
Auxiliary supply tracking positive input; tied to regulator output to synchronize start up with a
second supply; leave open for stand alone operation; 2µA internal pull down current
10
14
STN
Auxiliary supply tracking negative input; connect to output of a second supply to synchronize
start up; leave open for stand alone operation; 2µA internal pull up current
11, 12, 13
15, 16
PGND
Ground return of the regulator; connected to the source of the low-side synchronous NMOS
power FET
14, 15
18, 19, 20, 21,
22, 23
LX
Inductor drive pin; high current output whose average voltage equals the regulator output
voltage
16
24
VHI
Positive supply of high-side driver; boot strapped from VDRV to LX with an external 0.22µF
capacitor
17
25
VDRV
18
26
PG
Power good window comparator output; logic 1 when regulator output is within ±10% of target
output voltage
19
27
FB
Voltage feedback input; connected to external resistor divider between VOUT and SGND; a
125nA pull-up current forces VOUT to SGND in the event that FB is floating
20
28
EN
Chip enable, active high; a 2µA internal pull up current enables the device if the pin is left open;
a capacitor can be added at this pin to delay the start of converter
PIN FUNCTION
Ground return of the regulator; connected to the source of the low-side synchronous NMOS
power FET
Positive supply of low-side driver and input voltage for high side boot strap
Typical Performance Curves (20 Pin SO Package)
NOTE: The 28-Pin HTSSOP Package Offers Improved Performance
VIN=3.3V
VIN=3.3V
100
100
VO=2.5V
95
VO=1.8V
EFFICIENCY (%)
EFFICIENCY (%)
95
90
85
80
VO=1.2V
75
70
VO=1V
0
0.5
1
1.5
2
2.5
90
85
VO=1.8V
80
VO=1.2V
75
70
65
3
3.5
LOAD CURRENT IO (A)
FIGURE 1. EL7563CM EFFICIENCY vs IO
FN7296 Rev 2.00
May 13, 2005
VO=2.5V
4
60
0.1
0.6
1.1
1.6
2.1
2.6
3.1
3.6
4
IO (A)
FIGURE 2. EL7563CRE EFFICIENCY
Page 5 of 16
EL7563
Typical Performance Curves (20 Pin SO Package) (Continued)
NOTE: The 28-Pin HTSSOP Package Offers Improved Performance
VIN=3.3V
1.6
1.8
VO=1.8V
1.6
1.4
VO=1.2V
1
0.8
VO=1V
0.6
0.4
VO=2.5V
1.2
1.2
PLOSS (W)
POWER LOSS (W)
1.4
1
0.8
VO=1.2V
0.6
0.4
VO=2.5V
0.2
0.2
0
0
0.5
1
2
1.5
2.5
3
3.5
0
4
0
1
2
OUTPUT CURRENT IO (A)
FIGURE 3. EL7563CM CONVERTER POWER LOSS vs IO
2.505
0.6
VIN=3.6V
VIN=3.3V
0.2
VIN=3.3V
VO (V) (%)
OUTPUT VOLTAGE (V)
VO=2.5V
0.4
2.495
2.49
2.485
VIN=3V
2.48
VIN=3.6V
0
-0.2
-0.4
2.475
VIN=3V
-0.6
2.47
2.465
0.5
1
1.5
2
2.5
3
3.5
-0.8
4
0
0.5
1
1.5
FIGURE 5. EL7563CM LOAD REGULATION
50
3
3.5
4
CONDITION:
EL7563CRE THERMAL PAD SOLDERED TO 2-LAYER
PCB WITH 0.039” THICKNESS AND 1 OZ. COPPER ON
BOTH SIDES
45
WITH NO AIRFLOW
JA (°C/W)
THERMAL RESISTANCE (°C/W)
2.5
FIGURE 6. EL7563CRE LOAD REGULATION
TEST CONDITION:
CHIP IN THE CENTER OF COPPER AREA
46
2
IO (A)
LOAD CURRENT IO (A)
50
4
FIGURE 4. EL7563CRE TOTAL CONVERTER POWER LOSS
VO=2.5V
2.5
3
IO (A)
42
38
40
35
WITH 100 LFPM AIRFLOW
34
30
30
1 OZ. COPPER PCB USED
1
1.5
2
2.5
3
3.5
PCB COPPER HEAT-SINKING AREA (in2)
FIGURE 7. EL7563CM JA vs COPPER AREA
FN7296 Rev 2.00
May 13, 2005
4
25
0
1.5
2
2.5
3
3.5
4
PCB AREA (in2)
FIGURE 8. EL7563CRE THERMAL RESISTANCE vs PCB
AREA - NO AIR FLOW
Page 6 of 16
EL7563
Typical Performance Curves (20 Pin SO Package) (Continued)
NOTE: The 28-Pin HTSSOP Package Offers Improved Performance
JUNCTION TEMPERATURE RISE (°C)
60
45
40
50
35
NO AIRFLOW
100 LFM
30
20
25
20
15
200 LFM
10
0
30
TJ RISE
40
10
500 LPF
0
1
5
3
2
0
4
1
1.5
2
IO (A)
FIGURE 9. EL7563CM JUNCTION TEMPERATURE RISE ON
DEMO BOARD
900
800
IO=4A
700
330
FS (kHz)
OSCILLATOR FREQUENCY (kHz)
4
1000
350
320
310
3.5
FIGURE 10. EL7563CRE JUNCTION TEMPERATURE RISE
ON DEMO BOARD - NO AIR FLOW
360
340
3
2.5
IO (A)
IO=0A
600
500
400
300
300
290
200
280
-40
-20
20
0
40
60
100
100
80
200
300
400
500
600
700
800
900 1000
COSC (pF)
TEMPERATURE (°C)
FIGURE 12. SWITCHING FREQUENCY vs COSC
FIGURE 11. SWITCHING FREQUENCY vs TEMPERATURE
8
1.5
6
1.3
VIN=3.3V
VIN=3.6V
VTJ
ILMT (A)
7
VIN=3V
5
1.1
4
3
-40
0.9
-20
0
20
40
60
80
TJ (°C)
FIGURE 13. CURRENT LIMIT vs TJ
FN7296 Rev 2.00
May 13, 2005
100
120
0
25
50
75
100
125
150
JUNCTION TEMPERATURE (°C)
FIGURE 14. VTJ vs JUNCTION TEMPERATURE
Page 7 of 16
EL7563
Typical Performance Curves (20 Pin SO Package) (Continued)
NOTE: The 28-Pin HTSSOP Package Offers Improved Performance
VIN=3.3V, VO=1.8V, IO=0.2A-4A
VIN=3.3V, VO=1.8V, IO=4A
VIN
VLX
IO
iL
VO
VO
FIGURE 15. SWITCHING WAVEFORMS
FIGURE 16. TRANSIENT RESPONSE
VIN=3.3V, VO=1.8V, IO=0.2A
VIN=3.3V, VO=1.8V, IO=4A
VIN
VO
FIGURE 17. POWER-UP
FIGURE 18. POWER-DOWN
VIN=3.3V, VO=1.8V AT 4A
VIN=3.3V, VO=1.8V AT 4A
EN
EN
VO
VO
FIGURE 19. ENABLE
FN7296 Rev 2.00
May 13, 2005
FIGURE 20. DISABLE
Page 8 of 16
EL7563
Typical Performance Curves (20 Pin SO Package) (Continued)
NOTE: The 28-Pin HTSSOP Package Offers Improved Performance
VIN=3.3V, VO=1.8V, IO=4A TO SHORT
IO
VO
FIGURE 21. SHORT-CIRCUIT PROTECTION
Block Diagram
390pF
0.1µF
VREF
COSC
D2
VTJ
JUNCTION
TEMPERATURE
VOLTAGE
REFERENCE
OSCILLATOR
2.2nF
22
3.3V
PWM
CONTROLLER
POWER
FET
0.22µF
DRIVERS
POWER
TRACKING
0.1µF
VOUT
(2.5V, 4A)
330µF
PGND
EN
D1
4.7µH
POWER
FET
STN
D3
VIN
VDD
0.22µF
STP
0.22µF
VHI
CONTROLLER
SUPPLY
D4
VDRV
1.58k
2.2nF
1k
CURRENT
SENSE
VREF
-
PG
+
SGND
FN7296 Rev 2.00
May 13, 2005
FB
Page 9 of 16
EL7563
Applications Information
Circuit Description
General
The EL7563 is a fixed frequency, current mode controlled
DC/DC converter with integrated N-channel power
MOSFETs and a high precision reference. The device
incorporates all the active circuitry required to implement a
cost effective, user-programmable 4A synchronous stepdown regulator suitable for use in DSP core power supplies.
By combining fused-lead packaging technology with an
efficient synchronous switching architecture, high power
output (10W) can be realized without the use of discrete
external heat sinks.
PWM Controller
The EL7563 regulates output voltage through the use of
current-mode controlled pulse width modulation. The three
main elements in a PWM controller are the feedback loop and
reference, a pulse width modulator whose duty cycle is
controlled by the feedback error signal, and a filter which
averages the logic level modulator output. In a step-down
(buck) converter, the feedback loop forces the time-averaged
output of the modulator to equal the desired output voltage.
Unlike pure voltage-mode control systems, current-mode
control utilizes dual feedback loops to provide both output
voltage and inductor current information to the controller. The
voltage loop minimizes DC and transient errors in the output
voltage by adjusting the PWM duty-cycle in response to
changes in line or load conditions. Since the output voltage is
equal to the time-averaged of the modulator output, the
relatively large LC time constant found in power supply
applications generally results in low bandwidth and poor
transient response. By directly monitoring changes in inductor
current via a series sense resistor the controller's response
time is not entirely limited by the output LC filter and can react
more quickly to changes in line and load conditions. This feedforward characteristic also simplifies AC loop compensation
since it adds a zero to the overall loop response. Through
proper selection of the current-feedback to voltage-feedback
ratio the overall loop response will approach a one-pole
system. The resulting system offers several advantages over
traditional voltage control systems, including simpler loop
compensation, pulse by pulse current limiting, rapid response
to line variation and good load step response.
The heart of the controller is an input direct summing
comparator which sum voltage feedback, current feedback,
slope compensation ramp and power tracking signals
together. Slope compensation is required to prevent system
instability that occurs in current-mode topologies operating
at duty-cycles greater than 50% and is also used to define
the open-loop gain of the overall system. The slope
compensation is fixed internally and optimized for 500mA
inductor ripple current. The power tracking will not contribute
any input to the comparator steady-state operation. Current
FN7296 Rev 2.00
May 13, 2005
feedback is measured by the patented sensing scheme that
senses the inductor current flowing through the high-side
switch whenever it is conducting. At the beginning of each
oscillator period the high-side NMOS switch is turned on.
The comparator inputs are gated off for a minimum period of
time of about 150ns (LEB) after the high-side switch is
turned on to allow the system to settle. The Leading Edge
Blanking (LEB) period prevents the detection of erroneous
voltages at the comparator inputs due to switching noise. If
the inductor current exceeds the maximum current limit
(ILMAX) a secondary over-current comparator will terminate
the high-side switch on time. If ILMAX has not been reached,
the feedback voltage FB derived from the regulator output
voltage VOUT is then compared to the internal feedback
reference voltage. The resultant error voltage is summed
with the current feedback and slope compensation ramp.
The high-side switch remains on until all four comparator
inputs have summed to zero, at which time the high-side
switch is turned off and the low-side switch is turned on.
However, the maximum on-duty ratio of the high-side switch
is limited to 95%. In order to eliminate cross-conduction of
the high-side and low-side switches a 15ns break-beforemake delay is incorporated in the switch drive circuitry. The
output enable (EN) input allows the regulator output to be
disabled by an external logic control signal.
Output Voltage Setting
In general:
R 2

V OUT = 0.992V   1 + -------
R 1

However, due to the relatively low open loop gain of the
system, gain errors will occur as the output voltage and loopgain is changed. This is shown in the performance curves. A
100nA pull-up current from FB to VDD forces VOUT to GND
in the event that FB is floating.
NMOS Power FETs and Drive Circuitry
The EL7563 integrates low on-resistance (30m) NMOS
FETs to achieve high efficiency at 4A. In order to use an
NMOS switch for the high-side drive it is necessary to drive
the gate voltage above the source voltage (LX). This is
accomplished by bootstrapping the VHI pin above the LX
voltage with an external capacitor CVHI and internal switch
and diode. When the low-side switch is turned on and the LX
voltage is close to GND potential, capacitor CVHI is charged
through internal switch to VDRV, typically 6V with external
charge-pump. At the beginning of the next cycle the highside switch turns on and the LX pins begin to rise from GND
to VIN potential. As the LX pin rises the positive plate of
capacitor CVHI follows and eventually reaches a value of
VDRV+VIN, typically 9V, for VIN=3.3V. This voltage is then
level shifted and used to drive the gate of the high-side FET,
via the VHI pin. A value of 0.22µF for CVHI is recommended.
Page 10 of 16
EL7563
Reference
Junction Temperature Sensor
A 1.5% temperature compensated bandgap reference is
integrated in the EL7563. The external VREF capacitor acts
as the dominant pole of the amplifier and can be increased
in size to maximize transient noise rejection. A value of
0.1µF is recommended.
An internal temperature sensor continuously monitors die
temperature. In the event that die temperature exceeds the
thermal trip-point, the system is in fault state and will be shut
down. The upper and low trip-points are set to 135°C and
115°C respectively.
Oscillator
The VTJ pin is an accurate indication of the internal silicon
junction temperature (see performance curve.) The junction
temperature TJ (°C) can be deducted from the following
relation:
The system clock is generated by an internal relaxation
oscillator with a maximum duty-cycle of approximately 95%.
Operating frequency can be adjusted through the COSC pin
or can be driven by an external source. If the oscillator is
driven by an external source care must be taken in selecting
the ramp amplitude. Since CSLOPE value is derived from the
COSC ramp, changes to COSC ramp will change the
CSLOPE compensation ramp which determine the open-loop
gain of the system.
When external synchronization is required, always choose
COSC such that the free-running frequency is at least 20%
lower than that of sync source to accommodate component
and temperature variations. Figure 22 shows a typical
connection.
100pF
BAT54S
EXTERNAL
OSCILLATOR
390pF
1
20
2
19
3
18
5
16
6
EL7563
1.2 – VTJ
T J = 75 + ------------------------0.00384
Where VTJ is the voltage at VTJ pin in volts.
Power Good and Power On Reset
During power up the output regulator will be disabled until
VIN reaches a value of approximately 2.9V. About 300mV
hysteresis is present to eliminate noise-induced
oscillations.
Under-voltage and over-voltage conditions on the regulator
output are detected through an internal window comparator.
A logic high on the PG output indicates that the regulated
output voltage is within about +10% of the nominal selected
output voltage.
15
7
14
8
13
9
12
10
11
FIGURE 22. OSCILLATOR SYNCHRONIZATION
FN7296 Rev 2.00
May 13, 2005
Page 11 of 16
EL7563
Power Tracking
2. Offset Tracking
The power tracking pins STP and STN are the inputs to a
comparator, whose HI output forces the PWM controller to
skip switching cycle.
The intended start-up sequence is shown in Figure 24. In
this configuration, VC will not start until VP reaches a preset
value of:
1. Linear Tracking
RB
----------------------  V IN
RA + RB
In this application, it is always the case that the lower voltage
supply VC tracks the higher output supply VP. See Figure 23.
1
20
2
19
6
15
7
EL7563
8
VC
14
13
9
+
-
10
VP
12
11
1
20
2
19
6
15
VOUT
VC
TIME
7
EL7563
8
VP
14
13
9
+
-
10
12
11
FIGURE 23. LINEAR POWER TRACKING
1
20
2
19
6
15
VIN
7
RA
8
9
RB
10
EL7563
VC
14
13
STP
+
STN -
VP
12
11
1
20
2
19
6
15
VOUT
VC
TIME
7
EL7563
8
9
10
VP
14
13
STP
+
STN -
12
11
FIGURE 24. OFFSET POWER TRACKING
FN7296 Rev 2.00
May 13, 2005
Page 12 of 16
EL7563
The second way of offset tracking is to use the EN and
Power Good pins, as shown in Figure 25. In this
configuration, VP does not have to be larger than VC.
3. External Soft Start
An external soft start can be combined with auxiliary supply
tracking to provide desired soft start other than internally
preset soft start (Figure 26). The appropriate start-up time is:
VO
t s = R  C  --------V IN
1
EN 20
2
19
3
PG 18
5
16
6
EL7563
15
7
14
8
13
9
12
10
11
VC
VP
VC
1
EN 20
2
19
3
PG 18
5
16
6
EL7563
TIME
15
7
14
8
13
9
12
10
11
VP
FIGURE 25. OFFSET TRACKING
VIN
1
20
2
19
6
15
7
R
EL7563
8
9
10
VOUT
14
13
STP
STN
+
-
12
11
C
FIGURE 26. EXTERNAL SOFT START
FN7296 Rev 2.00
May 13, 2005
Page 13 of 16
EL7563
The EL7563CRE utilizes the 28-pin HTSSOP package. The
majority of heat is dissipated through the heat pad exposed
at the bottom of the package. Therefore, the heat pad needs
to be soldered to the PCB. The thermal resistance for this
package is as low as 29°C/W, better than that of the SO20.
Typical performance is shown in the curve section. The
actual junction temperature can be measured at VTJ pin.
Start-up Delay
A capacitor can be added to the EN pin to delay the
converter start-up (Figure 27) by utilizing the pull-up current.
The delay time is approximately:
t d  ms  = 1200  C  F 
Thermal Management
Since the thermal performance of the IC is heavily
dependent on the board layout, the system designer should
exercise care during the design phase to ensure that the IC
will operate under the worst-case environmental conditions.
The EL7563CM utilizes “fused lead” packaging technology in
conjunction with the system board layout to achieve a lower
thermal resistance than typically found in standard SO20
packages. By fusing (or connecting) multiple external leads
to the die substrate within the package, a very conductive
heat path is created to the outside of the package. This
conductive heat path MUST then be connected to a heat
sinking area on the PCB in order to dissipate heat out and
away from the device. The conductive paths for the SO20
package are the fused leads: # 6, 7, 11, 12, and 13. If a
sufficient amount of PCB metal area is connected to the
fused package leads, a junction-to-ambient resistance of
43°C/W can be achieved (compared to 85°C/W for a
standard SO20 package). The general relationship between
PCB heat-sinking metal area and the thermal resistance for
this package is shown in the Performance Curves section of
this data sheet. It can be readily seen that the thermal
resistance for this package approaches an asymptotic value
of approximately 43°C/W without any airflow, and 33°C/W
with 100 LFPM airflow. Additional information can be found
in Application Note #8 (Measuring the Thermal Resistance
of Power Surface-Mount Packages). For a thermal shutdown
die junction temperature of 135°C, and power dissipation of
1.5W, the ambient temperature can be as high as 70°C
without airflow. With 100 LFPM airflow, the ambient
temperature can be extended to 85°C.
1
20
2
19
6
15
Layout Considerations
The layout is very important for the converter to function
properly. Power Ground ( ) and Signal Ground (
)
should be separated to ensure that the high pulse current in
the Power Ground never interferes with the sensitive signals
connected to Signal Ground. They should only be connected
at one point (normally at the negative side of either the input
or output capacitor.)
The trace connected to the FB pin is the most sensitive
trace. It needs to be as short as possible and in a “quiet”
place, preferably between PGND or SGND traces.
In addition, the bypass capacitor connected to the VDD pin
needs to be as close to the pin as possible.
The heat of the chip is mainly dissipated through the PGND
pins and through the heat pad at the bottom of the CRE
package. Maximizing the copper area around these pins is
preferable. In addition, a solid ground plane is always helpful
for the EMI performance.
The demo board is a good example of layout based on these
principles. Please refer to the EL7563 Application Brief for
the layout.
C
7
EL7563
8
9
10
VOUT
14
VIN
13
STP
+
STN -
12
VO
td
11
TIME
FIGURE 27. START-UP DELAY
FN7296 Rev 2.00
May 13, 2005
Page 14 of 16
EL7563
SO Package Outline Drawing
FN7296 Rev 2.00
May 13, 2005
Page 15 of 16
EL7563
HTSSOP Package Outline Drawing
NOTE: The package drawing shown here may not be the latest version. To check the latest revision, please refer to the Intersil website at
<http://www.intersil.com/design/packages/index.asp>
© Copyright Intersil Americas LLC 2003-2005. All Rights Reserved.
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For additional products, see www.intersil.com/en/products.html
Intersil products are manufactured, assembled and tested utilizing ISO9001 quality systems as noted
in the quality certifications found at www.intersil.com/en/support/qualandreliability.html
Intersil products are sold by description only. Intersil may modify the circuit design and/or specifications of products at any time without notice, provided that such
modification does not, in Intersil's sole judgment, affect the form, fit or function of the product. Accordingly, the reader is cautioned to verify that datasheets are
current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its
subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or
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For information regarding Intersil Corporation and its products, see www.intersil.com
FN7296 Rev 2.00
May 13, 2005
Page 16 of 16