Intersil ISL6150IB-T Negative voltage hot plug controller Datasheet

ISL6140, ISL6150
Features
The ISL6140 is an 8 Ld, negative voltage hot plug
controller that allows a board to be safely inserted and
removed from a live backplane. Inrush current is limited
to a programmable value by controlling the gate voltage
of an external N-channel pass transistor. The pass
transistor is turned off if the input voltage is less than the
undervoltage threshold, or greater than the overvoltage
threshold. A programmable electronic circuit breaker
protects the system against shorts. The active low
PWRGD signal can be used to directly enable a power
module (with a low enable input)
• Low Side External NFET Switch
The ISL6150 is the same part, but with an active high
PWRGD signal.
• Operates from -10V to -80V (-100V absolute max
rating) or +10V to +80V (+100V absolute max
rating)
• Programmable Inrush Current
• Programmable Electronic Circuit Breaker
(overcurrent shutdown)
• Programmable Overvoltage Protection
• Programmable Undervoltage Lockout
• Power Good Control Output
- PWRGD Active High: (H Version) ISL6150
- PWRGD active Low: (L Version) ISL6140
• Pb-free available (RoHS compliant)
Applications
• VoIP (Voice over Internet Protocol) Servers
• Telecom systems at -48V
• Negative Power Supply Control
• +24V Wireless Base Station Power
Related Literature
• ISL6140/50EVAL1 Board Set, AN9967
• ISL6116 Hot Plug Controller, FN9100
NOTE: See www.intersil.com/hotplug for more information.
Typical Application
(RL and CL are the Load)
GND
GND
R4
R3 = 18kΩ (5%)
CL = 100µF (100V)
VDD
UV
R5
PWRGD
ISL6140
OV
R6
VEE
SENSE
GATE
C1
R2
DRAIN
R3
(LOAD)
C2
CL
RL
-48V IN
R1
Q1
-48VOUT
R5 = 9.09kΩ (1%)
R4 = 562kΩ (1%)
R6 = 10kΩ (1%)
C2 = 3.3nF (100V)
C1 = 150nF (25V)
Q1 = IRF530 (100V, 17A, 0.11Ω)
R2 = 10Ω (5%)
R1 = 0.02Ω (1%)
April 29, 2010
FN9039.4
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright Intersil Americas Inc. 2001, 2003, 2004, 2010. All Rights Reserved
All other trademarks mentioned are the property of their respective owners.
ISL6140, ISL6150
Negative Voltage Hot Plug Controller
ISL6140, ISL6150
Pin Configuration
ISL6140, ISL6150
(8 LD SOIC)
TOP VIEW
PWRGD/PWRGD 1
8 VDD
OV
2
7 DRAIN
UV
3
6 GATE
VEE
4
5 SENSE
ISL6140 has active Low (L version) PWRGD output pin
ISL6150 has active High (H version) PWRGD output pin
Ordering Information
PART
NUMBER
(Notes 2, 3)
PART MARKING
TEMP.
RANGE (°C)
0 to +70
PACKAGE
8 Ld SOIC (Pb-Free)
PKG.
DWG. #
ISL6140CBZ
ISL61 40CBZ
M8.15
ISL6140CBZ-T (Note 1)
ISL61 40CBZ
0 to +70
8 Ld SOIC (Pb-Free)
M8.15
ISL6140IBZ-T (Note 1)
ISL61 40IBZ
-40 to +85
8 Ld SOIC (Pb-Free)
M8.15
ISL6140IBZ
ISL61 40IBZ
-40 to +85
ISL6150CB
ISL 6150CB
0 to +70
8 Ld SOIC (Pb-Free)
M8.15
8 Ld SOIC
M8.15
ISL6150CBZ
ISL61 50CBZ
0 to +70
8 Ld SOIC (Pb-Free)
M8.15
ISL6150CBZ-T (Note 1)
ISL61 50CBZ
0 to +70
8 Ld SOIC (Pb-Free)
M8.15
ISL6150IB-T (Note 1)
ISL 6150IB
0 to +70
8 Ld SOIC
M8.15
ISL6150IBZ
ISL61 50IBZ
-40 to +85
8 Ld SOIC (Pb-Free)
M8.15
ISL6150IBZ-T (Note 1)
ISL61 50IBZ
-40 to +85
8 Ld SOIC (Pb-Free)
M8.15
NOTES:
1. Please refer to TB347 for details on reel specifications.
2. These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach
materials, and 100% matte tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both
SnPb and Pb-free soldering operations). Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that
meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
3. For Moisture Sensitivity Level (MSL), please see device information page forISL6140. For more information on MSL please see
techbrief TB363.
Pin Description
PWRGD (ISL6140; L Version) Pin 1
This digital output is an open-drain pull-down device.
The Power Good comparator looks at the DRAIN pin
voltage compared to the internal VPG reference (VPG is
nominal 1.7V); this essentially measures the voltage
drop across the external FET and sense resistor. If the
voltage drop is small (<1.7V is normal), the PWRGD
pin pulls low (to VEE); this can be used as an active
low enable for an external module. If the voltage drop
is too large (>1.7V indicates some kind of short or
overload condition), the pull-down device shuts off,
and the pin becomes high impedance. Typically, an
external pull-up of some kind is used to pull the pin
high (many brick regulators have a pull-up function
built in).
2
PWRGD (ISL6150; H Version) Pin 1
This digital output is a variation of an open-drain
pull-down device. The power good comparator is the
same as described above, but the polarity of the output
is reversed, as follows:
If the voltage drop across the FET is too large (>1.7V),
the open drain pull-down device will turn on, and sink
current to the DRAIN pin. If the voltage drop is small
(<1.7V), a 2nd pull-down device in series with a 6.2k
resistor (nominal) sinks current to VEE; if the external
pull-up current is low enough (<1mA, for example),
the voltage drop across the resistor will be big enough
to look like a logic high signal (in this example,
1mA*6.2kΩ = 6.2V). This pin can thus be used as an
active high enable signal for an external module.
FN9039.4
ISL6140, ISL6150
Note that for both versions, although this is a digital
pin functionally, the logic high level is determined by
the external pull-up device, and the power supply to
which it is connected; the IC will not clamp it below the
VDD voltage. Therefore, if the external device does not
have its own clamp, or if it would be damaged by a
high voltage, then an external clamp might be
necessary.
OV (OVERVOLTAGE) Pin 2
This analog input compares the voltage on the pin to
an internal voltage reference (nominal 1.223V). When
the input goes above the reference (low to high
transition), that signifies an OV (overvoltage)
condition, and the GATE pin is immediately pulled low
to shut off the external FET. Since there is 20mV of
nominal hysteresis built in, the GATE will remain off
until the OV pin drops below a 1.203V (nominal) high
to low threshold. A typical application will use an
external resistor divider from VDD to VEE, to set the OV
level as desired; a three-resistor divider can set both
OV and UV.
UV (Undervoltage) Pin 3
This analog input compares the voltage on the pin to
an internal voltage reference (nominal 1.223V). When
the input goes below the reference (high to low
transition), that signifies an UV (Under-Voltage)
condition, and the GATE pin is immediately pulled low
to shut off the external FET. Since there is 20mV of
nominal hysteresis built in, the GATE will remain off
until the UV pin rises above a 1.243V (nominal) low to
high threshold. A typical application will use an
external resistor divider from VDD to VEE, to set the UV
level as desired; a three-resistor divider can set both
OV and UV.
If there is an overcurrent condition, the GATE pin is
latched off, and the UV pin is then used to reset the
overcurrent latch; the pin must be externally pulled
below its trip point, and brought back up (toggled) in
order to turn the GATE back on (assuming the fault
condition has disappeared).
VEE Pin 4
This is the most Negative Supply Voltage, such as in a 48V system. Most of the other signals are referenced
3
relative to this pin, even though it may be far away
from what is considered a GND reference.
SENSE Pin 5
This analog input measures the voltage drop across an
external sense resistor (between SENSE and VEE), to
determine if the current exceeds an overcurrent trip
point, equal to nominal (50mV/RSENSE). Noise spikes
of less than 2µs are filtered out; if longer spikes need
to be filtered, an additional RC time constant can be
added to stretch the time (see Figure 29; note that the
FET must be able to handle the high currents for the
additional time). To disable the overcurrent function,
connect the SENSE pin to VEE.
GATE Pin 6
This analog output drives the gate of the external FET
used as a pass transistor. The GATE pin is high (FET is
on) when UV pin is high (above its trip point); the OV
pin is low (below its trip point), and there is no
overcurrent condition (VSENSE - VEE <50mV). If any of
the 3 conditions are violated, the GATE pin will be
pulled low, to shut off the FET.
The Gate is driven high by a weak (-45µA nominal)
pull-up current source, in order to slowly turn on the
FET. It is driven low by a strong (32mA nominal) pulldown device, in order to shut off the FET very quickly
in the event of an overcurrent or shorted condition.
DRAIN Pin 7
This analog input compares the voltage of the external
FET DRAIN to the internal VPG reference (nominal
1.7V), for the Power Good function.
Note that the Power Good comparator does NOT turn
off the GATE pin. However, whenever the GATE is
turned off (by OV, UV or SENSE), the Power Good
Comparator will usually then switch to the
power-NOT-good state, since an off FET will have the
supply voltage across it.
VDD Pin 8
This is the most positive power supply pin. It can range
from +10 to +80V (Relative to VEE). If operation down
near 10V is expected, the user should carefully choose
a FET to match up with the reduced GATE voltage
shown in the specification table.
FN9039.4
ISL6140, ISL6150
.
Absolute Maximum Ratings
Thermal Information
Supply Voltage (VDD to VEE) . . . . . . . . . . . . . -0.3V to 100V
DRAIN, PWRGD, PWRGD Voltage . . . . . . . . . . -0.3V to 100V
UV, OV Input Voltage . . . . . . . . . . . . . . . . . . . -0.3V to 60V
SENSE, GATE Voltage . . . . . . . . . . . . . . . . . . . -0.3V to 20V
ESD Rating
Human Body Model (Per MIL-STD-883 Method 3015.7 . . .2000V
Thermal Resistance (Typical, Note 4)
θJA (°C/W)
8 Lead SOIC . . . . . . . . . . . . . . . . . . . . . . .
95
Maximum Junction Temperature (Plastic Package) . . +150°C
Maximum Storage Temperature Range . . . -65°C to +150°C
Pb-Free Reflow Profile . . . . . . . . . . . . . . . . . .see link below
http://www.intersil.com/pbfree/Pb-FreeReflow.asp
Operating Conditions
Temperature Range (Industrial) . . . . . . . . . . -40°C to +85°C
Temperature Range (Commercial) . . . . . . . . . 0°C to +70°C
Supply Voltage Range (Typical) . . . . . . . . . . . 36V to +72V
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact
product reliability and result in failures not covered by warranty.
NOTES:
4. θJA is measured with the component mounted on a high effective thermal conductivity test board in free air. See Tech Brief
TB379 for details.
Electrical Specifications
VDD = +48V, VEE = +0V Unless Otherwise Specified. All tests are over the full temperature
range; either Commercial (0°C to +70°C) or Industrial (-40°C to +85°C). Typical specs are
at +25°C. Boldface limits apply over the operating temperature range, -40°C to
+85°C.
PART NUMBER
OR GRADE
PARAMETER
SYMBOL
TEST
CONDITIONS
TEST
MIN
MAX
LEVEL OR
NOTES (Note 7) TYP (Note 7) UNITS
DC PARAMETRIC
Supply Operating Range
VDD
Supply Current
IDD
Gate Pin Pull-Up Current
Gate Pin Pull-Down Current
10
-
80
V
UV = 3V; OV = VEE;
SENSE = VEE; VDD = 80V
0.6
0.9
1.3
mA
IPU
Gate Drive on, VGATE = VEE
-30
-45
-60
µA
IPD
Gate Drive off; any fault
condition
24
32
70
mA
(VGATE - VEE), 17V ≤ VDD ≤ 80V
10
14
15
V
5.4
6.2
15
V
40
50
60
mV
-
0
-0.5
µA
GATE PIN
External Gate Drive
Δ-VGATE
(VGATE - VEE), 10V ≤ VDD ≤ 17V
5
SENSE PIN
Circuit Breaker Trip Voltage
VCB
SENSE Pin Current
ISENSE
VCB = (VSENSE - VEE)
VSENSE = 50mV
UV PIN
UV Pin High Threshold Voltage
VUVH
UV Low to High Transition
1.213
1.243
1.272
V
UV Pin Low Threshold Voltage
VUVL
UV High to Low Transition
1.198
1.223
1.247
V
7
20
50
mV
-
-0.05
-0.5
µA
UV Pin Hysteresis
VUVHY
UV Pin Input Current
IINUV
VUV = VEE
OV Pin High Threshold Voltage
VOVH
OV Low to High Transition
1.198
1.223
1.247
V
OV Pin Low Threshold Voltage
VOVL
OV High to Low Transition
1.165
1.203
1.232
V
7
20
50
mV
-
-0.05
-0.5
µA
OV PIN
OV Pin Hysteresis
VOVHY
OV Pin Input Current
IINOV
4
VOV = VEE
FN9039.4
ISL6140, ISL6150
Electrical Specifications
VDD = +48V, VEE = +0V Unless Otherwise Specified. All tests are over the full temperature
range; either Commercial (0°C to +70°C) or Industrial (-40°C to +85°C). Typical specs are
at +25°C. Boldface limits apply over the operating temperature range, -40°C to
+85°C. (Continued)
PART NUMBER
OR GRADE
PARAMETER
SYMBOL
TEST
CONDITIONS
TEST
MIN
MAX
LEVEL OR
NOTES (Note 7) TYP (Note 7) UNITS
DRAIN PIN
Power Good Threshold (L to H)
VPGLH
VDRAIN - VEE, Low to High
Transition
1.55
1.70
1.87
V
Power Good Threshold (H to L)
VPGHL
VDRAIN - VEE, High to Low
Transition
1.10
1.25
1.42
V
Power Good Threshold Hysteresis
VPGHY
0.30
0.45
0.60
V
Drain Input Bias Current
IDRAIN
10
35
60
µA
(VDRAIN - VEE) < VPG
IOUT = 1mA
-
0.28
0.50
V
IOUT = 3mA
-
0.88
1.20
IOUT = 5mA
-
1.45
1.95
V
VDRAIN = 48V, VPWRGD = 80V
-
0.05
10
µA
VDRAIN = 5V, IOUT = 1mA
-
0.80
1.0
V
(VDRAIN - VEE) < VPG
3.5
6.2
9.0
kΩ
VDRAIN = 48V
ISL6140 (PWRGD PIN: L VERSION)
PWRGD Output Low Voltage
VOL
Output Leakage
IOH
ISL6150 (PWRGD PIN: H VERSION)
PWRGD Output Low Voltage
(PWRGD-DRAIN)
VOL
PWRGD Output Impedance
ROUT
AC TIMING
OV High to GATE Low
tPHLOV
(Figures 1, 3A)
0.6
1.6
3.0
µs
OV Low to GATE High
tPLHOV
(Figures 1, 3A)
1.0
7.8
12.0
µs
UV Low to GATE Low
tPHLUV
(Figures 1, 3B)
0.6
1.3
3.0
µs
UV High to GATE High
tPLHUV
(Figures 1, 3B)
1.0
8.4
12.0
µs
2
3
4
µs
SENSE High to GATE Low
tPHLSENSE (Figures 1, 2)
ISL6140 (L VERSION)
DRAIN Low to PWRGD Low
tPHLPG
(Figures 1, 4A)
0.1
0.9
2.0
µs
DRAIN High to PWRGD High
tPLHPG
(Figures 1, 4A)
0.1
0.7
2.0
µs
DRAIN Low to (PWRGD-DRAIN)
High
tPHLPG
(Figures 1, 4B)
6
0.1
0.9
2.0
µs
DRAIN High to (PWRGD-DRAIN)
Low
tPLHPG
(Figures 1, 4B)
6
0.1
0.8
2.0
µs
ISL6150 (H VERSION)
NOTES:
5. Typical value depends on VDD voltage; see Figure 13, “VGATE vs VDD” (<20V).
6. PWRGD is referenced to DRAIN; VPWRGD-VDRAIN = 0V.
7. Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established
by characterization and are not production tested.
5
FN9039.4
ISL6140, ISL6150
Test Circuit and Timing Diagrams
R = 5k
5V
+ 48V
PWRGD
VOV
OV
UV
VEE
VUV
8
1
2
3
ISL6140
ISL6150
4
7
6
5
VDD
DRAIN
tPHLSENSE
GATE
GATE
VSENSE
FIGURE 2. SENSE TO GATE TIMING
2V
2V
1.223V
1.203V
0V
13V
GATE
0V
1V
SENSE
FIGURE 1. TYPICAL TEST CIRCUIT
OV
50mV
SENSE
VDRAIN
tPHLOV
tPLHOV
1V
1V
UV
0V
13V
GATE
0V
FIGURE 3A. OV TO GATE TIMING
1.223V
tPHLUV
1.243V
tPLHUV
1V
1V
FIGURE 3B. UV TO GATE TIMING
FIGURE 3. OV AND UV TO GATE TIMING
DRAIN
1.8V
1.4V
tPHLPG
tPLHPG
PWRGD
DRAIN
1.0V
1.8V
1.4V
tPLHPG
1.0V
FIGURE 4A. DRAIN TO PWRGD TIMING (ISL6140)
PWRGD
1.0V
tPHLPG
1.0V
FIGURE 4B. DRAIN TO PWRGD TIMING (ISL6150)
FIGURE 4. DRAIN TO PWRGD/PWRGD TIMING
6
FN9039.4
ISL6140, ISL6150
ISL6140/ISL6150 Block Diagram
GND
GND
8 VDD
R4
VUVL, VOVH
(1.223V)
+
1 PWRGD (6150)
1 PWRGD (6140)
VEE
R5
+
-
VCC
VPG (1.7V)
VUVL, VOVH PWRGD/PWRGD
VCB (50mV) OUTPUT DRIVE
VCC (INTERNAL
VOLTAGE) AND
REFERENCE
GENERATOR
3 UV
-
VEE
2 OV
+
LOGIC AND
VCB (50mV) +
+
VEE
R6
4 VEE
+
-
GATE DRIVE
+
-
5 SENSE
6 GATE
VPG (1.7V)
VEE
7 DRAIN
LOAD
R3 C 2
R2
CL
C1
-48V IN
R1
Typical Values for a representative
system; which assumes:
36V to 72V supply range; 48 nominal; UV = 37V;
OV = 71V
1A of typical current draw; 2.5A overcurrent
100µF of load capacitance (CL); equivalent RL of 48Ω
(R = V/I = 48V/1A)
R1: 0.02Ω (1%)
R2: 10Ω (5%)
Q1
RL
-48VOUT
Applications: Quick Guide to
Choosing Component Values
(See Block Diagram for reference)
This section will describe the minimum components
needed for a typical application, and will show how to
select component values. (Note that “typical” values
may only be good for this application; the user may
have to select some component values to match the
system). Each block will then have more detailed
explanation of how it works, and alternatives.
R4, R5, R6 - together set the Under-Voltage (UV) and
overvoltage (OV) trip points. When the power supply
ramps up and down, these trip points (and their 20mV
nominal hysteresis) will determine when the gate is
allowed to turn on and off (the UV and OV do not affect
the PWRGD output). The input power supply is divided
down such that when each pin is equal to the trip point
(nominal is 1.223V), the comparator will switch.
R3: 18kΩ (5%)
R4: 562kΩ (1%)
R5: 9.09kΩ (1%)
R6: 10kΩ (1%)
C1: 150nF (25V)
C2: 3.3nF (100V)
Q1: IRF530 (100V, 17A, 0.11Ω)
VUV = 1.223 (R4 + R5 + R6)/(R5 + R6)
VOV = 1.223 (R4 + R5 + R6)/(R6)
The values of R4 = 562k, R5 = 9.09k, and R6 = 10k
will give trip points of UV = 37V and OV = 71V.
Q1 - is the FET that connects the input supply voltage
to the output load, when properly enabled. It needs to
7
FN9039.4
ISL6140, ISL6150
be selected based on several criteria: maximum
voltage expected on the input supply (including
transients) as well as transients on the output side;
maximum current expected; power dissipation and/or
safe-operating-area considerations (due to the quick
overcurrent latch, power dissipation is usually not a
problem compared to systems where current limiting is
used; however, worst case power is usually at a level
just below the overcurrent shutdown). Other
considerations include the gate voltage threshold which
affects the rDS(ON) (which in turn, affects the voltage
drop across the FET during normal operation), and the
maximum gate voltage allowed (the IC clamp output is
clamped to ~14V).
R1 - is the overcurrent sense resistor; if the input
current is high enough, such that the voltage drop
across R1 exceeds the SENSE comparator trip point
(50mV nominal), the GATE pin will go low, turning off
the FET, to protect the load from the excessive current.
A typical value for R1 is 0.02Ω; this sets an overcurrent
trip point of I = V/R = 0.05/0.02 = 2.5A. So, to choose
R1, the user must first determine at what level of
current it should trip. Take into account worst case
variations for the trip point (50mV ±10mV = ±20%),
and the R1 resistance (typically 1% or 5%). Note that
under normal conditions, there will be a voltage drop
across the resistor (V = IR), so the higher the resistor
value, the bigger the voltage drop. Also note that the
overcurrent should be set above the inrush current
level (plus the load current); otherwise, it will latch off
during that time (the alternative is to lower the in-rush
current further). One rule of thumb is to set the
overcurrent 2-3 times higher than the normal current
(see Equation 1).
R 1 = V ⁄ I OC = 0.05V/I OC ( typical = 0.02Ω )
(EQ. 1)
CL - is the sum of all load capacitances, including the
load’s input capacitance itself. Its value is usually
determined by the needs of the load circuitry, and not
the hot plug (although there can be interaction). For
example, if the load is a regulator, then the capacitance
may be chosen based on the input requirements of
that circuit (holding regulation under current spikes or
loading, filtering noise, etc.) The value chosen will then
affect how the inrush current is controlled. Note that in
the case of a regulator, there may be capacitors on the
output of that circuit as well; these need to be added
into the capacitance calculation during inrush (unless
the regulator is delayed from operation by the PWRGD
signal, for example).
RL - is the equivalent resistive value of the load; it
determines the normal operation current delivered
through the FET. It also affects some dynamic
conditions (such as the discharge time of the load
capacitors during a power-down). A typical value might
be 48Ω (I = V/R = 48/48 = 1A).
R2, C1, R3, C2 - are related to the gate driver, as it
controls the inrush current.
8
R2 prevents high frequency oscillations; 10Ω is a
typical value. R2 = 10Ω.
R3 and C2 act as a feedback network to control the
inrush current. I inrush = (Igate*CL)/C2, where CL is
the load capacitance (including module input
capacitance), and Igate is the gate pin charging
current, nominally 45µA. So choose a value of
acceptable inrush for the system, and then solve for
C2. So I = 45µA*(CL/C2). Or C2 = (45µA*CL)/I.
C1 and R3 prevent Q1 from turning on momentarily
when power is first applied. Without them, C2 would
pull the gate of Q1 up to a voltage roughly equal to
VEE*C2/CGS(Q1) (where CGS is the FET gate-source
capacitance) before the ISL6140 could power up and
actively pull the gate low. Place C1 in parallel with the
gate capacitance of Q1; isolate them from C2 by R3.
C1 = (VINMAX - VTH)/VTH*(C2+CGD) where VTH is
the FET’s minimum gate threshold, Vinmax is the
maximum operating input voltage, and Cgd is the FET
gate-drain capacitance.
R3 = (VINMAX + ΔVGATE)/5mA its value is not
critical; a typical value is 18kΩ.
Applications: Inrush Current
The primary function of the ISL6140 hot plug controller
is to control the inrush current. When a board is
plugged into a live backplane, the input capacitors of
the board’s power supply circuit can produce large
current transients as they charge up. This can cause
glitches on the system power supply (which can affect
other boards!), as well as possibly cause some
permanent damage to the power supply.
The key to allowing boards to be inserted into a live
backplane then is to turn on the power to the board in
a controlled manner, usually by limiting the current
allowed to flow through a FET switch, until the input
capacitors are fully charged. At that point, the FET is
fully on, for the smallest voltage drop across it.
In addition to controlling the in-rush current, the
ISL6140 also protects the board against overcurrent,
overvoltage, undervoltage, and can signal when the
output voltage is within its expected range (PWRGD).
Note that although this IC was designed for -48V
systems, it can also be used as a low-side switch for
positive 48V systems; the operation and components
are usually similar. One possible difference is the kind
of level shifting that may be needed to interface logic
signals to the UV input (to reset the latch) or PWRGD
output. For example, many of the IC functions are
referenced to the IC substrate, connected to the VEE
pin. But this pin may be considered -48V or GND,
depending upon the polarity of the system. And input
or output logic (running at 5V or 3.3V or even lower)
might be externally referenced to either VDD or VEE of
the IC, instead of GND.
FN9039.4
ISL6140, ISL6150
Applications: Overcurrent
CORRECT
V(t0) = 20mV (V = IR = 1A*20mΩ)
Vi = 100mV (V = IR = 5A*20mΩ)
If R7 = 100Ω, then C3 is around 1µF.
INCORRECT
Note that the FET must be rated to handle the higher
current for the longer time, since the IC is not doing
current limiting; the RC is just delaying the overcurrent
shutdown.
TO SENSE
AND VEE
Applications: OV and UV
CURRENT
SENSE RESISTOR
FIGURE 5. SENSE RESISTOR
Physical layout of R1 SENSE resistor is critical to avoid
the possibility of false overcurrent occurrences. Since it
is in the main input-to-output path, the traces should
be wide enough to support both the normal current,
and up to the overcurrent trip point. Ideally trace
routing between the R1 resistor and the ISL6140 and
ISL6150 (pin 4 (VEE) and pin 5 (SENSE) is direct and
as short as possible with zero current in the sense lines
(see Figure 5).
There is a short filter (3µs nominal) on the
comparator; current spikes shorter than this will be
ignored. Any longer pulse will shut down the output,
requiring the user to either power-down the system
(below the UV voltage), or pull the UV pin below its
trip point (usually with an external transistor).
If current pulses longer than the 3µs are expected, and
need to be filtered, then an additional resistor and
capacitor can be added. As shown in Figure 29, R7 and
C3 act as a low-pass filter such that the voltage on the
SENSE pin won’t rise as fast, effectively delaying the
shut-down. Since the ISL6140/ISL6150 has essentially
zero current on the SENSE pin, there is no voltage drop
or error associated with the extra resistor. R7 is
recommended to be small, 100Ω is a good value.
The delay time is approximated by the added RC time
constant, modified by a factor relative to the trip point
(see Equation 2).
t = – R*C*In [1 - (V(t) - V(t 0 ) ) ⁄ ( V i – V ( t 0 ) ) ]
(EQ. 2)
where V(t) is the trip voltage (nominally 50mV); V(t0)
is the nominal voltage drop across the sense resistor
before the overcurrent condition; Vi is the voltage drop
across the sense resistor while the overcurrent is
applied.
For example: a system has a normal 1A current load,
and a 20mΩ sense resistor, for a 2.5A overcurrent. It
needs to filter out a 50µs current pulse at 5A.
Therefore:
V(t) = 50mV (from spec)
9
The UV and OV input pins are high impedance, so the
value of the external resistor divider is not critical with
respect to input current. Therefore, the next
consideration is total current; the resistors will always
draw current, equal to the supply voltage divided by
the total of R4 + R5 + R6; so the values should be
chosen high enough to get an acceptable current.
However, to the extent that the noise on the power
supply can be transmitted to the pins, the resistor
values might be chosen to be lower. A filter capacitor
from UV to VEE or OV to UV is a possibility, if certain
transients need to be filtered. (Note that even some
transients which will momentarily shut off the gate
might recover fast enough such that the gate or the
output current does not even see the interruption).
Finally, take into account whether the resistor values
are readily available, or need to be custom ordered.
Tolerances of 1% are recommended for accuracy. Note
that for a typical 48V system (with a 36V to 72V
range), the 36V or 72V is being divided down to
1.223V, a significant scaling factor. For UV, the ratio is
roughly 30x; every 3mV change on the UV pin
represents roughly 0.1V change of power supply
voltage. Conversely, an error of 3mV (due to the
resistors, for example) results in an error of 0.1V for
the supply trip point. The OV ratio is around 60. So the
accuracy of the resistors comes into play.
The hysteresis of the comparators (20mV nominal)
is also multiplied by the scale factor of 30 for the UV
pin (30 * 20mV = 0.6V of hysteresis at the power
supply) and 60 for the OV pin (60*20mV = 1.2V of
hysteresis at the power supply).
With the three resistors, the UV equation is based on
the simple resistor divider:
1.223 = VUV*(R5 + R6)/(R4 + R5 + R6) or
VUV = 1.223 (R4 + R5 + R6)/(R5 + R6)
Similarly, for OV:
1.223 = VOV*(R6)/(R4 + R5 + R6) or
VOV = 1.223 (R4 + R5 + R6)/(R6)
Note that there are two equations, but 3 unknowns.
Because of the scale factor, R4 has to be much bigger
than the other two; chose its value first, to set the
current (for example, 50V/500kΩ draws 100µA), and
then the other two will be in the 10kΩ range. Solve the
two equations for two unknowns. Note that some
iteration may be necessary to select values that meet
FN9039.4
ISL6140, ISL6150
The PWRGD/PWRGD outputs are typically used to
directly enable a power module, such as a DC/DC
converter. The PWRGD (ISL6140) is used for modules
with active low enable (L version); PWRGD (ISL6150)
for those with active high enable (H version). The
modules usually have a pull-up device built-in, as well
as an internal clamp. If not, an external pull-up resistor
may be needed, since the output is open drain. If the
pin is not used, it can be left open.
For both versions, the PG comparator compares the
DRAIN pin to VEE (connected to the source of the FET);
if the voltage drop exceeds VPG (1.7V nominal), that
implies the drop across the FET is too high, and the
PWRGD pin should go in-active (power-NO-GOOD).
10
+
-
+
CL
Q2
VEE
DRAIN
ACTIVE
LOW
ENABLE
MODULE
VIN- VOUT-
FIGURE 6. ACTIVE LOW ENABLE MODULE
When the DRAIN is too high, the Q2 DMOS will shut off
(high impedance), and the pin will be pulled high by
the external module (or an optional pull-up resistor or
equivalent), disabling the module. If a pull-up resistor
is used, it can be connected to any supply voltage that
doesn’t exceed the IC pin maximum ratings on the
high end, but is high enough to give acceptable logic
levels to whatever signal it is driving. An external
clamp may be used to limit the range.
VDD
(SECTION OF) ISL6140
(L VERSION)
VPG
(1.7V)
+
-
PWRGD
R12
OPTO
PWRGD
Q2
VEE
DRAIN
FIGURE 7. ACTIVE LOW ENABLE OPTO-ISOLATOR
The PWRGD can also drive an opto-coupler (such as a
4N25), as shown in Figure 7 or LED (Figure 8). In both
cases, they are on (active) when power is good.
Resistors R12 or R13 are chosen, based on the supply
voltage, and the amount of current needed by the
loads.
VDD
(SECTION OF) ISL6140
(L VERSION)
VPG
(1.7V)
+
-
PWRGD
R13
LED (GREEN)
+
Applications: PWRGD/PWRGD
ON/OFF
VPG (1.7V)
Q2
-
Note that this is not a requirement; if the IC gets
powered at the same time as the rest of the board, it
should be able to properly control the inrush current.
But if finer control is needed, there are many variables
involved to consider: the number of pins in the
connector; the lengths of the pins; the amount of
mechanical play in the pin-to-connector interface; the
amount of extra time versus the shorter pin length; the
amount of input capacitance versus the ability of the
power supply to charge it; the manufacturing cost
adder (if any) of different length pins; etc.
PWRGD
-
The advantage of doing this: the VDD and VEE pin
connections are made first. The IC is powered up, but
since the top of the resistor divider is still open, both
the UV and OV pins are pulled low to VEE, which will
keep the gate off. This allows the IC time to get
initialized, and also allows the power supply to charge
up any input capacitance. By the time the resistor
divider makes contact, the power supply voltage on the
card is presumably stabilized, and the IC ready to
respond; when the UV pin reaches the proper voltage,
the IC will turn on the GATE of the FET, and starts the
controlled inrush current charging.
VIN+VOUT+
(SECTION OF) ISL6140
(L VERSION)
+
Note that the top of the resistor dividers is shown in
Figure 29 as GND (Short pin). In a system where cards
are plugged into a backplane (or any other case where
pins are plugged into an edge connector) the user may
want to take advantage of the order in which pins
make contact. Typically, pins on either end of the card
make contact first (although you may not know which
end is first). If you combine that with designating a pin
near the center as the short pin GND, and make it
shorter than the rest, then it should be the last pin to
make contact.
VDD
-
The three resistors (R4, R5, R6) is the recommended
approach for most cases. But if acceptable values can’t
be found, then consider 2 separate resistor dividers
(one for each pin; both from VDD to VEE). This also
allows the user to adjust or trim either trip point
independently.
ISL6140 (L version; Figure 6): Under normal
conditions (DRAIN < VPG), the Q2 DMOS will turn on,
pulling PWRGD low, enabling the module.
+
the requirement, and are also readily available
standard values.
VEE
DRAIN
FIGURE 8. ACTIVE LOW ENABLE WITH LED
FN9039.4
ISL6140, ISL6150
ISL6150 (H version; Figure 9): Under normal
conditions (DRAIN < VPG), the Q3 DMOS will be on,
shorting the bottom of the internal resistor to VEE, and
turning Q2 off. If the pull-up current from the external
module is high enough, the voltage drop across the
6.2kΩ resistor will look like a logic high (relative to
DRAIN). Note that the module is only referenced to
DRAIN, not VEE (but under normal conditions, the FET
is on, and the DRAIN and VEE are almost the same
voltage).
When the DRAIN voltage is high compared to VPG, Q3
DMOS turns off, and the resistor and Q2 clamp the
PWRGD pin to one diode drop (~0.7V) above the
DRAIN pin. This should be able to pull low against the
module pull-up current, and disable the module.
VDD
VIN+VOUT+
(SECTION OF) ISL6150
(H VERSION)
RPG
VPG
6.2k
(1.7V)
+
Q2
Q3
PWRGD
ON/OFF
CL
+
ACTIVE HIGH
ENABLE
MODULE
+
-
VEE
DRAIN
VIN- VOUT-
FIGURE 9. ACTIVE HIGH ENABLE MODULE
Applications: GATE Pin
To help protect the external FET, the output of the
GATE pin is internally clamped; up to an 80V supply, it
will not be any higher than 15V (nominal 14V). From
about 18V down to 10V, the GATE voltage will be
around 4V below the supply voltage; at 10V supply, the
minimum GATE voltage is 5.4V (worst case is at
-40°C).
Applications: Optional
Components
In addition to the typical application, and the variations
already mentioned, there are a few other possible
components that might be used in specific cases. See
Figure 29 for some possibilities.
If the input power supply exceeds the 100V absolute
maximum rating, even for a short transient, that could
cause permanent damage to the IC, as well as other
components on the board. If this cannot be
guaranteed, a voltage suppressor (such as the
SMAT70A, D1) is recommended. When placed from
VDD to VEE on the board, it will clamp the voltage.
If transients on the input power supply occur when the
supply is near either the OV or UV trip points, the GATE
could turn on or off momentarily. One possible solution
is to add a filter cap C4 to the VDD pin, through
isolation resistor R10. A large value of R10 is better for
the filtering, but be aware of the voltage drop across it.
For example, a 1kΩ resistor, with 1mA of IDD would
11
have 1V across it and dissipate 1mW. Since the UV and
OV comparators are referenced with respect to the VEE
supply, they should not be affected. But the GATE
clamp voltage could be offset by the voltage across the
extra resistor.
If there are negative transients on the DRAIN pin,
blocking diodes may help limit the amount of current
injected into the IC substrate. General purpose diodes
(such as 1N4148) may be used. Note that the ISL6140
(L version) requires one diode, while the ISL6150
(H version) requires two diodes. One consequence of
the added diodes it that the VPG voltage is offset by
each diode drop.
The switch SW1 is shown as a simple pushbutton. It
can be replaced by an active switch, such as an NPN or
NFET; the principle is the same; pull the UV node below
its trip point, and then release it (toggle low). To
connect an NFET, for example, the drain goes to UV;
the source to VEE, and the gate is the input; if it goes
high (relative to VEE), it turns the NFET on, and UV is
pulled low. Just make sure the NFET resistance is low
compared to the resistor divider, so that it has no
problem pulling down against it.
R8 is a pull-up resistor for PWRGD, if there is no other
component acting as a pull-up device. The value of R8
is determined by how much current you want when
pulled low (also affected by the VDD voltage); and you
want to pull it low enough for a good logic low level. An
LED can also be placed in series with R8, if desired. In
that case, the criteria is the LED brightness versus
current.
R7 and C3 are used to delay the overcurrent shutdown,
as described in the OV and UV section.
Applications: “Brick”
Regulators
One of the typical loads used are DC/DC regulators,
some commonly known as “brick” regulators, (partly
due to their shape, and because it can be considered a
“building block” of a system). For a given input voltage
range, there are usually whole families of different
output voltages and current ranges. There are also
various standardized sizes and pinouts, starting with
the original “full” brick, and since getting smaller
(half-bricks and quarter-bricks are now common).
Other common features may include: all components
(except some filter capacitors) are self-contained in a
molded plastic package; external pins for connections;
and often an ENABLE input pin to turn it on or off. A hot
plug IC, such as the ISL6140, is often used to gate
power to a brick, as well as turn it on.
Many bricks have both logic polarities available (Enable
Hi or Lo input); select the ISL6140 (L version) and
ISL6150 (H version) to match. There is little difference
between them, although the L version output is usually
simpler to interface.
FN9039.4
ISL6140, ISL6150
The Enable input often has a pull-up resistor or current
source, or equivalent built in; care must be taken in
the ISL6150 (H version) output that the given current
will create a high enough input voltage (remember that
current through the RPG 6.2kΩ resistor generates the
high voltage level; (see Figure 9).
The input capacitance of the brick is chosen to match
its system requirements, such as filtering noise, and
maintaining regulation under varying loads. Note that
this input capacitance appears as the load capacitance
of the ISL6140/ISL6150.
The brick’s output capacitance is also determined by
the system, including load regulation considerations.
However, it can affect the ISL6140 and ISL6150,
depending upon how it is enabled. For example, if the
PWRGD signal is not used to enable the brick, the
following could occur. Sometime during the inrush
current time, as the main power supply starts charging
the brick input capacitors, the brick itself will start
working, and start charging its output capacitors and
load; that current has to be added to the inrush
current. In some cases, the sum could exceed the
overcurrent shutdown, which would shut down the
whole system! Therefore, whenever practical, it is
advantageous to use the PWRGD output to keep the
brick off at least until the input caps are charged up,
and then start-up the brick to charge its output caps.
Typical brick regulators include models such as Lucent
JW050A1-E or Vicor VI-J30-CY. These are nominal -48V
input, and 5V outputs, with some isolation between the
input and output.
12
Applications: Layout
Considerations
For the minimum application, there are only 6
resistors, 2 capacitors, one IC and one FET. A sample
layout is shown in Figure 30. It assumes the IC is
8-SOIC; the FET is in a D2PAK (or similar SMD-220
package).
Although GND planes are common with multi-level
PCBs, for a -48V system, the -48V rails (both input and
output) act more like a GND than the top 0V rail
(mainly because the IC signals are mostly referenced
to the lower rail). So if separate planes for each voltage
are not an option, consider prioritizing the bottom rails
first.
Note that with the placement shown, most of the signal
lines are short, and there should not be much
interaction between them.
Although decoupling capacitors across the IC supply
pins are often recommended in general, this
application may not need one, nor even tolerate one.
For one thing, a decoupling cap would add to (or be
swamped out by) any other input capacitance; it also
needs to be charged up when power is applied. But
more importantly, there are no high speed (or any)
input signals to the IC that need to be conditioned. If
still desired, consider the isolation resistor R10, as
shown in Figure 29.
FN9039.4
ISL6140, ISL6150
1.0
15
0.8
12
VGATE (V)
IDD CURRENT (mA)
Typical Performance Curves
0.6
0.4
6
3
0.2
0.0
9
0
20
40
60
80
0
100
0
20
40
1.0
15
0.8
12
0.6
0.4
9
6
2
4
6
8
10
12
14
16
0
18 20
0
2
4
6
8
VDD VOLTAGE (V)
12
14
16
18 20
FIGURE 13. VGATE vs VDD (<20V)
0.95
14.5
0.93
14.3
GATE VOLTAGE (V)
CURRENT (mA)
10
VDD (V)
FIGURE 12. IDD vs VDD (<20V)
0.91
0.89
14.1
13.9
13.7
0.87
0.85
-40
100
3
0.2
0
80
FIGURE 11. VGATE vs VDD
VGATE (V)
IDD CURRENT (mA)
FIGURE 10. IDD vs VDD
0.0
60
VDD (V)
VDD VOLTAGE (V)
10
60
TEMPERATURE (°C)
FIGURE 14. IDD CURRENT (AT VDD = 80V)
13
110
13.5
-40
10
60
110
TEMPERATURE (°C)
FIGURE 15. GATE VOLTAGE (AT VDD = 80V)
FN9039.4
ISL6140, ISL6150
14.5
7.5
14.0
7.0
GATE VOLTAGE (V)
GATE VOLTAGE (V)
Typical Performance Curves (Continued)
13.5
13.0
12.5
6.5
6.0
5.5
12.0
-40
10
60
5.0
110
-40
TEMPERATURE (°C)
60
110
TEMPERATURE (°C)
FIGURE 16. GATE VOLTAGE (AT VDD = 17V)
FIGURE 17. GATE VOLTAGE (AT VDD = 10V)
-0.050
40
CURRENT (mA)
45
CURRENT (mA)
-0.048
-0.052
-0.054
-0.056
35
30
25
-0.058
-40
10
60
20
-40
110
TEMPERATURE (°C)
60
110
FIGURE 19. GATE PULL-DOWN CURRENT
7.5
0.32
7.0
IMPEDANCE (kΩ)
0.34
0.30
0.28
0.26
0.24
-40
10
TEMPERATURE (°C)
FIGURE 18. GATE PULL-UP CURRENT
VOLTAGE (V)
10
6.5
6.0
5.5
10
60
TEMPERATURE (°C)
FIGURE 20. PWRGD (ISL6140) VOL (AT 1mA)
VOLTAGE
14
110
5.0
-40
10
60
110
TEMPERATURE (°C)
FIGURE 21. PWRGD (ISL6150) IMPEDANCE (kΩ)
FN9039.4
ISL6140, ISL6150
1.90
1.35
1.85
1.30
TRIP VOLTAGE (V)
TRIP VOLTAGE (V)
Typical Performance Curves (Continued)
1.80
1.75
1.70
1.65
-40
1.25
1.20
1.15
10
60
1.10
-40
110
TEMPERATURE (°C)
110
FIGURE 23. DRAIN/PG DOWN TRIP VOLTAGE
0.55
0.055
0.53
0.053
TRIP VOLTAGE (V)
VOLTAGE (V)
60
TEMPERATURE (°C)
FIGURE 22. DRAIN/PG UP TRIP VOLTAGE
0.51
0.49
0.47
0.45
-40
10
0.051
0.049
0.047
10
60
110
TEMPERATURE (°C)
FIGURE 24. DRAIN/PG HYSTERESIS VOLTAGE
15
0.045
-40
10
60
110
TEMPERATURE (°C)
FIGURE 25. SENSE TRIP VOLTAGE
FN9039.4
ISL6140, ISL6150
Inrush Current
In the example in Figure 26, the supply voltage is 48V
and the load resistor (RL) is 620Ω, for around 80mA.
The load capacitance is 100μF (100V). The Sense
Resistor (R1) is 0.02Ω (trip point at 2.5A; well above
the inrush current here).
Note that the load current starts at 0 (FET off); reaches
a peak of ~850mA as the GATE voltage ramps and
turns on the FET slowly, and then settles out at 80mA,
once the CL is fully charged to the 48V. The width of
the inrush current pulse is 8ms wide. For comparison,
with the same conditions, but without the
gate-controlled FET, the current was over 20A, during a
130µs pulse.
VDD = 60V
OV = 54V
VDD
UV = 38V
OFF
13V
ON
GATE
OFF
FIGURE 27. POWER SUPPLY RAMP
Load
LOAD
CURRENT
Current
Overcurrent at 2.3A
In Figure 28, an Electronic Load Generator was used to
ramp the load current; no load resistor or capacitor
was connected. The sense Resistor R1 is 0.02Ω; that
should make the nominal overcurrent trip point 2.5A.
GATE
GATE
48V
48V
PWRGD-BAR
PWRGD-bar
FIGURE 26. INRUSH CURRENT
Power Supply Ramp
Figure 27 shows the power supply voltage (to the VDD
pin, with respect to GND at the VEE pin) ramping up. In
this case, the values chosen were R4 = 562k;
R5 = 5.9k; R6 = 13.3k; that sets the UV trip point
around 38V, and the OV trip point to 54V. Note that the
GATE starts at 0V, and stays there until the UV trip
point (38V) is exceeded; then it ramps (slowly, based
on the external components chosen) up to around 13V,
where it is clamped; it stays there until the power
supply exceeds the OV trip point at 54V (the GATE
shut-off is much faster than the turn-on). The total
time scale is 2 seconds; the VDD ramp speed was
simply based on the inherent characteristic of the
particular power supply used.
The GATE is high (clamped to around 13V), keeping
the FET on, as the current starts to ramp up from zero;
the GATE starts to go low (to shut off the FET) when
the load current hits 2.3A. Note that it takes only 44µs
for the GATE to shut off the FET (when the load current
equals zero).
Keep in mind that the tolerance of the sense resistor
(1% here) and the IC overcurrent trip voltage (VCB)
affect the accuracy of the trip point; that’s why the trip
point doesn’t necessarily equal the 2.5A design target.
Load Current
LOAD CURRENT
2.3A
2.3
A
R1 =
R1
=0.02Ω
0.02Ωoh
48V
NO CAP
48V
No cap
GATE
GATE
FIGURE 28. OVERCURRENT AT 2.3A
16
FN9039.4
ISL6140, ISL6150
GND
GND
GND
(SHORT PIN)
R10*
R4
R11
R8*
VDD
UV
ISL6140 (L)
SW1
G
OV
NFET*
(INSTEAD
OF SW1)
C4*
R5
D1*
R12
VEE
SENSE
GATE
DRAIN
R6*
D2*
C3*
C1
R1
CL*
D3*
R3
R7*
-VIN
PWRGD
R2
C2
Q1
-VOUT
FIGURE 29. ISL6140/50 OPTIONAL COMPONENTS (SHOWN WITH *)
Optional Components
D1 is a voltage suppressor; SMAT70A or equivalent.
D2 and D3 are DRAIN diodes; the ISL6150 (H version)
uses both D2 and D3; the ISL6140 (L version) uses
just D2. If neither is used, short the path of either, to
connect the DRAIN pin to C2 and Q1. The 1N4148 is a
typical diode.
SW1 is a push-button switch, that can manually reset
the fault latch after an overcurrent shutdown. It can
also be replaced by a transistor switch.
R10 and C4 are used to filter the VDD voltage, such
that small transients on the input supply do not trigger
UV or OV.
R7 and C3 are used to delay the overcurrent shutdown.
R7 should be shorted, if not used. See the overcurrent
section for more details.
the new thresholds with a rising and falling input are
shown in Equation 3 and 4:
R5 • R6 + R4 • R6 + R4 • R5
Vuv ( RISING ) = VUVH • ⎛ ------------------------------------------------------------------------------⎞
⎝
⎠
R5 • R6
(EQ. 3)
R5 • R6 + R4 • R6 + R4 • R5
R4
Vuv ( falling ) = VUVL • ⎛ ------------------------------------------------------------------------------⎞ – Vgate • ⎛ --------⎞
⎝
⎠
⎝ R6⎠
R5 • R6
(EQ. 4)
Since R6 is connected directly to the GATE output, it
will reduce the available gate current, which will reduce
the dv/dt across the MOSFET and hence the inrush
current. The value of R6 should be kept as high as
possible (greater than 500k recommended) so that it
does not drag down the GATE voltage below the value
required to ensure the MOSFET is fully enhanced.
Figure 30 shows a sample component placement and
routing for the typical application shown in Figure 31.
R8 is a pull-up resistor for PWRGD, if there is no other
component acting as a pull-up device. An LED can also
be placed in series with R8, if desired (see Figure 8).
CL is any extra output Load capacitance, which can
also be considered input capacitance for the external
module.
R6 is used to add more hysteresis to the UV threshold,
which already has a built-in 20mV hysteresis. With R6,
17
FN9039.4
ISL6140, ISL6150
GND
GND
C2
VDD 8
1 PG
R6
2 OV
R5
3 UV
G6
4 VEE
S5
G
R3
D7
U1
DRAIN
FET
R2
R4
C1
S
R1
-48VIN
-48VOUT
FIGURE 30. SAMPLE LAYOUT (NOT TO SCALE)
NOTES:
.
GND
1. Layout scale is approximate; routing lines are just for
illustration purposes; they do not necessarily conform to
normal PCB design rules. High current buses are wider,
shown with parallel lines.
GND
R4
VDD
UV
R5
2. Approximate size of the above layout is 1.6 x 0.6 inches;
almost half of the area is just the FET (D2PAK or similar
SMD-220 package).
PWRGD
ISL6140
OV
R6
VEE
SENSE
GATE
C1
R2
DRAIN
R3
(LOAD)
C2
CL
RL
-48V IN
R1
Q1
-48VOUT
FIGURE 31. TYPICAL APPLICATION
3. R1 sense resistor is size 2512; all other R’s and C’s shown
are 0805; they can all potentially use smaller footprints, if
desired.
4. The RL and CL are not shown on the layout.
5. R4 uses a via to connect to GND on the bottom of the
board; all other routing can be on top level. (It’s even
possible to eliminate the via, for an all top-level route).
6. PWRGD signal is not used here.
7. BOM (Bill Of Materials)
R1 = 0.02Ω (5%)
R2 = 10Ω (5%)
R3 = 18kΩ (5%)
R4 = 562kΩ (1%)
R5 = 9.09kΩ (1%)
R6 = 10kΩ (1%)
C1 = 150nF (25V)
C2 = 3.3nF (100V)
Q1 = IRF530 (100V, 17A, 0.11Ω)
18
FN9039.4
ISL6140, ISL6150
Small Outline Plastic Packages (SOIC)
M8.15 (JEDEC MS-012-AA ISSUE C)
N
INDEX
AREA
8 LEAD NARROW BODY SMALL OUTLINE PLASTIC PACKAGE
0.25(0.010) M
H
B M
INCHES
E
SYMBOL
-B1
2
3
L
SEATING PLANE
-A-
A
D
h x 45°
-C-
e
A1
B
0.25(0.010) M
C
0.10(0.004)
C A M
MIN
MAX
MIN
MAX
NOTES
A
0.0532
0.0688
1.35
1.75
-
A1
0.0040
0.0098
0.10
0.25
-
B
0.013
0.020
0.33
0.51
9
C
0.0075
0.0098
0.19
0.25
-
D
0.1890
0.1968
4.80
5.00
3
E
0.1497
0.1574
3.80
4.00
4
e
α
B S
0.050 BSC
1.27 BSC
-
H
0.2284
0.2440
5.80
6.20
-
h
0.0099
0.0196
0.25
0.50
5
L
0.016
0.050
0.40
1.27
6
N
α
NOTES:
MILLIMETERS
8
0°
8
8°
0°
1. Symbols are defined in the “MO Series Symbol List” in Section 2.2 of
Publication Number 95.
7
8°
Rev. 1 6/05
2. Dimensioning and tolerancing per ANSI Y14.5M-1982.
3. Dimension “D” does not include mold flash, protrusions or gate burrs.
Mold flash, protrusion and gate burrs shall not exceed 0.15mm (0.006
inch) per side.
4. Dimension “E” does not include interlead flash or protrusions. Interlead flash and protrusions shall not exceed 0.25mm (0.010 inch) per
side.
5. The chamfer on the body is optional. If it is not present, a visual index
feature must be located within the crosshatched area.
6. “L” is the length of terminal for soldering to a substrate.
7. “N” is the number of terminal positions.
8. Terminal numbers are shown for reference only.
9. The lead width “B”, as measured 0.36mm (0.014 inch) or greater
above the seating plane, shall not exceed a maximum value of
0.61mm (0.024 inch).
10. Controlling dimension: MILLIMETER. Converted inch dimensions
are not necessarily exact.
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19
FN9039.4