AD AD811AR-20 High performance video op amp Datasheet

a
High Performance
Video Op Amp
AD811
APPLICATIONS
Video Crosspoint Switchers, Multimedia Broadcast
Systems
HDTV Compatible Systems
Video Line Drivers, Distribution Amplifiers
ADC/DAC Buffers
DC Restoration Circuits
Medical—Ultrasound, PET, Gamma and Counter
Applications
PRODUCT DESCRIPTION
The AD811 is a wideband current-feedback operational amplifier, optimized for broadcast quality video systems. The –3 dB
bandwidth of 120 MHz at a gain of +2 and differential gain and
phase of 0.01% and 0.01° (RL = 150 Ω) make the AD811 an
excellent choice for all video systems. The AD811 is designed to
meet a stringent 0.1 dB gain flatness specification to a bandwidth of 35 MHz (G = +2) in addition to the low differential
gain and phase errors. This performance is achieved whether
driving one or two back terminated 75 Ω cables, with a low
power supply current of 16.5 mA. Furthermore, the AD811 is
specified over a power supply range of ± 4.5 V to ± 18 V.
NC 1
8 NC
–IN 2
7 +V S
+IN 3
–VS 4
6 OUTPUT
0.08
0.07
0.06
0.16
0.14
0.12
0.10
0.05
PHASE
0.04
0.08
0.06
0.03
GAIN
0.02
0.04
NC 4
NC 5
NC
NC
NC
18 NC
AD811
–IN 6
NC 7
+IN 8
5 NC
17 NC
16 +V S
15 NC
14 OUTPUT
9 10 11 12 13
NC = NO CONNECT
NC
NC
NC
NC
AD811
NC = NO CONNECT
16-Lead SOIC (R-16) Package 20-Lead SOIC (R-20) Package
NC 1
16 NC
NC
1
20 NC
NC 2
15 NC
NC 2
19 NC
14 +V S
NC 3
18 NC
13 NC
–IN 4
17 +V S
12 OUTPUT
NC
6
11 NC
+IN 6
–VS 7
10 NC
NC 7
14 NC
9 NC
–VS 8
13 NC
–IN
3
NC 4
+IN
NC
5
AD811
NC 8
16 NC
5
NC 9
NC = NO CONNECT
15 OUTPUT
AD811
NC 10
12 NC
11 NC
NC = NO CONNECT
The AD811 is also excellent for pulsed applications where transient response is critical. It can achieve a maximum slew rate of
greater than 2500 V/µs with a settling time of less than 25 ns to
0.1% on a 2 volt step and 65 ns to 0.01% on a 10 volt step.
The AD811 is ideal as an ADC or DAC buffer in data acquisition systems due to its low distortion up to 10 MHz and its wide
unity gain bandwidth. Because the AD811 is a current feedback
amplifier, this bandwidth can be maintained over a wide range
of gains. The AD811 also offers low voltage and current noise of
1.9 nV/√Hz and 20 pA/√Hz, respectively, and excellent dc accuracy for wide dynamic range applications.
12
G = +2
RL = 150V
RG = RFB
9
VS = 615V
6
GAIN – dB
0.18
RF = 649V
FC = 3.58MHz
100 IRE
MODULATED RAMP
RL = 150V
DIFFERENTIAL PHASE – Degrees
0.09
NC
NC
3 2 1 20 19
0.20
0.10
DIFFERENTIAL GAIN – %
CONNECTION DIAGRAMS
20-Lead LCC (E-20A) Package
8-Lead Plastic (N-8)
Cerdip (Q-8)
SOIC (SO-8) Packages
–VS
FEATURES
High Speed
140 MHz Bandwidth (3 dB, G = +1)
120 MHz Bandwidth (3 dB, G = +2)
35 MHz Bandwidth (0.1 dB, G = +2)
2500 V/␮s Slew Rate
25 ns Settling Time to 0.1% (For a 2 V Step)
65 ns Settling Time to 0.01% (For a 10 V Step)
Excellent Video Performance (RL =150 ⍀)
0.01% Differential Gain, 0.01ⴗ Differential Phase
Voltage Noise of 1.9 nV√Hz
Low Distortion: THD = –74 dB @ 10 MHz
Excellent DC Precision
3 mV max Input Offset Voltage
Flexible Operation
Specified for ⴞ5 V and ⴞ15 V Operation
ⴞ2.3 V Output Swing into a 75 ⍀ Load (VS = ⴞ5 V)
3
VS = 65V
0
–3
0.02
0.01
–6
5
6
7
8
9
10
11
12
13
14
15
SUPPLY VOLTAGE – 6Volts
1M
10M
100M
FREQUENCY – Hz
REV. D
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700
World Wide Web Site: http://www.analog.com
Fax: 781/326-8703
© Analog Devices, Inc., 1999
AD811–SPECIFICATIONS (@ T = +25ⴗC and V = ⴞ15 V dc, R
A
S
LOAD
AD811J/A1
Typ
Max
Conditions
VS
DYNAMIC PERFORMANCE
Small Signal Bandwidth (No Peaking)
–3 dB
G = +1
G = +2
G = +2
G = +10
0.1 dB Flat
G = +2
RFB = 562 Ω
RFB = 649 Ω
RFB = 562 Ω
RFB = 511 Ω
± 15 V
± 15 V
±5 V
± 15 V
140
120
80
100
140
120
80
100
MHz
MHz
MHz
MHz
RFB = 562 Ω
RFB = 649 Ω
VOUT = 20 V p-p
VOUT = 4 V p-p
VOUT = 20 V p-p
10 V Step, AV = –1
±5 V
± 15 V
± 15 V
±5 V
± 15 V
± 15 V
2 V Step, AV = –1
RFB = 649, AV = +2
f = 3.58 MHz
f = 3.58 MHz
VOUT = 2 V p-p, AV = +2
@ fC = 10 MHz
±5 V
± 15 V
± 15 V
± 15 V
± 15 V
±5 V
± 15 V
25
35
40
400
2500
50
65
25
3.5
0.01
0.01
–74
36
43
25
35
40
400
2500
50
65
25
3.5
0.01
0.01
–74
36
43
MHz
MHz
MHz
V/µs
V/µs
ns
ns
ns
ns
%
Degree
dBc
dBm
dBm
± 5 V, ± 15 V
0.5
Full Power Bandwidth
Slew Rate
Settling Time to 0.1%
Settling Time to 0.01%
Settling Time to 0.1%
Rise Time, Fall Time
Differential Gain
Differential Phase
THD @ fC = 10 MHz
Third Order Intercept4
INPUT OFFSET VOLTAGE
TMIN to TMAX
Offset Voltage Drift
3
5
0.5
5
INPUT BIAS CURRENT
–Input
TMIN to TMAX
+Input
± 5 V, ± 15 V
2
± 5 V, ± 15 V
2
TMIN to TMAX
TRANSRESISTANCE
Min
AD811S2
Typ
Max
Model
3
Min
= 150 Ω unless otherwise noted)
TMIN to TMAX
VOUT = ± 10 V
RL = ∞
RL = 200 Ω
VOUT = ± 2.5 V
RL = 150 Ω
3
5
mV
mV
µV/°C
5
30
10
25
µA
µA
µA
µA
5
5
15
10
20
2
2
Units
± 15 V
± 15 V
0.75
0.5
1.5
0.75
0.75
0.5
±5 V
0.25
0.4
0.125 0.4
MΩ
±5 V
± 15 V
56
60
60
66
1
50
56
3
70
0.3
0.4
2
2
1.5
0.75
MΩ
MΩ
COMMON-MODE REJECTION
VOS (vs. Common Mode)
TMIN to TMAX
TMIN to TMAX
Input Current (vs. Common Mode)
VCM = ± 2.5
VCM = ± 10 V
TMIN to TMAX
POWER SUPPLY REJECTION
VOS
+Input Current
–Input Current
VS = ± 4.5 V to ± 18 V
TMIN to TMAX
TMIN to TMAX
TMIN to TMAX
INPUT VOLTAGE NOISE
f = 1 kHz
1.9
1.9
nV/√Hz
INPUT CURRENT NOISE
f = 1 kHz
20
20
pA/√Hz
± 2.9
± 12
100
150
9
± 2.9
± 12
100
150
9
V
V
mA
mA
Ω
1.5
14
7.5
±3
± 13
1.5
14
7.5
±3
± 13
MΩ
Ω
pF
V
V
OUTPUT CHARACTERISTICS
Voltage Swing, Useful Operating Range5
Output Current
Short-Circuit Current
Output Resistance
INPUT CHARACTERISTICS
+Input Resistance
–Input Resistance
Input Capacitance
Common-Mode Voltage Range
±5 V
± 15 V
TJ = +25°C
(Open Loop @ 5 MHz)
+Input
POWER SUPPLY
Operating Range
Quiescent Current
TRANSISTOR COUNT
60
±5 V
± 15 V
±5 V
± 15 V
# of Transistors
± 4.5
14.5
16.5
40
60
± 18
16.0
18.0
60
66
1
3
dB
dB
µA/V
70
0.3
0.4
2
2
dB
µA/V
µA/V
± 4.5
14.5
16.5
± 18
16.0
18.0
V
mA
mA
40
NOTES
1
The AD811JR is specified with ± 5 V power supplies only, with operation up to ± 12 volts.
2
See Analog Devices’ military data sheet for 883B tested specifications.
3
FPBW = slew rate/(2 π VPEAK).
4
Output power level, tested at a closed loop gain of two.
5
Useful operating range is defined as the output voltage at which linearity begins to degrade.
Specifications subject to change without notice.
–2–
REV. D
AD811
ABSOLUTE MAXIMUM RATINGS 1
MAXIMUM POWER DISSIPATION
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ± 18 V
AD811JR Grade Only . . . . . . . . . . . . . . . . . . . . . . . . . ± 12 V
Internal Power Dissipation2 . . . . . . . . Observe Derating Curves
Output Short Circuit Duration . . . . . Observe Derating Curves
Common-Mode Input Voltage . . . . . . . . . . . . . . . . . . . . . ± VS
Differential Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . . ± 6 V
Storage Temperature Range (Q, E) . . . . . . . . –65°C to +150°C
Storage Temperature Range (N, R) . . . . . . . . –65°C to +125°C
Operating Temperature Range
AD811J . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0°C to +70°C
AD811A . . . . . . . . . . . . . . . . . . . . . . . . . . . . –40°C to +85°C
AD811S . . . . . . . . . . . . . . . . . . . . . . . . . . . –55°C to +125°C
Lead Temperature Range (Soldering 60 sec) . . . . . . . . +300°C
The maximum power that can be safely dissipated by the
AD811 is limited by the associated rise in junction temperature.
For the plastic packages, the maximum safe junction temperature is +145°C. For the cerdip and LCC packages, the maximum junction temperature is +175°C. If these maximums are
exceeded momentarily, proper circuit operation will be restored
as soon as the die temperature is reduced. Leaving the device in
the “overheated” condition for an extended period can result in
device burnout. To ensure proper operation, it is important to
observe the derating curves in Figures 17 and 18.
While the AD811 is internally short circuit protected, this may
not be sufficient to guarantee that the maximum junction temperature is not exceeded under all conditions. One important
example is when the amplifier is driving a reverse terminated
75 Ω cable and the cable’s far end is shorted to a power supply.
With power supplies of ± 12 volts (or less) at an ambient temperature of +25°C or less, if the cable is shorted to a supply rail,
then the amplifier will not be destroyed, even if this condition
persists for an extended period.
NOTES
1
Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute maximum rating
conditions for extended periods may affect device reliability.
2
8-Lead Plastic Package: θJA = 90°C/W
8-Lead Cerdip Package: θJA = 110°C/W
8-Lead SOIC Package: θJA = 155°C/W
16-Lead SOIC Package: θJA = 85°C/W
20-Lead SOIC Package: θJA = 80°C/W
20-Lead LCC Package: θJA = 70°C/W
ESD SUSCEPTIBILITY
ESD (electrostatic discharge) sensitive device. Electrostatic
charges as high as 4000 volts, which readily accumulate on the
human body and on test equipment, can discharge without
detection. Although the AD811 features proprietary ESD protection circuitry, permanent damage may still occur on these
devices if they are subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended
to avoid any performance degradation or loss of functionality.
ORDERING GUIDE
Model
AD811AN
AD811AR-16
AD811AR-20
AD811JR
AD811SQ/883B
5962-9313101MPA
AD811SE/883B
5962-9313101M2A
AD811JR-REEL
AD811JR-REEL7
AD811AR-16-REEL
AD811AR-16-REEL7
AD811AR-20-REEL
AD811ACHIPS
AD811SCHIPS
Temperature
Range
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
0°C to +70°C
–55°C to +125°C
–55°C to +125°C
–55°C to +125°C
–55°C to +125°C
0°C to +70°C
0°C to +70°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–40°C to +85°C
–55°C to +125°C
Package
Option*
N-8
R-16
R-20
SO-8
Q-8
Q-8
E-20A
E-20A
SO-8
SO-8
R-16
R-16
R-20
Die
Die
METALIZATION PHOTOGRAPH
Contact Factory for Latest Dimensions.
Dimensions Shown in Inches and (mm).
*E = Ceramic Leadless Chip Carrier; N = Plastic DIP; Q = Cerdip; SO (R) =
Small Outline IC (SOIC).
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the AD811 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.
REV. D
–3–
WARNING!
ESD SENSITIVE DEVICE
AD811–Typical Performance Characteristics
MAGNITUDE OF THE OUTPUT VOLTAGE – 6 Volts
COMMON-MODE VOLTAGE RANGE – 6Volts
20
TA = +258C
15
10
5
0
0
5
10
20
TA = +258C
15
NO LOAD
10
RL = 150V
5
0
20
15
0
5
SUPPLY VOLTAGE – 6Volts
Figure 1. Input Common-Mode Voltage Range vs. Supply
21
QUIESCENT SUPPLY CURRENT – mA
30
OUTPUT VOLTAGE – Volts p–p
20
Figure 4. Output Voltage Swing vs. Supply
35
VS = 615V
25
20
15
VS = 65V
10
5
0
10
1k
100
LOAD RESISTANCE – V
18
VS = 615V
15
12
VS = 65V
9
6
3
–60
10k
Figure 2. Output Voltage Swing vs. Resistive Load
–40
–20
0
20
60
40
80
100
JUNCTION TEMPERATURE – 8C
120
140
Figure 5. Quiescent Supply Current vs. Junction
Temperature
10
10
NONINVERTING INPUT
65 TO 615V
8
INPUT OFFSET VOLTAGE – mV
INPUT BIAS CURRENT – mA
15
10
SUPPLY VOLTAGE – 6 Volts
0
VS = 65V
INVERTING
INPUT
–10
VS = 615V
–20
6
VS = 65V
4
2
0
–2
VS = 615V
–4
–6
–8
–30
–60
–40
–20
0
20
40
60
80
JUNCTION TEMPERATURE – 8C
100
120
–10
–60
140
–40
–20
0
20
40
60
80
100
120
140
JUNCTION TEMPERATURE – 8C
Figure 3. Input Bias Current vs. Junction Temperature
Figure 6. Input Offset Voltage vs. Junction Temperature
–4–
REV. D
AD811
2.0
250
200
VS = 615V
150
VS = 65V
–40
–20
0
20
40
60
80
100
JUNCTION TEMPERATURE – 8C
120
0
–60 –40
140
–20
0
20
40
60
80
100
JUNCTION TEMPERATURE – 8C
100
10
NOISE VOLTAGE – nV/ Hz
VS = 65V
1
0.1
VS = 615V
0.01
10k
NONINVERTING CURRENT VS = 65 TO 15V
INVERTING CURRENT VS = 65 TO 15V
10
10
VOLTAGE NOISE VS = 615V
VOLTAGE NOISE VS = 65V
1
100k
1M
10M
100M
100
10
FREQUENCY – Hz
1k
FREQUENCY – Hz
200
10
VO = 1V p–p
VS= 615V
RL= 150V
GAIN = +2
OVERSHOOT
VO = 1V p–p
RL = 150V
20
GAIN = +2
2
–3dB BANDWIDTH – MHz
40
VS = 615V
OVERSHOOT – %
RISETIME – ns
160
60
6
1.0k
1.2k
1.4k
800
VALUE OF FEEDBACK RESISTOR (RFB) – V
120
8
6
BANDWIDTH
80
4
40
0
600
1
100k
10
RISE TIME
8
2
PEAKING
0
400
1.6k
600
1.0k
1.2k
1.4k
800
VALUE OF FEEDBACK RESISTOR (RFB) – V
0
1.6k
Figure 12. 3 dB Bandwidth and Peaking vs. Value of RFB
Figure 9. Rise Time and Overshoot vs. Value of
Feedback Resistor, RFB
REV. D
10k
Figure 11. Input Noise vs. Frequency
Figure 8. Closed-Loop Output Resistance vs. Frequency
4
140
100
GAIN = +2
RFB = 649V
0
400
120
Figure 10. Transresistance vs. Junction Temperature
Figure 7. Short Circuit Current vs. Junction Temperature
CLOSED-LOOP OUTPUT RESISTANCE – V
VS = 65V
RL = 150V
VOUT = 62.5V
0.5
NOISE CURRENT – pA/ Hz
50
–60
1.0
PEAKING – dB
100
1.5
TRANSRESISTANCE – MV
SHORT CIRCUIT CURRENT – mA
VS = 615V
RL = 200V
VOUT = 610V
–5–
AD811
110
25
649V
VIN
649V
VOUT
OUTPUT VOLTAGE – Volts p–p
100
90
CMRR – dB
150V
150V
80
70
VS = 615V
60
VS = 65V
50
VS = 615V
20
15
GAIN = +10
OUTPUT LEVEL FOR 3% THD
10
VS = 65V
5
40
30
1k
10k
100k
FREQUENCY – Hz
1M
0
100k
10M
Figure 13. Common-Mode Rejection vs. Frequency
100M
Figure 16. Large Signal Frequency Response
80
–50
VS = 615V
60
RF = 649V
AV = +2
HARMONIC DISTORTION – dBc
70
PSRR – dB
1M
10M
FREQUENCY – Hz
VS = 65V
50
CURVES ARE FOR WORST
CASE CONDITION WHERE
ONE SUPPLY IS VARIED
WHILE THE OTHER IS
HELD CONSTANT.
40
30
20
VOUT = 2V p–p
RL = 100V
GAIN = +2
–70
65V SUPPLIES
–90
2ND HARMONIC
615V SUPPLIES
3RD HARMONIC
–110
2ND HARMONIC
10
3RD HARMONIC
5
–130
1k
10k
100k
FREQUENCY – Hz
1M
10M
1k
Figure 14. Power Supply Rejection vs. Frequency
1M
10M
3.4
3.2
TJ MAX = +1458C
16-LEAD SOIC
TOTAL POWER DISSIPATION – Watts
TOTAL POWER DISSIPATION – Watts
100k
FREQUENCY – Hz
Figure 17. Harmonic Distortion vs. Frequency
2.5
2.0
20-LEAD SOIC
1.5
10k
8-LEAD MINI-DIP
1.0
8-LEAD SOIC
0.5
–50 –40 –30 –20 –10 0 10 20 30 40 50
AMBIENT TEMPERATURE – 8C
60 70
80
3.0
2.6
Figure 15. Maximum Power Dissipation vs. Temperature
for Plastic Packages
20-LEAD LCC
2.4
2.2
2.0
1.8
1.6
1.4
8-LEAD CERDIP
1.2
1.0
0.8
0.6
0.4
–60
90
TJ MAX = +1758C
2.8
–40
–20
0
20
40
60
80
AMBIENT TEMPERATURE – 8C
100
120
140
Figure 18. Maximum Power Dissipation vs. Temperature
for Hermetic Packages
–6–
REV. D
Typical Characteristics, Noninverting Connection–AD811
9
RFB
G = +1
RL = 150V
RG =
6
+VS
RG
AD811
VIN
RL
+
VS = 615V
RFB = 750V
3
VOUT TO
TEKTRONIX
P6201 FET
PROBE
GAIN – dB
0.1mF
HP8130
50V
PULSE
GENERATOR
0
–3
VS = 65V
RFB = 619V
–6
0.1mF
–VS
–9
–12
1M
10M
FREQUENCY – Hz
100M
Figure 22. Closed-Loop Gain vs. Frequency, Gain = +1
Figure 19. Noninverting Amplifier Connection
26
1V
10ns
G = +10
RL = 150V
23
VS = 615V
RFB = 511V
100
VIN
90
GAIN – dB
20
17
VS = 65V
R FB = 442V
14
VOUT 10
0%
11
1V
8
1M
Figure 20. Small Signal Pulse Response, Gain = +1
100mV
10ns
1V
20ns
100
VIN
90
VOUT 10
90
VOUT 10
0%
0%
1V
10V
Figure 21. Small Signal Pulse Response, Gain = +10
REV. D
100M
Figure 23. Closed-Loop Gain vs. Frequency, Gain = +10
100
VIN
10M
FREQUENCY – Hz
Figure 24. Large Signal Pulse Response, Gain = +10
–7–
AD811–Typical Characteristics, Inverting Connection
6
RFB
+VS
HP8130
PULSE
GENERATOR
VOUT TO
TEKTRONIX
P6201 FET
PROBE
0
GAIN – dB
0.1mF
RG
VIN
VS = 615V
RFB = 590V
G = –1
RL = 150V
3
AD811
RL
–3
V S = 65V
RFB = 562V
–6
–9
0.1mF
–12
1M
10M
FREQUENCY – Hz
–VS
Figure 25. Inverting Amplifier Connection
100M
Figure 28. Closed-Loop Gain vs. Frequency, Gain = –1
26
1V
10ns
G = –10
RL = 150V
23
VS = 615V
RFB = 511V
100
VIN
90
GAIN – dB
20
17
VS = 65V
RFB = 442V
14
VOUT 10
0%
11
1V
8
1M
1V
10ns
20ns
100
100
VIN
100M
Figure 29. Closed-Loop Gain vs. Frequency, Gain = –10
Figure 26. Small Signal Pulse Response, Gain = –1
100mV
10M
FREQUENCY – Hz
VIN
90
90
VOUT 10
VOUT 10
0%
0%
10V
1V
Figure 30. Large Signal Pulse Response, Gain = –10
Figure 27. Small Signal Pulse Response, Gain = –10
–8–
REV. D
AD811
Achieving the Flattest Gain Response at High Frequency
APPLICATIONS
General Design Considerations
Achieving and maintaining gain flatness of better than 0.1 dB at
frequencies above 10 MHz requires careful consideration of
several issues.
The AD811 is a current feedback amplifier optimized for use in
high performance video and data acquisition applications. Since
it uses a current feedback architecture, its closed-loop –3 dB
bandwidth is dependent on the magnitude of the feedback resistor. The desired closed-loop gain and bandwidth are obtained
by varying the feedback resistor (RFB) to tune the bandwidth,
and varying the gain resistor (RG) to get the correct gain. Table I
contains recommended resistor values for a variety of useful
closed-loop gains and supply voltages.
Choice of Feedback and Gain Resistors
Because of the above-mentioned relationship between the 3 dB
bandwidth and the feedback resistor, the fine scale gain flatness
will, to some extent, vary with feedback resistor tolerance. It is,
therefore, recommended that resistors with a 1% tolerance be
used if it is desired to maintain flatness over a wide range of
production lots. In addition, resistors of different construction
have different associated parasitic capacitance and inductance.
Metal-film resistors were used for the bulk of the characterization for this data sheet. It is possible that values other than those
indicated will be optimal for other resistor types.
Table I. –3 dB Bandwidth vs. Closed-Loop Gain and
Resistance Values
VS = ⴞ15 V
Closed-Loop
Gain
RFB
RG
–3 dB BW
(MHz)
+1
+2
+10
–1
–10
750 Ω
649 Ω
511 Ω
590 Ω
511 Ω
649 Ω
56.2 Ω
590 Ω
51.1 Ω
140
120
100
115
95
VS = ⴞ5 V
Closed-Loop
Gain
RFB
RG
–3 dB BW
(MHz)
+1
+2
+10
–1
–10
619 Ω
562 Ω
442 Ω
562 Ω
442 Ω
562 Ω
48.7 Ω
562 Ω
44.2 Ω
80
80
65
75
65
VS = ⴞ10 V
Closed-Loop
Gain
RFB
RG
–3 dB BW
(MHz)
+1
+2
+10
–1
–10
649 Ω
590 Ω
499 Ω
590 Ω
499 Ω
590 Ω
49.9 Ω
590 Ω
49.9 Ω
105
105
80
105
80
Printed Circuit Board Layout Considerations
As to be expected for a wideband amplifier, PC board parasitics
can affect the overall closed loop performance. Of concern are
stray capacitances at the output and the inverting input nodes. If
a ground plane is to be used on the same side of the board as
the signal traces, a space (3/16" is plenty) should be left around
the signal lines to minimize coupling. Additionally, signal lines
connecting the feedback and gain resistors should be short
enough so that their associated inductance does not cause
high frequency gain errors. Line lengths less than 1/4" are
recommended.
Quality of Coaxial Cable
Optimum flatness when driving a coax cable is possible only
when the driven cable is terminated at each end with a resistor
matching its characteristic impedance. If the coax was ideal,
then the resulting flatness would not be affected by the length of
the cable. While outstanding results can be achieved using inexpensive cables, it should be noted that some variation in flatness
due to varying cable lengths may be experienced.
Power Supply Bypassing
Adequate power supply bypassing can be critical when optimizing the performance of a high frequency circuit. Inductance in
the power supply leads can form resonant circuits that produce
peaking in the amplifier’s response. In addition, if large current
transients must be delivered to the load, then bypass capacitors
(typically greater than 1 µF) will be required to provide the best
settling time and lowest distortion. Although the recommended
0.1 µF power supply bypass capacitors will be sufficient in many
applications, more elaborate bypassing (such as using two paralleled capacitors) may be required in some cases.
Figures 11 and 12 illustrate the relationship between the feedback resistor and the frequency and time domain response characteristics for a closed-loop gain of +2. (The response at other
gains will be similar.)
The 3 dB bandwidth is somewhat dependent on the power supply
voltage. As the supply voltage is decreased for example, the
magnitude of internal junction capacitances is increased, causing
a reduction in closed-loop bandwidth. To compensate for this,
smaller values of feedback resistor are used at lower supply
voltages.
REV. D
–9–
AD811
Driving Capacitive Loads
100
The feedback and gain resistor values in Table I will result in
very flat closed-loop responses in applications where the load
capacitances are below 10 pF. Capacitances greater than this
will result in increased peaking and overshoot, although not
necessarily in a sustained oscillation.
RFB
70
60
50
40
30
20
10
0
10
100
LOAD CAPACITANCE – pF
1000
Figure 33. Recommended Value of Series Resistor vs. the
Amount of Capacitive Load
Figure 33 shows recommended resistor values for different load
capacitances. Refer again to Figure 32 for an example of the
results of this method. Note that it may be necessary to adjust
the gain setting resistor, RG, to correct for the attenuation which
results due to the divider formed by the series resistor, RS, and
the load resistance.
+VS
0.1mF
RG
RS (OPTIONAL)
VOUT
AD811
VIN
80
VALUE OF RS – V
There are at least two very effective ways to compensate for this
effect. One way is to increase the magnitude of the feedback
resistor, which lowers the 3 dB frequency. The other method is
to include a small resistor in series with the output of the amplifier to isolate it from the load capacitance. The results of these
two techniques are illustrated in Figure 32. Using a 1.5 kΩ
feedback resistor, the output ripple is less than 0.5 dB when driving 100 pF. The main disadvantage of this method is that it
sacrifices a little bit of gain flatness for increased capacitive load
drive capability. With the second method, using a series resistor,
the loss of flatness does not occur.
G = +2
VS = 615V
RS VALUE SPECIFIED
IS FOR FLATTEST
FREQUENCY RESPONSE
90
CL
RL
RT
0.1mF
Applications which require driving a large load capacitance at a
high slew rate are often limited by the output current available
from the driving amplifier. For example, an amplifier limited to
25 mA output current cannot drive a 500 pF load at a slew rate
greater than 50 V/µs. However, because of the AD811’s 100 mA
output current, a slew rate of 200 V/µs is achievable when driving this same 500 pF capacitor (see Figure 34).
–VS
2V
Figure 31. Recommended Connection for Driving a Large
Capacitive Load
100ns
100
VIN
12
90
RFB = 1.5kV
RS = 0
9
GAIN – dB
6
VOUT 10
3
G = +2
VS = 615V
RL = 10kV
CL = 100pF
0
0%
RFB = 649V
RS = 30V
5V
Figure 34. Output Waveform of an AD811 Driving a
500 pF Load. Gain = +2, RFB = 649 Ω, RS = 15 Ω,
RS = 10 kΩ
–3
–6
1M
10M
FREQUENCY – Hz
100M
Figure 32. Performance Comparison of Two Methods for
Driving a Capacitive Load
–10–
REV. D
AD811
Operation as a Video Line Driver
1V
The AD811 has been designed to offer outstanding performance at closed-loop gains of one or greater, while driving
multiple reverse-terminated video loads. The lowest differential
gain and phase errors will be obtained when using ± 15 volt
power supplies. With ± 12 volt supplies, there will be an insignificant increase in these errors and a slight improvement in
gain flatness. Due to power dissipation considerations, ± 12 volt
supplies are recommended for optimum video performance.
Excellent performance can be achieved at much lower supplies
as well.
100
VIN
0%
Another important consideration when driving multiple cables
is the high frequency isolation between the outputs of the
cables. Due to its low output impedance, the AD811 achieves
better than 40 dB of output to output isolation at 5 MHz driving back terminated 75 Ω cables.
Figure 37. Small Signal Pulse Response, Gain = +2,
VS = ± 15 V
RF= 649V
FC= 3.58MHz
100 IRE
MODULATED
RAMP
DIFFERENTIAL GAIN – %
0.09
VOUT #1
75V
+VS
1V
0.10
75V CABLE
649V
90
VOUT 10
The closed-loop gain vs. frequency at different supply voltages
is shown in Figure 36. Figure 37 is an oscilloscope photograph
of an AD811 line driver’s pulse response with ± 15 volt supplies.
The differential gain and phase error vs. supply are plotted in
Figures 38 and 39, respectively.
649V
75V
0.1mF
0.08
0.07
0.06
a. DRIVING A SINGLE, BACK TERMINATED,
0.05
75V COAX CABLE
b. DRIVING TWO PARALLEL,
0.04
BACK TERMINATED, COAX CABLES
0.03
75V CABLE
VIN
AD811
75V CABLE
10ns
VOUT #2
b
0.02
75V
0.01
75V
a
75V
5
6
7
8
9
10
11
12
SUPPLY VOLTAGE – 6Volts
13
14
15
0.1mF
Figure 38. Differential Gain Error vs. Supply Voltage for
the Video Line Driver of Figure 35
–VS
Figure 35. A Video Line Driver Operating at a Gain of +2
0.20
G = +2
RL = 150V
RG = RFB
9
DIFFERENTIAL PHASE – Degrees
12
VS = 615V
RFB = 649V
GAIN – dB
6
3
VS = 65V
RFB = 562V
0
RF = 649V
FC = 3.58MHz
100 IRE
MODULATED
RAMP
0.18
0.16
0.14
0.12
a. DRIVING A SINGLE, BACK TERMINATED,
0.10
b. DRIVING TWO PARALLEL,
75V COAX CABLE
BACK TERMINATED, COAX CABLES
0.08
b
0.06
0.04
–3
0.02
–6
1
10
FREQUENCY – MHz
5
100
6
7
8
9
10
11
12
SUPPLY VOLTAGE – 6Volts
13
14
15
Figure 39. Differential Phase Error vs. Supply Voltage for
the Video Line Driver of Figure 35
Figure 36. Closed-Loop Gain vs. Frequency, Gain = +2
REV. D
a
–11–
AD811
The gain can be increased to 20 dB (×10) by raising R8 and R9
to 1.27 kΩ, with a corresponding decrease in –3 dB bandwidth
to about 25 MHz. The maximum output voltage under these
conditions will be increased to ± 9 V using ± 12 V supplies.
An 80 MHz Voltage-Controlled Amplifier Circuit
The voltage-controlled amplifier (VCA) circuit of Figure 40
shows the AD811 being used with the AD834, a 500 MHz,
4-quadrant multiplier. The AD834 multiplies the signal input
by the dc control voltage, VG. The AD834 outputs are in the
form of differential currents from a pair of open collectors,
ensuring that the full bandwidth of the multiplier (which exceeds 500 MHz) is available for certain applications. Here,
the AD811 op amp provides a buffered, single-ended groundreferenced output. Using feedback resistors R8 and R9 of
511 Ω, the overall gain ranges from –70 dB, for VG = 0 dB to
+12 dB, (a numerical gain of four), when VG = +1 V. The overall transfer function of the VCA is:
The gain-control input voltage, VG, may be a positive or negative ground-referenced voltage, or fully differential, depending
on the user’s choice of connections at Pins 7 and 8. A positive
value of VG results in an overall noninverting response. Reversing the sign of VG simply causes the sign of the overall response
to invert. In fact, although this circuit has been classified as a
voltage-controlled amplifier, it is also quite useful as a generalpurpose four-quadrant multiplier, with good load-driving capabilities and fully-symmetrical responses from X- and Y-inputs.
VOUT = 4 (X1 – X2)(Y1 – Y2)
The AD811 and AD834 can both be operated from power
supply voltages of ± 5 V. While it is not necessary to power them
from the same supplies, the common-mode voltage at W1 and
W2 must be biased within the common-mode range of the
AD811’s input stage. To achieve the lowest differential gain and
phase errors, it is recommended that the AD811 be operated
from power supply voltages of ± 10 volts or greater. This VCA
circuit is designed to operate from a ± 12 volt dual power
supply.
which reduces to VOUT = 4 VG VIN using the labeling conventions shown in Figure 40. The circuit’s –3 dB bandwidth of
80 MHz, is maintained essentially constant—independent of
gain. The response can be maintained flat to within ± 0.1 dB
from dc to 40 MHz at full gain with the addition of an optional
capacitor of about 0.3 pF across the feedback resistor R8. The
circuit produces a full-scale output of ± 4 V for a ± 1 V input,
and can drive a reverse-terminated load of 50 Ω or 75 Ω to ±2 V.
FB
+12V
C1
0.1mF
+
R1 100V
–
R2 100V
R8*
VG
8
7
X2
X1 +V S
6
5
W1
R4
182V
R6
294V
U1
AD834
Y1
Y2
–VS
1
2
3
U3
AD811
W2
4
VOUT
R7
294V
R5
182V
RL
VIN
R9*
R3
249V
C2
0.1mF
–12V
*R8 = R9 = 511V FOR X4 GAIN
= 1.27kV FOR X10 GAIN
FB
Figure 40. An 80 MHz Voltage-Controlled Amplifier
–12–
REV. D
AD811
A Video Keyer Circuit
By using two AD834 multipliers, an AD811, and a 1 V dc
source, a special form of a two-input VCA circuit called a
video keyer can be assembled. “Keying” is the term used in
reference to blending two or more video sources under the
control of a third signal or signals to create such special effects
as dissolves and overlays. The circuit shown in Figure 41 is a
two-input keyer, with video inputs VA and VB, and a control
input VG. The transfer function (with VOUT at the load) is
given by:
VOUT = G VA + (1–G) VB
where G is a dimensionless variable (actually, just the gain of
the “A” signal path) that ranges from 0 when VG = 0, to 1
when VG = +1 V. Thus, VOUT varies continuously between VA
and VB as G varies from 0 to 1.
Circuit operation is straightforward. Consider first the signal
path through U1, which handles video input VA. Its gain is
clearly zero when VG = 0 and the scaling we have chosen
ensures that it is unity when VG = +1 V; this takes care of the
first term of the transfer function. On the other hand, the VG
input to U2 is taken to the inverting input X2 while X1 is
biased at an accurate +1 V. Thus, when VG = 0, the response
to video input VB is already at its full-scale value of unity,
whereas when VG = +1 V, the differential input X1–X2 is zero.
This generates the second term.
The bias currents required at the output of the multipliers are
provided by R8 and R9. A dc-level-shifting network comprising
R10/R12 and R11/R13 ensures that the input nodes of the
AD811 are positioned at a voltage within its common-mode
range. At high frequencies C1 and C2 bypass R10 and R11
respectively. R14 is included to lower the HF loop gain, and is
needed because the voltage-to-current conversion in the
AD834s, via the Y2 inputs, results in an effective value of the
feedback resistance of 250 Ω; this is only about half the value
required for optimum flatness in the AD811’s response. (Note
that this resistance is unaffected by G: when G = 1, all the
feedback is via U1, while when G = 0 it is all via U2). R14
reduces the fractional amount of output current from the multipliers into the current-summing inverting input of the AD811,
by sharing it with R8. This resistor can be used to adjust the
bandwidth and damping factor to best suit the application.
To generate the 1 V dc needed for the “1–G” term an AD589
reference supplies 1.225 V ± 25 mV to a voltage divider consisting of resistors R2 through R4. Potentiometer R3 should be
adjusted to provide exactly +1 V at the X1 input.
In this case, we have shown an arrangement using dual supplies
of ± 5 V for both the AD834 and the AD811. Also, the overall
gain in this case is arranged to be unity at the load, when it is
driven from a reverse-terminated 75 Ω line. This means that the
“dual VCA” has to operate at a maximum gain of 2, rather
C1
0.1mF
+5V
R7
45.3V
R5
113V
VG
R14
SEE TEXT
R10
2.49kV
TO PIN 6
AD811
R6
226V
(0 TO +1V dc)
SETUP FOR DRIVING
REVERSE-TERMINATED LOAD
ZO
VOUT
ZO
200V
TO Y2
8
X2
7
6
X1 +VS
5
W1
200V
+5V
R1
1.87kV
R2
174V
INSET
U1
AD834
U4
AD589
Y1
Y2
–VS
1
2
3
R8
29.4V
R12
6.98kV
+5V
W2
4
VA
FB
(61V FS)
C3
0.1mF
–5V
+5V
R3
100V
R4
1.02kV
8
X2
7
6
X1 +V S
R9
29.4V
5
W1
R13
6.98kV
Y1
Y2
–VS
W2
1
2
3
4
U3
AD811
R11
2.49kV
–5V
Figure 41. A Practical Video Keyer Circuit
–13–
VOUT
C4
0.1mF
LOAD
GND
FB
(61V FS)
REV. D
LOAD
GND
C2
0.1mF
U1
AD834
VB
–5V
–5V
AD811
than 4 as in the VCA circuit of Figure 40. However, this cannot
be achieved by lowering the feedback resistor, since below a
critical value (not much less than 500 Ω) the AD811’s peaking
may be unacceptable. This is because the dominant pole in the
open-loop ac response of a current-feedback amplifier is controlled by this feedback resistor. It would be possible to operate
at a gain of X4 and then attenuate the signal at the output.
Instead, we have chosen to attenuate the signals by 6 dB at the
input to the AD811; this is the function of R8 through R11.
R14 = 49.9V
0
CLOSED-LOOP GAIN – dB
–10
Figure 42 is a plot of the ac response of the feedback keyer,
when driving a reverse terminated 50 Ω cable. Output noise and
adjacent channel feedthrough, with either channel fully off and
the other fully on, is about –50 dB to 10 MHz. The feedthrough
at 100 MHz is limited primarily by board layout. For VG = +1 V,
the –3 dB bandwidth is 15 MHz when using a 137 Ω resistor for
R14 and 70 MHz with R14 = 49.9 Ω. For further information
regarding the design and operation of the VCA and video keyer
circuits, refer to the application note “Video VCA’s and Keyers
Using the AD834 & AD811” by Brunner, Clarke, and Gilbert,
available FREE from Analog Devices.
GAIN
R14 = 137V
–20
–30
–40
–50
ADJACENT
CHANNEL
FEEDTHROUGH
–60
–70
–80
10k
100k
1M
10M
100M
FREQUENCY – Hz
Figure 42. A Plot of the AC Response of the Video Keyer
–14–
REV. D
AD811
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
20-Lead LCC (E-20A) Package
0.082 ± 0.018
(2.085 ± 0.455)
0.39 (9.91)
MAX
8
0.350 ± 0.008 SQ
(8.89 ± 0.20) SQ
0.040 x 45°
(1.02 x 45°)
REF 3 PLCS
5
0.25 0.31
(6.35) (7.87)
1
4
0.025 ± 0.003
(0.635 ± 0.075)
PIN 1
0.30 (7.62)
REF
0.10 (2.54)
BSC
0.165 6 0.01
(4.19 6 0.25)
0.050
(1.27)
0.011 6 0.003
(0.28 6 0.08)
0.18 6 0.03
(4.57 6 0.75)
0.125 (3.18)
MIN
0.018 6 0.003
(0.46 6 0.08)
NO. 1 PIN
INDEX
0.035 6 0.01
(0.89 6 0.25)
0.020 x 45°
(0.51 x 45°)
REF
158
08
SEATING
PLANE
0.033
(0.84)
NOM
C1592b–0–8/99
8- Lead Plastic DIP (N) Package
16-Lead SOIC (R-16) Package
8-Lead Cerdip (Q) Package
9
16
0.005 (0.13)
MIN
0.055 (1.4)
MAX
8
0.299 (7.60)
0.291 (7.40)
5
PIN 1
1
0.419 (10.65)
0.404 (10.26)
PIN 1
0.310 (7.87)
0.220 (5.59)
8
1
4
0.100 (2.54) BSC
0.060 (1.52)
0.015 (0.38)
0.200.(5.08)
MAX
0.150
(3.81)
MIN
0.200 (5.08)
0.125 (3.18)
SEATING
0.023 (0.58) 0.070 (1.78) PLANE
0.014 (0.36) 0.030 (0.76)
0.010 (0.25)
0.004 (0.10)
0.015 (0.38)
0.008 (0.20)
15°
0°
0.107 (2.72)
0.089 (2.26)
0.413 (10.50)
0.398 (10.10)
0.320 (8.13)
0.290 (7.37)
0.405 (10.29) MAX
0.050 (1.27)
BSC
0.018 (0.46)
0.014 (0.36)
0.364 (9.246)
0.344 (8.738)
0.045 (1.15)
0.020 (0.50)
0.015 (0.38)
0.007 (1.18)
20-Lead Wide Body SOIC (R-20) Package
8-Lead SOIC (SO-8) Package
0.512 (13.00)
0.496 (12.60)
0.1968 (5.00)
0.1890 (4.80)
20
5
1
4
11
0.2440 (6.20)
0.2284 (5.80)
0.300 (7.60)
0.292 (7.40)
0.419 (10.65)
0.394 (10.00)
PIN 1
0.0196 (0.50)
3 458
0.0099 (0.25)
0.0500 (1.27)
BSC
0.0098 (0.25)
0.0040 (0.10)
SEATING
PLANE
10
1
0.0688 (1.75)
0.0532 (1.35)
0.0192 (0.49)
0.0138 (0.35)
88
0.0500 (1.27)
0.0098 (0.25) 08
0.0160 (0.41)
0.0075 (0.19)
0.50 (1.27)
BSC
0.019 (0.48)
0.014 (0.36)
0.104 (2.64)
0.093 (2.36)
0.011 (0.28)
0.004 (0.10)
0.015 (0.38)
0.007 (0.18)
REV. D
–15–
0.050 (1.27)
0.016 (0.40)
PRINTED IN U.S.A.
0.1574 (4.00)
0.1497 (3.80)
8