LINER LT3682 High efficiency, 65v 500ma synchronous Datasheet

LTC3630
High Efficiency, 65V
500mA Synchronous
Step-Down Converter
FEATURES
DESCRIPTION
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The LTC®3630 is a high efficiency step-down DC/DC
converter with internal high side and synchronous power
switches that draws only 12μA typical DC supply current
while maintaining a regulated output voltage at no load.
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Wide Operating Input Voltage Range: 4V to 65V
Synchronous Operation for Highest Efficiency
Internal High Side and Low Side Power MOSFETs
No Compensation Required
Adjustable 50mA to 500mA Maximum Output Current
Low Dropout Operation: 100% Duty Cycle
Low Quiescent Current: 12μA
Wide Output Range: 0.8V to VIN
0.8V ±1% Feedback Voltage Reference
Precise RUN Pin Threshold
Internal and External Soft-Start
Programmable 1.8V, 3.3V, 5V or Adjustable Output
Few External Components Required
Low Profile (0.75mm) 3mm × 5mm DFN and
Thermally-Enhanced MSE16 Packages
APPLICATIONS
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Industrial Control Supplies
Medical Devices
Distributed Power Systems
Portable Instruments
Battery-Operated Devices
Automotive
Avionics
The LTC3630 can supply up to 500mA load current and
features a programmable peak current limit that provides
a simple method for optimizing efficiency and for reducing output ripple and component size. The LTC3630’s
combination of Burst Mode® operation, integrated power
switches, low quiescent current, and programmable peak
current limit provides high efficiency over a broad range
of load currents.
With its wide input range of 4V to 65V, the LTC3630 is a
robust converter suited for regulating a wide variety of
power sources. Additionally, the LTC3630 includes a precise
run threshold and soft-start feature to guarantee that the
power system start-up is well-controlled in any environment. A feedback comparator output enables multiple
LTC3630s to be paralleled in higher current applications.
The LTC3630 is available in the thermally-enhanced
3mm × 5mm DFN and the MSE16 packages.
L, LT, LTC, LTM, Burst Mode, Linear Technology and the Linear logo are registered trademarks
and ThinSOT is a trademark of Linear Technology Corporation. All other trademarks are the
property of their respective owners.
TYPICAL APPLICATION
Efficiency vs Load Current
100
4V to 65V Input to 3.3V Output, 500mA Step-Down Converter
47μH
VIN
2.2μF
SW
LTC3630
RUN
VFB
SS
VPRG1 VPRG2
FBO
ISET
GND
VOUT
3.3V
100μF 500mA
w2
VIN = 12V
80
EFFICIENCY (%)
VIN
4V TO 65V
VOUT = 3.3V
90
70
VIN = 65V
60
50
3630 TA01a
40
30
0.1
ISET = 220kΩ||220pF
ISET = OPEN
1
10
100
LOAD CURRENT (mA)
1000
3630 TA01b
3630fb
1
LTC3630
ABSOLUTE MAXIMUM RATINGS
(Note 1)
VIN Supply Voltage ..................................... –0.3V to 70V
SW Voltage (DC) ........................... –0.3V to (VIN + 0.3V)
RUN Voltage................................................. –0.3V to 6V
SS, FBO, ISET Voltages ................................. –0.3V to 6V
VFB, VPRG1, VPRG2 Voltages ......................... –0.3V to 6V
Operating Junction Temperature Range (Notes 2, 3, 4)
LTC3630E, LTC3630I ......................... –40°C to 125°C
LTC3630H .......................................... –40°C to 150°C
LTC3630MP ....................................... –55°C to 150°C
Storage Temperature Range .................. –65°C to 150°C
Lead Temperature (Soldering, 10 sec)
MSOP ............................................................... 300°C
PIN CONFIGURATION
TOP VIEW
TOP VIEW
SW 1
16 GND
VIN 3
14 GND
RUN 5
VPRG2 6
VPRG1 7
GND 8
17
GND
12
11
10
9
FBO
ISET
SS
VFB
MSE PACKAGE
VARIATION: MSE16 (12)
16-LEAD PLASTIC MSOP
TJMAX = 150°C, θJA = 45°C/W, θJC = 10°C/W
EXPOSED PAD (PIN 17) IS GND, MUST BE SOLDERED TO PCB
SW
1
16 GND
NC
2
15 NC
VIN
3
NC
4
RUN
5
VPRG2
6
11 ISET
VPRG1
7
10 SS
GND
8
9
14 GND
17
GND
13 NC
12 FBO
VFB
DHC PACKAGE
16-LEAD (5mm w 3mm) PLASTIC DFN
(NOTE 6)
TJMAX = 150°C, θJA = 43°C/W, θJC = 5°C/W
EXPOSED PAD (PIN 17) IS GND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3630EMSE#PBF
LTC3630EMSE#TRPBF
3630
16-Lead Plastic MSOP
–40°C to 125°C
LTC3630IMSE#PBF
LTC3630IMSE#TRPBF
3630
16-Lead Plastic MSOP
–40°C to 125°C
LTC3630HMSE#PBF
LTC3630HMSE#TRPBF
3630
16-Lead Plastic MSOP
–40°C to 150°C
LTC3630MPMSE#PBF
LTC3630MPMSE#TRPBF
3630
16-Lead Plastic MSOP
–55°C to 150°C
LTC3630EDHC#PBF
LTC3630EDHC#TRPBF
3630
16-Lead (5mm × 3mm) Plastic DFN
–40°C to 125°C
LTC3630IDHC#PBF
LTC3630IDHC#TRPBF
3630
16-Lead (5mm × 3mm) Plastic DFN
–40°C to 125°C
LTC3630HDHC#PBF
LTC3630HDHC#TRPBF
3630
16-Lead (5mm × 3mm) Plastic DFN
–40°C to 150°C
LTC3630MPDHC#PBF
LTC3630MPDHC#TRPBF
3630
16-Lead (5mm × 3mm) Plastic DFN
–55°C to 150°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
3630fb
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LTC3630
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the specified operating
junction temperature range, otherwise specifications are at TA = 25°C (Note 2). VIN = 12V, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Input Supply (VIN)
VIN
Input Voltage Operating Range
4
65
V
VOUT
Output Voltage Operating Range
0.8
VIN
V
UVLO
VIN Undervoltage Lockout
3.65
3.5
150
3.85
3.70
V
V
mV
IQ
DC Supply Current (Note 5)
Active Mode
Sleep Mode
Shutdown Mode
165
12
5
270
20
10
μA
μA
μA
1.17
1.06
1.21
1.10
110
1.25
1.14
V
V
mV
V
V
VRUN
RUN Pin Threshold Voltage
l
l
VIN Rising
VIN Falling
Hysteresis
3.45
3.30
No Load
VRUN = 0V
RUN Rising
RUN Falling
Hysteresis
Output Supply (VFB)
VFB(ADJ)
Feedback Comparator Threshold Voltage
(Adjustable Output)
VFB Rising, VPRG1 = VPRG2 = 0V
LTC3630E, LTC3630I
LTC3630H, LTC3630MP
l
l
0.792
0.788
0.800
0.800
0.808
0.812
VFBH
Feedback Comparator Hysteresis
(Adjustable Output)
VFB Falling, VPRG1 = VPRG2 = 0V
l
2.5
5
7
mV
IFB
Feedback Pin Current
VFB = 1V, VPRG1 = 0V, VPRG2 = 0V
–10
0
10
nA
VFB(FIXED)
Feedback Comparator Threshold Voltages
(Fixed Output)
VFB Rising, VPRG1 = SS, VPRG2 = 0V
VFB Falling, VPRG1 = SS, VPRG2 = 0V
l
l
4.940
4.910
5.015
4.985
5.090
5.060
V
V
VFB Rising, VPRG1 = 0V, VPRG2 = SS
VFB Falling, VPRG1 = 0V, VPRG2 = SS
l
l
3.260
3.240
3.310
3.290
3.360
3.340
V
V
VFB Rising, VPRG1 = VPRG2 = SS
VFB Falling, VPRG1 = VPRG2 = SS
l
l
1.780
1.770
1.810
1.8
1.840
1.83
V
V
Feedback Voltage Line Regulation
VIN = 4V to 65V
IPEAK
Peak Current Comparator Threshold
ISET Floating
100k Resistor from ISET to GND
ISET Shorted to GND
RON
Power Switch On-Resistance
Top Switch
Bottom Switch
ISW = –200mA
ISW = 200mA
ΔVLINEREG
0.001
%/V
Operation
1
0.45
0.09
1.2
0.6
0.12
1.4
0.75
0.15
1.00
0.53
ILSW
Switch Pin Leakage Current
RUN = Open, VIN = 65V, SW = 0V
ISS
Soft-Start Pin Pull-Up Current
VSS < 2.5V
tINT(SS)
Internal Soft-Start Time
SS Pin Floating
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC3630 is tested under pulsed load conditions such that
TJ ≈ TA. The LTC3630E is guaranteed to meet performance specifications
from 0°C to 85°C. Specifications over the –40°C to 125°C operating
junction temperature range are assured by design, characterization and
correlation with statistical process controls. The LTC3630I is guaranteed
over the –40°C to 125°C operating junction temperature range, the
LTC3630H is guaranteed over the –40°C to 150°C operating junction
temperature range and the LTC3630MP is tested and guaranteed over the
–55°C to 150°C operating junction temperature range.
l
l
l
3
A
A
A
Ω
Ω
0.1
1
μA
5
6
μA
0.8
ms
High junction temperatures degrade operating lifetimes; operating lifetime
is derated for junction temperatures greater than 125°C. Note that the
maximum ambient temperature consistent with these specifications is
determined by specific operating conditions in conjunction with board
layout, the rated package thermal impedance and other environmental
factors.
Note 3: The junction temperature (TJ, in °C) is calculated from the ambient
temperature (TA, in °C) and power dissipation (PD, in Watts) according to
the formula:
TJ = TA + (PD • θJA)
where θJA is 43°C/W for the DFN or 45°C/W for the MSOP.
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LTC3630
ELECTRICAL CHARACTERISTICS
junction temperature may impair device reliability or permanently damage
the device. The overtemperature protection level is not production tested.
Note 5: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. See Applications Information.
Note 6: For application concerned with pin creepage and clearance
distances at high voltages, the MSOP package should be used. See
Applications Information.
Note that the maximum ambient temperature consistent with these
specifications is determined by specific operating conditions in
conjunction with board layout, the rated package thermal impedance and
other environmental factors.
Note 4: This IC includes over temperature protection that is intended to
protect the device during momentary overload conditions. The maximum
rated junction temperature will be exceeded when this protection is active.
Continuous operation above the specified absolute maximum operating
TYPICAL PERFORMANCE CHARACTERISTICS
Load Step Transient Response
Soft-Start Waveform
OUTPUT
VOLTAGE
2V/DIV
INDUCTOR
CURRENT
500mA/DIV
OUTPUT
VOLTAGE
50mV/DIV
OUTPUT
VOLTAGE
2V/DIV
LOAD
CURRENT
200mA/DIV
INDUCTOR
CURRENT
500mA/DIV
3630 G01
Efficiency and Power Loss
vs Load Current, VOUT = 5V
100
90
90
EFFICIENCY
10
VOUT = 5V
FIGURE 13 CIRCUIT
VIN = 12V
1
VIN = 65V
30
20
10
1
10
100
LOAD CURRENT (mA)
1000
3630 G04
EFFICIENCY (%)
EFFICIENCY (%)
100
POWER
80
EFFICIENCY
70
1000
60
50
100
POWER
40
10
VOUT = 3.3V
FIGURE 13 CIRCUIT
VIN = 12V
1
VIN = 65V
30
20
10
0
0.1
1
10
100
LOAD CURRENT (mA)
1000
3630 G05
EFFICIENCY
70
1000
60
50
100
POWER
40
POWER LOSS (mW)
50
40
VOUT = 1.8V
90 FIGURE 13 CIRCUIT
POWER LOSS (mW)
60
0
0.1
POWER LOSS (mW)
1000
Efficiency and Power Loss
vs Load Current, VOUT = 1.8V
100
80
70
3630 G03
VIN = 12V
200μs/DIV
VOUT = 5V
FIGURE 13 CIRCUIT
Efficiency and Power Loss
vs Load Current, VOUT = 3.3V
100
80
3630 G02
VIN = 12V
500μs/DIV
VOUT = 5V
FIGURE 13 CIRCUIT
EFFICIENCY (%)
COUT = 100μF
1ms/DIV
FIGURE 13 CIRCUIT
Short-Circuit Response
10
30
20
VIN = 12V 1
VIN = 65V
10
0
0.1
1
10
100
LOAD CURRENT (mA)
1000
3630 G06
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LTC3630
TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency vs Input Voltage
95
Line Regulation vs Input Voltage
0.05
VOUT = 5V
FIGURE 13 CIRCUIT
90
Load Regulation vs Load Current
5.04
FIGURE 13 CIRCUIT
ILOAD = 500mA
0.04
VIN = 12V
VOUT = 5V
FIGURE 13 CIRCUIT
5.03
0.03
80
75
70
0.02
0.01
0
–0.01
–0.02
ILOAD = 500mA
ILOAD = 100mA
ILOAD = 10mA
ILOAD = 1mA
60
55
20
10
50
40
INPUT VOLTAGE (V)
4.97
15
5
60
35
45
25
INPUT VOLTAGE (V)
125
5.5
5.3
5.2
5.1
5.0
4.9
4.8
4.7
4.6
4.5
–55
–25
65
35
5
95
TEMPERATURE (°C)
125
Peak Current Trip Threshold
vs RISET
PEAK CURRENT TRIP THRESHOLD (mA)
PEAK CURRENT TRIP THRESHOLD (mA)
800
600
400
200
50
100
150
200
250
RISET (kΩ)
3630 G13
RISET = 100kΩ
600
400
ISET = GND
200
–25
65
95
5
35
TEMPERATURE (°C)
125
155
3630 G12
Quiescent VIN Supply Current
vs Input Voltage
16
ISET = OPEN
SLEEP
14
1200
1000
800
ISET = 100k
600
400
ISET = 0V
200
0
10
40
30
50
20
INPUT VOLTAGE (V)
12
10
8
SHUTDOWN
6
4
2
0
0
0
800
0
–55
155
1400
1000
ISET OPEN
1000
Peak Current Trip Threshold
vs Input Voltage
1200
VIN = 12V
1200
3630 G11
3630 G10
500
Peak Current Trip Threshold
vs Temperature and ISET
1400
VIN = 12V
5.4
155
VIN = 12V
200
400
300
LOAD CURRENT (mA)
3630 G09
PEAK CURRENT TRIP THRESHOLD (mA)
0.798
100
0
3630 G08
FEEDBACK COMPARATOR HYSTERESIS (mV)
FEEDBACK COMPARATOR TRIP VOLTAGE (V)
0.800
95
5
35
65
TEMPERATURE (°C)
4.96
65
55
Feedback Comparator Hysteresis
vs Temperature
0.802
1400
4.98
–0.05
30
VIN = 12V
–25
4.99
–0.04
Feedback Comparator Trip
Voltage vs Temperature
0.796
–55
5.00
–0.03
3630 G07
0.804
5.01
VIN SUPPLY CURRENT (μA)
65
5.02
OUTPUT VOLTAGE (V)
ΔVOUT/VOUT (%)
EFFICIENCY (%)
85
60
0
5
15
25
35
45
55
65
VIN VOLTAGE (V)
3630 G14
3630 G15
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LTC3630
TYPICAL PERFORMANCE CHARACTERISTICS
Quiescent VIN Supply Current
vs Temperature
Switch On-Resistance
vs Temperature
2.0
VIN = 12V
1.6
1.8
SLEEP
12
8
SHUTDOWN
4
1.6
1.4
1.2
TOP
1.0
0.8
BOTTOM
0.6
0.4
–25
65
5
95
35
TEMPERATURE (°C)
125
0
155
0
10
40
30
20
50
INPUT VOLTAGE (V)
3630 G16
BOTTOM
0.6
0.4
0
–55
60
–25
95
65
35
TEMPERATURE (°C)
5
125
155
3630 G18
Operating Waveforms
1.30
VIN = 65V
10
8
6
4
SW = 65V
2
0
–2
SW = 0V
–4
–6
–55 –25
0.8
RUN Comparator Threshold
Voltage vs Temperature
RUN COMPARATOR THRESHOLD (V)
SWITCH LEAKAGE CURRENT (μA)
12
TOP
1.0
3630 G17
Switch Leakage Current
vs Temperature
14
1.2
0.2
0.2
0
–55
VIN = 12V
1.4
SWITCH ON-RESISTANCE (Ω)
16
SWITCH ON-RESISTANCE (Ω)
VIN SUPPLY CURRENT (μA)
20
Switch On-Resistance
vs Input Voltage
95
65
35
TEMPERATURE (°C)
5
125
155
3630 G19
SWITCH
VOLTAGE
25V/DIV
1.25
3*4*/(
1.20
OUTPUT
VOLTAGE
50mV/DIV
INDUCTOR
CURRENT
500mA/DIV
1.15
'"--*/(
1.10
1.05
1.00
–55 –25
35
95
5
TEMPERATURE (°C)
125
155
VIN = 65V
10μs/DIV
VOUT = 5V
ILOAD = 350mA
FIGURE 13 CIRCUIT
3630 G21
-5t(
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LTC3630
PIN FUNCTIONS
SW (Pin 1): Switch Node Connection to Inductor. This
pin connects to the drains of the internal power MOSFET
switches.
NC (Pins 2, 4, 13, 15 DHC Package Only): No Internal
Connection. Leave these pins open.
VIN (Pin 3): Main Input Supply Pin. A ceramic bypass
capacitor should be tied between this pin and GND.
RUN (Pin 5): Run Control Input. A voltage on this pin
above 1.21V enables normal operation. Forcing this pin
below 0.7V shuts down the LTC3630, reducing quiescent
current to approximately 5μA. Optionally, connect to the
input supply through a resistor divider to set the undervoltage lockout. An internal 2M resistor and 2μA current
source pulls this pin up to an internal 5V reference. See
Applications Information.
VPRG2, VPRG1 (Pins 6, 7): Output Voltage Selection. Short
both pins to ground for an external resistive divider programmable output voltage. Short VPRG1 to SS and short
VPRG2 to ground for a 5V output voltage. Short VPRG1 to
ground and short VPRG2 to SS for a 3.3V output voltage.
Short both pins to SS for a 1.8V output voltage.
GND (Pins 8, 14, 16, Exposed Pad Pin 17): Ground. The
exposed backside pad must be soldered to the PCB ground
plane for optimal thermal performance.
SS (Pin 10): Soft-Start Control Input. A capacitor to
ground at this pin sets the output voltage ramp time. A
50μA current initially charges the soft-start capacitor until
switching begins, at which time the current is reduced to
its nominal value of 5μA. The output voltage ramp time
from zero to its regulated value is 1ms for every 16.5nF
of capacitance from SS to GND. If left floating, the ramp
time defaults to an internal 0.8ms soft-start.
ISET (Pin 11): Peak Current Set Input and Voltage Output
Ripple Filter. A resistor from this pin to ground sets the
peak current comparator threshold. Leave floating for the
maximum peak current (1.2A typical) or short to ground
for minimum peak current (0.12A typical). The maximum
output current is one-half the peak current. The 5μA current
that is sourced out of this pin when switching, is reduced
to 1μA in sleep. Optionally, a capacitor can be placed from
this pin to GND to trade off efficiency for light load output
voltage ripple. See Applications Information.
FBO (Pin 12): Feedback Comparator Output. Connect
to the VFB pins of additional LTC3630s to combine the
output current. The typical pull-up current is 20μA. The
typical pull- down impedance is 70Ω. See Applications
Information.
VFB (Pin 9): Output Voltage Feedback. When configured
for an adjustable output voltage, connect to an external
resistive divider to divide the output voltage down for
comparison to the 0.8V reference. For the fixed output
configuration, directly connect this pin to the output supply.
3630fb
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LTC3630
BLOCK DIAGRAM
1.3V
11
ACTIVE: 5μA
SLEEP: 1μA
ISET
VIN
3
CIN
PEAK CURRENT
COMPARATOR
+
5V
–
SLEEP, ACTIVE: 2μA
SHUTDOWN: 0μA
2M
5
VIN
+
RUN
+
1.21V
LOGIC
AND
SHOOTTHROUGH
PREVENTION
–
SW
L1
VOUT
1
COUT
GND
16
+
–
5V
REVERSE CURRENT
COMPARATOR
20μA
FEEDBACK
COMPARATOR
12
FBO
VOLTAGE
REFERENCE
+
+
–
70Ω
5V
START-UP: 50μA
NORMAL: 5μA
0.800V
R1
14
8
17
R2
GND
GND
VPRG2 VPRG1
GND
GND
SS
SS
GND
SS
GND
SS
VOUT
ADJUSTABLE
5V FIXED
3.3V FIXED
1.8V FIXED
R1
VFB
VPRG1
VPRG2
R2
1.0M ∞
4.2M 800k
2.5M 800k
1.0M 800k
SS
10
9
7
6
IMPLEMENT DIVIDER
EXTERNALLY FOR
ADJUSTABLE VERSION
3630 BD
3630fb
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LTC3630
OPERATION
(Refer to Block Diagram)
The LTC3630 is a synchronous step-down DC/DC converter with internal power switches that uses Burst Mode
control. The low quiescent current and high switching
frequency results in high efficiency across a wide range
of load currents. Burst Mode operation functions by using
short “burst” cycles to switch the inductor current through
the internal power MOSFETs, followed by a sleep cycle
where the power switches are off and the load current is
supplied by the output capacitor. During the sleep cycle,
the LTC3630 draws only 12μA of supply current. At light
loads, the burst cycles are a small percentage of the total
cycle time which minimizes the average supply current,
greatly improving efficiency. Figure 1 shows an example
of Burst Mode operation. The switching frequency and the
number of switching cycles during Burst Mode operation
are dependent on the inductor value, peak current, load
current, input voltage and output voltage.
SLEEP
CYCLE
BURST
CYCLE
SWITCHING
FREQUENCY
INDUCTOR
CURRENT
BURST
FREQUENCY
OUTPUT
VOLTAGE
ΔVOUT
3630 F01
Figure 1. Burst Mode Operation
Main Control Loop
The LTC3630 uses the VPRG1 and VPRG2 control pins to
connect internal feedback resistors to the VFB pin. This
enables fixed outputs of 1.8V, 3.3V or 5V without increasing component count, input supply current or exposure to
noise on the sensitive input to the feedback comparator.
External feedback resistors (adjustable mode) can still
be used by connecting both VPRG1 and VPRG2 to ground.
In adjustable mode the feedback comparator monitors
the voltage on the VFB pin and compares it to an internal 800mV reference. If this voltage is greater than the
reference, the comparator activates a sleep mode in which
the power switches and current comparators are disabled,
reducing the VIN pin supply current to only 12μA. As the
load current discharges the output capacitor, the voltage
on the VFB pin decreases. When this voltage falls 5mV
below the 800mV reference, the feedback comparator
trips and enables burst cycles.
At the beginning of the burst cycle, the internal high side
power switch (P-channel MOSFET) is turned on and the
inductor current begins to ramp up. The inductor current
increases until either the current exceeds the peak current comparator threshold or the voltage on the VFB pin
exceeds 800mV, at which time the high side power switch
is turned off and the low side power switch (N-channel
MOSFET) turns on. The inductor current ramps down until
the reverse current comparator trips, signaling that the
current is close to zero. If the voltage on the VFB pin is
still less than the 800mV reference, the high side power
switch is turned on again and another cycle commences.
The average current during a burst cycle will normally be
greater than the average load current. For this architecture,
the maximum average output current is equal to half of
the peak current.
The hysteretic nature of this control architecture results
in a switching frequency that is a function of the input
voltage, output voltage, and inductor value. This behavior
provides inherent short-circuit protection. If the output is
shorted to ground, the inductor current will decay very
slowly during a single switching cycle. Since the high side
switch turns on only when the inductor current is near
zero, the LTC3630 inherently switches at a lower frequency
during start-up or short-circuit conditions.
Start-Up and Shutdown
If the voltage on the RUN pin is less than 0.7V, the LTC3630
enters a shutdown mode in which all internal circuitry is
disabled, reducing the DC supply current to 5μA. When the
voltage on the RUN pin exceeds 1.21V, normal operation
of the main control loop is enabled. The RUN pin comparator has 110mV of internal hysteresis, and therefore
must fall below 1.1V to stop switching and disable the
main control loop.
3630fb
9
LTC3630
OPERATION
(Refer to Block Diagram)
An internal 0.8ms soft-start function limits the ramp rate
of the output voltage on start-up to prevent excessive input
supply droop. If a longer ramp time and consequently less
supply droop is desired, a capacitor can be placed from
the SS pin to ground. The 5μA current that is sourced
out of this pin will create a smooth voltage ramp on the
capacitor. If this ramp rate is slower than the internal
0.8ms soft-start, then the output voltage will be limited
by the ramp rate on the SS pin instead. The internal and
external soft-start functions are reset on start-up and after
an undervoltage event on the input supply.
Dropout Operation
The peak inductor current is not limited by the internal or
external soft-start functions; however, placing a capacitor
from the ISET pin to ground does provide this capability.
When the input supply decreases toward the output supply, the duty cycle increases to maintain regulation. The
P-channel MOSFET top switch in the LTC3630 allows the
duty cycle to increase all the way to 100%. At 100% duty
cycle, the P-channel MOSFET stays on continuously, providing output current equal to the peak current, which can
be greater than 1A. The power dissipation of the LTC3630
can increase dramatically during dropout operation especially at input voltages less than 10V. The increased power
dissipation is due to higher potential output current and
increased P-channel MOSFET on-resistance. See the Thermal Considerations section of the Applications Information
for a detailed example.
Peak Inductor Current Programming
Input Voltage and Overtemperature Protection
The peak current comparator nominally limits the peak
inductor current to 1.2A. This peak inductor current can
be adjusted by placing a resistor from the ISET pin to
ground. The 5μA current sourced out of this pin through
the resistor generates a voltage that adjusts the peak current comparator threshold.
When using the LTC3630, care must be taken not to
exceed any of the ratings specified in the Absolute Maximum Ratings section. As an added safeguard, however,
the LTC3630 incorporates an overtemperature shutdown
feature. If the junction temperature reaches approximately
180°C, the LTC3630 will enter thermal shutdown mode.
Both power switches will be turned off and the SW node
will become high impedance. After the part has cooled
below 160°C, it will restart. The overtemperature level is
not production tested.
During sleep mode, the current sourced out of the ISET pin
is reduced to 1μA. The ISET current is increased back to 5μA
on the first switching cycle after exiting sleep mode. The
ISET current reduction in sleep mode, along with adding
a filtering capacitor, CISET, from the ISET pin to ground,
provides a method of reducing light load output voltage
ripple at the expense of lower efficiency and slightly degraded load step transient response.
For applications requiring higher output current, the
LTC3630 provides a feedback comparator output pin (FBO)
for combining the output current of multiple LTC3630s. By
connecting the FBO pin of a “master” LTC3630 to the VFB
pin of one or more “slave” LTC3630s, the output currents
can be combined to source much more than 500mA.
The LTC3630 can provide a programmable undervoltage
lockout which can also serve as a precise input voltage
monitor by using a resistive divider from VIN to GND with
the tap connected to the RUN pin. Switching is enabled
when the RUN pin voltage exceeds 1.21V and is disabled
when dropping below 1.1V. Pulling the RUN pin below
700mV forces a low quiescent current shutdown (5μA).
Furthermore, if the input voltage falls below 3.5V typical (3.7V maximum), an internal undervoltage detector
disables switching.
When switching is disabled, the LTC3630 can safely sustain input voltages up to the absolute maximum rating of
70V. Input supply undervoltage events trigger a soft-start
reset, which results in a graceful recovery from an input
supply transient.
3630fb
10
LTC3630
APPLICATIONS INFORMATION
The basic LTC3630 application circuit is shown on the front
page of the data sheet. External component selection is
determined by the maximum load current requirement and
begins with the selection of the peak current programming
resistor, RISET. The inductor value L can then be determined,
followed by capacitors CIN and COUT.
Peak Current Resistor Selection
The peak current comparator has a guaranteed maximum
current limit of 1A (1.2A typical), which guarantees a
maximum average current of 500mA. For applications
that demand less current, the peak current threshold can
be reduced to as little as 100mA (120mA typical). This
lower peak current allows the use of lower value, smaller
components (input capacitor, output capacitor, and inductor), resulting in lower input supply ripple and a smaller
overall DC/DC converter.
The threshold can be easily programmed using a resistor (RISET) between the ISET pin and ground. The voltage
generated on the ISET pin by RISET and the internal 5μA
current source sets the peak current. The voltage on the
ISET pin is internally limited within the range of 0.1V to
1.0V. The value of resistor for a particular peak current can
be selected by using Figure 2 or the following equation:
RISET = IPEAK • 0.2 • 106
where 100mA < IPEAK < 1A.
The internal 5μA current source is reduced to 1μA in sleep
mode to maximize efficiency and to facilitate a trade-off
The typical peak current is internally limited to be within the
range of 120mA to 1.2A. Shorting the ISET pin to ground
programs the current limit to 120mA, and leaving it float
sets the current limit to the maximum value of 1.2A. When
selecting this resistor value, be aware that the maximum
average output current for this architecture is limited to
half of the peak current. Therefore, be sure to select a value
that sets the peak current with enough margin to provide
adequate load current under all conditions. Selecting the
peak current to be 2.2 times greater than the maximum
load current is a good starting point for most applications.
Inductor Selection
The inductor, input voltage, output voltage, and peak current determine the switching frequency during a burst
cycle of the LTC3630. For a given input voltage, output
voltage, and peak current, the inductor value sets the
switching frequency during a burst cycle when the output
is in regulation. Generally, switching between 50kHz and
250kHz yields high efficiency, and 200kHz is a good first
choice for many applications. The inductor value can be
determined by the following equation:
⎛ V
⎞ ⎛ V
⎞
L = ⎜ OUT ⎟ • ⎜1– OUT ⎟
VIN ⎠
⎝ f •IPEAK ⎠ ⎝
The variation in switching frequency during a burst cycle
with input voltage and inductance is shown in Figure 3. For
lower values of IPEAK, multiply the frequency in Figure 3
by 1.2A/IPEAK.
220
200
180
160
RISET (kΩ)
between efficiency and light load output voltage ripple, as
described in the CISET Selection section of the Applications Information. For maximum efficiency, minimize the
capacitance on the ISET pin and place the RISET resistor
as close to the pin as possible.
140
120
An additional constraint on the inductor value is the
LTC3630’s 150ns minimum on-time of the high side switch.
Therefore, in order to keep the current in the inductor
100
80
60
40
20
0
50 100 150 200 250 300 350 400 450 500
MAXIMUM LOAD CURRENT (mA)
3630 F02
Figure 2. RISET Selection
3630fb
11
LTC3630
APPLICATIONS INFORMATION
600
VOUT = 3.3V
ISET OPEN
1000
500
INDUCTOR VALUE (μH)
SWITCHING FREQUENCY (kHz)
L = 4.2μH
400
300
L = 10μH
200
L = 22μH
100
0
L = 47μH
0
10
20
30
100
L = 100μH
40
50
60
VIN INPUT VOLTAGE (V)
10
100
1000
PEAK INDUCTOR CURRENT (mA)
3630 F04
3630 F03
Figure 3. Switching Frequency for VOUT = 3.3V
Figure 4. Recommended Inductor Values for Maximum Efficiency
well-controlled, the inductor value must be chosen so that
it is larger than a minimum value which can be computed
as follows:
in Figure 4. The values in this range are a good compromise
between the trade-offs discussed above. For applications
where board area is not a limiting factor, inductors with
larger cores can be used, which extends the recommended
range of Figure 4 to larger values.
L>
VIN(MAX) • tON(MIN)
• 1.2
IPEAK
where VIN(MAX) is the maximum input supply voltage when
switching is enabled, tON(MIN) is 150ns, IPEAK is the peak
current, and the factor of 1.2 accounts for typical inductor
tolerance and variation over temperature. Inductor values
that violate the above equation will cause the peak current
to overshoot and permanent damage to the part may occur.
Although the above equation provides the minimum inductor value, higher efficiency is generally achieved with
a larger inductor value, which produces a lower switching
frequency. The inductor value chosen should also be large
enough to keep the inductor current from going very negative which is more of a concern at higher VOUT (>~12V). For
a given inductor type, however, as inductance is increased,
DC resistance (DCR) also increases. Higher DCR translates into higher copper losses and lower current rating,
both of which place an upper limit on the inductance. The
recommended range of inductor values for small surface
mount inductors as a function of peak current is shown
Inductor Core Selection
Once the value for L is known, the type of inductor must
be selected. High efficiency converters generally cannot
afford the core loss found in low cost powdered iron cores,
forcing the use of the more expensive ferrite cores. Actual
core loss is independent of core size for a fixed inductor
value but is very dependent of the inductance selected.
As the inductance increases, core losses decrease. Unfortunately, increased inductance requires more turns of
wire and therefore copper losses will increase.
Ferrite designs have very low core losses and are preferred at high switching frequencies, so design goals
can concentrate on copper loss and preventing saturation. Ferrite core material saturates “hard,” which means
that inductance collapses abruptly when the peak design
current is exceeded. This results in an abrupt increase in
inductor ripple current and consequently output voltage
ripple. Do not allow the core to saturate!
3630fb
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LTC3630
APPLICATIONS INFORMATION
Different core materials and shapes will change the size/
current and price/current relationship of an inductor. Toroid
or shielded pot cores in ferrite or permalloy materials are
small and do not radiate energy but generally cost more
than powdered iron core inductors with similar characteristics. The choice of which style inductor to use mainly
depends on the price versus size requirements and any
radiated field/EMI requirements. New designs for surface
mount inductors are available from Coiltronics, Coilcraft,
TDK, Toko, and Sumida.
CIN and COUT Selection
The input capacitor, CIN, is needed to filter the trapezoidal
current at the source of the top high side MOSFET. CIN
should be sized to provide the energy required to charge
the inductor without causing a large decrease in input
voltage (ΔVIN). The relationship between CIN and ΔVIN
is given by:
CIN >
L •IPEAK 2
2 • VIN • ΔVIN
It is recommended to use a larger value for CIN than
calculated by the above equation since capacitance decreases with applied voltage. In general, a 4.7μF X7R
ceramic capacitor is a good choice for CIN in most LTC3630
applications.
To minimize large ripple voltage, a low ESR input capacitor sized for the maximum RMS current should be used.
RMS current is given by:
V
VIN
IRMS = IOUT(MAX) • OUT •
–1
VIN
VOUT
This formula has a maximum at VIN = 2VOUT, where IRMS =
IOUT/2. This simple worst-case condition is commonly used
for design because even significant deviations do not offer
much relief. Note that ripple current ratings from capacitor
manufacturers are often based only on 2000 hours of life
which makes it advisable to further derate the capacitor,
or choose a capacitor rated at a higher temperature than
required. Several capacitors may also be paralleled to meet
size or height requirements in the design.
The output capacitor, COUT, filters the inductor’s ripple
current and stores energy to satisfy the load current when
the LTC3630 is in sleep. The output ripple has a lower limit
of VOUT/160 due to the 5mV typical hysteresis of the feedback comparator. The time delay of the comparator adds
an additional ripple voltage that is a function of the load
current. During this delay time, the LTC3630 continues to
switch and supply current to the output. The output ripple
can be approximated by:
⎞ 4 • 10 –6 VOUT
⎛I
ΔVOUT ≈ ⎜ PEAK – ILOAD ⎟ •
+
⎠ COUT
⎝ 2
160
The output ripple is a maximum at no load and approaches
lower limit of VOUT/160 at full load. Choose the output
capacitor COUT to limit the output voltage ripple ΔVOUT
using the following equation:
• 2 • 10 –6
I
COUT ≥ PEAK
V
ΔVOUT – OUT
160
The value of the output capacitor must be large enough
to accept the energy stored in the inductor without a large
change in output voltage during a single switching cycle.
Setting this voltage step equal to 1% of the output voltage,
the output capacitor must be:
⎛I
⎞
COUT > 50 • L • ⎜ PEAK ⎟
⎝ VOUT ⎠
2
Typically, a capacitor that satisfies the voltage ripple requirement is adequate to filter the inductor ripple. To avoid
overheating, the output capacitor must also be sized to
handle the ripple current generated by the inductor. The
worst-case ripple current in the output capacitor is given
3630fb
13
LTC3630
APPLICATIONS INFORMATION
by IRMS = IPEAK/2. Multiple capacitors placed in parallel
may be needed to meet the ESR and RMS current handling
requirements.
Dry tantalum, special polymer, aluminum electrolytic,
and ceramic capacitors are all available in surface mount
packages. Special polymer capacitors offer very low ESR
but have lower capacitance density than other types.
Tantalum capacitors have the highest capacitance density
but it is important only to use types that have been surge
tested for use in switching power supplies. Aluminum
electrolytic capacitors have significantly higher ESR but
can be used in cost-sensitive applications provided that
consideration is given to ripple current ratings and longterm reliability. Ceramic capacitors have excellent low ESR
characteristics but can have high voltage coefficient and
audible piezoelectric effects. The high quality factor (Q)
of ceramic capacitors in series with trace inductance can
also lead to significant input voltage ringing.
Ceramic Capacitors and Audible Noise
Higher value, lower cost ceramic capacitors are now becoming available in smaller case sizes. Their high ripple
current, high voltage rating, and low ESR make them ideal
for switching regulator applications. However, care must
be taken when these capacitors are used at the input and
output. When a ceramic capacitor is used at the input and
the power is supplied by a wall adapter through long wires,
a load step at the output can induce ringing at the input,
VIN. At best, this ringing can couple to the output and be
mistaken as loop instability. At worst, a sudden inrush
of current through the long wires can potentially cause
a voltage spike at VIN large enough to damage the part.
For application with inductive source impedance, such as
a long wire, an electrolytic capacitor or a ceramic capacitor
with a series resistor may be required in parallel with CIN
to dampen the ringing of the input supply. Figure 5 shows
this circuit and the typical values required to dampen the
ringing.
-5$
VIN
LIN
R"
LIN
CIN
$IN
3630 F05
t$IN
Figure 5. Series RC to Reduce VIN Ringing
Ceramic capacitors are also piezoelectric sensitive. The
LTC3630’s burst frequency depends on the load current,
and in some applications at light load the LTC3630 can
excite the ceramic capacitor at audio frequencies, generating audible noise. If the noise is unacceptable, use
a high performance tantalum or electrolytic capacitor at
the output.
Output Voltage Programming
The LTC3630 has three fixed output voltage modes that
can be selected with the VPRG1 and VPRG2 pins and an
adjustable mode. The fixed output modes use an internal
feedback divider which enables higher efficiency, higher
noise immunity, and lower output voltage ripple for 5V,
3.3V and 1.8V applications. To select the fixed 5V output
voltage, connect VPRG1 to SS and VPRG2 to GND. For 3.3V,
connect VPRG1 to GND and VPRG2 to SS. For 1.8V, connect
both VPRG1 and VPRG2 to SS. For any of the fixed output
voltage options, directly connect the VFB pin to VOUT.
For the adjustable output mode (VPRG1 = 0V, VPRG2 = 0V),
the output voltage is set by an external resistive divider
according to the following equation:
⎛ R1 ⎞
VOUT = 0.8V • ⎜1+ ⎟
⎝ R2 ⎠
3630fb
14
LTC3630
APPLICATIONS INFORMATION
The resistive divider allows the VFB pin to sense a fraction
of the output voltage as shown in Figure 6. The output
voltage can range from 0.8V to VIN. Be careful to keep the
divider resistors very close to the VFB pin to minimize the
trace length and noise pick-up on the sensitive VFB signal.
VOUT
LTC3630
R1
VFB
5V
4.2M
R2
0.8V
800k
SS
VPRG1
VPRG2
VOUT
R1
VFB
LTC3630
VPRG1
VPRG2
0.8V
3630 F07
R2
3630 F06
Figure 6. Setting the Output Voltage with External Resistors
To minimize the no-load supply current, resistor values in
the megohm range may be used; however, large resistor
values should be used with caution. The feedback divider
is the only load current when in shutdown. If PCB leakage
current to the output node or switch node exceeds the load
current, the output voltage will be pulled up. In normal
operation, this is generally a minor concern since the load
current is much greater than the leakage.
To avoid excessively large values of R1 in high output voltage applications (VOUT ≥ 10V), a combination of external
and internal resistors can be used to set the output voltage. This has an additional benefit of increasing the noise
immunity on the VFB pin. Figure 7 shows the LTC3630
with the VFB pin configured for a 5V fixed output with an
external divider to generate a higher output voltage. The
internal 5M resistance appears in parallel with R2, and the
value of R2 must be adjusted accordingly. R2 should be
chosen to be less than 200k to keep the output voltage
variation less than 1% due to the tolerance of the LTC3630’s
internal resistor.
Figure 7. Setting the Output Voltage with
External and Internal Resistors
RUN Pin and External Input Undervoltage Lockout
The RUN pin has two different threshold voltage levels.
Pulling the RUN pin below 0.7V puts the LTC3630 into a
low quiescent current shutdown mode (IQ ~ 5μA). When
the RUN pin is greater than 1.21V, the controller is enabled.
Figure 8 shows examples of configurations for driving the
RUN pin from logic.
SUPPLY
LTC3630
RUN
LTC3630
RUN
3630 F08
Figure 8. RUN Pin Interface to Logic
The RUN pin can alternatively be configured as a precise
undervoltage (UVLO) lockout on the VIN supply with a
resistive divider from VIN to ground. A simple resistive
divider can be used as shown in Figure 9 to meet specific
VIN voltage requirements.
5V
LTC3630
2M
SLEEP, ACTIVE: 2μA
SHUTDOWN: 0μA
VIN
R3
RUN
R4
3630 F09
Figure 9. Adjustable UV Lockout
3630fb
15
LTC3630
APPLICATIONS INFORMATION
The current that flows through the R3-R4 divider will
directly add to the shutdown, sleep, and active current
of the LTC3630, and care should be taken to minimize
the impact of this current on the overall efficiency of the
application circuit. To keep the variation of the rising VIN
UVLO threshold to less than 5% due to the internal pullup circuitry, the following equations should be used to
calculate R3 and R4:
RisingVIN UVLOThreshold
R3 ≤
40μA
R4 =
R3 • 1.21V
RisingVIN UVLOThreshold – 1.21V +R3 • 4μA
The falling UVLO threshold will be about 10% lower than
the rising VIN UVLO threshold due to the 110mV hysteresis
of the RUN comparator.
For applications that do not require a precise UVLO, the
RUN pin can be left floating. In this configuration, the UVLO
threshold is limited to the internal VIN UVLO thresholds as
shown in the Electrical Characteristics table.
Be aware that the RUN pin cannot be allowed to exceed
its absolute maximum rating of 6V. To keep the voltage
on the RUN pin from exceeding 6V, the following relation
should be satisfied:
VIN(MAX) < 4.5 • Rising VIN UVLO Threshold
To support a VIN(MAX) greater than 4.5x the external UVLO
threshold, an external 4.7V Zener diode should be used
in parallel with R4. See Figure 11.
Soft-Start
Soft-start is implemented by ramping the effective reference voltage from 0V to 0.8V. To increase the duration of
soft-start, place a capacitor from the SS pin to ground.
An internal 5μA pull-up current will charge this capacitor.
The value of the soft-start capacitor can be calculated by
the following equation:
CSS = Soft-Start Time •
The minimum soft-start time is limited to the internal softstart timer of 0.8ms. When the LTC3630 detects a fault
condition (input supply undervoltage or overtemperature)
or when the RUN pin falls below 1.1V, the SS pin is quickly
pulled to ground and the internal soft-start timer is reset.
This ensures an orderly restart when using an external
soft-start capacitor.
Note that the soft-start capacitor may not be the limiting
factor in the output voltage ramp. The maximum output
current, which is equal to half the peak current, must
charge the output capacitor from 0V to its regulated value.
For small peak currents or large output capacitors, this
ramp time can be significant. Therefore, the output voltage
ramp time from 0V to the regulated VOUT value is limited
to a minimum of:
Ramp Time ≥
2 • COUT
VOUT
IPEAK
CISET Selection
Once the peak current resistor, RISET, and inductor are selected to meet the load current and frequency requirements,
an optional capacitor, CISET, can be added in parallel with
RISET. This will boost efficiency at mid-loads and reduce
the output voltage ripple dependency on load current at the
expense of slightly degraded load step transient response.
The peak inductor current is controlled by the voltage on
the ISET pin. Current out of the ISET pin is 5μA while the
LTC3630 is switching and is reduced to 1μA during sleep
mode. The ISET current will return to 5μA on the first cycle
after sleep mode. Placing a parallel RC from the ISET pin to
ground filters the ISET voltage as the LTC3630 enters and
exits sleep mode which in turn will affect the output voltage ripple, efficiency and load step transient performance.
In general, when RISET is greater than 120k a CISET capacitor in the 100pF to 200pF range will improve most
performance parameters. When RISET is less than 100k,
the capacitance on the ISET pin should be minimized.
5μA
0.35V
3630fb
16
LTC3630
APPLICATIONS INFORMATION
Higher Current Applications
Efficiency Considerations
For applications that require more than 500mA, the
LTC3630 provides a feedback comparator output pin
(FBO) for driving additional LTC3630s. When the FBO pin
of a “master” LTC3630 is connected to the VFB pin of one
or more “slave” LTC3630s, the master controls the burst
cycle of the slaves.
The efficiency of a switching regulator is equal to the output
power divided by the input power times 100%. It is often
useful to analyze individual losses to determine what is
limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
Figure 10 shows an example of a 5V, 1A regulator using
two LTC3630s. The master is configured for a 5V fixed
output with external soft-start and the VIN UVLO level is
set by the RUN pin. Since the slaves are directly controlled
by the master, the SS pin of the slave should have minimal
capacitance and the RUN pin of the slave should be floating.
Furthermore, slaves should be configured for a 1.8V fixed
output (VPRG1 = VPRG2 = SS) to set the VFB pin threshold at
1.8V. The inductors L1 and L2 do not necessarily have to
be the same, but should both meet the criteria described
above in the Inductor Selection section.
VOUT
5V
COUT 1A
L1
VIN
CIN
R3
R4
SW
VIN
LTC3630
(MASTER)
VFB
RUN
SS
VPRG1
VPRG2
ISET
FBO
VFB
VIN
LTC3630
(SLAVE)
SW
RUN
SS
VPRG1
VPRG2
ISET
CSS
L2
FBO
3630 F10
Figure 10. 5V, 1A Regulator
Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage of input power.
Although all dissipative elements in the circuit produce
losses, two main sources usually account for most of
the losses: VIN operating current and I2R losses. The VIN
operating current dominates the efficiency loss at very
low load currents whereas the I2R loss dominates the
efficiency loss at medium to high load currents.
1. The VIN operating current comprises two components:
The DC supply current as given in the electrical characteristics and the internal MOSFET gate charge currents.
The gate charge current results from switching the gate
capacitance of the internal power MOSFET switches.
Each time the gate is switched from high to low to
high again, a packet of charge, ΔQ, moves from VIN to
ground. The resulting ΔQ/dt is the current out of VIN
that is typically larger than the DC bias current.
2. I2R losses are calculated from the resistances of the
internal switches, RSW and external inductor RL. When
switching, the average output current flowing through
the inductor is “chopped” between the high side PMOS
switch and the low side NMOS switch. Thus, the series
resistance looking back into the switch pin is a function
of the top and bottom switch RDS(ON) values and the
duty cycle (DC = VOUT/VIN) as follows:
RSW = (RDS(ON)TOP)DC + (RDS(ON)BOT) • (1 – DC)
The RDS(ON) for both the top and bottom MOSFETs can
be obtained from the Typical Performance Characteristics curves. Thus, to obtain the I2R losses, simply add
3630fb
17
LTC3630
APPLICATIONS INFORMATION
RSW to RL and multiply the result by the square of the
average output current:
I2R Loss = IO2(RSW + RL)
Other losses, including CIN and COUT ESR dissipative
losses and inductor core losses, generally account for
less than 2% of the total power loss.
Thermal Considerations
In most applications, the LTC3630 does not dissipate much
heat due to its high efficiency. But, in applications where
the LTC3630 is running at high ambient temperature with
low supply voltage and high duty cycles, such as dropout,
the heat dissipated may exceed the maximum junction
temperature of the part.
To prevent the LTC3630 from exceeding the maximum
junction temperature, the user will need to do some thermal
analysis. The goal of the thermal analysis is to determine
whether the power dissipated exceeds the maximum junction temperature of the part. The temperature rise from
ambient to junction is given by:
TR = PD • θJA
where PD is the power dissipated by the regulator and θJA
is the thermal resistance from the junction of the die to
the ambient temperature.
The junction temperature is given by:
TJ = TA + TR
Generally, the worst-case power dissipation is in dropout
at low input voltage. In dropout, the LTC3630 can provide
a DC current as high as the full 1.2A peak current to the
output. At low input voltage, this current flows through a
higher resistance MOSFET, which dissipates more power.
As an example, consider the LTC3630 in dropout at an input
voltage of 5V, a load current of 500mA and an ambient
temperature of 85°C. From the Typical Performance graphs
of Switch On-Resistance, the RDS(ON) of the top switch
at VIN = 5V and 100°C is approximately 1.9Ω. Therefore,
the power dissipated by the part is:
For the MSOP package the θJA is 45°C/W. Thus, the junction temperature of the regulator is:
45°C
= 106.4°C
W
which is below the maximum junction temperature of
150°C.
TJ = 85°C + 0.475W •
Note that the while the LTC3630 is in dropout, it can provide
output current that is equal to the peak current of the part.
This can increase the chip power dissipation dramatically
and may cause the internal overtemperature protection
circuitry to trigger at 180°C and shut down the LTC3630.
Design Example
As a design example, consider using the LTC3630 in an
application with the following specifications: VIN = 24V,
VIN(MAX) = 70V, VOUT = 3.3V, IOUT = 500mA, f = 200kHz.
Furthermore, assume for this example that switching
should start when VIN is greater than 12V.
First, calculate the inductor value that gives the required
switching frequency:
⎛
⎞ ⎛ 3.3V ⎞
3.3V
L =⎜
⎟ • ⎜1–
⎟ ≅ 10μH
⎝ 200kHz • 1.2A ⎠ ⎝ 24V ⎠
Next, verify that this value meets the LMIN requirement.
For this input voltage and peak current, the minimum
inductor value is:
LMIN =
24V • 150ns
≅ 3μH
1.2A
Therefore, the minimum inductor requirement is satisfied
and the 10μH inductor value may be used.
Next, CIN and COUT are selected. For this design, CIN should
be sized for a current rating of at least:
IRMS = 500mA •
3.3V
24V
•
– 1 ≅ 175mARMS
24V
3.3V
PD = (ILOAD)2 • RDS(ON) = (500mA)2 • 1.9Ω = 0.475W
3630fb
18
LTC3630
APPLICATIONS INFORMATION
The value of CIN is selected to keep the input from drooping less than 240mV (1%):
CIN >
10μH • 1.2A 2
≅ 2.2μF
2 • 24V • 240mV
COUT will be selected based on a value large enough to
satisfy the output voltage ripple requirement. For a 50mV
output ripple, the value of the output capacitor can be
calculated from:
COUT >
10μH • 1.2A 2
≅ 47μF
2 • 3.3V • 50mV
COUT also needs an ESR that will satisfy the output voltage
ripple requirement. The required ESR can be calculated
from:
ESR <
50mV
≅ 40mΩ
1.2A
A 47μF ceramic capacitor has significantly less ESR than
40mΩ.
The ISET pin should be left open in this example to select
maximum peak current (1.2A typical). Figure 11 shows a
complete schematic for this design example.
10μH
VIN
24V
200k
2.2μF
4.7V
21k
VIN
SW
LTC3630
VFB
RUN
SS
VPRG2
VPRG1
FBO
ISET
GND
VOUT
3.3V
500mA
47μF
3630 F11
Figure 11. 24V to 3.3V, 500mA Regulator at 200kHz
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of
the LTC3630. Check the following in your layout:
Since an output voltage of 3.3V is one of the standard
output configurations, the LTC3630 can be configured
by connecting VPRG1 to ground and VPRG2 to the SS pin.
1. Large switched currents flow in the power switches
and input capacitor. The loop formed by these components should be as small as possible. A ground plane
is recommended to minimize ground impedance.
The undervoltage lockout requirement on VIN can be satisfied with a resistive divider from VIN to the RUN pin (refer
to Figure 9). Calculate R3 and R4 as follows:
2. Connect the (+) terminal of the input capacitor, CIN, as
close as possible to the VIN pin. This capacitor provides
the AC current into the internal power MOSFETs.
12V
40μA
3. Keep the switching node, SW, away from all sensitive
small signal nodes. The rapid transitions on the switching
node can couple to high impedance nodes, in particular
VFB, and create increased output ripple.
R3 = 200k which is ≤
R4 =
200k • 1.21V
= 20.9k
12V – 1.21V + 200k • 4μA
Choose standard values for R3 = 200k, R4 = 21k. Note
that the VIN falling threshold will be 10% less than the
rising threshold or 11V.
Since the maximum VIN is more than 4.5x the UVLO threshold, a 4.7V Zener diode in parallel with R4 is required to
keep the maximum voltage on the RUN pin less than the
absolute maximum of 6V.
4. Flood all unused area on all layers with copper except
for the area under the inductor. Flooding with copper
will reduce the temperature rise of power components.
You can connect the copper areas to any DC net (VIN,
VOUT, GND, or any other DC rail in your system).
3630fb
19
LTC3630
APPLICATIONS INFORMATION
Pin Clearance/Creepage Considerations
The LTC3630 is available in two packages (MSE16 and
DHC) both with identical functionality. However, the 0.2mm
(minimum space) between pins and paddle on the DHCpackage may not provide sufficient PC board trace clearance
between high and low voltage pins in some higher voltage
applications. In applications where clearance is required,
the MSE16 package should be used. The MSE16 package
has removed pins between all the adjacent high voltage
and low voltage pins, providing 0.657mm clearance which
will be sufficient for most applications. For more information, refer to the printed circuit board design standards
described in IPC-2221 (www.ipc.org).
L1
VIN
VIN
VOUT
SW
R1
R3
RUN
R4
VFB
LTC3630
RISET
VIN
5V TO 65V
L1
33μH
CIN
4.7μF
SW
VIN
LTC3630
VFB
RUN
R2
ISET
ISET
CISET
CIN
COUT
SS
FBO
COUT
100μF
w2
RISET
220k
VOUT
5V
500mA
CISET
100pF
VPRG1
VPRG2
GND
CSS
3630 F13
FBO
SS
VPRG2
VPRG1
CIN: TDK C5750X7R2A-475M (2220)
COUT: 2 wAVX 1812D107MAT
L1: SUMIDA CDRH105RNP-330N
Figure 13. 5V to 65V Input to 5V Output,
High Efficiency, 500mA Regulator
GND
L1
VOUT
CIN
COUT
VIN
GND
VIAS TO GROUND PLANE
OUTLINE OF LOCAL GROUND PLANE
3630 F12
Figure 12. Example PCB Layout
3630fb
20
LTC3630
TYPICAL APPLICATIONS
4V to 24V Input to 3.3V Output,
250mA Regulator with External Soft-Start, Small Size
COUT
10μF
CSS
100nF
RISET
100k
VOUT = 3.3V
90
VOUT
3.3V
250mA
EFFICIENCY
80
1000
70
60
100
50
POWER
40
10
30
GND
POWER LOSS (mW)
CIN
2.2μF
VIN
SW
LTC3630
VFB
RUN
ISET
SS
FBO VPRG1
VPRG2
100
EFFICIENCY (%)
VIN
4V TO 24V
L1
10μH
Efficiency and Power Loss vs Load Current
3630 TA02a
1
20
10
CIN: MURATA GRM42-2X7R225K25D500
COUT: KEMET C1206C206K9PAC
L1: VISHAY IHLP2020BZ-100M-11
0
0
0.1
10
1
LOAD CURRENT (mA)
100
3630 TA02b
4V to 53V Input to –12V Output, Positive-to-Negative Converter
500
L1
22μH
CIN
4.7μF
100V
VIN
SW
LTC3630
RUN
VFB
ISET
FBO
SS
VPRG2
VPRG1
R1
200k
R2
147k
COUT
22μF
GND
CIN: KEMET C1210C475K5RAC
COUT: TDK C4532X7R1C226M
L1: COILCRAFT MSS1048-223ML
3630 TA03a
VOUT
–12V
MAXIMUM LOAD CURRENT (mA)
VIN
4V TO 53V
Maximum Load Current vs Input Voltage
VOUT = –12V
400
300
200
100
0
5
10
15
20 25 30 35 40
INPUT VOLTAGE (V)
45
50
3630 TA03b
3630fb
21
LTC3630
TYPICAL APPLICATIONS
5V to 65V Input to 5V Output,150mA Regulator
with 20kHz Minimum Burst Frequency
L1
33μH
CIN
2.2μF
VIN
SW
RUN
RISET
60.4k
COUT
10μF
LTC3630
VFB
30.1Ω
SS
ISET
VPRG2 VPRG1
FBO
IN
GND
OUT
LTC6994-1
V+
976k
CIN: TDK C3225X7R2AA225M
COUT: AVX 12063D106KAT
L1: COOPER BUSSMAN SD25-330
SET
196k
60
VOUT
5V
150mA
VIN = 3.3V
50
BURST FREQUENCY (kHz)
VIN
5V TO 65V
Burst Frequency vs Load Current
40
30
20kHz LIMIT
20
10
NO LIMIT
DIV
GND
100k
0
0
3630 TA04
1
10
LOAD CURRENT (mA)
100
3630 TA04b
12V to 65V Input to 12V Output with 100mA Input Current Limit
L1
22μH
CIN
2.2μF
VIN
R3
806k
R4
10k
500
SW
LTC3630
RUN
VFB
ISET
FBO
SS
VPRG1
VPRG2
R1
200k
R2
14.3k
GND
3630 TA05
VOUT
COUT 12V
22μF
MAXIMUM CURRENT (mA)
VIN
12V TO 65V
Maximum Input and Load Current vs Input Voltage
400
MAXIMUM LOAD CURRENT
300
200
MAXIMUM INPUT CURRENT
100
CIN: TDK C3225X7R2A225M
COUT: TAIYO YUDEN EMK316BJ226ML-T
L1:TDK SLF7045470MR75
0
10 15 20 25 30 35 40 45 50 55 60 65
INPUT VOLTAGE (V)
3630 TA05b
INPUT CURRENT LIMIT ~
VOUT
R4
s
2 R3R4
MAXIMUM LOAD CURRENT ~
VIN
R4
s
2 R3R4
3630fb
22
LTC3630
PACKAGE DESCRIPTION
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
MSE Package
Variation: MSE16 (12)
16-Lead Plastic MSOP with 4 Pins Removed
Exposed Die Pad
(Reference LTC DWG # 05-08-1871 Rev C)
BOTTOM VIEW OF
EXPOSED PAD OPTION
2.845 t 0.102
(.112 t .004)
5.23
(.206)
MIN
2.845 t 0.102
(.112 t .004)
0.889 t 0.127
(.035 t .005)
8
1
1.651 t 0.102
(.065 t .004)
1.651 t 0.102 3.20 – 3.45
(.065 t .004) (.126 – .136)
16
0.305 t 0.038
(.0120 t .0015)
TYP
0.50
(.0197)
1.0 BSC
(.039)
BSC
RECOMMENDED SOLDER PAD LAYOUT
0.254
(.010)
0.35
REF
4.039 t 0.102
(.159 t .004)
(NOTE 3)
0.12 REF
DETAIL “B”
CORNER TAIL IS PART OF
DETAIL “B” THE LEADFRAME FEATURE.
FOR REFERENCE ONLY
9
NO MEASUREMENT PURPOSE
0.280 t 0.076
(.011 t .003)
REF
16 14 121110 9
DETAIL “A”
0s – 6s TYP
3.00 t 0.102
(.118 t .004)
(NOTE 4)
4.90 t 0.152
(.193 t .006)
GAUGE PLANE
0.53 t 0.152
(.021 t .006)
1
DETAIL “A”
1.10
(.043)
MAX
0.18
(.007)
SEATING
PLANE
0.17 – 0.27
(.007 – .011)
TYP
3 567 8
1.0
(.039)
BSC
0.50
NOTE:
(.0197)
1. DIMENSIONS IN MILLIMETER/(INCH)
BSC
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL
NOT EXCEED 0.254mm (.010") PER SIDE.
0.86
(.034)
REF
0.1016 t 0.0508
(.004 t .002)
MSOP (MSE16(12)) 0911 REV C
3630fb
23
LTC3630
PACKAGE DESCRIPTION
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
DHC Package
16-Lead Plastic DFN (5mm w 3mm)
(Reference LTC DWG # 05-08-1706 Rev Ø)
0.65 t0.05
3.50 t0.05
1.65 t0.05
2.20 t0.05 (2 SIDES)
PACKAGE
OUTLINE
0.25 t 0.05
0.50 BSC
4.40 t0.05
(2 SIDES)
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
R = 0.115
TYP
5.00 t0.10
(2 SIDES)
R = 0.20
TYP
3.00 t0.10
(2 SIDES)
9
0.40 t0.10
16
1.65 t0.10
(2 SIDES)
PIN 1
TOP MARK
(SEE NOTE 6)
PIN 1
NOTCH
(DHC16) DFN 1103
8
0.200 REF
1
0.25 t0.05
0.50 BSC
0.75 t0.05
4.40 t0.10
(2 SIDES)
0.00 – 0.05
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING PROPOSED TO BE MADE VARIATION OF VERSION (WJED-1) IN JEDEC
PACKAGE OUTLINE MO-229
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE
TOP AND BOTTOM OF PACKAGE
3630fb
24
LTC3630
REVISION HISTORY
REV
DATE
DESCRIPTION
PAGE NUMBER
A
5/12
Circuit 3630 TA05: change 36V to 12V
22
B
6/12
Clarified Typical Application
26
3630fb
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
25
LTC3630
TYPICAL APPLICATION
4.5V to 65V Input to 3.3V Output, 1.5A Regulator
VX
22μF
SYNC/MODE INTVCC
PVIN
PGOOD
ROSC
105k
VIN*
4.5V TO 65V
L1
33μH
CIN
4.7μF
VIN
PVIN
RT
CITH
1nF RITH
4.32k
R1
200k
RUN
VFB
ISET
FBO
SS
VPRG1
VPRG2
R2
102k
R1
105k
R2
475k
CFB
10pF
CVCC
1μF
BOOST
CBST
0.22μF
LTC3603
SW
LTC3630
D1
ITH
SW
VFB
SW
RUN
SW
TRACK/SS
PGND
PGND
PGND
L1
2.2μH
VOUT
3.3V
1.5A
COUT
100μF
GND
3630 TA06
CIN: MURATA GCM32DR72A225KA64L
C1: TDK CGA6P1X7R1C226M
L1: COILCRAFT MSS1048T-333
COUT: TDK C3225X5ROJ107M
L2: VISHAY IHLP-2525CZ-01
*WHEN VIN > 15V, LTC3630 SWITCHES AND VX IS REGULATED TO 15V; WHEN
VIN < 15V, LTC3630 OPERATES IN DROPOUT AND VX FOLLOWS VIN
RELATED PARTS
PART NUMBER DESCRIPTION
COMMENTS
LTC3642
45V (Transient to 60V) 50mA Synchronous Step-Down
DC/DC Converter
VIN: 4.5V to 45V, VOUT(MIN) = 0.8V, IQ = 12μA, ISD = 3μA,
3 × 3 DFN-8, MSOP-8
LTC3631
45V (Transient to 60V) 100mA Synchronous Step-Down
DC/DC Converter
VIN: 4.5V to 45V, VOUT(MIN) = 0.8V, IQ = 12μA, ISD = 3μA,
3 × 3 DFN-8, MSOP-8
LTC3632
50V (Transient to 60V) 20mA Synchronous Step-Down
DC/DC Converter
VIN: 4.5V to 50V, VOUT(MIN) = 0.8V, IQ = 12μA, ISD = 3μA,
3 × 3 DFN-8, MSOP-8
LTC3103
15V, 300mA Synchronous Step-Down DC/DC Converter
with Ultralow Quiescent Current
VIN: 2.5V to 15V, VOUT(MIN) = 0.6V, IQ = 1.8μA, ISD = 1μA,
3 × 3 DFN-10, MSOP-10E
LTC3104
15V, 300mA Synchronous Step-Down DC/DC Converter
with Ultralow Quiescent Current and 10mA LDO
VIN: 2.5V to 15V, VOUT(MIN) = 0.6V, IQ = 2.6μA, ISD = 1μA,
4 × 3 DFN-14, MSOP-16E
LT3970
40V, 350mA, 2.2MHz High Efficiency Micropower Step-Down
DC/DC Converter with IQ = 2.5μA
VIN: 4.2V to 40V, VOUT(MIN) = 1.21V, IQ = 2.5μA, ISD < 1μA,
3 × 2 DFN-10, MSOP-10
LT3990
62V, 350mA, 2.2MHz High Efficiency Micropower Step-Down
DC/DC Converter with IQ = 2.5μA
VIN: 4.2V to 62V, VOUT(MIN) = 1.21V, IQ = 2.5μA, ISD < 1μA,
3 × 3 DFN-10, MSOP-16E
LT3971
38V, 1.2A, 2.2MHz High Efficiency Micropower Step-Down
DC/DC Converter with IQ = 2.8μA
VIN: 4.3V to 38V, VOUT(MIN) = 1.19V, IQ = 2.8μA, ISD < 1μA,
3 × 3 DFN-10, MSOP-10E
LT3991
55V, 1.2A, 2.2MHz High Efficiency Micropower Step-Down
DC/DC Converter with IQ = 2.8μA
VIN: 4.3V to 55V, VOUT(MIN) = 1.19V, IQ = 2.8μA, ISD < 1μA,
3 × 3 DFN-10, MSOP-10E
LT3682
36V, 60VMAX, 1A, 2.2MHz High Efficiency Micropower
Step-Down DC/DC Converter
VIN: 3.6V to 36V, VOUT(MIN) = 0.8V, IQ = 75μA, ISD < 1μA,
3 × 3 DFN-12
LTC3891
Low IQ, 60V Synchronous Step-Down Controller
VIN: 4V to 60V, VOUT(MIN) = 0.8V, IQ = 50μA, ISD = 14μA,
3 × 4 QFN-20, TSSOP-20E
3630fb
26 Linear Technology Corporation
LT 0612 REV B • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
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