AD AD603-EVALZ Low noise, 90 mhz variable gain amplifier Datasheet

Low Noise, 90 MHz
Variable Gain Amplifier
AD603
The decibel gain is linear in dB, accurately calibrated, and stable
over temperature and supply. The gain is controlled at a high
impedance (50 MΩ), low bias (200 nA) differential input; the
scaling is 25 mV/dB, requiring a gain control voltage of only
1 V to span the central 40 dB of the gain range. An overrange
and underrange of 1 dB is provided whatever the selected range.
The gain control response time is less than 1 μs for a 40 dB change.
FEATURES
Linear-in-dB gain control
Pin-programmable gain ranges
−11 dB to +31 dB with 90 MHz bandwidth
9 dB to 51 dB with 9 MHz bandwidth
Any intermediate range, for example −1 dB to +41 dB
with 30 MHz bandwidth
Bandwidth independent of variable gain
1.3 nV/√Hz input noise spectral density
±0.5 dB typical gain accuracy
The differential gain control interface allows the use of either
differential or single-ended positive or negative control voltages.
Several of these amplifiers may be cascaded and their gain
control gains offset to optimize the system SNR.
The AD603 can drive a load impedance as low as 100 Ω with
low distortion. For a 500 Ω load in shunt with 5 pF, the total
harmonic distortion for a ±1 V sinusoidal output at 10 MHz is
typically −60 dBc. The peak specified output is ±2.5 V minimum
into a 500 Ω load.
APPLICATIONS
RF/IF AGC amplifiers
Video gain controls
A/D range extensions
Signal measurements
The AD603 uses a patented proprietary circuit topology—the
X-AMP®. The X-AMP comprises a variable attenuator of 0 dB
to −42.14 dB followed by a fixed-gain amplifier. Because of the
attenuator, the amplifier never has to cope with large inputs and
can use negative feedback to define its (fixed) gain and dynamic
performance. The attenuator has an input resistance of 100 Ω,
laser trimmed to ±3%, and comprises a 7-stage R-2R ladder
network, resulting in an attenuation between tap points of
6.021 dB. A proprietary interpolation technique provides a
continuous gain control function that is linear in dB.
GENERAL DESCRIPTION
The AD603 is a low noise, voltage-controlled amplifier for use
in RF and IF AGC systems. It provides accurate, pin-selectable
gains of −11 dB to +31 dB with a bandwidth of 90 MHz or +9 dB to
51+ dB with a bandwidth of 9 MHz. Any intermediate gain
range may be arranged using one external resistor. The input
referred noise spectral density is only 1.3 nV/√Hz, and power
consumption is 125 mW at the recommended ±5 V supplies.
The AD603 is specified for operation from −40°C to +85°C.
FUNCTIONAL BLOCK DIAGRAM
VPOS 8
SCALING
REFERENCE
VNEG 6
PRECISION PASSIVE
INPUT ATTENUATOR
FIXED-GAIN
AMPLIFIER
GPOS 1
7
VOUT
5
FDBK
VG
GNEG
2
6.44kΩ*
AD603
GAINCONTROL
INTERFACE
694Ω*
0dB
VINP 3
–6.02dB
R
2R
–12.04dB –18.06dB –24.08dB
R
2R
R
2R
R
2R
–30.1dB
R
2R
–36.12dB –42.14dB
R
2R
R
R
20Ω*
COMM 4
00539-001
R-2R LADDER NETWORK
*NOMINAL VALUES.
Figure 1.
Rev. H
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
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Fax: 781.461.3113
©2007 Analog Devices, Inc. All rights reserved.
AD603
TABLE OF CONTENTS
Features .............................................................................................. 1
Applications....................................................................................... 1
Programming the Fixed-Gain Amplifier Using
Pin Strapping............................................................................... 12
General Description ......................................................................... 1
Using the AD603 in Cascade ........................................................ 14
Functional Block Diagram .............................................................. 1
Sequential Mode (Optimal SNR) ............................................. 14
Revision History ............................................................................... 2
Parallel Mode (Simplest Gain Control Interface) .................. 16
Specifications..................................................................................... 3
Low Gain Ripple Mode (Minimum Gain Error) ................... 16
Absolute Maximum Ratings............................................................ 4
Applications Information .............................................................. 17
ESD Caution.................................................................................. 4
A Low Noise AGC Amplifier.................................................... 17
Pin Configurations and Function Descriptions ........................... 5
Caution ........................................................................................ 18
Typical Performance Characteristics ............................................. 6
Evaluation Board ............................................................................ 19
Theory of Operation ...................................................................... 11
Outline Dimensions ....................................................................... 21
Noise Performance ..................................................................... 11
Ordering Guide .......................................................................... 21
The Gain Control Interface....................................................... 12
REVISION HISTORY
5/07—Rev. G to Rev. H
Changes to Layout ...........................................................................14
Changes to Layout ...........................................................................15
Changes to Layout ...........................................................................16
Inserted Evaluation Board Section, and Figure 48 to
Figure 51 ...........................................................................................19
Inserted Figure 52 and Table 4.......................................................20
Changes to Ordering Guide ...........................................................21
3/05—Rev. F to Rev. G
Updated Format.................................................................. Universal
Change to Features ............................................................................1
Changes to General Description .....................................................1
Change to Figure 1 ............................................................................1
Changes to Specifications .................................................................3
New Figure 4 and Renumbering Subsequent Figures...................6
Change to Figure 10 ..........................................................................7
Change to Figure 23 ..........................................................................9
Change to Figure 29 ........................................................................12
Updated Outline Dimensions ........................................................20
4/04—Rev. E to Rev. F
Changes to Specifications.................................................................2
Changes to Ordering Guide .............................................................3
8/03—Rev. D to Rev E
Updated Format.................................................................. Universal
Changes to Specifications.................................................................2
Changes to TPCs 2, 3, 4 ....................................................................4
Changes to Sequential Mode (Optimal S/N Ratio) section.........9
Change to Figure 8 ..........................................................................10
Updated Outline Dimensions........................................................14
Rev. H | Page 2 of 24
AD603
SPECIFICATIONS
@ TA = 25°C, VS = ±5 V, –500 mV ≤ VG ≤ +500 mV, GNEG = 0 V, –10 dB to +30 dB gain range, RL = 500 Ω, and CL = 5 pF, unless
otherwise noted.
Table 1.
Parameter
INPUT CHARACTERISTICS
Input Resistance
Input Capacitance
Input Noise Spectral Density 1
Noise Figure
1 dB Compression Point
Peak Input Voltage
OUTPUT CHARACTERISTICS
−3 dB Bandwidth
Slew Rate
Peak Output 2
Output Impedance
Output Short-Circuit Current
Group Delay Change vs. Gain
Group Delay Change vs. Frequency
Differential Gain
Differential Phase
Total Harmonic Distortion
Third-Order Intercept
ACCURACY
Gain Accuracy, f = 100 kHz; Gain (dB) = (40 VG + 10) dB
TMIN to TMAX
Gain, f = 10.7 MHz
Output Offset Voltage 3
TMIN to TMAX
Output Offset Variation vs. VG
TMIN to TMAX
GAIN CONTROL INTERFACE
Gain Scaling Factor
TMIN to TMAX
Conditions
Min
Typ
Max
Unit
Pin 3 to Pin 4
97
100
2
1.3
8.8
−11
±1.4
103
Ω
pF
nV/√Hz
dB
dBm
V
Input short-circuited
f = 10 MHz, gain = maximum, RS = 10 Ω
f = 10 MHz, gain = maximum, RS = 10 Ω
VOUT = 100 mV rms
RL ≥ 500 Ω
RL ≥ 500 Ω
f ≤ 10 MHz
f = 3 MHz; full gain range
VG = 0 V; f = 1 MHz to 10 MHz
f = 10 MHz, VOUT = 1 V rms
f = 40 MHz, gain = maximum, RS = 50 Ω
−500 mV ≤ VG ≤ +500 mV
VG = -0.5 V
VG = 0.0 V
VG = 0.5 V
VG = 0 V
−500 mV ≤ VG ≤ +500 mV
100 kHz
10.7 MHz
GNEG, GPOS Voltage Range 4
Input Bias Current
Input Offset Current
Differential Input Resistance
Response Rate
POWER SUPPLY
Specified Operating Range
Quiescent Current
TMIN to TMAX
±2.5
−1
−1.5
−10.3
+9.5
+29.3
−20
−30
−20
−30
39.4
38
38.7
−1.2
90
275
±3.0
2
50
±2
±2
0.2
0.2
−60
15
±0.5
−9.0
+10.5
+30.3
40
39.3
MHz
V/μs
V
Ω
mA
ns
ns
%
Degree
dBc
dBm
+1
+1.5
−8.0
+11.5
+31.3
+20
+30
+20
+30
dB
dB
dB
dB
dB
mV
mV
mV
mV
40.6
42
39.9
+2.0
dB/V
dB/V
dB/V
V
nA
nA
MΩ
dB/μs
±6.3
17
20
V
mA
mA
200
10
50
80
Pin 1 to Pin 2
Full 40 dB gain change
±4.75
12.5
1
±2
Typical open or short-circuited input; noise is lower when system is set to maximum gain and input is short-circuited. This figure includes the effects of both voltage
and current noise sources.
2
Using resistive loads of 500 Ω or greater or with the addition of a 1 kΩ pull-down resistor when driving lower loads.
3
The dc gain of the main amplifier in the AD603 is ×35.7; therefore, an input offset of 100 μV becomes a 3.57 mV output offset.
4
GNEG and GPOS, gain control, and voltage range are guaranteed to be within the range of −VS + 4.2 V to +VS − 3.4 V over the full temperature range of −40°C to +85°C.
Rev. H | Page 3 of 24
AD603
ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameter
Supply Voltage ±VS
Internal Voltage VINP (Pin 3)
GPOS, GNEG (Pin 1 and Pin2)
Internal Power Dissipation
Operating Temperature Range
AD603A
AD603S
Storage Temperature Range
Lead Temperature (Soldering, 60 sec)
Rating
±7.5 V
±2 V Continuous
±VS for 10 ms
±VS
400 mW
Table 3. Thermal Characteristics
Package Type
8-Lead SOIC
8-Lead CERDIP
ESD CAUTION
−40°C to +85°C
−55°C to +125°C
−65°C to +150°C
300°C
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
Rev. H | Page 4 of 24
θJA
155
140
θJC
33
15
Unit
°C/W
°C/W
AD603
PIN CONFIGURATIONS AND FUNCTION DESCRIPTIONS
VINP 3
8
VPOS
GPOS 1
7
VOUT
GNEG 2
6 VNEG
TOP VIEW
COMM 4 (Not to Scale) 5 FDBK
VPOS
7
Figure 3. 8-Lead CERDIP Pin Configuration
Table 4. Pin Function Descriptions
Mnemonic
GPOS
GNEG
VINP
COMM
FDBK
VNEG
VOUT
VPOS
8
VOUT
TOP VIEW
6 VNEG
(Not to Scale)
5 FDBK
COMM 4
VINP 3
Figure 2. 8-Lead SOIC Pin Configuration
Pin No.
1
2
3
4
5
6
7
8
AD603
00539-003
AD603
00539-002
GPOS 1
GNEG 2
Description
Gain Control Input High (Positive Voltage Increases Gain).
Gain Control Input Low (Negative Voltage Increases Gain).
Amplifier Input.
Amplifier Ground.
Connection to Feedback Network.
Negative Supply Input.
Amplifier Output.
Positive Supply Input.
Rev. H | Page 5 of 24
AD603
TYPICAL PERFORMANCE CHARACTERISTICS
@ TA = 25°C, VS = ±5 V, –500 mV ≤ VG ≤ +500 mV, GNEG = 0 V, –10 dB to +30 dB gain range, RL = 500 Ω, and CL = 5 pF, unless
otherwise noted.
225
3
180
2
135
1
0
GAIN (dB)
10.7MHz
10
100kHz
0
0
VG (V)
0.2
0.4
0.6
–45
–3
–90
–4
–135
–5
–180
Figure 4. Gain vs. VG at 100 kHz and 10.7 MHz
10M
FREQUENCY (Hz)
100M
45MHz
1.5
4
225
3
180
2
135
70MHz
0
10.7MHz
GAIN (dB)
GAIN ERROR (dB)
1
1.0
0.5
0
455kHz
–0.5
70MHz
–1.0
–0.4
–0.3
–0.2
–0.1
0
0.1
0.2
GAIN VOLTAGE (V)
0.3
0.4
0.5
–1
225
3
180
2
135
–45
–3
–90
–4
–135
–5
–180
–45
–3
–90
–4
–135
–5
–180
–6
100k
1M
10M
FREQUENCY (Hz)
100M
–225
GROUP DELAY (ns)
0
–2
PHASE (Degrees)
45
PHASE
10M
FREQUENCY (Hz)
100M
–225
7.4
7.2
7.0
6.8
6.6
00539-006
GAIN (dB)
–1
1M
7.6
90
GAIN
0
PHASE
Figure 8. Frequency and Phase Response vs. Gain
(Gain = 30 dB, PIN = −30 dBm)
4
0
45
–2
Figure 5. Gain Error vs. Gain Control Voltage at 455 kHz,
10.7 MHz, 45 MHz, 70 MHz
1
90
GAIN
–6
100k
00539-005
–1.5
–0.5
–225
Figure 7. Frequency and Phase Response vs. Gain
(Gain = 10 dB, PIN = −30 dBm)
2.5
2.0
1M
PHASE (Degrees)
–0.2
–2
Figure 6. Frequency and Phase Response vs. Gain
(Gain = −10 dB, PIN = −30 dBm)
6.4
–0.6
–0.4
–0.2
0
0.2
GAIN CONTROL VOLTAGE (V)
0.4
Figure 9. Group Delay vs. Gain Control Voltage
Rev. H | Page 6 of 24
0.6
00539-008
–0.4
0
PHASE
–6
100k
00539-004
–10
–0.6
45
–1
00539-009
GAIN (dB)
20
90
GAIN
PHASE (Degrees)
30
4
00539-007
40
AD603
–1.0
–1.2
8
3
AD603
100Ω
4
5
7
2
1
10×
PROBE
HP3585A
SPECTRUM
ANALYZER
511Ω
6
0.1µF
00539-010
–5V
DATEL
DVC 8500
–1.6
–1.8
–2.0
–2.2
–2.4
–2.6
–2.8
–3.0
–3.2
–3.4
Figure 10. Third-Order Intermodulation Distortion Test Setup
0
50
100
200
500
1000
LOAD RESISTANCE (Ω)
00539-013
+5V
HP3326A
DUALCHANNEL
SYNTHESIZER
NEGATIVE OUTPUT VOLTAGE (V)
–1.4
0.1µF
2000
Figure 13. Typical Output Voltage Swing vs. Load Resistance
(Negative Output Swing Limits First)
102
INPUT IMPEDANCE (Ω)
10dB/DIV
100
98
96
100k
Figure 11. Third-Order Intermodulation Distortion at 455 kHz
(10× Probe Used to HP3585A Spectrum Analyzer, Gain = 0 dB, PIN = 0 dBm)
1M
10M
FREQUENCY (Hz)
100M
00539-014
00539-011
94
Figure 14. Input Impedance vs. Frequency (Gain = −10 dB)
102
INPUT IMPEDANCE (Ω)
10dB/DIV
100
98
96
100k
Figure 12. Third-Order Intermodulation Distortion at 10.7 MHz
(10× Probe Used to HP3585A Spectrum Analyzer, Gain = 0 dB, PIN = 0 dBm)
Rev. H | Page 7 of 24
1M
10M
FREQUENCY (Hz)
100M
Figure 15. Input Impedance vs. Frequency (Gain = 10 dB)
00539-015
00539-012
94
AD603
3V
INPUT IMPEDANCE (Ω)
102
INPUT GND
100MV/DIV
100
1V
98
OUTPUT GND
1V/DIV
96
1M
10M
FREQUENCY (Hz)
100M
–2V
–49ns
50ns
451ns
00539-019
100k
00539-016
94
Figure 19. Output Stage Overload Recovery Time
(Input Is 500 ns Period, 50% Duty-Cycle Square Wave,
Output Is Captured Using Tektronix 11402 Digitizing Oscilloscope)
Figure 16. Input Impedance vs. Frequency (Gain = 30 dB)
3.5V
INPUT
500mV/DIV
1V
GND
100
90
500mV
OUTPUT
500mV/DIV
GND
–1.5V
–44ns
00539-017
200ns
1V
50ns
00539-020
10
0%
456ns
Figure 20. Transient Response, G = 0 dB
(Input Is 500 ns Period, 50% Duty-Cycle Square Wave,
Output Is Captured Using Tektronix 11402 Digitizing Oscilloscope)
Figure 17. Gain Control Channel Response Time
3.5V
4.5V
INPUT GND
1V/DIV
INPUT GND
100mV/DIV
500mV
500mV
OUTPUT GND
500mV/DIV
50ns
451ns
–1.5V
–44ns
50ns
456ns
00539-021
–500mV
–49ns
00539-018
OUTPUT GND
500mV/DIV
Figure 21. Transient Response, G = 20 dB
(Input Is 500 ns Period, 50% Duty-Cycle Square Wave,
Output Is Captured Using Tektronix 11402 Digitizing Oscilloscope)
Figure 18. Input Stage Overload Recovery Time
(Input Is 500 ns Period, 50% Duty-Cycle Square Wave,
Output Is Captured Using Tektronix 11402 Digitizing Oscilloscope)
Rev. H | Page 8 of 24
AD603
21
0
TA = 25°C
RS = 50Ω
TEST SETUP FIGURE 23
10MHz
19
–10
17
NOISE FIGURE (dB)
PSRR (dB)
–20
–30
–40
–50
–60
15
20MHz
13
11
9
1M
10M
FREQUENCY (Hz)
100M
5
30
00539-022
100k
Figure 22. PSRR vs. Frequency
(Worst Case Is Negative Supply PSRR, Shown Here)
31
32
33
34
35
36
GAIN (dB)
37
38
39
40
00539-025
7
Figure 25. Noise Figure in 0 dB/40 dB Mode
0
TA = 25°C
TEST SETUP FIGURE 23
–5
HP3326A
DUALCHANNEL
SYNTHESIZER
INPUT LEVEL (dBm)
0.1µF
+5V
8
3
AD603
100Ω
4
HP3585A
SPECTRUM
ANALYZER
50Ω
5
7
2
1
–10
–15
–20
6
00539-023
–5V
DATEL
DVC 8500
Figure 23. Test Setup Used for: Noise Figure, Third-Order Intercept, and
1 dB Compression Point Measurements
23
21
–25
10
20
TA = 25°C
TEST SETUP FIGURE 23
18
19
OUTPUT LEVEL (dBm)
30MHz
17
50MHz
15
13
30MHz
11
10MHz
16
40MHz
14
12
70MHz
9
10
5
20
21
22
23
24
25
26
GAIN (dB)
27
28
29
30
0
–20
Figure 24. Noise Figure in −10 dB/+30 dB Mode
–10
INPUT LEVEL (dBm)
0
00539-027
7
00539-024
NOISE FIGURE (dB)
70
Figure 26. 1 dB Compression Point, −10 dB/+30 dB Mode, Gain = 30 dB
TA = 25°C
RS = 50V
TEST SETUP FIGURE 23
70MHz
30
50
INPUT FREQUENCY (MHz)
00539-026
0.1µF
Figure 27. Third-Order Intercept −10 dB/+30 dB Mode, Gain = 10 dB
Rev. H | Page 9 of 24
AD603
20
18
16
40MHz
14
12
70MHz
10
8
–40
–30
INPUT LEVEL (dBm)
–20
00539-028
OUTPUT LEVEL (dBm)
30MHz
TA = 25°C
RS = 50Ω
RIN = 50Ω
RL = 100Ω
TEST SETUP FIGURE 23
Figure 28. Third-Order Intercept −10 dB/+30 dB Mode, Gain = 30 dB
Rev. H | Page 10 of 24
AD603
THEORY OF OPERATION
The AD603 comprises a fixed-gain amplifier, preceded by a
broadband passive attenuator of 0 dB to 42.14 dB, having a gain
control scaling factor of 40 dB per volt. The fixed gain is lasertrimmed in two ranges, to either 31.07 dB (×35.8) or 50 dB
(×358), or it may be set to any range in between using one
external resistor between Pin 5 and Pin 7. Somewhat higher
gain can be obtained by connecting the resistor from Pin 5 to
common, but the increase in output offset voltage limits the
maximum gain to about 60 dB. For any given range, the
bandwidth is independent of the voltage-controlled gain. This
system provides an underrange and overrange of 1.07 dB in all
cases; for example, the overall gain is −11.07 dB to +31.07 dB in
the maximum bandwidth mode (Pin 5 and Pin 7 strapped).
This X-AMP structure has many advantages over former
methods of gain control based on nonlinear elements. Most
importantly, the fixed-gain amplifier can use negative feedback
to increase its accuracy. Because large inputs are first attenuated,
the amplifier input is always small. For example, to deliver a
±1 V output in the −1 dB/+41 dB mode (that is, using a fixed
amplifier gain of 41.07 dB), its input is only 8.84 mV; therefore,
the distortion can be very low. Equally important, the smallsignal gain and phase response, and thus the pulse response, are
essentially independent of gain.
Figure 29 is a simplified schematic. The input attenuator is a
7-section R-2R ladder network, using untrimmed resistors of
nominally R = 62.5 Ω, which results in a characteristic resistance of
125 Ω ± 20%. A shunt resistor is included at the input and laser
trimmed to establish a more exact input resistance of 100 Ω ± 3%,
which ensures accurate operation (gain and HP corner frequency)
when used in conjunction with external resistors or capacitors.
The nominal maximum signal at input VINP is 1 V rms
(±1.4 V peak) when using the recommended ±5 V supplies,
although operation to ±2 V peak is permissible with some
increase in HF distortion and feedthrough. Pin 4 (COMM)
must be connected directly to the input ground; significant
impedance in this connection reduces the gain accuracy.
The signal applied at the input of the ladder network is attenuated
by 6.02 dB by each section; therefore, the attenuation to each of
the taps is progressively 0 dB, 6.02 dB, 12.04 dB, 18.06 dB,
24.08 dB, 30.1 dB, 36.12 dB, and 42.14 dB. A unique circuit
technique is employed to interpolate between these tap points,
indicated by the slider in Figure 29, thus providing continuous
attenuation from 0 dB to 42.14 dB. It helps in understanding the
AD603 to think in terms of a mechanical means for moving this
slider from left to right; in fact, its position is controlled by the
voltage between Pin 1 and Pin 2. The details of the gain control
interface are in the The Gain Control Interface section.
The gain is at all times very exactly determined, and a linear-indB relationship is automatically guaranteed by the exponential
nature of the attenuation in the ladder network (the X-AMP
principle). In practice, the gain deviates slightly from the ideal
law, by about ±0.2 dB peak (see, for example, Figure 5).
NOISE PERFORMANCE
An important advantage of the X-AMP is its superior noise
performance. The nominal resistance seen at inner tap points is
41.7 Ω (one third of 125 Ω), which exhibits a Johnson noise
spectral density (NSD) of 0.83 nV/√Hz (that is, √4kTR) at 27°C,
which is a large fraction of the total input noise. The first stage
of the amplifier contributes a further 1 nV/√Hz, for a total input
noise of 1.3 nV/√Hz. It is apparent that it is essential to use a
low resistance in the ladder network to achieve the very low
specified noise level. The source impedance of the signal
forms a voltage divider with the 100 Ω input resistance of the
AD603. In some applications, the resulting attenuation may
be unacceptable, requiring the use of an external buffer or
preamplifier to match a high impedance source to the low
impedance AD603.
The noise at maximum gain (that is, at the 0 dB tap) depends on
whether the input is short-circuited or open-circuited. When
short-circuited, the minimum NSD of slightly over 1 nV/√Hz is
achieved. When open-circuited, the resistance of 100 Ω looking
into the first tap generates 1.29 nV/√Hz, so the noise increases
to 1.63 nV/√Hz. (This last calculation would be important if the
AD603 were preceded by, for example, a 900 Ω resistor to allow
operation from inputs up to 10 V rms.) As the selected tap
moves away from the input, the dependence of the noise on
source impedance quickly diminishes.
Apart from the small variations just discussed, the signal-tonoise (SNR) at the output is essentially independent of the
attenuator setting. For example, on the −11 dB/+31 dB range,
the fixed gain of ×35.8 raises the output NSD to 46.5 nV/√Hz.
Therefore, for the maximum undistorted output of 1 V rms and
a 1 MHz bandwidth, the output SNR would be 86.6 dB, that is,
20 log(1 V/46.5 μV).
Rev. H | Page 11 of 24
AD603
VPOS 8
SCALING
REFERENCE
VNEG 6
PRECISION PASSIVE
INPUT ATTENUATOR
FIXED-GAIN
AMPLIFIER
GPOS 1
7
VOUT
5
FDBK
VG
GNEG 2
6.44kΩ*
AD603
GAINCONTROL
INTERFACE
694Ω*
0dB
VINP 3
–6.02dB
R
2R
–12.04dB –18.06dB –24.08dB
R
2R
R
2R
R
2R
–30.1dB
R
2R
–36.12dB –42.14dB
R
2R
R
20Ω*
R
COMM 4
00539-029
R-2R LADDER NETWORK
*NOMINAL VALUES.
Figure 29. Simplified Block Diagram
THE GAIN CONTROL INTERFACE
The attenuation is controlled through a differential, high
impedance (50 MΩ) input, with a scaling factor that is lasertrimmed to 40 dB per volt, that is, 25 mV/dB. An internal band
gap reference ensures stability of the scaling with respect to
supply and temperature variations.
When the differential input voltage VG = 0 V, the attenuator
slider is centered, providing an attenuation of 21.07 dB. For the
maximum bandwidth range, this results in an overall gain of
10 dB (= −21.07 dB + 31.07 dB). When the control input is
−500 mV, the gain is lowered by +20 dB (= 0.500 V × 40 dB/V)
to −10 dB; when set to +500 mV, the gain is increased by
+20 dB to +30 dB. When this interface is overdriven in either
direction, the gain approaches either −11.07 dB (= − 42.14 dB +
+31.07 dB) or 31.07 dB (= 0 + 31.07 dB), respectively. The only
constraint on the gain control voltage is that it be kept within
the common-mode range (−1.2 V to +2.0 V assuming +5 V
supplies) of the gain control interface.
The basic gain of the AD603 can therefore be calculated by
Gain (dB) = 40 VG +10
(1)
where VG is in volts. When Pin 5 and Pin 7 are strapped (see the
Programming the Fixed-Gain Amplifier Using Pin Strapping
section), the gain becomes
For example, if the gain is to be controlled by a DAC providing
a positive-only, ground-referenced output, the gain control low
(GNEG) pin should be biased to a fixed offset of 500 mV to set
the gain to −10 dB when gain control high (GPOS) is at zero,
and to 30 dB when at 1.00 V.
It is a simple matter to include a voltage divider to achieve other
scaling factors. When using an 8-bit DAC having an FS output
of 2.55 V (10 mV/bit), a divider ratio of 2 (generating 5 mV/bit)
results in a gain-setting resolution of 0.2 dB/bit. The use of such
offsets is valuable when two AD603s are cascaded, when
various options exist for optimizing the signal-to-noise profile,
as is shown in the Sequential Mode (Optimal SNR) section,
PROGRAMMING THE FIXED-GAIN AMPLIFIER
USING PIN STRAPPING
Access to the feedback network is provided at Pin 5 (FDBK).
The user may program the gain of the output amplifier of the
AD603 using this pin, as shown in Figure 30, Figure 31, and
Figure 32. There are three modes: in the default mode, FDBK
is unconnected, providing the range +9 dB/+51 dB; when VOUT
and FDBK are shorted, the gain is lowered to −11 dB/+31 dB;
and, when an external resistor is placed between VOUT and
FDBK, any intermediate gain can be achieved, for example,
−1 dB/+41 dB. Figure 33 shows the nominal maximum gain vs.
external resistor for this mode.
Gain (dB) = 40 VG + 20 for 0 to +40 dB
VC1
and
1
GPOS
VPOS 8
VPOS
AD603
VC2
(2)
The high impedance gain control input ensures minimal
loading when driving many amplifiers in multiple channel
or cascaded applications. The differential capability provides
flexibility in choosing the appropriate signal levels and
polarities for various control schemes.
Rev. H | Page 12 of 24
VIN
2
GNEG
VOUT 7
3
VINP
VNEG 6
4
COMM
FDBK 5
VOUT
VNEG
Figure 30. −10 dB to +30 dB; 90 MHz Bandwidth
00539-030
Gain (dB) = 40 VG + 30 for +10 to +50 dB
AD603
VPOS 8
VC1
1
GPOS
VC2
2
GNEG
VOUT 7
3
VINP
VNEG 6
4
COMM
FDBK 5
Optionally, when a resistor is placed from FDBK to COMM,
higher gains can be achieved. This fourth mode is of limited
value because of the low bandwidth and the elevated output
offsets; it is thus not included in Figure 30, Figure 31, or
Figure 32.
VPOS
AD603
VIN
VOUT
VNEG
2.15kΩ
The gain of this amplifier in the first two modes is set by the
ratio of on-chip laser-trimmed resistors. While the ratio of these
resistors is very accurate, the absolute value of these resistors
can vary by as much as ±20%. Therefore, when an external
resistor is connected in parallel with the nominal 6.44 kΩ ± 20%
internal resistor, the overall gain accuracy is somewhat poorer.
The worst-case error occurs at about 2 kΩ (see Figure 34).
00539-031
5.6pF
Figure 31. 0 dB to 40 dB; 30 MHz Bandwidth
VC1
1
GPOS
VPOS 8
VPOS
AD603
VC2
2
GNEG
VOUT 7
3
VINP
VNEG 6
4
COMM
FDBK 5
VOUT
1.2
VNEG
0.8
0.6
0.4
GAIN ERROR (dB)
18pF
Figure 32. 10 dB to 50 dB; 9 MHz to Set Gain
52
50
–1:VdB (OUT)
48
0.2
0
–0.2
–0.4
46
–0.6
–1.0
10
–2:VdB (OUT)
40
VdB (OUT) – VdB (OREF )
100
1k
10k
REXT (Ω)
38
100k
1M
00539-034
–0.8
VdB (OUT)
42
Figure 34. Worst-Case Gain Error, Assuming Internal Resistors Have a
Maximum Tolerance of −20% (Top Curve) or = 20% (Bottom Curve)
36
34
32
100
1k
10k
REXT (Ω)
100k
1M
00539-033
GAIN (dB)
44
30
10
–1:VdB (OUT) – (–1):VdB (OREF )
1.0
00539-032
VIN
Figure 33. Gain vs. REXT, Showing Worst-Case Limits Assuming Internal
Resistors Have a Maximum Tolerance of 20%
While the gain bandwidth product of the fixed-gain amplifier is
about 4 GHz, the actual bandwidth is not exactly related to the
maximum gain. This is because there is a slight enhancing of
the ac response magnitude on the maximum bandwidth range,
due to higher order poles in the open-loop gain function; this
mild peaking is not present on the higher gain ranges. Figure 30,
Figure 31, and Figure 32 show how an optional capacitor may
be added to extend the frequency response in high gain modes.
Rev. H | Page 13 of 24
AD603
USING THE AD603 IN CASCADE
Two or more AD603s can be connected in series to achieve
higher gain. Invariably, ac coupling must be used to prevent the
dc offset voltage at the output of each amplifier from overloading
the following amplifier at maximum gain. The required highpass coupling network is usually just a capacitor, chosen to set
the desired corner frequency in conjunction with the welldefined 100 Ω input resistance of the following amplifier.
Figure 35 shows the SNR over a gain range of −22 dB to +62 dB,
assuming an output of 1 V rms and a 1 MHz bandwidth. Figure 36,
Figure 37, and Figure 38 show the general connections to
accomplish this. Here, both the positive gain control inputs
(GPOS) are driven in parallel by a positive-only, ground-referenced
source with a range of 0 V to 2 V, while the negative gain
control inputs (GNEG) are biased by stable voltages to provide
the needed gain offsets. These voltages may be provided by
resistive dividers operating from a common voltage reference.
For two AD603s, the total gain control range becomes 84 dB
(2 × 42.14 dB); the overall −3 dB bandwidth of cascaded stages
is somewhat reduced. Depending on the pin strapping, the gain
and bandwidth for two cascaded amplifiers can range from
−22 dB to +62 dB (with a bandwidth of about 70 MHz) to
+22 dB to +102 dB (with a bandwidth of about 6 MHz).
90
85
80
75
SNR (dB)
There are several ways of connecting the gain control inputs
in cascaded operation. The choice depends on whether it is
important to achieve the highest possible instantaneous signalto-noise ratio (ISNR), or, alternatively, to minimize the ripple
in the gain error. The following examples feature the AD603
programmed for maximum bandwidth; the explanations apply
to other gain/bandwidth combinations with appropriate
changes to the arrangements for setting the maximum gain.
70
65
60
50
–0.2
SEQUENTIAL MODE (OPTIMAL SNR)
A1
GNEG
VG1
–51.07dB
31.07dB
–8.93dB
–42.14dB
GPOS
VG2
VO1 = 0.473V
VC = 0V
GNEG
31.07dB
OUTPUT
–20dB
VO2 = 1.526V
00539-036
–42.14dB
1.0
VC (V)
A2
–40.00dB
GPOS
0.6
Figure 36. AD603 Gain Control Input Calculations for Sequential Control Operation VC = 0 V
–11.07dB
0dB
GPOS
GNEG
VG1
31.07dB
31.07dB
–42.14dB
GPOS
VG2
VO1 = 0.473V
VC = 1.0V
GNEG
31.07dB
OUTPUT
20dB
VO2 = 1.526V
00539-037
0dB
INPUT
0dB
Figure 37. AD603 Gain Control Calculations for Sequential Control Operation VC = 1.0 V
0dB
GPOS
GNEG
VG1
VC = 2.0V
–28.93dB
31.07dB
VO1 = 0.473V
31.07dB
–2.14dB
GPOS
GNEG
VG2
31.07dB
OUTPUT
60dB
VO2 = 1.526V
Figure 38. AD603 Gain Control Input Calculations for Sequential Operation VC = 2.0 V
Rev. H | Page 14 of 24
00539-038
0dB
INPUT
0dB
1.4
1.8
2.2
Figure 35. SNR vs. Control Voltage, Sequential Control (1 MHz Bandwidth)
In the sequential mode of operation, the ISNR is maintained at
its highest level for as much of the gain control range as possible.
INPUT
0dB
0.2
00539-035
55
AD603
Gain (dB) = 40 VG + GO
(3)
Figure 41 is a plot of the SNR of the cascaded amplifiers vs. the
control voltage. Figure 42 is a plot of the gain error of the
cascaded stages vs. the control voltages.
70
60
20
10
0
A2
0.2
0.6
1.0
VC (V)
1.4
1.8
2.0
00539-040
–30
–0.2
Figure 40. Plot of Separate and Overall Gains in Sequential Control
90
+31.07dB
80
+31.07dB
+28.96dB
70
A2
*
–11.07dB
*
60
SNR (dB)
A1
–8.93dB
–11.07dB
1.526
0.5
0
1.0
20
1.50
40
2.0
60
VC (V)
62.14
*GAIN OFFSET OF 1.07dB, OR 26.75mV.
00539-039
0.473
0
–20
A1
30
–20
In the explanatory notes that follow, it is assumed that
the maximum bandwidth connections are used, for which
GO is −20 dB.
GAIN
(dB) –22.14
40
–10
where:
VG is the applied control voltage.
GO is determined by the gain range chosen.
+10dB
COMBINED
50
OVERALL GAIN (dB)
The gains are offset (Figure 39) such that the gain of A2 is
increased only after the gain of A1 has reached its maximum
value. Note that for a differential input of –600 mV or less, the
gain of a single amplifier (A1 or A2) is at its minimum value of
−11.07 dB; for a differential input of 600 mV or more, the gain
is at its maximum value of 31.07 dB. Control inputs beyond
these limits do not affect the gain and can be tolerated without
damage or foldover in the response. This is an important aspect
of the gain control response of the AD603. (See the Specifications
section for more details on the allowable voltage range.) The
gain is now
50
40
30
20
0.2
0.6
1.0
VC (V)
1.4
1.8
2.0
Figure 41. SNR for Cascaded Stages—Sequential Control
2.0
1.5
1.0
When VG = 2.0 V, the gain of A1 is pinned at 31.07 dB and that
of A2 is near its maximum value of 28.93 dB, resulting in an
overall gain of 60 dB (see Figure 38). This mode of operation is
further clarified in Figure 40, which is a plot of the separate
gains of A1 and A2 and the overall gain vs. the control voltage.
Rev. H | Page 15 of 24
0.5
0
–0.5
–1.0
–1.5
–2.0
–0.2
0
0.2
0.4
0.6
0.8
1.0 1.2
VC (V)
1.4
1.6
1.8
2.0
2.2
Figure 42. Gain Error for Cascaded Stages–Sequential Control
00539-042
When VG = 1.00 V, VG1 = 1.00 V − 0.473 V = 0.526 V, which sets
the gain of A1 to nearly its maximum value of +31.07 dB, while
VG2 = 1.00 V − 1.526 V = 0.526 V, which sets the gain of A2 to
nearly its minimum value of −11.07 dB. Close analysis shows
that the degree to which neither AD603 is completely pushed to
its maximum nor minimum gain exactly cancels in the overall
gain, which is now 20 dB (see Figure 37).
10
–0.2
GAIN ERROR (dB)
With reference to Figure 36, Figure 37, and Figure 38, note that
VG1 refers to the differential gain control input to A1, and VG2
refers to the differential gain control input to A2. When VG is
0 V, VG1 = −473 mV and thus the gain of A1 is −8.93 dB (recall
that the gain of each individual amplifier in the maximum
bandwidth mode is –10 dB for VG = −500 mV and 10 dB for VG
= 0 V); meanwhile, VG2 = −1.908 V so the gain of A2 is pinned
at −11.07 dB. The overall gain is therefore –20 dB (see Figure 36).
00539-041
Figure 39. Explanation of Offset Calibration for Sequential Control
AD603
LOW GAIN RIPPLE MODE (MINIMUM GAIN ERROR)
In this mode, the gain control of voltage is applied to both
inputs in parallel: the GPOS pins of both A1 and A2 are
connected to the control voltage and the GNEW inputs are
grounded. The gain scaling is then doubled to 80 dB/V,
requiring only a 1.00 V change for an 80 dB change of gain
Gain = (dB) = 80 VG + GO
(4)
where, as before, GO depends on the range selected; for example,
in the maximum bandwidth mode, GO is 20 dB. Alternatively,
the GNEG pins may be connected to an offset voltage of
0.500 V, in which case GO is −20 dB.
The amplitude of the gain ripple in this case is also doubled, as
shown in Figure 43, while the ISNR at the output of A2 now
decreases linearly as the gain increases, as shown in Figure 44.
2.0
1.5
As can be seen in Figure 42 and Figure 43, the error in the gain
is periodic, that is, it shows a small ripple. (Note that there is
also a variation in the output offset voltage, which is due to the
gain interpolation, but this is not exact in amplitude.) By
offsetting the gains of A1 and A2 by half the period of the ripple,
that is, by 3 dB, the residual gain errors of the two amplifiers
can be made to cancel. Figure 45 shows much lower gain ripple
when configured in this manner. Figure 46 plots the ISNR as a
function of gain; it is very similar to that in the parallel mode.
3.0
2.5
2.0
1.5
GAIN ERROR (dB)
PARALLEL MODE (SIMPLEST GAIN CONTROL
INTERFACE)
0
–0.5
–1.0
–2.0
0.5
–3.0
–0.1
0
0.1
0.2
0.3
0.4
–0.5
0.5 0.6
VC (V)
0.7
0.8
0.9
1.0
1.1
00539-045
–2.5
0
Figure 45. Gain Error for Cascaded Stages—Low Ripple Mode
–1.0
90
–1.5
0
0.2
0.4
0.6
0.8
1.0 1.2
VC (V)
1.4
1.6
1.8
2.0
2.2
00539-043
85
–2.0
–0.2
80
75
ISNR (dB)
Figure 43. Gain Error for Cascaded Stages—Parallel Control
90
85
70
65
80
60
75
55
70
50
–0.2
0
0.2
0.4
0.6
0.8
1.0
VC (V)
65
Figure 46. ISNR vs. Control Voltage—Low Ripple Mode
60
50
–0.2
0
0.2
0.4
0.6
VC (V)
0.8
1.0
1.2
00539-044
55
Figure 44. ISNR for Cascaded Stages—Parallel Control
Rev. H | Page 16 of 24
1.2
00539-046
GAIN ERROR (dB)
0.5
–1.5
1.0
ISNR (dB)
1.0
AD603
APPLICATIONS INFORMATION
A LOW NOISE AGC AMPLIFIER
The circuit operates as follows:
Figure 47 shows the ease with which the AD603 can be
connected as an AGC amplifier. The circuit illustrates many of
the points previously discussed: it uses few parts, has linear-indB gain, operates from a single supply, uses two cascaded amplifiers
in sequential gain mode for maximum SNR, and an external
resistor programs each gain of the amplifier. It also uses a
simple temperature-compensated detector.
• A1 and A2 are cascaded.
• Capacitor C1 and the 100 Ω of resistance at the input of A1
form a time constant of 10 μs.
• C2 blocks the small dc offset voltage at the output of A1
(which might otherwise saturate A2 at its maximum gain)
and introduces a high-pass corner at about 16 kHz,
eliminating low frequency noise.
The circuit operates from a single 10 V supply. Resistors R1, R2,
R3, and R4 bias the common pins of A1 and A2 at 5 V. The
common pin is a low impedance point and must have a low
impedance path to ground, provided here by the 100 μF tantalum
capacitors and the 0.1 μF ceramic capacitors.
A half-wave detector is used, based on Q1 and R8. The current
into capacitor, CAV, is the difference between the collector
current of Q2 (biased to be 300 μA at 300 K, 27°C) and the
collector current of Q1, which increases with the amplitude
of the output signal.
The cascaded amplifiers operate in sequential gain. Here, the
offset voltage between Pin 2 (GNEG) of A1 and A2 is 1.05 V
(42.14 dB × 25 mV/dB), provided by a voltage divider consisting of
Resistors R5, R6, and R7. Using standard values, the offset is not
exact, but it is not critical for this application.
The automatic gain control voltage, VAGC, is the time integral
of this error current. For VAGC (and thus the gain) to remain
insensitive to short-term amplitude fluctuations in the output
signal, the rectified current in Q1 must, on average, exactly
balance the current in Q2. If the output of A2 is too small to
do this, VAGC increases, causing the gain to increase until Q1
conducts sufficiently.
The gain of both A1 and A2 is programmed by Resistors R13
and R14, respectively, to be about 42 dB; therefore, the maximum
gain of the circuit is twice that, or 84 dB. The gain control range
can be shifted up by as much as 20 dB by appropriate choices of
R13 and R14.
Consider the case where R8 is zero and the output voltage VOUT
is a square wave at, for example, 455 kHz, which is well above
the corner frequency of the control loop.
10V
C7
0.1µF
10V
C1
0.1µF
J1
R1
2.49kΩ
+
C4
0.1µF
5
8
7
A2
AD603
10V
1
C 52
100µF
+
C6
0.1µF
Q1
2N3904
R8
806Ω
5
R11
3.83kΩ
5V
R1 2
4.99kΩ
C9
0.1µF
J2
7
2
C10
0.1µF
4
R3
2.49kΩ
R2
2.49kΩ
CAV
0.1µF
6
3
2
4
R1 4
2.49kΩ
C11
0.1µF
1
R4
2.49kΩ
AGC LINE
1V OFFSET FOR
SEQUENTIAL GAIN
R5
5.49kΩ
5.5V
R6
1.05kΩ
R7
3.48kΩ
6.5V
10V
00539-047
C 32
100µF
C2
0.1µF
6
A1
AD603
10V
10V
R1 0
1.24kΩ
Q2
2N3906
VAGC
C8
0.1µF
R1 3
2.49kΩ
8
3
R T1
100Ω
R9
1.54kΩ
THIS CAPACITOR SETS
AGC TIME CONSTANT
1 RT PR OVI D ES A 5 0Ω IN PU T I MPED A N C E.
2 C 3 A N D C 5 A R E TA NTA LU M.
Figure 47. A Low Noise AGC Amplifier
Rev. H | Page 17 of 24
AD603
During the time VOUT is negative with respect to the base
voltage of Q1, Q1 conducts; when VOUT is positive, it is cut off.
Because the average collector current of Q1 is forced to be
300 μA, and the square wave has a duty cycle of 1:1, Q1’s
collector current when conducting must be 600 μA. With R8
omitted, the peak amplitude of VOUT is forced to be just the VBE
of Q1 at 600 μA, typically about 700 mV, or 2 VBE peak-to-peak.
This voltage, the amplitude at which the output stabilizes, has a
strong negative temperature coefficient (TC), typically −1.7 mV/°C.
Although this may not be troublesome in some applications, the
correct value of R8 renders the output stable with temperature.
To understand this, note that the current in Q2 is made to be
proportional to absolute temperature (PTAT). For the moment,
continue to assume that the signal is a square wave.
When Q1 is conducting, VOUT is now the sum of VBE and a
voltage that is PTAT and that can be chosen to have an equal
but opposite TC to that of the VBE. This is actually nothing more
than an application of the band gap voltage reference principle.
When R8 is chosen such that the sum of the voltage across it
and the VBE of Q1 is close to the band gap voltage of about 1.2 V,
VOUT is stable over a wide range of temperatures, provided, of
course, that Q1 and Q2 share the same thermal environment.
Because the average emitter current is 600 μA during each half
cycle of the square wave, a resistor of 833 Ω adds a PTAT
voltage of 500 mV at 300 K, increasing by 1.66 mV/°C. In
practice, the optimum value depends on the type of transistor
used and, to a lesser extent, on the waveform for which the
temperature stability is to be optimized; for the inexpensive
2N3904/2N3906 pair and sine wave signals, the recommended
value is 806 Ω.
This resistor also serves to lower the peak current in Q1 when
more typical signals (usually sinusoidal) are involved, and the
1.8 kHz LP filter it forms with CAV helps to minimize distortion
due to ripple in VAGC. Note that the output amplitude under sine
wave conditions is higher than for a square wave because the
average value of the current for an ideal rectifier is 0.637 times
as large, causing the output amplitude to be 1.88 (= 1.2/0.637) V,
or 1.33 V rms. In practice, the somewhat nonideal rectifier
results in the sine-wave output being regulated to about
1.4 V rms, or 3.6 V p-p.
The bandwidth of the circuit exceeds 40 MHz. At 10.7 MHz, the
AGC threshold is 100 μV (−67 dBm) and its maximum gain is
83 dB (20 log 1.4 V/100 μV). The circuit holds its output at
1.4 V rms for inputs as low as −67 dBm to +15 dBm (82 dB),
where the input signal exceeds the maximum input rating of the
AD603. For a 30 dBm input at 10.7 MHz, the second harmonic
is 34 dB down from the fundamental, and the third harmonic is
35 dB down from the fundamental.
CAUTION
Careful component selection, circuit layout, power supply
decoupling, and shielding are needed to minimize the susceptibility
of the AD603 to interference from signals such as those from
radio and TV stations. In bench evaluation, it is recommended
to place all of the components into a shielded box and use
feedthrough decoupling networks for the supply voltage. Circuit
layout and construction are also critical because stray capacitances
and lead inductances can form resonant circuits and are a
potential source of circuit peaking, oscillation, or both.
Rev. H | Page 18 of 24
AD603
EVALUATION BOARD
The evaluation board of the AD603 enables simple bench-top
experimenting to be performed with easy control of the
AD603. Built-in flexibility allows convenient configuration to
accommodate most operating configurations. Figure 48 is a
photograph of the AD603 evaluation board.
The output is also ac-coupled and includes a 453 Ω series
resistor. The gain of the AD603 is adjusted by connecting a
voltage source between the GNEG and GPOS test loops. A 0 Ω
resistor, R5, is provided, permitting ground reference of the
differential gain control inputs. For other gain configurations, a
signal generator may be connected to the test loops, or the
GNEG and/or GPOS pins may be biased. Either pin may be
driven to either polarity within the common-mode limits of
−1.2 V to +2.0 V; therefore, to invert the gain slope, simply
consider the GPOS as the reference and drive the GNEG pin
positively with respect to GPOS.
00539-048
The AD603 includes built-in gain resistors selectable at the
FDBK pin. The board is shipped with the gain at minimum,
with a 0 Ω resistor installed in R3. For maximum gain, simply
remove R3. Because of the architecture of the AD603, the
bandwidth decreases by 10, but the gain range remains at 40 dB.
Intermediate gain values may be selected by installing a resistor
between the VOUT and FDBK pins.
Figure 48. AD603 Evaluation Board
Any dual-polarity power supply capable of providing 20 mA is
all that is required, in addition to whatever test equipment the
user wishes to perform the intended tests.
Referring to the schematic in Figure 49, the input to the VGA is
single-ended, ac-coupled, and terminated in 50 Ω to accommodate
most commonly available signal generators.
V+
C7
10µF
25V
V–
G1
+
+
V+
V–
G2 GND G3
Figure 50, Figure 51, and Figure 52 show the component and
circuit side copper patterns and silkscreen. A bill of materials is
shown in Table 5.
G4
C8
10µF
25V
V+
V+
R7
R1
R5
0Ω
GNEG
C2
0.1µF
R6
AD603
1
C4
2
C5
0.1µF
3
4
VIN
W2
R2
100Ω
GPOS
VPOS
GNEG
VOUT
VINP
VNEG
COMM
FDBK
8
7
6
VO
R4
453Ω
C6
0.1µF
VOUT
00539-050
C1
0.1µF
W1
Figure 50. Component Side Copper
R3
0Ω
5
V–
C9
C3
0.1µF
00539-049
GPOS
00539-051
Figure 49. Schematic of the AD603 Evaluation Board
Figure 51. Circuit Side Copper
Rev. H | Page 19 of 24
AD603
G1
G4
V+
GND
V–
C8
C7
+
R1
C1
C2
DUT
C3
R4
W1
VOUT
R2
C5
C9
W2
R5
C4
R6
GNEG
VIN
VO
+
R7
R3
C6
GPOS
Pb
AD603ARZ
EVALUATION BOARD
G3
RoHS
00539-052
G2
Figure 52. Component Side Silk Screen
Table 5. Bill of Materials
Qty
5
2
1
5
3
1
2
1
1
1
2
2
Name
Capacitors
Capacitors
IC
Test Loops
Test Loops
Resistor
Resistors
Resistor
Test Loop
Test Loop
Connectors
Headers
Description
0.1 μF, 16 V, 0603, X7R
10 μF, 25 V, C size tantalum
Variable gain amplifier
0.125” diameter, black
0.125” diameter, purple
100 Ω, 1%, 1/16 W, 0603
0 Ω, 5%, 1/10 W, 0603
453 Ω, 1/16 W, 1%, 0603
0.125” diameter, red
0.125” diameter, green
SMA female PC mount, RA
2-pin 025" sq., 0.1" spacing
Not inserted
Reference Designator
C1, C2, C3, C5, C6
C7, C8
DUT
G1, G2, G3, G4, GND
GNEG, GPOS, VO
R2
R3, R5
R4
+5V
−5 V
VIN, VOUT
W1, W2
C4, C9, R1, R6, R7
Rev. H | Page 20 of 24
Manufacturer
KEMET
Nichicon
Analog Devices, Inc.
Components Corp.
Components Corp.
Panasonic
Panasonic
Panasonic
Components Corp.
Components Corp.
Amphenol
Molex
Mfg Part Number
C0603C104K4RACTU
F931E106MCC
AD603ARZ
TP-104-01-00
TP-104-01-07
ERJ-3EKF1000V
ERJ-2GE0R00X
ERJ-3EKF4530V
TP-104-01-02
TP-104-01-05
901-143-6RFX
22-11-2032
AD603
OUTLINE DIMENSIONS
0.005 (0.13)
MIN
0.055 (1.40)
MAX
8
5
0.310 (7.87)
0.220 (5.59)
1
4
0.100 (2.54) BSC
0.320 (8.13)
0.290 (7.37)
0.405 (10.29) MAX
0.060 (1.52)
0.015 (0.38)
0.200 (5.08)
MAX
0.150 (3.81)
MIN
0.200 (5.08)
0.125 (3.18)
0.023 (0.58)
0.014 (0.36)
0.070 (1.78)
0.030 (0.76)
SEATING
PLANE
15°
0°
0.015 (0.38)
0.008 (0.20)
CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETER DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN.
Figure 53. 8-Lead Ceramic Dual In-Line Package [CERDIP]
(Q-8)
Dimensions shown in inches and (millimeters)
5.00 (0.1968)
4.80 (0.1890)
8
1
5
4
1.27 (0.0500)
BSC
0.25 (0.0098)
0.10 (0.0040)
COPLANARITY
0.10
SEATING
PLANE
6.20 (0.2441)
5.80 (0.2284)
1.75 (0.0688)
1.35 (0.0532)
0.51 (0.0201)
0.31 (0.0122)
0.50 (0.0196)
0.25 (0.0099)
45°
8°
0°
0.25 (0.0098)
0.17 (0.0067)
1.27 (0.0500)
0.40 (0.0157)
COMPLIANT TO JEDEC STANDARDS MS-012-A A
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN.
012407-A
4.00 (0.1574)
3.80 (0.1497)
Figure 54. 8-Lead Standard Small Outline Package [SOIC-N]
Narrow Body (R-8)
Dimensions shown in millimeters and (inches)
ORDERING GUIDE
Model
AD603AR
AD603AR-REEL
AD603AR-REEL7
AD603ARZ 1
AD603ARZ-REEL1
AD603ARZ-REEL71
AD603AQ
AD603SQ/883B 2
AD603-EVALZ1
AD603ACHIPS
1
2
Temperature Range
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−55°C to +125°C
Package Description
8-Lead SOIC
8-Lead SOIC, 13" Reel
8-Lead SOIC, 7" Reel
8-Lead SOIC
8-Lead SOIC, 13" Reel
8-Lead SOIC, 7" Reel
8-Lead CERDIP
8-Lead CERDIP
Evaluation Board
DIE
Z = RoHS Compliant Part.
Refer to AD603 Military data sheet. Also available as 5962-9457203MPA.
Rev. H | Page 21 of 24
Package Option
R-8
R-8
R-8
R-8
R-8
R-8
Q-8
Q-8
AD603
NOTES
Rev. H | Page 22 of 24
AD603
NOTES
Rev. H | Page 23 of 24
AD603
NOTES
©2007 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
C00539-0-5/07(H)
T
T
Rev. H | Page 24 of 24