AD ADP3204 3-phase imvp-ii and imvp-iii core controller for mobile cpus Datasheet

a
FEATURES
Pin Selectable 1-, 2-, or 3-Phase Operation
Static and Dynamic Current Sharing Characteristics
Backward Compatible to IMVP-II
Superior Load Transient Response with ADOPT®
Analog Devices’ Optimal Positioning Technology
Noise-Blanking for Speed and Stability
Synchronous Rectifier Control Extends Battery Life
Smooth Output Transition During VID Code Change
Cycle-by-Cycle Current Limiting
Hiccup or Latched Overload Protection
Transient-Glitch-Free Power Good
Soft Start Eliminates Power-On In-Rush Current Surge
Two-Level Overvoltage and Reverse Voltage
Protection
APPLICATIONS
IMVP-II and IMVP-III Core DC-to-DC Converters
Fixed Voltage Mobile CPU Core DC-to-DC Converters
Notebook/Laptop Power Supplies
Programmable Output Power Supplies
3-Phase IMVP-II and IMVP-III
Core Controller for Mobile CPUs
ADP3204*
FUNCTIONAL BLOCK DIAGRAM
VCC
HYSSET
ADP3204
DSHIFT
VR
BSHIFT
DPRSHIFT
HYSTERESIS
SETTING
AND
SHIFT-MUX
BOM
DPSLP
DPRSLP
OUT3
OUT2
OUT1
CS3
CLIM
CS2
CURRENT
SENSE
MUX
EN
CS1
CS+
CS–
CORE
RAMP
REG
VID4
VID3
5-BIT VID
DAC
AND
FIXED
REF
VID
MUX
AND
REG
VID2
VID1
VID0
DPRSLP
VID
GEN
BOM
DPSLP
DACOUT
DPSLP
DPRSLP
DACRAMP
BOM
GENERAL DESCRIPTION
The ADP3204 is a 1-, 2-, or 3-phase hysteretic peak current
dc-to-dc buck converter controller dedicated to power a mobile
processor’s core. The optimized low voltage design is powered
from the 3.3 V system supply. The nominal output voltage is
set by a 5-bit VID code. To accommodate the transition time
required by the newest processors, the ADP3204 features
high speed operation to allow a minimized inductor size that
results in the fastest change of current to the output. To
further allow for the minimum number of output capacitors
to be used, the ADP3204 features active voltage positioning
with ADOPT optimal compensation to ensure a superior
load transient response. The output signals interface with a
maximum of three ADP3415 MOSFET drivers that are
optimized for high speed and high efficiency for driving both the
top and bottom MOSFETs of the buck converter. The
ADP3204 is capable of controlling the synchronous rectifiers to
extend battery lifetime in light load conditions.
PHASE
SPLITTER
VID TRANSIENT
DETECTOR AND
SHIFT SELECTOR
VR
COREGD MONITOR
COREFB
SS-HICCUP TIMER
AND OCP
DPRSLP
DRVLSD
PWRGD
ENABLE UVLO-MAIN BIAS
SD
SS
SR CONTROL
PWRGD BLANKER
OVP AND RVP
CLAMP
PM MODULE
GND
ADOPT is a trademark of Analog Devices, Inc.
*Protected by U.S.Patent No. 5,969,657; other patents pending.
REV. 0
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties that
may result from its use. No license is granted by implication or otherwise
under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700
www.analog.com
Fax: 781/326-8703
© Analog Devices, Inc., 2002
ⱕ T ⱕ 100°C, High (H) = VCC, Low (L) = 0 V, VCC = 3.3 V, SD = H, V
ADP3204–SPECIFICATIONS1 (0°C
V (V
), V = V = V = 1.25 V, C
= 100 pF, R = R = R
A
DAC
DACOUT
COREFB
REG
CS–
VID
DACRAMP
OUT1
OUT2
=
=
OUT3
100 k, COUT1 = COUT2 = COUT3 =10 pF, CSS = 0.047 F, RPWRGD = 680 to 1.2 V, RCLAMP = 5.1 k to VCC, HYSSET, BSHIFT, DSHIFT, and
DPRSHIFT are open, BOM = H, DPSLP = H, DPRLP = L, unless otherwise noted.) Current sunk by a pin has a positive sign, sourced by a pin has a
negative sign. Negative sign is disregarded for min and max values.
Parameter
Symbol
SUPPLY-UVLO-SHUTDOWN
Normal Supply Current
UVLO Supply Current
Shutdown Supply Current
ICC
ICCUVLO
ICCSD
UVLO Threshold
VCCH
VCCL
UVLO Hysteresis
Shutdown Threshold
(CMOS Input)
POWER GOOD
Core Feedback Threshold Voltage
Power Good Output Voltage
(Open-Drain Output)
Masking Time2
SOFT START/HICCUP TIMER
Charge/Discharge Current
Min
SD = L, 3.0 V ≤ VCC ≤ 3.6 V
SD = H
VCC ramping up, VSS = 0 V
VCC ramping down,
VSS floating
Typ
Max
Unit
7
11
425
mA
␮A
␮A
2.95
V
V
70
2.60
VCCHYS
55
mV
VSDTH
VCC/2
V
VCOREFBH
VPWRGD
0.9 V < VDAC < 1.675 V
VCOREFB ramping up
VCOREFB ramping down
VCOREFB ramping up
VCOREFB ramping down
VCOREFB = VDACOUT
VCOREFB = 0.8 VDACOUT
tPWRGDMSK3
ISS
Soft Start Enable/Hiccup
Termination Threshold
VSSEN
Soft Start Termination/Hiccup
Enable Threshold
VSSTERM
VID DAC
VID Input Threshold
(CMOS Inputs)
VID Input Current
(Internal Active Pull-Up)
Output Voltage
Accuracy
Conditions
VSS = 0 V
VSS = 0.5 V
VREG = 1.25 V,
VRAMP = VCOREFB = 1.27 V
VSS ramping down
VRAMP = VCOREFB = 1.27 V
VSS ramping up
1.12 VDAC
1.10 VDAC
0.88 VDAC
0.86 VDAC
0.95 VCC
0
100
1.14 VDAC
1.12 VDAC
0.90 VDAC
0.88 VDAC
VCC
0.8
␮A
␮A
–55
1.2
1.70
VVID0..4
V
V
V
V
V
V
␮s
200
300
mV
2.00
2.25
V
VCC/2
V
85
␮A
IVID0..4
VID0 to VID4 = L
VDAC
⌬VDAC/VDAC
See VID Code, Table 1
0.600
1.750
V
1.750 V ≥ VDAC ≥ 0.850 V
0.825 V ≥ VDAC ≥ 0.600 V
CDACRAMP = 100 pF
CDACRAMP = 1 nF
–1.0
–8.5
+1.0
+8.5
%
mV
␮s
␮s
k⍀
Settling Time
tDACS4
DACRAMP Inner Resistance5
RDACRAMP
–2–
3.5
25
10
REV. 0
ADP3204
Parameter
Symbol
Conditions
CORE COMPARATOR
Input Offset Voltage (Ramp-Reg)
Input Bias Current
Output Voltage
(OUT1, OUT2, and OUT3)
Propagation Delay Time
VCOREOS
IREG, IRAMP
VOUT_H
VOUT_L
tRMPOUT_PD6
VREG = 1.25 V
VREG = VRAMP = 1.25 V
VCC = 3.0 V
VCC = 3.6 V
TA = 25°C
TA = Full Range
Rise and Fall Time
(OUT1, OUT2, and OUT3)
Noise Blanking Time
CURRENT LIMIT COMPARATOR
Input Offset Voltage
Input Bias Current
Propagation Delay Time
CURRENT SENSE
MULTIPLEXER
Trans-Resistance
Hysteresis Reference Voltage
CURRENT LIMIT SETTING
Hysteresis Current
REV. 0
Typ
Max
± 1.5
±1
Unit
OUT L-H Transition
OUT H-L Transition
35
45
7
7
70
130
mV
␮A
V
V
ns
ns
ns
ns
ns
ns
VCLIMOS
ICS+, ICS–
tCLPD6
VCS– = 1.25 V
VCS+ = 1.25 V
TA = 25° C
TA = Full Range
±1
–3
55
65
mV
␮A
ns
ns
RCS1–CS+,
RCS2–CS+,
RCS3–CS+
MUX switch is ON
MUX switch is OFF
150
50
⍀
M⍀
tOUT_R7
tOUT_F7
tBLNK
Common-Mode Voltage Range
HYSTERESIS SETTING
Hysteresis Current
Min
VCS1 = VCS2 = VCS3
IRAMP_H,
–ICS+_H
VREG = 1.25 V
VRAMP = 1.23 V
IHYSSET = 10 ␮A
IHYSSET = 100 ␮A
VRAMP = 1.27 V
IHYSSET = 10 ␮A
IHYSSET = 100 ␮A
2.5
0
0
VRAMP = 1.23 V
VREG = VCS– = VCOREFB = 1.25 V
VCS+ = 1.23 V
IHYSSET = 10 ␮A
IHYSSET = 100 ␮A
VCS+ = 1.27 V
IHYSSET = 10 ␮A
IHYSSET = 100 ␮A
VCS+ = 1.23 V, BOM = L
–3–
2
V
–8
–85
–10
–100
–12
–115
␮A
␮A
8
85
10
100
VDAC
12
115
␮A
␮A
V
–27
–270
–31.5 –36
–301.5 –333
␮A
␮A
–18
–180
–21.5 –25
–201.5 –223
␮A
␮A
VHYSSET
ICS–
3.0
0.4
ADP3204
Parameter
Symbol
Conditions
Min
SHIFT SETTING
Battery-Shift Current
IRAMPB, ICS+B
VVID = 1.25 V
IBSHIFT = –100 µA, BOM = L
DPSLP = H
–92.5
Battery-Shift Reference Voltage
VBSHIFT
Deep Sleep-Shift Current
IRAMPD, ICS+D
Deep Sleep-Shift Reference
Voltage
Deeper Sleep-Shift Current
VDSHIFT
Deeper Sleep-Shift Reference
Voltage
SHIFT CONTROL INPUTS
BOM Threshold
(CMOS Input)
DPSLP Threshold
(CMOS Input)
DPRSLP Mode Threshold8
(CMOS Input)
LOW SIDE DRIVE CONTROL
Output Voltage (CMOS Output)
Output Current
IREGDPR
ICOREFBDPR8
Typ
–100
Max
Unit
–107.5
mA
VDAC
VVID = 1.25 V
IDSHIFT = –100 µA, BOM = H
DPSLP = L
–92.5
–100
V
–107.5
VDAC
IDPRSHIFT = –100 µA, DPRSLP = H
VVID = 1.25 V,
IDPRSHIFT = –100 µA,
DPRSLP = H
–90
110
–100
130
mA
V
–110
150
µA
µA
VDPRSHIFT
VDAC
V
VBOM
VCC/2
V
VDSLP
VCC/2
V
VDPRSLP
VCC/2
V
VDRVLSD
IDRVLSD
OVER/REVERSE VOLTAGE
PROTECTION CORE FEEDBACK
Overvoltage Threshold
VCOREFB, OVP9
Reverse-Voltage Threshold
VCOREFB, RVP9
Output Current
ICLAMP
(Open-Drain Output)
DPRSLP = H
DPRSLP = L
DPRSLP = H, VDRVLSD = 1.5 V
DPRSLP = L, VDRVLSD = 1.5 V
VCOREFB
VCOREFB
VCOREFB = 2.2 V, VCLAMP = 1.5 V
VCOREFB = VDAC, VCLAMP = 1.5 V
0
0.7 VCC
+0.4
–0.4
0.4
VCC
2.0
–0.3
10
2
6
V
V
mA
mA
V
V
µA
mA
NOTES
1
All limits at temperature extremes are guaranteed via correlation using standard Statistical Quality Control (SQC) methods.
2
Two test conditions: 1) PWRGD is OK but forced to fail by applying an out-of-the-Core Good-window voltage (V COREFB, BAD = 1.0 V at VVID = 1.25 V setting) to the
COREFB pin right after the moment that BOM or DPRSLP is asserted/de-asserted. PWRGD should not fail immediately only with the specified blanking delay
time. 2) PWRGD is forced to fail (V COREFB, BAD = 1.0 V at VVID = 1.25 V setting) but gets into the Core Good-window (V COREFB, GOOD = 1.25 V) right after the moment
that BOM or DPRSLP is asserted/de-asserted. PWRGD should not go high immediately only with the specified blanking delay time.
3
Guaranteed by design
4
Measured from 50% of VID code transition amplitude to the point where V DACOUT settles within ± 1% of its steady state value.
5
Measured between DACRAMP and DACOUT pins.
6
40 mVpp amplitude impulse with 20 mV overdrive. Measured from the input threshold intercept point to 50% of the output voltage swing.
7
Measured between the 30% and 70% points of the output voltage swing.
8
DPRSLP circuit meets the minimum 30 ns DPRSLPVR signal assertion requirement; guaranteed by design.
9
COREFB pin has a resistor divider to GND whose resistance is 41.3 k⍀ (typ), guaranteed by design.
–4–
REV. 0
ADP3204
Junction to Air Thermal Resistance (θJA) . . . . . . . . . . . 98°C/W
Storage Temperature Range . . . . . . . . . . . . . –65°C to +150°C
Lead Temperature (Soldering, 10 sec) . . . . . . . . . . . . . 300°C
ABSOLUTE MAXIMUM RATINGS*
Input Supply Voltage (VCC) . . . . . . . . . . . . . . . –0.3 V to +7 V
All Other Inputs/Outputs . . . . . . . . . . . . –0.3 V to VCC + 0.3 V
Junction Temperature Range . . . . . . . . . . . . . . 0°C to +150°C
*This is a stress rating only; operation beyond these limits can cause the device to
be permanently damaged.
ORDERING GUIDE
Temperature
Range
Model
ADP3204JCP-REEL 0ºC to 100ºC
ADP3204JCP-REEL7 0ºC to 100ºC
Package
Description
Package
Option
Quantity
per Reel
LFCSP-32
LFCSP-32
CP-32
CP-32
5000
1500
Table I. VID CODE
VID4
VID3
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
VID2
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
VID1
VID0
VOUT
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
1.750
1.700
1.650
1.600
1.550
1.500
1.450
1.400
1.350
1.300
1.250
1.200
1.150
1.100
1.050
1.000
0.975
0.950
0.925
0.900
0.875
0.850
0.825
0.800
0.775
0.750
0.725
0.700
0.675
0.650
0.625
0.600
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection. Although the
ADP3204 features proprietary ESD protection circuitry, permanent damage may occur on devices
subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended
to avoid performance degradation or loss of functionality.
REV. 0
–5–
ADP3204
RAMP
REG
CS–
CS+
DSHIFT
32 31 30
HYSSET
BSHIFT
DPRSHIFT
PIN CONFIGURATION
29 28 27 26 25
VID4 1
24 VCC
PIN 1
IDENTIFIER
VID3 2
23 CS3
22 CS2
VID2 3
VID1 4
21 CS1
ADP3204
VID0 5
20 OUT3
TOP VIEW
(Not to Scale)
BOM 6
19 OUT2
18 OUT1
DPSLP 7
17 GND
DACOUT
DACRAMP
SS
14 15 16
COREFB
SD
12 13
DRVLSD
10 11
CLAMP
9
PWRGD
DPRSLP 8
PIN FUNCTION DESCRIPTIONS
Pin
Mnemonic
Function
1–5
VID[4:0]
Voltage Identification Inputs. These are the VID inputs for logic control of the programmed
reference voltage that appears at the DACOUT pin, and, via external component configuration, is used for setting the output voltage regulation point. The VID pins have a specified internal
pull-up current that, if left open, will default the pins to a logic high state. The VID code does not set
the DAC output voltage directly but through a transparent latch that is clocked by the BOM pin’s
GMUXSEL signal rising and falling edge.
6
BOM
Battery Optimized Mode Control (Active Low). This digital input pin corresponds to the system’s
GMUXSEL signal that corresponds to Battery Optimized Mode of the CPU operation in its active
low state and Performance Optimized Mode (POM) in its deactivated high state. The signal also
controls the optimal positioning of the core voltage regulation level by offsetting it downward in
Battery Optimized Mode according to the functionality of the BSHIFT and RAMP pins. It is also
used to initiate a masking period for the PWRGD signal whenever a GMUXSEL signal transition occurs.
7
DPSLP
Deep Sleep Mode Control (Active Low). This is a digital input pin corresponding to the system’s
STPCPU signal that, in its active state, corresponds to Deep Sleep Mode of the CPU operation,
which is a subset operating mode of either BOM or POM operation. The signal controls the optimal
positioning of the core voltage regulation level by offsetting it downward according to the functionality of the DSHIFT and RAMP pins.
8
DPRSLP
Deeper Sleep Mode Control (Active High). This is a digital input pin corresponding to the system’s
DPRSLPVR signal corresponding to Deeper Sleep Mode of the CPU operation. When the signal
when it is activated it controls the DAC output voltage by disconnecting the VID signals from the
DAC input and setting a specified internal Deeper Sleep code instead. At de-assertion of the DPRSLPVR
signal, the DAC output voltage returns to the voltage level determined by the externalVID code.
The DPRSLPVR signal is also used to initiate a blanking period for the PWRGD signal to disable its
response to a pending dynamic core voltage change that corresponds to the VID code transition.
9
PWRGD
Power Good (Active High). This open-drain output pin, via the assistance of an external pull-up
resistor to the desired voltage, indicates that the core voltage is within the specified tolerance
of the VID programmed value, or else is in a VID transition state as indicated by a recent state
transition of either the BOM or DPRSLP pins. PWRGD is deactivated (pulled low) when the IC is
disabled in UVLO mode, or starting up, or the COREFB voltage is out of the core power-good
window. The open-drain output allows external wired ANDing (logical NORing) with other open
drain/collector power-good indicators.
10
SD
Shutdown (Active Low). This is a digital input pin coming from a system signal that, in its active
state shuts down the IC operation, placing the IC in its lowest quiescent current state for maximum
power savings.
–6–
REV. 0
ADP3204
PIN FUNCTION DESCRIPTIONS (continued)
Pin
Mnemonic
Function
11
CLAMP
Clamp (Active High). This is open-drain output pin, via the assistance of an external pull-up
resistor, indicates that the core voltage should be clamped for its protection. To allow the highest
level of protection, the CLAMP signal is developed using both a redundant reference and a redundant feedback path with respect to those of the main regulation loop. In a preferred and more
conservative configuration, the core voltage is clamped by an external FET. The initial protection
function is served when it is activated by detection of either an overvoltage or a reverse-voltage
condition on the COREFB pin. Due to loss of the latched signal at IC power-off, a backup protection function is served by connecting the pull-up resistor to a system “ALWAYS” regulator output
(e.g., V5_ALWAYS). If the external FET is used, this implementation will keep the core voltage
clamped until the ADP3204 has power reapplied, thus keeping protection for the CPU even after a
hard-failure power-down and restart (e.g., a shorted top or bottom FET).
12
DRVLSD
Drive-Low Shutdown (Active Low). In its active state, this digital output pin indicates that the
lower FET of the core VR should be disabled. In the suggested application schematic, this pin is
directly connected to the pin of the same name on the ADP3415 or other driver IC. Drive-low
shutdown is normally activated by the DPRSLP signal corresponding to a light load condition, but a
number of dynamic conditions can override the control of this pin as needed.
13
SS
Soft Start. The output of this analog I/O pin is a controlled current source used to charge or
discharge an external grounded capacitor; the input is the detected voltage that is indicative of
elapsed time. The pin controls the soft start time of the IC as well as the hiccup cycle time during
overload, including but not limited to short circuit. Hiccup operation was added to reduce short
circuit power dissipation by more than an order of magnitude, while still allowing an automatic
restart when the failure mode ceased. The hiccup operation can be overwritten and changed to
latched-off operation by clamping the SS pin voltage to a voltage level somewhere above ~ 0.2 V.
In this configuration, the controller does not restart after a hiccup cycle is initiated, but stays latched off.
14
COREFB
Core Feedback. This high impedance analog input pin is used to monitor the output voltage for
setting the proper state of the PWRGD and CLAMP pins. It is generally recommended to RC-filter
the noise from the monitored core voltage, as suggested by the application schematic.
15
DACRAMP
DAC Output Ramp Rate Setting. The rate at which the DAC output voltage can ramp up or down
from one voltage to another when the VID code changes can be controlled by an external
DACRAMP capacitor connected from this pin to the DACOUT pin. The time constant of the
DACOUT voltage variation is determined by the internal resistance appearing across the
DACRAMP and DACOUT pins, and the capacitance of the DACRAMP capacitor. Not having any
DACRAMP capacitor connected to these pins results in the fastest rate. Use of the DACRAMP rate
control and the Deeper Sleep Shift adjustment features are exclusive.
16
DACOUT
Digital-to-Analog Converter Output of the VID input. This output voltage is the VID controlled
reference voltage whose primary function is to determine the output voltage regulation point.
17
GND
Ground
18–20
OUT1–3
Outputs to Driver 1–3. These digital output pins are used to command the state of theswitched
nodes via the drivers. They should be connected to the IN pin of the drivers of the appropriate channels.
21
CS1
Current Sense, Channel 1. This high impedance analog input pin is used for providing negative
feedback of the current information for the first channel.
22
CS2
Current Sense, Channel 2. This high impedance analog input pin is used to provide negative
feedback of the current information for the second channel. The pin is also used to determine
whether the chip is acting as a single or a multiphase controller. If the CS2 pin is tied to VCC but
not to a sense resistor, then three-phase operation is disabled. In this condition, the second phase
output signal (OUT2) is not switching but stays static low; the first and third phase output signals
(OUT1 and OUT2) are switching in phase. It’s the user’s discretion to use only one or both of the
two signals to drive a single- or dual-channel power stage.
23
CS3
Current Sense, Channel 3. This high impedance analog input pin is used to provide negative
feedback of the current information for the third channel. The pin is also used to determine
whether the chip is acting as a dual- or three-phase controller. If the pin is tied to VCC but not to a
sense resistor, then three-phase operation is disabled; the chip works as a dual-phase controller. In
this condition, the third phase output signal (OUT3) is not switching but stays static low; the first
and second phase output signals (OUT1, OUT2) are interleaved out-of-phase signals. In singlephase operation, CS3 should be left open instead of being tied to VCC.
REV. 0
–7–
ADP3204
PIN FUNCTION DESCRIPTIONS (continued)
Pin
Mnemonic
Function
24
VCC
Power Supply. This should be connected to the system’s 3.3 V power supply output.
25
RAMP
Regulation Ramp Feedback Input. The RAMP pin voltage is compared against the REG pin for
cycle-by-cycle switching response. Several switched current sources also appear at this input: the
cycle-by-cycle hysteresis-setting switched current programmed by the HYSSET pin, the BOM shift
current programmed by the BSHIFT pin, and the Deep Sleep shift current programmed by the
DSHIFT pin. The external resistive termination at this pin sets the magnitude of the hysteresis
applied to the regulation loop.
26
REG
Regulation Voltage Summing Input. This is a high impedance analog input pin into which the
voltage reference of the feedback loop allows the summing of both the DACOUT voltage and the
core voltage for programming the output resistance of the core voltage regulator. This is also the pin
at which an optimized transient response can be tailored using Analog Devices’ patented ADOPT
design technique.
27
CS+
Current Limit Positive Sense. This high impedance analog I/O pin is multiplexed between either of
the three current-sense inputs during the high state of the OUT pin of the respective channel.
During the common off-time of both channels, the pin voltage reflects the average of the three
channels. The multiplexed current sense signal is passed to the core comparator through an external
resistive termination connected from this pin to the RAMP pin. The external (RAMP) resistor sets
the magnitude of the hysteresis applied to the regulation loop.
28
CS–
Current Limit Negative Sense. This high impedance analog input pin which is normally Kelvin
connected to the negative node of the current sense resistor(s) via a current-limit programming
resistor. A hysteretically-controlled current—three times the current programmed at the HYSSET
pin—also flows out of this pin and develops a current-limit-setting voltage across that resistor, which
must then be matched by the inductor current flowing in the current sensing resistor in order to
trigger the current limit function. When triggered, the current flowing out of this pin is reduced to
two-thirds of its previous value, producing hysteresis in the current limiting function.
29
HYSSET
Hysteresis Set. This is an analog I/O pin whose output is the VID reference voltage and whose input
is a current that is programmed by an external resistance to ground. The current is used in the IC to
set the hysteretic currents for the Core Comparator and the Current Limit Comparator. Modification of the resistance will affect both the hysteresis of the feedback regulation, and the current limit
set point and hysteresis.
30
DSHIFT
Deep Sleep Shift. This is an analog I/O pin whose output is the VID reference voltage and whose
input is a current that is programmed by an external resistance to ground. The current is used in the
IC to set a switched bias current out of the RAMP pin, depending on whether it is activated by the
DSLP signal. When activated, this added bias current creates a downward shift of the regulated core
voltage to a predetermined optimum level for regulation corresponding to Deep Sleep Mode of
CPU operation. The use of the DACOUT voltage as the reference makes the Deep Sleep offset a
fixed percentage of the VID setting, as required by specifications.
31
BSHIFT
Battery Optimized Mode (BOM) Shift. This is an analog I/O pin whose output is the VID reference
voltage and whose input current is programmed by an external resistance to ground. The current is
used in the IC to set a switched bias current out of the RAMP pin, depending on whether it is
activated by the BOM signal. When activated, this added bias current creates a downward shift
of the regulated core voltage to a predetermined optimum level for regulation corresponding to
Battery Optimized Mode of CPU operation. The use of the DACOUT voltage as the reference
makes the DSHIFT a fixed percentage of the VID setting, as required by specifications.
–8–
REV. 0
ADP3204
PIN FUNCTION DESCRIPTIONS (continued)
Pin
Mnemonic
Function
32
DPRSHIFT
Deeper Sleep Shift. This is an analog I/O pin whose output is a fixed voltage reference and whose
input current is programmed by an external resistor to ground. The current is used to set two
switched bias currents that flow into both the REG and COREFB pins, depending on the DPRSLP
signal. When activated, the REG pin bias current creates an upward shift of the regulated core
voltage from the internally set (default) Deeper Sleep value to the voltage level specified by the CPU
Deeper Sleep operation. The COREFB bias current creates the same amount of downward shift of
the COREFB voltage is. The shifted back COREFB voltage compared against the internally set
Deeper Sleep voltage to create Power Good information. Use of the Deeper Sleep Shift adjustment
and the DACRAMP rate control features are exclusive.
V_5S
D3
BAR43S
C33
0.1F
C43
3.3F
V_DC
R106
2.7
C32
1F
2x0.1F
ADP3415
1 IN
C2
0.1µF
RA243 1%
C3
RDPRSHIFT
REG 26
RAMP 25
CS– 28
CS+ 27
DSHIFT 30
HYSSET 29
BSHIFT 31
DPRSHIFT 32
CS3
23
1 IN
CS2
22
2 SD
3 DRVLSD
VID1
CS1
21
VID0
OUT3
20
BOM
OUT2
19
DPSLP
OUT1
18
GND
17
16 DACOUT
14 COREFB
13 SS
12 DRVLSD
11 CLAMP
10 SD
DPRSLP
15 DACRAMP
VR_ VID2
VR_PWRGD
2x0.1F
R17
0
SW
8
4 DLY
GND
7
5 VCC
DRVL
6
JP1
RD
R9
1.05k
1% 1.05k
1%
C6
1000pF
C9
10pF
V_5S
C41
3.3F
R16
0
Q4
IR7811W
L2
0.32H
RCS2
3m
R14
D5
MBRS130LT3 10
C21
0.018F
Q12
IR7822
D1
BAR43S
C13
0.1F
V_DC
R104
2.7
ADP3415
R90
300k
Q3
IR7811W
Q11
IR7822
CSS
0.047F
C27 . . . C30
4x10F
BST 10
DRVH 9
CORE_ON
VCC-VID (1.2V)
V_DC
C32
1F
ADP3415
24
4
ADP3204
C23
0.1F
VCC
VID3
9 PWRGD
Q14
IR7822
D2
BAR43S
R102
2.7
VID2
BAR43S
R15
10
C31
0.018F
C42
3.3F
3
D90
RCS3
3m
D6
MBRS130LT3
Q13
IR7822
V_5S
2
3k
L3
0.32H
4700pF
VR_ VID3
V_3S
GND 7
DRVL 6
COC
OPTIONAL: DEEPERSLEEP
VOLTAGE ADJUSTMENT
VR_ VID1
5
VR_ VID0
220k 6
GMUXSEL
(IMVP-II ONLY)
7
STP_CPU
8
DPRSLPVR
4 DLY
5 VCC
Q6
IR7811W
RB 54.9k
C4
10pF
10pF
VID4
8
RC
1.5k 1%
RDSHIFT 8.87k 1%
1
SW
R18
0
RCL 576 1%
RBSHIFT 21.5k 1%
VR_ VID4
C1
1µF
Q5
IR7811W
DRVH 9
3 DRVLSD
R4 2.7
RHYSSET 21.5k 1%
BST 10
2 SD
V_3S
C37 . . . C40
4x10F
1
IN
2
SD
3
DRVLSD
4
DLY
GND
7
5
VCC
DRVL
6
C12
1F
2x0.1F
C17 . . . C20
4x10F
BST 10
Q1
IR7811W
DRVH 9
SW 8
Q2
IR7811W
L1
0.32H
RCS1
3m
R23
D4
MBRS130LT3 10
SET: LATCHED OCP
OPEN: HICCUP OCP
Q9
IR7822
C11
0.018F
Q10
IR7822
V5
R20
5.1k
Figure 1. Typical Application
REV. 0
–9–
VCC CORE
Q7
IR7807V
C44...C49
6x220F
ADP3204–Typical Performance Characteristics
10000
HIGH
1000
UVLO MODE
PWRGD
SUPPLY CURRENT – A
NORMAL OPERATING MODE
100
SHUTDOWN MODE
LOW
10
0
20
40
60
AMBIENT TEMPERATURE – C
80
0.15
100
0.05
0.1
0
0.05
0.1
0.15
RELATIVE CORE VOLTAGE – %
TPC 1. Supply Current vs. Temperature
TPC 4. Power Good vs. Relative Core Voltage Variation
1.77
10
+1%
SOFT START TIME – ms
DAC OUTPUT – V
1.76
FULL SCALE
1.75
1.74
1
0.1
0.01
1%
1.73
0
20
40
60
AMBIENT TEMPERATURE – C
80
0.001
100
0.615
108.0
DAC OUTPUT – V
NORMALIZED BLANKING TIME – %
110.0
+8.5mV
0.605
ZERO SCALE VID [11111]
0.595
100
TPC 5. Soft Start Timing vs. Timing Capacitor
0.620
0.600
10
SOFT START CAPACITANCE – nF
TPC 2. DAC Output Voltage vs. Temperature
0.610
1
0.1
8.5mV
0.590
0.585
106.0
104.0
102.0
100.0
98.0
96.0
94.0
92.0
0.580
0
20
40
60
80
90.0
100
0
AMBIENT TEMPERATURE – C
TPC 3. DAC Output Voltage vs. Temperature
20
40
60
80
AMBIENT TEMPERATURE – C
100
TPC 6. PWRGD Blanking Time vs. Temperature
Normalized to 25°C
–10–
REV. 0
ADP3204
0
110
ICS – CURRENT LIMIT THRESHOLD – A
HYSTERESIS CURRENT – A
OUT = HIGH, RHYS = 17k
OUT = HIGH, RHYS = 170k
0
OUT = LOW, RHYS = 170k
OUT = LOW, RHYS = 17k
OUT = LOW, RHYSSET = 170k
–50
OUT = HIGH, RHYSSET = 170k
–100
–150
–200
OUT = LOW, RHYSSET = 17k
–250
–300
OUT = HIGH, RHYSSET = 17k
–350
–110
0
20
40
60
80
0
100
TPC 7. Core Hysteresis Current vs. Temperature
Power Conversion Control Architecture
Driving of the individual channels is accomplished using external
drivers, such as the ADP3415. One PWM interface pin per
channel, OUT1, OUT2, and OUT3, is provided. A separate
pin, DRVLSD, commands the driver to enable or disable synchronous rectifier operation during the off time of each channel. The
same DRVLSD pin is connected to all three drivers.
The ADP3204 utilizes hysteretic control. The resistor from
the HYSSET pin to ground sets up a current that is switched
bidirectionally into a resistor interconnected between the RAMP
and CS+ pins. The switching of this current sets the hysteresis.
In a multichannel configuration, the hysteretic control requires
multiplexing information in all channels. The inductor current
of the channel that is driven high is controlled against the upper
hysteresis limit. During the common offtime of the channels,
the inductor currents are averaged together and compared against
the lower hysteresis limit. This proprietary offtime averaging
technique serves to eliminate a systematic offset that otherwise
appears in a fully multiplexed hysteretic control system.
REV. 0
60
80
100
Feedback/Current Sensing
Featuring a new proprietary 1-, 2-, or 3-channel buck converter
hysteretic control architecture developed by Analog Devices, the
ADP3204 is the optimal core voltage control solution for both
IMVP-II and IMVP-III generation microprocessors. The complex
multitiered regulation requirements of either IMVP specification are easily implemented with the highly integrated functionality of this controller.
As with all ADI products for core voltage control, the controller
is compatible with ADOPT compensation, which provides the
optimum output voltage containment within a specified voltage
window or along a specified load line using the fewest possible
output capacitors. The inductor ripple current is kept at a fixed
programmable value while the output voltage is regulated with
fully programmable voltage positioning parameters, which can
be tuned to optimize the design for any particular CPU regulation specification. By controlling the ripple current rather than
the ripple voltage, the frequency variations associated with
changes in output impedance for standard ripple regulators will
not appear.
40
TPC 8. Current Limit Threshold vs. Temperature
THEORY OF OPERATION
Overview
Compensation
20
AMBIENT TEMPERATURE – C
TEMPERATURE – C
Accurate current sensing is needed to accomplish output voltage
positioning accurately, which, in turn, is required to allow the
minimum number of output capacitors to be used to contain
transients. A current sense resistor is used between each inductor
and the output capacitors. To allow the control to operate
without amplifiers, the negative feedback signal is multiplexed
from the inductor or upstream side of the current sense resistors,
and a positive feedback signal, if needed for load-line tuning, is
taken from the output or downstream side.
Output Voltage Programming by VID, Offsets, and Load Line
In the IMVP-II and IMVP-III specifications, the output voltage
is a function of both the core current (according to a specified
load line) and the system operating mode (i.e., performance or
battery optimized, normal or deep sleep clocking state, or deeper
sleep). The VID code programs the “nominal” core voltage.
The core voltage decreases as a function of load current along
the load line, which is synonymous with an output resistance of
the power converter. The core voltage is also offset by a dc
value—usually specified as a percentage—depending on the
operating mode. The voltage offset is also called a “shift.”
Two pins, BSHIFT and DSHIFT, are used to program the
magnitude of the voltage shifts. The voltage shifts are accomplished by injecting current at the node of the negative input pin
of the feedback comparator. Resistive termination at the pins
determines the magnitude of the voltage “shifts.”
Two other pins, BOM and DPSLP, are used to activate the
respective two shifts only in their active low states. In the
ADP3204, the shifts are mutually exclusive, with the Deep
Sleep shift (controlled by the DPSLP and DSHIFT pins) being
the dominant one. Another pin, DPRSLP, eliminates both
shifts only in its active high state. Its assertion corresponds to
the Deeper Sleep operating mode.
Current Limiting
The current programmed at the HYSSET pin and a resistor
from the CS– pin to the common node of the current sense
resistors set the current limit. If the current limit threshold is
triggered, a hysteresis is applied to the threshold so that hysteretic
control is maintained during a current limited operating mode.
–11–
ADP3204
SoftStart and Hiccup
A capacitor from the SS pin to ground determines both the soft
start time and the frequency at which hiccup will occur under a
continuous short circuit or overload.
System Signal Interface
Several pins of the ADP3204 are meant to connect directly to
system signals. The VID pins connect to the system VID
control signals. The DPRSLP pin connects to the system’s
DPRSLPVR signal. The DPSLP pin connects to the system’s
DPSLP or STPCPU signal. The BOM signal connects to the
system’s GMUXSEL signal. In an IMVP-II system, the
GMUXSEL signal precedes any VID code change with a few
nanoseconds, while in an IMVP-III system, it follows it with a
maximum 12 µs delay. To comply with both specifications, the
ADP3204 has a VID register in front of the DAC inputs that is
written by a short pulse generated at the rising or falling edge of
the GMUXSEL signal. In an IMVP-II configuration, if the
external VID multiplex settling time is longer than the internal
VID register’s write pulsewidth, then the insertion of an external
RC delay network in the GMUXSEL signal path (in front of the
BOM pin) is recommended. The Intel specification calls for
maximum 200 ns VID code setup time. This specification can
be met with a simple RC network that consists of only a 220 kΩ
resistor and no external capacitor, just the BOM pin’s capacitance.
Undervoltage Lockout
configuration, the latched off state of the system would be
indicative of a system failure. The overvoltage/reverse voltage
protective means is via not allowing the continued application of
energy to the CPU core. The design objective should be, however,
to ensure that the CPU core could safely absorb the remaining
energy in the power converter, since this energy is not clamped
as in the preferred configuration.
LAYOUT CONSIDERATIONS
Advantages in PCB Layout
Analog Devices provides ADP3204/3415 as a dedicated threephase power management solution for IMVP-III Intel P4
mobile core supply.
This three-phase solution separates the controller (ADP3204)
and the MOSFET drivers (ADP3415). Today, most motherboards
only leave small pieces of PCB area for the power management
circuit. Therefore, the separation of the controller and the
MOSFET drivers gives much greater freedom in layout than
any single chip solution.
Meanwhile, the separation also provides the freedom to place
the analog controller in a relatively quiet area in the motherboard.
This can minimize the susceptibility of the controller to injected
noise. Any single chip solution with a high speed loop design
will suffer larger susceptibility to jitter that appears as modulation
of the output voltage.
The ADP3204’s supply pin, VCC, has undervoltage lockout
(UVLO) functionality to ensure that if the supply voltage is too
low to maintain proper operation, the IC will remain off and in
a low current state.
The ADP3204 maximizes the integration of IMVP-III features.
Therefore, no additional externally implemented functions are
required to comply with IMVP-III specifications. This saves
PCB area for component placement on the motherboard.
Overvoltage Protection (OVP) and Reverse Voltage
Protection (RVP)
PCB Layout Consideration for ADP3204/3415
The ADP3204 features a comprehensive redundantly monitored
OVP and RVP implementation to protect the CPU core against
an excessive or reverse voltage, e.g., as might be induced by a
component or connection failure in the control or power stage.
Two pins are associated with the OVP/RVP circuitry—a pin for
output voltage feedback, COREFB, which is also used for
power good monitoring but not for voltage regulation, and an
output pin, CLAMP.
The CLAMP pin defaults to a low state at startup of the
ADP3204 and remains low until an overvoltage or reverse
voltage condition is detected. If either condition is detected, the
CLAMP signal is asserted and latched high.
For maximum and fastest protection, the CLAMP pin should
be used to drive the gate of a power MOSFET whose drain
source is connected across the CPU core voltage. Detection of
overvoltage or reverse voltage will clamp the core voltage to
essentially zero, thus quickly removing the fault condition and
preventing further energy from being applied to the CPU core.
For a less comprehensively protective and less costly solution,
the CLAMP pin may be used to latch the disconnection of
input power. The latch should be powered whenever any input
power source is present. Typically, such a latching circuit is
already present in a system design, so it becomes only a matter
of allowing the CLAMP pin to also trigger the latch. In this
The following guidelines are recommended for optimal performance of the ADP3204 and ADP3415 in a power converter.
The circuitry is considered in three parts: the power switching
circuitry, the output filter, and the control circuitry.
Placement Overview
1. For ideal component placement, the output filter capacitors
will divide the power switching circuitry from the control
section. As an approximate guideline considered on a singlesided PCB, the best layout would have components aligned
in the following order: ADP3415, MOSFETs and input
capacitor, output inductor, current sense resistor, output
capacitors, control components, and ADP3204. Note that
the ADP3204 and ADP3415 are completely separated for an
ideal layout, which is impossible with a single-chip solution.
This keeps the noisy switched power section isolated from
the precision control section and gives more freedom in the
layout of the power switching circuitry.
2. Whenever a power dissipating component (e.g., a power
MOSFET) is soldered to a PCB, the liberal use of vias, both
directly on the mounting pad if possible and immediately
surrounding it, is recommended. Two important reasons for
this are: improvement of the current rating through the vias
(if it is a current path) and improved thermal performance,
especially if there is opportunity to spread the heat with a
plane on the opposite side of the PCB.
–12–
REV. 0
ADP3204
Power Switching Circuitry
Control Circuitry
ADP3415, MOSFETs, and Input Capacitors
ADP3204, Control Components
3.
Locate the ADP3415 near the MOSFETs so that the loop
inductance in the path of the top gate drive returned to the
SW pin is small, and similarly for the bottom gate drive
whose return path is the ground plane. The GND pin
should have at least one very close via into the ground plane.
12. If the ADP3204 cannot be placed as previously recommended, care should be taken to keep the device and
surrounding components away from radiation sources
(e.g., from power inductors) and capacitive coupling from
noisy power nodes.
4.
Locate the input bypass MLC capacitors close to the MOSFETs
so that the physical area of the loop enclosed in the electrical
path through the bypass capacitor and around through the
top and bottom MOSFETs (drain-source) is small and wide.
This is the switching power path loop.
13. Noise immunity can be improved by the use of a devoted
signal ground plane for the power controller and its
surrounding components. Space for a ground plane might
readily be available on a signal plane of the PCB since it is
often unused in the vicinity of the power controller.
5.
Make provisions for thermal management of all the MOSFETs.
Heavy copper and wide traces to ground and power planes
will help to pull the heat out. Heat sinking by a metal tap
soldered in the power plane near the MOSFETs will help.
Even just a small airflow can help tremendously. Paralleled
MOSFETs to achieve a given resistance will help spread the heat.
14. If critical signal lines (i.e., signals from the current sense
resistor leading back to the ADP3204) must cross through
power circuitry, it is best if a signal ground plane can be
interposed between those signal lines and the traces of the
power circuitry. This serves as a shield to minimize noise
injection into the signals.
6.
An external Schottky diode (across the bottom MOSFET)
may increase efficiency by a small amount (< ~1%), depending on its forward voltage drop compared to the MOSFET’s
body diode at a given current; a MOSFET with a built-in
Schottky is more effective. For an external Schottky, it should
be placed next to the bottom MOSFET or it may not be
effective at all.
15. Absolutely avoid crossing any signal lines over the switching
power path loop, described previously.
7.
The VCC bypass capacitor should be close to the VCC pin
and connected on either a very short trace to the GND pin
or to the GND plane.
Output Filter
Output Inductor and Capacitors, Current Sense Resistor
8.
Locate the current sense resistors very near to the output
voltage plane.
9.
The load-side heads of sense resistors should join as closely
as possible for accurate current signal measurement of
each phase.
17. The RC filter used for the current sense signal should be
located near the control components as this serves the
dual purpose of filtering out the effect of the current sense
resistors’ parasitic inductance and the noise picked up along
the routing of the signal. The former purpose is achieved by
having the time constant of the RC filters approximately
matched to that of the sense resistors, and is important
for maintaining the accuracy of the current signal.
APPLICATION INFORMATION
Theoretical Background
10. PCB trace resistances from the current sense resistors to
the regulation point should be minimized, known (calculated or measured), and compensated for as part of the
design if it is significant. (Remote sensing is not sufficient
for relieving this requirement.) A square section of 1-ounce
copper trace has a resistance of ~0.5 mΩ, which adds to
the specified DC output resistance of the power converter.
The output capacitors should similarly be close to the
regulation point and well tied into power planes as impedance here will add to the “AC output resistance” (i.e., the
ESR) that is implicitly specified as well.
This application section presents the theoretical background for
multiphase dc-to-dc converters using the ADP320x family of
controllers for mobile CPUs. Members of that family control
multiphase ripple regulators (also called hysteretic regulators) in
a configuration that allows employing ADOPT, Analog Devices’
optimal voltage positioning technique to implement the desired
output voltage and load line, both statically and dynamically, as
required by Intel’s IMVP-II and IMVP-III specifications.
Single-Phase Hysteretic Regulator with ADOPT
11. Whenever high currents must be routed between PCB
layers, vias should be used liberally to create parallel current
paths so that the resistance and inductance are minimized
and the via current rating is not exceeded.
REV. 0
16. Accurate voltage positioning depends on accurate current
sensing, so the control signals that monitor the voltage
differentially across the current sense resistor should be
Kelvin-connected. Please refer to ADI Evaluation Board of
the ADP3204 and its documentation for control signal
connection with sense resistors.
Figure 2 shows the conventional single-phase hysteretic
regulator and the characteristic waveforms. The operation is as
follows. During the time the upper transistor, Q1, is turned on,
the inductor current, IL, and also the output voltage, VOUT,
increase. When VOUT reaches the upper threshold of the
hysteretic comparator, Q1 is turned off, Q2 is turned on, and
the inductor current and the output voltage decrease. The cycle
repeats after VOUT reaches the lower threshold of the hysteretic
comparator.
–13–
ADP3204
and
VOUT
VIN
2
VH
Q1
L
VSW
VOUT
IL
CO
Q2
COC =
+
RE
LOAD
t
VSW
RCS =
t
IL
t
Since there is no voltage error amplifier in the hysteretic
regulator, its response to any change in the load current or the
input voltage is virtually instantaneous. Therefore, the hysteretic
regulator represents the fastest possible dc-to-dc converter. A
slight disadvantage of the conventional hysteretic regulator is
that its frequency varies with the input and output voltages. In a
typical mobile CPU converter application, the worst-case
frequency variation due to the input voltage variation is in the
order of 30%, which is usually acceptable. In the simplest
implementation of the hysteretic converter, shown in Figure 2,
the frequency also varies proportionally with the ESR, RE, of the
output capacitor. Since the initial value is often poorly controlled, and the ESR of electrolytic capacitors also changes with
temperature and age, practical ESR variations can easily lead to
a frequency variation in the order of three to one. However, a
modification of the hysteretic topology eliminates the dependence of the operating frequency on the ESR. In addition, the
modification allows the optimal implementation, ADOPT, of
Intel’s IMVP-II and IMVP-III load-line specifications. Figure 3
shows the modified hysteretic regulator.
VIN
VSW
Q2
VH
L IL
RCS
COC
VOUT
CO
RC
+
RE
RD
VREF
Figure 3. Modified Hysteretic Regulator with ADOPT
The implementation requires adding a resistive divider (RC
and RD) between the reference voltage and the output, and
connecting the tap of the divider to the noninverting input
of the hysteretic comparator. A capacitor, COC, is placed
across the upper member (RC) of the divider.
It is easily shown that the output impedance of the converter can be no less than the ESR of the output capacitor.
A straightforward derivation demonstrates that the output
impedance of the converter in Figure 3 can be minimized to
equal the ESR, RE, when the following two equations are
valid (neglecting PCB trace resistance for now):
RE
R
= 1+ D
RCS
RC
RE
R
1+ D
RC
(3)
This is the ADOPT configuration and design procedure that
allows the maximum possible ESR to be used while meeting a
given load-line specification.
Figure 2. Conventional Hysteretic Regulator and
Its Characteristic Waveforms
Q1
(2)
From (Equation 2), the series resistance is:
VH
VREF
CO RE
RCS RD
(1)
It can be seen from Equation 3 that unless RD is zero or RC is
infinite, RCS will always be smaller than RE. An advantage of
the circuit in Figure 3 is that if we select the ratio RD/RC well
above unity, the additional dissipation introduced by the series
resistance RCS will be negligible. Another interesting feature of
the circuit in Figure 3 is that the ac voltage across the two
inputs of the hysteretic comparator is now equal only to the ac
voltage across RCS. This is due to the presence of the capacitor
COC, which effectively couples the ac component of the output
voltage to the noninverting input voltage of the comparator.
Since the comparator sees only the ac voltage across RCS, in the
circuit in Figure 3 the dependence of the switching frequency
on the ESR of the output capacitor is completely eliminated.
Equation 4 presents the expression for the switching frequency.
f =
RCS (VIN −VOUT )
VOUT
LVH
VIN
(4)
Multiphase Hysteretic Regulator with ADOPT
Multiphase converters have very important advantages, including reduced rms current in the input filter capacitor (allowing
the use of a smaller and less expensive device), distributed heat
dissipation (reducing the hot spot temperature and increasing
reliability), higher total power capability, increased equivalent
frequency without increased switching losses (allowing the use
of smaller equivalent inductances, and thereby shortening the
load transient time), and reduced ripple current in the output
capacitor (reducing the output ripple voltage and allowing the
use of a smaller and less expensive output capacitor). Also, they
have some disadvantages, which should be considered when
choosing the number of phases. Those disadvantages include the
need for more switches and output inductors than in a single-phase
design (leading to higher cost than a single-phase solution, at
least below a certain power level), more complex control, and
the possibility of uneven current sharing among the phases.
The family of ADP320x controllers alleviates two of the above
disadvantages of multiphase converters. It presents a simple and
cost-effective control solution, and provides perfect current
sharing among the phases. Figure 4 shows a simplified block
diagram of a three-phase converter using the control principle
implemented with the ADP3204, the three-phase member of
the ADP320x family.
As Figure 4 shows, in the multiphase configuration, the ripple
current signal is multiplexed from all channels. During the on
time of any given channel, its current is compared to the upper
threshold of the hysteretic comparator. When the current
reaches the upper threshold, the control FET of that channel is
–14–
REV. 0
ADP3204
turned off. During the common off time of all channels, their
currents are averaged and compared to the lower threshold.
When the averaged channel current reaches the lower threshold,
the hysteretic comparator changes state again, and turns on the
control FET of the next channel, as selected by the phase
splitter logic. This control concept ensures that the peak
currents of all channels will be the same, and therefore the
channel currents will be perfectly balanced. The ADOPT
compensation can be used the same way as in the single-phase
version discussed previously.
Since due to second-order effects, the detailed design of a
multiphase converter with the ADP320x family is rather
complex, a design aid using MathSoft’s MathCAD™ program
has been developed. Please contact ADI for further information.
PHASE 1
L1
IL1
RCS1
PHASE 2
L2
IL2
RCS2
VIN
VOUT
RE
CO
PHASE 3
PHASE
SPLITTER
OUT 3
OUT 2
OUT 1
L3
IL3
HYSTERETIC
CORE
COMPARATOR
RCS3
CURRENT
SENSE MUX
CS1
CS2
CS3
COC
RD
VREF
Figure 4. 3-Phase Modified Hysteretic Regulator with ADOPT
MathCAD is a trademark of MathSoft.
REV. 0
–15–
RC
LOAD
ADP3204
OUTLINE DIMENSIONS
32-Lead Frame Chip Scale Package [LFCSP]
(CP-32)
5.00
BSC SQ
0.60 MAX
PIN 1
INDICATOR
0.50
BSC
4.75
BSC SQ
TOP
VIEW
0.50
0.40
0.30
32 1
3.25
3.10 SQ
2.95
BOTTOM
VIEW
17
16
9
8
3.50
REF
0.70 MAX
0.65 NOM
0.05 MAX
0.02 NOM
SEATING
PLANE
0.30
0.23
0.18
0.25 REF
COPLANARITY
0.08
COMPLIANT TO JEDEC STANDARDS MO-220-VHHD-2
PRINTED IN U.S.A.
1.00
0.90
0.80
PIN 1
INDICATOR
0.60 MAX
25
24
12 MAX
C02841–0–11/02(0)
Dimensions shown in millimeters
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