MIC2128 75V, Synchronous Buck Controller Featuring Adaptive On-Time Control with External Soft Start Features General Description • Hyper Speed Control® Architecture Enables: - High Input to Output Voltage Conversion Ratio Capability - Any Capacitor™ Stable - Ultra-Fast Load Transient Response • Wide 4.5V to 75V Input Voltage Range • Adjustable Output Voltage from 0.6V to 30V • 270 kHz to 800 kHz Programmable Switching Frequency • Built-in 5V Regulator for Single-Supply Operation • Auxiliary Bootstrap LDO for Improving System Efficiency • Internal Bootstrap Diode • Adjustable Soft-Start Time • Enable Input and Power Good Output • Programmable Current Limit • Hiccup Mode Short-Circuit Protection • Internal Compensation and Thermal Shutdown • Supports Safe Start-Up into a Prebiased Output The MIC2128 is a constant-frequency synchronous buck controller featuring a unique adaptive on-time control architecture with external soft start. The MIC2128 operates over an input voltage range from 4.5V to 75V.The output voltage is adjustable down to 0.6V with a guaranteed accuracy of ±1%. The device operates with programmable switching frequency from 270 kHz to 800 kHz. The MIC2128 features an external soft-start pin (SS) which allows the user to adjust output soft-start time to reduce inrush current from mains during start-up. The MIC2128 features an auxiliary bootstrap LDO which improves the system efficiency by supplying the MIC2128 internal circuit bias power and gate drivers from output of the converter. A logic level enable (EN) signal can be used to enable or disable the controller.The MIC2128 can start-up monotonically into a prebiased output. The MIC2128 features an open drain power good signal (PG) which signals when the output is in regulation and can be used for simple power supply sequencing purpose. The MIC2128 offers a full suite of protection features to ensure protection of the IC during Fault conditions. These include undervoltage lockout to ensure proper operation under power-sag conditions, “hiccup” mode short-circuit protection and thermal shutdown. The MIC2128 is available in a 16-pin 3 mm x 3 mm QFN package, with an operating junction temperature range from -40°C to +125°C. Applications • • • • Networking/Telecom Equipment Base Station, Servers Distributed Power Systems Industrial Power Supplies FB AGND VDD MIC2128 3 x 3 QFN* (Top View) VIN Package Types 16 15 14 13 PG 1 12 SS ILIM 2 11 FREQ EP SW 3 10 EN BST 4 5 6 7 8 DH PGND DL PVDD 9 EXTVDD * Includes Exposed Thermal Pad (EP); see Table 3-1. 2016 Microchip Technology Inc. DS20005620A-page 1 MIC2128 Typical Application VIN 4.5V to 75V VIN PVDD 4.7 µF 0.1 µF 2.2 µFX3 Q1 DH 2.2ё BST VDD L1 10 µH 0.1 µF 4.7 µF VOUT 5V@5A SW MIC2128 ILIM + C1 330 µF 1.5 kё Q2 DL 7.5 kё EN VIN 47 µF 0.1 µF PG 4.7 nF 18 kё SS 10 nF FB 1 kё 100 kё EXTVDD FREQ VIN AGND 60 kё VOUT 1 µF PGND Q1,Q2: SiR878ADP L1: SRP1265A-100M, Bourns C1: 10SVP330M Functional Block Diagram EXTVDD VDD PVDD EN VIN 9 15 8 10 16 LINEAR REGULATOR LINEAR REGULATOR UVLO 4 BST 5 DH 3 SW 7 DL 2 ILIM 6 PGND THERMAL SHUTDOWN FREQ 11 TON ESTIMATION Control Logic Negative Current Limit COMPENSATION FB 13 PVDD gm 1.3 µA SS 12 VREF 0.6V CURRENT LIMIT DETECTION PG 96 µA 1 0.9 VREF FB 14 AGND DS20005620A-page 2 2016 Microchip Technology Inc. MIC2128 1.0 ELECTRICAL CHARACTERISTICS Absolute Maximum Ratings † VIN, FREQ, ILIM, SW to PGND ..................................................................................................................... -0.3V to +76V VDD, PVDD, FB, PG, SS to AGND .................................................................................................................. -0.3V to +6V EXTVDD to AGND ....................................................................................................................................... -0.3V to +16V BST to SW ................................................................................................................................................... -0.3V to +6V BST to AGND .............................................................................................................................................. -0.3V to +82V EN to AGND ...................................................................................................................................... -0.3V to (VIN +0.3V) DH, DL to AGND .............................................................................................................................. -0.3V to (VDD +0.3V) PGND to AGND ........................................................................................................................................... -0.3V to +0.3V Junction Temperature .......................................................................................................................................... +150°C Storage Temperature (TS) ...................................................................................................................... -65°C to +150°C Lead Temperature (soldering, 10s) ........................................................................................................................ 260°C ESD Rating(1) ......................................................................................................................................................... 1000V † Notice: Stresses above those listed under “Maximum Ratings” may cause permanent damage to the device. This is a stress rating only and functional operation of the device at those or any other conditions above those indicated in the operational sections of this specification is not intended. Exposure to maximum rating conditions for extended periods may affect device reliability. Note 1: Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5 k in series with 100 pF. Operating Ratings(1) Supply Voltage (VIN) ..................................................................................................................................... 4.5V to 75V SW, FREQ, ILIM, EN........................................................................................................................................... 0V to VIN Junction Temperature (TJ) ..................................................................................................................... -40°C to +125°C Package Thermal Resistance (3 mm × 3 mm QFN-16) Junction to Ambient (JA) .................................................................................................................................. 50.8°C/W Junction to Case (JC)....................................................................................................................................... 25.3°C/W Note 1: The device is not ensured to function outside the operating range. 2016 Microchip Technology Inc. DS20005620A-page 3 MIC2128 ELECTRICAL CHARACTERISTICS (Note 1) Electrical Specifications: unless otherwise specified, VIN = 12V, VOUT = 1.2V; VBST - VSW = 5V, TA = +25°C. Boldface values indicate -40°C TJ +125°C (Note 2). Parameter Symbol Min. Typ. Max. Units VVIN 4.5 — 5.5 V Test Conditions Power Supply Input Input Voltage Range PVDD and VDD shorted to VIN (VPVDD = VVIN = VVDD) 5.5 — 75 Quiescent Supply Current IQ — 1.4 1.8 mA VFB = 1.5V, no switching Shutdown Supply Current IVIN(SHDN) — 0.1 5 µA EN = Low — 30 60 µA EN = Low, VIN = VDD = 5.5V 5.1 5.4 V VVIN = 7V to 75V, IPVDD = 10 mA PVDD,VDD and EXTVDD PVDD Output Voltage VPVDD 4.8 VDD UVLO Threshold VVDD_UVLO_Rise 3.7 4.2 4.5 V VDD rising VDD UVLO Hysteresis VVDD_UVLO_Hys — 600 — mV VDD falling EXTVDD Bypass Threshold VEXTVDD_Rise 4.4 4.6 4.85 V EXTVDD Bypass Hysteresis VEXTVDD_Hys — 200 — mV — 250 — mV 0.597 0.6 0.603 V 0.594 0.6 0.606 V -40°C TJ 125°C IFB — 50 500 nA VFB = 0.6V EN Logic Level High VEN_H 1.6 — — V EN Logic Level Low VEN_L — — 0.6 V VEN_Hys — 150 — mV IEN — 6 30 µA VEN = 12V f0 — 800 — kHz VFREQ = VVIN, VVIN = 12V 230 270 300 EXTVDD Dropout Voltage EXTVDD rising VEXTVDD = 5V, IPVDD = 25mA Reference Feedback Reference Voltage FB Bias Current VREF TJ = 25°C Enable Control EN Hysteresis EN Bias Current ON Timer Switching Frequency VFREQ = 33% of VVIN, VVIN = 12V Maximum Duty Cycle DMAX — 85 — % VFREQ = VVIN = 12V Minimum Duty Cycle DMIN — 0 — % VFB > 0.6V Minimum ON Time tON(MIN) — 80 — ns Minimum OFF Time tOFF(MIN) 150 230 350 ns ISS — 1.3 — µA VOFFSET -15 0 15 mV VFB = 0.59V ICL 80 96 110 µA VFB = 0.59V — 0.3 — µA/°C — 48 — mV Soft-Start Soft-Start Current Source Current Limit Current-Limit Comparator Offset ILIM Source Current ILIM Source Current Tempco Negative Current Limit Comparator Threshold Note 1: 2: — Specification for packaged product only. The application is fully functional at low VDD (supply of the control section) if the external MOSFETs have low voltage VTH. DS20005620A-page 4 2016 Microchip Technology Inc. MIC2128 ELECTRICAL CHARACTERISTICS (Note 1) Electrical Specifications: unless otherwise specified, VIN = 12V, VOUT = 1.2V; VBST - VSW = 5V, TA = +25°C. Boldface values indicate -40°C TJ +125°C (Note 2). Parameter Symbol Min. Typ. Max. Units DH On-Resistance, High State RDH(PULL-UP) — 2 3 DH On-Resistance, Low State RDH(PULL_DOWN) — 2 4 DL On-Resistance, High State RDL(PULL-UP) — 2 4 RDL(PULL_DOWN) — 0.36 0.6 Test Conditions FET Drivers DL On-Resistance, Low State SW, VIN and BST Leakage BST Leakage — — — 30 µA VIN Leakage — — — 50 µA SW Leakage — — — 50 µA PG Threshold Voltage VPG_Rise 85 — 95 %VOUT VFB rising PG Hysteresis VPG_Hys — 6 — %VOUT VFB falling PG Delay Time PG_R_DLY — 100 — µs VFB rising PG Low Voltage VOL_PG — 70 200 mV VFB < 90% × VNOM, IPG = 1 mA Overtemperature Shutdown TSHDN — 150 — °C Junction temperature rising Overtemperature Shutdown Hysteresis TSHDN_Hys — 15 — °C Power Good (PG) Thermal Protection Note 1: 2: Specification for packaged product only. The application is fully functional at low VDD (supply of the control section) if the external MOSFETs have low voltage VTH. 2016 Microchip Technology Inc. DS20005620A-page 5 MIC2128 2.0 TYPICAL CHARACTERISTIC CURVES The graphs and tables provided following this note are a statistical summary based on a limited number of samples and are provided for informational purposes only. The performance characteristics listed herein are not tested or guaranteed. In some graphs or tables, the data presented may be outside the specified operating range (e.g., outside specified power supply range) and therefore outside the warranted range. Note: Note: Unless otherwise indicated, VVIN = 12V, fSW = 300 kHz, RILIM = 1.5 k, L = 10 µH, VEXTVDD = VOUT, TA = +25°C (refer to the Typical Application circuit). 20 Input Current (ȝA) Input Supply Current (mA) 25 15 VOUT = 5V IOUT = 0A FSW = 300 kHz VEN = VVIN 10 5 0 6 350 340 330 320 310 300 290 280 270 260 250 VVIN = 48V, with resistor divider between VIN and AGND at FREQ pin (100 k and 60 k) EN = GND -50 12 18 24 30 36 42 48 54 60 66 72 78 -25 Input Voltage (V) FIGURE 2-1: Input Voltage. FIGURE 2-4: Temperature. Input Supply Current vs. 25 50 75 100 Input Shutdown Current vs. 5.4 25 5.3 EXTVDD = GND PVDD Voltage (V) Input Supply Current (mA) 30 20 15 VEXTVDD = VOUT VVIN = 48V IOUT = 0A FSW = 300 kHz VEN = VIN 10 5 0 5.2 5.1 5 IPVDD = 10 mA VEN = VVIN EXTVDD = GND 4.9 4.8 -50 -25 0 25 50 75 100 6 12 18 24 30 36 42 48 54 60 66 72 78 Temperature (°C) FIGURE 2-2: Temperature. Input Voltage (V) FIGURE 2-5: Input Supply Current vs. 600 5.4 500 5.3 PVDD Voltage (V) Input Current (ȝA) 0 Temperature (°C) 400 300 200 VVIN = 48V, with resistor divider between VIN and AGND at FREQ pin (100 k and 60 k) EN = GND 100 0 VVIN = 48V IPVDD = 10 mA VEN = VVIN 5.2 VEXTVDD EXTVDD = 12V 5.1 5 EXTVDD = GND 4.9 VEXTVDD = 5V 4.8 6 18 30 42 54 66 78 -50 -25 DS20005620A-page 6 Input Shutdown Current vs. 0 25 50 75 100 Temperature (°C) Input Voltage (V) FIGURE 2-3: Input Voltage. PVDD Line Regulation FIGURE 2-6: Temperature. PVDD Voltage vs. 2016 Microchip Technology Inc. MIC2128 Note: Unless otherwise indicated, VVIN = 12V, fSW = 300 kHz, RILIM = 1.5 k, L = 10 µH, VEXTVDD = VOUT, TA = +25°C (refer to the Typical Application circuit). 5.2 1.6 PVDD Voltage (V) EXTVDD = GND Enable Voltage (V) 5 VEXTVDD = 12V 4.8 VEXTVDD = 5V 4.6 4.4 4.2 VVIN = 48V VEN = VVIN 4 0 10 20 30 40 50 1.4 VEN rising 1.2 1.0 VEN falling 0.8 0.6 60 -50 -25 IPVDD (mA) FIGURE 2-7: PVDD Load Regulation. FIGURE 2-10: Temperature. 100 125 Enable Threshold vs. 5.4 VVDD rising 4.3 4.1 EN Current (ȝA) VDD Voltage (V) 75 5.6 4.5 3.9 VVDD falling 3.7 3.5 3.3 IVDD = 0 mA EXTVDD = GND 5.2 5.0 4.8 4.6 4.4 VVIN = 12V VEN = 5V 4.2 4.0 3.1 -50 -25 0 25 50 75 100 -50 125 -25 FIGURE 2-8: Temperature. VDD UVLO Threshold vs. Switching frequency (kHz) VEXTVDD rising 4.6 4.5 VEXTVDD falling 4.4 4.3 4.2 -50 -25 0 25 50 75 100 125 320 310 300 290 280 270 260 250 240 230 220 EXTVDD Threshold vs. 2016 Microchip Technology Inc. 50 75 100 125 Enable Bias Current vs. IOUT = 5A IOUT = 0A VOUT = 5V FSW_SETPOINT = 300 kHz VEXTVDD = VOUT VEN = VVIN 6 12 18 24 30 36 42 48 54 60 66 72 78 Input Voltage (V) Temperature (°C) FIGURE 2-9: Temperature. 25 FIGURE 2-11: Temperature 4.8 4.7 0 Temperature (°C) Temperature (°C) EXTVDD Voltage (V) 0 25 50 Temperature (°C) FIGURE 2-12: Input Voltage. Switching Frequency vs. DS20005620A-page 7 MIC2128 Note: Unless otherwise indicated, VVIN = 12V, fSW = 300 kHz, RILIM = 1.5 k, L = 10 µH, VEXTVDD = VOUT, TA = +25°C (refer to the Typical Application circuit). 310 1.38 TA = -40°C Switching Frequency (kHz) 305 1.36 295 SS Source Current (ȝA) 300 TA = 25°C 290 285 280 VVIN = 48V VOUT = 5V FSW_SETPOINT = 300 kHz VEXTVDD = VOUT VEN = VVIN TA = 85°C 275 270 265 0 1 2 3 4 1.34 1.32 1.30 1.28 1.26 1.24 1.22 1.20 5 -50 -25 0 Load Current (A) Switching Frequency vs. FIGURE 2-16: Temperature. 140 606 130 604 Feedback Voltage (mV) ILIM Source Current (ȝA) FIGURE 2-13: Load Current. 120 110 100 90 80 70 50 75 100 125 SS Source Current vs. 602 600 598 596 594 -50 -25 0 25 50 75 100 125 -50 -25 0 Temperature (°C) FIGURE 2-14: Temperature. ILIM Source Current vs. FIGURE 2-17: Temperature. PG Threshold/VREF Ratio 1.2 1.0 0.8 0.6 0.4 0.2 0.0 -50 -25 0 25 50 50 75 100 125 75 100 125 FIGURE 2-15: Current Limit Comparator Offset vs. Temperature. Feedback Voltage vs. 95 94 93 92 91 90 89 88 87 86 85 -50 -25 Temperature (°C) DS20005620A-page 8 25 Temperature (°C) 1.4 Current Limit Comparator Offset Voltgae (mV) 25 Temperature (°C) 0 25 50 75 100 125 Temperature (°C) FIGURE 2-18: vs. Temperature. PG Threshold/VREF Ratio 2016 Microchip Technology Inc. MIC2128 100 90 80 70 60 50 40 30 20 10 0 VOUT=5V VOUT = 5V Efficiency (%) Efficiency (%) Note: Unless otherwise indicated, VVIN = 12V, fSW = 300 kHz, RILIM = 1.5 k, L = 10 µH, VEXTVDD = VOUT, TA = +25°C (refer to the Typical Application circuit). VOUT = 3.3V VOUT=3.3V VOUT = 2.5V VOUT=2.5V VOUT=1.8V VOUT = 1.8V VOUT = 1.5V VOUT=1.5V VOUT=1.2V VOUT = 1.2V VOUT = 1.0V VOUT=1.0V 0 0.5 1 1.5 2 2.5 3 3.5 4 4.5 100 90 80 70 60 50 40 30 20 10 0 5 VOUT=5V VOUT = 5V VOUT=3.3V VOUT = 3.3V VOUT=2.5V VOUT = 2.5V VOUT=1.8V VOUT = 1.8V VOUT = 1.5V VOUT=1.5V VOUT = 1.2V VOUT=1.2V VOUT=1V VOUT = 1.0V 0 0.5 1 Output Current (A) VOUT = 5V VOUT=5V VOUT = 3.3V VOUT=3.3V VOUT = 2.5V VOUT=2.5V VOUT=1.8V VOUT = 1.8V VOUT=1.5V VOUT = 1.5V VOUT=1.2V VOUT = 1.2V VOUT=1.0V VOUT = 1.0V 1 1.5 2 2.5 3 3.5 4 4.5 5 VOUT=3.3V VOUT = 3.3V VOUT=2.5V VOUT = 2.5V VOUT=1.8V V = 1.8V OUT VOUT=1.5V VOUT = 1.5V VOUT=1.2V VOUT = 1.2V VOUT=1.0V VOUT = 1.0V 2 2.5 3 3.5 4 Output Current (A) FIGURE 2-21: Efficiency vs. Output Current (Input Voltage = 36V). 2016 Microchip Technology Inc. 4.5 5 0.5 1 1.5 2 2.5 3 3.5 4 4.5 5 FIGURE 2-23: Efficiency vs. Output Current (Input Voltage = 60V). VOUT=5V VOUT = 5V 1.5 4 VOUT=5V V OUT = 5V VOUT=3.3V V OUT = 3.3V VOUT=2.5V VOUT = 2.5V VOUT=1.8V V OUT = 1.8V VOUT=1.5V V OUT = 1.5V VOUT=1.2V VOUT = 1.2V VOUT=1.0V V OUT = 1.0V 0 Efficiency (%) Efficiency (%) 100 90 80 70 60 50 40 30 20 10 0 1 3.5 Output Current (A) FIGURE 2-20: Efficiency vs. Output Current (Input Voltage = 24V). 0.5 3 100 90 80 70 60 50 40 30 20 10 0 Output Current (A) 0 2.5 FIGURE 2-22: Efficiency vs. Output Current (Input Voltage = 48V). Efficiency (%) Efficiency (%) 100 90 80 70 60 50 40 30 20 10 0 0.5 2 Output Current (A) FIGURE 2-19: Efficiency vs. Output Current (Input Voltage=12V). 0 1.5 4.5 5 100 90 80 70 60 50 40 30 20 10 0 VOUT=5V VOUT = 5V VOUT=3.3V VOUT = 3.3V VOUT = 2.5V VOUT=2.5V VOUT=1.8V VOUT = 1.8V VOUT=1.5V VOUT = 1.5V VOUT=1.2V VOUT = 1.2V VOUT=1.0V VOUT = 1.0V 0 0.5 1 1.5 2 2.5 3 3.5 4 4.5 5 Output Current (A) FIGURE 2-24: Efficiency vs. Output Current (Input Voltage = 75V). DS20005620A-page 9 MIC2128 Note: Unless otherwise indicated, VVIN = 12V, fSW = 300 kHz, RILIM = 1.5 k, L = 10 µH, VEXTVDD = VOUT, TA = +25°C (refer to the Typical Application circuit). 5.1091 Output Voltage (V) 5.0991 VVIN 20V/div 5.0891 VIN = 12V VIN=12V 5.0791 VIN = 24V VIN=24V 5.0691 VSW 50V/div VIN=36V VIN = 36V 5.0591 VIN=48V VIN = 48V 5.0491 VIN=60V VIN = 60V VOUT 2V/div VIN = 75V VIN=75V 5.0391 0 0.5 1 1.5 2 2.5 3 3.5 4 4.5 5 Output Current (A) FIGURE 2-25: (VOUT = 5V). 10 ms/div FIGURE 2-28: VVIN Turn-On with Pre-biased Output. Load Regulation VEN 2V/div VVIN 20V/div VSW 20V/div VOUT 2V/div VPG 5V/div VOUT 2V/div IL 5A/div FIGURE 2-26: IL 5A/div 10 ms/div VVIN Turn-On. FIGURE 2-29: 4 ms/div EN Turn-On/Turn-Off. VEN 2V/div VVIN 20V/div VSW 20V/div VOUT 2V/div 10 ms/div FIGURE 2-27: DS20005620A-page 10 VVIN Turn-Off. IL 5A/div VOUT 2V/div VPG 5V/div IL 5A/div FIGURE 2-30: 2_ms/div EN Turn-On Delay. 2016 Microchip Technology Inc. MIC2128 Note: Unless otherwise indicated, VVIN = 48V, fSW = 300 kHz, RILIM = 1.5 k, L = 10 µH, VEXTVDD = VOUT, TA = +25°C (refer to the Typical Application circuit). VEN 2V/div VVDD 1V/div VOUT 2V/div VPG 5V/div 200 µs/div FIGURE 2-31: IL 5A/div EN Turn-Off Delay. VOUT 2V/div VSW 5V/div FIGURE 2-34: Rising. 20 ms/div VDD UVLO Threshold- VVDD 2V/div VEN 2V/div VOUT 2V/div VOUT 2V/div VSW 5V/div VSW 50V/div 4 ms/div FIGURE 2-32: Output. EN Turn-On with Prebiased VEN 1V/div 100 ms/div FIGURE 2-35: Falling. VDD UVLO Threshold- VEN 2V/div VOUT 500 mV/div VOUT 2V/div IL 5A/div VSW 50V/div FIGURE 2-33: 10 ms/div Enable Thresholds. 2016 Microchip Technology Inc. 4 ms/div FIGURE 2-36: Enable into Output Short. DS20005620A-page 11 MIC2128 Note: Unless otherwise indicated, VVIN = 48V, fSW = 300 kHz, RILIM = 1.5 k, L = 10 µH, VEXTVDD = VOUT, TA = +25°C (refer to the Typical Application circuit). VOUT 2V/div VVIN 20V/div VOUT 500 mV/div IL 5A/div IL 5A/div 4 ms/div 10 ms/div FIGURE 2-37: Power-Up into Output Short. FIGURE 2-40: Circuit. Recovery from Output Short VOUT 200 mV/div AC coupled VOUT 2V/div IOUT 5A/div IOUT 5A/div 4ms/div FIGURE 2-38: Threshold. Output Current Limit Load Transient Response IOUT 2A/div IL 5A/div 2 ms/div DS20005620A-page 12 FIGURE 2-41: (IOUT = 0A to 5A). VOUT 200 mV/div AC coupled VOUT 2V/div FIGURE 2-39: 200 µs/div Output Short Circuit. 200 µs/div FIGURE 2-42: Load Transient Response (IOUT = 0A to 2.5A). 2016 Microchip Technology Inc. MIC2128 Note: Unless otherwise indicated, VVIN = 48V, fSW = 300 kHz, RILIM = 1.5 k, L = 10 µH, VEXTVDD = VOUT, TA = +25°C (refer to the Typical Application circuit). VOUT 50 mV/div AC coupled VOUT 200 mV/div AC coupled VSW 50V/div IOUT 2A/div IL 5A/div 2 µs/div 200 µs/div FIGURE 2-43: Load Transient Response (IOUT = 2.5A to 5A). FIGURE 2-46: Load. Switching Waveform at No VOUT 50 mV/div AC coupled IL 5A/div VSW 50V/div VSW 50V/div VDH 50V/div IL 5A/div VDL 5 mV/div 2 µs/div 2 µs/div FIGURE 2-44: Load. Switching Waveform at No FIGURE 2-47: Load. Switching Waveform at Full IL 5A/div VSW 50V/div VDH 50V/div VDL 5V/div 2 µs/div FIGURE 2-45: Load. Switching Waveform at Full 2016 Microchip Technology Inc. DS20005620A-page 13 MIC2128 3.0 PIN DESCRIPTION The descriptions of the pins are listed in Table 3-1. TABLE 3-1: 3.1 PIN FUNCTION TABLE MIC2128 Symbol 1 PG Pin Function Open-drain Power Good Output pin 2 ILIM Current Limit setting resistor connection pin 3 SW Switch Pin and Current Sense Input for negative current limit 4 BST Bootstrap Capacitor Connection Pin 5 DH High-side N-MOSFET gate driver Output 6 PGND 7 DL 8 PVDD 9 EXTVDD Power Ground Low-side N-MOSFET gate driver output. Internal high voltage LDO Output of the MIC2128 Supply to the internal low voltage LDO 10 EN 11 FREQ Enable Input 12 SS Soft-Start time setting capacitor connection pin 13 FB Feedback Input Analog ground Switching Frequency Programming Input 14 AGND 15 VDD Supply for the MIC2128 internal analog circuits 16 VIN Supply Input for the internal high voltage LDO 17 EP Exposed Pad Power Good Output Pin (PG) Connect PG to VDD through a pull up resistor. PG is low when the FB voltage is 10% below the 0.6V reference voltage. 3.2 Current Limit Pin (ILIM) Connect a resistor from ILIM to SW to set current limit. Refer to Section 4.3 “Current Limit (ILIM)” for more details. 3.3 Switch Pin (SW) SW pin provides return path for the high-side N-MOSFET gate driver when DH is low and is also used to sense low-side MOSFET current by monitoring the SW node voltage for negative current limit function. Connect SW to the pin where high-side MOSFET source and the low-side MOSFET drain terminal are connected together. 3.4 Bootstrap Capacitor Pin (BST) BST capacitor acts as supply for the high-side N-MOSFET driver. Connect a minimum of 0.1 µF low ESR ceramic capacitor between BST and SW. Refer to Section 4.5 “High-Side MOSFET Gate Drive (DH)” section for more details. 3.5 High-Side N-MOSFET Gate Driver Output Pin (DH) High-side N-MOSFET gate driver Output. Connect DH to the gate of external high-side N-MOSFET. 3.6 Power Ground Pin (PGND) PGND provides return path for the internal low-side N-MOSFET gate driver output and also acts as reference for current limit comparator. Connect PGND to the external low-side N-MOSFET source terminal and to the return terminal of PVDD bypass capacitor. 3.7 Low-Side N-MOSFET Gate Driver Output Pin (DL) Low-side N-MOSFET gate driver output. Connect to the gate terminal of the external low-side N-MOSFET. DS20005620A-page 14 2016 Microchip Technology Inc. MIC2128 3.8 Internal High Voltage LDO Output Pin (PVDD) Internal high voltage LDO Output of the MIC2128. PVDD is the supply for the low-side MOSFET driver and for floating high-side MOSFET driver. Connect a minimum of 4.7 µF low ESR ceramic capacitor from PVDD to PGND. 3.9 EXTVDD Supply to the internal low voltage LDO. Connect EXTVDD to the output of the Buck converter if it is between 4.7V to 14V to improve system efficiency. Bypass EXTVDD with a minimum of 1 µF low ESR ceramic capacitor. 3.10 Enable Input Pin (EN) EN is a logic input. Connect to logic high to enable the converter and connect to logic low to disable the converter. 3.11 Switching Frequency Programming Input Pin (FREQ) Switching Frequency Programming Input. Connect to mid-point of the resistor divider formed between VIN and AGND to set switching frequency of the converter. Tie FREQ to VIN to set the switching frequency to 800 kHz. Refer to Section 5.1 “Setting the Switching Frequency” for more details. 3.12 3.13 Feedback Input Pin (FB) FB is input to the transconductance amplifier of the control loop. The control loop regulates the FB voltage to 0.6V. Connect FB node to mid-point of the resistor divider between output and AGND. 3.14 Analog Ground Pin (AGND) AGND is reference to the analog control circuits inside the MIC2128. Connect AGND to PGND at one point on PCB. 3.15 Bias Voltage Pin (VDD) Supply for the MIC2128 internal analog circuits. Connect VDD to PVDD of the MIC2128 through a low pass filter. Connect a minimum of 4.7 µF low ESR ceramic capacitor from VDD to AGND for decoupling. 3.16 Input Voltage Pin (VIN) Supply Input to the internal high voltage LDO. Connect to the main power source and bypass to PGND with a minimum of 0.1 µF low ESR ceramic capacitor. 3.17 Exposed Pad (EP) Connect to AGND copper plane to improve thermal performance of the MIC2128. Soft-Start Time Setting Capacitor Connection Pin (SS) Soft-Start time setting capacitor connection pin. Connect a ceramic capacitor from SS to AGND to set the output Soft-Start time. Refer to Section 5.3 “Setting the Soft-Start Time” section for further details. 2016 Microchip Technology Inc. DS20005620A-page 15 MIC2128 4.0 FUNCTIONAL DESCRIPTION The MIC2128 is an adaptive on-time synchronous buck controller designed to cover a wide range of input voltage applications ranging from 4.5V to 75V. An adaptive on-time control scheme is employed to get fast transient response and to obtain high voltage conversion ratios at constant switching frequency. Overcurrent protection is implemented by sensing low-side MOSFET's RDS(ON) which eliminates lossy current sense resistor. The device features external soft-start, enable input, UVLO, power good output (PG), secondary bootstrap LDO and thermal shutdown. 4.1 Theory of Operation The MIC2128 is an adaptive on-time synchronous buck controller which operates based on ripple at feedback node. The output voltage is sensed by the MIC2128 feedback pin (FB) and is compared to a 0.6V reference voltage (VREF) at the low-gain transconductance error amplifier (gm) as shown in the Functional Block Diagram. Figure 4-1 shows the MIC2128 control loop timing during steady-state operation. The error amplifier behaves as short circuit for the ripple voltage frequency on the FB pin which causes the error amplifier output voltage ripple to follow the feedback voltage ripple. When the transconductance error amplifier output (VgM) is below the reference voltage of the comparator, which is same as the error amplifier reference (VREF), the comparator triggers and generates an on-time event. The on-time period is predetermined by the fixed tON estimator circuitry which is given by the following Equation 4-1: EQUATION 4-1: The maximum duty cycle can be calculated using the following Equation 4-2: EQUATION 4-2: t SW – t OFF MIN 230ns D MAX = --------------------------------------- = 1 – --------------t SW tSW Where: tSW = Switching period, equal to 1/fSW It is not recommended to use the MIC2128 with an OFF time close to tOFF(MIN) during steady-state operation. The adaptive on-time control scheme results in a constant switching frequency over wide range of input voltage and load current. The actual ON time and resulting switching frequency varies with the different rising and falling times of the external MOSFETs. The minimum controllable ON time (tON(MIN)) results in a lower switching frequency than the target switching frequency in high VIN to VOUT ratio applications. The equation below shows the output-to-input voltage ratio, below which the MIC2128 lowers the switching frequency in order to regulate the output-to-set value. EQUATION 4-3: V OUT ------------- = t ON MIN f SW V IN Where: VOUT = Output voltage VIN = Input voltage fSW = Switching frequency tON(MIN) = Minimum controllable ON time (80 ns typ.) V OUT t ON ESTIMATED = -------------------------V VIN f SW Where: ȴIL IL VOUT = Output voltage VVIN = Power stage input voltage fSW = Switching frequency At the end of the ON time, the internal high-side driver turns off the high-side MOSFET and the low-side driver turns on the low-side MOSFET. The OFF time of the high-side MOSFET depends on the feedback voltage. When the feedback voltage decreases, the output of the gm amplifier (VgM) also decreases. When the output of the gm amplifier (VgM) is below the reference voltage of the comparator (which is same as the error amplifier reference (VREF)) the OFF time ends and ON time is triggered. If the OFF time determined by the feedback voltage is less than the minimum OFF time (tOFF(MIN))of the MIC2128, which is about 230 ns (typical), the MIC2128 control logic applies the tOFF(MIN) instead. DS20005620A-page 16 ȴVOUT = ESR*ȴIL VOUT ȴVFB =ȴVOUT *(VREF/VOUT) VREF VFB ȴVFB VREF VgM MIC2127 Triggers ON-Time event if the error amplifier output (VgM) is below VREF VDH Estimated ON-Time FIGURE 4-1: Timing. MIC2128 Control Loop 2016 Microchip Technology Inc. MIC2128 Figure 4-2 shows operation of the MIC2128 during load transient. The output voltage drops due to a sudden increase in load, which results in the error amplifier output (VgM) falling below VREF. This causes the comparator to trigger an on-time event. At the end of the ON time, a minimum OFF time tOFF(MIN) is generated to charge the bootstrap capacitor. The next ON time is triggered immediately after the tOFF(MIN) if the error amplifier output voltage (VgM) is still below VREF due to the low feedback voltage. This operation results in higher switching frequency during load transients. The switching frequency returns to the nominal set frequency once the output stabilizes at new load current level. The output recovery time is fast and the output voltage deviation is small in the MIC2128 converter due to the varying duty cycle and switching frequency. Full Load IL current ripple if the ESR of the output capacitor is very low. For these applications, ripple injection is required to ensure proper operation. Refer to Section 5.8 “Ripple Injection” for details about the ripple injection technique. 4.2 Soft Start reduces the power supply inrush current at start-up by controlling the output voltage rise time. The MIC2128 features SS pin which allows the user to set the soft-start time by connecting a capacitor from the SS pin to AGND. An internal current source of 1.3 µA charges this capacitor and generates a linear voltage which is used as the reference for the internal error amplifier during Soft Start. Once the voltage on this SS capacitor is above the internal fixed reference of 0.6V, the error amplifier uses the fixed 0.6V as reference instead of the voltage on the external SS capacitor. 4.3 No Load Soft Start (SS) Current Limit (ILIM) The MIC2128 uses the low-side MOSFET RDS(ON) to sense inductor current. In each switching cycle of the MIC2128 converter, the inductor current is sensed by monitoring the voltage across the low-side MOSFET during the OFF period of the switching cycle during which low-side MOSFET is ON. An internal current source of 96 µA generates a voltage across the external current limit setting resistor RCL as show in the Figure 4-3. VOUT VREF VFB VIN VREF MIC2128 VgM DH L1 SW Control Logic DL PGND VDH CURRENT LIMIT DETECTION toff(MIN) FIGURE 4-2: Response. RCL ICL ILIM MIC2128 Load Transient Unlike true current-mode control, the MIC2128 uses the output voltage ripple to trigger an on-time event. In order to meet the stability requirements, the MIC2128 feedback voltage ripple should be in phase with the inductor current ripple and large enough to be sensed by the internal error amplifier. The recommended feedback voltage ripple is 20 mV~100 mV over the full input voltage range. If a low-ESR output capacitor is selected, then the feedback voltage ripple may be too small to be sensed by the internal error amplifier. Also, the output voltage ripple and the feedback voltage ripple are not necessarily in phase with the inductor 2016 Microchip Technology Inc. FIGURE 4-3: Circuit. MIC2128 Current Limiting The ILIM pin voltage (VILIM) is the difference of the voltage across the low-side MOSFET and the voltage across the resistor (VCL). The sensed voltage VILIM is compared with the power ground (PGND) after a blanking time of 150 ns. If the absolute value of the voltage drop across the low-side MOSFET is greater than the absolute value of the voltage across the current setting resistor (VCL), the MIC2128 triggers the current limit event. Consecutive DS20005620A-page 17 MIC2128 eight current limit events trigger the hiccup mode. Once the controller enters into hiccup mode, it initiates a soft-start sequence after a hiccup timeout of 4 ms (typical). Both the high-side and low-side MOSFETs are turned off during hiccup timeout. The hiccup sequence including the soft start reduces the stress on the switching FETs and protects the load and supply from severe short conditions. The current limit can be programmed by using the following Equation 4-4. EQUATION 4-4: RCL IL PP I + ---------------- R DS ON + V OFFSET CLIM 2 = -------------------------------------------------------------------------------------------------I CL Where: ILIM = Load current limit RDS (ON) = On-resistance of low-side power MOSFET ILPP = Inductor peak-to-peak ripple current VOFFSET = Current-limit comparator offset (15 mV max.) ICL = Current-limit source current (96 µA typ) Since MOSFET RDS(ON) varies from 30% to 40% with temperature, it is recommended to consider the RDS(ON) variation while calculating RCL in the above equation to avoid false current limiting due to increased MOSFET junction temperature rise. It is also recommended to connect SW pin directly to the drain of the low-side MOSFET to accurately sense the MOSFETs RDS(ON). To improve the current limit variation, the MIC2128 adjusts the internal current limit source current (ICL) at a rate of 0.3 µA/°C when the MIC2128 junction temperature changes to compensate the RDS(ON) variation of external low-side MOSFET. The effectiveness of this method depends on the thermal gradient between the MIC2128 and the external low-side MOSFET. The lower the thermal gradient, the better the current limit variation. EQUATION 4-5: 48mV I NLIM = -------------------R DS ON Where: INLIM = Negative current limit RDS (ON) = On-resistance of low-side power MOSFET 4.5 High-Side MOSFET Gate Drive (DH) The MIC2128's high-side drive circuit is designed to switch an N-Channel external MOSFET. The MIC2128 Functional Block diagram shows a bootstrap diode between PVDD and BST pins. This circuit supplies energy to the high-side drive circuit. A low ESR ceramic capacitor should be connected between BST and SW pins (refer Typical Application Circuit).The capacitor between BST and SW pins, CBST, is charged while the low-side MOSFET is on. When the high-side MOSFET driver is turned on, energy from CBST is used to turn the MOSFET on. A minimum of 0.1 µF low ESR ceramic capacitor is recommended between BST and SW pins. The required value of CBS can be calculated using the following Equation 4-6. EQUATION 4-6: Q G_HS C BST = --------------- VBST Where: QG_HS = High-side MOSFET total gate charge VBST = Drop across the CBST, generally 50 mV to 100 mV A small resistor in series with CBST, can be used to slow down the turn-on time of the high-side N-channel MOSFET. 4.6 Low-Side MOSFET Gate Drive (DL) A small capacitor (CCL) can be connected from the ILIM pin to PGND to filter the switch node ringing during the Off-time allowing a better current sensing. The time constant of RCL and CCL should be less than the minimum off time. The MIC2128's low-side drive circuit is designed to switch an N-Channel external MOSFET. The internal low-side MOSFET driver is powered by PVDD. Connect a minimum of 4.7 µF low-ESR ceramic capacitor to supply the transient gate current of the external MOSFET. 4.4 4.7 Negative Current Limit The MIC2128 implements negative current limit by sensing the SW voltage when the low-side FET is ON. If the SW node voltage exceeds 48 mV typical, the device turns off the low-side FET for 500 ns. Negative current limit value is shown in Equation 4-5. DS20005620A-page 18 Auxiliary Bootstrap LDO (EXTVDD) The MIC2128 features an auxiliary bootstrap LDO which improves the system efficiency by supplying the MIC2128 internal circuit bias power and gate drivers from the converter output voltage. This LDO is enabled when the voltage on the EXTVDD pin is above 4.6V (typical) and at the same time the main LDO which operates from VIN is disabled to reduce power consumption. 2016 Microchip Technology Inc. MIC2128 5.0 APPLICATIONS INFORMATION 5.2 5.1 Setting the Switching Frequency The output voltage can be adjusted using a resistor divider from output to AGND whose mid-point is connected to FB pin as shown the Figure 5-3. The MIC2128 is an adjustable-frequency, synchronous buck controller featuring a unique adaptive on-time control architecture. The switching frequency can be adjusted between 270 kHz and 800 kHz by changing the resistor divider network between VIN and AGND pins consisting of R1 and R2 as shown in Figure 5-1. Output Voltage Setting MIC2128 MIC2128 VOUT R1 VIN 16 COMPENSATION VIN 4.5V to 75V 13 gm FB R1 11 VREF 0.6V R2 14 AGND FIGURE 5-3: FIGURE 5-1: Adjustment. Equation 5-1 frequency: Switching Frequency shows the estimated switching Output Voltage Adjustment. The output voltage Equation 5-2: can be calculated using EQUATION 5-2: R1 V OUT = V REF 1 + ------ R 2 EQUATION 5-1: Where: f SW_ADJ VREF R2 = fO ------------------R1 + R2 fO is the switching frequency when R1 is 100 k and R2 being open; fO is typically 800 kHz. For more precise setting, it is recommended to use Figure 5-2: 800 Switching Frequency (kHz) R2 SOFTSTART Comparator FREQ VIN = 24V 700 600 VIN = 48V 500 VIN = 75V 400 = 0.6V The maximum output voltage that can be programmed using the MIC2128 is limited to 30V, if not limited by the maximum duty cycle (see Equation 4-2). A typical value of R1 is less than 30 k. If R1 is too large, it may allow noise to be introduced into the voltage feedback loop and also increases the offset between the set output voltage and actual output voltage because of the error amplifier bias current. If R1 is too small in value, it will decrease the efficiency of the power supply, especially at light loads. Once R1 is selected, R2 can be calculated using Equation 5-3. EQUATION 5-3: VOUT = 5V R1 = 100 k IOUT = 5A 300 200 50 500 5000 R2 (k) FIGURE 5-2: R1 R 2 = ----------------------V OUT ------------- – 1 V REF Switching Frequency vs. R2. 2016 Microchip Technology Inc. DS20005620A-page 19 MIC2128 5.3 5.4.1 Setting the Soft-Start Time The output Soft-Start time can be set by connecting a capacitor from SS to AGND from 2 ms to 100 ms as shown in Figure 5-4. MIC2128 HIGH-SIDE MOSFET POWER LOSSES The total power loss in the high-side MOSFET (PHSFET) is the sum of the power losses because of conduction (PCONDUCTION), switching (PSW), reverse recovery charge of low-side MOSFET body diode (PQrr) and MOSFET's output capacitance discharge as calculated in the Equation 5-5. EQUATION 5-5: PHSFET = PCONDUCTION HS + PSW HS + P Qrr + P COSS 2 P CONDUCTION HS = I RMS HS R DS ON_HS 13 COMPENSATION FB P SW HS = 0.5 VIN ILOAD tR + t F f SW ISS 1.3 µA gm 12 PQrr = V IN Q rr f SW SS CSS Comparator Where: VREF 0.6V FIGURE 5-4: 1 2 P COSS = --- C OSS HS + COSS HS V IN f SW 2 Setting the Soft-Start Time. The value of the capacitor can be calculated using Equation 5-4. EQUATION 5-4: ISS tSS C SS = -------------------V REF Where: CSS = Capacitor from SS pin to AGND ISS = Internal Soft-Start current (1.3 µA typical) tSS = Output Soft-Start time VREF = 0.6V RDS(ON_HS) = VIN = On-resistance of the high-side MOSFET Operating input voltage ILOAD = Load current fSW = Operating switching frequency Qrr = Reverse recovery charge of low-side MOSFET body diode or of external diode across low-side MOSFET COSS(HS) = Effective high-side MOSFET output capacitance COSS(LS) = Effective low-side capacitance IRMS(HS) = RMS current of the high-side MOSFET which can be calculated using Equation 5-6. tR, tF = The high-side MOSFET turn-on and turn-off transition times which can be approximated by Equation 5-8 and Equation 5-9 MOSFET output EQUATION 5-6: 5.4 MOSFET Selection Important parameters for MOSFET selection are: • Voltage rating • On-resistance • Total gate charge The voltage rating for the high-side and low-side MOSFETs are essentially equal to the power stage input voltage VIN. A safety factor of 30% should be added to the VIN(MAX) while selecting the voltage rating of the MOSFETs to account for voltage spikes due to circuit parasitic elements. DS20005620A-page 20 I RMS HS = I LOAD D ILOAD is the load current and D is the operating duty cycle, given by Equation 5-7. EQUATION 5-7: VOUT D = ------------V IN 2016 Microchip Technology Inc. MIC2128 EQUATION 5-12: EQUATION 5-8: Q SW HS R DH PULL_UP + RHS GATE t R = -----------------------------------------------------------------------------------------------------V DD – V TH ILOAD is the load current and D is the operating duty cycle. EQUATION 5-9: Q SW HS R DH PULL_DOWN + RHS GATE t F = ------------------------------------------------------------------------------------------------------------V TH Where: RDH(PULL-UP) I RMS LS = I LOAD 1 – D = High-side gate driver pull-up resistance RDH(PULL-DOWN) = High-side gate driver pull-down resistance RHS(GATE) = High-side MOSFET gate resistance VTH = Gate threshold voltage of the high-side MOSFET QSW(HS) = Switching gate charge of the high-side MOSFET which can be approximated by Equation 5-10. 5.5 Inductor Selection Inductance value, saturation and RMS currents are required to select the output inductor. The input and output voltages and the inductance value determine the peak-to-peak inductor ripple current. The lower the inductance value, the higher the peak-to-peak ripple current through the inductor, which increases the core losses in the inductor. Higher inductor ripple current also requires more output capacitance to smooth out the ripple current. The greater the inductance value, the lower the peak-to-peak ripple current, which results in a larger and more expensive inductor. A good compromise between size, loss and cost is to set the inductor ripple current to be equal to 30% of the maximum output current. EQUATION 5-10: QGS HS Q SW HS = -------------------- + QGD HS 2 Where: The inductance value is calculated by Equation 5-13: EQUATION 5-13: QGS(HS) = High-side MOSFET gate to source charge QGD(HS) = High-side MOSFET gate to drain charge VOUT VIN – VOUT L = -----------------------------------------------------V IN f SW 0.3 IFL Where: 5.4.2 LOW-SIDE MOSFET POWER LOSSES The total power loss in the low-side MOSFET (PLSFET) is the sum of the power losses because of conduction (PCONDUCTION(LS)) and body diode conduction during the dead time (PDT) as calculated in Equation 5-11. EQUATION 5-11: PLSFET = PCONDUCTION LS + P DT VIN = Input voltage fSW = Switching frequency IFL = Full load current VOUT = Output voltage For a selected Inductor, the peak-to-peak inductor ripple current ripple can be calculated using Equation 5-14. EQUATION 5-14: 2 P CONDUCTION LS = I RMS LS RDS ON_LS P DT = 2 V F I LOAD t DT f SW Where: RDS(ON_LS) = On-resistance of the low-side MOSFET VF = Low-side MOSFET body diode forward voltage drop tDT = Dead time which is approximately 20 ns fSW = Switching Frequency IRMS(LS) = RMS current of the low-side MOSFET which can be calculated using Equation 5-12 2016 Microchip Technology Inc. V V – V V IN f SW L OUT IN OUT I L_PP = ----------------------------------------------------- The peak inductor current is equal to the load current plus one half of the peak-to-peak inductor current ripple which is shown in Equation 5-15. EQUATION 5-15: I L_PP IL_PK = I LOAD + ---------------2 DS20005620A-page 21 MIC2128 The RMS and saturation current ratings of the selected inductor should be at least equal to the RMS current and saturation current calculated in the Equation 5-16 and Equation 5-17. EQUATION 5-19: VOUT_PP ESR ------------------------- I L_PP Where: EQUATION 5-16: 2 I L_RMS = 2 I L_PP I LOAD(MAX) + -----------------------12 Where: ILOAD(MAX) VOUT_PP = Peak-to-peak output voltage ripple IL_PP = Peak-to-peak inductor current ripple The required output capacitance to meet steady state output ripple can be calculated using Equation 5-20. = Maximum load current EQUATION 5-20: I L_PP C OUT = -------------------------------------------------8 f SW V OUT_PP EQUATION 5-17: Where: RCL I CL + 15mV IL_SAT = --------------------------------------------R DS(ON) Where: RCL = Current limit resistor COUT = Output capacitance value ICL = Current-Limit Source Current (96 µA typical) fSW = Switching frequency RDS (ON) = On-resistance of low-side power MOSFET Maximizing the efficiency requires the proper selection of core material and minimizing the winding resistance. Use of ferrite materials is recommended in the higher switching frequency applications. Lower cost iron powder cores may be used but the increase in core loss reduces the efficiency of the power supply. This is especially noticeable at low output power. The winding resistance decreases efficiency at the higher output current levels. The winding resistance must be minimized although this usually comes at the expense of a larger inductor. The power dissipated in the inductor is equal to the sum of the core and copper losses. At higher output loads, the core losses are usually insignificant and can be ignored. At lower output currents, the core losses can be a significant contributor. Core loss information is usually available from the magnetic’s vendor. The amount of copper loss in the inductor is calculated by Equation 5-18. As described in Section 4.1 “Theory of Operation”, the MIC2128 requires at least 20 mV peak-to-peak ripple at the FB pin to ensure that the gm amplifier and the comparator behave properly. Also, the output voltage ripple should be in phase with the inductor current. Therefore, the output voltage ripple caused by the output capacitor’s value should be much smaller than the ripple caused by the output capacitor ESR. If low-ESR capacitors, such as ceramic capacitors, are selected as the output capacitors, a ripple injection circuit should be used to provide the enough feedback-voltage ripple. Refer to the Section 5.8 “Ripple Injection” for details. The voltage rating of the capacitor should be twice the output voltage for a tantalum and 20% greater for aluminum electrolytic, ceramic or OS-CON. The output capacitor RMS current is calculated in Equation 5-21. EQUATION 5-21: I L_PP I C_OUT(RMS) = ---------------12 EQUATION 5-18: 2 P INDUCTOR CU = I L_RMS R DCR 5.6 Output Capacitor Selection The main parameters for selecting the output capacitor are capacitance value, voltage rating and RMS current rating. The type of the output capacitor is usually determined by its equivalent series resistance (ESR). Recommended capacitor types are ceramic, tantalum, low-ESR aluminum electrolytic, OS-CON and POSCAP. The output capacitor ESR also affects the control loop from a stability point of view. The maximum value of ESR can be calculated using Equation 5-19. DS20005620A-page 22 The power dissipated in the output capacitor is shown in Equation 5-22. EQUATION 5-22: 2 P DIS(C_OUT) = IC_OUT(RMS) ESRC_OUT 5.7 Input Capacitor Selection The input capacitor reduces peak current drawn from the power supply and reduces noise and voltage ripple on the input. The input voltage ripple depends on the input capacitance and ESR. The input capacitance and ESR values can be calculated using Equation 5-23. 2016 Microchip Technology Inc. MIC2128 EQUATION 5-23: EQUATION 5-26: R R2 + R1 I LOAD D 1 – D C IN = ------------------------------------------------ fSW V IN_C 2 VFB PP = ----------------- ESR I L_PP V IN_ESR ESRC_IN = ----------------------I L_PK IL_PP is the peak-to-peak value of the inductor current ripple. Where: ILOAD = Load Current IL_PK = Peak Inductor Current VINC = Input ripple due to capacitance VINESR = Input ripple due to input capacitor ESR η = Power conversion efficiency SW L R1 MIC2128 COUT FB ESR R2 The input capacitor should be rated for ripple current rating and voltage rating. The RMS value of input capacitor current is determined at the maximum output current. The RMS current rating of the input capacitor should be greater than or equal to the input capacitor RMS current calculated using the Equation 5-24. EQUATION 5-24: I C_IN(RMS) = I LOAD(MAX) D 1 – D The power dissipated in the input capacitor is calculated using Equation 5-25. FIGURE 5-5: 2. The output voltage ripple can be fed into the FB pin through a feed forward capacitor, CFF in this case, as shown in Figure 5-6. The typical CFF value is between 1 nF and 100 nF. With the feed forward capacitor, the feedback voltage ripple is very close to the output voltage ripple which is shown in Equation 5-27. V FB PP = ESR I L_PP 2 PDISS(C_IN) = I C_IN(RMS) ESR C_IN Ripple Injection The minimum recommended ripple at the FB pin for proper operation of the MIC2128 error amplifier and comparator is 20 mV. However, the output voltage ripple is generally designed as 1% to 2% of the output voltage. For low output voltages, such as a 1V, the output voltage ripple is only 10 mV to 20 mV, and the feedback voltage ripple is less than 20 mV. If the feedback voltage ripple is so small that the gm amplifier and comparator cannot sense it, then the MIC2128 loses control and the output voltage is not regulated. In order to have sufficient VFB ripple, ripple injection method should be applied for low output voltage ripple applications. The applications are divided into three situations according to the amount of the feedback voltage ripple: 1. Inadequate ripple at the feedback voltage due to the small ESR of the output capacitor. EQUATION 5-27: EQUATION 5-25: 5.8 Enough Ripple at FB. SW L R1 MIC2128 FB CFF COUT ESR R2 FIGURE 5-6: 3. Inadequate Ripple at FB. Virtually no ripple at the FB pin voltage due to the very-low ESR of the output capacitors: In this case, additional ripple can be injected into the FB pin from the switching node SW via a resistor RINJ and a capacitor CINJ, as shown in Figure 5-7. Enough ripple at the feedback due to the large ESR of the output capacitor (Figure 5-5). The converter is stable without any additional ripple injection at the FB node. The feedback voltage ripple is given by Equation 5-26. 2016 Microchip Technology Inc. DS20005620A-page 23 MIC2128 5.9 L SW RINJ MIC2128 R1 CFF CINJ FB COUT ESR R2 FIGURE 5-7: Invisible Ripple at FB. The injected ripple at the FB pin in this case is given by the Equation 5-28:. EQUATION 5-28: V 1 – D CFF RINJ f SW OUT VFB PP = ------------------------------------------ In the above Equation 5-28, it is assumed that the time constant associated with the CFF meets the criteria shown in Equation 5-29. Power Dissipation in MIC2128 The MIC2128 features two Low Dropout Regulators (LDOs) to supply power at the PVDD pin from either VIN or EXTVDD depending on the voltage at the EXTVDD pin. PVDD powers MOSFET drivers and VDD pin, which is recommended to connect to PVDD through a low pass filter, powers the internal circuitry. In the applications where the output voltage is 5V and above (up to 14V), it is recommended to connect EXTVDD to the output to reduce the power dissipation in the MIC2128, to reduce the MIC2128 junction temperature and to improve the system efficiency. The power dissipation in the MIC2128 depends on the internal LDO being in use, gate charge of the external MOSFETs and switching frequency. The power dissipation and the junction temperature of the MIC2128 can be estimated using the Equation 5-31, Equation 5-32 and Equation 5-33. Power dissipation in the MIC2128 when EXTVDD is not used: EQUATION 5-31: P IC = V IN ISW + IQ EQUATION 5-29: T SW C FF R 1 R 2 RINJ The process of sizing the ripple injection resistor and capacitors is: 1. Select CINJ in the range of 47 nF to 100 nF, which can be considered as short for a wide range of the frequencies. Select CFF in the range of 0.47 nF to 10 nF, if R1 and R2 are in k range. Select RINJ according to Equation 5-30. 2. 3. EQUATION 5-30: R INJ Where: V OUT 1 – D = ------------------------------------------------------CFF fSW V FB PP VOUT = Output voltage D = Duty cycle fSW = Switching frequency VFB(pp) = Feedback Ripple Once all the ripple injection component values are calculated, ensure that the criteria shown in the Equation 5-29 is met. DS20005620A-page 24 Power dissipation in the MIC2128 when EXTVDD is used: EQUATION 5-32: PIC = V EXTVDD I SW + I Q I SW = QG f SW Q G = QG_HS + Q G_LS Where: ISW = Switching current into the VIN pin IQ = Quiescent current (1.4 mA typ) QG = Total gate charge of the external MOSFETs which is sum of the gate charge of high-side MOSFET (QG_HS) and the low-side MOSFET (QG_LS) at 5V gate to source voltage. Gate charge information can be obtained from the MOSFETs datasheet. VEXTVDD = Voltage at the EXTVDD pin (4.6 ≤ VEXTVDD ≤ 14 V typ.) The junction temperature of the MIC2128 can be estimated using Equation 5-33. 2016 Microchip Technology Inc. MIC2128 EQUATION 5-33: T J = P IC JA + T A Where: TJ = Junction temperature PIC = Power dissipation θJA = Junction Ambient Thermal resistance (50.8°C/W) The maximum recommended operating junction temperature for the MIC2128 is 125°C. Using the output voltage of the same switching regulator, when it is in between 4.6V (typ.) to 14V, as the voltage at the EXTVDD pin, significantly reduces the power dissipation inside the MIC2128. This reduces the junction temperature rise as illustrated below. For the typical case of VVIN = 48V, VOUT = 5V, the maximum ambient temperature = 85°C and 10 mA of ISW. The condition where EXTVDD is not used is shown in Equation 5-34: EQUATION 5-34: · P IC = 48V 10mA + 1.5 mA PIC = 0.552W T J = 0.552W 50.8 C W + 85 C T J = 113 C When the 5V output is used as the input to the EXTVDD pin, the MIC2128 junction temperature reduces from 113°C to 88°C as calculated in Equation 5-35: EQUATION 5-35: PIC = 5V 10mA + 1.5mA PIC = 0.058W T J = 0.058W 50.8 C W + 85 C T J = 88 C 2016 Microchip Technology Inc. DS20005620A-page 25 MIC2128 6.0 PCB LAYOUT GUIDELINES PCB layout is critical to achieve reliable, stable and efficient performance. The following guidelines should be followed to ensure proper operation of the MIC2128converter. 6.1 IC • The ceramic bypass capacitors, which are connected to the VDD and PVDD pins, must be located right at the IC. Use wide traces to connect to the VDD, PVDD and AGND, PGND pins respectively. • The signal ground pin (AGND) must be connected directly to the ground planes. • Place the IC close to the point-of-load (POL). • Signal and power grounds should be kept separate and connected at only one location. 6.2 Input Capacitor • Place the input ceramic capacitors as close as possible to the MOSFETs. • Place several vias to the ground plane close to the input capacitor ground terminal. 6.3 6.4 Output Capacitor • Use a copper plane to connect the output capacitor ground terminal to the input capacitor ground terminal. • The feedback trace should be separate from the power trace and connected as close as possible to the output capacitor. Sensing a long high-current load trace can degrade the DC load regulation. 6.5 MOSFETs • MOSFET gate drive traces must be short and wide. The ground plane should be the connection between the MOSFET source and PGND. • Chose a low-side MOSFET with a high CGS/CGD ratio and a low internal gate resistance to minimize the effect of dv/dt inducted turn-on. • Use a 4.5V VGS rated MOSFET. Its higher gate threshold voltage is more immune to glitches than a 2.5V or 3.3V rated MOSFET. Inductor • Keep the inductor connection to the switch node (SW) short. • Do not route any digital lines underneath or close to the inductor. • Keep the switch node (SW) away from the feedback (FB) pin. • The SW pin should be connected directly to the drain of the low-side MOSFET to accurately sense the voltage across the low-side MOSFET. DS20005620A-page 26 2016 Microchip Technology Inc. MIC2128 7.0 PACKAGING INFORMATION 7.1 Package Marking Information 16-Pin QFN (3 x 3 mm) Example WWNNN Legend: XX...X WW NNN e3 * Customer-specific information Week code (week of January 1 is week ‘01’) Alphanumeric traceability code Pb-free JEDEC designator for Matte Tin (Sn) This package is Pb-free. The Pb-free JEDEC designator ( e3 ) can be found on the outer packaging for this package. ●, ▲, ▼ Pin one index is identified by a dot, delta up, or delta down (triangle mark). Note: In the event the full Microchip part number cannot be marked on one line, it will be carried over to the next line, thus limiting the number of available characters for customer-specific information. Package may or may not include the corporate logo. Underbar (_) symbol may not be to scale. 2016 Microchip Technology Inc. DS20005620A-page 27 MIC2128 Note: For the most current package drawings, please see the Microchip Packaging Specification located at http://www.microchip.com/packaging. DS20005620A-page 28 2016 Microchip Technology Inc. MIC2128 APPENDIX A: REVISION HISTORY Revision A (September 2016) • Original release of this document. 2016 Microchip Technology Inc. DS20005620A-page 29 MIC2128 PRODUCT IDENTIFICATION SYSTEM To order or obtain information, e.g., on pricing or delivery, refer to the factory or the listed sales office. X PART NO. Device XX Temperature Package Code Device: MIC2128: 75V, Synchronous Buck Controller Featuring Adaptive On-Time Control with External Soft Start Temperature: Y = Pb-Free with Industrial Temperature Grade (-40°C to +125°C) Package: ML = 16-Lead, 3x3 mm QFN DS20005620A-page 30 Examples: a) MIC2128YML: Synchronous Buck Controller, -40°C to +125°C junction temperature range, 16LD QFN package 2016 Microchip Technology Inc. Note the following details of the code protection feature on Microchip devices: • Microchip products meet the specification contained in their particular Microchip Data Sheet. • Microchip believes that its family of products is one of the most secure families of its kind on the market today, when used in the intended manner and under normal conditions. • There are dishonest and possibly illegal methods used to breach the code protection feature. All of these methods, to our knowledge, require using the Microchip products in a manner outside the operating specifications contained in Microchip’s Data Sheets. Most likely, the person doing so is engaged in theft of intellectual property. • Microchip is willing to work with the customer who is concerned about the integrity of their code. • Neither Microchip nor any other semiconductor manufacturer can guarantee the security of their code. Code protection does not mean that we are guaranteeing the product as “unbreakable.” Code protection is constantly evolving. We at Microchip are committed to continuously improving the code protection features of our products. Attempts to break Microchip’s code protection feature may be a violation of the Digital Millennium Copyright Act. If such acts allow unauthorized access to your software or other copyrighted work, you may have a right to sue for relief under that Act. Information contained in this publication regarding device applications and the like is provided only for your convenience and may be superseded by updates. It is your responsibility to ensure that your application meets with your specifications. MICROCHIP MAKES NO REPRESENTATIONS OR WARRANTIES OF ANY KIND WHETHER EXPRESS OR IMPLIED, WRITTEN OR ORAL, STATUTORY OR OTHERWISE, RELATED TO THE INFORMATION, INCLUDING BUT NOT LIMITED TO ITS CONDITION, QUALITY, PERFORMANCE, MERCHANTABILITY OR FITNESS FOR PURPOSE. Microchip disclaims all liability arising from this information and its use. Use of Microchip devices in life support and/or safety applications is entirely at the buyer’s risk, and the buyer agrees to defend, indemnify and hold harmless Microchip from any and all damages, claims, suits, or expenses resulting from such use. No licenses are conveyed, implicitly or otherwise, under any Microchip intellectual property rights unless otherwise stated. Trademarks The Microchip name and logo, the Microchip logo, AnyRate, dsPIC, FlashFlex, flexPWR, Heldo, JukeBlox, KeeLoq, KeeLoq logo, Kleer, LANCheck, LINK MD, MediaLB, MOST, MOST logo, MPLAB, OptoLyzer, PIC, PICSTART, PIC32 logo, RightTouch, SpyNIC, SST, SST Logo, SuperFlash and UNI/O are registered trademarks of Microchip Technology Incorporated in the U.S.A. and other countries. ClockWorks, The Embedded Control Solutions Company, ETHERSYNCH, Hyper Speed Control, HyperLight Load, IntelliMOS, mTouch, Precision Edge, and QUIET-WIRE are registered trademarks of Microchip Technology Incorporated in the U.S.A. Analog-for-the-Digital Age, Any Capacitor, AnyIn, AnyOut, BodyCom, chipKIT, chipKIT logo, CodeGuard, dsPICDEM, dsPICDEM.net, Dynamic Average Matching, DAM, ECAN, EtherGREEN, In-Circuit Serial Programming, ICSP, Inter-Chip Connectivity, JitterBlocker, KleerNet, KleerNet logo, MiWi, motorBench, MPASM, MPF, MPLAB Certified logo, MPLIB, MPLINK, MultiTRAK, NetDetach, Omniscient Code Generation, PICDEM, PICDEM.net, PICkit, PICtail, PureSilicon, RightTouch logo, REAL ICE, Ripple Blocker, Serial Quad I/O, SQI, SuperSwitcher, SuperSwitcher II, Total Endurance, TSHARC, USBCheck, VariSense, ViewSpan, WiperLock, Wireless DNA, and ZENA are trademarks of Microchip Technology Incorporated in the U.S.A. and other countries. SQTP is a service mark of Microchip Technology Incorporated in the U.S.A. Microchip received ISO/TS-16949:2009 certification for its worldwide headquarters, design and wafer fabrication facilities in Chandler and Tempe, Arizona; Gresham, Oregon and design centers in California and India. The Company’s quality system processes and procedures are for its PIC® MCUs and dsPIC® DSCs, KEELOQ® code hopping devices, Serial EEPROMs, microperipherals, nonvolatile memory and analog products. In addition, Microchip’s quality system for the design and manufacture of development systems is ISO 9001:2000 certified. QUALITY MANAGEMENT SYSTEM CERTIFIED BY DNV == ISO/TS 16949 == 2016 Microchip Technology Inc. Silicon Storage Technology is a registered trademark of Microchip Technology Inc. in other countries. GestIC is a registered trademarks of Microchip Technology Germany II GmbH & Co. KG, a subsidiary of Microchip Technology Inc., in other countries. All other trademarks mentioned herein are property of their respective companies. © 2016, Microchip Technology Incorporated, Printed in the U.S.A., All Rights Reserved. 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