Not Recommended For New Designs bq24751B www.ti.com .......................................................................................................................................................... SLUS835A – JULY 2008 – REVISED MARCH 2009 Host-Controlled Multi-Chemistry Battery Charger with Low Iq and System Power Selector – < 10 µA Off-State Battery Discharge Current – < 1.5 mA Off-State Input Quiescent Current FEATURES 1 APPLICATIONS • • • • • • Notebook and Ultra-Mobile Computers Portable Data Capture Terminals Portable Printers Medical Diagnostics Equipment Battery Bay Chargers Battery Back-up Systems DESCRIPTION The bq24751B is a high-efficiency, synchronous battery charger with integrated compensation and system power selector logic, offering low component count for space-constrained multi-chemistry battery charging applications. Ratiometric charge current and voltage programming allows gor high regulation accuracies, and can be either hardwired with resistors or programmed by the system power-management microcontroller using a DAC or GPIOs. PGND LODRV REGN HIDRV PH BTST PVCC The bq24751B charges two, three, or four series Li+ cells, supporting up to 10 A of charge current, and is available in a 28-pin, 5x5-mm thin QFN package. 28 27 26 25 24 23 22 LEARN 2 20 CELLS ACP 3 bq24751B 19 SRP ACDRV 4 28 LD QFN 18 SRN ACDET 5 TOP VIEW 17 BAT ACSET 6 16 SRSET ACOP 7 15 IADAPT BATDRV ACGOOD 10 11 12 13 14 VADJ 9 VDAC 8 VREF 21 ACN AGND CHGEN 1 OVPSET • NMOS-NMOS Synchronous Buck Converter with 300 kHz Frequency and >95% Efficiency • 30-ns Minimum Driver Dead-time and 99.5% Maximum Effective Duty Cycle • High-Accuracy Voltage and Current Regulation – ±0.5% Charge Voltage Accuracy – ±3% Charge Current Accuracy – ±3% Adapter Current Accuracy – ±2% Input Current Sense Amp Accuracy • Integration – Automatic System Power Selection From AC/DC Adapter or Battery – Internal Loop Compensation – Internal Soft Start • Safety – Programmable Input Overvoltage Protection (OVP) – Dynamic Power Management (DPM) with Status Indicator – Programmable Inrush Adapter Power (ACOP) and Overcurrent (ACOC) Limits – Reverse-Conduction Protection Input FET • Supports Two, Three, or Four Li+ Cells • 5 – 24 V AC/DC-Adapter Operating Range • Analog Inputs with Ratiometric Programming via Resistors or DAC/GPIO Host Control – Charge Voltage (4-4.512 V/cell) – Charge Current (up to 10 A, with 10-mΩ Sense Resistor) – Adapter Current Limit (DPM) • Status and Monitoring Outputs – AC/DC Adapter Present with Programmable Voltage Threshold – Current Drawn from Input Source • Battery Learn Cycle Control • Supports Any Battery Chemistry: Li+, NiCd, NiMH, Lead Acid, etc. • Charge Enable • 28-pin, 5x5-mm QFN package • Energy Star Low Iq 2 1 2 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. All trademarks are the property of their respective owners. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 2008–2009, Texas Instruments Incorporated Not Recommended For New Designs bq24751B SLUS835A – JULY 2008 – REVISED MARCH 2009 .......................................................................................................................................................... www.ti.com These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. DESCRIPTION (CONTINUED) The bq24751B controls external switches to prevent battery discharge back to the input, connect the adapter to the system, and to connect the battery to the system using 6-V gate drives for better system efficiency. For maximum system safety, inrush-power limiting provides instantaneous response to high input voltage multiplied by current. This AC Overpower protection (ACOP) feature limits the input-switch power to the programmed level on the ACOP pin, and latches off if the high-power condition persists to prevent overheating. The bq24751B features Dynamic Power Management (DPM) and input power limiting. These features reduce battery charge current when the input power limit is reached to avoid overloading the AC adapter when supplying the load and the battery charger simultaneously. A highly-accurate current-sense amplifier enables precise measurement of input current from the AC adapter to monitor the overall system power. ADAPTER + SYSTEM R10 2Ω ADAPTER - C1 2.2 µF P Q1 (ACFET) SI4435 C6 10 µF RAC 0.010 Ω P C7 10 µF Q2 (ACFET) SI4435 C3 C2 0.1 µF 0.1 µF ACN PVCC C8 0.1 µF ACP /ACDRV ACDET R2 66.5 kΩ 1% VREF R5 10 kΩ /ACGOOD bq24751B 422 kΩ 1% L1 8.2 µH PH OVPSET REGN R4 71 kΩ 1% C9 0.1 µF LODRV ACSET N C4 0.1 µF C14 0.1 µF SRP LEARN SRN CELLS BAT C15 0.1 µF /CHGEN ACOP VDAC ADC PACK- PGND VREF DAC C12 10 µF C11 10 µF C13 0.1 µF Q5 FDS6680A SRSET DAC GPIO RSR 0.010 Ω PACK+ D1 BAT54 C10 1 µF HOST P BTST /ACGOOD R3 Q4 FDS6680A HIDRV AGND N 432 kΩ 1% Q3(BATFET) SI4435 /BATDRV R1 C16 0.47 µF VADJ IADAPT PowerPad C5 100 pF (1) Pull-up rail could be either VREF or other system rail. (2) SRSET/ACSET could come from either DAC or resistor dividers. VIN = 20 V, VBAT = 3-cell Li-Ion, Icharge = 3 A, Iadapter_limit = 4 A Figure 1. Typical System Schematic, Voltage and Current Programmed by DAC 2 Submit Documentation Feedback Copyright © 2008–2009, Texas Instruments Incorporated Product Folder Link(s) :bq24751B Not Recommended For New Designs bq24751B www.ti.com .......................................................................................................................................................... SLUS835A – JULY 2008 – REVISED MARCH 2009 ADAPTER + SYSTEM R10 2Ω ADAPTER - C1 2.2 µF P Q1 (ACFET) SI4435 C6 10 µF RAC 0.010 Ω P C7 10 µF Q2 (ACFET) SI4435 C3 C2 0.1 µF ACN 0.1 µF PVCC C8 0.1 µF ACP /ACDRV ACDET R2 66.5 kΩ 1% R5 10 kΩ bq24751B /ACGOOD 422 kΩ 1% VREF R4 71 kΩ 1% R8 100 kΩ REGN R7 100 kΩ R11 C9 0.1 µF C10 1 µF LODRV ACSET C14 0.1 µF N SRP LEARN GPIO SRN CELLS BAT C15 0.1 µF /CHGEN VREF ACOP VDAC C16 0.47 µF REGN VADJ ADC PACK- PGND VREF C4 0.1 µF HOST C12 10 µF C11 10 µF C13 0.1 µF Q5 FDS6680A SRSET 43 kΩ R9 66.5 kΩ RSR 0.010 Ω PACK+ D1 BAT54 OVPSET VREF L1 8.2 µH PH P BTST /ACGOOD R3 Q4 FDS6680A HIDRV AGND VREF N 432 kΩ 1% Q3(BATFET) SI4435 /BATDRV R1 IADAPT PowerPad C5 100 pF (1) Pull-up rail could be either VREF or other system rail. (2) SRSET/ACSET could come from either DAC or resistor dividers. VIN = 20 V, VBAT = 3-cell Li-Ion, Icharge = 3 A, Iadapter_limit = 4 A Figure 2. Typical System Schematic, Voltage and Current Programmed by Resistor ORDERING INFORMATION PART NUMBER PACKAGE bq24751B 28-PIN 5 x 5 mm QFN ORDERING NUMBER (Tape and Reel) QUANTITY bq24751BRHDR 3000 bq24751BRHDT 250 PACKAGE THERMAL DATA (1) (2) PACKAGE θJA TA = 70°C POWER RATING DERATING FACTOR ABOVE TA = 70°C QFN – RHD (1) (2) 39°C/W 2.36 W 0.028 W/°C For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI Web site at www.ti.com. This data is based on using the JEDEC High-K board and the exposed die pad is connected to a Cu pad on the board. This is connected to the ground plane by a 2x3 via matrix. Submit Documentation Feedback Copyright © 2008–2009, Texas Instruments Incorporated Product Folder Link(s) :bq24751B 3 Not Recommended For New Designs bq24751B SLUS835A – JULY 2008 – REVISED MARCH 2009 .......................................................................................................................................................... www.ti.com Table 1. PIN FUNCTIONS – 28-PIN QFN PIN NAME DESCRIPTION NO. CHGEN 1 Charge enable active-low logic input. LO enables charge. HI disables charge. ACN 2 Adapter current sense resistor, negative input. A 0.1-µF ceramic capacitor is placed from ACN to ACP to provide differential-mode filtering. An optional 0.1-µF ceramic capacitor is placed from ACN pin to AGND for common-mode filtering. ACP 3 Adapter current sense resistor, positive input. A 0.1-µF ceramic capacitor is placed from ACN to ACP to provide differential-mode filtering. A 0.1-µF ceramic capacitor is placed from ACP pin to AGND for common-mode filtering. ACDRV 4 AC adapter to system-switch driver output. Connect directly to the gate of the ACFET P-channel power MOSFET and the reverse conduction blocking P-channel power MOSFET. Connect both FETs as common-source. Connect the ACFET drain to the system-load side. The PVCC should be connected to the common-source node to ensure that the driver logic is always active when needed. If needed, an optional capacitor from gate to source of the ACFET is used to slow down the ON and OFF times. The internal gate drive is asymmetrical, allowing a quick turn-off and slower turn-on in addition to the internal break-before-make logic with respect to the BATDRV. The output goes into linear regulation mode when the input sensed current exceeds the ACOC threshold. ACDRV is latched off after ACOP voltage exceeds 2 V, to protect the charging system from an ACFET-overpower condition. ACDET 5 Adapter detected voltage set input. Program the adapter detect threshold by connecting a resistor divider from adapter input to ACDET pin to AGND pin. Adapter voltage is detected if ACDET-pin voltage is greater than 2.4 V. The IADAPT current sense amplifier is active when the ACDET pin voltage is greater than 0.6 V. 6 Adapter current set input. The voltage ratio of ACSET voltage versus VDAC voltage programs the input current regulation set-point during Dynamic Power Management (DPM). Program by connecting a resistor divider from VDAC to ACSET to AGND; or by connecting the output of an external DAC to the ACSET pin and connect the DAC supply to the VDAC pin. 7 Input power limit set input. Program the input overpower time constant by placing a ceramic capacitor from ACOP to AGND. The capacitor sets the time that the input current limit, ACOC, can be sustained before exceeding the power-MOSFET power limit. When the ACOP voltage exceeds 2 V, then the ACDRV latches off to protect the charge system from an overpower condition, ACOP. Reset latch by toggling ACDET or PVCC_UVLO. OVPSET 8 Set input over voltage protection threshold. Charge is disabled and ACDRV is turned off if adapter input voltage is higher than the OVPSET programmed threshold. Input overvoltage, ACOV, disables charge and ACDRV when OVPSET > 3.1 V. ACOV does not latch. Program the overvoltage protection threshold by connecting a resistor divider from adapter input to OVPSET pin to AGND pin. AGND 9 Analog ground. Ground connection for low-current sensitive analog and digital signals. On PCB layout, connect to the analog ground plane, and only connect to PGND through the PowerPad underneath the IC. VREF 10 3.3-V regulated voltage output. Place a 0.1-µF ceramic capacitor from VREF to AGND pin close to the IC. This voltage could be used for ratiometric programming of voltage and current regulation. Do not apply a external voltage source on this pin. VDAC 11 Charge voltage set reference input. Connect the VREF or external DAC voltage source to the VDAC pin. Battery voltage, charge current, and input current are programmed as a ratio of the VDAC pin voltage versus the VADJ, SRSET, and ACSET pin voltages, respectively. Place resistor dividers from VDAC to VADJ, SRSET, and ACSET pins to AGND for programming. A DAC could be used by connecting the DAC supply to VDAC and connecting the output to VADJ, SRSET, or ACSET. VADJ 12 Charge voltage set input. The voltage ratio of VADJ voltage versus VDAC voltage programs the battery voltage regulation set-point. Program by connecting a resistor divider from VDAC to VADJ, to AGND; or, by connecting the output of an external DAC to VADJ, and connect the DAC supply to VDAC. VADJ connected to REGN programs the default of 4.2 V per cell. ACGOOD 13 Valid adapter active-low detect logic open-drain output. Pulled low when Input voltage is above programmed ACDET. Connect a 10-kΩ pullup resistor from ACGOOD to VREF, or to a different pullup-supply rail. BATDRV 14 Battery to system switch driver output. Gate drive for the battery to system load BAT PMOS power FET to isolate the system from the battery to prevent current flow from the system to the battery, while allowing a low impedance path from battery to system and while discharging the battery pack to the system load. Connect this pin directly to the gate of the input BAT P-channel power MOSFET. Connect the source of the FET to the system load voltage node. Connect the drain of the FET to the battery pack positive node. An optional capacitor is placed from the gate to the source to slow down the switching times. The internal gate drive is asymmetrical to allow a quick turn-off and slower turn-on, in addition to the internal break-before-make logic with respect to ACDRV. IADAPT 15 Adapter current sense amplifier output. IADAPT voltage is 20 times the differential voltage across ACP-ACN. Place a 100-pF or less ceramic decoupling capacitor from IADAPT to AGND. SRSET 16 Charge current set input. The voltage ratio of SRSET voltage versus VDAC voltage programs the charge current regulation set-point. Program by connecting a resistor divider from VDAC to SRSET to AGND; or by connecting the output of an external DAC to SRSET pin and connect the DAC supply to VDAC pin. BAT 17 Battery voltage remote sense. Directly connect a kelvin sense trace from the battery pack positive terminal to the BAT pin to accurately sense the battery pack voltage. Place a 0.1-µF capacitor from BAT to AGND close to the IC to filter high-frequency noise. ACSET ACOP 4 Submit Documentation Feedback Copyright © 2008–2009, Texas Instruments Incorporated Product Folder Link(s) :bq24751B Not Recommended For New Designs bq24751B www.ti.com .......................................................................................................................................................... SLUS835A – JULY 2008 – REVISED MARCH 2009 Table 1. PIN FUNCTIONS – 28-PIN QFN (continued) PIN NAME DESCRIPTION NO. SRN 18 Charge current sense resistor, negative input. A 0.1-µF ceramic capacitor is placed from SRN to SRP to provide differential-mode filtering. An optional 0.1-µF ceramic capacitor is placed from SRN pin to AGND for common-mode filtering. SRP 19 Charge current sense resistor, positive input. A 0.1-µF ceramic capacitor is placed from SRN to SRP to provide differential-mode filtering. A 0.1-µF ceramic capacitor is placed from SRP pin to AGND for common-mode filtering. CELLS 20 2, 3 or 4 cells selection logic input. Logic low programs 3 cell. Logic high programs 4 cell. Floating programs 2 cell. LEARN 21 Learn mode logic input control pin — logic high to override system selector when adapter is present, the battery is discharged to recalibrate the battery-pack gas gauge. When adapter is present and LEARN is high, battery charging is disabled, the adapter is disconnected (ACDRV is off), and the battery is connected to system (BATDRV is on). System selector automatically switches to adapter if battery is discharged below LOWBAT (3 V). When adapter is present and LEARN is low, the adapter is connected to system in normal selector logic (ACDRV is on and BATDRV is off), allowing battery charging. If adapter is not present, the battery is always connected to the system (ACDRV is off and BATDRV is on). PGND 22 Power ground. Ground connection for high-current power converter node. On PCB layout, connect directly to source of low-side power MOSFET, to ground connection of in put and output capacitors of the charger. Only connect to AGND through the PowerPad underneath the IC. LODRV 23 PWM low side driver output. Connect to the gate of the low-side power MOSFET with a short trace. REGN 24 PWM low side driver positive 6-V supply output. Connect a 1-µF ceramic capacitor from REGN to PGND, close to the IC. Use for high-side driver bootstrap voltage by connecting a small-signal Schottky diode from REGN to BTST. REGN is disabled when CHGEN is high. PH 25 PWM high side driver negative supply. Connect to the phase switching node (junction of the low-side power MOSFET drain, high-side power MOSFET source, and output inductor). Connect the 0.1-µF bootstrap capacitor from from PH to BTST. HIDRV 26 PWM high side driver output. Connect to the gate of the high-side power MOSFET with a short trace. BTST 27 PWM high side driver positive supply. Connect a 0.1-µF bootstrap ceramic capacitor from BTST to PH. Connect a small bootstrap Schottky diode from REGN to BTST. PVCC 28 IC power positive supply. Connect to the common-source (diode-OR) point: source of high-side P-channel MOSFET and source of reverse-blocking power P-channel MOSFET. Place a 0.1-µF ceramic capacitor from PVCC to PGND pin close to the IC. PowerPad Exposed pad beneath the IC. AGND and PGND star-connected only at the PowerPad plane. Always solder PowerPad to the board, and have vias on the PowerPad plane connecting to AGND and PGND planes. It also serves as a thermal pad to dissipate the heat. ABSOLUTE MAXIMUM RATINGS over operating free-air temperature range (unless otherwise noted) (1) (2) VALUE PVCC, ACP, ACN, SRP, SRN, BAT, BATDRV, ACDRV Voltage range Maximum difference voltage –0.3 to 30 PH –1 to 30 REGN, LODRV, VREF, VDAC, VADJ, ACSET, SRSET, ACDET, ACOP, CHGEN, CELLS, STAT, ACGOOD, LEARN, OVPSET –0.3 to 7 VREF, IADAPT –0.3 to 36 ACP–ACN, SRP–SRN, AGND–PGND –0.5 to 0.5 Junction temperature range, TJ –40 to 155 –55 to 155 (2) V –0.3 to 3.6 BTST, HIDRV with respect to AGND and PGND Storage temperature range, Tstg (1) UNIT V °C Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under recommended operating conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. All voltages are with respect to GND if not specified. Currents are positive into, negative out of the specified terminal. Consult Packaging Section of the data book for thermal limitations and considerations of packages. Submit Documentation Feedback Copyright © 2008–2009, Texas Instruments Incorporated Product Folder Link(s) :bq24751B 5 Not Recommended For New Designs bq24751B SLUS835A – JULY 2008 – REVISED MARCH 2009 .......................................................................................................................................................... www.ti.com RECOMMENDED OPERATING CONDITIONS over operating free-air temperature range (unless otherwise noted) MIN NOM MAX –1 24 V PVCC, ACP, ACN, SRP, SRN, BAT, BATDRV, ACDRV 0 24 V REGN, LODRV 0 6.5 V VDAC, IADAPT 0 3.6 V PH Voltage range VREF UNIT 3.3 V ACSET, SRSET, TS, ACDET, ACOP, CHGEN, CELLS, ACGOOD, LEARN, OVPSET 0 5.5 VADJ 0 6.5 V BTST, HIDRV with respect to AGND and PGND 0 30 V AGND, PGND –0.3 0.3 V ACP–ACN, SRP–SRN –0.3 0.3 V Junction temperature range, TJ –40 125 Storage temperature range, Tstg –55 150 Maximum difference voltage V °C ELECTRICAL CHARACTERISTICS 7 V ≤ VPVCC ≤ 24 V, 0°C < TJ < 125°C, typical values are at TA = 25°C, with respect to AGND (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT OPERATING CONDITIONS VPVCC_OP PVCC Input voltage operating range 5 24 V CHARGE VOLTAGE REGULATION VBAT_REG_RNG BAT voltage regulation range VVDAC_OP VDAC reference voltage range VADJ_OP VADJ voltage range 4-4.512 V per cell, times 2,3,4 cells Charge voltage regulation set to default to 4.2 V per cell 18.048 V 3.6 V V 0 REGN –0.5% 0.5% 12 V, 12.6 V, 13.536 V –0.5% 0.5% 16 V, 16.8 V, 18.048 V –0.5% 0.5% VADJ connected to REGN, 8.4 V, 12.6 V, 16.8 V –0.5% 0.5% 0 100 0 VDAC –3% 3% 8 V, 8.4 V, 9.024 V Charge voltage regulation accuracy 8 2.6 CHARGE CURRENT REGULATION VIREG_CHG Charge current regulation differential voltage range VSRSET_OP SRSET voltage range VIREG_CHG = VSRP – VSRN VIREG_CHG = 40–100 mV Charge current regulation accuracy VIREG_CHG = 20 mV –5% 5% VIREG_CHG = 5 mV –25% 25% VIREG_CHG = 1.5 mV (VBAT>4V) –33% 33% 0 100 0 VDAC –3% 3% mV V INPUT CURRENT REGULATION VIREG_DPM Adapter current regulation differential voltage range VACSET_OP ACSET voltage range VIREG_DPM = VACP – VACN VIREG_DPM = 40–100 mV Input current regulation accuracy VIREG_DPM = 20 mV –5% 5% VIREG_DPM = 5 mV –25% 25% VIREG_DPM = 1.5 mV –33% 33% 3.267 mV V VREF REGULATOR VVREF_REG VREF regulator voltage VACDET > 0.6 V, 0-30 mA IVREF_LIM VREF current limit VVREF = 0 V, VACDET > 0.6 V 6 Submit Documentation Feedback 35 3.3 3.333 80 V mA Copyright © 2008–2009, Texas Instruments Incorporated Product Folder Link(s) :bq24751B Not Recommended For New Designs bq24751B www.ti.com .......................................................................................................................................................... SLUS835A – JULY 2008 – REVISED MARCH 2009 ELECTRICAL CHARACTERISTICS (continued) 7 V ≤ VPVCC ≤ 24 V, 0°C < TJ < 125°C, typical values are at TA = 25°C, with respect to AGND (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP 5.9 MAX UNIT REGN REGULATOR VREGN_REG REGN regulator voltage VACDET > 0.6 V, 0-75 mA, PVCC > 10 V 5.6 6.2 V IREGN_LIM REGN current limit VREGN = 0 V, VACDET > 0.6 V 90 135 mA 0 24 0 2 ADAPTER CURRENT SENSE AMPLIFIER VACP/N_OP Input common mode range VIADAPT IADAPT output voltage range IIADAPT IADAPT output current AIADAPT Current sense amplifier voltage gain Voltage on ACP/ACN 0 AIADAPT = VIADAPT / VIREG_DPM Adapter current sense accuracy 1 20 VIREG_DPM = 40–100 mV –2% –3% 3% VIREG_DPM = 5 mV –25% 25% VIREG_DPM = 1.5 mV –33% 33% Output current limit VIADAPT = 0 V CIADAPT_MAX Maximum output load capacitance For stability with 0 mA to 1 mA load mA V/V VIREG_DPM = 20 mV IIADAPT_LIM V 2% 1 mA 100 pF ACDET COMPARATOR VACDET_CHG ACDET adapter-detect rising threshold Min voltage to enable charging, VACDET rising VACDET_CHG_HYS ACDET falling hysteresis VACDET falling 2.376 2.40 2.424 ACDET rising deglitch VACDET rising 518 700 908 ms ACDET falling deglitch VACDET falling 7 9 11 ms 0.56 0.62 0.68 40 V mV VACDET_BIAS ACDET enable-bias rising threshold Min voltage to enable all bias, VACDET rising VACDET_BIAS_HYS Adapter present falling hysteresis VACDET falling 20 mV V ACDET rising deglitch VACDET rising 10 µs ACDET falling deglitch VACDET falling 10 µs PVCC / BAT COMPARATOR (REVERSE DISCHARGING PROTECTION) VPVCC-BAT_OP Differential Voltage from PVCC to BAT VPVCC-BAT_FALL PVCC to BAT falling threshold VPVCC-BAT__HYS PVCC to BAT hysteresis –20 VPVCC – VBAT to turn off ACFET 140 185 24 V 240 mV 50 PVCC to BAT Rising Deglitch VPVCC – VBAT > VPVCC-BAT_RISE PVCC to BAT Falling Deglitch VPVCC – VBAT < VPVCC-BAT_FALL 7 9 mV 11 ms µs 6 INPUT UNDERVOLTAGE LOCK-OUT COMPARATOR (UVLO) UVLO AC Undervoltage rising threshold UVLO(HYS) AC Undervoltage hysteresis, falling Measured on PVCC 3.5 4 4.5 260 V mV AC LOWVOLTAGE COMPARATOR (ACLOWV) VACLOWV AC lowvoltage rising threshold AC lowvoltage falling threshold 3.6 Measure on ACP pin V 3 ACN / BAT COMPARATOR VACN-BAT_FALL ACN to BAT falling threshold VACN-BAT_HYS ACN to BAT hysteresis VACN – VBAT to turn on BATDRV 175 285 340 mV 50 mV ACN to BAT rising deglitch VACN – VBAT > VACN-BAT_RISE 20 µs ACN to BAT falling deglitch VACN – VBAT < VACN-BAT_FALL 6 µs BAT OVERVOLTAGE COMPARATOR VOV_RISE Overvoltage rising threshold As percentage of VBAT_REG 104% VOV_FALL Overvoltage falling threshold As percentage of VBAT_REG 102% Submit Documentation Feedback Copyright © 2008–2009, Texas Instruments Incorporated Product Folder Link(s) :bq24751B 7 Not Recommended For New Designs bq24751B SLUS835A – JULY 2008 – REVISED MARCH 2009 .......................................................................................................................................................... www.ti.com ELECTRICAL CHARACTERISTICS (continued) 7 V ≤ VPVCC ≤ 24 V, 0°C < TJ < 125°C, typical values are at TA = 25°C, with respect to AGND (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX 2.9 3.045 UNIT BAT SHORT (UNDERVOLTAGE) COMPARATOR VBAT_SHORT_FALL BAT short falling threshold VBAT falling VBAT_SHORT_HYS BAT short hysteresis VBAT rising 2.755 250 V/cell BAT short rising deglitch VBAT > VBAT_SHORT+VBAT_SHORT_HYS Detection delay 1.5 BAT short falling deglitch VBAT < VBAT_SHORT 1.5 BATSHORT EXIT delay to turn on BATFET and turn off ACFET LEARN=HIGH 600 ms BATSHORT ENTRY delay to turn off BATFET and turn on ACFET LEARN=HIGH 10 µs mV/cell s CHARGE OVERCURRENT COMPARATOR VOC Charge overcurrent falling threshold OCP Floor Minimum current Limit (SRP-SRN) OCP Threshold As percentage of IREG_CHG 145% 50 mV falling threshold 0.1xSRSET/VDAC falling 33.75 mV rising threshold 0.1xSRSET/VDAC rising 42.5 mV CHARGE UNDERCURRENT COMPARATOR (SYNCHRONOUS TO NON-SYNCHRONOUS TRANSITION) VISYNSET_FALL Charge undercurrent falling threshold VISYNSET_HYS Charge undercurrent rising hysteresis Charge undercurrent, falling-current deglitch Changing from synchronous to non-sysnchronous 9.75 13 16.25 8 mV mV 20 µs VIREG_DPM < VISYNSET Charge undercurrent, rising-current deglitch 640 INPUT OVERPOWER COMPARATOR (ACOP) VACOC VACOC_CEILING ACOC Gain for initial ACOC current limit limit (Percentage of programmed VIREG_DPM) Begins 700 ms after ACDET, Input current limited to this threshold for fault protection 150 % VIREG_DPM Maximum ACOC input current limit (VACP–VACN)max Internally limited ceiling, VACOC_MAX = (VACP–VACN)max 100 mV ACOP Latch Blankout Time with ACOC active (begins 700 ms after ACDET) Begins 700 ms after ACDET (does not allow ACOP latch-off, and no ACOP source current) VACOP ACOP pin latch-off threshold voltage (See ACOP in Terminal Functions table) KACOP Gain for ACOP Source Current when in ACOC Current source on when in ACOC limit. Function of voltage across power FET IACOP_SOURCE = KACOP ×(VPVCC -VACP) IACOP_SINK ACOP Sink Current when not in ACOC ACOP Latch is reset by going below ACDET or UVLO Current sink on when not in ACOC VACN-SHORT ACN Short protection threshold latching ACN < 2.4 V, ACDET > 2.4 V 2 ms 1.95 2 2.05 V 18 µA / V 5 µA 2.4 V INPUT OVERVOLTAGE COMPARATOR (ACOV) AC Overvoltage rising threshold on OVPSET (See OVPSET in Table 1) VACOV VACOV_HYS Measured on OVPSET 3.007 3.1 AC Overvoltage rising deglitch 1.3 AC Overvoltage falling deglitch 1.3 3.193 V ms THERMAL SHUTDOWN COMPARATOR TSHUT Thermal shutdown rising temperature TSHUT_HYS Thermal shutdown hysteresis, falling Temperature Increasing 155 °C 20 °C BATTERY SWITCH (BATDRV) DRIVER RDS(off)_BAT BATFET Turn-off resistance VACN > 5 V 160 Ω RDS(on)_BAT BATFET Turn-on resistance VACN > 5 V 3 kΩ V/BATDRV_REG BATFET drive voltage V/BATDRV_REG = VACN – VBATDRV when VACN > 5 V and BATFET is on BATFET Power-up delay Delay to turn off BATFET after adapter is detected (after VACDET > 2.4 V) 8 Submit Documentation Feedback 6.5 518 700 V 908 ms Copyright © 2008–2009, Texas Instruments Incorporated Product Folder Link(s) :bq24751B Not Recommended For New Designs bq24751B www.ti.com .......................................................................................................................................................... SLUS835A – JULY 2008 – REVISED MARCH 2009 ELECTRICAL CHARACTERISTICS (continued) 7 V ≤ VPVCC ≤ 24 V, 0°C < TJ < 125°C, typical values are at TA = 25°C, with respect to AGND (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT AC SWITCH (ACDRV) DRIVER RDS(off)_AC ACFET turn-off resistance VPVCC > 5 V 80 Ω RDS(on)_AC ACFET turn-on resistance VPVCC > 5 V 2.5 kΩ V/ACDRV_REG ACFET drive voltage V/ACDRV_REG = VPVCC – VACDRV when VPVCC > 5 V and ACFET is on ACFET Power-up Delay Delay to turn on ACFET after adapter is detected (after VACDET > 2.4 V) 6.5 518 700 V 908 ms AC / BAT MOSFET DRIVERS TIMING Dead time when switching between ACDRV and BATDRV Driver dead time µs 10 PWM HIGH SIDE DRIVER (HIDRV) RDS(on)_HI High side driver turn-on resistance VBTST – VPH = 5.5 V, tested at 100 mA 3 6 Ω RDS(off)_HI High side driver turn-off resistance VBTST – VPH = 5.5 V, tested at 100 mA 0.7 1.4 Ω VBTST_REFRESH Bootstrap refresh comparator threshold voltage VBTST – VPH when low side refresh pulse is requested 4 V PWM LOW SIDE DRIVER (LODRV) RDS(on)_LO Low side driver turn-on resistance REGN = 6 V, tested at 100 mA 3 6 Ω RDS(off)_LO Low side driver turn-off resistance REGN = 6 V, tested at 100 mA 0.6 1.2 Ω PWM DRIVERS TIMING Driver Dead Time — Dead time when switching between LODRV and HIDRV. No load at LODRV and HIDRV 30 ns PWM OSCILLATOR FSW PWM switching frequency VRAMP_HEIGHT PWM ramp height 240 As percentage of PVCC 300 360 6.6 kHz %PVCC QUIESCENT CURRENT IOFF_STATE Total off-state quiescent current into pins SRP, SRN, BAT, BTST, PH, PVCC, ACP, ACN VBAT = 16.8 V, VACDET < 0.6 V, VPVCC > 5 V, TJ = 0 to 85°C IBATQ_CD Total quiescent current into pins: SRP, SRN, BAT, BTST, PH Adapter present, VACDET > 2.4 V, charge disabled IAC Adapter quiescent current VPVCC = 20 V, charge disabled 7 10 µA 100 200 µA 1 1.5 mA INTERNAL SOFT START (8 steps to regulation current) Soft start steps Soft start step time 8 step 1.7 ms CHARGER SECTION POWER-UP SEQUENCING Charge-enable delay after power-up Delay from when adapter is detected to when the charger is allowed to turn on 518 700 908 ms LOGIC INPUT PIN CHARACTERISTICS (CHGEN, LEARN) VIN_LO Input low threshold voltage VIN_HI Input high threshold voltage 0.8 2.1 IBIAS Input bias current VCHGEN = 0 to VREGN tCHGEN_DEGLITCH Charge enable deglitch time ACDET > 2.4 V, CHGEN rising 1 2 V µA ms LOGIC INPUT PIN CHARACTERISTICS (CELLS) VIN_LO Input low threshold voltage, 3 cells CELLS voltage falling edge VIN_MID Input mid threshold voltage, 2 cells CELLS voltage rising for MIN, CELLS voltage falling for MAX 0.5 0.8 VIN_HI Input high threshold voltage, 4 cells CELLS voltage rising 2.5 IBIAS_FLOAT Input bias float current for 2-cell selection VCHGEN = 0 to VREGN –1 1.8 V 1 µA OPEN-DRAIN LOGIC OUTPUT PIN CHARACTERISTICS (ACGOOD) VOUT_LO 0.5 V Delay, ACGOOD falling Output low saturation voltage Sink Current = 5 mA 518 700 908 ms Delay, ACGOOD rising 7 9 11 ms Submit Documentation Feedback Copyright © 2008–2009, Texas Instruments Incorporated Product Folder Link(s) :bq24751B 9 Not Recommended For New Designs bq24751B SLUS835A – JULY 2008 – REVISED MARCH 2009 .......................................................................................................................................................... www.ti.com TYPICAL CHARACTERISTIC Table of Graphs (1) Figure VREF Load and Line Regulation vs Load Current Figure 3 REGN Load and Line Regulation vs Load Current Figure 4 BAT Voltage vs VADJ/VDAC Ratio Figure 5 Charge Current vs SRSET/VDAC Ratio Figure 6 Input Current vs ACSET/VDAC Ratio Figure 7 BAT Voltage Regulation Accuracy vs Charge Current Figure 8 BAT Voltage Regulation Accuracy Figure 9 Charge Current Regulation Accuracy Figure 10 Input Current Regulation (DPM) Accuracy Figure 11 VIADAPT Input Current Sense Amplifier Accuracy Input Regulation Current (DPM), and Charge Current Figure 12 vs System Current Figure 13 Transient System Load (DPM) Response Figure 14 Charge Current Regulation vs BAT Voltage Figure 15 Efficiency vs Battery Charge Current Figure 16 Battery Removal (from Constant Current Mode) Figure 17 ACDRV and BATDRV Startup Figure 18 REF and REGN Startup Figure 19 System Selector on Adapter Insertion with 390-µF SYS-to-PGND System Capacitor Figure 20 System Selector on Adapter Removal with 390-µF SYS-to-PGND System Capacitor Figure 21 System Selector LEARN Turn-On with 390-µF SYS-to-PGND System Capacitor Figure 22 System Selector LEARN Turn-Off with 390-µF SYS-to-PGND System Capacitor Figure 23 System selector on Adapter Insertion Figure 24 Selector Gate Drive Voltages, 700 ms delay after ACDET Figure 25 Charger on Adapter Removal Figure 26 Charge Enable / Disable and Current Soft-Start Figure 27 Nonsynchronous to Synchronous Transition Figure 28 Synchronous to Nonsynchronous Transition Figure 29 Near 100% Duty Cycle Bootstrap Recharge Pulse Figure 30 Battery Shorted Charger Response, Over Current Protection (OCP) and Charge Current Regulation Figure 31 Continuous Conduction Mode (CCM) Switching Waveforms Figure 32 Discontinuous Conduction Mode (DCM) Switching Waveforms Figure 33 (1) 10 Test results based on Figure 2 application schematic. VIN = 20 V, VBAT = 3-cell Li-Ion, ICHG = 3 A, IADAPTER_LIMIT = 4 A, TA = 25°C, unless otherwise specified. Submit Documentation Feedback Copyright © 2008–2009, Texas Instruments Incorporated Product Folder Link(s) :bq24751B Not Recommended For New Designs bq24751B www.ti.com .......................................................................................................................................................... SLUS835A – JULY 2008 – REVISED MARCH 2009 VREF LOAD AND LINE REGULATION vs Load Current REGN LOAD AND LINE REGULATION vs LOAD CURRENT 0 0.50 -0.50 Regulation Error - % Regulation Error - % 0.40 0.30 PVCC = 10 V 0.20 0.10 0 -1 -1.50 PVCC = 10 V -2 PVCC = 20 V -0.10 -2.50 -0.20 -3 PVCC = 20 V 0 10 20 30 VREF - Load Current - mA 40 50 0 20 Figure 4. BAT VOLTAGE vs VADJ/VDAC RATIO CHARGE CURRENT vs SRSET/VDAC RATIO 70 80 10 VADJ = 0 -VDAC, 4-Cell, No Load 17.8 SRSET Varied, 4-Cell, Vbat = 16 V 9 Charge Current Regulation - A 18 17.6 17.4 17.2 17 16.8 16.6 16.4 8 7 6 5 4 3 2 1 16.2 0 16 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 0 1 0.1 0.2 VADJ/VDAC Ratio 0.3 0.4 0.5 0.6 0.7 SRSET/VDAC Ratio 0.8 0.9 Figure 5. Figure 6. INPUT CURRENT vs ACSET/VDAC RATIO BAT VOLTAGE REGULATION ACCURACY vs CHARGE CURRENT 1 0.2 10 ACSET Varied, 4-Cell, Vbat = 16 V 8 Vreg = 16.8 V Regulation Error - % 9 Input Current Regulation - A 30 40 50 60 REGN - Load Current - mA Figure 3. 18.2 Voltage Regulation - V 10 7 6 5 4 3 0.1 0 -0.1 2 1 -0.2 0 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 ACSET/VDAC Ratio 0.8 0.9 1 0 Figure 7. 2000 4000 Charge Current - mA 6000 8000 Figure 8. Submit Documentation Feedback Copyright © 2008–2009, Texas Instruments Incorporated Product Folder Link(s) :bq24751B 11 Not Recommended For New Designs bq24751B SLUS835A – JULY 2008 – REVISED MARCH 2009 .......................................................................................................................................................... www.ti.com BAT VOLTAGE REGULATION ACCURACY CHARGE CURRENT REGULATION ACCURACY 0.10 2 SRSET Varied 0 0.06 -1 0.04 Regulation Error - % Regulation Error - % 4-Cell, VBAT = 16 V 1 VADJ = 0 -VDAC 0.08 4-Cell, no load 0.02 0 -0.02 -0.04 -0.06 -2 -3 -4 -5 -6 -7 -8 -0.08 -9 -0.10 16.5 -10 17 17.5 18 18.5 0 19 2 4 I(CHRG) - Setpoint - A V(BAT) - Setpoint - V 8 Figure 9. Figure 10. INPUT CURRENT REGULATION (DPM) ACCURACY VIADAPT INPUT CURRENT SENSE AMPLIFIER ACCURACY 5 10 ACSET Varied 9 0 8 7 4-Cell, VBAT = 16 V 6 Percent Error Regulation Error - % 6 5 4 3 2 VI = 20 V, CHG = EN -5 VI = 20 V, CHG = DIS -10 -15 1 0 -20 -1 -2 -25 Iadapt Amplifier Gain 0 1 2 3 4 Input Current Regulation Setpoint - A 5 6 0 1 2 3 4 5 6 I(ACPWR) - A 7 8 9 Figure 11. Figure 12. INPUT REGULATION CURRENT (DPM), AND CHARGE CURRENT vs SYSTEM CURRENT TRANSIENT SYSTEM LOAD (DPM) RESPONSE 10 5 VI = 20 V, 4-Cell, Vbat = 16 V Ichrg and Iin - A 4 Input Current 3 2 System Current Charge Current 1 0 0 1 2 System Current - A 3 4 Figure 13. 12 Figure 14. Submit Documentation Feedback Copyright © 2008–2009, Texas Instruments Incorporated Product Folder Link(s) :bq24751B Not Recommended For New Designs bq24751B www.ti.com .......................................................................................................................................................... SLUS835A – JULY 2008 – REVISED MARCH 2009 CHARGE CURRENT REGULATION vs BAT VOLTAGE EFFICIENCY vs BATTERY CHARGE CURRENT 5 100 Vin = 20 V, Ichrg_set = 4 A, TA = 20°C 4 Cell Efficiency - % Charge Current - A 4 3 2 V(BAT) = 16.8 V Vreg = 12.6 V 90 Vreg = 8.4 V 80 1 0 0 2 4 6 8 10 12 14 Battery Voltage - V 16 18 70 0 2000 6000 4000 Battery Charge Current - mA Figure 16. BATTERY REMOVAL ACDRV AND BATDRV STARTUP Ch1 2 V/div Figure 15. 8000 Ch2 20 V/div VACDET Ch3 5 V/div VBATDRV Ch4 5 V/div VACDRV VACGOOD t − Time = 100 ms/div Figure 18. REF AND REGN STARTUP SYSTEM SELECTOR ON ADAPTER INSERTION WITH 390 µF SYS-TO-PGND SYSTEM CAPACITOR Ch4 10 V/div Ch3 Ch2 Ch1 10 V/div 20 V/div 20 V/div Figure 17. VBAT VSYS VACDRV VBATDRV t − Time = 400 ms/div Figure 19. Figure 20. Submit Documentation Feedback Copyright © 2008–2009, Texas Instruments Incorporated Product Folder Link(s) :bq24751B 13 Not Recommended For New Designs bq24751B SLUS835A – JULY 2008 – REVISED MARCH 2009 .......................................................................................................................................................... www.ti.com Ch4 10 V/div Ch3 Ch2 Ch1 10 V/div 20 V/div 20 V/div SYSTEM SELECTOR ON ADAPTER REMOVAL WITH 390 µF SYS-TO-PGND SYSTEM CAPACITOR SYSTEM SELECTOR LEARN TURN-ON WITH 390 µF SYS-TO-PGND SYSTEM CAPACITOR VBAT VSYS VACDRV VBATDRV t − Time = 2 ms/div SYSTEM SELECTOR LEARN TURN-OFF WITH 390 µF SYS-TO-PGND SYSTEM CAPACITOR SYSTEM SELECTOR ON ADAPTER INSERTION Ch4 5 A/div Ch1 7.2 V Figure 22. Ch3 Ch2 Ch1 5 V/div 20 V/div 20 V/div Figure 21. VACPWR VACDRV VACGOOD IL SELECTOR GATE DRIVE VOLTAGES, 700 MS DELAY AFTER ACDET CHARGER on ADAPTER REMOVAL VACDRV VIN VBAT VACOP IL IIN t − Time = 200 ms/div t − Time = 1 ms/div Figure 25. 14 Ch1 12.6 V Ch3 2 A/div VSYS Ch1 Ch4 5 V/div 5 V/div Figure 24. Ch1 10.8 V Ch2 Ch3 Ch4 5 A/div 500 mV/div 5 V/div Ch1 5 V/div t − Time = 400 ms/div Figure 23. Figure 26. Submit Documentation Feedback Copyright © 2008–2009, Texas Instruments Incorporated Product Folder Link(s) :bq24751B Not Recommended For New Designs bq24751B www.ti.com .......................................................................................................................................................... SLUS835A – JULY 2008 – REVISED MARCH 2009 Ch1 10 V/div NONSYNCHRONOUS TO SYNCHRONOUS TRANSITION VCHGEN Ch2 Ch3 Ch4 2 A/div 5 V/div 10 V/div Ch3 2 A/div Ch2 20 V/div Ch4 1 V/div Ch1 10 V/div CHARGE ENABLE / DISABLE AND CURRENT SOFT-START VBAT VPH VHIDRV VPH VLDDRV IL IBAT t − Time = 4 ms/div t − Time = 4 ms/div Figure 28. SYNCHRONOUS TO NONSYNCHRONOUS TRANSITION NEAR 100% DUTY CYCLE BOOTSTRAP RECHARGE PULSE Ch2 Ch1 20 V/div 20 V/div Figure 27. Ch2 10 V/div VPH Ch4 Ch3 2 A/div 5 V/div Ch3 2 A/div Ch4 5 V/div VLODRV IL VPH VHIDRV VLODRV IL t − Time = 4 ms/div Figure 30. BATTERY SHORTED CHARGER RESPONSE, OVERCURRENT PROTECTION (OCP) AND CHARGE CURRENT REGULATION CONTINUOUS CONDUCTION MODE (CCM) SWITCHING WAVEFORMS Ch1 20 V/div Figure 29. VBAT Ch3 2 A/div Ch3 5 V/div Ch2 20 V/div Ch4 10 V/div t − Time = 2 ms/div Ch4 5 A/div IL VPH VHIDRV VLODRV IL t − Time = 400 ms/div t − Time = 1 ms/div Figure 31. Figure 32. Submit Documentation Feedback Copyright © 2008–2009, Texas Instruments Incorporated Product Folder Link(s) :bq24751B 15 Not Recommended For New Designs bq24751B SLUS835A – JULY 2008 – REVISED MARCH 2009 .......................................................................................................................................................... www.ti.com Ch4 2 A/div Ch3 5 V/div Ch2 20 V/div Ch1 20 V/div DISCONTINUOUS CONDUCTION MODE (DCM) SWITCHING WAVEFORMS VPH VHIDRV VLODRV IL t − Time = 1 ms/div Figure 33. 16 Submit Documentation Feedback Copyright © 2008–2009, Texas Instruments Incorporated Product Folder Link(s) :bq24751B Not Recommended For New Designs bq24751B www.ti.com .......................................................................................................................................................... SLUS835A – JULY 2008 – REVISED MARCH 2009 FUNCTIONAL BLOCK DIAGRAM 700 ms ENA_BIAS - 0.6 V + - 2.4 V ADAPTER DETECTED ACGOOD Delay Rising + ACDET VREF PVCC ACOP + BAT ACOPDET + PVCC-6V 185 mV + - S Q R Q PVCC PVCC- BAT - 5 µA 2V ENA_SNK PVCC-6V LDO - _ + Isrc=K*V(PVCC-ACP) K=18 µA/V ENA_SRC PVCC ENA_BIAS ACOP_LATCH SYSTEM POWER SELECTOR LOGIC ACDET PVCC_UVLO CHGEN ACDRV PVCC-6V ACN ENA_BIAS VREF 3.3 V LDO VREFGOOD EAI PVCC EAO BATDRV ACN-6V UVLO ACP ACFET_ON FBO + V(ACP-ACN) - IIN_REG - COMP ERROR AMPLIFIER + ACN + V(ACN-BAT) + _ + 1V VBAT_REG BTST CHGEN - - 285 mV BAT LEARN IIN_ER BAT_OVP BAT_ER LEVEL SHIFTER CHG_OCP + 20uA HIDRV ACOV ACOP SRP V(ACN-BAT) 3.5 mA + – SRN V(SRP-SRN) - 20x IBAT_ REG BAT_SHORT ICH_ER PH DC-DC CONVERTER PWM LOGIC UVLO + 20 µA PVCC ACLOWV 3.5 mA REGN 6V LDO VREFGOOD SYNCH CHGEN BTST CHRG_ON - REFRESH CBTST LODRV + V(SRP - SRN) + + 4V _ SYNCH - ACSET PH 13 mV +- IC Tj + 155°C - PGND TSHUT SRSET VBATSET IBATSET IINSET VADJ BAT + 104% X VBAT_REG - VBAT_REG BAT_OVP ACP ACN IBAT_REG RATIO IIN_REG PROGRAM V(SRP-SRN) + 145% X IBAT_REG - + V(IADAPT) IADAPT CHG_OCP - ACP VDAC + 20x - ACOV 3.0 V + ACLOWV + - 3.1 V +- OVPSET CELLS PVCC - UVLO + BAT 4 V +- AGND – BAT_SHORT + 2.9 V/cell + – bq24751B Submit Documentation Feedback Copyright © 2008–2009, Texas Instruments Incorporated Product Folder Link(s) :bq24751B 17 Not Recommended For New Designs bq24751B SLUS835A – JULY 2008 – REVISED MARCH 2009 .......................................................................................................................................................... www.ti.com DETAILED DESCRIPTION Battery Voltage Regulation The bq24751B uses a high-accuracy voltage regulator for charging voltage. Internal default battery voltage setting VBATT = 4.2 V × cell count. The regulation voltage is ratiometric with respect to VADC. The ratio of VADJ and VDAC provides extra 12.5% adjust range on VBATT regulation voltage. By limiting the adjust range to 12.5% of the regulation voltage, the external resistor mismatch error is reduced from ±1% to ±0.1%. Therefore, an overall voltage accuracy as good as 0.5% is maintained, while using 1% mismatched resistors. Ratiometric conversion also allows compatibility with D/As or microcontrollers (µC). The battery voltage is programmed through VADJ and VDAC using Equation 1. é æ V VBATT = cell count ´ ê 4 V + ç 0.512 ´ VADJ VVDAC êë è öù ÷ú ø úû (1) The input voltage range of VDAC is between 2.6 V and 3.6 V. VADJ is set between 0 and VDAC. VBATT defaults to 4.2 V × cell count when VADJ is connected to REGN. The CELLS pin is the logic input for selecting the cell count. Connect CELLS to the appropriate voltage level to charge 2,3, or 4 Li+ cells, as shown in Table 2. When charging other cell chemistries, use CELLS to select an output voltage range for the charger. Table 2. Cell-Count Selection CELLS CELL COUNT Float 2 AGND 3 VREF 4 The per-cell charge-termination voltage is a function of the battery chemistry. Consult the battery manufacturer to determine this voltage. The BAT pin is used to sense the battery voltage for voltage regulation and should be connected as close to the battery as possible, or directly on the output capacitor. A 0.1-µF ceramic capacitor from BAT to AGND is recommended to be as close to the BAT pin as possible to decouple high-frequency noise. Battery Current Regulation The SRSET input sets the maximum charge current. Battery current is sensed by resistor RSR connected between SRP and SRN. The full-scale differential voltage between SRP and SRN is 100 mV. Thus, for a 0.010-Ω sense resistor, the maximum charging current is 10 A. SRSET is ratiometric with respect to VDAC using Equation 2: V I CHARGE + SRSET 0.10 VVDAC R SR (2) The input voltage range of SRSET is between 0 and VDAC, up to 3.6 V. The SRP and SRN pins are used to sense across RSR, with a default value of 10 mΩ. However, resistors of other values can also be used. A larger sense-resistor value yields a larger sense voltage, and a higher regulation accuracy. However, this is at the expense of a higher conduction loss. Input Adapter Current Regulation The total input current from an AC adapter or other DC sources is a function of the system supply current and the battery charging current. System current normally fluctuates as portions of the systems are powered up or down. Without Dynamic Power Management (DPM), the source must be able to supply the maximum system current and the maximum charger input current simultaneously. By using DPM, the input current regulator reduces the charging current when the input current exceeds the input current limit set by ACSET. The current capacity of the AC adapter can be lowered, reducing system cost. Similar to setting battery-regulation current, adapter current is sensed by resistor RAC connected between ACP and ACN. Its maximum value is set by ACSET, which is ratiometric with respect to VDAC, using Equation 3. 18 Submit Documentation Feedback Copyright © 2008–2009, Texas Instruments Incorporated Product Folder Link(s) :bq24751B Not Recommended For New Designs bq24751B www.ti.com .......................................................................................................................................................... SLUS835A – JULY 2008 – REVISED MARCH 2009 V I ADAPTER + ACSET VVDAC 0.10 R AC (3) The input voltage range of ACSET is between 0 and VDAC, up to 3.6 V. The ACP and ACN pins are used to sense RAC with a default value of 10 mΩ. However, resistors of other values can also be used. A larger sense-resistor value yields a larger sense voltage, and a higher regulation accuracy. However, this is at the expense of a higher conduction loss. Adapter Detect and Power Up An external resistor voltage divider attenuates the adapter voltage to the ACDET pin. The adapter detect threshold should typically be programmed to a value greater than the maximum battery voltage and lower than the minimum-allowed adapter voltage. The ACDET divider should be placed before the ACFET in order to sense the true adapter input voltage whether the ACFET is on or off. Before the adapter is detected, BATFET stays on and ACFET turns off. If PVCC is below 4 V, the device is disabled. If ACDET is below 0.6 V but PVCC is above 4 V, part of the bias is enabled, including a crude bandgap reference, ACFET drive and BATFET drive. IADAPT is disabled and pulled down to GND. The total quiescent current is less than 10 µA. When ACDET rises above 0.6 V and PVCC is above 4 V, all the bias circuits are enabled and VREF rises to 3.3 V, and the REGN output rises to 6 V when CHGEN is LOW. IADAPT becomes valid to proportionally reflect the adapter current. When ACDET keeps rising and passes 2.4 V, a valid AC adapter is present. 700 ms later, the following occurs: • ACGOOD is pulled high through the external pull-up resistor to the host digital voltage rail; • ACFET is allowed to turn on and BATFET turns off consequently; (refer to System Power Selector) • Charging begins if all the conditions are satisfied. (refer to Enable and Disable Charging) Enable and Disable Charging The following conditions must be valid before the charge function is enabled: • CHGEN is LOW • PVCC > UVLO • Adapter is detected • Adapter voltage is higher than BAT + 185 mV • Adapter is not over voltage (ACOV) • 700 ms delay is complete after the adapter is detected plus 10 ms ACOC time • Thermal Shut (TSHUT) is not valid • TS is within the temperature qualification window • VDAC > 2.4 V • LEARN is low System Power Selector The bq24751B automatically switches between connecting the adapter or battery power to the system load. By default, the battery is connected to the system during power up or when a valid adapter is not present. When the adapter is detected, the battery is first disconnected from the system, then the adapter is connected. An automatic break-before-make algorithm prevents shoot-through currents when the selector transistors switch. The ACDRV signal drives a pair of back-to-back p-channel power MOSFETs (with sources connected together and to PVCC) connected between the adapter and ACP. The FET connected to the adapter prevents reverse discharge from the battery to the adapter when it is turned off. The p-channel FET with the drain connected to the adapter input provides reverse battery discharge protection when off; and also minimizes system power dissipation, with its low RDS(on), compared to a Schottky diode. The other p-channel FET connected to ACP separates the battery from the adapter, and provides both ACOC current limit and ACOP power limit to the system. The BATDRV signal controls a p-channel power MOSFET placed between BAT and the system. Submit Documentation Feedback Copyright © 2008–2009, Texas Instruments Incorporated Product Folder Link(s) :bq24751B 19 Not Recommended For New Designs bq24751B SLUS835A – JULY 2008 – REVISED MARCH 2009 .......................................................................................................................................................... www.ti.com When the adapter is not detected, the ACDRV output is pulled to PVCC to turn off the ACFET, disconnecting the adapter from system. BATDRV stays at ACN – 6 V to connect the battery to system. At 700 ms after adapter is detected, the system begins to switch from the battery to the adapter. The ACN voltage must be 285 mV above BAT to enable the switching. The break-before-make logic turns off both ACFET and BATFET for 10µs before ACFET turns on. This isolates the battery from shoot-through current or any large discharging current. The BATDRV output is pulled up to ACN and the ACDRV pin is set to PVCC – 6 V by an internal regulator to turn on the p-channel ACFET, connecting the adapter to the system. When the adapter is removed, the system waits till ACN drops back to within 285 mV above BAT to switch from the adapter back to the battery. The break-before-make logic ensures a 10-µs dead time. The ACDRV output is pulled up to PVCC and the BATDRV pin is set to ACN – 6 V by an internal regulator to turn on the p-channel BATFET, connecting the battery to the system. Asymmetrical gate drive for the ACDRV and BATDRV drivers provides fast turn-off and slow turn-on of the ACFET and BATFET to help the break-before-make logic and to allow a soft-start at turn-on of either FET. The soft-start time can be further increased, by putting a capacitor from gate to source of the p-channel power MOSFETs. Battery Learn Cycles A battery Learn cycle can be implemented using the LEARN pin. A logic low on LEARN keeps the system power selector logic in its default states dependant on the adapter. If adapter is not detected, then; the ACFET is kept off, and the BATFET is kept on. If the adapter is detected, the BATFET is kept off, and the ACFET is kept on. When the LEARN pin is at logic high, the system power selector logic is overridden, keeping the ACFET off and the BATFET on when the adapter is present. This is used to allow the battery to discharge in order to calibrate the battery gas gauge over a complete discharge/charge cycle. Charge turns off when LEARN is high. The controller automatically exits the learn cycle when BAT < 2.9 V per cell. BATDRV turns off and ACDRV turns on. Automatic Internal Soft-Start Charger Current The charger automatically soft-starts the charger regulation current every time the charger is enabled to ensure there is no overshoot or stress on the output capacitors or the power converter. The soft-start consists of stepping-up the charge regulation current into 8 evenly-divided steps up to the programmed charge current. Each step lasts approximately 1 ms, for a typical rise time of 8 ms. No external components are needed for this function. Converter Operation The synchronous-buck PWM converter uses a fixed-frequency (300 kHz) voltage mode with a feed-forward control scheme. A Type-III compensation network allows the use of ceramic capacitors at the output of the converter. The compensation input stage is internally connected between the feedback output (FBO) and the error-amplifier input (EAI). The feedback compensation stage is connected between the error amplifier input (EAI) and error amplifier output (EAO). The LC output filter is selected for a nominal resonant frequency of 8 kHz–12.5 kHz. fo + The resonant frequency, fo, is given by: CO = C11 + C12 LO = L1 1 2p ǸLoC o where (from Figure 1 schematic) An internal sawtooth ramp is compared to the internal EAO error-control signal to vary the duty cycle of the converter. The ramp height is one-fifteenth of the input adapter voltage, making it always directly proportional to the input adapter voltage. This cancels out any loop-gain variation due to a change in input voltage, and simplifies the loop compensation. The ramp is offset by 200 mV in order to allow a 0% duty cycle when the EAO signal is below the ramp. The EAO signal is also allowed to exceed the sawtooth ramp signal in order to operate with a 100% duty-cycle PWM request. Internal gate-drive logic allows a 99.98% duty-cycle while ensuring that 20 Submit Documentation Feedback Copyright © 2008–2009, Texas Instruments Incorporated Product Folder Link(s) :bq24751B Not Recommended For New Designs bq24751B www.ti.com .......................................................................................................................................................... SLUS835A – JULY 2008 – REVISED MARCH 2009 the N-channel upper device always has enough voltage to stay fully on. If the BTST-to-PH voltage falls below 4 V for more than 3 cycles, the high-side N-channel power MOSFET is turned off and the low-side N-channel power MOSFET is turned on to pull the PH node down and recharge the BTST capacitor. Then the high-side driver returns to 100% duty-cycle operation until the (BTST-PH) voltage is detected falling low again due to leakage current discharging the BTST capacitor below 4 V, and the reset pulse is reissued. The 300-kHz fixed-frequency oscillator tightly controls the switching frequency under all conditions of input voltage, battery voltage, charge current, and temperature. This simplifies output-filter design, and keeps it out of the audible noise region. The charge-current sense resistor RSR should be designed with at least half or more of the total output capacitance placed before the sense resistor, contacting both sense resistor and the output inductor; and the other half, or remaining capacitance placed after the sense resistor. The output capacitance should be divided and placed on both sides of the charge-current sense resistor. A ratio of 50:50 percent gives the best performance; but the node in which the output inductor and sense resistor connect should have a minimum of 50% of the total capacitance. This capacitance provides sufficient filtering to remove the switching noise and give better current-sense accuracy. The Type-III compensation provides phase boost near the cross-over frequency, giving sufficient phase margin. Synchronous and Non-Synchronous Operation The charger operates in non-synchronous mode when the sensed charge current is below the ISYNSET internal setting value. Otherwise, the charger operates in synchronous mode. During synchronous mode, the low-side N-channel power MOSFET is on when the high-side N-channel power MOSFET is off. The internal gate-drive logic uses break-before-make switching to prevent shoot-through currents. During the 30-ns dead time where both FETs are off, the back-diode of the low-side power MOSFET conducts the inductor current. Having the low-side FET turn-on keeps the power dissipation low, and allows safe charging at high currents. During synchronous mode, the inductor current always flows, and the device operates in Continuous Conduction Mode (CCM), creating a fixed two-pole system. During non-synchronous operation, after the high-side n-channel power MOSFET turns off, and after the break-before-make dead-time, the low-side n-channel power MOSFET turns on for approximately 80 ns, then the low-side power MOSFET turns off and stays off until the beginning of the next cycle, when the high-side power MOSFET is turned on again. The 80-ns low-side MOSFET on-time is required to ensure that the bootstrap capacitor is always recharged and able to keep the high-side power MOSFET on during the next cycle. This is important for battery chargers, where unlike regular dc-dc converters, there is a battery load that maintains a voltage and can both source and sink current. The 80-ns low-side pulse pulls the PH node (connection between high and low-side MOSFET) down, allowing the bootstrap capacitor to recharge up to the REGN LDO value. After the 80 ns, the low-side MOSFET is kept off to prevent negative inductor current from flowing. The inductor current is blocked by the turned-off low-side MOSFET, and the inductor current becomes discontinuous. This mode is called Discontinuous Conduction Mode (DCM). During the DCM mode, the loop response automatically changes and has a single-pole system at which the pole is proportional to the load current, because the converter does not sink current, and only the load provides a current sink. This means that at low currents, the loop response is slower, because there is less sinking current available to discharge the output voltage. At low currents during non-synchronous operation, there may be a small amount of negative inductor current during the 80-ns recharge pulse. The charge should be low enough to be absorbed by the input capacitance. Whenever BTST – PH < 4 V, the 80-ns recharge pulse occurs on LODRV, the high-side MOSFET does not turn on, and the low-side MOSFET does not turn on (only 80-ns recharge pulse). In the bq24751B, VISYNSET=ISYN×RSR is internally set to 13mV as the charge-current threshold at which the charger changes from non-synchronous operation to synchronous operation. The low-side driver turns on for only 80 ns to charge the boost capacitor. This is important to prevent negative inductor current, which may cause a boost effect in which the input voltage increases as power is transferred from the battery to the input capacitors. This boost effect can lead to an overvoltage on the PVCC node and potentially damage the system. The inductor ripple current is given by Submit Documentation Feedback Copyright © 2008–2009, Texas Instruments Incorporated Product Folder Link(s) :bq24751B 21 Not Recommended For New Designs bq24751B SLUS835A – JULY 2008 – REVISED MARCH 2009 .......................................................................................................................................................... www.ti.com IRIPPLE_MAX 2 £ ISYN £ IRIPPLE_ MAX and (VIN - VBAT )´ IRIPPLE = VBAT 1 ´ VIN fs L VIN ´ (1- D )´ D ´ = 1 fs L (4) where VIN = adapter voltage VBAT = BAT voltage fS = switching frequency L = output inductor D = VBAT/VIN, duty-cycle IRIPPLE_MAX happens when the duty cycle (D) value is close to 0.5 at given VIN,fS, and L. The ISYNSET comparator, or charge undercurrent comparator, compares the voltage between SRP-SRN and the internal threshold. The threshold is set to 13 mV on the falling edge with an 8-mV hysteresis on the rising edge with a 10% variation. High Accuracy IADAPT Using Current Sense Amplifier (CSA) An industry-standard, high-accuracy current sense amplifier (CSA) is used by the host or some discrete logic to monitor the input current through the analog voltage output of the IADAPT pin. The CSA amplifies the sensed input voltage of ACP – ACN by 20x through the IADAPT pin. The IADAPT output is a voltage source 20 times the input differential voltage. When PVCC is above 5 V and ACDET is above 0.6 V, IADAPT no longer stays at ground, but becomes active. If the user wants to lower the voltage, they can use a resistor divider from IOUT to AGND, and still achieve accuracy over temperature as the resistors can be matched according to their thermal coefficients. A 200-pF capacitor connected on the output is recommended for decoupling high-frequency noise. An additional RC filter is optional, after the 200-pF capacitor, if additional filtering is desired. Note that adding filtering also adds additional response delay. Input Overvoltage Protection (ACOV) ACOV provides protection to prevent system damage due to high input voltage. Once the adapter voltage is above the programmable OVPSET voltage (3.1 V), charge is disabled, the adapter is disconnected from the system by turning off ACDRV, and the battery is connected to the system by turning on BATDRV. ACOV is not latched—normal operation resumes when the OVPSET voltage returns below 3.1 V. Input Undervoltage Lockout (UVLO) The system must have 5 V minimum of PVCC voltage for proper operation. This PVCC voltage can come from either the input adapter or the battery, using a diode-OR input. When the PVCC voltage is below 5 V, the bias circuits REGN, VREF, and the gate drive bias to ACFET and BATFET stay inactive, even with ACDET above 0.6 V. AC Lowvoltage (ACLOWV) ACLOWV clears the break-before-make protection latch when ACP < 3V in additon to UVLO clearing this latch when PVCC < UVLO. It ensures the BATDRV is off when ACP < 3V, and thus this function allows the ACDRV to turn on the ACFET again when ACP < 3V or PVCC <UVLO. 22 Submit Documentation Feedback Copyright © 2008–2009, Texas Instruments Incorporated Product Folder Link(s) :bq24751B Not Recommended For New Designs bq24751B www.ti.com .......................................................................................................................................................... SLUS835A – JULY 2008 – REVISED MARCH 2009 Battery Overvoltage Protection The converter stops switching when BAT voltage goes above 104% of the regulation voltage. The converter will not allow the high-side FET to turn on until the BAT voltage goes below 102% of the regulation voltage. This allows one-cycle response to an overvoltage condition, such as when the load is removed or the battery is disconnected. A 10-mA current sink from BAT to PGND is on only during charge, and allows discharging the stored output-inductor energy into the output capacitors. Battery Shorted (Battery Undervoltage) Protection The bq24751B has a BAT SHORT comparator monitoring the output battery voltage (BAT). If the voltage falls below 2.9 V per cell (5.8 V for 2 cells, 8.7 V for 3 cells, 11.6 V for 4 cells), a battery-short status is detected. Below the BAT_SHORT threshold, the charger reduces the charge current to 1/8th of the programmed charging current (0.1×SRSET/VDAC)/8 = C/8 down to zero volts on BAT pin.. This lower current is used as a pre-charge current for over-discharged battery packs. Above the BAT_SHORT threshold (plus hysteresis, the charge current resumes at the programmed value (0.1×SRSET/VDAC). The BAT_SHORT comparator also serves as a depleted-battery alarm during a LEARN cycle. If the selector is in a LEARN cycle, and the battery voltage falls bellow the BAT_SHORT threshold, the selector disconnects the battery from the system and connects the adapter to the system in order to protect the battery pack. If battery voltage increases, and LEARN is still logic high, then the selector disconnects the adapter from the system and reconnects the battery to the system. Charge Overcurrent Protection The charger has a secondary overcurrent protection feature. It monitors the charge current, and prevents the current from exceeding 145% of regulated charge current. The high-side gate drive turns off when the overcurrent is detected, and automatically resumes when the current falls below the overcurrent threshold. Thermal Shutdown Protection The QFN package has low thermal impedance, which provides good thermal conduction from the silicon to the ambient, to keep junction temperatures low. As an added level of protection, the charger converter turns off and self-protects when the junction temperature exceeds the TSHUT threshold of 155°C. The charger stays off until the junction temperature falls below 135°C. Adapter Detected Status Register (ACGOOD Pin) One status output is available, and it requires an external pullup resistor to pull the pin to the system digital rail for a high level. ACGOOD goes low when ACDET is above 2.4 V and the 700-ms delay time is over. It indicates that the adapter voltage is high enough for normal operation. Input Overpower Protection (ACOP) The ACOC/ACOP circuit provides a reliable layer of safety protection that can complement other safety measures. ACOC/ACOP helps to protect from input current surge due to various conditions including: • Adapter insertion and system selector connecting adapter to system where system capacitors need to charge • Learn mode exit when adapter is reconnected to the system; system load overcurrent surge • System shorted to ground • Battery shorted to ground • Phase shorted to ground • High-side FET shorted from drain to source (SYSTEM shorted to PH) • BATFET shorted from drain to source (SYSTEM shorted to BAT) Several examples of the circuit protecting from these fault conditions are shown below. For designs using the selector functions, an input overcurrent (ACOC) and input overpower protection function (ACOP) is provided. The threshold is set by an external capacitor from the ACOP pin to AGND. After the adapter Submit Documentation Feedback Copyright © 2008–2009, Texas Instruments Incorporated Product Folder Link(s) :bq24751B 23 Not Recommended For New Designs bq24751B SLUS835A – JULY 2008 – REVISED MARCH 2009 .......................................................................................................................................................... www.ti.com is detected (ACDET pin > 2.4V), there is a 700-ms delay before ACGOOD is asserted low, and Q3 (BATFET) is turned-off. Then Q1/Q2 (ACFET) are turned on by the ACDRV pin. When Q1/Q2 (ACFET) are turned on, the ACFET allows operation in linear-regulation mode to limit the maximum input current, ACOC, to a safe level. The ACOC current limit is 1.5 times the programmed DPM input current limit set by the ratio of SRSET/VDAC. The maximum allowable current limit is 100 mV across ACP – ACN (10 A for a 10-mΩ sense resistor). The first 2 ms after the ACDRV signal begins to turn on, ACOC may limit the current; but the controller is not allowed to latch off in order to allow a reasonable time for the system voltage to rise. After 2 ms, ACOP is enabled. ACOP allows the ACFET to latch off before the ACFET can be damaged by excessive thermal dissipation. The controller only latches if the ACOP pin voltage exceeds 2 V with respect to AGND. In ACOP, a current source begins to charge the ACOP capacitor when the input current is being limited by ACOC. This current source is proportional to the voltage across the source-drain of the ACFET (VPVCC-ACP) by an 18-µA/V ratio. This dependency allows faster capacitor charging if the voltage is larger (more power dissipation). It allows the time to be programmed by the ACOP capacitor selected. If the controller is not limiting current, a fixed 5-µA sink current into the ACOP pin to discharge the ACOP capacitor. This charge and discharge effect depends on whether there is a current-limit condition, and has a memory effect that averages the power over time, protecting the system from potentially hazardous repetitive faults. Whenever the ACOP threshold is exceeded, the charge is disabled and the adapter is disconnected from the system to protect the ACFET and the whole system. If the ACFET is latched off, the BATFET is turned on to connect the battery to the system. The capacitor provides a predictable time to limit the power dissipation of the ACFET. Since the input current is constant at the ACOC current limit, the designer can calculate the power dissipation on the ACFET. The ACOC current Limit threshold is equal to Power = Id × Vsd = IACOC _ LIM × V(PVCC - ACP) The time it takes to charge to 2V can be calculated from C C ACOP × 2V × DVACOP Dt = ACOP = i ACOP 18mA/V × V(PVCC - ACP) . (5) An ACOP fault latch off can only be cleared by bringing the ACDET pin voltage below 2.4 V, then above 2.4 V (i.e. remove adapter and reinsert), or by reducing the PVCC voltage below the UVLO threshold and raising it. Conditions for ACOP Latch Off: 702ms after ACDET (adapter detected), and a. ACOP voltage > 2V. The ACOP pin charges the ceramic capacitor when in an ACOC current-limit condition. The ACOP pin discharges the capacitor when not in ACOC current-limit. b. ACOP protects from a single-pulse ACOC condition depending on duration and source-drain voltage of ACFET. Larger voltage across ACFET creates more power dissipation so latch-off protection occurs faster, by increasing the current source out of ACOP pin. c. Memory effect (capacitor charging and discharging) allows protection from repetitive ACOC conditions, depending on duration and frequency. (Figure 35) d. In short conditions when the system is shorted to ground (ACN < 2.4 V) after the initial 2-ms ACDET. 24 Submit Documentation Feedback Copyright © 2008–2009, Texas Instruments Incorporated Product Folder Link(s) :bq24751B Not Recommended For New Designs bq24751B www.ti.com .......................................................................................................................................................... SLUS835A – JULY 2008 – REVISED MARCH 2009 In all cases, after 700ms delay, have input overcurrent protection, ACOC, by linearly limiting input current. Threshold is equal to the lower of Idpm*1.5, or 10A. ACOC, No Latch-off ACOC, with ACOP Latch-off, 700ms delay after ACDET, before allow ACDRV to turn-on After Latch-Off, Latch can only clear by: Latch-off time accumulates only when in current limit regulation, ACOC. The time before latch-off is programmable with Cacop, and is inversely proportional to source-drain voltage of ACFET (power). Cacop charge/discharge per time also provides memory for power averaging over time. 1) bringing ACDET below 2.4V, then above 2. 4V; or 2) bringing PVCC below UVLO, then above UVLO. 700ms 2ms 8ms Allow Charge to Turn-on Vin Vadapter ACDET 0V ACGOOD BATDRV ACDRV Vadapter Vsystem Vbattery Ilim = 1.5xIdpm (100 mV max Across ACP_ACN) Input Current Allow Charge Charge Current V(ACOP) A. ACFET overpower protection; initial current limit allows safe soft-start without system voltage droop. Figure 34. ACOC Protection During Adapter Insertion Submit Documentation Feedback Copyright © 2008–2009, Texas Instruments Incorporated Product Folder Link(s) :bq24751B 25 Not Recommended For New Designs bq24751B SLUS835A – JULY 2008 – REVISED MARCH 2009 .......................................................................................................................................................... www.ti.com Ilim = 1.5xIdpm Iin ACOC_REG V(PVCC-ACP) LATCH-OFF Iacop_pin LATCH-OFF 2V Memory Effect Averages Power V(ACOP) ACDRV_ON ON OFF LATCH-OFF Figure 35. ACOC Protection and ACOP Latch Off with Memory Effect Example 26 Submit Documentation Feedback Copyright © 2008–2009, Texas Instruments Incorporated Product Folder Link(s) :bq24751B Not Recommended For New Designs bq24751B www.ti.com .......................................................................................................................................................... SLUS835A – JULY 2008 – REVISED MARCH 2009 ADAPTER+ ADAPTER- R10 2Ω C1 2.2 µF P Q1 (ACFET) SI4435 RAC 0.010 Ω P Q2 (ACFET) SI4435 C3 C2 0.1 µF ACSET 0.1 µF ACDRV_ON ACDRV ACOC ERROR AMPLIFIER & DRIVER Regulation IDPM Reference IDPM_PRG RatioLowest of metric 1.5xIDPM_PRG Program or 10A (100mV) (100mV_max) + ACOCREG = REGULATING ACP IIN Differential Amp CSA V(ACP-ACN) + - ACN IADAPT + PVCC C8 0.1 µF VDS Differential Amp V(PVCC-ACP) Isrc=K*V(PVCC-ACP) K=18 µA/V REF=3.3V ACOP Adaptor Over Power Comparator ACOPDET + - ENA_SRC ACOP Cacop 0.47 µF ENA_SNK 1µs Deglitch 5 µA + - Rising-edge Set & Reset inputs 2V ACDET PVCC_UVLO /ACDRV & / BATDRV breakbefore-make logic ACOPDETDG S Q R Q ACDET 700 ms Delay Turn -off ACDRV To clear latch fault , user must remove adapter and reinsert, or PVCC brought below then above input UVLO threshold Figure 36. ACOC / ACOP Circuit Functional Block Diagram Table 3. Component List for Typical System Circuit of Figure 1 PART DESIGNATOR QTY DESCRIPTION Q1, Q2, Q3 3 P-channel MOSFET, –30 V, –6 A, SO-8, Vishay-Siliconix, Si4435 Q4, Q5 2 N-channel MOSFET, 30 V, 12.5 A, SO-8, Fairchild, FDS6680A D1 1 Diode, Dual Schottky, 30 V, 200 mA, SOT23, Fairchild, BAT54C RAC, RSR 2 Sense Resistor, 10 mΩ, 1%, 1 W, 2010, Vishay-Dale, WSL2010R0100F L1 1 Inductor, 8.2 µH, 8.5 A, 24.8 mΩ, Vishay-Dale, IHLP5050CE-01 C1 1 Capacitor, Ceramic, 2.2 µF, 25 V, 20%, X5R, 1206, Panasonic, ECJ-3YB1E225M C6, C7, C11, C12 4 Capacitor, Ceramic, 10 µF, 35 V, 20%, X5R, 1206, Panasonic, ECJ-3YB1E106M C4, C10 2 Capacitor, Ceramic, 1 µF, 25 V, 10%, X7R, 2012, TDK, C2012X7R1E105K C2, C3, C8, C9, C13, C14, C15 7 Capacitor, Ceramic, 0.1 µF, 50 V, 10%, X7R, 0805, Kemet, C0805C104K5RACTU C5 1 Capacitor, Ceramic, 100 pF, 25 V, 10%, X7R, 0805, Kemet C16 1 Capacitor, Ceramic, 0.47 µF, 25 V, 10%, X7R, 0805, Kemet R1 1 Resistor, Chip, 432 kΩ, 1/16 W, 1%, 0402 R2 1 Resistor, Chip, 66.5 kΩ, 1/16 W, 1%, 0402 R3 1 Resistor, Chip, 422 kΩ, 1/16 W, 1%, 0402 R4 1 Resistor, Chip, 71 kΩ, 1/16 W, 5%, 0402 R10 1 Resistor, Chip, 2 Ω, 1 W, 5%, 2010 Submit Documentation Feedback Copyright © 2008–2009, Texas Instruments Incorporated Product Folder Link(s) :bq24751B 27 Not Recommended For New Designs bq24751B SLUS835A – JULY 2008 – REVISED MARCH 2009 .......................................................................................................................................................... www.ti.com APPLICATION INFORMATION Input Capacitance Calculation During the adapter hot plug-in, the ACDRV has not been enabled. The AC switch is off and the simplified equivalent circuit of the input is shown in Figure 37. IIN Ri Vi VIN Li Rc Ci Vc Figure 37. Simplified Equivalent Circuit During Adapter Insertion The voltage on the charger input side VIN is given by: V IN(t) + I IN(t) RC ) VCi(t) + Vie Ri t 2L i ƪ Ri * RC sin wt ) cos wt wL i ƫ (6) in which, R t = Ri + R C w = 1 - L i Ci Rt t 2Li VCi (t) = Vi - Vie æ Rt ö ç ÷ è 2Li ø 2 IIN (t) = Vi Ri t 2Li e sin w t wL i æ Rt ö s in w t + cosw t ÷ ç è 2w L i ø (7) The damping conditions is: Ri ) Rc u 2 28 Ǹ Li Ci (8) Submit Documentation Feedback Copyright © 2008–2009, Texas Instruments Incorporated Product Folder Link(s) :bq24751B Not Recommended For New Designs bq24751B www.ti.com .......................................................................................................................................................... SLUS835A – JULY 2008 – REVISED MARCH 2009 Figure 38 (a) demonstrates a higher Ci helps dampen the voltage spike. Figure 38 (b) demonstrates the effect of the input stray inductance Li upon the input voltage spike. Figure 38 (c) shows how increased resistance helps to suppress the input voltage spike. 35 35 Ci = 20 mF Ci = 40 mF 30 Input Capacitor Voltage - V Input Capacitor Voltage - V Li = 5 mH Ri = 0.21 W Li = 9.3 mH 30 25 20 15 10 5 0 Li = 12 mH Ri = 0.15 W Ci = 40 mF 25 20 15 10 5 0 0.5 1 1.5 2 2.5 3 3.5 Time - ms (a) Vc with various Ci values 4 4.5 0 5 0 0.5 1 1.5 2 2.5 3 3.5 Time - ms (b) Vc with various Li values 4 4.5 5 35 Ri = 0.15 W Li = 9.3 mH Ci = 40 mF Input Capacitor Voltage - V 30 Ri = 0.5 W 25 20 15 10 5 0 0 0.5 1 1.5 2 2.5 3 Time - ms 3.5 4 4.5 5 (c) Vc with various Ri values Figure 38. Parametric Study Of The Input Voltage As shown in Figure 38, minimizing the input stray inductance, increasing the input capacitance, and adding resistance (including using higher ESR capacitors) helps suppress the input voltage spike. However, a user often cannot control input stray inductance and increasing capacitance can increase costs. Therefore, the most efficient and cost-effective approach is to add an external resistor. Submit Documentation Feedback Copyright © 2008–2009, Texas Instruments Incorporated Product Folder Link(s) :bq24751B 29 Not Recommended For New Designs bq24751B SLUS835A – JULY 2008 – REVISED MARCH 2009 .......................................................................................................................................................... www.ti.com Figure 39 depicts the recommended input filter design. The measured input voltage and current waveforms are shown in Figure 40. The input voltage spike has been well damped by adding a 2Ω resistor, while keeping the capacitance low. VIN 2W (0.5 W, 1210 anti-surge) 2.2 mF (25 V, 1210) VPVCC Rext C1 0.1 mF (50 V, 0805, very close to PVCC) C2 Figure 39. Recommended Input Filter Design 30 Figure 40. Adapter DC Side Hot Plug-in Test Waveforms Submit Documentation Feedback Copyright © 2008–2009, Texas Instruments Incorporated Product Folder Link(s) :bq24751B Not Recommended For New Designs bq24751B www.ti.com .......................................................................................................................................................... SLUS835A – JULY 2008 – REVISED MARCH 2009 PCB Layout Design Guideline 1. It is critical that the exposed power pad on the backside of the IC package be soldered to the PCB ground. Ensure that there are sufficient thermal vias directly under the IC, connecting to the ground plane on the other layers. 2. The control stage and the power stage should be routed separately. At each layer, the signal ground and the power ground are connected only at the power pad. 3. The AC current-sense resistor must be connected to ACP (pin 3) and ACN (pin 2) with a Kelvin contact. The area of this loop must be minimized. An additional 0.1µF decoupling capacitor for ACN is required to futther reduce noise. The decoupling capacitors for these pins should be placed as close to the IC as possible. 4. The charge-current sense resistor must be connected to SRP (pin 19), SRN (pin 18) with a Kelvin contact. The area of this loop must be minimized. An additional 0.1µF decoupling capacitor for SRN is required to futther reduce noise. The decoupling capacitors for these pins should be placed as close to the IC as possible. 5. Decoupling capacitors for PVCC (pin 28), VREF (pin 10), REGN (pin 24) should be placed underneath the IC (on the bottom layer) with the interconnections to the IC as short as possible. 6. Decoupling capacitors for BAT (pin 17), IADAPT (pin 15) must be placed close to the corresponding IC pins with the interconnections to the IC as short as possible. 7. Decoupling capacitor CX for the charger input must be placed close to the Q4 drain and Q5 source. Figure 41 shows the recommended component placement with trace and via locations. For the QFN information, please refer to the following links: SCBA017 and SLUA271 (a) Top Layer (b) Bottom Layer Figure 41. Layout Example Submit Documentation Feedback Copyright © 2008–2009, Texas Instruments Incorporated Product Folder Link(s) :bq24751B 31 PACKAGE OPTION ADDENDUM www.ti.com 28-Mar-2011 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Drawing Pins Package Qty Eco Plan (2) Lead/ Ball Finish MSL Peak Temp (3) BQ24751BRHDR NRND VQFN RHD 28 3000 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR BQ24751BRHDRG4 NRND VQFN RHD 28 3000 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR BQ24751BRHDT NRND VQFN RHD 28 250 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR BQ24751BRHDTG4 NRND VQFN RHD 28 250 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR Samples (Requires Login) (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material) (3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. 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Addendum-Page 1 PACKAGE MATERIALS INFORMATION www.ti.com 14-Jul-2012 TAPE AND REEL INFORMATION *All dimensions are nominal Device Package Package Pins Type Drawing SPQ Reel Reel A0 Diameter Width (mm) (mm) W1 (mm) B0 (mm) K0 (mm) P1 (mm) W Pin1 (mm) Quadrant BQ24751BRHDR VQFN RHD 28 3000 330.0 12.4 5.3 5.3 1.5 8.0 12.0 Q2 BQ24751BRHDT VQFN RHD 28 250 180.0 12.4 5.3 5.3 1.5 8.0 12.0 Q2 Pack Materials-Page 1 PACKAGE MATERIALS INFORMATION www.ti.com 14-Jul-2012 *All dimensions are nominal Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm) BQ24751BRHDR VQFN RHD 28 3000 367.0 367.0 35.0 BQ24751BRHDT VQFN RHD 28 250 210.0 185.0 35.0 Pack Materials-Page 2 IMPORTANT NOTICE Texas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, enhancements, improvements and other changes to its semiconductor products and services per JESD46C and to discontinue any product or service per JESD48B. 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