ON NCP3020A Synchronous pwm controller Datasheet

NCP3020A, NCP3020B
Product Preview
Synchronous PWM Controller
The NCP3020 is a PWM device designed to operate from a wide
input range and is capable of producing an output voltage as low as
0.6 V. The NCP3020A provides integrated gate drivers and an
internally set 300 kHz oscillator (600 kHz for the NCP3020B). The
NCP3020 has an externally compensated transconductance error
amplifier with an internally fixed soft−start. Protection features
include lossless current limit and Short Circuit Protection, Output
Overvoltage protection, Output Undervoltage protection, and Input
Undervoltage Lockout. The NCP3020 is currently available in a
SOIC−8 package.
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8
1
SOIC−8 NB
CASE 751
Features
•
•
•
•
•
•
•
•
•
•
Input Voltage Range from 4.5 V to 28 V
Switching and Synchronous Rectifier Drives with >1.0 A Capability
300 kHz Operation (NCP3020B – 600 kHz)
0.6 ± 1.0% Reference Voltage
Internally Programmed 6.8 ms Soft−Start (NCP3020B – 3.4 ms)
Current Limit and Short Circuit Protection
Transconductance Amplifier with External Compensation
Input Undervoltage Lockout
Output Overvoltage and UndervoltageDetection
This is a Pb−Free Device
MARKING DIAGRAM
8
1
XXXXX
A
L
Y
W
G
4.5−28 V
VCC
BST
L0
VSW
LSDR
Vout
VSW
GND
ORDERING INFORMATION
RFB1
Q2
LSDR
FB
CC2
HSDR
FB
GND
Q1
HSDR
COMP
BST
VCC
COMP
CBST
CC1
= Specific Device Code
= Assembly Location
= Wafer Lot
= Year
= Work Week
= Pb−Free Package
PIN CONNECTIONS
CIN
RC
XXXXX
ALYW
G
RFB2
RISET
Figure 1. Typical Application Circuit
C0
Device
Package
Shipping†
NCP3020ADR2G
SOIC−8 2500 / Tape & Reel
(Pb−Free)
NCP3020BDR2G
SOIC−8 2500 / Tape & Reel
(Pb−Free)
†For information on tape and reel specifications,
including part orientation and tape sizes, please
refer to our Tape and Reel Packaging Specifications
Brochure, BRD8011/D.
This document contains information on a product under development. ON Semiconductor
reserves the right to change or discontinue this product without notice.
© Semiconductor Components Industries, LLC, 2009
November, 2009 − Rev. P1
1
Publication Order Number:
NCP3020/D
NCP3020A, NCP3020B
VCC
INTERNAL BIAS
BST
POR/STARTUP
VC
THERMAL SD
BOOST
CLAMP
CLK/
DMAX/
SOFT
START
OSCILLATOR
GATE
DRIVE
LOGIC
RAMP
1.5 V
LEVEL
SHIFT
HSDR
VCC
VSW
CURRENT
LIMIT
SAMPLE &
HOLD
ISET
+
−
COMP
REF
OTA
VC
LSDR
PWM
COMP
FB
+
−
+
−
BST_CHRG
OOV
OUV
GND
Figure 2. NCP3020 Block Diagram
PIN FUNCTION DESCRIPTION
Pin
Pin Name
Description
1
VCC
The VCC pin is the main voltage supply input. It is also used in conjunction with the VSW pin to sense current
in the high side MOSFET.
2
COMP
The COMP pin connects to the output of the Operational Transconductance Amplifier (OTA) and the positive
terminal of the PWM comparator. This pin is used in conjunction with the FB pin to compensate the voltage
mode control feedback loop.
3
FB
4
GND
Ground Pin
5
LSDR
The LSDR pin is connected to the output of the low side driver which connects to the gate of the low side
N−FET. It is also used to set the threshold of the current limit circuit (ISET) by connecting a resistor from LSDR
to GND.
6
VSW
The VSW pin is the return path for the high side driver. It is also used in conjunction with the VCC pin to sense
current in the high side MOSFET.
7
HSDR
The HSDR pin is connected to the output of the high side driver which connects to the gate of the high side
N−FET.
8
BST
The BST pin is the supply rail for the gate drivers. A capacitor must be connected between this pin and the
VSW pin.
The FB pin is connected to the inverting input of the OTA. This pin is used in conjunction with the COMP pin to
compensate the voltage mode control feedback loop.
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NCP3020A, NCP3020B
ABSOLUTE MAXIMUM RATINGS (measured vs. GND pin 8, unless otherwise noted)
Rating
Symbol
VMAX
VMIN
Unit
BST
45
−0.3
V
BST−VSW
13.2
−0.3
V
COMP
5.5
−0.3
V
FB
5.5
−0.3
V
High−Side Driver Output
HSDR
40
−0.3
V
Low−Side Driver Output
High Side Drive Boost Pin
Boost to VSW differential voltage
COMP
Feedback
LSDR
13.2
−0.3
V
Main Supply Voltage Input
VCC
40
−0.3
V
Switch Node Voltage
VSW
40
−0.6
V
Maximum Average Current
VCC, BST, HSDRV, LSDRV, VSW, GND
Imax
Operating Junction Temperature Range (Note 1)
TJ
−40 to +140
°C
TJ(MAX)
+150
°C
Storage Temperature Range
Tstg
−55 to +150
°C
Thermal Characteristics (Note 2)
TSSOP−14 Plastic Package
Maximum Power Dissipation @ TA = 25°C
Thermal Resistance Junction−to−Air
PD
RqJA
0.5
190
W
°C/W
RF
260 Peak
°C
Moisture Sensitivity Level (Note 4)
MSL
3
−
ESD Withstand Voltage (Note 5)
Human Body Model
Machine Model
Vesd
2.0
200
kV
V
100
mA
Maximum Junction Temperature
Lead Temperature Soldering (10 sec): Reflow (SMD styles only) Pb−Free
(Note 3)
Latch−up Current (TA = 85°C) (Note 6)
Lu
130
mA
Stresses exceeding Maximum Ratings may damage the device. Maximum Ratings are stress ratings only. Functional operation above the
Recommended Operating Conditions is not implied. Extended exposure to stresses above the Recommended Operating Conditions may affect
device reliability.
1. The maximum package power dissipation limit must not be exceeded.
PD +
T J(max) * T A
R qJA
2.
3.
4.
5.
When mounted on minimum recommended FR−4 or G−10 board
60−180 seconds minimum above 237°C.
Moisture Sensitivity Level (MSL): 3 per IPC/JEDEC standard: J−STD−020A.
This device series contains ESD protection and exceeds the following tests:
Human Body Model (HBM) per JEDEC standard: JESD22−A114.
Machine Model (MM) per JEDEC standard: JESD22−A115.
6. Latch−up Current Maximum Rating: per JEDEC standard: JESD78
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NCP3020A, NCP3020B
ELECTRICAL CHARACTERISTICS (−40°C < TJ < +125°C, VCC = 12 V, for min/max values unless otherwise noted)
Characteristic
Max
Unit
28
V
5.1
8.0
mA
−
6.2
11
mA
VFB = 0.55 V, Switching, VCC = 4.5 V
−
6.8
TBD
mA
VFB = 0.55 V, Switching, VCC = 28 V
−
8.3
TBD
mA
UVLO Rising Threshold
VCC Rising Edge
4.0
4.3
4.7
V
UVLO Falling Threshold
VCC Falling Edge
3.5
3.9
4.3
V
Input Voltage Range
Conditions
Min
−
4.5
VFB = 0.55 V, Switching, VCC = 4.5 V
−
VFB = 0.55 V, Switching, VCC = 28 V
Typ
SUPPLY CURRENT
VCC Supply Current
VCC Supply Current
NCP3020A
NCP3020B
UNDER VOLTAGE LOCKOUT
OSCILLATOR
Oscillator Frequency
Oscillator Frequency
NCP3020A
NCP3020B
Ramp−Amplitude Voltage
TJ = +25°C, 4.5 V v VCC v 28 V
250
300
350
kHz
TJ = −40°C to +125°C, 4.5 V v VCC v 28 V
240
300
360
kHz
TJ = +25°C, 4.5 V v VCC v 28 V
TBD
600
TBD
kHz
TJ = −40°C to +125°C, 4.5 V v VCC v 28 V
TBD
600
TBD
kHz
Vpeak − Valley
−
1.5
−
V
0.46
0.71
0.85
V
Ramp Valley Voltage
PWM
(Note 7)
Minimum Duty Cycle
Maximum Duty Cycle
Soft Start Ramp Time
NCP3020A
NCP3020B
VFB = VCOMP
−
7.0
−
%
80
84
−
%
−
−
6.8
3.4
−
−
ms
ERROR AMPLIFIER (GM)
0.9
1.3
1.7
mS
Open Loop dc Gain
(Notes 7 and 9)
−
70
−
dB
Output Source Current
VFB = 545 mV
45
70
100
mA
Output Sink Current
VFB = 655 mV
45
70
100
mA
−
0.5
500
nA
TJ = 25 C
4.5 V < VIN < 28 V, −40°C < TJ < +125°C
0.594
0.588
0.6
0.6
0.606
0.612
V
V
COMP High Voltage
VFB = 0 V
4.0
4.4
5.0
V
COMP Low Voltage
VFB = 2 V
−
65
200
mV
Feedback OOV Threshold
0.68
0.75
0.82
V
Feedback OUV Threshold
0.42
0.45
0.48
V
TBD
12.5
TBD
mA
TBD
298
TBD
mV
Transconductance
FB Input Bias Current
Feedback Voltage
OUTPUT VOLTAGE FAULTS
OVERCURRENT
ISET Source Current
Current Limit Set Voltage (Note 8)
RSET = 22.5 kW
7. Guaranteed by design.
8. The voltage sensed across the high side MOSFET during conduction.
9. This assumes 100 pF capacitance to ground on the COMP Pin and a typical internal Ro of > 10 MW.
10. This is not a protection feature.
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NCP3020A, NCP3020B
ELECTRICAL CHARACTERISTICS (−40°C < TJ < +125°C, VCC = 12 V, for min/max values unless otherwise noted)
Characteristic
Conditions
Min
Typ
Max
Unit
VCC = 8 V, VBST = 7.5 V
VSW = GND
100 mA out of HSDR pin
5.0
10.5
20
W
−
0.7
−
A
VCC = 8 V, VBST = 7.5 V
VSW = GND
100 mA into HSDR pin
2.5
5.0
10.5
W
−
1.5
−
A
VCC = 8 V, VBST = 7.5 V
VSW = GND
100 mA out of LSDR pin
5.0
8.9
15
W
−
0.85
−
A
VCC = 8 V, VBST = 7.5 V
VSW = GND
100 mA into LSDR pin
1.0
2.8
6.0
W
−
2.7
−
A
HSDRV falling to LSDRV Rising
Delay
VCC = 8 V, VBST = 7.5 V
60
76
100
ns
LSDRV Falling to HSDRV Rising
Delay
VCC = 8 V, VBST = 7.5 V
60
84
100
ns
VIN = 12 V, VSW = GND, VCOMP = 1.3 V
6.0
7.5
9.6
V
Thermal Shutdown
(Notes 7 and 10)
−
175
−
°C
Hysteresis
(Notes 7 and 10)
−
20
−
°C
GATE DRIVERS AND BOOST CLAMP
HSDRV Pullup Resistance
HSDRV Source
HSDRV Pulldown Resistance
HSDRV Sink
LSDRV Pullup Resistance
LSDRV Source
LSDRV Pulldown Resistance
LSDRV Sink
Boost Clamp Voltage
THERMAL SHUTDOWN
7. Guaranteed by design.
8. The voltage sensed across the high side MOSFET during conduction.
9. This assumes 100 pF capacitance to ground on the COMP Pin and a typical internal Ro of > 10 MW.
10. This is not a protection feature.
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NCP3020A, NCP3020B
TYPICAL PERFORMANCE CHARACTERISTICS
3.28
95
9V
3.275
85
12 V
18 V
3.27
15 V
80
75
9V
3.265
3.26
70
65
60
15 V
18 V
Vout (V)
EFFICIENCY (%)
90
3.255
Typical Application Circuit
Figure 37
0
2
4
8
6
12 V
3.25
10
Typical Application Circuit
Figure 37
0
2
4
Iout (A)
8
6
10
Iout (A)
Figure 3. Efficiency vs. Output Current and Input
Voltage
Figure 4. Load Regulation vs. Input Voltage
Input = 9 V, Output = 3.3 V, Load = 10 A
C4 (Green) = VIN, C2 (Red) = VOUT
C1 (Yellow) = VSW, C3 (Blue) = HSDR
Input = 18 V, Output = 3.3 V, Load = 10 A
C4 (Green) = VIN, C2 (Red) = VOUT
C1 (Yellow) = VSW, C3 (Blue) = HSDR
Figure 5. Switching Waveforms (VIN = 9 V)
Figure 6. Switching Waveforms (VIN = 18 V)
606
340
NCP3020A
330
604
320
VCC = 12 V, 28 V
fSW (kHz)
VFB (mV)
602
600
598
VCC = 5 V
VCC = 12 V, 28 V
300
VCC = 5 V
290
280
596
594
−40 −25 −10
310
270
5
20
35
50
65
80
260
−40 −25 −10
95 110 125
TEMPERATURE (°C)
5
20
35
50
65
80
95 110 125
TEMPERATURE (°C)
Figure 7. Feedback Reference Voltage vs. Input
Voltage and Temperature
Figure 8. Switching Frequency vs. Input Voltage
and Temperature
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NCP3020A, NCP3020B
TYPICAL PERFORMANCE CHARACTERISTICS
1.50
660
1.45
NCP3020B
640
1.40
VCC = 12 V, 28 V
600
gm (ms)
fSW (kHz)
VCC = 5 V
1.35
620
VCC = 5 V
580
1.30
VCC = 12 V, 28 V
1.25
1.20
1.15
1.10
560
1.05
540
−40 −25 −10
5
20
35
50
65
80
1.00
−40 −25 −10
95 110 125
5
TEMPERATURE (°C)
20
35
50
65
80
95 110 125
TEMPERATURE (°C)
Figure 9. Switching Frequency vs. Input Voltage
and Temperature
Figure 10. Transconductance vs. Input Voltage
and Temperature
800
4.4
760
4.3
720
UVLO Rising
OOV (mV)
4.2
UVLO (V)
OOV, VCC = 5 − 28 V
680
4.1
4.0
640
600
560
520
480
3.9
UVLO Falling
3.8
−40 −25 −10
5
20
35
50
OOV, VCC = 5 − 28 V
440
65
80
400
−40 −25 −10
95 110 125
TEMPERATURE (°C)
5
20
35
50
65
80
95 110 125
TEMPERATURE (°C)
Figure 11. Input Undervoltage Lockout vs.
Temperature
Figure 12. Output Overvoltage Lockout vs. Input
Voltage and Temperature
7.5
ICC, SWITCHING (mA)
7.0
VCC = 28 V
6.5
6.0
VCC = 12 V
5.5
TBD
VCC = 5 V
5.0
4.5
4.0
−40 −25 −10
NCP3020A
5
20
35
50
65
80
95 110 125
TEMPERATURE (°C)
Figure 13. Supply Current vs. Input Voltage and
Temperature
Figure 14. Supply Current vs. Input Voltage and
Temperature
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NCP3020A, NCP3020B
TYPICAL PERFORMANCE CHARACTERISTICS
7.5
750
700
650
NCP3020A tSoft−Start (ms)
900
850
800
VALLEY VOLTAGE (mV)
8.0
VCC = 5 − 28 V
600
550
500
450
400
−40 −25 −10
5
20
35
50
65
80
6.0
5.5
NCP3020A
VCC = 5 V
7.0
5.0
VCC = 12 V, 28 V
6.5
4.5
6.0
4.0
NCP3020B
VCC = 5 V
5.5
5.0
95 110 125
−40 −25 −10
TEMPERATURE (°C)
3.5
VCC = 12 V, 28 V
5
20
35
50
65
80
3.0
95 110 125
TEMPERATURE (°C)
Figure 15. Ramp Valley Voltage vs. Input Voltage
and Temperature
Figure 16. Soft−Start Time vs. Input Voltage and
Temperature
14
ISET (mA)
13.8
13.6
VCC = 12 V, 28 V
13.4
VCC = 5 V
13.2
13
−40 −25 −10
5
20
35
50
65
80
Input = 12 V, Output = 3.3 V, Load = 5 A
C1 (Yellow) = VIN, C4 (Green) = VOUT
C2 (Red) = HSDR, C3 (Blue) = LSDR
95 110 125
TEMPERATURE (°C)
Figure 18. Soft−Start Waveforms
Figure 17. Current Limit Set Current vs.
Temperature
Input = 12 V
C1 (Yellow) = FB, C3 (Blue) = LSDR
C2 (Red) = HSDR, C4 (Green) = VIN
Input = 12 V, Output = 3.3 V, Load = 5 A
C1 (Yellow) = VIN, C4 (Green) = VOUT
C2 (Red) = HSDR, C3 (Blue) = LSDR
Figure 19. Shutdown Waveforms
Figure 20. Startup into a Current Limit
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NCP3020B tSoft−Start (ms)
1000
950
NCP3020A, NCP3020B
DETAILED DESCRIPTION
OVERVIEW
the high−side and low−side MOSFET gate drives to prevent
cross conduction of the power MOSFET’s.
The NCP3020A/B operates as a 300/600 kHz, voltage
mode, pulse width modulated, (PWM) synchronous buck
converter. It drives high−side and low−side N−channel power
MOSFETs. The NCP3020 incorporates an internal boost
circuit consisting of a boost clamp and boost diode to provide
supply voltage for the high side MOSFET gate driver. The
NCP3020 also integrates several protection features including
input undervoltage lockout (UVLO), output undervoltage
(OUV), output overvoltage (OOV), adjustable high−side
current limit (ISET and ILIM), and thermal shutdown (TSD).
The operational transconductance amplifier (OTA)
provides a high gain error signal from Vout which is
compared to the internal 1.5 V pk-pk ramp signal to set the
duty cycle converter using the PWM comparator. The high
side switch is turned on by the positive edge of the clock
cycle going into the PWM comparator and flip flop
following a non-overlap time. The high side switch is turned
off when the PWM comparator output is tripped by the
modulator ramp signal reaching a threshold level
established by the error amplifier. The gate driver stage
incorporates symmetrical fixed non− overlap time between
POR and UVLO
The device contains an internal Power On Reset (POR) and
input Undervoltage Lockout (UVLO) that inhibits the internal
logic and the output stage from operating until VCC reaches its
respective predefined voltage levels (4.3 V typical).
Startup and Shutdown
Once enable is asserted the device begins its startup
process. Closed−loop soft−start begins after a 400 ms delay
wherein the boost capacitor is charged, and the current limit
threshold is set. During the 400 ms delay the OTA output is
set to 700 mV which is just below the low point of the
internal ramp. This is done to reduce delays and to ensure a
consistent pre−soft−start condition. The device increases the
internal reference from 0 V to 0.6 V in 32 discrete steps
while maintaining closed loop regulation at each step. Some
overshoot may be evident at the start of each step depending
on the voltage loop phase margin and bandwidth. See
Figure 21. The total soft−start time is 6.8 ms for the
NCP3020A and 3.4 ms for the NCP3020B.
0.6 V Output Voltage
18.75 mV Steps
32 Voltage Steps
Internal Reference Voltage
Internal Ramp
OTA Output
0 .7V
0V
Output Voltage
Internal Reference Voltage
0 1 2 3 4 5 6 7 8 9 1 1 1 1 1 1 1 1 1 1 2 2 2 2 2 2 2 2 2 2 3 3 3
0 1 2 3 4 5 6 7 8 9 0 1 2 3 4 5 6 7 8 9 0 1 2
Switch
Node
OTA Output
Figure 21. Soft−Start Details
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NCP3020A, NCP3020B
OOV and OUV
the output is considered “undervoltage” and the device will
initiate a restart. When the feedback pin voltage rises
between the reference voltages of comparator 1 and
comparator 2 (0.45 < VFB < 0.75), then the output voltage
is considered “Power Good.” Finally, if the feedback voltage
is greater than comparator 1 (VFB > 0.75 V), the output
voltage is considered “overvoltage,” and the device will
latch off. To clear a latch fault, input voltage must be
recycled. Graphical representation of the OOV and OUV is
shown in Figures 24 and 25.
The output voltage of the buck converter is monitored at
the Feedback pin of the output power stage. Two
comparators are placed on the feedback node of the OTA to
monitor the operating window of the feedback voltage as
shown in Figures 22 and 23. All comparator outputs are
ignored during the soft−start sequence as soft−start is
regulated by the OTA and false trips would be generated.
After the soft−start period has ended, if the feedback is
below the reference voltage of comparator 2 (VFB < 0.45 V),
Soft Start Complete
Vref*125%
Comparator 1
Restart
LOGIC
FB
Latch off
Vref*75%
Comparator 2
Vref = 0.6 V
Figure 22. OOV and OUV Circuit Diagram
OOVP & Power Good = 0
Hysteresis = 5 mV
Voov = Vref * 125%
Power Not good High
Power Good = 1
Vref = 0.6 V
Power Good = 1
Hysteresis = 5 mV
OUVP & Power Good = 0
Figure 23. OOV and OUV Window Diagram
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Power Not Good Low
Voov = Vref * 75%
NCP3020A, NCP3020B
0.75 V (vref *125%)
0.6 V (vref *100%)
0.45 V (vref *75%)
FB Voltage
Latch off
Reinitiate Softstart
Softstart Complete
Figure 24. Powerup Sequence and Overvoltage Latch
0.75 V (vref *125%)
0.6 V (vref *100%)
0.45 V (vref * 75%)
FB Voltage
Latch off
Reinitiate Softstart
Softstart Complete
Figure 25. Powerup Sequence and Undervoltage Soft−Start
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NCP3020A, NCP3020B
CURRENT LIMIT AND CURRENT LIMIT SET
ILimit block consists of a voltage comparator circuit which
compares the differential voltage across the VCC Pin and the
VSW Pin with a resistor settable voltage reference. The sense
portion of the circuit is only active while the HS MOSFET
is turned ON.
Overview
The device uses the voltage drop across the High Side
MOSFET during the on time to sense inductor current. The
VIN
VCC
Ilim Out
VSense
HSDR
Itrip Ref
VSW
Switch
Cap
CONTROL
Iset
12.5uA
LSDR
6
Vset
DAC /
COUNTER
RSet
Itrip Ref−63 Steps, 3.255 mV/step
Figure 26. Iset / ILimit Block Diagram
Current Limit Set
a maximum sense voltage of 176 mV. During the Iset period
prior to Soft−Start, the DAC counter increments the
reference on the ISET comparator until it crosses the VSET
voltage and holds the DAC reference output to that count
value. This voltage is translated to the ILimit comparator
during the ISense portion of the switching cycle through the
switch cap circuit. See Figure 26.
The ILimit comparator reference is set during the startup
sequence by forcing a 12.5 mA current through the low side
gate drive resistor. The gate drive output will rise to a voltage
level shown in the equation below:
V set + I set * R set
(eq. 1)
Where ISET is 12.5 mA and RSET is the gate to source resistor
on the low side MOSFET.
This resistor is normally installed to prevent MOSFET
leakage from causing unwanted turn on of the low side
MOSFET. In this case, the resistor is also used to set the
ILimit trip level reference through the ILimit DAC. The Iset
process takes approximately 350 ms to complete prior to
Soft−Start stepping. The scaled voltage level across the ISET
resistor is converted to a 6 bit digital value and stored as the
trip value. The binary ILimit value is scaled and converted to
the analog ILimit reference voltage through a DAC counter.
The DAC has 54 steps in 3.255 mV increments equating to
Current Sense Cycle
Figure 27 shows how the current is sampled as it relates
to the switching cycle. Current level 1 in Figure 27
represents a condition that will not cause a fault. Current
level 2 represents a condition that will cause a fault. The
sense circuit is allowed to operate below the 3/4 point of a
given switching cycle. A given switching cycle’s 3/4 Ton
time is defined by the prior cycle’s Ton and is quantized in
10 ns steps. A fault occurs if the sensed MOSFET voltage
exceeds the DAC reference within the 3/4 time window of
the switching cycle.
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NCP3020A, NCP3020B
Trip:
Vsense > Itrip Ref at 3/4 Point
No Trip:
Vsense < Itrip Ref at 3/4 Point
Itrip Ref
Vsense
¾
¾
Current Level 1
Ton−1
Ton−2
Current Level 2
3/4 Point Determined by
Prior Cycle
1/4
1/2
1/4
1/2
3/4
3/4
Ton−1
Ton
Each switching cycle’s Ton is counted in 10 nS time steps. The 3/4 sample time
value is held and used for the following cycle’s limit sample time
Figure 27. ILimit Trip Point Description
Soft−Start Current limit
Boost Clamp Functionality
During soft−start the ISET value is doubled to allow for
inrush current to charge the output capacitance. The DAC
reference is set back to its normal value after soft−start has
completed.
The boost circuit requires an external capacitor connected
between the BST and VSW pins to store charge for supplying
the high and low−side gate driver voltage. This clamp circuit
limits the driver voltage to typically 7.5 V when VIN > 9 V,
otherwise this internal regulator is in dropout and typically
VIN − 1.25 V.
The boost circuit regulates the gate driver output voltage
and acts as a switching diode. A simplified diagram of the
boost circuit is shown in Figure 28. While the switch node
is grounded, the sampling circuit samples the voltage at the
boost pin, and regulates the boost capacitor voltage. The
sampling circuit stores the boost voltage while the VSW is
high and the linear regulator output transistor is reversed
biased.
VSW Ringing
The ILimit block can lose accuracy if there is excessive
VSW voltage ringing that extends beyond the 1/2 point of the
high−side transistor on−time. Proper snubber design and
keeping the ratio of ripple current and load current in the
10−30% range can help alleviate this as well.
Current Limit
A current limit trip results in completion of one switching
cycle and subsequently half of another cycle Ton to account
for negative inductor current that might have caused
negative potentials on the output. Subsequently the power
MOSFETs are both turned off and a 4 soft−start time period
wait passes before another soft−start cycle is attempted.
VIN
8.9V
Iave vs Trip Point
The average load trip current versus RSET value is shown
the equation below:
I AveTRIP +
I set
R set
R DS(on)
*
ƪ
1 V IN * V OUT
4
L
Switch
Sampling
Circuit
ƫ
V OUT
1
V IN
F SW
BST
VSW
LSDR
(eq. 2)
Where:
L = inductance (H)
ISET = 12.5 mA
RSET = gate to source resistance (W)
RDS(on) = On resistance of the HS MOSFET (W)
VIN = Input Voltage (V)
VOUT = Output Voltage (V)
FSW = Switching Frequency (Hz)
Figure 28. Boost Circuit
Reduced sampling time occurs at high duty cycles where
the low side MOSFET is off for the majority of the switching
period. Reduced sampling time causes errors in the
regulated voltage on the boost pin. High duty cycle / input
voltage induced sampling errors can result in increased
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13
NCP3020A, NCP3020B
Conditions where maximum boost ripple voltage could
damage the device or $12 V gate rated MOSFETs can be
seen in Region 3 (Orange). Placing a boost capacitor that is
no greater than 10X the input capacitance of the high side
MOSFET on the boost pin limits the maximum boost
voltage < 12 V. The typical drive waveforms for Regions 1,
2 and 3 (green, yellow, and orange) regions of Figure 29 are
shown in Figure 30.
boost ripple voltage or higher than desired DC boost voltage.
Figure 29 outlines all operating regions.
The recommended operating conditions are shown in
Region 1 (Green) where a 0.1 mF, 25 V ceramic capacitor
can be placed on the boost pin without causing damage to the
device or MOSFETS. Larger boost ripple voltage occurring
over several switching cycles is shown in Region 2 (Yellow).
The boost ripple frequency is dependant on the output
capacitance selected. The ripple voltage will not damage the
device or $12 V gate rated MOSFETs.
Boost Voltage Levels
Normal Operation
Increased Boost Ripple
(Still in Specification)
Increased Boost Ripple
Capacitor Optimization
Required
28
Region 3
26
24
22V
In p u t V o lt a g e
22
20
18
Region 2
Maxi
mum
Max
Duty
Duty
Cycle
Cycle
16
Region 1
14
12
10
11.5V
8
71%
6
4
2
5
10
15
20
25
30
35
40
45
50
55
60
65
70
75
80
85
90
Duty Cycle
Figure 29. Safe Operating Area for Boost Voltage with a 0.1 mF Capacitor
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14
NCP3020A, NCP3020B
VIN
7.5V
VBOOST
7.5V
0V
Maximum
Normal
VIN
7.5V
VBOOST
7.5V
0V
Maximum
Normal
VIN
7.5V
VBOOST
7.5V
0V
Figure 30. Typical Waveforms for the Green, Yellow, and Orange Region
To illustrate, a 0.1 mF boost capacitor operating at > 80% duty cycle and > 22.5 V input voltage will exceed the specifications
for the driver supply voltage. See Figure 31.
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15
NCP3020A, NCP3020B
Boost Voltage
18
Voltage Ripple
Maximum Allowable Voltage
Maximum Boost Voltage
16
14
Boost Voltage (V)
12
10
8
6
4
2
0
4.5
6.5
8.5
10.5
12.5
14.5
16.5
18.5
Input Voltage (V)
20.5
22.5
24.5
26.5
(Clarity on Boost Max and Ripple Def)
Figure 31. Boost Voltage at 80% Duty Cycle
Inductor Selection
D+
When selecting the inductor, it is important to know the
input and output requirements. Some example conditions
are listed below to assist in the process.
D+
Example Value
Input Voltage
(VIN)
9 V to 18 V
Nominal Input Voltage
(VIN)
12 V
(VOUT)
3.3V
Output Voltage
Input ripple voltage
(VINRIPPLE)
300 mV
Output ripple voltage
(VOUTRIPPLE)
50 mV
Output current rating
(IOUT)
10A
Operating frequency
(Fsw)
300 kHz
V IN * V HSD ) V LSD
³ 27.5% +
1
T
T OFF
T
[D+
(eq. 4)
V OUT
V IN
(eq. 5)
3.3 V
12 V
The ratio of ripple current to maximum output current
simplifies the equations used for inductor selection. The
formula for this is given in Equation 6.
ra +
DI
(eq. 6)
I OUT
The designer should employ a rule of thumb where the
percentage of ripple current in the inductor lies between
10% and 40%. When using ceramic output capacitors the
ripple current can be greater thus a user might select a higher
ripple current, but when using electrolytic capacitors a lower
ripple current will result in lower output ripple. Now,
acceptable values of inductance for a design can be
calculated using Equation 7.
A buck converter produces input voltage (VIN) pulses that
are LC filtered to produce a lower dc output voltage (VOUT).
The output voltage can be changed by modifying the on time
relative to the switching period (T) or switching frequency.
The ratio of high side switch on time to the switching period
is called duty cycle (D). Duty cycle can also be calculated
using VOUT, VIN, the low side switch voltage drop VLSD,
and the High side switch voltage drop VHSD.
F+
T
(* D Ǔ +
V OUT ) V LSD
Table 1. DESIGN PARAMETERS
Design Parameter
T ON
L+
(eq. 3)
+
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V OUT
I OUT @ ra @ F SW
@ (1 * D) ³ 3.3 mH
12 V
10 A @ 32.5% @ 300 kHz
(eq. 7)
@ (1 * 27.5%)
NCP3020A, NCP3020B
expense of higher ripple current. The peak−to−peak ripple
current for the NCP3020A is given by the following
equation:
L, INDUCTANCE (mH)
The relationship between ra and L for this design example
is shown in Figure 32.
18
17
16
18 V
15
14
15 V
13
12
11
10
12 V
9
8
7
6
5
4
9V
3
2
1
0
10%
15%
20%
25%
I PP +
Vout = 3.3 V
30%
35%
40%
LP CU + I RMS 2 @ DCR
To keep within the bounds of the parts maximum rating,
calculate the RMS current and peak current.
2
LP tot + LP CU_DC ) LP CU_AC ) LP Core (eq. 13)
(eq. 8)
Component Selection
2
Input Capacitor Selection
ra
32.5%
Ǔ
I PK + I OUT @ ǒ1 ) Ǔ ³ 11.63 A + 10 A @ ǒ1 )
2
The input capacitor has to sustain the ripple current
produced during the on time of the upper MOSFET, so it
must have a low ESR to minimize the losses. The RMS value
of this ripple is:
2
(eq. 9)
An inductor for this example would be around 3.3 mH and
should support an rms current of 10.04 A and a peak current
of 11.63 A.
The final selection of an output inductor has both
mechanical and electrical considerations. From a
mechanical perspective, smaller inductor values generally
correspond to smaller physical size. Since the inductor is
often one of the largest components in the regulation system,
a minimum inductor value is particularly important in
space−constrained applications. From an electrical
perspective, the maximum current slew rate through the
output inductor for a buck regulator is given by Equation 10.
SlewRate LOUT +
V IN * V OUT
L OUT
³ 1.5
(eq. 12)
The core losses and ac copper losses will depend on the
geometry of the selected core, core material, and wire used.
Most vendors will provide the appropriate information to
make accurate calculations of the power dissipation then the
total inductor losses can be capture buy the equation below:
Figure 32. Ripple Current Ration vs. Inductance
Ǹ1 ) ra12 ³ 10.04 A
+ 10 A @ Ǹ1 ) 32.5%
12
(eq. 11)
L OUT @ F SW
Ipp is the peak to peak current of the inductor. From this
equation it is clear that the ripple current increases as LOUT
decreases, emphasizing the trade−off between dynamic
response and ripple current.
The power dissipation of an inductor consists of both
copper and core losses. The copper losses can be further
categorized into dc losses and ac losses. A good first order
approximation of the inductor losses can be made using the
DC resistance as they usually contribute to 90% of the losses
of the inductor shown below:
VIN, (V)
I RMS + I OUT @
V OUT(1 * D)
Iin RMS + I OUT @ ǸD @ (1 * D)
(eq. 14)
D is the duty cycle, IinRMS is the input RMS current, and
IOUT is the load current.
The equation reaches its maximum value with D = 0.5.
Loss in the input capacitors can be calculated with the
following equation:
P CIN + ESR CIN @ ǒIIN RMSǓ
2
(eq. 15)
PCIN is the power loss in the input capacitors and ESRCIN
is the effective series resistance of the input capacitance.
Due to large dI/dt through the input capacitors, electrolytic
or ceramics should be used. If a tantalum must be used, it
must by surge protected. Otherwise, capacitor failure could
occur.
12 V * 3.3 V
A
+
ms
5.8 mH
(eq. 10)
This equation implies that larger inductor values limit the
regulator’s ability to slew current through the output
inductor in response to output load transients. Consequently,
output capacitors must supply the load current until the
inductor current reaches the output load current level. This
results in larger values of output capacitance to maintain
tight output voltage regulation. In contrast, smaller values of
inductance increase the regulator’s maximum achievable
slew rate and decrease the necessary capacitance, at the
Input Start−up Current
To calculate the input startup current, the following
equation can be used.
I INRUSH +
C OUT @ V OUT
t SS
(eq. 16)
Iinrush is the input current during startup, COUT is the total
output capacitance, VOUT is the desired output voltage, and
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NCP3020A, NCP3020B
tSS is the soft start interval. If the inrush current is higher than
the steady state input current during max load, then the input
fuse should be rated accordingly, if one is used.
2
DV OUT−DISCHG +
Output Capacitor Selection
ǒ
DV OUT−CHG +
(eq. 17)
Ǔ
1
8 @ F SW @ Co
ESL @ I PP @ F SW
V ESLOFF +
D
ESL @ I PP @ F SW
(1 * D )
(eq. 23)
C OUT @ V OUT
Power dissipation, package size, and the thermal
environment drive MOSFET selection. To adequately select
the correct MOSFETs, the design must first predict its power
dissipation. Once the dissipation is known, the thermal
impedance can be calculated to prevent the specified
maximum junction temperatures from being exceeded at the
highest ambient temperature.
Power dissipation has two primary contributors:
conduction losses and switching losses. The control or
high−side MOSFET will display both switching and
conduction losses. The synchronous or low−side MOSFET
will exhibit only conduction losses because it switches into
nearly zero voltage. However, the body diode in the
synchronous MOSFET will suffer diode losses during the
non−overlap time of the gate drivers.
Starting with the high−side or control MOSFET, the
power dissipation can be approximated from:
(eq. 18)
(eq. 19)
P D_CONTROL + P COND ) P SW_TOT
(eq. 20)
(eq. 24)
The first term is the conduction loss of the high−side
MOSFET while it is on.
The output capacitor is a basic component for the fast
response of the power supply. In fact, during load transient,
for the first few microseconds it supplies the current to the
load. The controller immediately recognizes the load
transient and sets the duty cycle to maximum, but the current
slope is limited by the inductor value.
During a load step transient the output voltage initially
drops due to the current variation inside the capacitor and the
ESR (neglecting the effect of the effective series inductance
(ESL)).
DV OUT−ESR + DI TRAN @ ESR Co
ǒI TRANǓ @ LOUT
Power MOSFET Selection
The ESL of capacitors depends on the technology chosen
but tends to range from 1 nH to 20 nH where ceramic
capacitors have the lowest inductance and electrolytic
capacitors then to have the highest. The calculated
contributing voltage ripple from ESL is shown for the switch
on and switch off below:
V ESLON +
(eq. 22)
2
The maximum allowable output voltage ripple is a
combination of the ripple current selected, the output
capacitance selected, the equivalent series inductance (ESL)
and ESR.
The main component of the ripple voltage is usually due
to the ESR of the output capacitor and the capacitance
selected.
V ESR_C + I O @ ra @ ESR Co )
C OUT @ ǒV IN * V OUTǓ
In a typical converter design, the ESR of the output capacitor
bank dominates the transient response. It should be noted
that DVOUT−DISCHARGE and DVOUT−ESR are out of
phase with each other, and the larger of these two voltages
will determine the maximum deviation of the output voltage
(neglecting the effect of the ESL).
Conversely during a load release, the output voltage can
increase as the energy stored in the inductor dumps into the
output capacitor. The ESR contribution from Equation 18
still applies in addition to the output capacitor charge which
is approximated by the following equation:
The important factors to consider when selecting an
output capacitor is dc voltage rating, ripple current rating,
output ripple voltage requirements, and transient response
requirements.
The output capacitor must be rated to handle the ripple
current at full load with proper derating. The RMS ratings
given in datasheets are generally for lower switching
frequency than used in switch mode power supplies but a
multiplier is usually given for higher frequency operation.
The RMS current for the output capacitor can be calculated
below:
ra
Co RMS + I O @
Ǹ12
ǒI TRANǓ @ LOUT
2
P COND + ǒI RMS_CONTROLǓ @ R DS(on)_CONTROL (eq. 25)
Using the ra term from Equation 6, IRMS becomes:
I RMS_CONTROL + I OUT @
Ǹ
D@
ǒ1 )12ra Ǔ
2
(eq. 26)
The second term from Equation 24 is the total switching
loss and can be approximated from the following equations.
P SW_TOT + P SW ) P DS ) P RR
(eq. 21)
(eq. 27)
The first term for total switching losses from Equation 27
includes the losses associated with turning the control
MOSFET on and off and the corresponding overlap in drain
voltage and current.
A minimum capacitor value is required to sustain the
current during the load transient without discharging it. The
voltage drop due to output capacitor discharge is
approximated by the following equation:
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NCP3020A, NCP3020B
P SW + P TON ) P TOFF
IG1: output current from the high−side gate drive (HSDR)
IG2: output current from the low−side gate drive (LSDR)
ƒSW: switching frequency of the converter. NCP3020A is
300 kHz and NCP3020B is 600 kHz
VBST: gate drive voltage for the high−side drive, typically
7.5 V.
QGD: gate charge plateau region, commonly specified in the
MOSFET datasheet
VTH: gate−to−source voltage at the gate charge plateau
region
QOSS: MOSFET output gate charge specified in the data
sheet
QRR: reverse recovery charge of the low−side or
synchronous MOSFET, specified in the datasheet
RDS(on)_CONTROL: on resistance of the high−side, or
control, MOSFET
RDS(on)_SYNC: on resistance of the low−side, or
synchronous, MOSFET
NOLLH: dead time between the LSDR turning off and the
HSDR turning on, typically 90 ns
NOLHL: dead time between the HSDR turning off and the
LSDR turning on, typically 80 ns
(eq. 28)
+ 1 @ ǒI OUT @ V IN @ f SWǓ @ ǒt ON ) t OFFǓ
2
where:
t ON +
Q GD
I G1
+
Q GD
(eq. 29)
ǒV BST * V THǓńǒR HSPU ) R GǓ
and:
t OFF +
Q GD
I G2
+
Q GD
ǒV BST * V THǓńǒR HSPU ) R GǓ
(eq. 30)
Next, the MOSFET output capacitance losses are caused
by both the control and synchronous MOSFET but are
dissipated only in the control MOSFET.
P DS + 1 @ Q OSS @ V IN @ f SW
2
(eq. 31)
Finally the loss due to the reverse recovery time of the
body diode in the synchronous MOSFET is shown as
follows:
P DS + Q RR @ V IN @ f SW
(eq. 32)
The low−side or synchronous MOSFET turns on into zero
volts so switching losses are negligible. Its power
dissipation only consists of conduction loss due to RDS(on)
and body diode loss during the non−overlap periods.
P D_SYNC + P COND ) P BODY
Once the MOSFET power dissipations are determined,
the designer can calculate the required thermal impedance
for each device to maintain a specified junction temperature
at the worst case ambient temperature. The formula for
calculating the junction temperature with the package in free
air is:
(eq. 33)
Conduction loss in the low−side or synchronous
MOSFET is described as follows:
T J + T A ) P D @ R qJA
2
P COND + ǒI RMS_SYNCǓ @ R DS(on)_SYNC (eq. 34)
where:
I RMS_CONTROL + I OUT @
Ǹ
ǒ
Ǔ
2
(1 * D) @ 1 ) ra
12
TJ: Junction Temperature
TA: Ambient Temperature
PD: Power Dissipation of the MOSFET under analysis
RqJA: Thermal Resistance Junction−to−Ambient of the
MOSFET’s package
(eq. 35)
The body diode losses can be approximated as:
As with any power design, proper laboratory testing
should be performed to insure the design will dissipate the
required power under worst case operating conditions.
Variables considered during testing should include
maximum ambient temperature, minimum airflow,
maximum input voltage, maximum loading, and component
variations (i.e. worst case MOSFET RDS(on)).
P BODY + V FD @ I OUT @ f SW @ ǒNOL LH ) NOL HLǓ (eq. 36)
Vth
Figure 33. MOSFET Switching Characteristics
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NCP3020A, NCP3020B
NOLHL
NOLLH
High−Side
Logic Signal
Low−Side
Logic Signal
td(on)
tf
RDSmax
High−Side
MOSFET
RDS(on)min
tr
td(off)
tr
tf
RDSmax
Low−Side
MOSFET
RDS(on)min
td(on)
td(off)
Figure 34. MOSFETs Timing Diagram
response. The goal of the compensation circuit is to provide
a loop gain function with the highest crossing frequency and
adequate phase margin (minimally 45°). The transfer
function of the power stage (the output LC filter) is a double
pole system. The resonance frequency of this filter is
expressed as follows:
Another consideration during MOSFET selection is their
delay times. Turn−on and turn−off times must be short
enough to prevent cross conduction. If not, there will be
conduction from the input through both MOSFETs to
ground. Therefore, the following conditions must be met.
t d(ON)_CONTROL ) NOL LH u t d(OFF)_SYNC ) t f_SYNC
f P0 +
(eq. 37)
and
t (ON)_SYNC ) NOL HL u t d(OFF)_CONTROL ) t f _CONTROL
1
2 @ p @ ǸL @ C OUT
(eq. 38)
Parasitic Equivalent Series Resistance (ESR) of the
output filter capacitor introduces a high frequency zero to
the filter network. Its value can be calculated by using the
following equation:
The MOSFET parameters, td(ON), tr, td(OFF) and tf are can
be found in their appropriate datasheets for specific
conditions. NOLLH and NOLHL are the dead times which
were described earlier and are 90 ns and 80 ns, respectively.
f Z0 +
Feedback and Compensation
The NCP3020 is a voltage mode buck convertor with a
transconductance error amplifier compensated by an
external compensation network. Compensation is needed to
achieve accurate output voltage regulation and fast transient
1
2 @ p @ C OUT @ ESR
(eq. 39)
The main loop zero crossover frequency f0 can be chosen
to be 1/10 − 1/5 of the switching frequency. Table 2 shows
the three methods of compensation.
Table 2. COMPENSATION TYPES
Zero Crossover Frequency Condition
Compensation Type
Typical Output Capacitor Type
fP0 < fZ0 < f0 < fS/2
Type II (PI)
Electrolytic, Tantalum
fP0 < f0 < fZ0 < fS/2
Type III (PID) Method I
Tantalum, Ceramic
fP0 < f0 < fS/2 < fZ0
Type III (PID) Method II
Ceramic
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NCP3020A, NCP3020B
Compensation Type II (PI)
This compensation is suitable for electrolytic capacitors.
Components of the PI compensation (Figure 35) network
can be specified by the following equations:
f Z1 + 0.75 @ f P0
(eq. 44)
f Z2 + f P0
(eq. 45)
f P2 + f Z0
(eq. 46)
fS
f P3 +
(eq. 47)
2
Method II is better suited for ceramic capacitors that
typically have the lowest ESR available:
Figure 35. PI Compensation (II Type)
R C1 +
2 @ p @ f 0 @ L @ V RAMP @ V OUT
ESR @ V IN @ V ref @ gm
(eq. 41)
C C2 +
1
p @ R C1 @ f S
(eq. 42)
V ref
@ R2
(eq. 48)
f P2 + f 0 @
sin q max
Ǹ11 *) sin
q max
(eq. 49)
f Z1 + 0.5 @ f Z2
(eq. 50)
f P3 + 0.5 @ f S
(eq. 51)
R C1 u u
1
0.75 @ 2 @ p @ f P0 @ R C1
V OUT * V ref
sinq max
Ǹ11 )* sin
q max
The remaining calculations are the same for both methods.
(eq. 40)
C C1 +
R1 +
f Z2 + f 0 @
1
2 @ p @ f Z1 @ R C1
(eq. 53)
C C2 +
1
2 @ p @ f P3 @ R C1
(eq. 54)
C FB1 +
R1 +
Tantalum and ceramics capacitors have lower ESR than
electrolytic, so the zero of the output LC filter goes to a
higher frequency above the zero crossover frequency. This
requires a PID compensation network as shown in
Figure 36.
There are two methods to select the zeros and poles of this
compensation network. Method I is ideal for tantalum
output capacitors, which have a higher ESR than ceramic:
R2 +
2 @ p @ f 0 @ L @ V RAMP @ C OUT
V IN @ R C1
(eq. 55)
1
2p @ C FB1 @ f P2
(eq. 56)
1
2 @ p @ C FB1 @ f Z2
(eq. 57)
R FB1 +
Compensation Type III (PID)
(eq. 52)
C C1 +
(eq. 43)
VRAMP is the peak−to−peak voltage of the oscillator ramp
and gm is the transconductance error amplifier gain.
Capacitor CC2 is optional.
2
gm
V
ref
V OUT * V ref
@ R1
(eq. 58)
If the equation in Equation 59 is not true, then a higher value
of RC1 must be selected.
R1 @ R2 @ R FB1
R1 @ R FB1 ) R2 @ R FB1 @ R1 @ R2
Figure 36. PID Compensation (III Type)
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21
u
1
(eq. 59)
gm
NCP3020A, NCP3020B
TYPICAL APPLICATION CIRCUIT
9−18 V
C IN−1/2
CIN−3/4
CIN−5
COMP
RC
C c1
VCC
BST
NCP3020A
C BST
HSDR
RG
Q1
RGS
LO
3.3 V
VSW
Q2
LSDR
FB
C c2
D1
RFB1
C FB
R ISET
GND
R FB3
R FB2
Figure 37. Typical Application, VIN = 9 − 18 V, VOUT = 3.3 V, IOUT = 10 A
Reference Designator
Value
CIN−1
470 mF
CIN−2
470 mF
CIN−3
22 mF
CIN−4
22 mF
CIN−5
1 mF
CC1
82 nF
CC2
1.2 nF
CFB
6.8 nF
COUT1
470 mF
COUT2
22 mF
COUT3
22 mF
CBST
0.1 mF
RC
750 W
RG
8.06 W
RGS
10 kW
RISET
22.1 kW
RFB1
4.53 kW
RFB2
1.0 kW
RFB3
1.0 kW
Q1
NTMFS4841N
Q2
NTMFS4935
D1
BAT54
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C OUT −1
C OUT −2/3
NCP3020A, NCP3020B
PACKAGE DIMENSIONS
SOIC−8 NB
CASE 751−07
ISSUE AJ
−X−
NOTES:
1. DIMENSIONING AND TOLERANCING PER
ANSI Y14.5M, 1982.
2. CONTROLLING DIMENSION: MILLIMETER.
3. DIMENSION A AND B DO NOT INCLUDE
MOLD PROTRUSION.
4. MAXIMUM MOLD PROTRUSION 0.15 (0.006)
PER SIDE.
5. DIMENSION D DOES NOT INCLUDE DAMBAR
PROTRUSION. ALLOWABLE DAMBAR
PROTRUSION SHALL BE 0.127 (0.005) TOTAL
IN EXCESS OF THE D DIMENSION AT
MAXIMUM MATERIAL CONDITION.
6. 751−01 THRU 751−06 ARE OBSOLETE. NEW
STANDARD IS 751−07.
A
8
5
S
B
0.25 (0.010)
M
Y
M
1
4
−Y−
K
G
C
N
DIM
A
B
C
D
G
H
J
K
M
N
S
X 45 _
SEATING
PLANE
−Z−
0.10 (0.004)
H
D
0.25 (0.010)
M
Z Y
S
X
M
J
S
MILLIMETERS
MIN
MAX
4.80
5.00
3.80
4.00
1.35
1.75
0.33
0.51
1.27 BSC
0.10
0.25
0.19
0.25
0.40
1.27
0_
8_
0.25
0.50
5.80
6.20
INCHES
MIN
MAX
0.189
0.197
0.150
0.157
0.053
0.069
0.013
0.020
0.050 BSC
0.004
0.010
0.007
0.010
0.016
0.050
0 _
8 _
0.010
0.020
0.228
0.244
SOLDERING FOOTPRINT*
1.52
0.060
7.0
0.275
4.0
0.155
0.6
0.024
1.270
0.050
SCALE 6:1
mm Ǔ
ǒinches
*For additional information on our Pb−Free strategy and soldering
details, please download the ON Semiconductor Soldering and
Mounting Techniques Reference Manual, SOLDERRM/D.
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