ES EW D ® EL7562 I GNS N FO R D E D 32 MEN EL75Sheet COM SEEData E R NOT April 30, 2004 Monolithic 2Amp DC-DC Step-Down Regulator FN7295 Features The EL7562 is an integrated, synchronous step-down regulator with output voltage adjustable from 1.0V to 3.8V. It is capable of delivering 2A continuous current at up to 95% efficiency. The EL7562 operates at a constant frequency pulse width modulation (PWM) mode, making external synchronization possible. Patented on-chip resistorless current sensing enables current mode control, which provides cycle-by-cycle current limiting, over-current protection, and excellent step load response. The EL7562 is available in a fused-lead 16-pin QSOP package. With proper external components, the whole converter fits into a less than 0.5 in2 area. The minimal external components and small size make this EL7562 ideal for desktop and portable applications. • Integrated synchronous MOSFETs and current mode controller • 2A continuous output current • Up to 95% efficiency • 3.3V or 5V nominal input voltage • Adjustable output from 1V to 3.8V • Cycle-by-cycle current limit • Precision reference • ±0.5% load and line regulation • Adjustable switching frequency to 1MHz • Oscillator synchronization possible • Internal soft-start The EL7562 is specified for operation over the 0°C to +70°C temperature range. • Over-temperature protection Pinout • 16-pin QSOP package EL7562 (16-PIN QSOP) TOP VIEW • Under-voltage lockout Applications • DSP, CPU core and IO supplies • Logic/Bus supplies • Portable equipment C3 C4 0.1µF 270pF R3 1 SGND PGND 16 2 COSC VREF 15 3 VDD 39Ω C1 100µF VIN (4.5V- 5.5V) C2 0.1µF FB 14 4 PGND VDRV 13 5 PGND LX 12 6 VIN LX 11 7 VIN VHI 10 8 EN PGND 9 • DC-DC converter modules C5 • GTL + Bus power supply 0.1µF R2 R1 2.37kΩ 1kΩ C6 0.1µF Ordering Information VO (3.3V, C7 2A) 100µF PART NUMBER PACKAGE TAPE & REEL PKG. DWG. # EL7562CU 16-Pin QSOP - MDP0040 EL7562CU-T7 16-Pin QSOP 7” MDP0040 EL7562CU-T13 16-Pin QSOP 13” MDP0040 Please refer to page 4 for 3.3V input Application Diagram Manufactured under U.S. Patent No. 57,323,974 1 CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 321-724-7143 | Intersil (and design) is a registered trademark of Intersil Americas Inc. Copyright © Intersil Americas Inc. 2004. All Rights Reserved. Elantec is a registered trademark of Elantec Semiconductor, Inc. All other trademarks mentioned are the property of their respective owners. Absolute Maximum Ratings (TA = 25°C) Supply Voltage between VIN or VDD and GND . . . . . . . . . . . . +6.5V VLX Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . VIN +0.3V Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . GND -0.3V, VDD +0.3V VHI Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . GND -0.3V, VLX +6V Storage Temperature . . . . . . . . . . . . . . . . . . . . . . . .-65°C to +150°C Operating Ambient Temperature . . . . . . . . . . . . . . . . . 0°C to +70°C Operating Junction Temperature . . . . . . . . . . . . . . . . . . . . . . +135°C CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied. IMPORTANT NOTE: All parameters having Min/Max specifications are guaranteed. Typical values are for information purposes only. Unless otherwise noted, all tests are at the specified temperature and are pulsed tests, therefore: TJ = TC = TA DC Electrical Specifications PARAMETER VDD = VIN = 5V, TA = TJ = 25°C, COSC = 270pF, unless otherwise specified. DESCRIPTION VREF Reference Accuracy VREFTC Reference Temperature Coefficient VREFLOAD Reference Load Regulation VRAMP Oscillator Ramp Amplitude IOSC_CHG Oscillator Charge Current IOSC_DIS CONDITIONS MIN TYP MAX UNIT 1.24 1.26 1.28 V 50 0 < IREF < 50µA ppm/°C -1 % 1.15 V 0.1V < VOSC < 1.25V 200 µA Oscillator Discharge Current 0.1V < VOSC < 1.25V 8 mA IVDD+VDRV VDD+VDRV Supply Current VEN = 4V, FOSC = 120kHz IVDD_OFF VDD Standby Current EN = 0 VDD_OFF VDD for Shutdown VDD_ON VDD for Startup TOT Over-temperature Threshold 135 °C THYS Over-temperature Hysteresis 20 °C ILEAK Internal FET Leakage Current ILMAX Peak Current Limit RDSON FET On Resistance RDSONTC RDSON Tempco VFB Output Initial Accuracy ILOAD = 0A VFB_LINE Output Line Regulation VIN = 5V, ∆VIN = 10%, ILOAD = 0A 0.5 % VFB_LOAD Output Load Regulation 0.1A < ILOAD < 1A 0.5 % VFB_TC Output Temperature Stability -40°C < TA < 85°C, ILOAD = 0.5A ±1 % IFB Feedback Input Pull Up Current VFB = 0V 100 VEN_HI EN Input High Level (Note) VEN_LO EN Input Low Level IEN Enable Pull Up Current 6.5 mA 1.5 mA 2.5 2.7 V 2.6 3 V 1 EN = 0, LX = 5V (low FET), LX = 0V (high FET) 20 3 Wafer level test only 60 0.970 120 0.985 1.000 200 4 -4 mΩ mΩ/°C V nA V 1 VEN = 0 µA A 0.2 NOTE: VEN_HI is typically 2/3 of VDD. For VDD = 3.3V, VEN_HI is 2.2V typical. 2 2 -2.5 V µA Closed-Loop AC Electrical Specifications PARAMETER VS = VIN = 5V, TA = TJ = 25°C, COSC = 270pF, unless otherwise specified. DESCRIPTION CONDITIONS MIN TYP MAX UNIT 493 580 667 kHz FOSC Oscillator Initial Accuracy tSYNC Minimum Oscillator Sync Width 25 ns MSS Soft-start Slope 0.5 V/ms tBRM FET Break Before Make Delay 15 ns tLEB High Side FET Minimum On Time 150 ns DMAX Maximum Duty Cycle 95 % Pin Descriptions PIN NUMBER PIN NAME 1 SGND Control circuit negative supply 2 COSC Oscillator timing capacitor; FOSC can be approximated by: FOSC (kHz) = 0.1843/COSC, COSC in µF 3 VDD 4 PGND Ground return of the regulator; connected to the source of the low-side synchronous NMOS power FET 5 PGND Ground return of the regulator; connected to the source of the low-side synchronous NMOS power FET 6 VIN Power supply input of the regulator; connected to the drain of the high-side NMOS power FET 7 VIN Power supply input of the regulator; connected to the drain of the high-side NMOS power FET 8 EN Chip enable, active high; a 2µA internal pull-up current enables the device if the pin is left open 9 PGND 10 VHI Positive supply of the high-side driver 11 LX Inductor drive pin; high current digital output whose average voltage equals the regulator output voltage 12 LX Inductor drive pin; high current digital output whose average voltage equals the regulator output voltage 13 VDRV 14 FB 15 VREF Bandgap reference bypass capacitor; typically 0.1µF to GND 16 PGND Ground return of the regulator PIN FUNCTION Control circuit positive supply Ground return of the regulator Positive supply of the low-side driver and input voltage for the high-side boot strap Voltage feedback input; connected to an external resistor divider between VOUT and GND; a 125nA pull-up current forces VOUT to VS in the event that FB is floating 3 Application Diagram for 3.3V Input C3 C4 0.1µF 270pF 1 SGND PGND 16 C5 2 COSC VREF 15 0.1µF R3 3 VDD FB 14 39Ω C1 C2 100µF 0.1µF VIN (3V-3.6V) 4 PGND VDRV 13 5 PGND LX 12 D3 D4 C8 C9 0.1µF 0.1µF VO (2.5V, 2A) L1 6 VIN LX 11 7 VIN VHI 10 8 EN PGND 9 EL7562 (16-Pin QSOP) 4 D2 C6 0.1µF 4.7µF C7 R2 100µF 1.54kΩ R1 1kΩ Typical Performance Curves 100 Efficiency vs IO VIN=5V 0.7 VO=2.5 95 VO=3.3 0.6 VO=1.8 80 VO=1.5 Power Loss (W) Efficiency (%) 90 85 VO=1.2 75 70 FS=500kH 65 Power Loss vs IO VIN=5V 0.5 VO=3.3 0.4 VO=2.5 0.3 VO=1.8 VO=1.2 0.2 VO=1.5 0.1 L=Coilcraft DO3316P60 0.1 0 1 2 0 0.5 Load Current IO (A) Efficiency vs IO VO=3.3V Efficiency (%) 0.8 1.5 2 Load Regulation VO=3.3V 0.6 VIN=5V 90 85 VIN=5.5 80 75 70 VIN=5.5 0.4 0.2 VIN=5V 0 VIN=4.5 -0.2 -0.4 65 60 -0.6 0 0.5 1 1.5 0 2 0.5 Load Current IO (A) 1 Load Current IO (A) Line Regulation VO=3.3V VREF vs Temperature 1.258 1.256 0.4 IO=0.1A 1.254 VREF (V) 0.2 VO (%) 2 VIN=4.5 95 0.6 1.5 Load Current IO (A) Output Voltage (%) 100 1 IO=1A 0 IO=2A -0.2 1.252 1.25 1.248 1.246 -0.4 -0.6 4.5 1.244 1.242 4.7 4.9 5.1 VIN (V) 5 5.3 5.5 0 10 20 30 40 50 60 70 Temperature (°C) 80 90 100 110 Typical Performance Curves (Continued) Input Current vs Temperature (Enable connected to GND) Oscillator Frequency vs Temperature 0.96 390 0.92 380 375 370 VIN=5V 0.9 VIN=4.5 0.88 0.86 0.84 365 0.82 360 0.8 0 10 20 30 40 50 60 70 80 90 100 110 Temperature (°C) 1400 1200 1000 800 600 400 200 0 0 200 400 600 COSC (pF) 6 0 10 20 30 40 50 60 70 Temperature (°C) Switching Frequency vs COSC FS (kHz) VIN=5.5 0.94 385 Input Current (A) Oscillator Frequency (kHz) COSC=390p 800 1000 80 90 100 110 Block Diagram 0.1µF VREF Junction Temperature Voltage Reference 270pF COSC VDRV Oscillator VHI Controller Supply 39Ω 5V VDD VIN 0.1µF Power 0.1µF PWM Controlle FET 4.7µH Drivers VOUT Power 100µF FET PGND EN 2370Ω 1kΩ Current Sense SGND FB Applications Information Circuit Description General The EL7562 is a fixed frequency, current mode controlled DC-DC converter with integrated N-channel power MOSFETs and a high precision reference. The device incorporates all the active circuitry required to implement a cost effective, user-programmable 2A synchronous stepdown regulator suitable for use in DSP core power supplies. Theory of Operation The EL7562 is composed of 5 major blocks: 1. PWM Controller 2. NMOS Power FETs and Drive Circuitry 3. Bandgap Reference 4. Oscillator 5. Thermal Shut-down PWM Controller The EL7562 regulates output voltage through the use of current-mode controlled pulse width modulation. The three 7 main elements in a PWM controller are the feedback loop and reference, a pulse width modulator whose duty cycle is controlled by the feedback error signal, and a filter which averages the logic level modulator output. In a step-down (buck) converter, the feedback loop forces the timeaveraged output of the modulator to equal the desired output voltage. Unlike pure voltage-mode control systems, currentmode control utilizes dual feedback loops to provide both output voltage and inductor current information to the controller. The voltage loop minimizes DC and transient errors in the output voltage by adjusting the PWM duty-cycle in response to changes in line or load conditions. Since the output voltage is equal to the time-averaged of the modulator output, the relatively large LC time constant found in power supply applications generally results in low bandwidth and poor transient response. By directly monitoring changes in inductor current via a series sense resistor the controller's response time is not entirely limited by the output LC filter and can react more quickly to changes in line and load conditions. This feed-forward characteristic also simplifies AC loop compensation since it adds a zero to the overall loop response. Through proper selection of the currentfeedback to voltage-feedback ratio the overall loop response will approach a one-pole system. The resulting system offers several advantages over traditional voltage control systems, including simpler loop compensation, pulse by pulse current limiting, rapid response to line variation and good load step response. The heart of the controller is an input direct summing comparator which sum voltage feedback, current feedback, slope compensation ramp and power tracking signals together. Slope compensation is required to prevent system instability that occurs in current-mode topologies operating at duty-cycles greater than 50% and is also used to define the open-loop gain of the overall system. The slope compensation is fixed internally and optimized for 500mA inductor ripple current. The power tracking will not contribute any input to the comparator steady-state operation. Current feedback is measured by the patented sensing scheme that senses the inductor current flowing through the high-side switch whenever it is conducting. At the beginning of each oscillator period the high-side NMOS switch is turned on. The comparator inputs are gated off for a minimum period of time of about 150ns (LEB) after the high-side switch is turned on to allow the system to settle. The Leading Edge Blanking (LEB) period prevents the detection of erroneous voltages at the comparator inputs due to switching noise. If the inductor current exceeds the maximum current limit (ILMAX) a secondary over-current comparator will terminate the high-side switch on time. If ILMAX has not been reached, the feedback voltage FB derived from the regulator output voltage VOUT is then compared to the internal feedback reference voltage. The resultant error voltage is summed with the current feedback and slope compensation ramp. The high-side switch remains on until all four comparator inputs have summed to zero, at which time the high-side switch is turned off and the low-side switch is turned on. However, the maximum on-duty ratio of the high-side switch is limited to 95%. In order to eliminate cross-conduction of the high-side and low-side switches a 15ns break-beforemake delay is incorporated in the switch drive circuitry. The output enable (EN) input allows the regulator output to be disabled by an external logic control signal. Output Voltage Setting In general: R V OUT = 0.985 × 1 + ------2- R 1 For VIN = 5V R V OUT = 0.975 × 1 + ------2- R 1 For VIN = 3.3V However, due to the relatively low open loop gain of the system, gain errors will occur as the output voltage and loopgain is changed. This is shown in the performance curves. A 8 100nA pull-up current from FB to VDD forces VOUT to GND in the event that FB is floating. NMOS Power FETs and Drive Circuitry The EL7562 integrates low on-resistance (60mΩ) NMOS FETs to achieve high efficiency at 2A. In order to use an NMOS switch for the high-side drive it is necessary to drive the gate voltage above the source voltage (LX). This is accomplished by bootstrapping the VHI pin above the LX voltage with an external capacitor CVHI and internal switch and diode. When the low-side switch is turned on and the LX voltage is close to GND potential, capacitor CVHI is charged through internal switch to VDRV, typically 5V. At the beginning of the next cycle the high-side switch turns on and the LX pins begin to rise from GND to VIN potential. As the LX pin rises the positive plate of capacitor CVHI follows and eventually reaches a value of VDRV+VIN, typically 10V, for VDRV=VIN=5V. This voltage is then level shifted and used to drive the gate of the high-side FET, via the VHI pin. A value of 0.1µF for CVHI is recommended. Reference A 1.5% temperature compensated bandgap reference is integrated in the EL7562. The external VREF capacitor acts as the dominant pole of the amplifier and can be increased in size to maximize transient noise rejection. A value of 0.1µF is recommended. Oscillator The system clock is generated by an internal relaxation oscillator with a maximum duty-cycle of approximately 95%. Operating frequency can be adjusted through the COSC pin or can be driven by an external source. If the oscillator is driven by an external source care must be taken in selecting the ramp amplitude. Since CSLOPE value is derived from the COSC ramp, changes to COSC ramp will change the CSLOPE compensation ramp which determine the open-loop gain of the system. When external synchronization is required, always choose COSC such that the free-running frequency is at least 20% lower than that of sync source to accommodate component Layout Considerations and temperature variations. Figure 1 shows a typical connection. 100p BAT54 External Oscillato 1 16 2 15 3 14 6 11 7 10 8 9 EL7562 The layout is very important for the converter to function properly. Power Ground ( ) and Signal Ground (--) should be separated to ensure that the high pulse current in the Power Ground never interferes with the sensitive signals connected to Signal Ground. They should only be connected at one point (normally at the negative side of either the input or output capacitor). The trace connected to pin 14 (FB) is the most sensitive trace. It needs to be as short as possible and in a “quiet” place, preferably between PGND or SGND traces. In addition, the bypass capacitor connected to the VDD pin needs to be as close to the pin as possible. The heat of the chip is mainly dissipated through the PGND pins. Maximizing the copper area around these pins is preferable. In addition, a solid ground plane is always helpful for the EMI performance. FIGURE 1. Oscillator Synchronization Thermal Shut-down An internal temperature sensor continuously monitors die temperature. In the event that die temperature exceeds the thermal trip-point, the system is in fault state and will be shut down. The upper and low trip-points are set to 135°C and 115°C respectively. The demo board is a good example of layout based on these principles. Please refer to the EL7562 Application Brief for the layout. Start-up Delay A capacitor can be added to the EN pin to delay the converter start-up (Figure 2) by utilizing the pull-up current. The delay time is approximately: t d ( ms ) = 1200 × C ( µF ) C 1 1 2 1 3 1 6 1 7 1 8 VOU VIN VO td 9 EL7562 TIME FIGURE 2. Start-up Delay All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems. Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries. For information regarding Intersil Corporation and its products, see www.intersil.com 9