LINER LTC3890EGN-1TRPBF 60v low iq, dual, 2-phase synchronous step-down dc/dc controller Datasheet

LTC3890-1
60V Low IQ,
Dual, 2-Phase Synchronous
Step-Down DC/DC Controller
DESCRIPTION
FEATURES
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The LTC®3890-1 is a high performance dual step-down
switching regulator DC/DC controller that drives all Nchannel synchronous power MOSFET stages. A constant
frequency current mode architecture allows a phase-lockable frequency of up to 850kHz. Power loss and supply
noise are minimized by operating the two controller output
stages out-of-phase.
Wide VIN Range: 4V to 60V (65V Abs Max)
Low Operating IQ: 50μA (One Channel On)
Wide Output Voltage Range: 0.8V ≤ VOUT ≤ 24V
RSENSE or DCR Current Sensing
Out-of-Phase Controllers Reduce Required Input
Capacitance and Power Supply Induced Noise
Phase-Lockable Frequency (75kHz to 850kHz)
Programmable Fixed Frequency (50kHz to 900kHz)
Selectable Continuous, Pulse Skipping or Low Ripple
Burst Mode® Operation at Light Loads
Very Low Dropout Operation: 99% Duty Cycle
Adjustable Output Voltage Soft-Start or Tracking
Power Good Output Voltage Monitor
Output Overvoltage Protection
Low Shutdown IQ: <14μA
Internal LDO Powers Gate Drive from VIN or EXTVCC
No Current Foldback During Start-Up
Narrow SSOP Package
The 50μA no-load quiescent current extends operating life
in battery-powered systems. OPTI-LOOP® compensation
allows the transient response to be optimized over a wide
range of output capacitance and ESR values. A wide 4V
to 60V input supply range encompasses a wide range of
intermediate bus voltages and battery chemistries.
Independent TRACK/SS pins for each controller ramp the
output voltages during start-up. Current foldback limits
MOSFET heat dissipation during short-circuit conditions.
The PLLIN/MODE pin selects among Burst Mode operation,
pulse skipping mode, or continuous conduction mode at
light loads.
APPLICATIONS
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For a leadless 32-pin QFN package with additional features
of adjustable current limit, clock out, phase modulation
and two PGOOD outputs, see the LTC3890 data sheet.
Automotive Always-On Systems
Battery Operated Digital Devices
Distributed DC Power Systems
L, LT, LTC, LTM, Linear Technology, Burst Mode, OPTI-LOOP and the Linear logo are
registered trademarks of Linear Technology Corporation. All other trademarks are the
property of their respective owners. Protected by U.S. Patents including 5481178, 5705919,
5929620, 6100678, 6144194, 6177787, 6304066, 6580258, 7230497.
TYPICAL APPLICATION
Efficiency and Power Loss
vs Output Current (Buck)
High Efficiency Dual 8.5V/3.3V Output Step-Down Converter
VIN
9V TO 60V
22μF
4.7μF
VIN
TG2
4.7μH
BOOST2
SW1
SW2
BG1
BG2
LTC3890-1
0.1μF
8μH
PGND
SENSE1+
SENSE2+
SENSE1–
VFB1
ITH1
SENSE2–
VFB2
ITH2
1000pF
31.6k
34.8k
TRACK/SS1 SGND TRACK/SS2
0.1μF
0.1μF
1000
70
100
60
50
10
40
20
0.01Ω
100k
470μF
80
30
0.008Ω
VOUT1
3.3V
5A
10000
100k
1000pF
10.5k
VOUT2
8.5V
3A
330μF
POWER LOSS (mW)
BOOST1
EFFICIENCY (%)
TG1
0.1μF
100
VIN = 12V
90 VOUT = 3.3V
INTVCC
1
10
0
0.0001
0.001
0.01
0.1
1
OUTPUT CURRENT (A)
0.1
10
38901 TA01b
34.8k
38901 TA01
38901f
1
LTC3890-1
ABSOLUTE MAXIMUM RATINGS
PIN CONFIGURATION
(Note 1)
Input Supply Voltage (VIN) ......................... –0.3V to 65V
Topside Driver Voltages
BOOST1, BOOST2 ................................. –0.3V to 71V
Switch Voltage (SW1, SW2) ........................ –5V to 65V
(BOOST1-SW1), (BOOST2-SW2) ................ –0.3V to 6V
RUN1, RUN2 ............................................... –0.3V to 8V
Maximum Current Sourced into Pin from
Source >8V .......................................................100μA
SENSE1+, SENSE2+, SENSE1–
SENSE2– Voltages...................................... –0.3V to 28V
PLLIN/MODE, FREQ Voltages .............. –0.3V to INTVCC
EXTVCC ...................................................... –0.3V to 14V
ITH1, ITH2, VFB1, VFB2 Voltages ................... –0.3V to 6V
PGOOD1 Voltage ......................................... –0.3V to 6V
TRACK/SS1, TRACK/SS2 Voltages .............. –0.3V to 6V
Operating Junction Temperature Range
(Note 2) ................................................. –40°C to 125°C
Maximum Junction Temperature (Note 3) ............ 125°C
Storage Temperature Range................... –65°C to 150°C
TOP VIEW
ITH1
1
28 TRACK/SS1
VFB1
2
27 PGOOD1
SENSE1+
3
26 TG1
SENSE1–
4
25 SW1
FREQ
5
24 BOOST1
PLLIN/MODE
6
23 BG1
SGND
7
22 VIN
RUN1
8
21 PGND
RUN2
9
20 EXTVCC
SENSE2–
10
19 INTVCC
SENSE2+ 11
18 BG2
VFB2 12
17 BOOST2
ITH2 13
16 SW2
TRACK/SS2 14
15 TG2
GN PACKAGE
28-LEAD PLASTIC SSOP
TJMAX = 125°C, θJA = 90°C/W
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3890EGN-1#PBF
LTC3890EGN-1#TRPBF
LTC3890GN-1
28-Lead Plastic SSOP
–40°C to 125°C
LTC3890IGN-1#PBF
LTC3890IGN-1#TRPBF
LTC3890GN-1
28-Lead Plastic SSOP
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
junction temperature range, otherwise specifications are at TJ = 25°C. VIN = 12V, VRUN1,2 = 5V, EXTVCC = 0V unless otherwise noted.
SYMBOL
PARAMETER
VIN
Input Supply Operating Voltage Range
VFB1,2
Regulated Feedback Voltage
CONDITIONS
MIN
TYP
4
ITH1,2 Voltage = 1.2V (Note 4)
–40°C to 125°C
–40°C to 85°C
IFB1,2
Feedback Current
(Note 4)
VREFLNREG
Reference Voltage Line Regulation
VIN = 4.5V to 60V (Note 4)
VLOADREG
Output Voltage Load Regulation
(Note4)
Measured in Servo Loop,
ΔITH Voltage = 1.2V to 0.7V
(Note4)
Measured in Servo Loop,
ΔITH Voltage = 1.2V to 2V
l
0.788
0.792
0.800
0.800
MAX
UNITS
60
V
0.812
0.808
V
V
±5
±50
nA
0.002
0.02
%/V
l
0.01
0.1
%
l
–0.01
–0.1
%
38901f
2
LTC3890-1
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
junction temperature range, otherwise specifications are at TJ = 25°C. VIN = 12V, VRUN1,2 = 5V, EXTVCC = 0V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
gm1,2
Transconductance Amplifier gm
ITH1,2 = 1.2V, Sink/Source = 5μA (Note 4)
IQ
Input DC Supply Current
(Note 5)
Pulse Skip or Forced Continuous Mode
(One Channel On)
UVLO
MIN
TYP
MAX
UNITS
2
mmho
RUN1 = 5V and RUN2 = 0V or
RUN1 = 0V and RUN2 = 5V,
VFB1 = 0.83V (No Load)
2
mA
Pulse Skip or Forced Continuous Mode
(Both Channels On)
RUN1,2 = 5V, VFB1,2 = 0.83V (No Load)
2
mA
Sleep Mode (One Channel On)
RUN1 = 5V and RUN2 = 0V or
RUN1 = 0V and RUN2 = 5V,
VFB1 = 0.83V (No Load)
50
75
μA
Sleep Mode (Both Channels On)
RUN1,2 = 5V, VFB1,2 = 0.83V (No Load)
60
100
μA
Shutdown
RUN1,2 = 0V
14
25
μA
Undervoltage Lockout
INTVCC Ramping Up
INTVCC Ramping Down
3.6
4.0
3.8
4.2
4.0
V
V
7
10
13
%
±1
μA
±1
μA
μA
VOVL
Feedback Overvoltage Protection
Measured at VFB1,2, Relative to Regulated VFB1,2
ISENSE+
ISENSE–
SENSE+ Pin Current
Each Channel
SENSE– Pins Current
Each Channel
VSENSE– < INTVCC – 0.5V
VSENSE– > INTVCC + 0.5V
l
l
700
DFMAX
Maximum Duty Factor
In Dropout
98
99
ITRACK/SS1,2
Soft-Start Charge Current
VTRACK1,2 = 0V
0.7
1.0
1.4
μA
VRUN1 On
VRUN2 On
RUN1 Pin On Threshold
RUN2 Pin On Threshold
VRUN1 Rising
VRUN2 Rising
1.15
1.20
1.21
1.25
1.27
1.30
V
V
l
l
VRUN1,2 Hyst RUN Pin Hysteresis
VSENSE(MAX)
Maximum Current Sense Threshold
%
50
VFB1,2 = 0.7V, VSENSE1–,2– = 3.3V, ILIM = 0
l
64
75
mV
85
mV
Gate Driver
TG1,2
Pull-Up On-Resistance
Pull-Down On-Resistance
2.5
1.5
Ω
Ω
BG1,2
Pull-Up On-Resistance
Pull-Down On-Resistance
2.4
1.1
Ω
Ω
TG1,2 tr
TG1,2 tf
TG Transition Time:
Rise Time
Fall Time
(Note 6)
CLOAD = 3300pF
CLOAD = 3300pF
25
25
ns
ns
BG1,2 tr
BG1,2 tf
BG Transition Time:
Rise Time
Fall Time
(Note 6)
CLOAD = 3300pF
CLOAD = 3300pF
25
25
ns
ns
TG/BG t1D
Top Gate Off to Bottom Gate On Delay
Synchronous Switch-On Delay Time
CLOAD = 3300pF Each Driver
30
ns
BG/TG t1D
Bottom Gate Off to Top Gate On Delay
Top Switch-On Delay Time
CLOAD = 3300pF Each Driver
30
ns
tON(MIN)
Minimum On-Time
(Note 7)
95
ns
INTVCC Linear Regulator
VINTVCCVIN
Internal VCC Voltage
6V < VIN < 60V, VEXTVCC = 0V
VLDOVIN
INTVCC Load Regulation
ICC = 0mA to 50mA, VEXTVCC = 0V
VINTVCCEXT
Internal VCC Voltage
6V < VEXTVCC < 13V
4.85
4.85
5.1
5.35
V
0.7
1.1
%
5.1
5.35
V
38901f
3
LTC3890-1
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
junction temperature range, otherwise specifications are at TJ = 25°C. VIN = 12V, VRUN1,2 = 5V, EXTVCC = 0V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
VLDOEXT
INTVCC Load Regulation
ICC = 0mA to 50mA, VEXTVCC = 8.5V
VEXTVCC
EXTVCC Switchover Voltage
EXTVCC Ramping Positive
VLDOHYS
EXTVCC Hysteresis
4.5
TYP
MAX
0.6
1.1
4.7
4.9
250
UNITS
%
V
mV
Oscillator and Phase-Locked Loop
f25kΩ
Programmable Frequency
RFREQ = 25k, PLLIN/MODE = DC Voltage
f65kΩ
Programmable Frequency
RFREQ = 65k, PLLIN/MODE = DC Voltage
f105kΩ
Programmable Frequency
RFREQ = 105k, PLLIN/MODE = DC Voltage
fLOW
Low Fixed Frequency
VFREQ = 0V, PLLIN/MODE = DC Voltage
fHIGH
High Fixed Frequency
VFREQ = INTVCC, PLLIN/MODE = DC Voltage
fSYNC
Synchronizable Frequency
PLLIN/MODE = External Clock
105
kHz
375
440
505
320
350
380
kHz
485
535
585
kHz
850
kHz
835
l
75
kHz
kHz
PGOOD1 Output
VPGL
PGOOD1 Voltage Low
IPGOOD = 2mA
IPGOOD
PGOOD1 Leakage Current
VPGOOD = 5V
VPG
PGOOD1 Trip Level
VFB with Respect to Set Regulated Voltage
VFB Ramping Negative
Hysteresis
–13
VFB with Respect to Set Regulated Voltage
VFB Ramping Positive
Hysteresis
7
tPG
Delay for Reporting a Fault
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Ratings for extended periods may affect device reliability and
lifetime.
Note 2: The LTC3890E-1 is guaranteed to meet performance specifications
from 0°C to 125°C. Specifications over the –40°C to 125°C operating
junction temperature range are assured by design, characterization and
correlation with statistical process controls. The LTC3890I-1 is guaranteed
over the full –40°C to 125°C operating junction temperature range.
Note 3: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formula:
TJ = TA + (PD • 90°C/W)
0.2
0.4
V
±1
μA
–10
2.5
–7
%
%
10
2.5
13
%
%
25
μs
Note 4: The LTC3890-1 is tested in a feedback loop that servos VITH1,2 to
a specified voltage and measures the resultant VFB1,2. The specification at
85°C is not tested in production. This specification is assured by design,
characterization and correlation to production testing at 125°C.
Note 5: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. See Applications information.
Note 6: Rise and fall times are measured using 10% and 90% levels. Delay
times are measured using 50% levels.
Note 7: The minimum on-time condition is specified for an inductor peakto-peak ripple current ≥ of IMAX (See Minimum On-Time Considerations in
the Applications Information section).
38901f
4
LTC3890-1
TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency and Power Loss
vs Output Current
BURST EFFICIENCY
90
1000
CCM LOSS
100
60
50
BURST LOSS
PULSE-SKIPPING
LOSS
40
10
30
CCM EFFICIENCY
20
PULSE-SKIPPING
EFFICIENCY
10
0
0.0001
0.001
0.01
0.1
1
OUTPUT CURRENT (A)
EFFICIENCY (%)
70
98
VOUT = 3.3V
70
60
50
40
10
90
VOUT1 = 3.3V
88
82
ILOAD = 2A
80
10
0
Load Step
Pulse-Skipping Mode
VOUT
100mV/DIV
ACCOUPLED
VOUT
100mV/DIV
ACCOUPLED
IL
2A/DIV
IL
2A/DIV
IL
2A/DIV
38901 G04
50μs/DIV
VIN = 12V
VOUT = 3.3V
FIGURE 13 CIRCUIT
Inductor Current at Light Load
FORCED
CONTINUOUS
MODE
38901 G05
38901 G06
50μs/DIV
VIN = 12V
VOUT = 3.3V
FIGURE 13 CIRCUIT
VIN = 12V
VOUT = 3.3V
FIGURE 13 CIRCUIT
Soft Start-Up
Tracking Start-Up
VOUT2
2V/DIV
Burst Mode
OPERATION
1A/DIV
38901 G03
Load Step
Forced Continuous Mode
VOUT
100mV/DIV
ACCOUPLED
50μs/DIV
5 10 15 20 25 30 35 40 45 50 55 60
INPUT VOLTAGE (V)
FIGURE 13 CIRCUIT
38901 G02
FIGURE 13 CIRCUIT
Load Step
Burst Mode Operation
92
84
Burst Mode OPERATION
VIN = 12V
0
0.0001 0.001
0.01
0.1
1
OUTPUT CURRENT (A)
0.1
10
94
86
20
1
VOUT2 = 8.5V
96
30
38901 G01
FIGURE 13 CIRCUIT
VOUT = 8.5V
80
POWER LOSS (mW)
EFFICIENCY (%)
80
100
100
EFFICIENCY (%)
VIN = 12V
90 VOUT = 3.3V
Efficiency vs Input Voltage
Efficiency vs Output Current
10000
100
VOUT2
2V/DIV
VOUT1
2V/DIV
VOUT1
2V/DIV
PULSE-SKIPPING
MODE
5μs/DIV
VIN = 12V
VOUT = 3.3V
ILOAD = 200μA
38901 G07
2ms/DIV
FIGURE 13 CIRCUIT
38901 G08
2ms/DIV
FIGURE 13 CIRCUIT
38901 G09
38901f
5
LTC3890-1
TYPICAL PERFORMANCE CHARACTERISTICS
Total Input Supply Current
vs Input Voltage
EXTVCC Switchover and INTVCC
Voltages vs Temperature
INTVCC Line Regulation
5.2
6.0
300
SUPPLY CURRENT (μA)
200
300μA
150
100
NO LOAD
50
5.6
INTVCC VOLTAGE (A)
EXTVCC AND INTVCC VOLTAGE (V)
5.8
250
5.4
INTVCC
5.2
5.0
EXTVCC RISING
4.8
EXTVCC FALLING
4.6
5.1
5.0
4.9
4.4
4.2
0
4.0
–45
5 10 15 20 25 30 35 40 45 50 55 60 65
INPUT VOLTAGE (V)
4.8
–20
55
30
5
80
TEMPERATURE (°C)
38901 G10
MAXIMUM CURRENT SENSE VOLTAGE (mV)
–100
60
SENSE– CURRENT (A)
CURRENT SENSE THESHOLD (mV)
Maximum Current Sense
Threshold vs Duty Cycle
0
40
Burst Mode
OPERATION
20
0
–200
–300
–400
–500
–600
–20
–700
FORCED CONTINUOUS MODE
–40
0
0.2
0.4
0.6 0.8
VITH (V)
1.0
1.2
–800
1.4
0
5
10
15
20
INPUT VOLTAGE (V)
38901 G13
90
85
80
75
70
65
60
25
Foldback Current Limit
75
40
30
20
INTVCC vs Load Current
5.20
VIN = 12V
5.15
70
INVCC VOLTAGE (V)
QUIESCENT CURRENT (μA)
70
50
10 20 30 40 50 60 70 80 90 100
DUTY CYCLE (%)
38901 G15
Quiescent Current vs Temperature
80
60
0
38901 G14
80
65
60
55
50
5.10
EXTVCC = 0V
5.05
EXTVCC = 8.5V
5.00
45
10
0
38901 G12
SENSE– Pin Input Bias Current
80
PULSE SKIPPING MODE
5 10 15 20 25 30 35 40 45 50 55 60 65
INPUT VOLTAGE (V)
130
38901 G11
Maximum Current Sense Voltage
vs ITH Voltage
MAXIMUM CURRENT SENSE VOLTAGE (mV)
105
0
100 200 300 400 500 600
FEEDBACK VOLTAGE (MV)
700 800
3890 G16
40
–45 –20
4.95
80
55
30
TEMPERATURE (°C)
5
105
130
38901 G17
0
20 40 60 80 100 120 140 160 180 200
LOAD CURRENT (mA)
38901 G18
38901f
6
LTC3890-1
TYPICAL PERFORMANCE CHARACTERISTICS
TRACK/SS Pull-Up Current
vs Temperature
Regulated Feedback Voltage
vs Temperature
Shutdown (RUN) Threshold
vs Temperature
1.10
800
REGULATED FEEDBACK VOLTAGE (mV)
1.40
1.05
RUN PIN VOLTAGE (V)
TRACK/SS CURRENT (μA)
1.35
1.00
0.95
1.30
RUN1 RISING
RUN2 RISING
1.25
1.20
1.15
RUN1 FALLING
1.10
RUN2 FALLING
1.05
0.90
–45
–20
80
5
30
55
TEMPERATURE (°C)
105
1.00
–45 –20
130
55
30
80
5
TEMPERATURE (°C)
105
38901 G19
802
800
798
796
794
792
–45 –20
130
80
55
30
TEMPERATURE (°C)
5
105
130
38901 G21
Shutdown Current vs Input
Voltage
Oscillator Frequency
vs Temperature
30
600
25
550
FREQUENCY (kHz)
SHUTDOWN CURRENT (μA)
SENSE – CURRENT (μA)
804
38901 G20
SENSE– Pin Total Input Bias Current
vs Temperature
50
0
VOUT < INTVCC – 0.5V
–50
–100
–150
–200
–250
–300
–350
–400
–450
–500
–550
–600
VOUT > INTVCC – 0.5V
–650
–700
–750
–800
–45 –25 –5 15 35 55 75 95 115
TEMPERATURE (°C)
806
20
15
10
FREQ = INTVCC
500
450
400
FREQ = GND
350
5
300
–45 –20
0
5 10 15 20 25 30 35 40 45 50 55 60 65
INPUT VOLTAGE (V)
55
30
80
5
TEMPERATURE (°C)
105
130
38901 G22
38901 G23
Undervoltage Lockout Threshold
vs Temperature
Oscillator Frequency vs Input
Voltage
356
OSCILLATOR FREQUENCY (kHz)
4.3
INTVCC VOLTAGE (V)
4.2
RISING
4.0
3.9
FALLING
3.8
3.7
3.6
Shutdown Current vs Temperature
22
FREQ = GND
354
352
350
348
346
–20
55
30
5
80
TEMPERATURE (°C)
105
130
38901 G25
18
16
14
12
10
3.5
3.4
–45
VIN = 12V
20
SHUTDOWN CURRENT (μA)
4.4
4.1
38901 G24
344
5 10 15 20 25 30 35 40 45 50 55 60 65
INPUT VOLTAGE (V)
38901 G26
8
–45
–20
5
30
55
80
TEMPERATURE (°C)
105
130
38901 G27
38901f
7
LTC3890-1
PIN FUNCTIONS
ITH1, ITH2 (Pin 1, Pin 13): Error Amplifier Outputs and
Switching Regulator Compensation Points. Each associated channel’s current comparator trip point increases
with this control voltage.
VFB1, VFB2 (Pin 2, Pin 12): Receives the remotely sensed
feedback voltage for each controller from an external
resistive divider across the output.
SENSE1+,
SENSE2+
(Pin 3, Pin 11): The (+) input to the
differential current comparators are normally connected
to DCR sensing networks or current sensing resistors.
The ITH pin voltage and controlled offsets between the
SENSE– and SENSE+ pins in conjunction with RSENSE set
the current trip threshold.
SENSE1–, SENSE2– (Pin 4, Pin 10): The (–) Input to the
Differential Current Comparators. When greater than
INTVCC – 0.5V, the SENSE– pin supplies current to the
current comparator.
FREQ (Pin 5): The Frequency Control Pin for the Internal
VCO. Connecting the pin to GND forces the VCO to a fixed
low frequency of 350kHz. Connecting the pin to INTVCC
forces the VCO to a fixed high frequency of 535kHz.
Other frequencies between 50kHz and 900kHz can be
programmed using a resistor between FREQ and GND.
An internal 20μA pull-up current develops the voltage to
be used by the VCO to control the frequency.
PLLIN/MODE (Pin 6): External Synchronization Input to
Phase Detector and Forced Continuous Mode Input. When
an external clock is applied to this pin, the phase-locked
loop will force the rising TG1 signal to be synchronized
with the rising edge of the external clock. When not synchronizing to an external clock, this input, which acts on
both controllers, determines how the LTC3890-1 operates
at light loads. Pulling this pin to ground selects Burst Mode
operation. An internal 100k resistor to ground also invokes
Burst Mode Operation when the pin is floated. Tying this
pin to INTVCC forces continuous inductor current operation.
Tying this pin to a voltage greater than 1.2V and less than
INTVCC – 1.3V selects pulse skipping operation.
SGND (Pin 7): Small-signal ground common to both
controllers, must be routed separately from high current
grounds to the common (–) terminals of the CIN capacitors.
RUN1, RUN2 (Pin 8, Pin 9): Digital Run Control Inputs for
Each Controller. Forcing either of these pins below 1.2V
shuts down that controller. Forcing both of these pins
below 0.7V shuts down the entire LTC3890-1, reducing
quiescent current to approximately 14μA.
INTVCC (Pin 19): Output of the Internal Linear Low Dropout
Regulator. The driver and control circuits are powered
from this voltage source. Must be decoupled to power
ground with a minimum of 4.7μF ceramic or other low
ESR capacitor. Do not use the INTVCC pin for any other
purpose.
EXTVCC (Pin 20): External Power Input to an Internal LDO
Connected to INTVCC. This LDO supplies INTVCC power,
bypassing the internal LDO powered from VIN whenever
EXTVCC is higher than 4.7V. See EXTVCC Connection in
the Applications Information section. Do not exceed 14V
on this pin.
PGND (Pin 21): Driver Power Ground. Connects to the
sources of bottom (synchronous) N-channel MOSFETs
and the (–) terminal(s) of CIN.
VIN (Pin 22): Main Supply Pin. A bypass capacitor should
be tied between this pin and the signal ground pin.
38901f
8
LTC3890-1
PIN FUNCTIONS
BG1, BG2 (Pin 23, Pin 18): High Current Gate Drives for
Bottom (Synchronous) N-Channel MOSFETs. Voltage
swing at these pins is from ground to INTVCC.
PGOOD1 (Pin 27): Open-Drain Logic Output. PGOOD1 is
pulled to ground when the voltage on the VFB1 pin is not
within ±10% of its set point.
BOOST1, BOOST2 (Pin 24, Pin 17): Bootstrapped Supplies
to the Topside Floating Drivers. Capacitors are connected
between the BOOST and SW pins and Schottky diodes are
tied between the BOOST and INTVCC pins. Voltage swing
at the BOOST pins is from INTVCC to (VIN + INTVCC).
TRACK/SS1, TRACK/SS2 (Pin 28, Pin 14): External
Tracking and Soft-Start Input. The LTC3890-1 regulates
the VFB1,2 voltage to the smaller of 0.8V or the voltage
on the TRACK/SS1,2 pin. An internal 1μA pull-up current
source is connected to this pin. A capacitor to ground at
this pin sets the ramp time to final regulated output voltage. Alternatively, a resistor divider on another voltage
supply connected to this pin allows the LTC3890-1 output
to track the other supply during start-up.
SW1, SW2 (Pin 25, Pin 16): Switch Node Connections
to Inductors.
TG1, TG2 (Pin 26, Pin 15): High Current Gate Drives for
Top N-Channel MOSFETs. These are the outputs of floating drivers with a voltage swing equal to INTVCC – 0.5V
superimposed on the switch node voltage SW.
38901f
9
LTC3890-1
FUNCTIONAL DIAGRAM
VIN
INTVCC
DUPLICATE FOR SECOND
CONTROLLER CHANNEL
PGOOD1
BOOST
DROP
OUT
DET
0.88V
+
–
VFB1
S
Q
R
Q
+
0.72V
CB
TG
TOP
D
BOT
TOP ON
SWITCH
LOGIC BOT
SHDN
CIN
SW
INTVCC
BG
COUT
–
PGND
20μA
FREQ
DB
VOUT
VCO
CLK2
0.425V
CLK1
+
SLEEP
RSENSE
L
–
ICMP
PFD
+
–
CLP
–+
+–
IR
–
SENSE+
+
3mV
SYNC
DET
PLLIN/MODE
2.7V
0.65V
SENSE–
100k
SLOPE COMP
VFB
EA
–
OV
–
5.1V
5.1V
LDO
EN
LDO
EN
7μA (RUN1)
0.5μA (RUN2) SHDN
RST
2(VFB)
+
–
0.80V
TRACK/SS
RA
+
EXTVCC
4.7V
RB
+
VIN
0.88V
CC
ITH
CC2
FOLDBACK
11V
CSS
SHDN
SGND
INTVCC
RUN
RC
1μA TRACK/SS
38901 FD
38901f
10
LTC3890-1
OPERATION (Refer to the Functional Diagram)
Main Control Loop
The LTC3890-1 uses a constant frequency, current mode
step-down architecture with the two controller channels
operating 180 degrees out of phase. During normal operation, each external top MOSFET is turned on when the
clock for that channel sets the RS latch, and is turned off
when the main current comparator, ICMP, resets the RS
latch. The peak inductor current at which ICMP trips and
resets the latch is controlled by the voltage on the ITH pin,
which is the output of the error amplifier, EA. The error
amplifier compares the output voltage feedback signal at
the VFB pin, (which is generated with an external resistor
divider connected across the output voltage, VOUT , to
ground) to the internal 0.800V reference voltage. When
the load current increases, it causes a slight decrease
in VFB relative to the reference, which causes the EA to
increase the ITH voltage until the average inductor current
matches the new load current.
After the top MOSFET is turned off each cycle, the bottom MOSFET is turned on until either the inductor current
starts to reverse, as indicated by the current comparator
IR, or the beginning of the next clock cycle.
INTVCC/EXTVCC Power
Power for the top and bottom MOSFET drivers and most
other internal circuitry is derived from the INTVCC pin.
When the EXTVCC pin is left open or tied to a voltage less
than 4.7V, the VIN LDO (low dropout linear regulator) supplies 5.1V from VIN to INTVCC. If EXTVCC is taken above
4.7V, the VIN LDO is turned off and an EXTVCC LDO is
turned on. Once enabled, the EXTVCC LDO supplies 5.1V
from EXTVCC to INTVCC. Using the EXTVCC pin allows
the INTVCC power to be derived from a high efficiency
external source such as one of the LTC3890-1 switching
regulator outputs.
Each top MOSFET driver is biased from the floating bootstrap capacitor CB, which normally recharges during each
cycle through an external diode when the top MOSFET
turns off. If the input voltage, VIN, decreases to a voltage
close to VOUT , the loop may enter dropout and attempt
to turn on the top MOSFET continuously. The dropout
detector detects this and forces the top MOSFET off for
about one-twelfth of the clock period every tenth cycle to
allow CB to recharge.
Shutdown and Start-Up (RUN1, RUN2 and
TRACK/ SS1, TRACK/SS2 Pins)
The two channels of the LTC3890-1 can be independently
shut down using the RUN1 and RUN2 pins. Pulling either
of these pins below 1.2V shuts down the main control loop
for that controller. Pulling both pins below 0.7V disables
both controllers and most internal circuits, including the
INTVCC LDOs. In this state, the LTC3890-1 draws only
14μA of quiescent current.
Releasing either RUN pin allows a small internal current to
pull up the pin to enable that controller. The RUN1 pin has a
7μA pull-up current while the RUN2 pin has a smaller 0.5μA.
The 7μA current on RUN1 is designed to be large enough
so that the RUN1 pin can be safely floated (to always enable the controller) without worry of condensation or other
small board leakage pulling the pin down. This is ideal for
always-on applications where one or both controllers are
enabled continuously and never shut down.
The RUN pin may be externally pulled up or driven directly
by logic. When driving the RUN pin with a low impedance
source, do not exceed the absolute maximum rating of
8V. The RUN pin has an internal 11V voltage clamp that
allows the RUN pin to be connected through a resistor to a
higher voltage (for example, VIN), so long as the maximum
current into the RUN pin does not exceed 100μA.
The start-up of each controller’s output voltage VOUT is
controlled by the voltage on the TRACK/SS pin for that
channel. When the voltage on the TRACK/SS pin is less
than the 0.8V internal reference, the LTC3890-1 regulates
the VFB voltage to the TRACK/SS pin voltage instead of
the 0.8V reference. This allows the TRACK/SS pin to be
used to program a soft-start by connecting an external
capacitor from the TRACK/SS pin to SGND. An internal
1μA pull-up current charges this capacitor creating a
voltage ramp on the TRACK/SS pin. As the TRACK/SS
voltage rises linearly from 0V to 0.8V (and beyond up to
5V), the output voltage VOUT rises smoothly from zero to
its final value.
Alternatively the TRACK/SS pin can be used to cause the
start-up of VOUT to track that of another supply. Typically,
this requires connecting to the TRACK/SS pin an external
resistor divider from the other supply to ground (see Applications Information section).
38901f
11
LTC3890-1
OPERATION (Refer to the Functional Diagram)
Light Load Current Operation (Burst Mode Operation,
Pulse Skipping, or Forced Continuous Mode)
(PLLIN/MODE Pin)
The LTC3890-1 can be enabled to enter high efficiency
Burst Mode operation, constant frequency pulse skipping
mode, or forced continuous conduction mode at low load
currents. To select Burst Mode operation, tie the PLLIN/
MODE pin to a DC voltage below 0.8V (e.g., SGND). To
select forced continuous operation, tie the PLLIN/MODE
pin to INTVCC. To select pulse-skipping mode, tie the
PLLIN/MODE pin to a DC voltage greater than 1.2V and
less than INTVCC – 1.3V.
When a controller is enabled for Burst Mode operation, the
minimum peak current in the inductor is set to approximately 25% of the maximum sense voltage even though
the voltage on the ITH pin indicates a lower value. If the
average inductor current is higher than the load current,
the error amplifier, EA, will decrease the voltage on the
ITH pin. When the ITH voltage drops below 0.425V, the
internal sleep signal goes high (enabling sleep mode)
and both external MOSFETs are turned off. The ITH pin is
then disconnected from the output of the EA and parked
at 0.450V.
In sleep mode, much of the internal circuitry is turned off,
reducing the quiescent current that the LTC3890-1 draws.
If one channel is shut down and the other channel is in
sleep mode, the LTC3890-1 draws only 50μA of quiescent
current. If both channels are in sleep mode, the LTC3890-1
draws only 60μA of quiescent current. In sleep mode,
the load current is supplied by the output capacitor. As
the output voltage decreases, the EA’s output begins to
rise. When the output voltage drops enough, the ITH pin
is reconnected to the output of the EA, the sleep signal
goes low, and the controller resumes normal operation
by turning on the top external MOSFET on the next cycle
of the internal oscillator.
When a controller is enabled for Burst Mode operation,
the inductor current is not allowed to reverse. The reverse
current comparator, IR, turns off the bottom external
MOSFET just before the inductor current reaches zero,
preventing it from reversing and going negative. Thus,
the controller operates in discontinuous operation.
In forced continuous operation or clocked by an external
clock source to use the phase-locked loop (see Frequency
Selection and Phase-Locked Loop section), the inductor current is allowed to reverse at light loads or under
large transient conditions. The peak inductor current
is determined by the voltage on the ITH pin, just as in
normal operation. In this mode, the efficiency at light
loads is lower than in Burst Mode operation. However,
continuous operation has the advantage of lower output
voltage ripple and less interference to audio circuitry. In
forced continuous mode, the output ripple is independent
of load current.
When the PLLIN/MODE pin is connected for pulse skipping
mode, the LTC3890-1 operates in PWM pulse skipping
mode at light loads. In this mode, constant frequency
operation is maintained down to approximately 1% of
designed maximum output current. At very light loads, the
current comparator, ICMP, may remain tripped for several
cycles and force the external top MOSFET to stay off for
the same number of cycles (i.e., skipping pulses). The
inductor current is not allowed to reverse (discontinuous
operation). This mode, like forced continuous operation,
exhibits low output ripple as well as low audio noise and
reduced RF interference as compared to Burst Mode
operation. It provides higher low current efficiency than
forced continuous mode, but not nearly as high as Burst
Mode operation.
Frequency Selection and Phase-Locked Loop
(FREQ and PLLIN/MODE Pins)
The selection of switching frequency is a trade-off between
efficiency and component size. Low frequency operation
increases efficiency by reducing MOSFET switching
losses, but requires larger inductance and/or capacitance
to maintain low output ripple voltage.
The switching frequency of the LTC3890-1’s controllers
can be selected using the FREQ pin.
38901f
12
LTC3890-1
OPERATION (Refer to the Functional Diagram)
If the PLLIN/MODE pin is not being driven by an external
clock source, the FREQ pin can be tied to SGND, tied to
INTVCC or programmed through an external resistor.
Tying FREQ to SGND selects 350kHz while tying FREQ
to INTVCC selects 535kHz. Placing a resistor between
FREQ and SGND allows the frequency to be programmed
between 50kHz and 900kHz, as shown in Figure 10.
A phase-locked loop (PLL) is available on the LTC3890-1
to synchronize the internal oscillator to an external clock
source that is connected to the PLLIN/MODE pin. The
LTC3890-1’s phase detector adjusts the voltage (through
an internal lowpass filter) of the VCO input to align the
turn-on of controller 1’s external top MOSFET to the rising edge of the synchronizing signal. Thus, the turn-on of
controller 2’s external top MOSFET is 180 degrees out of
phase to the rising edge of the external clock source.
The VCO input voltage is prebiased to the operating frequency set by the FREQ pin before the external clock is
applied. A resistor connected between the FREQ pin and
SGND can prebias VCO’s input voltage to the desired
frequency. If prebiased near the external clock frequency,
the PLL loop only needs to make slight changes to the
VCO input in order to synchronize the rising edge of the
external clock’s to the rising edge of TG1. The ability to
prebias the loop filter allows the PLL to lock-in rapidly
without deviating far from the desired frequency.
The typical capture range of the phase-locked loop is
from approximately 55kHz to 1MHz, with a guarantee
to be between 75kHz and 850kHz. In other words, the
LTC3890-1’s PLL is guaranteed to lock to an external clock
source whose frequency is between 75kHz and 850kHz.
The typical input clock thresholds on the PLLIN/MODE
pin are 1.6V (rising) and 1.2V (falling).
Output Overvoltage Protection
An overvoltage comparator guards against transient overshoots as well as other more serious conditions that may
overvoltage the output. When the VFB pin rises by more
than 10% above its regulation point of 0.800V, the top
MOSFET is turned off and the bottom MOSFET is turned
on until the overvoltage condition is cleared.
Power Good (PGOOD1 Pin)
The PGOOD1 pin is connected to an open drain of an internal
N-channel MOSFET. The MOSFET turns on and pulls the
PGOOD1 pin low when the corresponding VFB1 pin voltage is not within ±10% of the 0.8V reference voltage. The
PGOOD1 pin is also pulled low when the corresponding
RUN1 pin is low (shut down). When the VFB1 pin voltage
is within the ±10% requirement, the MOSFET is turned
off and the pin is allowed to be pulled up by an external
resistor to a source no greater than 6V.
Foldback Current
When the output voltage falls to less than 70% of its
nominal level, foldback current limiting is activated, progressively lowering the peak current limit in proportion to
the severity of the overcurrent or short-circuit condition.
Foldback current limiting is disabled during the soft-start
interval (as long as the VFB voltage is keeping up with the
TRACK/SS voltage).
Theory and Benefits of 2-Phase Operation
Why the need for 2-phase operation? Up until the 2-phase
family, constant-frequency dual switching regulators
operated both channels in phase (i.e., single phase
operation). This means that both switches turned on at
the same time, causing current pulses of up to twice the
amplitude of those for one regulator to be drawn from the
input capacitor and battery. These large amplitude current
pulses increased the total RMS current flowing from the
input capacitor, requiring the use of more expensive input
capacitors and increasing both EMI and losses in the input
capacitor and battery.
With 2-phase operation, the two channels of the dual
switching regulator are operated 180 degrees out of phase.
This effectively interleaves the current pulses drawn by the
switches, greatly reducing the overlap time where they add
38901f
13
LTC3890-1
OPERATION (Refer to the Functional Diagram)
Figure 1 compares the input waveforms for a representative
single-phase dual switching regulator to the LTC3890-1
2-phase dual switching regulator. An actual measurement of the RMS input current under these conditions
shows that 2-phase operation dropped the input current
from 2.53ARMS to 1.55ARMS. While this is an impressive
reduction in itself, remember that the power losses are
proportional to IRMS2, meaning that the actual power
wasted is reduced by a factor of 2.66. The reduced input
ripple voltage also means less power is lost in the input
power path, which could include batteries, switches,
trace/connector resistances and protection circuitry.
Improvements in both conducted and radiated EMI also
directly accrue as a result of the reduced RMS input current and voltage.
It can readily be seen that the advantages of 2-phase
operation are not just limited to a narrow operating range,
for most applications is that 2-phase operation will reduce
the input capacitor requirement to that for just one channel
operating at maximum current and 50% duty cycle.
The schematic on the first page is a basic LTC3890-1 application circuit. External component selection is driven
by the load requirement, and begins with the selection of
RSENSE and the inductor value. Next, the power MOSFETs
are selected. Finally, CIN and COUT are selected
Of course, the improvement afforded by 2-phase operation is a function of the dual switching regulator’s relative
duty cycles which, in turn, are dependent upon the input
voltage VIN (Duty Cycle = VOUT/VIN). Figure 2 shows how
the RMS input current varies for single phase and 2-phase
operation for 3.3V and 5V regulators over a wide input
voltage range.
3.0
SINGLE PHASE
DUAL CONTROLLER
2.5
INPUT RMS CURRENT (A)
together. The result is a significant reduction in total RMS
input current, which in turn allows less expensive input
capacitors to be used, reduces shielding requirements for
EMI and improves real world operating efficiency.
2.0
1.5
2-PHASE
DUAL CONTROLLER
1.0
0.5
0
VO1 = 5V/3A
VO2 = 3.3V/3A
0
10
20
30
INPUT VOLTAGE (V)
40
38901 F02
Figure 2. RMS Input Current Comparison
5V SWITCH
20V/DIV
3.3V SWITCH
20V/DIV
INPUT CURRENT
5A/DIV
INPUT VOLTAGE
500mV/DIV
IIN(MEAS) = 2.53ARMS
IIN(MEAS) = 1.55ARMS
38901 F01
Figure 1. Input Waveforms Comparing Single-Phase (a) and 2-Phase (b) Operation for Dual Switching Regulators
Converting 12V to 5V and 3.3V at 3A Each. The Reduced Input Ripple with the 2-Phase Regulator Allows
Less Expensive Input Capacitors, Reduces Shielding Requirements for EMI and Improves Efficiency
38901f
14
LTC3890-1
APPLICATIONS INFORMATION
The Typical Application on the first page is a basic
LTC3890-1 application circuit. LTC3890-1 can be configured to use either DCR (inductor resistance) sensing or low
value resistor sensing. The choice between the two current
sensing schemes is largely a design trade-off between
cost, power consumption, and accuracy. DCR sensing
is becoming popular because it saves expensive current
sensing resistors and is more power efficient, especially
in high current applications. However, current sensing
resistors provide the most accurate current limits for the
controller. Other external component selection is driven
by the load requirement, and begins with the selection of
RSENSE (if RSENSE is used) and inductor value. Next, the
power MOSFETs and Schottky diodes are selected. Finally,
input and output capacitors are selected.
SENSE+ and SENSE– Pins
SENSE+
SENSE–
and
pins are the inputs to the curThe
rent comparators. The common mode voltage range on
these pins is 0V to 28V (abs max), enabling the LTC3857
to regulate output voltages up to a nominal 24V (allowing
margin for tolerances and transients).
programmed current limit unpredictable. If inductor DCR
sensing is used (Figure 4b), sense resistor R1 should be
placed close to the switching node, to prevent noise from
coupling into sensitive small-signal nodes.
TO SENSE FILTER,
NEXT TO THE CONTROLLER
COUT
38901 F03
INDUCTOR OR RSENSE
Figure 3. Sense Lines Placement with Inductor or Sense Resistor
VIN
INTVCC
BOOST
TG
VOUT
LTC3890-1
BG
SENSE+
PLACE CAPACITOR NEAR
SENSE PINS
SENSE–
SGND
38901 F04a
mode range, drawing at most ±1μA. This high impedance
allows the current comparators to be used in inductor
DCR sensing.
Filter components mutual to the sense lines should be
placed close to the LTC3890-1, and the sense lines should
run close together to a Kelvin connection underneath the
current sense element (shown in Figure 3). Sensing current elsewhere can effectively add parasitic inductance
and capacitance to the current sense element, degrading
the information at the sense terminals and making the
RSENSE
SW
The SENSE+ pin is high impedance over the full common
The impedance of the SENSE– pin changes depending on
the common mode voltage. When SENSE– is less than
INTVCC – 0.5V, a small current of less than 1μA flows out
of the pin. When SENSE– is above INTVCC + 0.5V, a higher
current (~700μA) flows into the pin. Between INTVCC
– 0.5V and INTVCC + 0.5V, the current transitions from
the smaller current to the higher current.
VIN
(4a) Using a Resistor to Sense Current
VIN
INTVCC
VIN
BOOST
INDUCTOR
TG
L
DCR
VOUT
SW
LTC3890-1
BG
R1
SENSE+
C1*
R2
SENSE–
SGND
*PLACE C1 NEAR
SENSE PINS
(R1||R2) • C1 =
L
DCR
RSENSE(EQ) = DCR
R2
R1 + R2
38901 F04b
(4b) Using the Inductor DCR to Sense Current
Figure 4. Current Sensing Methods
38901f
15
LTC3890-1
APPLICATIONS INFORMATION
Low Value Resistor Current Sensing
A typical sensing circuit using a discrete resistor is shown
in Figure 4a. RSENSE is chosen based on the required
output current.
The current comparator has a maximum threshold
VSENSE(MAX). The current comparator threshold voltage
sets the peak of the inductor current, yielding a maximum
average output current, IMAX, equal to the peak value less
half the peak-to-peak ripple current, ΔIL. To calculate the
sense resistor value, use the equation:
RSENSE =
VSENSE(MAX)
IMAX +
ΔIL
2
To ensure that the application will deliver full load current
over the full operating temperature range, choose the
minimum value for the Maximum Current Sense Threshold
(VSENSE(MAX)).
When using the controller in very low dropout conditions,
the maximum output current level will be reduced due
to the internal compensation required to meet stability
criterion for buck regulators operating at greater than
50% duty factor. A curve is provided in the Typical Performance Characteristics section to estimate this reduction
in peak inductor current depending upon the operating
duty factor.
Inductor DCR Sensing
For applications requiring the highest possible efficiency
at high load currents, the LTC3890-1 is capable of sensing
the voltage drop across the inductor DCR, as shown in
Figure 4b. The DCR of the inductor represents the small
amount of DC resistance of the copper wire, which can be
less than 1mΩ for today’s low value, high current inductors.
In a high current application requiring such an inductor,
power loss through a sense resistor would cost several
points of efficiency compared to inductor DCR sensing.
If the external (R1||R2) • C1 time constant is chosen to be
exactly equal to the L/DCR time constant, the voltage drop
across the external capacitor is equal to the drop across
the inductor DCR multiplied by R2/(R1 + R2). R2 scales the
voltage across the sense terminals for applications where
the DCR is greater than the target sense resistor value.
To properly dimension the external filter components, the
DCR of the inductor must be known. It can be measured
using a good RLC meter, but the DCR tolerance is not
always the same and varies with temperature; consult the
manufacturers’ data sheets for detailed information.
Using the inductor ripple current value from the Inductor
Value Calculation section, the target sense resistor value
is:
RSENSE(EQUIV) =
VSENSE(MAX)
IMAX +
ΔIL
2
To ensure that the application will deliver full load current
over the full operating temperature range, choose the
minimum value for the Maximum Current Sense Threshold
(VSENSE(MAX)).
Next, determine the DCR of the inductor. When provided,
use the manufacturer’s maximum value, usually given at
20°C. Increase this value to account for the temperature
coefficient of copper resistance, which is approximately
0.4%/°C. A conservative value for TL(MAX) is 100°C.
To scale the maximum inductor DCR to the desired sense
resistor value (RD), use the divider ratio:
RD =
RSENSE(EQUIV )
DCRMAX at TL(MAX )
C1 is usually selected to be in the range of 0.1μF to 0.47μF.
This forces R1|| R2 to around 2k, reducing error that might
have been caused by the SENSE+ pin’s ±1μA current.
38901f
16
LTC3890-1
APPLICATIONS INFORMATION
The equivalent resistance R1|| R2 is scaled to the room
temperature inductance and maximum DCR:
R1|| R2 =
L
DCR at 20°C • C1
(
)
The sense resistor values are:
R1 =
R1 • RD
R1|| R2
; R2 =
RD
1 – RD
The maximum power loss in R1 is related to duty cycle,
and will occur in continuous mode at the maximum input
voltage:
PLOSS R1 =
( VIN(MAX) – VOUT ) • VOUT
R1
Ensure that R1 has a power rating higher than this value.
If high efficiency is necessary at light loads, consider this
power loss when deciding whether to use DCR sensing or
sense resistors. Light load power loss can be modestly
higher with a DCR network than with a sense resistor, due
to the extra switching losses incurred through R1. However,
DCR sensing eliminates a sense resistor, reduces conduction losses and provides higher efficiency at heavy loads.
Peak efficiency is about the same with either method.
Inductor Value Calculation
The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the use of
smaller inductor and capacitor values. So why would anyone
ever choose to operate at lower frequencies with larger
components? The answer is efficiency. A higher frequency
generally results in lower efficiency because of MOSFET
switching and gate charge losses. In addition to this basic
trade-off, the effect of inductor value on ripple current and
low current operation must also be considered.
The inductor value has a direct effect on ripple current. The
inductor ripple current, ΔIL, decreases with higher inductance or higher frequency and increases with higher VIN:
IL =
V 1
VOUT 1– OUT ( f) (L) VIN Accepting larger values of ΔIL allows the use of low
inductances, but results in higher output voltage ripple
and greater core losses. A reasonable starting point for
setting ripple current is ΔIL =0.3(IMAX). The maximum
ΔIL occurs at the maximum input voltage.
The inductor value also has secondary effects. The transition to Burst Mode operation begins when the average
inductor current required results in a peak current below
25% of the current limit determined by RSENSE. Lower
inductor values (higher ΔIL) will cause this to occur at
lower load currents, which can cause a dip in efficiency in
the upper range of low current operation. In Burst Mode
operation, lower inductance values will cause the burst
frequency to decrease.
Inductor Core Selection
Once the value for L is known, the type of inductor must
be selected. High efficiency converters generally cannot
afford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite or molypermalloy
cores. Actual core loss is independent of core size for a
fixed inductor value, but it is very dependent on inductance
value selected. As inductance increases, core losses go
down. Unfortunately, increased inductance requires more
turns of wire and therefore copper losses will increase.
Ferrite designs have very low core loss and are preferred
for high switching frequencies, so design goals can
concentrate on copper loss and preventing saturation.
Ferrite core material saturates hard, which means that
inductance collapses abruptly when the peak design
current is exceeded. This results in an abrupt increase
in inductor ripple current and consequent output voltage
ripple. Do not allow the core to saturate!
Power MOSFET and Schottky Diode
(Optional) Selection
Two external power MOSFETs must be selected for each
controller in the LTC3890-1: one N-channel MOSFET for
the top (main) switch, and one N-channel MOSFET for the
bottom (synchronous) switch.
38901f
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LTC3890-1
APPLICATIONS INFORMATION
The peak-to-peak drive levels are set by the INTVCC
voltage. This voltage is typically 5.1V during start-up
(see EXTVCC Pin Connection). Consequently, logic-level
threshold MOSFETs must be used in most applications.
Pay close attention to the BVDSS specification for the
MOSFETs as well.
Selection criteria for the power MOSFETs include the
on-resistance, RDS(ON), Miller capacitance, CMILLER, input
voltage and maximum output current. Miller capacitance,
CMILLER, can be approximated from the gate charge curve
usually provided on the MOSFET manufacturers’ data
sheet. CMILLER is equal to the increase in gate charge
along the horizontal axis while the curve is approximately
flat divided by the specified change in VDS. This result is
then multiplied by the ratio of the application applied VDS
to the Gate charge curve specified VDS. When the IC is
operating in continuous mode the duty cycles for the top
and bottom MOSFETs are given by:
Main Switch Duty Cycle =
VOUT
VIN
Synchronous Switch Duty Cycle =
VIN − VOUT
VIN
The MOSFET power dissipations at maximum output
current are given by:
PMAIN =
PSYNC =
2
VOUT
IMAX ) (1+ ) RDS(ON) +
(
VIN
2 I
( VIN) MAX
(RDR ) (CMILLER ) •
2
1 1
+
( f)
VINTVCC – VTHMIN VTHMIN 2
VIN – VOUT
IMAX ) (1+ ) RDS(ON)
(
VIN
where δ is the temperature dependency of RDS(ON) and
RDR (approximately 2Ω) is the effective driver resistance
at the MOSFET’s Miller threshold voltage. V THMIN is the
typical MOSFET minimum threshold voltage.
Both MOSFETs have I2R losses while the topside N-channel
equation includes an additional term for transition losses,
which are highest at high input voltages. For VIN < 20V
the high current efficiency generally improves with larger
MOSFETs, while for VIN > 20V the transition losses rapidly
increase to the point that the use of a higher RDS(ON) device
with lower CMILLER actually provides higher efficiency. The
synchronous MOSFET losses are greatest at high input
voltage when the top switch duty factor is low or during
a short-circuit when the synchronous switch is on close
to 100% of the period.
The term (1+ δ) is generally given for a MOSFET in the
form of a normalized RDS(ON) vs Temperature curve, but
δ = 0.005/°C can be used as an approximation for low
voltage MOSFETs.
The optional Schottky diodes D3 and D4 shown in
Figure 11 conduct during the dead-time between the
conduction of the two power MOSFETs. This prevents
the body diode of the bottom MOSFET from turning on,
storing charge during the dead-time and requiring a
reverse recovery period that could cost as much as 3%
in efficiency at high VIN. A 1A to 3A Schottky is generally
a good compromise for both regions of operation due
to the relatively small average current. Larger diodes
result in additional transition losses due to their larger
junction capacitance.
CIN and COUT Selection
The selection of CIN is simplified by the 2-phase architecture and its impact on the worst-case RMS current drawn
through the input network (battery/fuse/capacitor). It can
be shown that the worst-case capacitor RMS current occurs
when only one controller is operating. The controller with
the highest (VOUT)(IOUT) product needs to be used in the
formula shown in Equation 1 to determine the maximum
38901f
18
LTC3890-1
APPLICATIONS INFORMATION
RMS capacitor current requirement. Increasing the output current drawn from the other controller will actually
decrease the input RMS ripple current from its maximum
value. The out-of-phase technique typically reduces the
input capacitor’s RMS ripple current by a factor of 30%
to 70% when compared to a single phase power supply
solution.
In continuous mode, the source current of the top MOSFET
is a square wave of duty cycle (VOUT)/(VIN). To prevent
large voltage transients, a low ESR capacitor sized for the
maximum RMS current of one channel must be used. The
maximum RMS capacitor current is given by:
CIN Required IRMS ≈
1/ 2
IMAX ⎡
VOUT ) ( VIN – VOUT )⎤⎦ (1)
(
⎣
VIN
This formula has a maximum at VIN = 2VOUT , where IRMS
= IOUT/2. This simple worst-case condition is commonly
used for design because even significant deviations do not
offer much relief. Note that capacitor manufacturers’ ripple
current ratings are often based on only 2000 hours of life.
This makes it advisable to further derate the capacitor, or
to choose a capacitor rated at a higher temperature than
required. Several capacitors may be paralleled to meet
size or height requirements in the design. Due to the high
operating frequency of the LTC3890-1, ceramic capacitors
can also be used for CIN. Always consult the manufacturer
if there is any question.
The benefit of the LTC3890-1 2-phase operation can be
calculated by using Equation 1 for the higher power controller and then calculating the loss that would have resulted
if both controller channels switched on at the same time.
The total RMS power lost is lower when both controllers
are operating due to the reduced overlap of current pulses
required through the input capacitor’s ESR. This is why
the input capacitor’s requirement calculated above for the
worst-case controller is adequate for the dual controller
design. Also, the input protection fuse resistance, battery
resistance, and PC board trace resistance losses are also
reduced due to the reduced peak currents in a 2-phase
system. The overall benefit of a multiphase design will
only be fully realized when the source impedance of the
power supply/battery is included in the efficiency testing.
The drains of the top MOSFETs should be placed within
1cm of each other and share a common CIN(s). Separating
the drains and CIN may produce undesirable voltage and
current resonances at VIN.
A small (0.1μF to 1μF) bypass capacitor between the chip
VIN pin and ground, placed close to the LTC3890-1, is
also suggested. A 10Ω resistor placed between CIN (C1)
and the VIN pin provides further isolation between the
two channels.
The selection of COUT is driven by the effective series
resistance (ESR). Typically, once the ESR requirement
is satisfied, the capacitance is adequate for filtering. The
output ripple (ΔVOUT) is approximated by:
1
VOUT IL ESR +
8 • f • COUT where f is the operating frequency, COUT is the output
capacitance and ΔIL is the ripple current in the inductor.
The output ripple is highest at maximum input voltage
since ΔIL increases with input voltage.
Setting Output Voltage
The LTC3890-1 output voltages are each set by an external feedback resistor divider carefully placed across the
output, as shown in Figure 5. The regulated output voltage
is determined by:
R VOUT = 0.8V 1+ B RA To improve the frequency response, a feedforward capacitor, CFF , may be used. Great care should be taken to
route the VFB line away from noise sources, such as the
inductor or the SW line.
VOUT
1/2 LTC3890-1
RB
CFF
VFB
RA
38901 F05
Figure 5. Setting Output Voltage
38901f
19
LTC3890-1
APPLICATIONS INFORMATION
Tracking and Soft-Start (TRACK/SS Pins)
Soft-start is enabled by simply connecting a capacitor
from the TRACK/SS pin to ground, as shown in Figure 6.
An internal 1μA current source charges the capacitor,
providing a linear ramping voltage at the TRACK/SS pin.
The LTC3890-1 will regulate the VFB pin (and hence VOUT)
according to the voltage on the TRACK/SS pin, allowing
VOUT to rise smoothly from 0V to its final regulated value.
The total soft-start time will be approximately:
tSS = CSS •
OUTPUT VOLTAGE
VX(MASTER)
VOUT(SLAVE)
38901 F07a
TIME
(7a) Coincident Tracking
VX(MASTER)
OUTPUT VOLTAGE
The start-up of each VOUT is controlled by the voltage on
the respective TRACK/SS pin. When the voltage on the
TRACK/SS pin is less than the internal 0.8V reference, the
LTC3890-1 regulates the VFB pin voltage to the voltage on
the TRACK/SS pin instead of 0.8V. The TRACK/SS pin can
be used to program an external soft-start function or to
allow VOUT to track another supply during start-up.
VOUT(SLAVE)
0.8V
1μA
38901 F07b
TIME
1/2 LTC3890-1
TRACK/SS
(7b) Ratiometric Tracking
CSS
Figure 7. Two Different Modes of Output Voltage Tracking
SGND
38901 F06
Vx VOUT
Figure 6. Using the TRACK/SS Pin to Program Soft-Start
Alternatively, the TRACK/SS pin can be used to track two
(or more) supplies during start-up, as shown qualitatively
in Figures 7a and 7b. To do this, a resistor divider should
be connected from the master supply (V X) to the TRACK/
SS pin of the slave supply (VOUT), as shown in Figure 8.
During start-up VOUT will track V X according to the ratio
set by the resistor divider:
+ RTRACKB
R
VX
RA
=
• TRACKA
VOUT RTRACKA
RA + RB
For coincident tracking (VOUT = V X during start-up):
RA = RTRACKA
RB = RTRACKB
RB
1/2 LTC3890-1
VFB
RA
RTRACKB
TRACK/SS
RTRACKA
38901 F08
Figure 8. Using the TRACK/SS Pin for Tracking
INTVCC Regulators
The LTC3890-1 features two separate internal P-channel
low dropout linear regulators (LDO) that supply power
at the INTVCC pin from either the VIN supply pin or the
EXTVCC pin depending on the connection of the EXTVCC
pin. INTVCC powers the gate drivers and much of the
LTC3890-1’s internal circuitry. The VIN LDO and the EXTVCC
38901f
20
LTC3890-1
APPLICATIONS INFORMATION
LDO regulate INTVCC to 5.1V. Each of these can supply a
peak current of 50mA and must be bypassed to ground
with a minimum of 4.7μF ceramic capacitor. No matter
what type of bulk capacitor is used, an additional 1μF
ceramic capacitor placed directly adjacent to the INTVCC
and PGND IC pins is highly recommended. Good bypassing
is needed to supply the high transient currents required
by the MOSFET gate drivers and to prevent interaction
between the channels.
High input voltage applications in which large MOSFETs
are being driven at high frequencies may cause the maximum junction temperature rating for the LTC3890-1 to
be exceeded. The INTVCC current, which is dominated by
the gate charge current, may be supplied by either the VIN
LDO or the EXTVCC LDO. When the voltage on the EXTVCC pin is less than 4.7V, the VIN LDO is enabled. Power
dissipation for the IC in this case is highest and is equal
to VIN • IINTVCC. The gate charge current is dependent
on operating frequency as discussed in the Efficiency
Considerations section. The junction temperature can be
estimated by using the equations given in Note 3 of the
Electrical Characteristics. For example, the LTC3890-1
INTVCC current is limited to less than 15mA from a 40V
supply when not using the EXTVCC supply at 70°C ambient temperature:
TJ = 70°C + (15mA)(40V)(90°C/W) = 125°C
To prevent the maximum junction temperature from being exceeded, the input supply current must be checked
while operating in forced continuous mode (PLLIN/MODE
= INTVCC) at maximum VIN.
When the voltage applied to EXTVCC rises above 4.7V, the
VIN LDO is turned off and the EXTVCC LDO is enabled. The
EXTVCC LDO remains on as long as the voltage applied to
EXTVCC remains above 4.5V. The EXTVCC LDO attempts
to regulate the INTVCC voltage to 5.1V, so while EXTVCC
is less than 5.1V, the LDO is in dropout and the INTVCC
voltage is approximately equal to EXTVCC. When EXTVCC
is greater than 5.1V, up to an absolute maximum of 14V,
INTVCC is regulated to 5.1V.
Using the EXTVCC LDO allows the MOSFET driver and
control power to be derived from one of the LTC3890-1’s
switching regulator outputs (4.7V ≤ VOUT ≤ 14V) during
normal operation and from the VIN LDO when the output is out of regulation (e.g., start-up, short-circuit). If
more current is required through the EXTVCC LDO than
is specified, an external Schottky diode can be added
between the EXTVCC and INTVCC pins. In this case, do
not apply more than 6V to the EXTVCC pin and make sure
that EXTVCC ≤ VIN.
Significant efficiency and thermal gains can be realized
by powering INTVCC from the output, since the VIN current resulting from the driver and control currents will be
scaled by a factor of (Duty Cycle)/(Switcher Efficiency).
For 5V to 14V regulator outputs, this means connecting
the EXTVCC pin directly to VOUT . Tying the EXTVCC pin to
an 8.5V supply reduces the junction temperature in the
previous example from 125°C to:
TJ = 70°C + (15mA)(8.5V)(90°C/W) = 82°C
However, for 3.3V and other low voltage outputs, additional circuitry is required to derive INTVCC power from
the output.
The following list summarizes the four possible connections for EXTVCC:
1. EXTVCC Left Open (or Grounded). This will cause
INTVCC to be powered from the internal 5.1V regulator
resulting in an efficiency penalty of up to 10% at high
input voltages.
2. EXTVCC Connected directly to VOUT . This is the normal
connection for a 5V to 14V regulator and provides the
highest efficiency.
3. EXTVCC Connected to an External supply. If an external supply is available in the 5V to 14V range, it may
be used to power EXTVCC providing it is compatible
with the MOSFET gate drive requirements. Ensure that
EXTVCC < VIN.
4. EXTVCC Connected to an Output-Derived Boost Network.
For 3.3V and other low voltage regulators, efficiency
gains can still be realized by connecting EXTVCC to an
output-derived voltage that has been boosted to greater
than 4.7V. This can be done with the capacitive charge
pump shown in Figure 9. Ensure that EXTVCC < VIN.
38901f
21
LTC3890-1
APPLICATIONS INFORMATION
CIN
BAT85
VIN
BAT85
MTOP
NDS7002
TG1
1/2 LTC3890-1
EXTVCC
L
BAT85
RSENSE
VOUT
SW
MBOT
BG1
D
PGND
COUT
38901 F09
Figure 9. Capacitive Charge Pump for EXTVCC
Topside MOSFET Driver Supply (CB, DB)
External bootstrap capacitors, CB, connected to the BOOST
pins supply the gate drive voltages for the topside MOSFETs.
Capacitor CB in the Functional Diagram is charged though
external diode DB from INTVCC when the SW pin is low.
When one of the topside MOSFETs is to be turned on, the
driver places the CB voltage across the gate-source of the
desired MOSFET. This enhances the top MOSFET switch
and turns it on. The switch node voltage, SW, rises to VIN
and the BOOST pin follows. With the topside MOSFET on,
the boost voltage is above the input supply: VBOOST = VIN
+ VINTVCC. The value of the boost capacitor, CB, needs
to be 100 times that of the total input capacitance of the
topside MOSFET(s). The reverse breakdown of the external
Schottky diode must be greater than VIN(MAX).
When adjusting the gate drive level, the final arbiter is the
total input current for the regulator. If a change is made
and the input current decreases, then the efficiency has
improved. If there is no change in input current, then there
is no change in efficiency.
Fault Conditions: Current Limit and Current Foldback
The LTC3890-1 includes current foldback to help limit
load current when the output is shorted to ground. If the
output voltage falls below 70% of its nominal output level,
then the maximum sense voltage is progressively lowered
from 100% to 45% of its maximum selected value. Under
short-circuit conditions with very low duty cycles, the
LTC3890-1 will begin cycle skipping in order to limit the
short-circuit current. In this situation the bottom MOSFET
will be dissipating most of the power but less than in normal
operation. The short-circuit ripple current is determined by
the minimum on-time. tON(MIN), of the LTC3890-1 (≈90ns),
the input voltage and inductor value:
V IL(SC) = tON(MIN) IN L The resulting average short-circuit current is:
1
ISC = 45% • ILIM(MAX) – IL(SC)
2
Fault Conditions: Overvoltage Protection (Crowbar)
The overvoltage crowbar is designed to blow a system
input fuse when the output voltage of the regulator rises
much higher than nominal levels. The crowbar causes huge
currents to flow, that blow the fuse to protect against a
shorted top MOSFET if the short occurs while the controller is operating.
A comparator monitors the output for overvoltage conditions. The comparator detects faults greater than 10%
above the nominal output voltage. When this condition
is sensed, the top MOSFET is turned off and the bottom
MOSFET is turned on until the overvoltage condition is
cleared. The bottom MOSFET remains on continuously
for as long as the overvoltage condition persists; if VOUT
returns to a safe level, normal operation automatically
resumes.
A shorted top MOSFET will result in a high current condition
which will open the system fuse. The switching regulator
will regulate properly with a leaky top MOSFET by altering
the duty cycle to accommodate the leakage.
38901f
22
LTC3890-1
APPLICATIONS INFORMATION
Phase-Locked Loop and Frequency Synchronization
1000
900
The LTC3890-1 has an internal phase-locked loop (PLL)
comprised of a phase frequency detector, a lowpass filter,
and a voltage-controlled oscillator (VCO). This allows the
turn-on of the top MOSFET of controller 1 to be locked to
the rising edge of an external clock signal applied to the
PLLIN/MODE pin. The turn-on of controller 2’s top MOSFET
is thus 180 degrees out of phase with the external clock.
The phase detector is an edge sensitive digital type that
provides zero degrees phase shift between the external
and internal oscillators. This type of phase detector does
not exhibit false lock to harmonics of the external clock.
If the external clock frequency is greater than the internal oscillator’s frequency, fOSC, then current is sourced
continuously from the phase detector output, pulling up
the VCO input. When the external clock frequency is less
than fOSC, current is sunk continuously, pulling down the
VCO input. If the external and internal frequencies are the
same but exhibit a phase difference, the current sources
turn on for an amount of time corresponding to the phase
difference. The voltage at the VCO input is adjusted until
the phase and frequency of the internal and external oscillators are identical. At the stable operating point, the phase
detector output is high impedance and the internal filter
capacitor, CLP, holds the voltage at the VCO input.
Note that the LTC3890-1 can only be synchronized to
an external clock whose frequency is within range of
the LTC3890-1’s internal VCO, which is nominally 55kHz
to 1MHz. This is guaranteed to be between 75kHz and
850kHz.
Typically, the external clock (on the PLLIN/MODE pin)
input high threshold is 1.6V, while the input low threshold
is 1.1V.
Rapid phase locking can be achieved by using the FREQ pin
to set a free-running frequency near the desired synchronization frequency. The VCO’s input voltage is prebiased
at a frequency corresponding to the frequency set by the
FREQ pin. Once prebiased, the PLL only needs to adjust
the frequency slightly to achieve phase lock and synchronization. Although it is not required that the free-running
frequency be near external clock frequency, doing so will
FREQUENCY (kHz)
800
700
600
500
400
300
200
100
0
15 25 35 45 55 65 75 85 95 105 115 125
FREQ PIN RESISTOR (kΩ)
38901 F10
Figure 10. Relationship Between Oscillator Frequency
and Resistor Value at the FREQ Pin
prevent the operating frequency from passing through a
large range of frequencies as the PLL locks.
Table 2 summarizes the different states in which the FREQ
pin can be used.
Table 2
FREQ PIN
PLLIN/MODE PIN
FREQUENCY
0V
DC Voltage
350kHz
INTVCC
DC Voltage
535kHz
Resistor
DC Voltage
50kHz to 900kHz
Any of the Above
External Clock
Phase Locked to
External Clock
Minimum On-Time Considerations
Minimum on-time, tON(MIN), is the smallest time duration that the LTC3890-1 is capable of turning on the top
MOSFET. It is determined by internal timing delays and the
gate charge required to turn on the top MOSFET. Low duty
cycle applications may approach this minimum on-time
limit and care should be taken to ensure that:
tON(MIN) <
VOUT
VIN f
()
If the duty cycle falls below what can be accommodated
by the minimum on-time, the controller will begin to skip
cycles. The output voltage will continue to be regulated,
but the ripple voltage and current will increase.
38901f
23
LTC3890-1
APPLICATIONS INFORMATION
The minimum on-time for the LTC3890-1 is approximately
90ns. However, as the peak sense voltage decreases the
minimum on-time gradually increases up to about TBDns.
This is of particular concern in forced continuous applications with low ripple current at light loads. If the duty cycle
drops below the minimum on-time limit in this situation,
a significant amount of cycle skipping can occur with
correspondingly larger current and voltage ripple.
Efficiency Considerations
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Percent efficiency can
be expressed as:
%Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage of input power.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC3890-1 circuits: 1) IC VIN current, 2) INTVCC
regulator current, 3) I2R losses, 4) topside MOSFET
transition losses.
1. The VIN current is the DC supply current given in the
Electrical Characteristics table, which excludes MOSFET
driver and control currents. VIN current typically results
in a small (<0.1%) loss.
2. INTVCC current is the sum of the MOSFET driver and
control currents. The MOSFET driver current results
from switching the gate capacitance of the power
MOSFETs. Each time a MOSFET gate is switched from
low to high to low again, a packet of charge, dQ, moves
from INTVCC to ground. The resulting dQ/dt is a current
out of INTVCC that is typically much larger than the
control circuit current. In continuous mode, IGATECHG
= f(QT + QB), where QT and QB are the gate charges of
the topside and bottom side MOSFETs.
Supplying INTVCC from an output-derived source power
through EXTVCC will scale the VIN current required
for the driver and control circuits by a factor of (Duty
Cycle)/(Efficiency). For example, in a 20V to 5V application, 10mA of INTVCC current results in approximately
2.5mA of VIN current. This reduces the midcurrent loss
from 10% or more (if the driver was powered directly
from VIN) to only a few percent.
3. I2R losses are predicted from the DC resistances of the
fuse (if used), MOSFET, inductor, current sense resistor, and input and output capacitor ESR. In continuous
mode the average output current flows through L and
RSENSE, but is chopped between the topside MOSFET
and the synchronous MOSFET. If the two MOSFETs
have approximately the same RDS(ON), then the resistance of one MOSFET can simply be summed with the
resistances of L, RSENSE and ESR to obtain I2R losses.
For example, if each RDS(ON) = 30mΩ, RL = 50mΩ,
RSENSE = 10mΩ and RESR = 40mΩ (sum of both input
and output capacitance losses), then the total resistance
is 130mΩ. This results in losses ranging from 3% to
13% as the output current increases from 1A to 5A for
a 5V output, or a 4% to 20% loss for a 3.3V output.
Efficiency varies as the inverse square of VOUT for the
same external components and output power level. The
combined effects of increasingly lower output voltages
and higher currents required by high performance digital
systems is not doubling but quadrupling the importance
of loss terms in the switching regulator system!
4. Transition losses apply only to the topside MOSFET(s),
and become significant only when operating at high
input voltages (typically 15V or greater). Transition
losses can be estimated from:
Transition Loss = (1.7) • VIN • 2 • IO(MAX) • CRSS • f
Other hidden losses such as copper trace and internal
battery resistances can account for an additional 5%
to 10% efficiency degradation in portable systems. It
is very important to include these system level losses
38901f
24
LTC3890-1
APPLICATIONS INFORMATION
during the design phase. The internal battery and fuse
resistance losses can be minimized by making sure that
CIN has adequate charge storage and very low ESR at
the switching frequency. A 25W supply will typically
require a minimum of 20μF to 40μF of capacitance
having a maximum of 20mΩ to 50mΩ of ESR. The
LTC3890-1 2-phase architecture typically halves this
input capacitance requirement over competing solutions. Other losses including Schottky conduction losses
during dead-time and inductor core losses generally
account for less than 2% total additional loss.
Checking Transient Response
The regulator loop response can be checked by looking at
the load current transient response. Switching regulators
take several cycles to respond to a step in DC (resistive)
load current. When a load step occurs, VOUT shifts by
an amount equal to ΔILOAD (ESR), where ESR is the effective series resistance of COUT . ΔILOAD also begins to
charge or discharge COUT generating the feedback error
signal that forces the regulator to adapt to the current
change and return VOUT to its steady-state value. During
this recovery time VOUT can be monitored for excessive
overshoot or ringing, which would indicate a stability
problem. OPTI-LOOP compensation allows the transient
response to be optimized over a wide range of output
capacitance and ESR values. The availability of the ITH pin
not only allows optimization of control loop behavior, but
it also provides a DC coupled and AC filtered closed-loop
response test point. The DC step, rise time and settling
at this test point truly reflects the closed-loop response.
Assuming a predominantly second order system, phase
margin and/or damping factor can be estimated using the
percentage of overshoot seen at this pin. The bandwidth
can also be estimated by examining the rise time at the
pin. The ITH external components shown in Figure 13
circuit will provide an adequate starting point for most
applications.
The ITH series RC-CC filter sets the dominant pole-zero
loop compensation. The values can be modified slightly
(from 0.5 to 2 times their suggested values) to optimize
transient response once the final PC layout is done and
the particular output capacitor type and value have been
determined. The output capacitors need to be selected
because the various types and values determine the loop
gain and phase. An output current pulse of 20% to 80%
of full-load current having a rise time of 1μs to 10μs will
produce output voltage and ITH pin waveforms that will
give a sense of the overall loop stability without breaking
the feedback loop.
Placing a power MOSFET directly across the output
capacitor and driving the gate with an appropriate signal
generator is a practical way to produce a realistic load step
condition. The initial output voltage step resulting from
the step change in output current may not be within the
bandwidth of the feedback loop, so this signal cannot be
used to determine phase margin. This is why it is better to
look at the ITH pin signal which is in the feedback loop and
is the filtered and compensated control loop response.
The gain of the loop will be increased by increasing RC
and the bandwidth of the loop will be increased by decreasing CC. If RC is increased by the same factor that CC
is decreased, the zero frequency will be kept the same,
thereby keeping the phase shift the same in the most
critical frequency range of the feedback loop. The output
voltage settling behavior is related to the stability of the
closed-loop system and will demonstrate the actual overall
supply performance.
A second, more severe transient is caused by switching
in loads with large (>1μF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel
with COUT , causing a rapid drop in VOUT . No regulator can
alter its delivery of current quickly enough to prevent this
sudden step change in output voltage if the load switch
resistance is low and it is driven quickly. If the ratio of
CLOAD to COUT is greater than 1:50, the switch rise time
should be controlled so that the load rise time is limited
to approximately 25 • CLOAD. Thus a 10μF capacitor would
require a 250μs rise time, limiting the charging current
to about 200mA.
38901f
25
LTC3890-1
APPLICATIONS INFORMATION
Design Example
As a design example for one channel, assume VIN =
12V(nominal), VIN = 22V (max), VOUT = 3.3V, IMAX = 5A,
VSENSE(MAX) = 75mV and f = 350kHz.
The inductance value is chosen first based on a 30% ripple
current assumption. The highest value of ripple current
occurs at the maximum input voltage. Tie the FREQ pin
to GND, generating 350kHz operation. The minimum
inductance for 30% ripple current is:
V
V
IL = OUT 1– OUT ( f) (L) VIN(NOM) A 4.7μH inductor will produce 29% ripple current. The
peak inductor current will be the maximum DC value plus
one half the ripple current, or 5.73A. Increasing the ripple
current will also help ensure that the minimum on-time
of 95ns is not violated. The minimum on-time occurs at
maximum VIN:
tON(MIN) =
VOUT
VIN(MAX ) ( f)
=
3.3V
= 429ns
22V (350kHz )
The equivalent RSENSE resistor value can be calculated by
using the minimum value for the maximum current sense
threshold (43mV):
RSENSE ≤
64mV
≈ 0.01Ω
5.73A
Choosing 1% resistors: RA = 25k and RB = 78.7k yields
an output voltage of 3.32V.
The power dissipation on the topside MOSFET can be easily
estimated. Choosing a Fairchild FDS6982S dual MOSFET
results in: RDS(ON) = 0.035Ω/0.022Ω, CMILLER = 215pF. At
maximum input voltage with T(estimated) = 50°C:
PMAIN =
2
3.3V
5A ) 1+ (0.005) (50°C – 25°C)
(
22V
2
(0.035) + (22V) 5A2 (2.5) (215pF ) •
1 1
+
(350kHz ) = 331mW
5V – 2.3V 2.3V A short-circuit to ground will result in a folded back current of:
ISC =
34mV 1 95ns (22V ) –
= 3.18A
0.01 2 4.7μH with a typical value of RDS(ON) and δ = (0.005/°C)(25°C)
= 0.125. The resulting power dissipated in the bottom
MOSFET is:
2
PSYNC = (3.18A ) (1.125) (0.022Ω)
= 250mW
which is less than under full-load conditions.
CIN is chosen for an RMS current rating of at least 3A at
temperature assuming only this channel is on. COUT is
chosen with an ESR of 0.02Ω for low output ripple. The
output ripple in continuous mode will be highest at the
maximum input voltage. The output voltage ripple due to
ESR is approximately:
VORIPPLE = RESR (ΔIL) = 0.02Ω(1.45A) = 29mVP-P
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of
the IC. These items are also illustrated graphically in the
layout diagram of Figure 11. Figure 12 illustrates the current
waveforms present in the various branches of the 2-phase
synchronous regulators operating in the continuous mode.
Check the following in your layout:
1. Are the top N-channel MOSFETs MTOP1 and MTOP2
located within 1cm of each other with a common drain
connection at CIN? Do not attempt to split the input
decoupling for the two channels as it can cause a large
resonant loop.
2. Are the signal and power grounds kept separate? The
combined IC signal ground pin and the ground return
of CINTVCC must return to the combined COUT (–) terminals. The path formed by the top N-channel MOSFET,
Schottky diode and the CIN capacitor should have short
38901f
26
LTC3890-1
APPLICATIONS INFORMATION
leads and PC trace lengths. The output capacitor (–)
terminals should be connected as close as possible
to the (–) terminals of the input capacitor by placing
the capacitors next to each other and away from the
Schottky loop described above.
3. Do the LTC3890-1 VFB pins’ resistive dividers connect to
the (+) terminals of COUT? The resistive divider must be
connected between the (+) terminal of COUT and signal
ground. The feedback resistor connections should not
be along the high current input feeds from the input
capacitor(s).
4. Are the SENSE– and SENSE+ leads routed together
with minimum PC trace spacing? The filter capacitor
between SENSE+ and SENSE– should be as close as
possible to the IC. Ensure accurate current sensing with
Kelvin connections at the SENSE resistor.
5. Is the INTVCC decoupling capacitor connected close
to the IC, between the INTVCC and the power ground
pins? This capacitor carries the MOSFET drivers’ current peaks. An additional 1μF ceramic capacitor placed
immediately next to the INTVCC and PGND pins can
help improve noise performance substantially.
6. Keep the switching nodes (SW1, SW2), top gate nodes
(TG1, TG2), and boost nodes (BOOST1, BOOST2) away
from sensitive small-signal nodes, especially from
the opposites channel’s voltage and current sensing
feedback pins. All of these nodes have very large and
fast moving signals and therefore should be kept on
the output side of the LTC3890-1 and occupy minimum
PC trace area.
7. Use a modified star ground technique: a low impedance,
large copper area central grounding point on the same
side of the PC board as the input and output capacitors
with tie-ins for the bottom of the INTVCC decoupling
capacitor, the bottom of the voltage feedback resistive
divider and the SGND pin of the IC.
PC Board Layout Debugging
Start with one controller on at a time. It is helpful to use
a DC-50MHz current probe to monitor the current in the
inductor while testing the circuit. Monitor the output
switching node (SW pin) to synchronize the oscilloscope to
the internal oscillator and probe the actual output voltage
as well. Check for proper performance over the operating
voltage and current range expected in the application. The
frequency of operation should be maintained over the input
voltage range down to dropout and until the output load
drops below the low current operation threshold—typically 15% of the maximum designed current level in Burst
Mode operation.
The duty cycle percentage should be maintained from cycle
to cycle in a well-designed, low noise PCB implementation.
Variation in the duty cycle at a subharmonic rate can suggest noise pickup at the current or voltage sensing inputs
or inadequate loop compensation. Overcompensation of
the loop can be used to tame a poor PC layout if regulator bandwidth optimization is not required. Only after
each controller is checked for its individual performance
should both controllers be turned on at the same time.
A particularly difficult region of operation is when one
controller channel is nearing its current comparator trip
point when the other channel is turning on its top MOSFET.
This occurs around 50% duty cycle on either channel due
to the phasing of the internal clocks and may cause minor
duty cycle jitter.
Reduce VIN from its nominal level to verify operation
of the regulator in dropout. Check the operation of the
undervoltage lockout circuit by further lowering VIN while
monitoring the outputs to verify operation.
Investigate whether any problems exist only at higher output currents or only at higher input voltages. If problems
coincide with high input voltages and low output currents,
look for capacitive coupling between the BOOST, SW, TG,
38901f
27
LTC3890-1
APPLICATIONS INFORMATION
and possibly BG connections and the sensitive voltage
and current pins. The capacitor placed across the current
sensing pins needs to be placed immediately adjacent to
the pins of the IC. This capacitor helps to minimize the
effects of differential noise injection due to high frequency
capacitive coupling. If problems are encountered with
high current output loading at lower input voltages, look
for inductive coupling between CIN, Schottky and the top
MOSFET components to the sensitive current and voltage
sensing traces. In addition, investigate common ground
path voltage pickup between these components and the
SGND pin of the IC.
An embarrassing problem, which can be missed in an
otherwise properly working switching regulator, results
when the current sensing leads are hooked up backwards.
The output voltage under this improper hookup will still
be maintained but the advantages of current mode control
will not be realized. Compensation of the voltage loop will
be much more sensitive to component selection. This
behavior can be investigated by temporarily shorting out
the current sensing resistor—don’t worry, the regulator
will still maintain control of the output voltage.
38901f
28
LTC3890-1
APPLICATIONS INFORMATION
ITH1
TRACK/SS1
VFB1
PGOOD1
RPU1
VPULL-UP
PGOOD1
L1
SENSE1+
RSENSE
VOUT1
TG1
SW1
SENSE1–
LTC3890-1
BOOST1
CB1
M1
M2
D1
BG1
FREQ
fIN
PLLIN/MODE
RUN1
RIN
VIN
CVIN
EXTVCC
SENSE2–
INTVCC
SENSE2+
BG2
VFB2
COUT1
+
PGND
RUN2
SGND
1μF
CERAMIC
GND
+
CINTVCC
VIN
COUT2
1μF
CERAMIC
M3
BOOST2
+
CIN
M4
D2
CB2
ITH2
TRACK/SS2
SW2
RSENSE
TG2
VOUT2
L2
38901 F11
Figure 11. Recommended Printed Circuit Layout Diagram
38901f
29
LTC3890-1
APPLICATIONS INFORMATION
SW1
L1
D1
RSENSE1
VOUT1
COUT1
RL1
VIN
RIN
CIN
SW2
BOLD LINES INDICATE
HIGH SWITCHING
CURRENT. KEEP LINES
TO A MINIMUM LENGTH.
D2
L2
RSENSE2
VOUT2
COUT2
RL2
38901 F12
Figure 12. Branch Current Waveforms
38901f
30
LTC3890-1
TYPICAL APPLICATIONS
SENSE1+
C1
1nF
RB1
100k
RA1
31.6k
INTVCC
100k
SENSE1–
PGOOD1
VFB1
CITH1A 100pF
BG1
RITH1
34.8k
CITH1 1000pF
MBOT1
SW1
BOOST1
ITH1
CB1
0.1μF
LTC3890-1
CSS1 0.01μF
L1
4.7μH
TRACK/SS1
TG1
RSENSE1
8mΩ
VOUT1
3.3V
5A
COUT1
470μF
MTOP1
D1
VIN
PLLIN/MODE
VOUT2
RFREQ
41.2k
SGND
EXTVCC
RUN1
RUN2
FREQ
INTVCC
CINT
4.7μF
PGND
D2
TG2
CSS2 0.01μF
MTOP2
CB2 0.1μF
TRACK/SS2
RITH2
34.8k
CITH2 470pF
VIN
9V TO 60V
CIN
220μF
L2
8μH
BOOST2
RSENSE2
10mΩ
VOUT2
8.5V
3A
SW2
ITH2
COUT2
330μF
RA2
10.5k
VFB2
RB2
100k
MBOT2
BG2
SENSE2–
C2
1nF
SENSE2+
38901 TA07a
MTOP1, MTOP2, MBOT1, MBOT2: RJK0651DPB
L1: COILCRAFT SER1360-472KL
L2: COILCRAFT SER1360-802KL
COUT1: SANYO 6TPE470M
COUT2: SANYO 10TPE330M
D1, D2: DFLS1100
Figure 13. High Efficiency Dual 8.5V/3.3V Step-Down Converter
Efficiency and Power Loss
vs Output Current
Efficiency vs Load Current
10000
VIN = 12V
90 VOUT = 3.3V
BURST EFFICIENCY
90
1000
CCM LOSS
100
60
50
40
PULSE-SKIPPING
LOSS
BURST LOSS
10
30
20
10
0
0.0001
CCM EFFICIENCY
PULSE-SKIPPING
EFFICIENCY
0.001
0.01
0.1
1
OUTPUT CURRENT (A)
1
0.1
10
38901 TA07b
EEFICIENCY (%)
70
80
POWER LOSS (mW)
EEFICIENCY (%)
80
Efficiency vs Input Voltage
100
100
VOUT = 8.5V
98
VOUT = 3.3V
70
60
50
40
94
92
90
VOUT1 = 3.3V
88
30
86
20
84
10
82
VIN = 12V
0
0.0001 0.001
0.01
0.1
1
OUTPUT CURRENT (A)
VOUT2 = 8.5V
96
EEFICIENCY (%)
100
ILOAD = 2A
80
10
38901 TA07c
0
5 10 15 20 25 30 35 40 45 50 55 60
INPUT VOLTAGE (V)
38901 TA07d
38901f
31
LTC3890-1
TYPICAL APPLICATIONS
High Efficiency 8.5V Dual-Phase Step-Down Converter
SENSE1+
RB1
100k
C1
1nF
RA1
10.5k
SENSE1–
INTVCC
100k
PGOOD1
VFB1
MBOT1
CITH1A 100pF
BG1
L1
8μH
VOUT1
8.5V
6A
SW1
RITH1 34.8k
CITH1 C
SS1 0.01μF
470pF
RSENSE1
10mΩ
BOOST1
ITH1
TRACK/SS1
COUT1
330μF
CB1
0.1μF
LTC3890-1
TG1
MTOP1
D1
INTVCC RMODE
100k
VIN
CIN 9V TO 60V
220μF
VIN
PLLIN/MODE
INTVCC
CINT
4.7μF
SGND
VIN
RRUN
V
1000k OUT
EXTVCC
RUN1
RUN2
FREQ
RFREQ 41.2k
PGND
D2
TG2
BOOST2
ITH2
VFB2
MTOP2
CB2 0.1μF
TRACK/SS2
CITH2
100pF
L2
8μH
RSENSE2
10mΩ
SW2
BG2
COUT2
330μF
MBOT2
SENSE2–
C2
1nF
SENSE2+
38901 TA03
MTOP1, MTOP2, MBOT1, MBOT2: RJK0651DPB
L1, L2: COILCRAFT SER1360-802KL
COUT1, COUT2: SANYO 10TPE330M
D1, D2: DFLS1100
38901f
32
LTC3890-1
TYPICAL APPLICATIONS
High Efficiency Dual 12V/5V Step-Down Converter
SENSE1+
RB1
100k
C1
1nF
RA1
6.98k
SENSE1–
INTVCC
100k
PGOOD1
VFB1
CITH1A 100pF
CITH1 470pF
BG1
RITH1
34.8k
MBOT1
SW1
TRACK/SS1
COUT1
180μF
VOUT1
12V
3A
CB1
0.47μF
LTC3890-1
CSS1 0.01μF
L1
8μH
BOOST1
ITH1
RSENSE1
9mΩ
TG1
MTOP1
D1
VIN
PLLIN/MODE
SGND
EXTVCC
RUN1
RUN2
FREQ
RFREQ
41.2k
INTVCC
VIN
12.5V TO 60V
CINT
4.7μF
PGND
D2
TG2
MTOP2
CB2 0.47μF
CSS2 0.01μF
TRACK/SS2
CITH2 470pF
CIN
220μF
RITH2
20k
BOOST2
L2
4.7μH
RSENSE2
10mΩ
VOUT2
5V
5A
COUT2
470μF
SW2
ITH2
BG2
MBOT2
RA2
18.7k
VFB2
SENSE2–
RB2
100k
C2
1nF
SENSE2+
38901 TA04
MTOP1, MTOP2, MBOT1, MBOT2: RJK0651DPB
L1: COILCRAFT SER1360-802KL
L2: COILCRAFT SER1360-472KL
COUT1: 16SVP180MX
COUT2: SANYO 6TPE470M
D1, D2: DFLS1100
38901f
33
LTC3890-1
TYPICAL APPLICATIONS
High Efficiency Dual 24V/5V Step-Down Converter
RB1
487k
SENSE1+
C1
1nF
CF1 33pF
RA1
16.9k
PGOOD1
VFB1
CITH1A 100pF
CITH1 680pF
SENSE1–
INTVCC
100k
BG1
RITH1
46k
MBOT1
L1
22μH
RSENSE1
25mΩ
VOUT1
24V
1A
SW1
BOOST1
ITH1
TRACK/SS1
COUT1
22μF
×2 CERAMIC
CB1
0.47μF
LTC3890-1
CSS1 0.01μF
TG1
MTOP1
D1
VIN
PLLIN/MODE
SGND
EXTVCC
RUN1
RUN2
FREQ
RFREQ
60k
INTVCC
VIN
28V TO 60V
CINT
4.7μF
PGND
D2
TG2
MTOP2
CB2 0.47μF
CSS2 0.01μF
CITH2 470pF
CIN
220μF
TRACK/SS2
RITH2
20k
BOOST2
L2
4.7μH
RSENSE2
10mΩ
VOUT2
5V
5A
SW2
ITH2
BG2
RA2
18.7k
COUT2
470μF
MBOT2
VFB2
RB2 100k
SENSE2–
C2
1nF
SENSE2+
38901 TA05
MTOP1, MTOP2, MBOT1, MBOT2: RJK0651DPB
L1: SUMIDA CDR7D43MN
L2: COILCRAFT SER1360-472KL
COUT1: KEMET T525D476MO16E035
COUT2: SANYO 6TPE470M
D1, D2: DFLS1100
38901f
34
LTC3890-1
PACKAGE DESCRIPTION
GN Package
28-Lead Plastic SSOP (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1641)
.386 – .393*
(9.804 – 9.982)
.045 p.005
28 27 26 25 24 23 22 21 20 19 18 17 1615
.254 MIN
.033
(0.838)
REF
.150 – .165
.229 – .244
(5.817 – 6.198)
.0165 p.0015
.150 – .157**
(3.810 – 3.988)
.0250 BSC
1
RECOMMENDED SOLDER PAD LAYOUT
.015 p .004
s 45o
(0.38 p 0.10)
.0075 – .0098
(0.19 – 0.25)
2 3
4
5 6
7
8
9 10 11 12 13 14
.0532 – .0688
(1.35 – 1.75)
.004 – .0098
(0.102 – 0.249)
0o – 8o TYP
.016 – .050
(0.406 – 1.270)
NOTE:
1. CONTROLLING DIMENSION: INCHES
INCHES
2. DIMENSIONS ARE IN
(MILLIMETERS)
.008 – .012
(0.203 – 0.305)
TYP
.0250
(0.635)
BSC
GN28 (SSOP) 0204
3. DRAWING NOT TO SCALE
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
38901f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
35
LTC3890-1
TYPICAL APPLICATION
High Efficiency Dual 12V/3.3V Step-Down Converter
SENSE1+
C1
1nF
RB1
100k
RA1
6.98k
SENSE1–
INTVCC
100k
PGOOD1
VFB1
CITH1A 100pF
BG1
RITH1
34.8k
CITH1 470pF
MBOT1
SW1
BOOST1
ITH1
TRACK/SS1
COUT1
180μF
VOUT1
12V
3A
CB1
0.47μF
LTC3890-1
CSS1 0.01μF
L1
8μH
RSENSE1
9mΩ
TG1
MTOP1
D1
VIN
PLLIN/MODE
SGND
EXTVCC
RUN1
RUN2
FREQ
RFREQ
41.2k
INTVCC
D2
TG2
MTOP2
CB2 0.47μF
TRACK/SS2
RITH2
34.8k
BOOST2
L2
4.7μH
RSENSE2
10mΩ
VOUT2
3.3V
5A
COUT2
470μF
SW2
ITH2
CITH2A
100pF
RA2
31.6k
VIN
12.5V TO 60V
CINT
4.7μF
PGND
CSS2 0.01μF
CITH2 1000pF
CIN
220μF
BG2
MBOT2
VFB2
MTOP1, MTOP2, MBOT1, MBOT2: RJK0651DPB
L1: COILCRAFT SER1360-802KL
L2: COILCRAFT SER1360-472KL
COUT1: 16SVP180MX
COUT2: SANYO 6TPE470M
D1, D2: DFLS1100
SENSE2–
RB2
100k
C2
1nF
SENSE2+
38901 TA06
RELATED PARTS
PART NUMBER
DESCRIPTION
LTC3857/LTC3857-1 Low IQ, Dual Output 2-Phase Synchronous Step-Down
DC/DC Controllers with 99% Duty Cycle
LTC3858/LTC3858-1 Low IQ, Dual Output 2-Phase Synchronous Step-Down
DC/DC Controllers with 99% Duty Cycle
LTC3868/LTC3868-1 Low IQ, Dual Output 2-Phase Synchronous Step-Down
DC/DC Controller with 99% Duty Cycle
LTC3834/LTC3834-1 Low IQ, Synchronous Step-Down DC/DC Controller with
99% Duty Cycle
LTC3835/LTC3835-1 Low IQ, Synchronous Step-Down DC/DC Controller with
99% Duty Cycle
LT3845
Low IQ, High Voltage Synchronous Step-Down DC/DC
Controller
COMMENTS
Phase-Lockable Fixed Frequency 50kHz to 900kHz, 4V ≤ VIN ≤ 38V,
0.8V ≤ VOUT ≤ 24V, IQ = 50μA
Phase-Lockable Fixed Frequency 50kHz to 900kHz, 4V ≤ VIN ≤ 38V,
0.8V ≤ VOUT ≤ 24V, IQ = 170μA
Phase-Lockable Fixed Frequency 50kHz to 900kHz, 4V ≤ VIN ≤ 24V,
0.8V ≤ VOUT ≤ 14V, IQ = 170μA
Phase-Lockable Fixed Frequency 140kHz to 650kHz, 4V ≤ VIN ≤ 36V,
0.8V ≤ VOUT ≤ 10V, IQ = 30μA
Phase-Lockable Fixed Frequency 140kHz to 650kHz, 4V ≤ VIN ≤ 36V,
0.8V ≤ VOUT ≤ 10V, IQ = 80μA
Adjustable Fixed Frequency 100kHz to 500kHz, 4V ≤ VIN ≤ 60V,
1.23V ≤ VOUT ≤ 36V, IQ = 120μA, TSSOP-16
38901f
36 Linear Technology Corporation
LT 0210 • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900
●
FAX: (408) 434-0507 ● www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2010
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