LINER LTC3803HS6-5 Constant frequency current mode flyback dc/dc controller in thinsot Datasheet

LTC3803-5
Constant Frequency
Current Mode Flyback
DC/DC Controller in ThinSOT
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FEATURES
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DESCRIPTIO
VIN and VOUT Limited Only by External Components
4.8V Undervoltage Lockout Threshold
Operating Junction Temperature from –40°C to
150°C
Adjustable Slope Compensation
Internal Soft-Start
Constant Frequency 200kHz Operation
±1.5% Reference Accuracy
Current Mode Operation for Excellent Line and Load
Transient Response
No Minimum Load Requirement
Low Quiescent Current: 240µA
Low Profile (1mm) SOT-23 Package
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APPLICATIO S
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42V and 12V Automotive Power Supplies
Telecom Power Supplies
Auxiliary/Housekeeping Power Supplies
Power Over Ethernet
The LTC®3803-5 is a constant frequency current mode
flyback controller optimized for driving 4.5V and 6V-rated
N-channel MOSFETs in high input voltage applications.
The LTC3803-5 operates from inputs as low as 5V. Constant frequency operation is maintained down to very light
loads, resulting in less low frequency noise generation
over a wide range of load currents. Slope compensation
can be programmed with an external resistor.
The LTC3803-5 provides ±1.5% output voltage accuracy
and consumes only 240µA of quiescent current. Groundreferenced current sensing allows LTC3803-5-based converters to accept input supplies beyond the LTC3803-5’s
absolute maximum VCC. For simplicity, the LTC3803-5
can be powered from a high VIN through a resistor, due to
its internal 8V shunt regulator. An internal undervoltage
lockout shuts down the IC when the input voltage falls
below 3.2V, guaranteeing at least 3.2V of gate drive to the
external MOSFET.
The LTC3803-5 is available in a low profile (1mm) 6-lead
SOT-23 (ThinSOTTM) package.
, LTC and LT are registered trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
ThinSOT is a trademark of Linear Technology Corporation.
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TYPICAL APPLICATIO
Efficiency and Power Loss
vs Output Power
Dual Output Wide Input Range Converter
VPH5-0155
7.5k
LTC3803-5
PHM25NQ10T
1µF
100V
ITH/RUN NGATE
GND
8.06k
VFB
VCC
SENSE
1µF
100V
4.7k
B3100
0.012Ω
85 VIN = 12V
2.5
80
2.0
VIN = 24V
75
6.5V/1.2A
70
47µF
10V
65
0.1µF
ALL CAPACITORS ARE X7R, TDK
38035 TA01
1.5
1.0
VIN = 48V
60
57.6k
3.0
VIN = 8V
POWER LOSS (W)
22µF
10V
MMBTA42
10nF
90
13V/0.3A
20mA MIN
LOAD
1µF
100V
3x
22k
PDZ9.1B
10MQ100N
EFFICIENCY (%)
VIN
6V TO 50V
0.5
VIN = 12V
0
2
6
8
4
OUTPUT POWER (W)
10
0
12
38035 TA01b
38035f
1
LTC3803-5
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ABSOLUTE MAXIMUM RATINGS
PACKAGE/ORDER INFORMATION
(Note 1)
VCC to GND (Current Fed) .................... 25mA into VCC*
NGATE Voltage ......................................... – 0.3V to VCC
VFB, ITH/RUN Voltages ..............................– 0.3V to 3.5V
SENSE Voltage ........................................... – 0.3V to 1V
NGATE Peak Output Current (<10µs) ........................ 1A
Operating Junction Temperature Range (Note 2)
LTC3803E-5 ....................................... – 40°C to 85°C
LTC3803H-5 (Note 3) ....................... – 40°C to 150°C
Storage Temperature Range ................. – 65°C to 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
*LTC3803-5 internal clamp circuit self regulates VCC voltage to 8V.
ORDER PART
NUMBER
TOP VIEW
LTC3803HS6-5
LTC3803ES6-5
6 NGATE
ITH/RUN 1
5 VCC
GND 2
VFB 3
4 SENSE
S6 PART
MARKING
S6 PACKAGE
6-LEAD PLASTIC TSOT-23
TJMAX = 150°C, θJA = 165°C/W
LTBMH
LTBPF
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
LTC3803E-5: The ● indicates specifications which apply over the full –40°C
to 85°C operating junction temperature range, otherwise specifications are at TJ = 25°C. VCC = 5V, unless otherwise noted. (Note 2)
SYMBOL
VTURNON
VTURNOFF
VHYST
VCLAMP1mA
VCLAMP25mA
ICC
VITHSHDN
IITHSTART
VFB
PARAMETER
VCC Turn On Voltage
VCC Turn Off Voltage
VCC Hysteresis
VCC Shunt Regulator Voltage
VCC Shunt Regulator Voltage
Input DC Supply Current
Normal Operation
Undervoltage
Shutdown Threshold (at ITH/RUN)
Start-Up Current Source
Regulated Feedback Voltage
gm
∆VO(LINE)
∆VO(LOAD)
Error Amplifier Transconductance
Output Voltage Line Regulation
Output Voltage Load Regulation
IFB
fOSC
DCON(MIN)
DCON(MAX)
tRISE
tFALL
VIMAX
ISLMAX
tSFST
VFB Input Current
Oscillator Frequency
Minimum Switch On Duty Cycle
Maximum Switch On Duty Cycle
Gate Drive Rise Time
Gate Drive Fall Time
Peak Current Sense Voltage
Peak Slope Compensation Output Current
Soft-Start Time
CONDITIONS
●
●
VTURNON – VTURNOFF
ICC = 1mA, VITH/RUN = 0V
ICC = 25mA, VITH/RUN = 0V
(Note 4)
VITH/RUN = 1.3V
VCC = VTURNON – 100mV
VCC = VTURNON + 100mV
VITH/RUN = 0V
0°C ≤ TJ ≤ 85°C (Note 5)
–40°C ≤ TJ ≤ 85°C (Note 5)
ITH/RUN Pin Load = ±5µA (Note 5)
VTURNOFF < VCC < VCLAMP (Note 5)
ITH/RUN Sinking 5µA (Note 5)
ITH/RUN Sourcing 5µA (Note 5)
(Note 5)
VITH/RUN = 1.3V
VITH/RUN = 1.3V, VFB = 0.8V
VITH/RUN = 1.3V, VFB = 0.8V
CLOAD = 3000pF
CLOAD = 3000pF
RSL = 0 (Note 6)
(Note 7)
●
●
●
MIN
4
3.3
0.05
6.2
6.3
●
●
●
●
0.12
0.07
0.788
0.780
200
170
70
●
90
TYP
4.8
4
0.8
8
8.1
MAX
5.7
4.9
240
40
0.28
0.34
0.800
0.800
333
0.1
3
3
10
200
6.5
80
40
40
100
5
0.7
350
90
0.45
0.8
0.812
0.816
500
9.9
10.3
50
230
8.5
90
115
UNITS
V
V
V
V
V
µA
µA
V
µA
V
V
µA/V
mV/V
mV/µA
mV/µA
nA
kHz
%
%
ns
ns
mV
µA
ms
38035f
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LTC3803-5
ELECTRICAL CHARACTERISTICS
LTC3803H-5: The ● indicates specifications which apply over the full –40°C
to 150°C operating junction temperature range, otherwise specifications are at TA = 25°C. VCC = 5V, unless otherwise noted.
(Notes 2, 3)
SYMBOL
VTURNON
VTURNOFF
VHYST
VCLAMP1mA
VCLAMP25mA
ICC
VITHSHDN
IITHSTART
VFB
PARAMETER
VCC Turn On Voltage
VCC Turn Off Voltage
VCC Hysteresis
VCC Shunt Regulator Voltage
VCC Shunt Regulator Voltage
Input DC Supply Current
Normal Operation
Undervoltage
Shutdown Threshold (at ITH/RUN)
Start-Up Current Source
Regulated Feedback Voltage
gm
∆VO(LINE)
∆VO(LOAD)
Error Amplifier Transconductance
Output Voltage Line Regulation
Output Voltage Load Regulation
IFB
fOSC
DCON(MIN)
DCON(MAX)
tRISE
tFALL
VIMAX
ISLMAX
tSFST
VFB Input Current
Oscillator Frequency
Minimum Switch On Duty Cycle
Maximum Switch On Duty Cycle
Gate Drive Rise Time
Gate Drive Fall Time
Peak Current Sense Voltage
Peak Slope Compensation Output Current
Soft-Start Time
CONDITIONS
●
●
VTURNON – VTURNOFF
ICC = 1mA, VITH/RUN = 0V
ICC = 25mA, VITH/RUN = 0V
(Note 4)
VITH/RUN = 1.3V
VCC = VTURNON – 100mV
VCC = VTURNON + 100mV
VITH/RUN = 0V
0°C ≤ TJ ≤ 85°C (Note 5)
–40°C ≤ TJ ≤ 150°C (Note 5)
ITH/RUN Pin Load = ±5µA (Note 5)
VTURNOFF < VCC < VCLAMP (Note 5)
ITH/RUN Sinking 5µA (Note 5)
ITH/RUN Sourcing 5µA (Note 5)
(Note 5)
VITH/RUN = 1.3V
VITH/RUN = 1.3V, VFB = 0.8V
VITH/RUN = 1.3V, VFB = 0.8V
CLOAD = 3000pF
CLOAD = 3000pF
RSL = 0 (Note 6)
(Note 7)
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: The LTC3803H-5 is guaranteed to meet specifications from –40°C
to 150°C. The LTC3803E-5 is guaranteed to meet specifications from 0°C
to 85°C with specifications over the –40°C to 85°C temperature range
assured by design, characterization and correlation with statistical process
controls.
Junction temperature (TJ) is calculated from the ambient temperature TA
and the power dissipation PD in the LTC3803-5 using the formula:
TJ = TA + (PD • 230°C/W)
Note 3: High junction temperatures degrade operating lifetimes. Operating
●
●
●
MIN
3.9
3.2
0.05
6.2
6.3
●
●
●
●
0.08
0.07
0.788
0.780
200
170
70
●
85
TYP
4.8
4
0.8
8
8.1
MAX
5.7
4.9
240
40
0.28
0.34
0.800
0.800
333
0.1
3
3
10
200
6.5
80
40
40
100
5
0.7
350
100
0.45
1
0.812
0.820
500
10.4
10.7
50
230
8.5
90
115
UNITS
V
V
V
V
V
µA
µA
V
µA
V
V
µA/V
mV/V
mV/µA
mV/µA
nA
kHz
%
%
ns
ns
mV
µA
ms
lifetime at junction temperatures greater than 125°C is derated to 1000
hours.
Note 4: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency.
Note 5: The LTC3803-5 is tested in a feedback loop that servos VFB to the
output of the error amplifier while maintaining ITH/RUN at the midpoint of
the current limit range.
Note 6: Peak current sense voltage is reduced dependent on duty cycle
and an optional external resistor in series with the SENSE pin (RSL). For
details, refer to the programmable slope compensation feature in the
Applications Information section.
Note 7: Guaranteed by design.
38035f
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LTC3803-5
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TYPICAL PERFOR A CE CHARACTERISTICS
812
808
VFB VOLTAGE (mV)
VFB VOLTAGE (mV)
808
804
800
796
812
TA = 25°C
VCC ≤ VCLAMP1mA
804
800
796
792
792
788
–50 –30 –10 10 30 50 70 90 110 130 150
TEMPERATURE (°C)
788
4.0
220
VCC = 5V
4.5
5.0 5.5 6.0
6.5 7.0
VCC SUPPLY VOLTAGE (V)
788
7.5
0
205
200
195
190
10
15
ICC (mA)
20
220
TA = 25°C
210
205
200
195
190
185
185
180
–50 –30 –10 10 30 50 70 90 110 130 150
TEMPERATURE (°C)
180
TA = 25°C
215
210
205
200
195
190
185
180
4.0
4.5
6.5 7.0
5.0 5.5 6.0
VCC SUPPLY VOLTAGE (V)
38035 G04
7.5
0
5
15
10
ICC (mA)
38035 G05
VCC Undervoltage Lockout
Thresholds vs Temperature
10.5
5.5
10.0
20
25
38035 G06
VCC Shunt Regulator Voltage
vs Temperature
6.0
25
Oscillator Frequency
vs VCC Shunt Regulator Current
OSCILLATOR FREQUENCY (kHz)
210
5
38035 G03
215
OSCILLATOR FREQUENCY (kHz)
OSCILLATOR FREQUENCY (kHz)
796
Oscillator Frequency
vs Supply Voltage
215
ICC Supply Current
vs Temperature
300
VCC = 5V
VITH/RUN = 1.3V
280
9.5
VTURNON
VCC (V)
VOLTS
800
38035 F02
Oscillator Frequency
vs Temperature
5.0
804
792
38035 G01
220
TA = 25°C
808
VFB VOLTAGE (mV)
VCC = 5V
SUPPLY CURRENT (µA)
812
Reference Voltage
vs VCC Shunt Regulator Current
Reference Voltage
vs Supply Voltage
Reference Voltage vs Temperature
4.5
9.0
ICC = 25mA
8.5
ICC = 1mA
4.0
8.0
VTURNOFF
260
240
220
3.5
7.5
3.0
–50 –30 –10 10 30 50 70 90 110 130 150
TEMPERATURE (°C)
7.0
–50 –30 –10 10 30 50 70 90 110 130 150
TEMPERATURE (°C)
3803 G07
38035 G08
200
–50 –30 –10 10 30 50 70 90 110 130 150
TEMPERATURE (°C)
38035 G08
38035f
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LTC3803-5
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TYPICAL PERFOR A CE CHARACTERISTICS
Start-Up ICC Supply Current
vs Temperature
ITH/RUN Start-Up Current Source
vs Temperature
500
VCC = VTURNON – 0.1V
1000
SHUTDOWN THRESHOLD (mV)
50
40
30
20
10
ITH/RUN PIN CURRENT SOURCE (nA)
450
60
400
350
300
250
200
150
100
50
0
–50 –30 –10 10 30 50 70 90 110 130 150
TEMPERATURE (°C)
0
–50 –30 –10 10 30 50 70 90 110 130 150
TEMPERATURE (°C)
Peak Current Sense Voltage
vs Temperature
120
VCC = VTURNON + 0.1V
900 VITH/RUN = 0V
800
700
600
500
400
300
200
100
0
–50 –30 –10 10 30 50 70 90 110 130 150
TEMPERATURE (°C)
3803 G11
38035 G10
38035 G12
Soft-Start Time vs Temperature
1.4
VCC = 5V
115
VCC = 5V
1.2
110
SOFT-START TIME (ms)
SENSE PIN VOLTAGE (mV)
START-UP SUPPLY CURRENT (µA)
70
ITH/RUN Shutdown Threshold
vs Temperature
105
100
95
90
1.0
0.8
0.6
0.4
0.2
85
80
–50 –30 –10 10 30 50 70 90 110 130 150
TEMPERATURE (°C)
38035 G13
0
–50 –30 –10 10 30 50 70 90 110 130 150
TEMPERATURE (°C)
38035 G14
38035f
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LTC3803-5
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PI FU CTIO S
ITH/RUN (Pin 1): This pin performs two functions. It
serves as the error amplifier compensation point as well as
the run/shutdown control input. Nominal voltage range is
0.7V to 1.9V. Forcing this pin below 0.28V causes the
LTC3803-5 to shut down. In shutdown mode, the NGATE
pin is held low.
SENSE (Pin 4): This pin performs two functions. It monitors switch current by reading the voltage across an
external current sense resistor to ground. It also injects a
current ramp that develops slope compensation voltage
across an optional external programming resistor.
GND (Pin 2): Ground Pin.
VCC (Pin 5): Supply Pin. Must be closely decoupled to GND
(Pin 2).
VFB (Pin 3): Receives the feedback voltage from an external resistive divider across the output.
NGATE (Pin 6): Gate Drive for the External N-Channel
MOSFET. This pin swings from 0V to VCC.
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BLOCK DIAGRA
5
VCC
0.3µA 0.28V
800mV
REFERENCE
VCC
SHUNT
REGULATOR
+
SHUTDOWN
COMPARATOR
VCC < VTURNON
–
SHUTDOWN
SOFTSTART
CLAMP
+
–
ERROR
AMPLIFIER
CURRENT
COMPARATOR
3
GND
2
VCC
R
+
VFB
UNDERVOLTAGE
LOCKOUT
Q
S
–
20mV
1.2V
200kHz
OSCILLATOR
SWITCHING
LOGIC AND
BLANKING
CIRCUIT
GATE
DRIVER NGATE
SLOPE
COMP
CURRENT
RAMP
SENSE
1
6
4
ITH/RUN
38035 BD
38035f
6
LTC3803-5
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OPERATIO
The LTC3803-5 is a constant frequency current mode
controller for flyback, SEPIC and DC/DC boost converter
applications in a tiny ThinSOT package. The LTC3803-5 is
designed so that none of its pins need to come in contact
with the input or output voltages of the power supply
circuit of which it is a part, allowing the conversion of
voltages well beyond the LTC3803-5’s absolute maximum
ratings.
Main Control Loop
Due to space limitations, the basics of current mode
DC/DC conversion will not be discussed here; instead, the
reader is referred to the detailed treatment in Application
Note 19, or in texts such as Abraham Pressman’s Switching Power Supply Design.
Please refer to the Block Diagram and the Typical Application on the front page of this data sheet. An external
resistive voltage divider presents a fraction of the output
voltage to the VFB pin. The divider must be designed so that
when the output is at the desired voltage, the VFB pin
voltage will equal the 800mV from the internal reference.
If the load current increases, the output voltage will
decrease slightly, causing the VFB pin voltage to fall below
800mV. The error amplifier responds by feeding current
into the ITH/RUN pin. If the load current decreases, the VFB
voltage will rise above 800mV and the error amplifier will
sink current away from the ITH/RUN pin.
The voltage at the ITH/RUN pin commands the pulse-width
modulator formed by the oscillator, current comparator
and RS latch. Specifically, the voltage at the ITH/RUN pin
sets the current comparator’s trip threshold. The current
comparator monitors the voltage across a current sense
resistor in series with the source terminal of the external
MOSFET. The LTC3803-5 turns on the external power
MOSFET when the internal free-running 200kHz oscillator
sets the RS latch. It turns off the MOSFET when the current
comparator resets the latch or when 80% duty cycle is
reached, whichever happens first. In this way, the peak
current levels through the flyback transformer’s primary
and secondary are controlled by the ITH/RUN voltage.
Since the ITH/RUN voltage is increased by the error amplifier whenever the output voltage is below nominal, and
decreased whenever output voltage exceeds nominal, the
voltage regulation loop is closed. For example, whenever
the load current increases, output voltage will decrease
slightly, and sensing this, the error amplifier raises the
ITH/RUN voltage by sourcing current into the ITH/RUN pin,
raising the current comparator threshold, thus increasing
the peak currents through the transformer primary and
secondary. This delivers more current to the load, bringing the output voltage back up.
The ITH/RUN pin serves as the compensation point for the
control loop. Typically, an external series RC network is
connected from ITH/RUN to ground and is chosen for
optimal response to load and line transients. The impedance of this RC network converts the output current of the
error amplifier to the ITH/RUN voltage which sets the
current comparator threshold and commands considerable influence over the dynamics of the voltage regulation
loop.
Start-Up/Shutdown
The LTC3803-5 has two shutdown mechanisms to disable
and enable operation: an undervoltage lockout on the VCC
supply pin voltage, and a forced shutdown whenever
external circuitry drives the ITH/RUN pin low. The LTC38035 transitions into and out of shutdown according to the
state diagram (Figure 1).
LTC3803-5
SHUT DOWN
VCC < VTURNOFF
(NOMINALLY 4V)
V
> VITHSHDN
VITH/RUN < VITHSHDN ITH/RUN
AND VCC > VTURNON
(NOMINALLY 0.28V)
(NOMINALLY 4.8V)
LTC3803-5
ENABLED
38035 F01
Figure 1. Start-Up/Shutdown State Diagram
38035f
7
LTC3803-5
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OPERATIO
The undervoltage lockout (UVLO) mechanism prevents
the LTC3803-5 from trying to drive a MOSFET with insufficient VGS. The voltage at the VCC pin must exceed
VTURNON (nominally 4.8V) at least momentarily to enable
LTC3803-5 operation. The VCC voltage is then allowed to
fall to VTURNOFF (nominally 4V) before undervoltage lockout disables the LTC3803-5.
The ITH/RUN pin can be driven below VSHDN (nominally
0.28V) to force the LTC3803-5 into shutdown. An internal
0.3µA current source always tries to pull this pin towards
VCC. When the ITH/RUN pin voltage is allowed to exceed
VSHDN, and VCC exceeds VTURNON, the LTC3803-5 begins
to operate and an internal clamp immediately pulls the
ITH/RUN pin up to about 0.7V. In operation, the ITH/RUN
pin voltage will vary from roughly 0.7V to 1.9V to represent current comparator thresholds from zero to maximum.
Internal Soft-Start
An internal soft-start feature is enabled whenever the
LTC3803-5 comes out of shutdown. Specifically, the
ITH/RUN voltage is clamped and is prevented from reaching maximum until roughly 0.7ms has passed. This
allows the input and output currents of LTC3803-5based power supplies to rise in a smooth and controlled
manner on start-up.
Powering the LTC3803-5
In the simplest case, the LTC3803-5 can be powered from
a high voltage supply through a resistor. A built-in shunt
regulator from the VCC pin to GND will draw as much
current as needed through this resistor to regulate the VCC
voltage to around 8V as long as the VCC pin is not forced
to sink more than 25mA. This shunt regulator is always
active, even when the LTC3803-5 is in shutdown, since it
serves the vital function of protecting the VCC pin from
seeing too much voltage.
The VCC pin must be bypassed to ground immediately
adjacent to the IC pins with a ceramic or tantalum capacitor. Proper supply bypassing is necessary to supply the
high transient currents required by the MOSFET gate
driver. 10µF is a good starting point.
Adjustable Slope Compensation
The LTC3803-5 injects a 5µA peak current ramp out
through its SENSE pin which can be used for slope
compensation in designs that require it. This current ramp
is approximately linear and begins at zero current at 6.5%
duty cycle, reaching peak current at 80% duty cycle.
Additional details are provided in the Applications Information section.
38035f
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LTC3803-5
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APPLICATIO S I FOR ATIO
Many LTC3803-5 application circuits can be derived from
the topology shown in Figure 2.
The LTC3803-5 itself imposes no limits on allowed power
output, input voltage VIN or desired regulated output
voltage VOUT; these are all determined by the ratings on the
external power components. The key factors are: Q1’s
maximum drain-source voltage (BVDSS), on-resistance
(RDS(ON)) and maximum drain current, T1’s saturation
flux level and winding insulation breakdown voltages, CIN
and COUT’s maximum working voltage, ESR, and maximum ripple current ratings, and D1 and RSENSE’s power
ratings.
VIN
D1
T1
VOUT
•
RVCC
CIN LPRI
LSEC
COUT
•
5
CVCC
1
CC
2
VCC
ITH/RUN NGATE
LTC3803-5
GND
SENSE
6
4
VFB
R1
3
Q1
RSL
RSENSE
R2
38035 F02
Figure 2. Typical LTC3803-5 Application Circuit
TRANSFORMER DESIGN CONSIDERATIONS
Transformer specification and design is perhaps the most
critical part of applying the LTC3803-5 successfully. In
addition to the usual list of caveats dealing with high
frequency power transformer design, the following should
prove useful.
Turns Ratios
Due to the use of the external feedback resistor divider
ratio to set output voltage, the user has relative freedom in
selecting transformer turns ratio to suit a given application. Simple ratios of small integers, e.g., 1:1, 2:1, 3:2, etc.
can be employed which yield more freedom in setting total
turns and mutual inductance. Simple integer turns ratios
also facilitate the use of “off-the-shelf” configurable transformers such as the Coiltronics VERSA-PACTM series in
applications with high input to output voltage ratios. For
example, if a 6-winding VERSA-PAC is used with three
windings in series on the primary and three windings in
parallel on the secondary, a 3:1 turns ratio will be achieved.
Turns ratio can be chosen on the basis of desired duty
cycle. However, remember that the input supply voltage
plus the secondary-to-primary referred version of the
flyback pulse (including leakage spike) must not exceed
the allowed external MOSFET breakdown rating.
SELECTING FEEDBACK RESISTOR DIVIDER VALUES
Leakage Inductance
The regulated output voltage is determined by the resistor
divider across VOUT (R1 and R2 in Figure 2). The ratio of
R2 to R1 needed to produce a desired VOUT can be
calculated:
Transformer leakage inductance (on either the primary or
secondary) causes a voltage spike to occur after the
output switch (Q1) turn-off. This is increasingly prominent at higher load currents, where more stored energy
must be dissipated. In some cases a “snubber” circuit will
be required to avoid overvoltage breakdown at the
MOSFET’s drain node. Application Note 19 is a good
reference on snubber design.
R2 =
VOUT – 0.8 V
• R1
0.8 V
Choose resistance values for R1 and R2 to be as large as
possible in order to minimize any efficiency loss due to the
static current drawn from VOUT, but just small enough so
that when VOUT is in regulation, the error caused by the
nonzero input current to the VFB pin is less than 1%. A
good rule of thumb is to choose R1 to be 80k or less.
A bifilar or similar winding technique is a good way to
minimize troublesome leakage inductances. However,
remember that this will limit the primary-to-secondary
breakdown voltage, so bifilar winding is not always
practical.
VERSA-PAC is a trademark of Coiltronics, Inc.
38035f
9
LTC3803-5
U
W
U U
APPLICATIO S I FOR ATIO
CURRENT SENSE RESISTOR CONSIDERATIONS
The external current sense resistor (RSENSE in Figure 2)
allows the user to optimize the current limit behavior for
the particular application. As the current sense resistor is
varied from several ohms down to tens of milliohms, peak
switch current goes from a fraction of an ampere to several
amperes. Care must be taken to ensure proper circuit
operation, especially with small current sense resistor
values.
For example, a peak switch current of 5A requires a sense
resistor of 0.020Ω. Note that the instantaneous peak
power in the sense resistor is 0.5W and it must be rated
accordingly. The LTC3803-5 has only a single sense line
to this resistor. Therefore, any parasitic resistance in the
ground side connection of the sense resistor will increase
its apparent value. In the case of a 0.020Ω sense resistor,
one milliohm of parasitic resistance will cause a 5%
reduction in peak switch current. So the resistance of
printed circuit copper traces and vias cannot necessarily
be ignored.
PROGRAMMABLE SLOPE COMPENSATION
The LTC3803-5 injects a ramping current through its
SENSE pin into an external slope compensation resistor
(RSL in Figure 2). This current ramp starts at zero right
after the NGATE pin has been high for the LTC3803-5’s
minimum duty cycle of 6.5%. The current rises linearly
towards a peak of 5µA at the maximum duty cycle of 80%,
shutting off once the NGATE pin goes low. A series resistor
(RSL) connecting the SENSE pin to the current sense
resistor (RSENSE) thus develops a ramping voltage drop.
From the perspective of the SENSE pin, this ramping
voltage adds to the voltage across the sense resistor,
effectively reducing the current comparator threshold in
proportion to duty cycle. This stabilizes the control loop
against subharmonic oscillation. The amount of reduction
in the current comparator threshold (∆VSENSE) can be
calculated using the following equation:
∆VSENSE =
Duty Cycle – 6.5%
• 5µA • RSL
73.5%
Note: LTC3803-5 enforces 6.5% < Duty Cycle < 80%.
A good starting value for RSL is 5.9k, which gives a 30mV
drop in current comparator threshold at 80% duty cycle.
Designs not needing slope compensation may replace RSL
with a short circuit.
VCC SHUNT REGULATOR
An internal shunt regulator allows the LTC3803-5 to be
powered through a single dropping resistor from VIN to
VCC, in conjunction with a bypass capacitor, CVCC, that
closely decouples VCC to GND (see Figure 3). The shunt
regulator can draw up to 25mA through the VCC pin to
GND to drop enough voltage across RVCC to regulate VCC
to around 8V. For applications where VIN is low enough
such that the static power dissipation in RVCC is acceptable, using the VCC shunt regulator is the simplest way to
power the LTC3803-5.
EXTERNAL PREREGULATOR
The circuit in Figure 4 shows another way to power the
LTC3803-5. An external series preregulator consisting of
series pass transistor Q1, Zener diode D1, and bias resistor RB brings VCC above the VCC turn-on threshold, enabling the LTC3803-5.
8V TO
75 VIN
VIN
RVCC
LTC3803-5
RB
100k
LTC3803-5
VCC
VCC
CVCC
Q1
MMBTA42
D1
6.8V
GND
CVCC
0.1µF
GND
38035 F04
38035 F03
Figure 3. Powering the LTC3803-5
Via the Internal Shunt Regulator
Figure 4. Powering the LTC3803-5
with an External Preregulator
38035f
10
LTC3803-5
U
TYPICAL APPLICATIO S
2W Isolated Housekeeping Telecom Converter
BAS516
PRIMARY SIDE
10V, 100mA
OUTPUT
T1
•
2.2µF
1µF
VIN
36V TO 75V
•
22k
806Ω
2.2µF
BAS516
9.2k
1nF
BAS516
1k
1
LTC3803-5
6
ITH/RUN NGATE
2
5
3
VCC
GND
VFB
SENSE
220k
•
SECONDARY SIDE
10V, 100mA
OUTPUT
SECONDARY
SIDE GROUND
FDC2512
T1: PULSE ENGINEERING PA0648
OR TYCO TTI8698
5.6k
4
1µF
0.1Ω
PRIMARY GROUND
38035 TA03
U
PACKAGE DESCRIPTIO
S6 Package
6-Lead Plastic TSOT-23
(Reference LTC DWG # 05-08-1636)
0.62
MAX
2.90 BSC
(NOTE 4)
0.95
REF
1.22 REF
3.85 MAX 2.62 REF
1.4 MIN
2.80 BSC
1.50 – 1.75
(NOTE 4)
PIN ONE ID
RECOMMENDED SOLDER PAD LAYOUT
PER IPC CALCULATOR
0.30 – 0.45
6 PLCS (NOTE 3)
0.95 BSC
0.80 – 0.90
0.20 BSC
0.01 – 0.10
1.00 MAX
DATUM ‘A’
0.30 – 0.50 REF
0.09 – 0.20
(NOTE 3)
1.90 BSC
S6 TSOT-23 0302
NOTE:
1. DIMENSIONS ARE IN MILLIMETERS
2. DRAWING NOT TO SCALE
3. DIMENSIONS ARE INCLUSIVE OF PLATING
4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
5. MOLD FLASH SHALL NOT EXCEED 0.254mm
6. JEDEC PACKAGE REFERENCE IS MO-193
38035f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
11
LTC3803-5
U
TYPICAL APPLICATIO S
90% Efficient Synchronous Flyback Converter
VIN
36V TO 72V
•
Q2
CIN
270k
Synchronous Flyback 5VOUT
Synchronous Flyback 3.3VOUT
VOUT*
3.3V
1.5A
T1
92
91
CO
•
91
D1
2
8.06k
3
ITH/RUN
6
GATE
LTC3803-5
5
VCC
GND
VFB
SENSE
25.5k*
RFB
VOUT
T1: PULSE ENGINEERING PA1006
Q1: FAIRCHILD FDC2512
Q2: VISHAY Si9803
4
Q1
90
•
0.1µF
560
5k
1µF
10V
90
EFFICIENCY (%)
33k 1
EFFICIENCY (%)
1n
89
89
88
87
38035 TA04a
RCS
86
RCS: VISHAY OR IRC, 80mΩ
D1: PHILIPS BAS516
CIN: TDK 1µF, 100V, X5R *FOR 5V OUTPUT CHANGE
CO: TDK 100µF, 6.3V, X5R RFB TO 42.2k
88
0.5
1.5
1.0
OUTPUT CURRENT (A)
2.0
38035 TA04b
85
0.5
1.0
1.5
2.0
OUTPUT CURRENT (A)
2.5
38035 TA04c
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
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SOT-23 Constant Frequency Current Mode Boost DC/DC
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550kHz Switching Frequency, 2.4V to 9.8V VIN Range
LT1950
Current Mode PWM Controller
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LT1952
Current Mode PWM Controller
Synchronous Controller for Forward Converters from
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LT3420
Photoflash Capacitor Charger with Automatic Refresh
Specialized Flyback Charges High Voltage Photoflash Capacitors
Quickly and Efficiently
LT3468/LT3468-1
Photoflash Capacitor Charger in 5-Pin SOT-23
Minimal Component Count, Uses Small Transformers;
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Synchronous Flyback Controller
High Efficiency (89%); Multiple Output with
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6A N-Channel MOSFET Driver
Gate Drive Adjustable from 5V to 8V, Adjustable Blanking
Prevents Ringing, 10-Lead MSSOP Package
Burst Mode is a registered trademark of Linear Technology Corporation. No RSENSE is a trademark of Linear Technology Corporation.
38035f
12
Linear Technology Corporation
LT/TP 1104 1K • PRINTED IN THE USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2004