CHAMP CM6800XIP Low start-up current pfc/pwm controller combo Datasheet

CM6800
LOW START-UP CURRENT PFC/PWM CONTROLLER COMBO
GENERAL DESCRIPTION
FEATURES
The CM6800 is a controller for power factor corrected,
‹
switched mode power suppliers. Power Factor Correction
Patent Number #5,565,761, #5,747,977, #5,742,151,
#5,804,950, #5,798,635
(PFC) allows the use of smaller, lower cost bulk
‹
Pin to pin compatible with ML4800 and FAN4800
capacitors, reduces power line loading and stress on the
‹
Additional folded-back current limit for PWM section.
switching FETs, and results in a power supply that fully
‹
23V Bi-CMOS process
compiles with IEC-1000-3-2 specifications. Intended as a
‹
VIN OK turn on PWM at 2.5V instead of 1.5V
BiCMOS version
‹
Internally synchronized leading edge PFC and trailing
of the industry-standard ML4824,
CM6800 includes circuits for the implementation of leading
edge,
average
current,
“boost”
type
power
factor
edge PWM in one IC
‹
correction and a trailing edge, pulse width modulator
Slew rate enhanced transconductance error amplifier for
ultra-fast PFC response
(PWM). Gate-driver with 1A capabilities minimizes the
‹
Low start-up current (100 μ A typ.)
need for external driver circuits. Low power requirements
‹
Low operating current (3.0mA type.)
improve efficiency and reduce component costs.
‹
Low total harmonic distortion, high PF
‹
An over-voltage comparator shuts down the PFC section
in the event of a sudden decrease in load. The PFC
Reduces ripple current in the storage capacitor between
the PFC and PWM sections
‹
section also includes peak current limiting and input
Average current, continuous or discontinuous boost
leading edge PFC
voltage brownout protection. The PWM section can be
‹
VCC OVP Comparator, Low Power Detect Comparator
operated in current or voltage mode, at up to 250kHz, and
‹
PWM configurable for current mode or voltage mode
includes an accurate 50% duty cycle limit to prevent
transformer saturation.
operation
‹
‹
CM6800 includes an additional folded-back current limit
Current fed gain modulator for improved noise immunity
Brown-out control, over-voltage protection, UVLO, and
soft start, and Reference OK
for PWM section to provide short circuit protection
function.
APPLICATIONS
‹
Desktop PC Power Supply
‹
Internet Server Power Supply
‹
IPC Power Supply
‹
UPS
‹
Battery Charger
‹
DC Motor Power Supply
‹
Monitor Power Supply
‹
‹
PIN CONFIGURATION
SOP-16 (S16) / PDIP-16 (P16)
Top View
1
IEAO
2
I AC
3
I SENSE
4
V RMS
Telecom System Power Supply
5
Distributed Power
2008/10/23 Rev. 2.1
VEAO
16
V FB
15
V REF
14
V CC
13
SS
PFC OUT
12
6
V DC
PW M OUT
11
7
RAM P1
GND
10
8
RAM P2
DC I LIMIT
Champion Microelectronic Corporation
9
Page 1
CM6800
LOW START-UP CURRENT PFC/PWM CONTROLLER COMBO
PIN DESCRIPTION
Pin No.
Symbol
Description
Min.
Operating Voltage
Typ.
Max.
Unit
1
IEAO
PFC transconductance current error amplifier output
0
4.25
V
2
IAC
PFC gain control reference input
0
1
mA
3
ISENSE
Current sense input to the PFC current limit comparator
-5
0.7
V
4
VRMS
Input for PFC RMS line voltage compensation
0
6
V
5
SS
Connection point for the PWM soft start capacitor
0
8
V
6
VDC
PWM voltage feedback input
0
8
V
7
RAMP 1
Oscillator timing node; timing set by RT CT
1.2
3.9
V
0
6
V
0
1
V
(RTCT)
8
When in current mode, this pin functions as the current sense
input; when in voltage mode, it is the PWM input from PFC
(PWM RAMP) output (feed forward ramp).
RAMP 2
9
DC ILIMIT
PWM current limit comparator input
10
GND
Ground
11
PWM OUT
PWM driver output
0
VCC
V
12
PFC OUT
PFC driver output
0
VCC
V
13
VCC
Positive supply
10
18
V
14
VREF
Buffered output for the internal 7.5V reference
15
VFB
PFC transconductance voltage error amplifier input
16
VEAO
2008/10/23 Rev. 2.1
PFC transconductance voltage error amplifier output
Champion Microelectronic Corporation
15
7.5
0
0
2.5
V
3
V
6
V
Page 2
CM6800
LOW START-UP CURRENT PFC/PWM CONTROLLER COMBO
BLOCK DIAGRAM (CM6800)
1
16
13
IEAO
VEAO
VCC
PFC OVP
17.9V
-
LOW POWER
DETECT
GMv
2
POWER
FACTOR
CORRECTOR
S
Q
R
Q
S
Q
R
Q
+
MPPFC
VCC
-
-1V
GAIN
VRMS
7
PFC CMP
.
-
+
PFC OUT
-
MODULATOR
4
3
GMi
+
IAC
14
REFERENCE
-
+
3.5K
.
+
2.75V
VREF
7.5V
.
TRI-FAULT
0.5V
-
2.5V
+
VCC OVP
-
VFB
+
+
15
-
0.3V
VCC
12
PFC ILIMIT
3.5K
ISENSE
MNPFC
GND
RAMP1
OSCILLATOR
CLK
350
SW SPST
PWM
CMP
-
1.5V
MPPWM
VDC
PWM OUT
Vcc
SS CMP
20uA
350
S
Q
R
Q
11
+
VFB
-
2.45V
+
.
SS
SW SPST
SW SPST
SW SPST
1.0V
MNPWM
+
5
VIN OK
GND
DC ILIMIT
PULSE
WIDTH
MODULATOR
VREF
Q
9
VCC
+
6
PWMOUT
LIMIT
-
8
PFCOUT
PWM
DUTY
DUTY CYCLE
RAMP2
S
DC ILIMIT
VCC
UVLO
GND
R
10
ORDERING INFORMATION
Part Number
CM6800GIP
Temperature Range
-40℃ to 125℃
16-Pin PDIP (P16)
Package
CM6800GIS
-40℃ to 125℃
16-Pin Wide SOP (S16)
CM6800XIP*
-40℃ to 125℃
16-Pin PDIP (P16)
CM6800XIS*
-40℃ to 125℃
16-Pin Wide SOP (S16)
*Note: G : Suffix for Pb Free Product
X : Suffix for Halogen Free Product
2008/10/23 Rev. 2.1
Champion Microelectronic Corporation
Page 3
CM6800
LOW START-UP CURRENT PFC/PWM CONTROLLER COMBO
ABSOLUTE MAXIMUM RATINGS
Absolute Maximum ratings are those values beyond which the device could be permanently damaged.
Parameter
VCC
IEAO
ISENSE Voltage
PFC OUT
PWMOUT
Voltage on Any Other Pin
IREF
IAC Input Current
Peak PFC OUT Current, Source or Sink
Peak PWM OUT Current, Source or Sink
Min.
Max.
Units
0
-5
GND – 0.3
GND – 0.3
20
7.5
0.7
VCC + 0.3
VCC + 0.3
V
V
V
V
V
GND – 0.3
VCC + 0.3
V
10
1
1
1
PFC OUT, PWM OUT Energy Per Cycle
1.5
mA
mA
A
A
μJ
Junction Temperature
150
℃
Storage Temperature Range
-65
150
℃
Operating Temperature Range
-40
125
℃
Lead Temperature (Soldering, 10 sec)
260
℃
Thermal Resistance (θJA)
Plastic DIP
Plastic SOIC
80
105
℃/W
Power Dissipation (PD) TA<50℃
800
mW
℃/W
ELECTRICAL CHARACTERISTICS Unless otherwise stated, these specifications apply Vcc=+15V, RT
= 30.16kΩ, CT =1000pF, TA=Operating Temperature Range (Note 1)
Symbol
Parameter
Test Conditions
CM6800
Min.
Typ.
Max.
Unit
Voltage Error Amplifier (gmv)
Input Voltage Range
Transconductance
0
VNONINV = VINV, VEAO = 3.75V
at room temp
Feedback Reference Voltage
Input Bias Current
Note 2
Output High Voltage
Source Current
Open Loop Gain
Power Supply Rejection Ratio
11V < VCC < 16.5V
μ mho
70
90
2.45
2.5
2.55
-1.0
-0.05
μA
5.8
6.0
V
VFB = 3V, VEAO = 6V
VFB = 1.5V, VEAO = 1.5V
V
50
Output Low Voltage
Sink Current
6
0.1
0.4
-35
-20
V
V
μA
30
40
μA
50
60
dB
50
60
dB
Current Error Amplifier (gmi)
Input Voltage Range
Transconductance
-1.5
VNONINV = VINV, VEAO = 3.75V
at room temp
50
Input Offset Voltage
-25
Output High Voltage
4.0
Output Low Voltage
2008/10/23 Rev. 2.1
85
V
100
μ mho
25
mV
4.25
1.0
Champion Microelectronic Corporation
0.7
V
1.2
Page 4
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CM6800
LOW START-UP CURRENT PFC/PWM CONTROLLER COMBO
ELECTRICAL CHARACTERISTICS (Conti.) Unless otherwise stated, these specifications apply
Vcc=+15V, RT = 30.16kΩ, CT =1000pF, TA=Operating Temperature Range (Note 1)
Symbol
Parameter
Test Conditions
Sink Current
ISENSE = +0.5V, IEAO = 4.0V
Source Current
ISENSE = -0.5V, IEAO = 1.5V
Open Loop Gain
Power Supply Rejection Ratio
11V < VCC < 16.5V
CM6800
Min.
Typ.
Max.
-65
-35
Unit
35
75
μA
μA
60
70
dB
60
75
dB
2.70
2.77
PFC OVP Comparator
Threshold Voltage
Hysteresis
230
2.85
V
290
mV
Low Power Detect Comparator
Threshold Voltage
0.2
0.3
0.4
V
17.5
17.9
18.5
V
1.40
1.5
1.65
V
2.65
2.75
2.85
V
2
4
ms
0.4
0.5
0.6
V
-1.10
-1.00
-0.90
V
20
100
mV
250
ns
VCC OVP Comparator
Threshold Voltage
Hysteresis
Tri-Fault Detect
Fault Detect HIGH
VFB=VFAULT DETECT LOW to
Time to Fault Detect HIGH
VFB=OPEN.470pF from VFB to GND
Fault Detect LOW
PFC ILIMIT Comparator
Threshold Voltage
(PFC ILIMIT VTH – Gain Modulator
Output)
Delay to Output (Note 4)
Overdrive Voltage = -100mV
DC ILIMIT Comparator
Threshold Voltage
Delay to Output (Note 4)
0.95
Overdrive Voltage = 100mV
1.0
1.05
250
V
ns
VIN OK Comparator
OK Threshold Voltage
Hysteresis
GAIN Modulator
IAC = 100 μ A, VRMS =0, VFB = 1V
at room temp
IAC = 100 μ A, VRMS = 1.1V, VFB = 1V
Gain (Note 3)
at room temp
IAC = 150 μ A, VRMS = 1.8V, VFB = 1V
at room temp
IAC = 300 μ A, VRMS = 3.3V, VFB = 1V
Bandwidth
Output Voltage =
3.5K*(ISENSE-IOFFSET)
2008/10/23 Rev. 2.1
at room temp
IAC = 100 μ A
IAC = 250 μ A, VRMS = 1.1V, VFB = 1V
Champion Microelectronic Corporation
2.35
2.45
2.55
V
0.95
1.2
1.4
V
0.70
0.84
0.95
1.80
2.00
2.20
0.90
1.00
1.10
0.25
0.32
0.40
10
0.84
0.90
MHz
0.95
Page 5
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CM6800
LOW START-UP CURRENT PFC/PWM CONTROLLER COMBO
ELECTRICAL CHARACTERISTICS
(Conti.) Unless otherwise stated, these specifications apply
Vcc=+15V, RT = 30.16kΩ, CT =1000pF, TA=Operating Temperature Range (Note 1)
Symbol
Parameter
Test Conditions
Initial Accuracy
Oscillator
TA = 25℃
Voltage Stability
11V < VCC < 16.5V
CM6800
Min.
Typ.
60
Max.
70
1
Temperature Stability
Line, Temp
52
Ramp Valley to Peak Voltage
%
74
2.5
PFC Dead Time (Note 4)
CT Discharge Current
Load Regulation
kHz
V
360
640
ns
VRAMP2 = 0V, VRAMP1 = 2.5V
6.5
15
mA
Reference
TA = 25℃, I(VREF) = 1mA
7.4
7.5
7.6
V
11V < VCC < 16.5V
0mA < I(VREF) < 7mA; TA = 0℃~70℃
10
25
mV
10
20
mV
0mA < I(VREF) < 5mA; TA = -40℃~85℃
10
20
mV
Output Voltage
Line Regulation
kHz
%
2
Total Variation
Unit
Temperature Stability
0.4
Total Variation
Line, Load, Temp
TJ = 125℃, 1000HRs
Long Term Stability
%
7.35
7.65
V
5
25
mV
0
%
PFC
Minimum Duty Cycle
VIEAO > 4.0V
Maximum Duty Cycle
VIEAO < 1.2V
Output Low Rdson
92
%
IOUT = -20mA at room temp
15
ohm
IOUT = -100mA at room temp
15
ohm
0.8
V
IOUT = 10mA, VCC = 9V at room temp
Output High Rdson
95
0.4
IOUT = 20mA at room temp
15
20
ohm
IOUT = 100mA at room temp
15
20
ohm
CL = 1000pF
50
Rise/Fall Time (Note 4)
ns
PWM
Duty Cycle Range
0-43
Output Low Rdson
0-49
%
IOUT = -20mA at room temp
15
ohm
IOUT = -100mA at room temp
15
ohm
0.8
V
IOUT = 10mA, VCC = 9V
Output High Rdson
0-47
0.4
IOUT = 20mA at room temp
15
20
ohm
IOUT = 100mA at room temp
15
20
ohm
Rise/Fall Time (Note 4)
CL = 1000pF
PWM Comparator Level Shift
50
1.4
ns
1.55
1.7
V
VCC = 12V, CL = 0 at room temp
100
150
μA
14V, CL = 0
3.0
7.0
mA
Supply
Start-Up Current
Operating Current
Undervoltage Lockout Threshold
CM6800
12.74
13
13.26
V
Undervoltage Lockout Hysteresis
CM6800
2.85
3.0
3.15
V
Note 1: Limits are guaranteed by 100% testing, sampling, or correlation with worst-case test conditions.
Note 2: Includes all bias currents to other circuits connected to the VFB pin.
Note 3: Gain = K x 5.375V; K = (ISENSE – IOFFSET) x [IAC (VEAO – 0.625)]-1; VEAOMAX = 6V
Note 4: Guaranteed by design, not 100% production test.
2008/10/23 Rev. 2.1
Champion Microelectronic Corporation
Page 6
CM6800
LOW START-UP CURRENT PFC/PWM CONTROLLER COMBO
TYPICAL PERFORMANCE CHARACTERISTIC
100
127
Transconductance (um ho)
Transconductance (umho)
120
113
106
99
92
85
78
71
90
80
70
60
50
40
30
20
10
64
0
57
2
2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8 2.9
-500
3
0
VFB (V)
500
ISENSE(m V)
Voltage Error Amplifier (gmv) Transconductance
Current Error Amplifier (gmi) Transconductance
0.4
2
0.35
1.8
0.3
1.6
0.25
1.4
0.2
1.2
Gain
Variable Gain Block Constant (K)
2.2
0.15
1
0.8
0.1
0.6
0.05
0.4
0
0
0.5
1
1.5
2
2.5
3
3.5
4
4.5
5
VRMS (V)
0.2
0
0
0.5
1
1.5
2
2.5
3
3.5
4
4.5
5
VRMS (V)
Gain Modulator Transfer Characteristic (K)
K=
2008/10/23 Rev. 2.1
-1
IGAINMOD − IOFFSET
mV
IAC x (6 - 0.625)
Gain
Gain =
Champion Microelectronic Corporation
ISENSE − IOFFSET
IAC
Page 7
CM6800
LOW START-UP CURRENT PFC/PWM CONTROLLER COMBO
Functional Description
The CM6800 consists of an average current controlled,
continuous boost Power Factor Correction (PFC) front end
and a synchronized Pulse Width Modulator (PWM) back
end. The PWM can be used in either current or voltage
mode. In voltage mode, feedforward from the PFC output
buss can be used to improve the PWM’s line regulation. In
either mode, the PWM stage uses conventional trailing
edge duty cycle modulation, while the PFC uses leading
edge modulation. This patented leading/trailing edge
modulation technique results in a higher usable PFC error
amplifier bandwidth, and can significantly reduce the size of
the PFC DC buss capacitor.
The synchronized of the PWM with the PFC simplifies the
PWM compensation due to the controlled ripple on the PFC
output capacitor (the PWM input capacitor). The PWM
section of the CM6800 runs at the same frequency as the
PFC.
In addition to power factor correction, a number of
protection features have been built into the CM6800. These
include soft-start, PFC overvoltage protection, peak current
limiting, brownout protection, duty cycle limiting, and
under-voltage lockout.
Since the boost converter topology in the CM6800 PFC is of
the current-averaging type, no slope compensation is
required.
PFC Section
Power Factor Correction
Power factor correction makes a nonlinear load look like a
resistive load to the AC line. For a resistor, the current
drawn from the line is in phase with and proportional to the
line voltage, so the power factor is unity (one). A common
class of nonlinear load is the input of most power supplies,
which use a bridge rectifier and capacitive input filter fed
from the line. The peak-charging effect, which occurs on
the input filter capacitor in these supplies, causes brief
high-amplitude pulses of current to flow from the power line,
rather than a sinusoidal current in phase with the line
voltage. Such supplies present a power factor to the line of
less than one (i.e. they cause significant current harmonics
of the power line frequency to appear at their input). If the
input current drawn by such a supply (or any other
nonlinear load) can be made to follow the input voltage in
instantaneous amplitude, it will appear resistive to the AC
line and a unity power factor will be achieved.
To hold the input current draw of a device drawing power
from the AC line in phase with and proportional to the input
voltage, a way must be found to prevent that device from
loading the line except in proportion to the instantaneous
line voltage. The PFC section of the CM6800 uses a
boost-mode DC-DC converter to accomplish this. The input
to the converter is the full wave rectified AC line voltage. No
bulk filtering is applied following the bridge rectifier, so the
input voltage to the boost converter ranges (at twice line
frequency) from zero volts to the peak value of the AC input
and back to zero. By forcing the boost converter to meet
two simultaneous conditions, it is possible to ensure that
the current drawn from the power line is proportional to the
input
2008/10/23 Rev. 2.1
line voltage. One of these conditions is that the output
voltage of the boost converter must be set higher than the
peak value of the line voltage. A commonly used value is
385VDC, to allow for a high line of 270VACrms. The other
condition is that the current drawn from the line at any given
instant must be proportional to the line voltage. Establishing
a suitable voltage control loop for the converter, which in turn
drives a current error amplifier and switching output driver
satisfies the first of these requirements. The second
requirement is met by using the rectified AC line voltage to
modulate the output of the voltage control loop. Such
modulation causes the current error amplifier to command a
power stage current that varies directly with the input voltage.
In order to prevent ripple, which will necessarily appear at the
output of boost circuit (typically about 10VP-P ripple at low
frequency on a 385V DC level), from introducing distortion
back through the voltage error amplifier, the bandwidth of the
voltage loop is deliberately kept low. A final refinement is to
adjust the overall gain of the PFC such to be proportional to
1/VIN^2, which linearizes the transfer function of the system
as the AC input to voltage varies.
Gain Modulator
Figure 1 shows a block diagram of the PFC section of the
CM6800. The gain modulator is the heart of the PFC, as it is
this circuit block which controls the response of the current
loop to line voltage waveform and frequency, rms line
voltage, and PFC output voltages. There are three inputs to
the gain modulator. These are:
1. A current representing the instantaneous input voltage
(amplitude and waveshape) to the PFC. The rectified AC
input sine wave is converted to a proportional current via a
resistor and is then fed into the gain modulator at IAC.
Sampling current in this way minimizes ground noise, as is
required in high power switching power conversion
environments. The gain modulator responds linearly to this
current.
2. A voltage proportional to the long-term RMS AC line
voltage, derived from the rectified line voltage after scaling
and filtering. This signal is presented to the gain modulator
at VRMS. The gain modulator’s output is inversely
2
proportional to VRMS (except at unusually low values of
VRMS where special gain contouring takes over, to limit
power dissipation of the circuit components under heavy
brownout conditions). The relationship between VRMS and
gain is called K, and is illustrated in the Typical
Performance Characteristics.
3. The output of the voltage error amplifier, VEAO. The gain
modulator responds linearly to variations in this voltage.
Champion Microelectronic Corporation
Page 8
CM6800
LOW START-UP CURRENT PFC/PWM CONTROLLER COMBO
The output of the gain modulator is a current signal, in the
form of a full wave rectified sinusoid at twice the line
frequency. This current is applied to the virtual-ground
(negative) input of the current error amplifier. In this way the
gain modulator forms the reference for the current error
loop, and ultimately controls the instantaneous current draw
of the PFC form the power line. The general for of the
output of the gain modulator is:
IGAINMOD =
IAC × VEAO
x 1V
VRMS2
(1)
More exactly, the output current of the gain modulator is
given by:
IGAINMOD = K x (VEAO – 0.625V) x IAC
-1
Where K is in units of V
Note that the output current of the gain modulator is limited
around 228.47 μ A and the maximum output voltage of the
gain modulator is limited to 228.47uA x 3.5K=0.8V. This
0.8V also will determine the maximum input power.
However, IGAINMOD cannot be measured directly from ISENSE.
ISENSE = IGAINMOD-IOFFSET and IOFFSET can only be measured
when VEAO is less than 0.5V and IGAINMOD is 0A. Typical
IOFFSET is around 60uA.
Selecting RAC for IAC pin
IAC pin is the input of the gain modulator. IAC also is a
current mirror input and it requires current input. By
selecting a proper resistor RAC, it will provide a good sine
wave current derived from the line voltage and it also helps
program the maximum input power and minimum input line
voltage.
RAC=Vin peak x 7.9K. For example, if the minimum line
voltage is 80VAC, the RAC=80 x 1.414 x 7.9K=894Kohm.
Current Error Amplifier, IEAO
The current error amplifier’s output controls the PFC duty
cycle to keep the average current through the boost
inductor a linear function of the line voltage. At the inverting
input to the current error amplifier, the output current of the
gain modulator is summed with a current which results from
a negative voltage being impressed upon the ISENSE pin.
The negative voltage on ISENSE represents the sum of all
currents flowing in the PFC circuit, and is typically derived
from a current sense resistor in series with the negative
terminal of the input bridge rectifier.
2008/10/23 Rev. 2.1
In higher power applications, two current transformers are
sometimes used, one to monitor the IF of the boost diode. As
stated above, the inverting input of the current error amplifier
is a virtual ground. Given this fact, and the arrangement of
the duty cycle modulator polarities internal to the PFC, an
increase in positive current from the gain modulator will
cause the output stage to increase its duty cycle until the
voltage on ISENSE is adequately negative to cancel this
increased current. Similarly, if the gain modulator’s output
decreases, the output duty cycle will decrease, to achieve a
less negative voltage on the ISENSE pin.
Cycle-By-Cycle Current Limiter and Selecting RS
The ISENSE pin, as well as being a part of the current feedback
loop, is a direct input to the cycle-by-cycle current limiter for
the PFC section. Should the input voltage at this pin ever be
more negative than –1V, the output of the PFC will be
disabled until the protection flip-flop is reset by the clock
pulse at the start of the next PFC power cycle.
RS is the sensing resistor of the PFC boost converter. During
the steady state, line input current x RS = IGAINMOD x 3.5K.
Since the maximum output voltage of the gain modulator is
IGAINMOD max x 3.5K= 0.8V during the steady state, RS x line
input current will be limited below 0.8V as well. Therefore, to
choose RS, we use the following equation:
RS =0.8V x Vinpeak/(2x Line Input power)
For example, if the minimum input voltage is 80VAC, and the
maximum input rms power is 200Watt, RS = (0.8V x 80V x
1.414)/(2 x 200) = 0.226 ohm.
PFC OVP
In the CM6800, PFC OVP comparator serves to protect the
power circuit from being subjected to excessive voltages if
the load should suddenly change. A resistor divider from the
high voltage DC output of the PFC is fed to VFB. When the
voltage on VFB exceeds 2.75V, the PFC output driver is shut
down. The PWM section will continue to operate. The OVP
comparator has 250mV of hysteresis, and the PFC will not
restart until the voltage at VFB drops below 2.50V. The VFB
power components and the CM6800 are within their safe
operating voltages, but not so low as to interfere with the
boost voltage regulation loop. Also, VCC OVP can be served
as a redundant PFCOVP protection. VCC OVP threshold is
17.9V with 1.5V hysteresis.
Champion Microelectronic Corporation
Page 9
CM6800
LOW START-UP CURRENT PFC/PWM CONTROLLER COMBO
1
16
VEAO
13
IEAO
VCC
PFC OVP
GMv
17.9V
-
LOW POWER
DETECT
0.5V
2.5V
GMi
PFC CMP
+
+
.
2
IAC
VRMS
4
3
7
-
14
REFERENCE
-
+
3.5K
VREF
7.5V
.
2.75V
POWER
FACTOR
CORRECTOR
TRI-FAULT
.
+
VCC OVP
-
VFB
+
+
18
-
0.3V
VCC
S
Q
R
Q
+
MPPFC
-
-1V
GAIN
+
-
MODULATOR
VCC
PFC OUT
S
Q
R
Q
12
PFC ILIMIT
ISENSE
RAMP1
3.5K
MNPFC
GND
OSCILLATOR
CLK
Figure 1. PFC Section Block Diagram
Error Amplifier Compensation
The PWM loading of the PFC can be modeled as a
negative resistor; an increase in input voltage to the PWM
causes a decrease in the input current. This response
dictates the proper compensation of the two
transconductance error amplifiers. Figure 2 shows the types
of compensation networks most commonly used for the
voltage and current error amplifiers, along with their
respective return points. The current loop compensation is
returned to VREF to produce a soft-start characteristic on the
PFC: as the reference voltage comes up from zero volts, it
creates a differentiated voltage on IEAO which prevents the
PFC from immediately demanding a full duty cycle on its
boost converter.
PFC Voltage Loop
There are two major concerns when compensating the
voltage loop error amplifier, VEAO; stability and transient
response. Optimizing interaction between transient
response and stability requires that the error amplifier’s
open-loop crossover frequency should be 1/2 that of the
line frequency, or 23Hz for a 47Hz line (lowest anticipated
international power frequency). The gain vs. input voltage
of the CM6800’s voltage error amplifier, VEAO has a
specially shaped non-linearity such that under steady-state
operating conditions the transconductance of the error
amplifier is at a local minimum. Rapid perturbation in line or
load conditions will cause the input to the voltage error
amplifier (VFB) to deviate from its 2.5V (nominal) value. If
this happens, the transconductance of the voltage error
amplifier will increase significantly, as shown in the Typical
Performance
Characteristics.
This
raises
the
gain-bandwidth product of the voltage loop, resulting in a
much more rapid voltage loop response to such
perturbations than would occur with a conventional linear
gain characteristics.
2008/10/23 Rev. 2.1
The Voltage Loop Gain (S)
ΔVOUT ΔVFB ΔVEAO
*
*
ΔVEAO ΔVOUT ΔVFB
PIN * 2.5V
≈
* GMV * ZCV
2
VOUTDC * ΔVEAO * S * CDC
=
ZCV: Compensation Net Work for the Voltage Loop
GMv: Transconductance of VEAO
PIN: Average PFC Input Power
VOUTDC: PFC Boost Output Voltage; typical designed value is
380V.
CDC: PFC Boost Output Capacitor
PFC Current Loop
The current amplifier, IEAO compensation is similar to that of
the voltage error amplifier, VEAO with exception of the choice
of crossover frequency. The crossover frequency of the
current amplifier should be at least 10 times that of the
voltage amplifier, to prevent interaction with the voltage loop.
It should also be limited to less than 1/6th that of the
switching frequency, e.g. 16.7kHz for a 100kHz switching
frequency.
The Current Loop Gain (S)
ΔVISENSE ΔDOFF ΔIEAO
*
*
ΔDOFF
ΔIEAO ΔISENSE
VOUTDC * RS
≈
* GMI * ZCI
S * L * 2.5V
=
Champion Microelectronic Corporation
Page 10
CM6800
LOW START-UP CURRENT PFC/PWM CONTROLLER COMBO
ZCI: Compensation Net Work for the Current Loop
GMI: Transconductance of IEAO
VOUTDC: PFC Boost Output Voltage; typical designed value
is 380V and we use the worst condition to calculate the ZCI
RS: The Sensing Resistor of the Boost Converter
2.5V: The Amplitude of the PFC Leading Modulation Ramp
L: The Boost Inductor
There is a modest degree of gain contouring applied to the
transfer characteristic of the current error amplifier, to
increase its speed of response to current-loop
perturbations. However, the boost inductor will usually be
the dominant factor in overall current loop response.
Therefore, this contouring is significantly less marked than
that of the voltage error amplifier. This is illustrated in the
Typical Performance Characteristics.
Figure 2. Compensation Network Connections for the
Voltage and Current Error Amplifiers
2008/10/23 Rev. 2.1
ISENSE Filter, the RC filter between RS and ISENSE :
There are 2 purposes to add a filter at ISENSE pin:
1.) Protection: During start up or inrush current
conditions, it will have a large voltage cross Rs
which is the sensing resistor of the PFC boost
converter. It requires the ISENSE Filter to attenuate
the energy.
2.) To reduce L, the Boost Inductor: The ISENSE Filter
also can reduce the Boost Inductor value since the
ISENSE Filter behaves like an integrator before going
ISENSE which is the input of the current error
amplifier, IEAO.
The ISENSE Filter is a RC filter. The resistor value of the ISENSE
Filter is between 100 ohm and 50 ohm because IOFFSET x the
resistor can generate an offset voltage of IEAO. By selecting
RFILTER equal to 50 ohm will keep the offset of the IEAO less
than 5mV. Usually, we design the pole of ISENSE Filter at
fpfc/6, one sixth of the PFC switching frequency. Therefore,
the boost inductor can be reduced 6 times without disturbing
the stability. Therefore, the capacitor of the ISENSE Filter,
CFILTER, will be around 283nF.
Figure 3. External Component Connections to VCC
Champion Microelectronic Corporation
Page 11
CM6800
LOW START-UP CURRENT PFC/PWM CONTROLLER COMBO
Oscillator (RAMP1)
The oscillator frequency is determined by the values of RT
and CT, which determine the ramp and off-time of the
oscillator output clock:
fOSC =
1
tRAMP + tDEADTIME
The dead time of the oscillator is derived from the following
equation:
tRAMP = CT x RT x In VREF − 1.25
VREF − 3.75
at VREF = 7.5V:
tRAMP = CT x RT x 0.51
The dead time of the oscillator may be determined using:
tDEADTIME =
2.5V
x CT = 943 x CT
2.65mA
The dead time is so small (tRAMP >> tDEADTIME ) that the
operating frequency can typically be approximately by:
fOSC =
1
tRAMP
EXAMPLE:
For the application circuit shown in the datasheet, with the
oscillator running at:
fOSC = 67.5kHz =
1
tRAMP
Solving for CT x RT yields 2.9 x 10-5. Selecting standard
components values, CT = 470pF, and RT = 61.9kΩ
The dead time of the oscillator adds to the Maximum PWM
Duty Cycle (it is an input to the Duty Cycle Limiter). With
zero oscillator dead time, the Maximum PWM Duty Cycle is
typically 45%. In many applications, care should be taken
that CT not be made so large as to extend the Maximum
Duty Cycle beyond 50%. This can be accomplished by
using a stable 390pF capacitor for CT.
PWM Section
Pulse Width Modulator
The PWM section of the CM6800 is straightforward, but
there are several points which should be noted. Foremost
among these is its inherent synchronization to the PFC
section of the device, from which it also derives its basic
timing. The PWM is capable of current-mode or
voltage-mode operation. In current-mode applications, the
PWM ramp (RAMP2) is usually derived directly from a
current sensing resistor or current transformer in the
primary of the output stage, and is thereby representative
2008/10/23 Rev. 2.1
of the current flowing in the converter’s output stage.
DCILIMIT, which provides cycle-by-cycle current limiting, is
typically connected to RAMP2 in such applications. For
voltage-mode, operation or certain specialized applications,
RAMP2 can be connected to a separate RC timing network
to generate a voltage ramp against which VDC will be
compared. Under these conditions, the use of voltage
feedforward from the PFC buss can assist in line regulation
accuracy and response. As in current mode operation, the
DC ILIMIT input is used for output stage overcurrent protection.
No voltage error amplifier is included in the PWM stage of
the CM6800, as this function is generally performed on the
output side of the PWM’s isolation boundary. To facilitate the
design of optocoupler feedback circuitry, an offset has been
built into the PWM’s RAMP2 input which allows VDC to
command a zero percent duty cycle for input voltages below
1.25V.
PWM Current Limit
The DC ILIMIT pin is a direct input to the cycle-by-cycle current
limiter for the PWM section. Should the input voltage at this
pin ever exceed 1V, the output flip-flop is reset by the clock
pulse at the start of the next PWM power cycle. Beside, the
cycle-by-cycle current, when the DC ILIMIT triggered the
cycle-by-cycle current, it also softly discharge the voltage of
soft start capacitor. It will limit PWM duty cycle mode.
Therefore, the power dissipation will be reduced during the
dead short condition.
VIN OK Comparator
The VIN OK comparator monitors the DC output of the PFC
and inhibits the PWM if this voltage on VFB is less than its
nominal 2.45V. Once this voltage reaches 2.45V, which
corresponds to the PFC output capacitor being charged to its
rated boost voltage, the soft-start begins.
PWM Control (RAMP2)
When the PWM section is used in current mode, RAMP2 is
generally used as the sampling point for a voltage
representing the current on the primary of the PWM’s output
transformer, derived either by a current sensing resistor or a
current transformer. In voltage mode, it is the input for a
ramp voltage generated by a second set of timing
components (RRAMP2, CRAMP2),that will have a minimum value
of zero volts and should have a peak value of approximately
5V. In voltage mode operation, feedforward from the PFC
output buss is an excellent way to derive the timing ramp for
the PWM stage.
Soft Start
Start-up of the PWM is controlled by the selection of the
external capacitor at SS. A current source of 20 μ A supplies
the charging current for the capacitor, and start-up of the
PWM begins at 1.25V. Start-up delay can be programmed by
the following equation:
CSS = tDELAY x
20 μA
1.25V
Champion Microelectronic Corporation
Page 12
CM6800
LOW START-UP CURRENT PFC/PWM CONTROLLER COMBO
where CSS is the required soft start capacitance, and the
tDEALY is the desired start-up delay.
It is important that the time constant of the PWM soft-start
allow the PFC time to generate sufficient output power for
the PWM section. The PWM start-up delay should be at
least 5ms.
If anything goes wrong, and VCC goes beyond 17.9V, the
PFC gate (pin 12) drive goes low and the PWM gate drive
(pin 11) remains function. The resistor’s value must be
chosen to meet the operating current requirement of the
CM6800 itself (5mA, max.) plus the current required by the
two gate driver outputs.
Solving for the minimum value of CSS:
CSS = 5ms x
20 μA = 80nF
1.25V
Caution should be exercised when using this minimum soft
start capacitance value because premature charging of the
SS capacitor and activation of the PWM section can result if
VFB is in the hysteresis band of the VIN OK comparator at
start-up. The magnitude of VFB at start-up is related both to
line voltage and nominal PFC output voltage. Typically, a
1.0 μ F soft start capacitor will allow time for VFB and PFC
out to reach their nominal values prior to activation of the
PWM section at line voltages between 90Vrms and
265Vrms.
Generating VCC
After turning on CM6800 at 13V, the operating voltage can
vary from 10V to 17.9V. The threshold voltage of VCC OVP
comparator is 17.9V. The hysteresis of VCC OVP is 1.5V.
When VCC see 17.9V, PFCOUT will be low, and PWM
section will not be disturbed. That’s the two ways to
generate VCC. One way is to use auxiliary power supply
around 15V, and the other way is to use bootstrap winding
to self-bias CM6800 system. The bootstrap winding can be
either taped from PFC boost choke or from the transformer
of the DC to DC stage.
2008/10/23 Rev. 2.1
The ratio of winding transformer for the bootstrap should be
set between 18V and 15V. A filter network is recommended
between VCC (pin 13) and bootstrap winding. The resistor of
the filter can be set as following.
RFILTER x IVCC ~ 2V, IVCC = IOP + (QPFCFET + QPWMFET ) x fsw
IOP = 3mA (typ.)
EXAMPLE:
With a wanting voltage called, VBIAS ,of 18V, a VCC of 15V
and the CM6800 driving a total gate charge of 90nC at
100kHz (e.g. 1 IRF840 MOSFET and 2 IRF820 MOSFET),
the gate driver current required is:
IGATEDRIVE = 100kHz x 90nC = 9mA
RBIAS =
VBIAS − VCC
ICC + IG
RBIAS =
18V − 15V
5mA + 9mA
Choose RBIAS = 214Ω
The CM6800 should be locally bypassed with a 1.0 μ F
ceramic capacitor. In most applications, an electrolytic
capacitor of between 47 μ F and 220 μ F is also required
across the part, both for filtering and as part of the start-up
bootstrap circuitry.
Champion Microelectronic Corporation
Page 13
CM6800
LOW START-UP CURRENT PFC/PWM CONTROLLER COMBO
Leading/Trailing Modulation
Conventional Pulse Width Modulation (PWM) techniques
employ trailing edge modulation in which the switch will turn
on right after the trailing edge of the system clock. The error
amplifier output is then compared with the modulating ramp
up. The effective duty cycle of the trailing edge modulation
is determined during the ON time of the switch. Figure 4
shows a typical trailing edge control scheme.
One of the advantages of this control technique is that it
required only one system clock. Switch 1(SW1) turns off and
switch 2 (SW2) turns on at the same instant to minimize the
momentary “no-load” period, thus lowering ripple voltage
generated by the switching action. With such synchronized
switching, the ripple voltage of the first stage is reduced.
Calculation and evaluation have shown that the 120Hz
component of the PFC’s output ripple voltage can be
reduced by as much as 30% using this method.
In case of leading edge modulation, the switch is turned
OFF right at the leading edge of the system clock. When
the modulating ramp reaches the level of the error amplifier
output voltage, the switch will be turned ON. The effective
duty-cycle of the leading edge modulation is determined
during OFF time of the switch. Figure 5 shows a leading
edge control scheme.
2008/10/23 Rev. 2.1
Champion Microelectronic Corporation
Page 14
CM6800
LOW START-UP CURRENT PFC/PWM CONTROLLER COMBO
APPLICATION CIRCUIT (Voltage Mode)
R5
IVIN_EMC
EMC FILTER
L2
R3
L3
RT1
IVIN
D4
PFC_VIN
IVIN
L1
IL1
PFC_VIN
IAC
D5
Q3
VIN
AC
R2
C3
C8
C2
R1
R10
R12
R14
C33
100n
D12
Q1
Q2N2222
PFC_DC
R13
Q2
Q2N904
R15
R23
R18
D7
1N4002
1N4002
Q4
R22
R25
10k
D10
R26
18k
PFC_Vout
R16A
MUR1100
C10
C23
470p
R24
75
D6
PFC_Vout
IC10
22
1N4148
R11
D5
IBOOT
R17A
22
VFB
C55A
C41
R65A
C30
R58
R64
R59
C43
IEAO
U2 CM6800/01/24
1
16
IEAO
VEAO
2
ISENSE
R60
3
VRMS
SS
VDC
C47 VREF
VRMS
5
SS
6
C44
R56
R57
C45
C48
C49
8
VCC
GND
RAMP2
ILIMIT
VREF
VCC
13
12
PFC-OUT
RAMP1
VREF
14
VCC
VDC PWM-OUT
7
VEAO
15
VFB
I-SENSE VREF
4
C46
R66
IAC
C52
C53
1u
100n
VCC
C54
11
10
PWM_OUT
R63
ZD2
9
C57
C56
R62
ILIMIT
ILIMIT
C50
R61
470
C51
R44
C4
ISO1
VDC
C14
PWM_IN
PFC_Vout
C7
10n
R27
100k
10n
C22
10n
R34
C38
4.7
R49
IL4
D9A
D8
D13
T1
D9B
L4
L5
R35
4.7
IC17
IC18
C17
C18
C40
R43
C39
ILOAD
C19
MUR1100
MUR1100
R46
PWM_Vout
C22
10n
R45
PWM_Rload
500m
U1
CM431
R48
C15
10n
VCC
IBIAS
C34
100n
D16
Q6
Q2N2222
PWM_DC
PWM_OUT
R32A
R32
VCC
1N4148
Q7
Q2N904
C31
R33
R28
T 2:3
Q3
22
R29
10k
ILIMIT
R31
ZD1
6.8V
2008/10/23 Rev. 2.1
Champion Microelectronic Corporation
Page 15
CM6800
LOW START-UP CURRENT PFC/PWM CONTROLLER COMBO
APPLICATION CIRCUIT (Current Mode)
R5
IVIN_EMC
EMC FILTER
L2
R3
L3
RT1
IVIN
D4
PFC_VIN
IVIN
L1
IL1
PFC_VIN
IAC
D5
Q3
VIN
AC
R2
C3
C8
C2
R1
R10
R12
R14
C33
100n
D12
Q1
Q2N2222
PFC_DC
R13
Q2
Q2N904
R15
R23
R18
D7
1N4002
1N4002
PFC_Vout
IC10
R22
R25
10k
D10
R26
18k
R16A
MUR1100
C10
C23
470p
R24
75
D6
PFC_Vout
Q4
22
1N4148
R11
D5
IBOOT
R17A
22
VFB
C55A
C41
R65A
C30
R58
R64
R59
C43
IEAO
U2 CM6800/01/24
1
16
IEAO
VEAO
2
ISENSE
3
VRMS
SS
R66
I-SENSE VREF
4
C46
VDC
C47 VREF
R57
5
SS
6
R67
C48
C49
PFC-OUT
VDC PWM-OUT
8
ILIMIT
C45
VCC
VRMS
7
R56
C44
VFB
RAMP1
GND
RAMP2
ILIMIT
VEAO
15
VREF
14
VREF
VCC
13
12
C52
C53
1u
100n
VCC
C54
11
10
PWM_OUT
R63
ZD2
9
C57
C56
R62
C50
ILIMIT
R61 470
ILIMIT
R60
IAC
C51
R68
R44
C4
ISO1
VDC
C14
PWM_IN
PFC_Vout
C7
10n
R27
100k
C22
10n
10n
R34
C38
4.7
R49
IL4
D9A
D8
D13
MUR1100
T1
D9B
MUR1100
L4
L5
R46
PWM_Vout
IC17
IC18
C17
C18
ILOAD
R35
4.7
C40
R43
C39
C19
C22
10n
R45
PWM_Rload
500m
U1
CM431
R48
C15
10n
VCC
IBIAS
C34
100n
D16
R32A
R32
VCC
1N4148
Q6
Q2N2222
PWM_DC
R28
Q7
Q2N904
22
C31
R33
T 2:3
PWM_OUT
Q3
R29
10k
ILIMIT
R31
ZD1
6.8V
2008/10/23 Rev. 2.1
Champion Microelectronic Corporation
Page 16
CM6800
LOW START-UP CURRENT PFC/PWM CONTROLLER COMBO
PACKAGE DIMENSION
16-PIN PDIP (P16)
PIN 1 ID
θ
θ
16-PIN SOP (S16), 0.300” Wide Body
PIN 1 ID
θ
2008/10/23 Rev. 2.1
Champion Microelectronic Corporation
Page 17
CM6800
LOW START-UP CURRENT PFC/PWM CONTROLLER COMBO
IMPORTANT NOTICE
Champion Microelectronic Corporation (CMC) reserves the right to make changes to its products or to
discontinue any integrated circuit product or service without notice, and advises its customers to obtain
the latest version of relevant information to verify, before placing orders, that the information being relied
on is current.
A few applications using integrated circuit products may involve potential risks of death, personal injury, or
severe property or environmental damage. CMC integrated circuit products are not designed, intended,
authorized, or warranted to be suitable for use in life-support applications, devices or systems or other
critical applications. Use of CMC products in such applications is understood to be fully at the risk of the
customer. In order to minimize risks associated with the customer’s applications, the customer should
provide adequate design and operating safeguards.
HsinChu Headquarter
Sales & Marketing
5F, No. 11, Park Avenue II,
Science-Based Industrial Park,
HsinChu City, Taiwan
7F-6, No.32, Sec. 1, Chenggong Rd.,
Nangang District, Taipei City 115, Taiwan
T E L : +886-3-567 9979
F A X : +886-3-567 9909
http://www.champion-micro.com
T E L : +886-2-2788 0558
F A X : +886-2-2788 2985
2008/10/23 Rev. 2.1
Champion Microelectronic Corporation
Page 18