LINER LTC3558EUD-TRPBF Linear usb battery charger with buck and buck-boost regulator Datasheet

LTC3558
Linear USB Battery Charger
with Buck and
Buck-Boost Regulators
FEATURES
DESCRIPTION
Battery Charger
n Standalone USB Charger
n Up to 950mA Charge Current Programmable via
Single Resistor
n HPWR Input Selects 20% or 100% of Programmed
Charge Current
n NTC Input for Temperature Qualified Charging
n Internal Timer Termination
n Bad Battery Detection
Switching Regulators (Buck and Buck-Boost)
n Up to 400mA Output Current per Regulator
n 2.25MHz Constant-Frequency Operation
n Power Saving Burst Mode® Operation
n Low Profile, 20-Lead, 3mm × 3mm QFN Package
The LTC®3558 is a USB battery charger with dual high efficiency switching regulators. The device is ideally suited
to power single-cell Li-Ion/Polymer based handheld applications needing multiple supply rails.
Battery charge current is programmed via the PROG pin
and the HPWR pin with capability up to 950mA of current
at the BAT pin. The CHRG pin allows battery status to be
monitored continuously during the charging process. An
internal timer controls charger termination.
The part includes monolithic synchronous buck and buckboost regulators that can provide up to 400mA of output
current each and operate at efficiencies greater than 90%
over the entire Li-Ion/Polymer battery range. The buckboost regulator can regulate its programmed output voltage
at its rated deliverable current over the entire Li-Ion range
without drop out, increasing battery runtime.
APPLICATIONS
n
n
The LTC3558 is offered in a low profile (0.75mm), thermally
enhanced, 20-lead (3mm × 3mm) QFN package.
SD/Flash-Based MP3 Players
Low Power Handheld Applications
, LT, LTC, LTM and Burst Mode are registered trademarks of Linear Technology
Corporation. All other trademarks are the property of their respective owners.
TYPICAL APPLICATION
USB Charger Plus Buck Regulator and Buck-Boost Regulator
USB (4.3V TO 5.5V)
BAT
VCC
1μF
+
PVIN1
PVIN2
1.74k
PROG
Demo Board
SINGLE
Li-lon CELL
(2.7V TO 4.2V)
10μF
1.2V AT 400mA
NTC
SW1
LTC3558
CHRG
4.7μH
FB1
SWAB2
SUSP
324k
10pF
10μF
649k
2.2μH
DIGITAL
CONTROL
HPWR
3.3V AT 400mA
SWCD2
VOUT2
MODE
EN1
324k
121k
22μF
33pF
EN2
FB2
105k
GND
EXPOSED
PAD
15k
330pF
10pF
VC2
3558 TA01
3558f
1
LTC3558
ABSOLUTE MAXIMUM RATINGS
PIN CONFIGURATION
(Note 1)
HPWR
NTC
PROG
CHRG
VCC
TOP VIEW
20 19 18 17 16
15 EN2
GND 1
14 VC2
13 FB2
BAT 2
21
MODE 3
12 SUSP
FB1 4
11 VOUT2
7
8
9 10
PVIN2
SWAB2
SWCD2
6
SW1
EN1 5
PVIN1
VCC (Transient);
t < 1ms and Duty Cycle < 1%....................... –0.3V to 7V
VCC (Static) .................................................. –0.3V to 6V
BAT, CHRG ................................................... –0.3V to 6V
PROG, SUSP .................................–0.3V to (VCC + 0.3V)
HPWR, NTC................... –0.3V to Max (VCC, BAT) + 0.3V
PROG Pin Current ...............................................1.25mA
BAT Pin Current ..........................................................1A
PVIN1, PVIN2 ..................................–0.3V to (BAT + 0.3V)
EN1, EN2, MODE, VOUT2 .............................. –0.3V to 6V
FB1, SW1 ......................... –0.3V to (PVIN1 + 0.3V) or 6V
FB2, VC2, SWAB2 ............. –0.3V to (PVIN2 + 0.3V) or 6V
SWCD2 ............................–0.3V to (VOUT2 + 0.3V) or 6V
ISW1 ...............................................................600mA DC
ISWAB2, ISWCD2, IVOUT2 ...................................750mA DC
Junction Temperature (Note 2) ............................. 125°C
Operating Temperature Range (Note 3).... –40°C to 85°C
Storage Temperature.............................. –65°C to 125°C
UD PACKAGE
20-LEAD (3mm × 3mm) PLASTIC QFN
TJMAX = 125°C, θJA = 68°C/W
EXPOSED PAD (PIN 21) IS GND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3558EUD#PBF
LTC3558EUD#TRPBF
LDCD
20-Lead (3mm × 3mm) Plastic QFN
–40°C to 85°C
Consult LTC Marketing for parts specified with wider operating temperature ranges.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
3558f
2
LTC3558
ELECTRICAL CHARACTERISTICS
The l denotes specifications that apply over the full operating temperature
range, otherwise specifications are at TA = 25°C. VCC = 5V, BAT = PVIN1 = PVIN2 = 3.6V, RPROG = 1.74k, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Battery Charger
VCC
Input Supply Voltage
IVCC
Battery Charger Quiescent Current
(Note 4)
BAT Regulated Output Voltage
VFLOAT
l
4.3
Standby Mode, Charge Terminated
Suspend Mode, VSUSP = 5V
4.179
4.165
440
84
5.5
V
285
8.5
4.200
4.200
460
92
–3.5
–2.5
–1.5
400
17
4.221
4.235
500
100
–7
–4
–3
μA
μA
V
V
mA
mA
μA
μA
μA
–50
4
–100
4.125
μA
V
IBAT
Constant-Current Mode Charge
Current
Battery Drain Current
VUVLO
Undervoltage Lockout Threshold
0°C ≤ TA ≤ 85°C
l
HPWR = 1
HPWR = 0
Standby Mode, Charger Terminated, EN1 = EN2 = 0
Shutdown, VCC < VUVLO, BAT = 4.2V, EN1 = EN2 = 0
Suspend Mode, SUSP = 5V, BAT = 4.2V, EN1 = EN2 = 0
VCC = 0V, EN1 = EN2 = 1, MODE = 1,
FB1 = FB2 = 0.85V, VOUT2 = 3.6V
BAT = 3.5V, VCC Rising
ΔVUVLO
Undervoltage Lockout Hysteresis
BAT = 3.5V
VDUVLO
BAT = 4.05V, (VCC – BAT) Falling
VPROG
Differential Undervoltage Lockout
Threshold
Differential Undervoltage Lockout
Hysteresis
PROG Pin Servo Voltage
hPROG
Ratio of IBAT to PROG Pin Current
ITRKL
Trickle Charge Current
BAT < VTRKL
36
VTRKL
Trickle Charge Threshold Voltage
BAT Rising
2.8
ΔVTRKL
Trickle Charge Hysteresis Voltage
ΔVRECHRG
Recharge Battery Threshold Voltage
Threshold Voltage Relative to VFLOAT
tRECHRG
Recharge Comparator Filter Time
BAT Falling
tTERM
Safety Timer Termination Period
BAT = VFLOAT
tBADBAT
Bad Battery Termination Time
BAT < VTRKL
hC/10
End-of-Charge Indication Current Ratio (Note 5)
tC/10
End-of-Charge Comparator Filter Time
IBAT Falling
2.2
ms
RON(CHG)
Battery Charger Power FET OnResistance (Between VCC and BAT)
Junction Temperature in Constant
Temperature Mode
IBAT = 190mA
500
mΩ
105
°C
VDIS
Cold Temperature Fault Threshold
Voltage
Hot Temperature Fault Threshold
Voltage
NTC Disable Threshold Voltage
INTC
NTC Leakage Current
Rising NTC Voltage
Hysteresis
Falling NTC Voltage
Hysteresis
Falling NTC Voltage
Hysteresis
VNTC = VCC = 5V
ICHG
ΔVDUVLO
TLIM
3.85
200
30
50
BAT = 4.05V
130
HPWR = 1
HPWR = 0
BAT < VTRKL
1.000
0.200
0.100
800
mV
70
mV
V
V
V
mA/mA
46
56
2.9
3
100
–75
–95
mV
mA
V
mV
–115
1.7
mV
ms
3.5
4
4.5
Hour
0.4
0.5
0.6
Hour
0.085
0.1
0.11
mA/mA
NTC
VCOLD
VHOT
75
33.4
l
0.7
–1
76.5
1.6
34.9
1.6
1.7
50
78
36.4
2.7
1
%VCC
%VCC
%VCC
%VCC
%VCC
mV
μA
3558f
3
LTC3558
ELECTRICAL CHARACTERISTICS
The l denotes specifications that apply over the full operating temperature
range, otherwise specifications are at TA = 25°C. VCC = 5V, BAT = PVIN1 = PVIN2 = 3.6V, RPROG = 1.74k, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Logic (HPWR, SUSP, CHRG, EN1, EN2, MODE)
VIL
Input Low Voltage
HPWR, SUSP, MODE, EN1, EN2 Pins
VIH
Input High Voltage
HPWR, SUSP, MODE, EN1, EN2 Pins
RDN
Logic Pin Pull-Down Resistance
HPWR, SUSP Pins
VCHRG
CHRG Pin Output Low Voltage
ICHRG = 5mA
ICHRG
CHRG Pin Input Current
BAT = 4.5V, VCHRG = 5V
0.4
1.2
l
1.9
V
V
4
6.3
MΩ
100
250
mV
0
1
μA
4.2
V
400
50
2
8
Buck Switching Regulator
l
PVIN1
Input Supply Voltage
IPVIN1
fOSC
Pulse Skip Input Current
Burst Mode Current
Shutdown Current
Supply Current in UVLO
PVIN1 Falling
PVIN1 Rising
Switching Frequency
ILIMSW1
Peak PMOS Current Limit
VFB1
Feedback Voltage
MODE = 0
IFB1
FB Input Current
FB1 = 0.85V
DMAX1
Maximum Duty Cycle
FB1 = 0V
RPMOS1
RDS(ON) of PMOS
ISW1 = 100mA
0.65
Ω
RNMOS1
RDS(ON) of NMOS
ISW1 = –100mA
0.75
Ω
RSW1(PD)
SW Pull-Down in Shutdown
13
kΩ
PVIN1 UVLO
FB1 = 0.85V, MODE = 0 (Note 6)
FB1 = 0.85V, MODE = 1 (Note 6)
EN1 = 0
PVIN1 = PVIN2 = 2V
2.7
1.91
220
35
0
4
2.45
2.55
2.25
2.70
2.59
μA
μA
μA
μA
V
V
MHz
550
800
1050
mA
780
800
820
mV
50
nA
l
l
l
MODE = 0
l
2.30
–50
l
100
%
Buck-Boost Switching Regulator
l
PVIN2
Input Supply Voltage
IPVIN2
VOUT2(LOW)
PWM Input Current
Burst Mode Input Current
Shutdown Current
Supply Current in UVLO
PVIN2 Falling
PVIN2 Rising
Minimum Regulated Buck-Boost VOUT
VOUT2(HIGH)
Maximum Regulated Buck-Boost VOUT
ILIMF2
Forward Current Limit (Switch A)
MODE = 0
IPEAK2(BURST)
Forward Current Limit (Switch A)
ILIMR2
Reverse Current Limit (Switch D)
IZERO2(BURST) Reverse Current Limit (Switch D)
PVIN2 UVLO
2.7
MODE = 0, IOUT = 0A, FB2 = 0.85V (Note 6)
MODE = 1, IOUT = 0A, FB2 = 0.85V (Note 6)
EN2 = 0, IOUT = 0A
PVIN1 = PVIN2 = 2V
l
l
2.30
V
400
30
1
8
μA
μA
μA
μA
V
V
V
2.70
2.75
5.45
5.60
l
580
700
820
mA
MODE = 1
l
180
250
320
mA
MODE = 0
l
325
450
575
mA
MODE = 1
l
–35
0
35
mA
2.7V < PVIN2 < 4.2V
2.75V < VOUT2 < 5.5V
VFB2
Maximum Deliverable Output Current
in Burst Mode Operation
Feedback Servo Voltage
IFB2
FB2 Input Current
FB2 = 0.85V
–50
fOSC
Switching Frequency
MODE = 0
1.91
IMAX2(BURST)
220
20
0
4
2.45
2.55
2.65
4.2
V
50
l
780
mA
800
2.25
820
mV
50
nA
2.59
MHz
3558f
4
LTC3558
ELECTRICAL CHARACTERISTICS
The l denotes specifications that apply over the full operating temperature
range, otherwise specifications are at TA = 25°C. VCC = 5V, BAT = PVIN1 = PVIN2 = 3.6V, RPROG = 1.74k, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
RDSP(ON)
PMOS RDS(ON)
VOUT = 3.6V
RDSN(ON)
NMOS RDS(ON)
ILEAK(P)
PMOS Switch Leakage
Switches A, D
–1
1
μA
ILEAK(N)
NMOS Switch Leakage
Switches B, C
–1
1
μA
DCBUCK(MAX)
Maximum Buck Duty Cycle
MODE = 0
DCBOOST(MAX) Maximum Boost Duty Cycle
MODE = 0
MIN
l
TYP
MAX
UNITS
0.6
Ω
0.6
Ω
100
%
75
%
tSS2
Soft-Start Time
0.5
ms
ROUT(PD)
VOUT Pull-Down in Shutdown
10
kΩ
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formula:
TJ = TA + (PD • θJA)
Note 3: The LTC3558E is guaranteed to meet specifications from 0°C to
85°C. Specifications over the –40°C to 85°C operating temperature range
are assured by design, characterization and correlation with statistical
process controls.
Note 4: VCC supply current does not include current through the PROG pin
or any current delivered to the BAT pin. Total input current is equal to this
specification plus 1.00125 • IBAT where IBAT is the charge current.
Note 5: IC/10 is expressed as a fraction of measured full charge current
with indicated PROG resistor.
Note 6: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency.
3558f
5
LTC3558
TYPICAL PERFORMANCE CHARACTERISTICS
Suspend State Supply and BAT
Currents vs Temperature
4.24
4.205
VCC = 5V
4.200
4.23
IVCC
8
Battery Regulation (Float) Voltage
vs Battery Charge Current,
Constant-Voltage Charging
Battery Regulation (Float)
Voltage vs Temperature
10
9
TA = 25°C, unless otherwise noted.
4.195
4.22
4.190
5
4
VFLOAT (V)
6
VCC = 5V
BAT = 4.2V
SUSP = 5V
EN1 = EN2 = 0V
4.185
4.21
VBAT (V)
CURRENT (μA)
7
4.180
4.20
4.175
4.19
4.170
4.18
4.165
3
2
IBAT
1
4.17
0
–55
4.16
–55 –35
–35
25
5
–15
45
TEMPERATURE (°C)
65
85
4.155
45
25
5
TEMPERATURE (°C)
65
–15
3558 G01
4.150
500
VCC = 5V
495 HPWR
= 5V
490 RPROG = 1.74k
485 EN1 = EN2 = 0V
Battery Charge Current vs Ambient
Temperature in Thermal Regulation
500
HPWR = 5V
VCC = 5V
RPROG = 1.74k
450
450
480
350
475
300
300
460
250
200
450
100
445
50
0
4.3 4.4 4.5 4.6 4.7 4.8 4.9 5.0 5.1 5.2 5.3 5.4 5.5
VCC (V)
HPWR = 0V
50
2
2.5
3.5
3
VBAT (V)
4
3.0
1.2
1.0
0.8
3.7
3.6
65
85
3558 G07
VPROG (V)
3.8
25
5
45
–15
TEMPERATURE (°C)
VCC = 5V
HPWR = 5V
RPROG = 1.74k
EN1 = EN2 = 0V
EN1 = EN2 = 0V
2.0
IBAT (μA)
VCC (V)
PROG Voltage
vs Battery Charge Current
BAT = 4.2V
FALLING
5 25 45 65 85 105 125
TEMPERATURE (°C)
3558 G06
2.5
RISING
4.0
3.5
–55 –35
0
–55 –35 –15
4.5
Battery Drain Current in Undervoltage
Lockout vs Temperature
BAT = 3.5V
3.9
VCC = 5V
HPWR = 5V
RPROG = 1.74k
EN1 = EN2 = 0
3558 G05
Battery Charger Undervoltage
Lockout Threshold vs Temperature
4.1
200
100
3558 G04
4.2
250
150
150
455
440
IBAT (mA)
400
350
IBAT (mA)
400
465
0 100 200 300 400 500 600 700 800 900 1000
IBAT (mA)
3558 G03
Battery Charge Current
vs Battery Voltage
500
IBAT (mA)
85
3558 G02
Battery Charge Current
vs Supply Voltage
470
VCC = 5V
HPWR = 5V
RPROG = 845Ω
EN1 = EN2 = 0V
4.160
BAT = 3.6V
1.5
0.6
1.0
0.4
0.5
0.2
0
–55 –35
0
25
5
45
–15
TEMPERATURE (°C)
65
85
3558 G08
0
50 100 150 200 250 300 350 400 450 500
IBAT (mA)
3558 G09
3558f
6
LTC3558
TYPICAL PERFORMANCE CHARACTERISTICS
Recharge Threshold
vs Temperature
115
111
Battery Charger FET
On-Resistance vs Temperature
SUSP/HPWR Pin Rising
Thresholds vs Temperature
1.2
700
VCC = 5V
VCC = 4V
IBAT = 200mA
EN1 = EN2 = 0V
650
107
1.0
95
91
THRESHOLD (V)
99
VCC = 5V
1.1
600
103
RDS(ON) (mΩ)
VRECHARGE (mV)
TA = 25°C, unless otherwise noted.
550
500
450
0.9
0.8
0.7
87
83
79
75
–55
–35
25
5
–15
45
TEMPERATURE (°C)
65
400
0.6
350
0.5
300
–55
85
–35
–15
5
25
45
65
PERCENT ERROR (%)
1.5
50
ICHRG (mA)
VOLTAGE (mV)
VCC = 5V
BAT = 3.8V
60
100
Timer Accuracy vs Supply Voltage
2.0
70
VCC = 5V
ICHRG = 5mA
60
40
30
40
20
20
10
0
–55 –35
25
5
45
–15
TEMPERATURE (°C)
65
1
2
4
3
CHRG (V)
3
2
1
0
–1
65
85
3558 G16
4.5
4.7
4.9
VCC (V)
5.1
5.3
5.5
Buck and Buck-Boost Regulator
Switching Frequency vs Temperature
VCC = 5V
RPROG = 0.845k
HPWR = 5V
2.325
FREQUENCY (MHz)
BAT (V)
4
CHRG (V)
PERCENT ERROR (%)
5
4.3
3558 G15
2.425
1000
800
600
400
200
0
5.0
4.5
4.0
3.5
3.0
5.0
4.0
3.0
2.0
1.0
0
IBAT (mA)
6
–15
5
25
45
TEMPERATURE (°C)
6
5
Complete Charge Cycle
2400mAh Battery
VCC = 5V
–35
0
3558 G14
Timer Accuracy vs Temperature
–2
–55
0.5
–1.0
0
3558 G13
7
1.0
–0.5
0
85
85
3558 G12
CHRG Pin I-V Curve
80
65
3558 G11
CHRG Pin Output Low Voltage
vs Temperature
120
45
25
5
TEMPERATURE (°C)
–15
TEMPERATURE (°C)
3558 G10
140
0.4
–55 –35
85
VCC = 0V, MODE = 0
BAT = PVIN1 = PVIN2
BAT = 4.2V
2.225
BAT = 2.7V
BAT = 3.6V
2.125
2.025
1.925
1.825
0
1
2
4
3
TIME (HOUR)
5
6
3558 G17
1.725
–55 –35 –15
5 25 45 65 85 105 125
TEMPERATURE (°C)
3558 G18
3558f
7
LTC3558
TYPICAL PERFORMANCE CHARACTERISTICS
Buck and Buck-Boost Regulator
Undervoltage Thresholds
vs Temperature
2.700
Buck and Buck-Boost Regulator
Enable Thresholds
vs Temperature
1200
BAT = PVIN1 = PVIN2
50
BAT = PVIN1 = PVIN2 = 3.6V
1100
2.650
2.600
RISING
900
2.550
2.500
FALLING
800
2.450
RISING
700
2.400
FALLING
600
2.350
2.250
–55 –35 –15
400
–55 –35 –15
5 25 45 65 85 105 125
TEMPERATURE (°C)
PVIN1 = 4.2V
35
PVIN1 = 2.7V
30
20
–55 –35 –15
5 25 45 65 85 105 125
TEMPERATURE (°C)
5 25 45 65 85 105 125
TEMPERATURE (°C)
3558 G20
3558 G19
Buck Regulator Input Current vs
Temperature, Pulse Skip Mode
400
40
25
500
2.300
FB1 = 0.85V
45
1000
VEN (V)
INPUT VOLTAGE (V)
Buck Regulator Input Current vs
Temperature, Burst Mode Operation
INPUT CURRENT (μA)
2.750
TA = 25°C, unless otherwise noted.
3558 G21
Buck Regulator PMOS RDS(0N)
vs Temperature
Buck Regulator NMOS RDS(0N)
vs Temperature
1300
1300
1200
1200
1100
1100
1000
1000
FB1 = 0.85V
PVIN1 = 4.2V
250
PVIN1 = 2.7V
200
150
100
–55 –35 –15
900
PVIN1 = 2.7V
800
PVIN1 = 4.2V
700
600
500
Buck Regulator Efficiency vs ILOAD
1.23
70
1.22
60
1.21
1.230
Burst Mode
OPERATION
1.20
1.220
PULSE SKIP
MODE
1.19
1.210
1.200
1.190
30
1.18
1.180
20
1.17
1.170
1.16
1.160
VOUT = 1.2V
PVIN1 = 2.7V
PVIN1 = 4.2V
0
0.1
1
10
ILOAD (mA)
100
1000
1.15
1
10
100
1000
ILOAD (mA)
3558 G25
ILOAD = 200mA
1.240
VOUT (V)
80
VOUT (V)
EFFICIENCY (%)
Buck Regulator Line Regulation
1.250
PVIN1 = 3.6V
1.24 VOUT = 1.2V
40
5 25 45 65 85 105 125
TEMPERATURE (°C)
3558 G24
Buck Regulator Load Regulation
Burst Mode
OPERATION
10
8
400
–55 –35 –15
5 25 45 65 85 105 125
TEMPERATURE (°C)
1.25
PULSE SKIP
MODE
PVIN1 = 4.2V
700
3558 G23
100
50
800
500
3558 G22
90
PVIN1 = 2.7V
900
600
400
–55 –35 –15
5 25 45 65 85 105 125
TEMPERATURE (°C)
RDS(ON) (mΩ)
300
RDS(ON) (mΩ)
INPUT CURRENT (μA)
350
3558 G26
1.150
2.700
3.000
3.600
3.300
PVIN1 (V)
3.900
4.200
3558 G27
3558f
LTC3558
TYPICAL PERFORMANCE CHARACTERISTICS
Buck Regulator
Pulse Skip Mode Operation
Buck Regulator Start-Up Transient
VOUT
500mV/DIV
INDUCTOR
CURRENT
IL = 200mA/
DIV
EN
2V/DIV
Buck Regulator
Burst Mode Operation
VOUT
20mV/
DIV (AC)
VOUT
20mV/
DIV (AC)
SW
2V/DIV
SW
2V/DIV
INDUCTOR
CURRENT
IL = 50mA/
DIV
INDUCTOR
CURRENT
IL = 60mA/
DIV
3558 G28
PVIN1 = 3.8V
LOAD = 10mA
Buck Regulator Transient
Response, Pulse Skip Mode
200ns/DIV
3558 G29
Buck Regulator Transient
Response, Burst Mode Operation
INDUCTOR
CURRENT
IL = 200mA/
DIV
INDUCTOR
CURRENT
IL = 200mA/
DIV
VOUT
50mV/
DIV (AC)
VOUT
50mV/
DIV (AC)
LOAD STEP
5mA TO
290mA
50μs/DIV
3558 G31
2μs/DIV
3558 G30
Buck-Boost Regulator Input
Current vs Temperature
30
Burst Mode OPERATION
FB2 = 0.85V
25
LOAD STEP
5mA TO
290mA
PVIN1 = 3.8V
PVIN1 = 3.8V
LOAD = 60mA
INPUT CURRENT (μA)
PVIN1 = 3.8V
50μs/DIV
PULSE SKIP MODE
LOAD = 6Ω
TA = 25°C, unless otherwise noted.
PVIN1 = 3.8V
50μs/DIV
PVIN2 = 4.2V
20
PVIN2 = 2.7V
15
10
3558 G32
5
–55 –35 –15
5 25 45 65 85 105 125
TEMPERATURE (°C)
3558 G33
Buck-Boost Regulator Input
Current vs Temperature
500
1100
PVIN2 = 4.2V
250
PVIN2 = 2.7V
600
1000
PVIN2 = 2.7V
900
RDS(ON) (mΩ)
350
550
500
450
400
PVIN2 = 4.2V
3558 G34
600
250
300
200
–55 –35 –15
200
–55 –35 –15
5 25 45 65 85 105 125
TEMPERATURE (°C)
3558 G35
PVIN2 = 2.7V
700
400
300
5 25 45 65 85 105 125
TEMPERATURE (°C)
800
500
350
150
100
–55 –35 –15
1200
750
650
400
200
800
700
RDS(ON) (mΩ)
INPUT CURRENT (μA)
PWM MODE
450 FB2 = 0.85V
300
Buck-Boost Regulator NMOS
RDS(ON) vs Temperature
Buck-Boost Regulator PMOS
RDS(ON) vs Temperature
PVIN2 = 4.2V
5 25 45 65 85 105 125
TEMPERATURE (°C)
3558 G36
3558f
9
LTC3558
TYPICAL PERFORMANCE CHARACTERISTICS
Buck-Boost Efficiency
vs Load Current
Buck-Boost Regulator
Efficiency vs Input Voltage
100
VOUT = 3.3V
100
ILOAD = 10mA
90
80
85
70
80
ILOAD = 100mA
75
ILOAD = 400mA
70
VOUT = 3.3V
90
ILOAD = 1mA
EFFICIENCY (%)
EFFICIENCY (%)
95
65
3.6V
4.2V
50
40
2.7V
20
Burst Mode
OPERATION
PWM MODE
55
50
2.700
3.000
3.600
3.300
PVIN2 (V)
3.900
PVIN2, Burst Mode
OPERATION
PVIN2, PWM MODE
3.6V
4.2V
10
0
0.10
4.200
2.7V
60
30
60
TA = 25°C, unless otherwise noted.
1
10
ILOAD (mA)
100
3558 G38
3558 G37
Buck-Boost Regulator
Load Regulation
3.36
3.35
Buck-Boost Regulator
Line Regulation
3.36
PVIN2 = 3.6V
3.35
3.34
3.34
3.33
3.33
3.32
3.32
Burst Mode OPERATION
3.31
PWM MODE
3.30
3.29
VOUT (V)
VOUT (V)
1000
3.31
3.30
3.29
3.28
3.28
3.27
3.27
3.26
3.26
3.25
3.25
3.24
0.10
1
10
ILOAD (mA)
100
1000
PWM MODE
ILOAD = 100mA
Burst Mode
OPERATION
ILOAD = 10mA
3.24
2.700
3.000
3.600
3.300
PVIN2 (V)
4.200
3.900
3558 G40
3558 G39
Buck-Boost Regulator Start-Up
Transient, Burst Mode Operation
Buck-Boost Regulator Start-Up
Transient, PWM Mode
PVIN2 = 3.6V
RLOAD = 16Ω
PVIN2 = 3.6V
RLOAD = 332Ω
VOUT
1V/DIV
VOUT
1V/DIV
INDUCTOR
CURRENT
IL = 200mA/DIV
INDUCTOR
CURRENT
IL = 200mA/DIV
EN2
1V/DIV
EN2
1V/DIV
100μs/DIV
3558 G41
100μs/DIV
3558 G42
3558f
10
LTC3558
PIN FUNCTIONS
GND (Pin 1): Ground. Connect to Exposed Pad (Pin 21).
BAT (Pin 2): Charge Current Output. Provides charge current to the battery and regulates final float voltage to 4.2V.
VC2 (Pin 14): Output of the Error Amplifier and Voltage
Compensation Node for the Buck-Boost Regulator. External Type I or Type III compensation (to FB2) connects
to this pin.
MODE (Pin 3): MODE Pin for Switching Regulators. When
held high, both regulators operate in Burst Mode Operation. When held low, the buck regulator operates in pulse
skip mode and the buck-boost regulator operates in PWM
mode. This pin is a high impedance input; do not float.
EN2 (Pin 15): Enable Input Pin for the Buck-Boost Regulator. This pin is a high impedance input; do not float.
Active high.
FB1 (Pin 4): Buck Regulator Feedback Voltage Pin. Receives feedback by a resistor divider connected across
the output.
EN1 (Pin 5): Enable Input Pin for the Buck Regulator. This
pin is a high impedance input; do not float. Active high.
SW1 (Pin 6): Buck Regulator Switching Node. External
inductor connects to this node.
PVIN1 (Pin 7): Input Supply Pin for Buck Regulator. Connect to BAT and PVIN2. A single 10μF input decoupling
capacitor to GND is required.
PVIN2 (Pin 8): Input Supply Pin for Buck-Boost Regulator.
Connect to BAT and PVIN1. A single 10μF input decoupling
capacitor to GND is required.
SWAB2 (Pin 9): Switch Node for Buck-Boost Regulator
Connected to the Internal Power Switches A and B. External
inductor connects between this node and SWCD2.
SWCD2 (Pin 10): Switch Node for Buck-Boost Regulator
Connected to the Internal Power Switches C and D. External
inductor connects between this node and SWAB2.
VOUT2 (Pin 11): Regulated Output Voltage for Buck-Boost
Regulator.
SUSP (Pin 12): Suspend Battery Charging Operation. A
voltage greater than 1.2V on this pin puts the battery charger in suspend mode, disables the charger and resets the
termination timer. A weak pull-down current is internally
applied to this pin to ensure it is low at power-up when
the input is not being driven externally.
FB2 (Pin 13): Buck-Boost Regulator Feedback Voltage
Pin. Receives feedback by a resistor divider connected
across the output.
HPWR (Pin 16): High Current Battery Charging Enabled.
A voltage greater than 1.2V at this pin programs the
BAT pin current at 100% of the maximum programmed
charge current. A voltage less than 0.4V sets the BAT pin
current to 20% of the maximum programmed charge
current. When used with a 1.74k PROG resistor, this pin
can toggle between low power and high power modes per
USB specification. A weak pull-down current is internally
applied to this pin to ensure it is low at power-up when
the input is not being driven externally.
NTC (Pin 17): Input to the NTC Thermistor Monitoring
Circuit. The NTC pin connects to a negative temperature
coefficient thermistor which is typically co-packaged with
the battery pack to determine if the battery is too hot or too
cold to charge. If the battery temperature is out of range,
charging is paused until the battery temperature re-enters
the valid range. A low drift bias resistor is required from
VCC to NTC and a thermistor is required from NTC to
ground. To disable the NTC function, the NTC pin should
be tied to ground.
PROG (Pin 18): Charge Current Program and Charge
Current Monitor Pin. Charge current is programmed by
connecting a resistor from PROG to ground. When charging in constant-current mode, the PROG pin servos to 1V
if the HPWR pin is pulled high, or 200mV if the HPWR pin
is pulled low. The voltage on this pin always represents
the BAT pin current through the following formula:
IBAT =
PROG • 800
RPROG
CHRG (Pin 19): Open-Drain Charge Status Output. The
CHRG pin indicates the status of the battery charger. Four
possible states are represented by CHRG charging, not
charging (i.e., the charge current is less than one-tenth
3558f
11
LTC3558
PIN FUNCTIONS
of the full-scale charge current), unresponsive battery
(i.e., the battery voltage remains below 2.9V after one half
hour of charging) and battery temperature out of range.
CHRG requires a pull-up resistor and/or LED to provide
indication.
VCC (Pin 20): Battery Charger Input. A 1μF decoupling
capacitor to GND is recommended.
Exposed Pad (Pin 21): Ground. The Exposed Pad must
be soldered to PCB ground to provide electrical contact
and rated thermal performance.
BLOCK DIAGRAM
20
VCC
BAT
BODY
MAXER
VCC
1x
800x
BAT
–
16
12
+
19
CHRG
HPWR
TA
CA
LOGIC
SUSP
TDIE
PROG
NTCA
17
5
15
4
18
NTC
NTC REF
3
2
PVIN1
MODE
BATTERY CHARGER
PVIN2
PVIN1
7
EN1
UNDERVOLTAGE
LOCKOUT
EN2
EN
–
FB1
OT
DIE
TEMPERATURE
0.8V
TDIE
+
MODE
CLK
MP
CONTROL
LOGIC
Gm
SW1
6
MN
BUCK REGULATOR
BANDGAP
OSCILLATOR
2.25MHz
13
PVIN2
VREF
= 0.8V
CLK
VOUT2
11
BUCK-BOOST REGULATOR
EN
MODE
CLK
–
FB2
0.8V
14
8
ERROR
+ AMP
VC2
A D
SWAB2
CONTROL
LOGIC
SWCD2
9
10
B C
VC2
GND
EXPOSED PAD
1
21
3558 BD
3558f
12
LTC3558
OPERATION
The LTC3558 is a linear battery charger with a monolithic
synchronous buck regulator and a monolithic synchronous buck-boost regulator. The buck regulator is internally compensated and needs no external compensation
components.
The battery charger employs a constant-current, constantvoltage charging algorithm and is capable of charging a
single Li-Ion battery at charging currents up to 950mA. The
user can program the maximum charging current available
at the BAT pin via a single PROG resistor. The actual BAT
pin current is set by the status of the HPWR pin.
For proper operation, the BAT, PVIN1 and PVIN2 pins
must be tied together, as shown in Figure 1. Current being delivered at the BAT pin is 500mA. Both
switching regulators are enabled. The sum of the
average input currents drawn by both switching regulators
is 200mA. This makes the effective battery charging current only 300mA. If the HPWR pin were tied LO, the BAT
pin current would be 100mA. With the switching regulator
conditions unchanged, this would cause the battery to
discharge at 100mA.
500mA
USB (5V)
300mA
BAT
VCC
+
200mA
PVIN1
PROG
SUSP
HIGH
HIGH
HIGH
LOW
+
PVIN2
LTC3558
RPROG
10μF
SWAB2
2.2μH
HPWR
EN1
EN2
MODE
SINGLE Li-lon
CELL 3.6V
SWCD2
VOUT2
SW1
VOUT1
3558 F01
Figure 1. For Proper Operation, the BAT, PVIN1 and PVIN2 Pins Must Be Tied Together
APPLICATIONS INFORMATION
Battery Charger Introduction
Input Current vs Charge Current
The LTC3558 has a linear battery charger designed to
charge single-cell lithium-ion batteries. The charger uses
a constant-current/constant-voltage charge algorithm
with a charge current programmable up to 950mA. Additional features include automatic recharge, an internal
termination timer, low-battery trickle charge conditioning,
bad-battery detection, and a thermistor sensor input for
out of temperature charge pausing.
The battery charger regulates the total current delivered
to the BAT pin; this is the charge current. To calculate the
total input current (i.e., the total current drawn from the
VCC pin), it is necessary to sum the battery charge current,
charger quiescent current and PROG pin current.
Furthermore, the battery charger is capable of operating
from a USB power source. In this application, charge
current can be programmed to a maximum of 100mA or
500mA per USB power specifications.
Undervoltage Lockout (UVLO)
The undervoltage lockout circuit monitors the input voltage (VCC) and disables the battery charger until VCC rises
above VUVLO (typically 4V). 200mV of hysteresis prevents
oscillations around the trip point. In addition, a differential
undervoltage lockout circuit disables the battery charger
3558f
13
LTC3558
APPLICATIONS INFORMATION
when VCC falls to within VDUVLO (typically 50mV) of the
BAT voltage.
Suspend Mode
The battery charger can also be disabled by pulling the
SUSP pin above 1.2V. In suspend mode, the battery
drain current is reduced to 1.5μA and the input current is
reduced to 8.5μA.
Charge Cycle Overview
When a battery charge cycle begins, the battery charger
first determines if the battery is deeply discharged. If the
battery voltage is below VTRKL, typically 2.9V, an automatic
trickle charge feature sets the battery charge current to
10% of the full-scale value.
Once the battery voltage is above 2.9V, the battery charger
begins charging in constant-current mode. When the
battery voltage approaches the 4.2V required to maintain
a full charge, otherwise known as the float voltage, the
charge current begins to decrease as the battery charger
switches into constant-voltage mode.
Trickle Charge and Defective Battery Detection
Any time the battery voltage is below VTRKL, the charger
goes into trickle charge mode and reduces the charge
current to 10% of the full-scale current. If the battery
voltage remains below VTRKL for more than 1/2 hour, the
charger latches the bad-battery state, automatically terminates, and indicates via the CHRG pin that the battery was
unresponsive. If for any reason the battery voltage rises
above VTRKL, the charger will resume charging. Since the
charger has latched the bad-battery state, if the battery
voltage then falls below VTRKL again but without rising past
VRECHRG first, the charger will immediately assume that
the battery is defective. To reset the charger (i.e., when
the dead battery is replaced with a new battery), simply
remove the input voltage and reapply it or put the part in
and out of suspend mode.
Charge Termination
The battery charger has a built-in safety timer that sets
the total charge time for 4 hours. Once the battery voltage
rises above VRECHRG (typically 4.105V) and the charger
enters constant-voltage mode, the 4-hour timer is started.
After the safety timer expires, charging of the battery will
discontinue and no more current will be delivered.
Automatic Recharge
After the battery charger terminates, it will remain off,
drawing only microamperes of current from the battery.
If the portable product remains in this state long enough,
the battery will eventually self discharge. To ensure that the
battery is always topped off, a charge cycle will automatically begin when the battery voltage falls below VRECHRG
(typically 4.105V). In the event that the safety timer is
running when the battery voltage falls below VRECHRG, it
will reset back to zero. To prevent brief excursions below
VRECHRG from resetting the safety timer, the battery voltage
must be below VRECHRG for more than 1.7ms. The charge
cycle and safety timer will also restart if the VCC UVLO or
DUVLO cycles low and then high (e.g., VCC is removed
and then replaced) or the charger enters and then exits
suspend mode.
Programming Charge Current
The PROG pin serves both as a charge current program
pin, and as a charge current monitor pin. By design, the
PROG pin current is 1/800th of the battery charge current.
Therefore, connecting a resistor from PROG to ground
programs the charge current while measuring the PROG pin
voltage allows the user to calculate the charge current.
Full-scale charge current is defined as 100% of the constant-current mode charge current programmed by the
PROG resistor. In constant-current mode, the PROG pin
servos to 1V if HPWR is high, which corresponds to charging at the full-scale charge current, or 200mV if HPWR
is low, which corresponds to charging at 20% of the fullscale charge current. Thus, the full-scale charge current
and desired program resistor for a given full-scale charge
current are calculated using the following equations:
ICHG =
800 V
RPROG
RPROG =
800 V
ICHG
3558f
14
LTC3558
APPLICATIONS INFORMATION
In any mode, the actual battery current can be determined
by monitoring the PROG pin voltage and using the following equation:
IBAT =
PROG
• 800
RPROG
Thermal Regulation
To prevent thermal damage to the IC or surrounding
components, an internal thermal feedback loop will automatically decrease the programmed charge current if the
die temperature rises to approximately 115°C. Thermal
regulation protects the battery charger from excessive
temperature due to high power operation or high ambient
thermal conditions and allows the user to push the limits
of the power handling capability with a given circuit board
design without risk of damaging the LTC3558 or external
components. The benefit of the LTC3558 battery charger
thermal regulation loop is that charge current can be set
according to actual conditions rather than worst-case
conditions with the assurance that the battery charger
will automatically reduce the current in worst-case conditions.
Charge Status Indication
The CHRG pin indicates the status of the battery charger.
Four possible states are represented by CHRG charging,
not charging, unresponsive battery and battery temperature
out of range.
The signal at the CHRG pin can be easily recognized as one
of the above four states by either a human or a microprocessor. The CHRG pin, which is an open-drain output, can
drive an indicator LED through a current limiting resistor
for human interfacing, or simply a pull-up resistor for
microprocessor interfacing.
To make the CHRG pin easily recognized by both humans
and microprocessors, the pin is either a low for charging,
a high for not charging, or it is switched at high frequency
(35kHz) to indicate the two possible faults: unresponsive
battery and battery temperature out of range.
When charging begins, CHRG is pulled low and remains
low for the duration of a normal charge cycle. When the
charge current has dropped to below 10% of the full-scale
current, the CHRG pin is released (high impedance). If
a fault occurs after the CHRG pin is released, the pin remains high impedance. However, if a fault occurs before
the CHRG pin is released, the pin is switched at 35kHz.
While switching, its duty cycle is modulated between a high
and low value at a very low frequency. The low and high
duty cycles are disparate enough to make an LED appear
to be on or off thus giving the appearance of “blinking”.
Each of the two faults has its own unique “blink” rate for
human recognition as well as two unique duty cycles for
microprocessor recognition.
Table 1 illustrates the four possible states of the CHRG
pin when the battery charger is active.
Table 1. CHRG Output Pin
STATUS
FREQUENCY
MODULATION
(BLINK)
FREQUENCY
Charging
0Hz
0 Hz (Lo-Z)
100%
IBAT < C/10
0Hz
0 Hz (Hi-Z)
0%
NTC Fault
35kHz
1.5Hz at 50%
6.25%, 93.75%
Bad Battery
35kHz
6.1Hz at 50%
12.5%, 87.5%
DUTY CYCLE
An NTC fault is represented by a 35kHz pulse train whose
duty cycle alternates between 6.25% and 93.75% at a
1.5Hz rate. A human will easily recognize the 1.5Hz rate as
a “slow” blinking which indicates the out of range battery
temperature while a microprocessor will be able to decode
either the 6.25% or 93.75% duty cycles as an NTC fault.
If a battery is found to be unresponsive to charging (i.e.,
its voltage remains below VTRKL for over 1/2 hour), the
CHRG pin gives the battery fault indication. For this fault,
a human would easily recognize the frantic 6.1Hz “fast”
blinking of the LED while a microprocessor would be able
to decode either the 12.5% or 87.5% duty cycles as a bad
battery fault.
Although very improbable, it is possible that a duty cycle
reading could be taken at the bright-dim transition (low
duty cycle to high duty cycle). When this happens the
duty cycle reading will be precisely 50%. If the duty cycle
reading is 50%, system software should disqualify it and
take a new duty cycle reading.
3558f
15
LTC3558
APPLICATIONS INFORMATION
NTC Thermistor
The battery temperature is measured by placing a negative temperature coefficient (NTC) thermistor close to the
battery pack. The NTC circuitry is shown in Figure 3.
To use this feature, connect the NTC thermistor, RNTC,
between the NTC pin and ground, and a bias resistor, RNOM,
from VCC to NTC. RNOM should be a 1% resistor with a
value equal to the value of the chosen NTC thermistor at
25°C (R25). A 100k thermistor is recommended since
thermistor current is not measured by the battery charger
and its current will have to be considered for compliance
with USB specifications.
The battery charger will pause charging when the resistance of the NTC thermistor drops to 0.54 times the
value of R25 or approximately 54k (for a Vishay “Curve
1” thermistor, this corresponds to approximately 40°C). If
the battery charger is in constant-voltage mode, the safety
timer will pause until the thermistor indicates a return to
a valid temperature.
As the temperature drops, the resistance of the NTC
thermistor rises. The battery charger is also designed
to pause charging when the value of the NTC thermistor
increases to 3.25 times the value of R25. For a Vishay
“Curve 1” thermistor, this resistance, 325k, corresponds
to approximately 0°C. The hot and cold comparators each
have approximately 3°C of hysteresis to prevent oscillation
about the trip point. Grounding the NTC pin disables all
NTC functionality.
DUVLO, UVLO AND SUSPEND
DISABLE MODE
NO
POWER
ON
IF SUSP < 0.4V AND
VCC > 4V AND
VCC > BAT + 130mV?
CHRG HIGH IMPEDANCE
YES
FAULT
NTC FAULT
STANDBY MODE
BATTERY CHARGING SUSPENDED
CHRG PULSES
NO CHARGE CURRENT
CHRG HIGH IMPEDANCE
NO FAULT
BAT b 2.9V
TRICKLE CHARGE MODE
1/10 FULL CHARGE CURRENT
CHRG STRONG PULL-DOWN
30 MINUTE TIMER BEGINS
2.9V < BAT < 4.105V
BAT > 2.9V
CONSTANT CURRENT MODE
FULL CHARGE CURRENT
CHRG STRONG PULL-DOWN
4-HOUR
TIMEOUT
30 MINUTE
TIMEOUT
DEFECTIVE BATTERY
NO CHARGE CURRENT
CHRG PULSES
CONSTANT VOLTAGE MODE
4-HOUR TERMINATION TIMER
BEGINS
BAT DROPS BELOW 4.105V
4-HOUR TERMINATION TIMER RESETS
3558 F02
Figure 2. State Diagram of Battery Charger Operation
3558f
16
LTC3558
APPLICATIONS INFORMATION
Alternate NTC Thermistors and Biasing
The battery charger provides temperature qualified
charging if a grounded thermistor and a bias resistor are
connected to the NTC pin. By using a bias resistor whose
value is equal to the room temperature resistance of the
thermistor (R25) the upper and lower temperatures are
pre-programmed to approximately 40°C and 0°C, respectively (assuming a Vishay “Curve 1” thermistor).
The upper and lower temperature thresholds can be adjusted by either a modification of the bias resistor value
or by adding a second adjustment resistor to the circuit.
If only the bias resistor is adjusted, then either the upper
or the lower threshold can be modified but not both. The
other trip point will be determined by the characteristics
of the thermistor. Using the bias resistor in addition to an
adjustment resistor, both the upper and the lower temperature trip points can be independently programmed with
the constraint that the difference between the upper and
lower temperature thresholds cannot decrease. Examples
of each technique are given below.
NTC thermistors have temperature characteristics which
are indicated on resistance-temperature conversion tables.
The Vishay-Dale thermistor NTHS0603N011-N1003F, used
in the following examples, has a nominal value of 100k
and follows the Vishay “Curve 1” resistance-temperature
characteristic.
In the explanation below, the following notation is used.
R25 = Value of the thermistor at 25°C
RNTC|COLD = Value of thermistor at the cold trip point
RNTC|HOT = Value of the thermistor at the hot trip point
rCOLD = Ratio of RNTC|COLD to R25
rHOT = Ratio of RNTC|HOT to R25
RNOM = Primary thermistor bias resistor (see Figure 3)
R1 = Optional temperature range adjustment resistor (see
Figure 4)
The trip points for the battery charger’s temperature qualification are internally programmed at 0.349 • VCC for the
hot threshold and 0.765 • VCC for the cold threshold.
Therefore, the hot trip point is set when:
RNTCHOT
|
RNOM + RNTCHOT
|
• VCC = 0.349 • VCC
and the cold trip point is set when:
RNTC|COLD
RNOM + RNTC|COLD
• VCC = 0.765 • VCC
Solving these equations for RNTC|COLD and RNTC|HOT
results in the following:
RNTC|HOT = 0.536 • RNOM
and
RNTC|COLD = 3.25 • RNOM
By setting RNOM equal to R25, the above equations result
in rHOT = 0.536 and rCOLD = 3.25. Referencing these ratios
to the Vishay Resistance-Temperature Curve 1 chart gives
a hot trip point of about 40°C and a cold trip point of about
0°C. The difference between the hot and cold trip points
is approximately 40°C.
By using a bias resistor, RNOM, different in value from
R25, the hot and cold trip points can be moved in either
direction. The temperature span will change somewhat due
to the nonlinear behavior of the thermistor. The following
equations can be used to easily calculate a new value for
the bias resistor:
r
RNOM = HOT • R25
0.536
RNOM =
rCOLD
• R25
3.25
where rHOT and rCOLD are the resistance ratios at the desired hot and cold trip points. Note that these equations
are linked. Therefore, only one of the two trip points can
be chosen, the other is determined by the default ratios
designed in the IC. Consider an example where a 60°C
hot trip point is desired.
From the Vishay Curve 1 R-T characteristics, rHOT is 0.2488
at 60°C. Using the above equation, RNOM should be set
3558f
17
LTC3558
APPLICATIONS INFORMATION
to 46.4k. With this value of RNOM, the cold trip point is
about 16°C. Notice that the span is now 44°C rather than
the previous 40°C.
The upper and lower temperature trip points can be independently programmed by using an additional bias resistor
as shown in Figure 4. The following formulas can be used
to compute the values of RNOM and R1:
RNOM =
rCOLD – rHOT
• R25
2.714
0.765 • VCC
(NTC RISING)
17
R1 = 0.536 • 105k – 0.4368 • 100k = 12.6k
20
–
+
NTC
3.266 – 0.4368
• 100k = 104.2k
2.714
the nearest 1% value is 105k.
NTC BLOCK
VCC
RNOM
100k
RNOM =
the nearest 1% value is 12.7k. The final solution is shown
in Figure 4 and results in an upper trip point of 45°C and
a lower trip point of 0°C.
R1 = 0.536 • RNOM – rHOT • R25
20
For example, to set the trip points to 0°C and 45°C with
a Vishay Curve 1 thermistor choose:
VCC
0.765 • VCC
(NTC RISING)
RNOM
105k
TOO_COLD
17
–
+
NTC
TOO_COLD
R1
12.7k
RNTC
100k
–
–
0.349 • VCC
(NTC FALLING)
+
RNTC
100k
TOO_HOT
0.349 • VCC
(NTC FALLING)
+
+
+
NTC_ENABLE
NTC_ENABLE
0.017 • VCC
(NTC FALLING)
TOO_HOT
–
0.017 • VCC
(NTC FALLING)
3558 F03
Figure 3. Typical NTC Thermistor Circuit
–
3558 F04
Figure 4. NTC Thermistor Circuit with Additional Bias Resistor
3558f
18
LTC3558
APPLICATIONS INFORMATION
USB and Wall Adapter Power
Although the battery charger is designed to draw power
from a USB port to charge Li-Ion batteries, a wall adapter
can also be used. Figure 5 shows an example of how to
combine wall adapter and USB power inputs. A P-channel
MOSFET, MP1, is used to prevent back conduction into
the USB port when a wall adapter is present and Schottky
diode, D1, is used to prevent USB power loss through the
1k pull-down resistor.
Typically, a wall adapter can supply significantly more
current than the 500mA-limited USB port. Therefore, an
N-channel MOSFET, MN1, and an extra program resistor are
used to increase the maximum charge current to 950mA
when the wall adapter is present.
5V WALL
ADAPTER
950mA ICHG
USB
POWER
500mA ICHG
BAT
BATTERY
CHARGER
VCC
MP1
PROG
MN1 1.65k
TA = 105°C – PDθ JA
TA = 105°C – ( VCC – VBAT ) • IBAT • θ JA
Example: Consider an LTC3558 operating from a USB port
providing 500mA to a 3.5V Li-Ion battery. The ambient
temperature above which the LTC3558 will begin to reduce
the 500mA charge current is approximately:
TA = 105°C – ( 5V – 3.5V ) • ( 500mA ) • 68°C / W
TA = 105°C – 0.75W • 68°C / W = 105°C – 51°C
TA = 54°C
IBAT
D1
current. It is not necessary to perform any worst-case
power dissipation scenarios because the LTC3558 will
automatically reduce the charge current to maintain the
die temperature at approximately 105°C. However, the
approximate ambient temperature at which the thermal
feedback begins to protect the IC is:
+
Li-Ion
BATTERY
The LTC3558 can be used above 70°C, but the charge current will be reduced from 500mA. The approximate current
at a given ambient temperature can be calculated:
IBAT =
1.74k
1k
3558 F05
Figure 5. Combining Wall Adapter and USB Power
Power Dissipation
The conditions that cause the LTC3558 to reduce charge
current through thermal feedback can be approximated
by considering the power dissipated in the IC. For high
charge currents, the LTC3558 power dissipation is
approximately:
PD = ( VCC – VBAT ) • IBAT
where PD is the power dissipated, VCC is the input supply
voltage, VBAT is the battery voltage, and IBAT is the charge
105°C – TA
( VCC – VBAT ) • θJA
Using the previous example with an ambient temperature of 88°C, the charge current will be reduced to
approximately:
IBAT =
105°C – 88°C
17°C
=
(5V – 3.5V ) • 68°C / W 102°C / A
IBAT = 167mA
Furthermore, the voltage at the PROG pin will change
proportionally with the charge current as discussed in
the Programming Charge Current section.
It is important to remember that LTC3558 applications do
not need to be designed for worst-case thermal conditions
since the IC will automatically reduce power dissipation
when the junction temperature reaches approximately
105°C.
3558f
19
LTC3558
APPLICATIONS INFORMATION
Battery Charger Stability Considerations
The LTC3558 battery charger contains two control loops: the
constant-voltage and constant-current loops. The constantvoltage loop is stable without any compensation when a
battery is connected with low impedance leads. Excessive
lead length, however, may add enough series inductance
to require a bypass capacitor of at least 1.5μF from BAT
to GND. Furthermore, a 4.7μF capacitor with a 0.2Ω to 1Ω
series resistor from BAT to GND is required to keep ripple
voltage low when the battery is disconnected.
High value capacitors with very low ESR (especially
ceramic) reduce the constant-voltage loop phase margin,
possibly resulting in instability. Ceramic capacitors up to
22μF may be used in parallel with a battery, but larger
ceramics should be decoupled with 0.2Ω to 1Ω of series
resistance.
In constant-current mode, the PROG pin is in the feedback
loop, not the battery. Because of the additional pole created
by the PROG pin capacitance, capacitance on this pin must
be kept to a minimum. With no additional capacitance on
the PROG pin, the charger is stable with program resistor
values as high as 25K. However, additional capacitance on
this node reduces the maximum allowed program resistor. The pole frequency at the PROG pin should be kept
above 100kHz. Therefore, if the PROG pin is loaded with a
capacitance, CPROG, the following equation should be used
to calculate the maximum resistance value for RPROG:
RPROG ≤
1
Average, rather than instantaneous, battery current may be
of interest to the user. For example, if a switching power
supply operating in low-current mode is connected in
parallel with the battery, the average current being pulled
out of the BAT pin is typically of more interest than the
instantaneous current pulses. In such a case, a simple RC
filter can be used on the PROG pin to measure the average
battery current as shown in Figure 6. A 10k resistor has
been added between the PROG pin and the filter capacitor
to ensure stability.
USB Inrush Limiting
When a USB cable is plugged into a portable product,
the inductance of the cable and the high-Q ceramic input
capacitor form an L-C resonant circuit. If there is not much
impedance in the cable, it is possible for the voltage at
the input of the product to reach as high as twice the
USB voltage (~10V) before it settles out. In fact, due to
the high voltage coefficient of many ceramic capacitors
(a nonlinearity), the voltage may even exceed twice the
USB voltage. To prevent excessive voltage from damaging the LTC3558 during a hot insertion, the soft connect
circuit in Figure 7 can be employed.
In the circuit of Figure 7, capacitor C1 holds MP1 off
when the cable is first connected. Eventually C1 begins
to charge up to the USB input voltage applying increasing
gate support to MP1. The long time constant of R1 and
C1 prevents the current from building up in the cable too
fast thus dampening out any resonant overshoot.
5
2π • 10 • CPROG
MP1
Si2333
VCC
LTC3558
10k
PROG
GND
RPROG
CFILTER
CHARGE
CURRENT
MONITOR
CIRCUITRY
5V USB
INPUT
C1
100nF
USB CABLE
R1
40k
C2
10μF
LTC3558
GND
3558 F06
3558 F07
Figure 6. Isolated Capacitive Load on PROG Pin and Filtering
Figure 7. USB Soft Connect Circuit
3558f
20
LTC3558
APPLICATIONS INFORMATION
Buck Switching Regulator General Information
Buck Switching Regulator
Output Voltage Programming
The LTC3558 contains a 2.25MHz constant-frequency
current mode buck switching regulator that can provide
up to 400mA. The switcher can be programmed for a
minimum output voltage of 0.8V and can be used to power
a microcontroller core, microcontroller I/O, memory or
other logic circuitry. The regulator supports 100% duty
cycle operation (dropout mode) when the input voltage
drops very close to the output voltage and is also capable
of operating in Burst Mode operation for highest efficiencies at light loads (Burst Mode operation is pin selectable).
The buck switching regulator also includes soft-start to
limit inrush current when powering on, short-circuit current protection, and switch node slew limiting circuitry to
reduce radiated EMI.
The buck switching regulator can be programmed for
output voltages greater than 0.8V. The output voltage
for the buck switching regulator is programmed using a
resistor divider from the switching regulator output connected to its feedback pin (FB1), as shown in Figure 8,
such that:
VOUT = 0.8(1 + R1/R2)
Typical values for R1 are in the range of 40k to 1M. The
capacitor CFB cancels the pole created by feedback resistors and the input capacitance of the FB pin and also
helps to improve transient response for output voltages
much greater than 0.8V. A variety of capacitor sizes can
be used for CFB but a value of 10pF is recommended for
most applications. Experimentation with capacitor sizes
between 2pF and 22pF may yield improved transient
response if so desired by the user.
A MODE pin sets the buck switching regulator in Burst
Mode operation or pulse skip operating mode. The regulator is enabled individually through its enable pin. The buck
regulator input supply (PVIN1) should be connected to the
battery pin (BAT) and PVIN2. This allows the undervoltage
lockout circuit on the BAT pin to disable the buck regulators
when the BAT voltage drops below 2.45V. Do not drive the
buck switching regulator from a voltage other than BAT.
A 10μF decoupling capacitor from the PVIN1 pin to GND
is recommended.
Buck Switching Regulator Operating Modes
The buck switching regulator includes two possible operating modes to meet the noise/power needs of a variety
of applications.
In pulse skip mode, an internal latch is set at the start of
every cycle, which turns on the main P-channel MOSFET
PVIN
EN
MP
PWM
CONTROL
MODE
SW
MN
L
VOUT
CO
CFB
R1
FB
0.8V
GND
R2
3558 F08
Figure 8. Buck Converter Application Circuit
3558f
21
LTC3558
APPLICATIONS INFORMATION
switch. During each cycle, a current comparator compares
the peak inductor current to the output of an error amplifier.
The output of the current comparator resets the internal
latch, which causes the main P-channel MOSFET switch to
turn off and the N-channel MOSFET synchronous rectifier
to turn on. The N-channel MOSFET synchronous rectifier
turns off at the end of the 2.25MHz cycle or if the current
through the N-channel MOSFET synchronous rectifier
drops to zero. Using this method of operation, the error
amplifier adjusts the peak inductor current to deliver the
required output power. All necessary compensation is
internal to the buck switching regulator requiring only a
single ceramic output capacitor for stability. At light loads
in pulse skip mode, the inductor current may reach zero
on each pulse which will turn off the N-channel MOSFET
synchronous rectifier. In this case, the switch node (SW1)
goes high impedance and the switch node voltage will
“ring”. This is discontinuous operation, and is normal behavior for a switching regulator. At very light loads in pulse
skip mode, the buck switching regulator will automatically
skip pulses as needed to maintain output regulation. At
high duty cycle (VOUT > PVIN1 /2) in pulse skip mode, it is
possible for the inductor current to reverse causing the
buck converter to switch continuously. Regulation and
low noise operation are maintained but the input supply
current will increase to a couple mA due to the continuous
gate switching.
During Burst Mode operation, the buck switching regulator automatically switches between fixed frequency PWM
operation and hysteretic control as a function of the load
current. At light loads the buck switching regulator controls
the inductor current directly and use a hysteretic control
loop to minimize both noise and switching losses. During
Burst Mode operation, the output capacitor is charged to a
voltage slightly higher than the regulation point. The buck
switching regulator then goes into sleep mode, during
which the output capacitor provides the load current. In
sleep mode, most of the switching regulator’s circuitry is
powered down, helping conserve battery power. When
the output voltage drops below a pre-determined value,
the buck switching regulator circuitry is powered on and
another burst cycle begins. The sleep time decreases as the
load current increases. Beyond a certain load current point
(about 1/4 rated output load current) the buck switching
regulator will switch to a low noise constant-frequency
PWM mode of operation, much the same as pulse skip
operation at high loads. For applications that can tolerate
some output ripple at low output currents, Burst Mode
operation provides better efficiency than pulse skip at
light loads.
The buck switching regulator allows mode transition onthe-fly, providing seamless transition between modes even
under load. This allows the user to switch back and forth
between modes to reduce output ripple or increase low
current efficiency as needed. Burst Mode operation is set
by driving the MODE pin high, while pulse skip mode is
achieved by driving the MODE pin low.
Buck Switching Regulator in Shutdown
The buck switching regulator is in shutdown when not
enabled for operation. In shutdown, all circuitry in the buck
switching regulator is disconnected from the regulator input
supply, leaving only a few nanoamps of leakage pulled to
ground through a 13k resistor on the switch (SW1) pin
when in shutdown.
Buck Switching Regulator Dropout Operation
It is possible for the buck switching regulator’s input voltage to approach its programmed output voltage (e.g., a
battery voltage of 3.4V with a programmed output voltage
of 3.3V). When this happens, the PMOS switch duty cycle
increases until it is turned on continuously at 100%. In this
dropout condition, the respective output voltage equals the
regulator’s input voltage minus the voltage drops across
the internal P-channel MOSFET and the inductor.
3558f
22
LTC3558
APPLICATIONS INFORMATION
Buck Switching Regulator Soft-Start Operation
Buck Switching Regulator Inductor Selection
Soft-start is accomplished by gradually increasing the peak
inductor current for each switching regulator over a 500μs
period. This allows an output to rise slowly, helping minimize the battery in-rush current required to charge up the
regulator’s output capacitor. A soft-start cycle occurs when
the buck switcher first turns on, or after a fault condition
has occurred (thermal shutdown or UVLO). A soft-start
cycle is not triggered by changing operating modes using
the MODE pin. This allows seamless output operation when
transitioning between operating modes.
The buck switching regulator is designed to work with
inductors in the range of 2.2μH to 10μH, but for most
applications a 4.7μH inductor is suggested. Larger value
inductors reduce ripple current which improves output
ripple voltage. Lower value inductors result in higher
ripple current which improves transient response time.
To maximize efficiency, choose an inductor with a low DC
resistance. For a 1.2V output efficiency is reduced about 2%
for every 100mΩ series resistance at 400mA load current,
and about 2% for every 300mΩ series resistance at 100mA
load current. Choose an inductor with a DC current rating
at least 1.5 times larger than the maximum load current to
ensure that the inductor does not saturate during normal
operation. If output short-circuit is a possible condition
the inductor should be rated to handle the maximum peak
current specified for the buck regulators.
Buck Switching Regulator
Switching Slew Rate Control
The buck switching regulator contains circuitry to limit
the slew rate of the switch node (SW1). This circuitry is
designed to transition the switch node over a period of a
couple of nanoseconds, significantly reducing radiated
EMI and conducted supply noise while maintaining high
efficiency.
Buck Switching Regulator Low Supply Operation
An undervoltage lockout (UVLO) circuit on PVIN1 shuts
down the step-down switching regulators when BAT drops
below 2.45V. This UVLO prevents the buck switching regulator from operating at low supply voltages where loss of
regulation or other undesirable operation may occur.
Different core materials and shapes will change the size/current and price/current relationship of an inductor. Toroid or
shielded pot cores in ferrite or permalloy materials are small
and don’t radiate much energy, but generally cost more
than powdered iron core inductors with similar electrical
characteristics. Inductors that are very thin or have a very
small volume typically have much higher DCR losses, and
will not give the best efficiency. The choice of which style
inductor to use often depends more on the price vs size,
performance, and any radiated EMI requirements than on
what the buck regulator requires to operate.
The inductor value also has an effect on Burst Mode
operation. Lower inductor values will cause Burst Mode
switching frequency to increase.
3558f
23
LTC3558
APPLICATIONS INFORMATION
Table 2 shows several inductors that work well with the
LTC3558 buck switching regulator. These inductors offer
a good compromise in current rating, DCR and physical
size. Consult each manufacturer for detailed information
on their entire selection of inductors.
Buck Switching Regulator
Input/Output Capacitor Selection
Low ESR (equivalent series resistance) ceramic capacitors
should be used at switching regulator outputs as well as
the switching regulator input supply. Ceramic capacitor
dielectrics are a compromise between high dielectric
constant and stability versus temperature and versus
DC bias voltage. The X5R/X7R dielectrics offer the best
compromise with high dielectric constant and acceptable
performance over temperature and under bias. Do not
use Y5V dielectrics. A 10μF output capacitor is sufficient
for most applications. For good transient response and
stability the output capacitor should retain at least 4μF
of capacitance over operating temperature and bias voltage. The buck switching regulator input supply should be
bypassed with a 10μF capacitor. Consult manufacturer
for detailed information on their selection and specifications of ceramic capacitors. Many manufacturers now
offer very thin (< 1mm tall) ceramic capacitors ideal for
use in height-restricted designs. Table 3 shows a list of
several ceramic capacitor manufacturers.
Table 3: Recommended Ceramic Capacitor Manufacturers
AVX
(803) 448-9411
www.avxcorp.com
Murata
(714) 852-2001
www.murata.com
Taiyo Yuden
(408) 537-4150
www.t-yuden.com
TDK
(888) 835-6646
www.tdk.com
Table 2. Recommended Inductors for Buck Switching Regulators
INDUCTOR TYPE
L
(μH)
MAX IDC
(A)
MAX DCR
(mΩ)
SIZE IN mm
(L × W × H)
MANUFACTURER
DE2818C
DE2812C
4.7
4.7
1.25
1.15
72*
130*
3 × 2.8 × 1.8
3 × 2.8 × 1.2
Toko
www.toko.com
CDRH3D16
4.7
0.9
110
4 × 4 × 1.8
Sumida
www.sumida.com
SD3118
SD3112
4.7
4.7
1.3
0.8
162
246
3.1 × 3.1 × 1.8
3.1 × 3.1 × 1.2
Cooper
www.cooperet.com
LPS3015
4.7
1.1
200
3 × 3 × 1.5
Coilcraft
www.coilcraft.com
*Typical DCR
3558f
24
LTC3558
APPLICATIONS INFORMATION
Buck-Boost Switching Regulator
The LTC3558 contains a 2.25MHz constant-frequency,
voltage mode, buck-boost switching regulator. The regulator provides up to 400mA of output load current. The
buck-boost switching regulator can be programmed for a
minimum output voltage of 2.75V and can be used to power
a microcontroller core, microcontroller I/O, memory, disk
drive, or other logic circuitry. To suit a variety of applications, different mode functions allow the user to trade off
noise for efficiency. Two modes are available to control the
operation of the buck-boost regulator. At moderate to heavy
loads, the constant-frequency PWM mode provides the
least noise switching solution. At lighter loads, Burst Mode
operation may be selected. Regulation is maintained by an
error amplifier that compares the divided output voltage
with a reference and adjusts the compensation voltage
accordingly until the FB2 voltage has stabilized at 0.8V. The
buck-boost switching regulator also includes soft-start to
limit inrush current and voltage overshoot when powering
on, short-circuit current protection, and switch node slew
limiting circuitry for reduced radiated EMI.
Buck-Boost Regulator PWM Operating Mode
In PWM mode, the voltage seen at the feedback node is
compared to a 0.8V reference. From the feedback voltage,
an error amplifier generates an error signal seen at the
VC2 pin. This error signal controls PWM waveforms that
modulate switches A (input PMOS), B (input NMOS), C
(output NMOS), and D (output PMOS). Switches A and
B operate synchronously, as do switches C and D. If the
input voltage is significantly greater than the programmed
output voltage, then the regulator will operate in buck
mode. In this case, switches A and B will be modulated,
with switch D always on (and switch C always off), to stepdown the input voltage to the programmed output. If the
input voltage is significantly less than the programmed
output voltage, then the converter will operate in boost
mode. In this case, switches C and D are modulated, with
switch A always on (and switch B always off), to step up
the input voltage to the programmed output. If the input
voltage is close to the programmed output voltage, then
the converter will operate in four-switch mode. While
operating in four-switch mode, switches turn on as per
the following sequence: switches A and D → switches A
and C → switches B and D → switches A and D.
Buck-Boost Regulator Burst Mode Operation
In Burst Mode operation, the switching regulator uses a
hysteretic feedback voltage algorithm to control the output
voltage. By limiting FET switching and using a hysteretic
control loop switching losses are greatly reduced. In
this mode, output current is limited to 50mA. While in
Burst Mode operation, the output capacitor is charged
to a voltage slightly higher than the regulation point. The
buck-boost converter then goes into a SLEEP state, during which the output capacitor provides the load current.
The output capacitor is charged by charging the inductor
until the input current reaches 250mA typical, and then
discharging the inductor until the reverse current reaches
0mA typical. This process of bursting current is repeated
until the feedback voltage has charged to the reference
voltage plus 6mV (806mV typical). In the SLEEP state,
most of the regulator’s circuitry is powered down, helping
to conserve battery power. When the feedback voltage
drops below the reference voltage minus 6mV (794mV
typical), the switching regulator circuitry is powered on
and another burst cycle begins. The duration for which the
regulator operates in SLEEP depends on the load current
and output capacitor value. The SLEEP time decreases
as the load current increases. The maximum deliverable
load current in Burst Mode operation is 50mA typical.
The buck-boost regulator may not enter SLEEP if the load
current is greater than 50mA. If the load current increases
beyond this point while in Burst Mode operation, the output may lose regulation. Burst Mode operation provides a
significant improvement in efficiency at light loads at the
expense of higher output ripple when compared to PWM
mode. For many noise-sensitive systems, Burst Mode
operation might be undesirable at certain times (i.e., during a transmit or receive cycle of a wireless device), but
highly desirable at others (i.e., when the device is in low
power standby mode).
3558f
25
LTC3558
APPLICATIONS INFORMATION
Buck-Boost Switching Regulator Output Voltage
Programming
The output filter zero is given by:
f FILTER _ ZERO =
The buck-boost switching regulator can be programmed
for output voltages greater than 2.75V and less than 5.45V.
To program the output voltage, a resistor divider is connected between VOUT2 and the feedback node (FB2) as
shown in Figure 9. The output voltage is given by VOUT2
= 0.8(1 + R1/R2).
2 • π • RESR • COUT
Hz
where RESR is the capacitor equivalent series resistance.
A troublesome feature in boost mode is the right-half
plane zero (RHP), and is given by:
f RHPZ =
LTC3558
VOUT2
R1
PVIN22
Hz
2 • π • IOUT • L • VOUT2
The loop gain is typically rolled off before the RHP zero
frequency.
FB2
R2
A simple Type I compensation network, as shown in Figure
10, can be incorporated to stabilize the loop, but at the
cost of reduced bandwidth and slower transient response.
To ensure proper phase margin, the loop requires to be
crossed over a decade before the LC double pole.
3558 F09
Figure 9. Programming the Buck-Boost Output Voltage Requires
a Resistor Divider Connected Between VOUT2 and FB2
The unity-gain frequency of the error amplifier with the
Type I compensation is given by:
Closing the Feedback Loop
The LTC3558 incorporates voltage mode PWM control. The
control to output gain varies with operation region (buck,
boost, buck-boost), but is usually no greater than 20. The
output filter exhibits a double pole response given by:
f FILTER _ POLE =
1
f UG =
1
Hz
2 • π • R1• CP1
1
Hz
2 • π • L • COUT
where COUT is the output filter capacitor.
VOUT2
+
ERROR
AMP
0.8V
R1
FB2
–
VC2
CP1
R2
3558 F10
Figure 10. Error Amplifier with Type I Compensation
3558f
26
LTC3558
APPLICATIONS INFORMATION
Most applications demand an improved transient response
to allow a smaller output filter capacitor. To achieve a higher
bandwidth, Type III compensation is required. Two zeros
are required to compensate for the double-pole response.
Type III compensation also reduces any VOUT2 overshoot
seen during a start-up condition. A Type III compensation circuit is shown in Figure 11 and yields the following
transfer function:
VC2
1
=
VOUT 2 R1 (C1 + C2)
•
(1 + sR2C2) [1 + s (R1 + R3)C3 ]
s ⎣⎡1 + sR2(C1|| C2)⎤⎦ (1 + sR3C3)
A Type III compensation network attempts to introduce
a phase bump at a higher frequency than the LC double
pole. This allows the system to cross unity gain after the
LC double pole, and achieve a higher bandwidth. While
attempting to cross over after the LC double pole, the
system must still cross over before the boost right-half
plane zero. If unity gain is not reached sufficiently before
the right-half plane zero, then the –180° of phase lag from
the LC double pole combined with the –90° of phase lag
from the right-half plane zero will result in negating the
phase bump of the compensator.
at the filter double pole. If they are placed at too low of a
frequency, they will introduce too much gain to the system
and the crossover frequency will be too high. The two high
frequency poles should be placed such that the system
crosses unity gain during the phase bump introduced
by the zeros and before the boost right-half plane zero
and such that the compensator bandwidth is less than
the bandwidth of the error amp (typically 900kHz). If the
gain of the compensation network is ever greater than
the gain of the error amplifier, then the error amplifier no
longer acts as an ideal op amp, and another pole will be
introduced at the same point.
Recommended Type III compensation components for a
3.3V output are:
R1: 324kΩ
RFB: 105kΩ
C1: 10pF
R2: 15kΩ
C2: 330pF
R3: 121kΩ
C3: 33pF
COUT : 22μF
The compensator zeros should be placed either before
or only slightly after the LC double pole such that their
positive phase contributions offset the –180° that occurs
LOUT : 2.2μH
VOUT2
+
ERROR
AMP
R3
0.8V
R1
C3
FB2
–
VC2
RFB
C2
R2
C1
3558 F11
Figure 11. Error Amplifier with Type III Compensation
3558f
27
LTC3558
APPLICATIONS INFORMATION
Input Current Limit
Buck-Boost Switching Regulator Inductor Selection
The input current limit comparator will shut the input PMOS
switch off once current exceeds 700mA typical. Before the
switch current limit, the average current limit amp (620mA
typical) will source current into the feedback pin to drop
the output voltage. The input current limit also protects
against a short-circuit condition at the VOUT2 pin.
The buck-boost switching regulator is designed to work
with inductors in the range of 1μH to 5μH. For most
applications, a 2.2μH inductor will suffice. Larger value
inductors reduce ripple current which improves output
ripple voltage. Lower value inductors result in higher
ripple current and improved transient response time.
To maximize efficiency, choose an inductor with a low
DC resistance and a DC current rating at least 1.5 times
larger than the maximum load current to ensure that the
inductor does not saturate during normal operation. If
output short-circuit is a possible condition, the inductor
current should be rated to handle up to the peak current
specified for the buck-boost regulator.
Reverse Current Limit
The reverse current limit comparator will shut the output
PMOS switch off once current returning from the output
exceeds 450mA typical.
Output Overvoltage Protection
If the feedback node were inadvertently shorted to ground,
then the output would increase indefinitely with the maximum current that could be sourced from the input supply.
The buck-boost regulator protects against this by shutting
off the input PMOS if the output voltage exceeds a 5.75V
maximum.
Buck-Boost Regulator Soft-Start Operation
Soft-start is accomplished by gradually increasing the
reference voltage over a 500μs typical period. A softstart cycle occurs whenever the buck-boost is enabled,
or after a fault condition has occurred (thermal shutdown
or UVLO). A soft-start cycle is not triggered by changing
operating modes. This allows seamless output operation
when transitioning between Burst Mode operation and
PWM mode operation.
The inductor value also affects Burst Mode operation.
Lower inductor values will cause Burst Mode switching
frequencies to increase.
Different core materials and shapes will change the size/current and price/current relationship of an inductor. Toroid
or shielded pot cores in ferrite or permalloy materials
are small and do not radiate much energy, but cost more
than powdered iron core inductors with similar electrical
characteristics. Inductors that are very thin or have a very
small volume typically have much higher core and DCR
losses and will not give the best efficiency.
Table 4 shows some inductors that work well with the
buck-boost regulator. These inductors offer a good compromise in current rating, DCR and physical size. Consult
each manufacturer for detailed information on their entire
selection of inductors.
Table 4. Recommended Inductors for the Buck-Boost Switching Regulator.
INDUCTOR TYPE
L
(μH)
MAX IDC
(A)
MAX DCR
(mΩ)
SIZE IN mm
(L × W × H)
DB3018C
D312C
DE2812C
DE2812C
2.4
2.2
2
2.7
1.31
1.14
1.4
1.2
80
140
81
87
3.8 × 3.8 × 1.4
3.6 × 3.6 × 1.2
3 × 3.2 × 1.2
3 × 3.2 × 1.2
Toko
www.toko.com
CDRH3D16
2.2
1.2
72
4 × 4 × 1.8
Sumida
www.sumida.com
SD12
2.2
1.8
74
5.2 × 5.2 × 1.2
Cooper
www.cooperet.com
MANUFACTURER
*Typical DCR
3558f
28
LTC3558
APPLICATIONS INFORMATION
Buck-Boost Switching Regulator Input/Output
Capacitor Selection
Low ESR (equivalent series resistance) ceramic capacitors
should be used at both the buck-boost regulator input
(PVIN2) and the output (VOUT2). It is recommended that the
input be bypassed with a 10μF capacitor. The output should
be bypassed with at least a 10μF capacitor if using Type I
compensation and 22μF if using Type III compensation.
The same selection criteria apply for the buck-boost
regulator input and output capacitors as described in the
Buck Switching Regulator Input/Output Capacitor Selection section.
PCB Layout Considerations
In order to deliver maximum charge current under all
conditions, it is critical that the backside of the LTC3558
be soldered to the PC board ground.
The LTC3558 has dual switching regulators. As with all
switching regulators, care must be taken while laying out
a PC board and placing components. The input decoupling
capacitors, the output capacitor and the inductors must all
be placed as close to the pins as possible and on the same
side of the board as the LTC3558. All connections must
also be made on the same layer. Place a local unbroken
ground plane below these components. Avoid routing
noisy high frequency lines such as those that connect to
switch pins over or parallel to lines that drive high impedance inputs.
3558f
29
LTC3558
TYPICAL APPLICATIONS
UP TO 500mA
USB
(4.3V TO 5.5V)
OR AC ADAPTER
VCC
110k
BAT
PVIN1
PVIN2
10μF
NTC
+
1
10μF
4.7μF
28.7K
100k (NTC)
NTH50603NO1
510Ω
SINGLE
Li-lon CELL
(2.7V TO 4.2V)
LTC3558
1.8V AT 400mA
4.7μH
SW1
CHRG
806k
1.74k
PROG
FB1
SUSP
SWAB2
EN1
EN2
10μF
649k
2.2μH
HPWR
DIGITAL
CONTROL
10pF
3.3V AT 400mA
SWCD2
VOUT2
619k
MODE
10μF
FB2
GND2
(EXPOSED
GND
PAD)
200k
15k
150pF
VC2
3558 TA02
Figure 12. Li-Ion to 3.3V at 400mA, 1.8V at 400mA and USB-Compatible Battery Charger
As shown in Figure 12, the LTC3558 can be operated
with no battery connected to the BAT pin. A 1Ω resistor
in series with a 4.7μF capacitor at the BAT pin ensures
battery charger stability. 10μF VCC decoupling capacitors
are required for proper operation of the DC/DC converters.
A three-resistor bias network for NTC sets hot and cold
trip points at approximately 55°C and 0°C.
The battery can be charged with up to 950mA of charge
current when powered from a 5V wall adaptor, as shown
in Figure 13. CHRG has a LED to provide a user with a
visual indication of battery charge status. The buck-boost
regulator starts up only after VOUT1 is up to approximately
0.7V. This provides a sequencing function which may be
desirable in applications where a microprocessor needs to
be powered up before peripherals. A Type III compensation
network improves the transient response of the buck-boost
switching regulator.
3558f
30
LTC3558
PACKAGE DESCRIPTION
UD Package
20-Lead Plastic QFN (3mm × 3mm)
(Reference LTC DWG # 05-08-1720 Rev A)
0.70 ±0.05
3.50 ± 0.05
(4 SIDES)
1.65 ± 0.05
2.10 ± 0.05
PACKAGE
OUTLINE
0.20 ±0.05
0.40 BSC
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED
3.00 ± 0.10
(4 SIDES)
BOTTOM VIEW—EXPOSED PAD
R = 0.115
TYP
0.75 ± 0.05
R = 0.05
TYP
PIN 1
TOP MARK
(NOTE 6)
PIN 1 NOTCH
R = 0.20 TYP
OR 0.25 × 45°
CHAMFER
19 20
0.40 ± 0.10
1
2
1.65 ± 0.10
(4-SIDES)
(UD20) QFN 0306 REV A
0.200 REF
0.00 – 0.05
NOTE:
1. DRAWING IS NOT A JEDEC PACKAGE OUTLINE
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
0.20 ± 0.05
0.40 BSC
3558f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
31
LTC3558
TYPICAL APPLICATIONS
UP TO 950mA
5V WALL
ADAPTER
100k
510Ω
VCC
BAT
PVIN1
NTC
PVIN2
1μF
10μF
1.2V AT 400mA
4.7μH
LTC3558
100k (NTC)
SW1
324k
CHRG
FB1
PROG
SWAB2
887Ω
DIGITAL
CONTROL
SUSP
HPWR
MODE
EN1
SINGLE
Li-lon CELL
(2.7V TO 4.2V)
+
10pF
10μF
649k
2.2μH
3.3V AT 400mA
SWCD2
VOUT2
121k
324k
22μF
33pF
EN2
FB2
105k
15k
330pF
10pF
VC2
3558 TA03
GND2
(EXPOSED
PAD)
GND
Figure 13. Battery Charger Can Charge a Battery with Up to 950mA When Powered From a Wall Adapter
RELATED PARTS
PART NUMBER DESCRIPTION
COMMENTS
LTC3550
Dual Input USB/AC Adapter Li-Ion Battery
Charger with Adjustable Output 600mA
Buck Converter
Synchronous Buck Converter, Efficiency: 93%, Adjustable Output at 600mA, Charge Current:
950mA Programmable, USB Compatible, Automatic Input Power Detection
and Selection
LTC3552
Standalone Linear Li-Ion Battery Charger
with Adjustable Output Dual Synchronous
Buck Converter
Synchronous Buck Converter, Efficiency: >90%, Adjustable Outputs at 800mA and
400mA, Charge Current Programmable Up to 950mA, USB Compatible, 5mm × 3mm
DFN-16 Package
LTC3552-1
Standalone Linear Li-Ion Battery Charger
with Dual Synchronous Buck Converter
Synchronous Buck Converter, Efficiency: >90%, Outputs 1.8V at 800mA and 1.575 at
400mA, Charge Current Programmable up to 950mA, USB Compatible
LTC3455
Dual DC/DC Converter with USB Power
Manager and Li-Ion Battery Charger
Seamless Transition Between Input Power Sources: Li-Ion Battery, USB and 5V Wall
Adapter, Two High Efficiency DC/DC Converters: Up to 96%, Full Featured Li-Ion Battery
Charger with Accurate USB Current Limiting (500mA/100mA) Pin-Selectable Burst Mode
Operation, Hot SwapTM Output for SDIO and Memory Cards, 4mm × 4mm QFN-24 Package
LTC3456
2-Cell, Multi-Output DC/DC Converter with Seamless Transition Between 2-Cell Battery, USB and AC Wall Adapter Input Power Sources,
USB Power Manager
Main Output: Fixed 3.3V Output, Core Output: Adjustable from 0.8V to VBATT(MIN), Hot Swap
Output for Memory Cards, Power Supply Sequencing: Main and Hot Swap Accurate USB
Current Limiting, High Frequency Operation: 1MHz, High Efficiency: Up to 92%, 4mm ×
4mm QFN-24 Package
LTC3559
USB Charger with Dual Buck Regulators
Adjustable, Synchronous Buck Converters, Efficiency >90%, Outputs: Down to 0.8V at
400mA Each, Charge Current Programmable Up to 950mA, USB-Compatible, 3mm × 3mm
QFN-16 Package
LTC4080
500mA Standalone Charger with 300mA
Synchronous Buck
Charges Single-Cell Li-Ion Batteries, Timer Termination + C/10, Thermal Regulation, Buck
Output: 0.8V to VBAT, Buck Input VIN: 2.7V to 5.5V, 3mm × 3mm DFN-10 Package
Hot Swap is a trademark of Linear Technology Corporation.
3558f
32 Linear Technology Corporation
LT 0408 • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900
●
FAX: (408) 434-0507 ● www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2008
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