TI1 ADS1220 Low-power, low-noise, 24-bit, analog-to-digital converter for small-signal sensor Datasheet

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SBAS501A – MAY 2013 – REVISED JULY 2013
Low-Power, Low-Noise, 24-Bit, Analog-to-Digital Converter for Small-Signal Sensors
FEATURES
DESCRIPTION
•
The ADS1220 is a precision, 24-bit, analog-to-digital
converter (ADC) offered in a leadless QFN-16 or a
TSSOP-16 package. The device features two
differential or four single-ended inputs through a very
flexible input multiplexer (mux), a low-noise,
programmable
gain
amplifier
(PGA),
two
programmable excitation current sources, an internal
reference, an oscillator, a low-side bridge switch, and
a precision temperature sensor. The many integrated
features and the simple control of the ADS1220
through an SPI-compatible interface ease precision
measurements of the most common sensor signals.
1
23
•
•
•
•
•
•
•
•
•
•
•
•
Low Current Consumption:
– Duty-Cycle Mode: 120 μA
– Normal Mode: 415 μA
Wide Supply Range: 2.3 V to 5.5 V
Programmable Gain: 1 V/V to 128 V/V
Programmable Data Rates: Up to 2 kSPS
50-Hz and 60-Hz Rejection at 20 SPS
Low-Noise PGA: 90 nVRMS at 20 SPS
Dual Matched Programmable Current Sources:
10 μA to 1500 μA
Internal Temperature Sensor:
0.5°C Error (max)
Low-Drift Internal Reference
Low-Drift Internal Oscillator
Two Differential or Four Single-Ended Inputs
SPI™-Compatible Interface
3,5 mm × 3,5 mm × 0,9 mm QFN Package
APPLICATIONS
•
•
•
•
Temperature Sensors:
– Thermocouples
– Resistance Temperature Detectors (RTDs)
– 2-, 3-, and 4-Wire RTD Excitation
Bridge Sensors
Portable Instrumentation
Factory Automation and Process Control
The device can perform conversions at data rates of
up to 2000 samples-per-second (SPS) with singlecycle settling. The internal PGA offers gains of up to
128 V/V. This PGA makes the ADS1220 ideallysuited for applications measuring small signals, such
as thermocouples, resistance temperature detectors
(RTDs), thermistors, and bridge sensors. The device
supports true bipolar analog supplies in the event that
single-ended signals referenced to ground must be
measured using the PGA. Alternatively, the device
can be configured to bypass the internal PGA while
still providing gains of up to 4 V/V, allowing for rail-torail input signals with no loss of signal integrity when
running from a single analog supply.
The device operates in either duty-cycle mode
(consuming 120 µA of current), normal mode
(consuming 415 µA of current), or turbo mode (for
highest data rates). The ADS1220 operates over a
temperature range of –40°C to +125°C.
REFP0
AVDD
REFN0
DVDD
10 A to
1.5 mA
Internal
Reference
AIN0/REFP1
Reference
Mux
Device
24-bit
ûADC
Digital Filter
and
SPI
Interface
Low Drift
Oscillator
Precision
Temp Sensor
CLK
DGND
AIN1
Mux
PGA
AIN2
AIN3/REFN1
AVSS
CS
SCLK
DIN
DOUT/DRDY
DRDY
1
2
3
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
SPI is a trademark of Motorola, Inc.
All other trademarks are the property of their respective owners.
UNLESS OTHERWISE NOTED this document contains
PRODUCTION DATA information current as of publication date.
Products conform to specifications per the terms of Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2013, Texas Instruments Incorporated
ADS1220
SBAS501A – MAY 2013 – REVISED JULY 2013
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This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
ORDERING INFORMATION
For the most current package and ordering information, see the Package Option Addendum at the end of this
document, or visit the device product folder at www.ti.com.
PRODUCT FAMILY
DEVICE
RESOLUTION (Bits)
MAXIMUM GAIN
MAXIMUM SAMPLE
RATE (SPS)
ADS1120
16
128
2000
ADS1220
24
128
2000
PACKAGE
DESIGNATOR
QFN-16
TSSOP-16
QFN-16
TSSOP-16
ABSOLUTE MAXIMUM RATINGS (1)
VALUE
UNIT
MIN
MAX
AVDD to AVSS
–0.3
+7
V
DVDD to DGND
–0.3
+7
V
AVSS to DGND
–2.8
+0.3
V
V
Analog input voltage
AIN0/REFP1, AIN1, AIN2, AIN3/REFN1, REFP0, REFN0
AVSS – 0.3
AVDD + 0.3
Digital input voltage
CS, SCLK, DIN, DOUT/DRDY, DRDY, CLK
Analog input current
Temperature
DGND – 0.3
DVDD + 0.3
Momentary
–100
+100
mA
Continuous
–10
+10
mA
Maximum junction, TJMax
(1)
+150
°C
–60
+150
°C
–2000
+2000
V
–500
+500
V
Storage, Tstg
Human body model (HBM)
Electrostatic discharge (ESD) JEDEC standard 22, test method A114-C.01, all pins
ratings
Charged device model (CDM)
JEDEC standard 22, test method C101, all pins
V
Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to absolute
maximum conditions for extended periods may affect device reliability.
THERMAL INFORMATION
ADS1220
THERMAL METRIC
(1)
QFN (RVA)
TSSOP (PW)
16 PINS
16 PINS
θJA
Junction-to-ambient thermal resistance
43.4
99.5
θJCtop
Junction-to-case (top) thermal resistance
47.3
35.2
θJB
Junction-to-board thermal resistance
18.4
44.3
ψJT
Junction-to-top characterization parameter
0.6
2.4
ψJB
Junction-to-board characterization parameter
18.4
43.8
θJCbot
Junction-to-case (bottom) thermal resistance
2.0
n/a
(1)
2
UNITS
°C/W
For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report, SPRA953.
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SBAS501A – MAY 2013 – REVISED JULY 2013
ELECTRICAL CHARACTERISTICS
Minimum and maximum specifications are at TA = –40°C to +125°C. Typical specifications are at TA = +25°C.
All specifications are at AVDD = 3.3 V, AVSS = 0 V, DVDD = 3.3 V, and DR = 20 SPS using external VREF = 2.5 V, unless
otherwise noted. (1)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
ANALOG INPUTS
Full-scale differential input voltage
range
VCM
±VREF / PGA (2)
VIN = (AINP – AINN)
Absolute input voltage
AINP or AINN, PGA disabled
Common-mode input voltage range
[VCM = (AINP + AINN) / 2]
PGA disabled (3)
(3)
PGA = 1...128
V
AVSS – 0.1
AVDD + 0.1
V
AVSS – 0.1
AVDD + 0.1
V
See the Low-Noise PGA section
Absolute input current
See the Typical Characteristics
Differential input current
See the Typical Characteristics
SYSTEM PERFORMANCE
Resolution
No missing codes
24
Normal mode
DR
Data rate
SPS
Duty-cycle mode
5, 11.25, 22.5, 44, 82.5, 150, 250
SPS
Turbo mode
40, 90, 180, 350, 660, 1200, 2000
SPS
Noise (input-referred)
INL
Integral nonlinearity
See the Noise Performance section
PGA = 1…128, VCM = 0.5 AVDD,
external reference, best fit
-15
PGA disabled, TA = +25°C,
differential inputs
VIO
Offset voltage (input-referred)
CMRR
PSRR
15
ppm
PGA = 1, TA = +25°C, differential inputs
–30
±4
µV
30
µV
±4
µV
PGA = 1…128, TA = –40°C to +85°C (4)
0.08
PGA = 1…128, TA = –40°C to +125°C
0.25
µV/°C
Offset match
Match between any two inputs
±20
µV
Gain error
PGA = 1…128, TA = +25°C
Gain drift
PGA = 1…128, TA = –40°C to +125°C (4)
Offset drift
NMRR
±6
±4
PGA = 2…128, TA = +25°C,
differential inputs
GE
Bits
20, 45, 90, 175, 330, 600, 1000
Normal-mode rejection ratio (5)
Common-mode rejection ratio
Power-supply rejection ratio
-0.1%
0.3
±0.015%
0.1%
1
4
µV/°C
ppm/°C
50 Hz ±3%, DR = 20 SPS, external CLK,
bit 50/60 = 10
105
dB
60 Hz ±3%, DR = 20 SPS, external CLK,
bit 50/60 = 11
105
dB
50 Hz or 60 Hz ±3%, DR = 20 SPS,
external CLK, Bit 50/60 = '01'
90
dB
At dc and PGA = 1
90
105
dB
fCM = 50 Hz, DR = 2000 SPS (4)
95
115
dB
fCM = 60 Hz, DR = 2000 SPS (4)
95
115
dB
AVDD at dc, VCM = 0.5 AVDD, PGA = 1
80
105
dB
100
115
dB
2.045
2.048
2.051
5
40
DVDD at dc, VCM = 0.5 AVDD, PGA = 1 (4)
INTERNAL VOLTAGE REFERENCE
Initial accuracy
TA = +25°C
Reference drift
TA = –40°C to +125°C (4)
V
ppm/°C
VOLTAGE REFERENCE INPUT
VREF
(1)
(2)
(3)
(4)
(5)
Reference input range
VREF = (REFPx – REFNx)
AVDD
V
Negative reference absolute input
REFNx to AVSS
AVSS – 0.1
0.75
2.5
REFPx – 0.75
V
Positive reference absolute input
REFPx to AVSS
REFNx + 0.75
AVDD + 0.1
Reference input current
REFN0 = AVSS, REFP0 = VREF
±10
V
nA
PGA disabled means the low-noise PGA is bypassed. Only gains of 1, 2, and 4 are possible in this case with the switched-capacitor
input structure. PGA = 1…128 denotes that the low-noise PGA is enabled and set to the respective gain setting.
Limited to [(AVDD – AVSS) – 0.4 V] / PGA, when the PGA is enabled.
See the Bypassing the PGA section for more information.
Minimum and maximum values are ensured by design and characterization data.
Minimum values are ensured by design.
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ELECTRICAL CHARACTERISTICS (continued)
Minimum and maximum specifications are at TA = –40°C to +125°C. Typical specifications are at TA = +25°C.
All specifications are at AVDD = 3.3 V, AVSS = 0 V, DVDD = 3.3 V, and DR = 20 SPS using external VREF = 2.5 V, unless
otherwise noted.(1)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
EXCITATION CURRENT SOURCES (IDACs)
Current settings
10, 50, 100, 250, 500, 1000, 1500
µA
Compliance voltage
All currents
AVDD – 0.9
Accuracy
All currents, each IDAC
Current match
Between IDACs
(not valid for 10-µA setting)
±0.3%
Temperature drift
Each IDAC
(not valid for 10-µA setting)
50
ppm/°C
Temperature drift matching
Between IDACs
(not valid for 10-µA setting)
10
ppm/°C
–6%
±1%
V
6%
CLOCK SOURCES
Internal oscillator accuracy
External clock
Normal mode
Frequency range
Duty cycle
–2%
±1%
0.5
4.096
40%
2%
4.5
MHz
60%
TEMPERATURE SENSOR
Temperature sensor resolution
Conversion resolution
TA = 0°C to +75°C
Temperature sensor accuracy
14
Temperature resolution
TA = –40°C to +125°C
Bits
0.03125
°C
–0.5
±0.25
0.5
–1
±0.5
1
0.0625
0.25
3.5
5.5
Ω
30
mA
vs analog supply voltage
°C
°C
°C/V
LOW-SIDE POWER SWITCH
RON
On resistance
Current through switch
DIGITAL INPUT/OUTPUT
VIH
High-level input voltage
0.7 DVDD
DVDD
V
VIL
Low-level input voltage
DGND – 0.3
0.3 DVDD
V
VOH
High-level output voltage
IOH = 3 mA
VOL
Low-level output voltage
IOL = 3 mA
IH
Input leakage, high
VIH = 5.5 V
IL
Input leakage, low
VIL = DGND
4
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0.8 DVDD
V
0.2 DVDD
V
–10
10
µA
–10
10
µA
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SBAS501A – MAY 2013 – REVISED JULY 2013
ELECTRICAL CHARACTERISTICS (continued)
Minimum and maximum specifications are at TA = –40°C to +125°C. Typical specifications are at TA = +25°C.
All specifications are at AVDD = 3.3 V, AVSS = 0 V, DVDD = 3.3 V, and DR = 20 SPS using external VREF = 2.5 V, unless
otherwise noted.(1)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
POWER-SUPPLY REQUIREMENTS
VDD
Supply voltage
Digital
DVDD to DGND
2.3
5.5
V
Analog,
unipolar
AVDD to AVSS, AVSS = DGND
2.3
5.5
V
AVDD to DGND
2.3
2.75
V
AVSS to DGND
–2.75
–2.3
V
3
µA
Analog,
bipolar
Power-down mode
0.1
Duty-cycle mode, PGA disabled
IAVDD
ICC
Supply current (6)
IDVDD
PD
Power dissipation (6)
65
µA
Normal mode, PGA disabled
240
Normal mode, PGA = 1…16
340
µA
Normal mode, PGA = 32
425
µA
Normal mode, PGA = 64, 128
510
µA
Turbo mode, PGA = 1…16
540
Power-down mode
0.3
Duty-cycle mode
55
Normal mode
75
Turbo mode
95
µA
Duty-cycle mode, PGA disabled
0.4
mW
Normal mode, PGA = 1…16
1.4
mW
Turbo mode, PGA = 1…16
2.1
mW
490
µA
µA
5
µA
µA
110
µA
TEMPERATURE RANGE
Tstg
(6)
Storage temperature
–60
+150
°C
Specified temperature
–40
+125
°C
Internal voltage reference selected, internal oscillator enabled, both IDACs turned off.
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SPI TIMING CHARACTERISTICS
tCSH
tCSSC
tSCLK
tSPWL
tDIHD
tCSDOD
§
§ §
DIN
tDOPD
tCSDOZ
Hi-Z
§ §
Hi-Z
§ §
DOUT/DRDY
§
tDIST
tSCCS
tSPWH
§
SCLK
§
§
CS
Figure 1. Serial Interface Timing
Timing Characteristics for Figure 1 (1)
PARAMETER
MIN
MAX
UNIT
tCSSC
CS low to first SCLK high: setup time
50
ns
tSCCS
Final SCLK falling edge to CS high
25
ns
tDIST
DIN setup time
50
ns
tDIHD
DIN hold time
25
tDOPD
SCLK rising edge to new data valid: propagation delay
(2)
0
ns
50
ns
tSCLK
SCLK period
150
ns
tSPWH
SCLK pulse width: high (2)
60
ns
tSPWL
SCLK pulse width: low (2)
60
ns
tCSDOZ
CS high to DOUT high impedance: propagation delay
50
ns
tCSDOD
CS low to DOUT driven: propagation delay
50
ns
tCSH
CS high pulse width
(1)
(2)
6
50
ns
At TA = –40°C to +125°C, DVDD = 2.3 V to 5.5 V, and DOUT load = 20 pF || 10 kΩ to DGND, unless otherwise noted.
If a complete command is not sent within 13955 × tMOD (normal mode, duty-cycle mode) or 27910 × tMOD (turbo mode), respectively, the
serial interface resets and the next SCLK pulse starts a new communication cycle. tMOD = 1 / fMOD. Modulator frequency (fMOD) is 256
kHz in normal and duty-cycle mode and 512 kHz in turbo mode when using the internal oscillator or an external 4.096-MHz clock.
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PIN CONFIGURATIONS
CS
SCLK
DIN
DOUT/DRDY
RVA PACKAGE
QFN-16
(TOP VIEW)
16
15
14
13
DGND
2
11 DVDD
AVSS
3
10 AVDD
AIN3/REFN1
4
9
AIN2
5
6
7
8
AIN1
12 DRDY
REFP0
1
REFN0
CLK
AIN0/REFP1
PIN DESCRIPTIONS (QFN PACKAGE)
NAME
PIN NO.
ANALOG OR DIGITAL
INPUT/OUTPUT
CLK
1
Digital input
DGND
2
Digital
Digital ground
AVSS
3
Analog
Negative analog power supply
AIN3/REFN1
4
Analog input
Differential or single-ended input; negative reference input
AIN2
5
Analog input
Differential or single-ended input
REFN0
6
Analog input
Negative reference input
REFP0
7
Analog input
Positive reference input
AIN1
8
Analog input
Differential or single-ended input
AIN0/REFP1
9
Analog input
Differential or single-ended input; positive reference input
AVDD
10
Analog
Positive analog power supply
DVDD
11
Digital
Positive digital power supply
DRDY
12
Digital output
Data ready; active low
DOUT/DRDY
13
Digital output
Serial data output combined with data ready; active low
DIN
14
Digital input
Serial data input
SCLK
15
Digital input
Serial clock input
CS
16
Digital input
Chip select; active low
Thermal pad
Thermal pad
—
DESCRIPTION
External clock source pin; connect to DGND if not used
Thermal power pad. Do not connect or only connect to AVSS.
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PW PACKAGE
TSSOP-16
(TOP VIEW)
SCLK
1
16
DIN
CS
2
15
DOUT/DRDY
CLK
3
14
DRDY
DGND
4
13
DVDD
AVSS
5
12
AVDD
AIN3/REFN1
6
11
AIN0/REFP1
AIN2
7
10
AIN1
REFN0
8
9
REFP0
PIN DESCRIPTIONS (TSSOP PACKAGE)
8
NAME
PIN NO.
ANALOG OR DIGITAL
INPUT/OUTPUT
SCLK
1
Digital input
Serial clock input
DESCRIPTION
CS
2
Digital input
Chip select; active low
CLK
3
Digital input
External clock source pin; connect to DGND if not used
DGND
4
Digital
Digital ground
AVSS
5
Analog
Negative analog power supply
AIN3/REFN1
6
Analog input
Differential or single-ended input; negative reference input
AIN2
7
Analog input
Differential or single-ended input
REFN0
8
Analog input
Negative reference input
REFP0
9
Analog input
Positive reference input
AIN1
10
Analog input
Differential or single-ended input
AIN0/REFP1
11
Analog input
Differential or single-ended input; positive reference input
AVDD
12
Analog
Positive analog power supply
DVDD
13
Digital
Positive digital power supply
DRDY
14
Digital output
Data ready; active low
DOUT/DRDY
15
Digital output
Serial data output combined with data ready; active low
DIN
16
Digital input
Serial data input
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SBAS501A – MAY 2013 – REVISED JULY 2013
TYPICAL CHARACTERISTICS
At TA = +25°C, AVDD = 3.3 V, and AVSS = 0 V using external VREF = 2.5 V, unless otherwise noted.
40
40
PGA = 1
PGA = 128
30
20
10
0
20
10
-10
-40
-20
0
20
40
60
80
100
120
Temperature (ƒC)
-40
-20
0
20
40
60
80
100
120
Temperature (ƒC)
C017
Figure 2. INPUT-REFERRED OFFSET VOLTAGE vs
TEMPERATURE (AVDD = 3.3 V)
C018
Figure 3. INPUT-REFERRED OFFSET VOLTAGE vs
TEMPERATURE (AVDD = 5.0 V)
500
500
PGA = 1
AVDD = 3.3 V
PGA = 128
400
PGA = 1
AVDD = 5.0 V
Gain Error (ppm of FS)
Gain Error (ppm of FS)
PGA Disabled
0
-10
PGA Disabled
300
200
100
0
PGA = 128
400
PGA Disabled
300
200
100
0
-40
-20
0
20
40
60
80
100
Temperature (ƒC)
120
-40
15
PGA = 32
INL (ppm of FS)
0
-5
AVDD = 3.3 V
External 2.5-V Reference
Normal Mode
-50
-25
0
25
50
75
VIN (% of FS)
60
80
100
120
C020
PGA = 32
10
5
-75
40
PGA = 1
PGA Disabled
PGA Disabled
-15
-100
20
Figure 5. GAIN ERROR vs TEMPERATURE
(AVDD = 5.0 V)
15
-10
0
Temperature (ƒC)
PGA = 1
10
-20
C019
Figure 4. GAIN ERROR vs TEMPERATURE
(AVDD = 3.3 V)
INL (ppm of FS)
AVDD = 5.0 V
PGA = 128
30
PGA Disabled
Offset Voltage (µV)
Offset Voltage (µV)
PGA = 1
AVDD = 3.3 V
5
0
-5
AVDD = 5.0 V
External 2.5-V Reference
Normal Mode
-10
100
-15
-100
C025
Figure 6. INTEGRAL NONLINEARITY vs
DIFFERENTIAL INPUT SIGNAL
(AVDD = 3.3 V, External Reference)
-75
-50
-25
0
25
50
75
VIN (% of FS)
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C029
Figure 7. INTEGRAL NONLINEARITY vs
DIFFERENTIAL INPUT SIGNAL
(AVDD = 5.0 V, External Reference)
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TYPICAL CHARACTERISTICS (continued)
At TA = +25°C, AVDD = 3.3 V, and AVSS = 0 V using external VREF = 2.5 V, unless otherwise noted.
20
15
PGA = 1
15
PGA = 32
PGA Disabled
PGA = 32
PGA Disabled
10
INL (ppm of FS)
10
INL (ppm of FS)
20
PGA = 1
5
0
-5
-10
5
0
-5
-10
AVDD = 3.3 V
Internal Reference
Normal Mode
-15
-20
-100
-75
-50
-25
0
25
50
75
VIN (% of FS)
AVDD = 5.0 V
Internal Reference
Normal Mode
-15
-20
-100
100
-25
0
25
50
75
100
C029
Figure 9. INTEGRAL NONLINEARITY vs
DIFFERENTIAL INPUT SIGNAL
(AVDD = 5.0 V, Internal Reference)
2.051
1000
AVDD = 3.3 V
Data from 5490 Devices
TA = +25°C
AVDD = 5.0 V
2.05
Reference Voltage (V)
800
Counts
-50
VIN (% of FS)
Figure 8. INTEGRAL NONLINEARITY vs
DIFFERENTIAL INPUT SIGNAL
(AVDD = 3.3 V, Internal Reference)
600
400
200
2.049
2.048
2.047
Initial Reference Voltage (V)
2.051
2.050
2.049
2.048
2.047
2.046
2.046
2.045
0
-75
C025
2.045
-40
-20
0
20
40
60
80
100
120
Temperature (ƒC)
C021
C042
Figure 10. INTERNAL REFERENCE VOLTAGE HISTOGRAM
Figure 11. INTERNAL REFERENCE VOLTAGE vs
TEMPERATURE
1
0
0.75
-20
0.5
-40
0.25
-60
PSRR (dB)
Frequency Error (%)
PGA = 1
0
-0.25
-0.5
-80
-100
-120
DVDD = 3.3 V
Normal Mode
-0.75
-140
-1
-160
-40
-20
0
20
40
60
Temperature (ƒC)
80
100
120
0.1
1
10
Frequency (kHz)
C002
Figure 12. INTERNAL OSCILLATOR ACCURACY vs
TEMPERATURE
10
PGA = 128
100
1000
C016
Figure 13. AVDD POWER-SUPPLY REJECTION RATIO vs
FREQUENCY
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TYPICAL CHARACTERISTICS (continued)
At TA = +25°C, AVDD = 3.3 V, and AVSS = 0 V using external VREF = 2.5 V, unless otherwise noted.
15
AIN0
AIN1
AIN2
AIN3
10
AVDD = 3.3 V
PGA Enabled
TA = -40°C
Absolute Input Current (nA)
Absolute Input Current (nA)
15
5
0
-5
-10
-15
AIN0
AIN1
AIN2
AIN3
10
5
0
-5
-10
-15
0.5
1
1.5
2
2.5
3
Absolute Input Voltage VAINx (V)
0.5
10
AIN0
AIN1
AIN2
AIN3
1.5
2
2.5
3
Absolute Input Voltage VAINx (V)
C031
Figure 15. ABSOLUTE INPUT CURRENT vs
ABSOLUTE INPUT VOLTAGE
(PGA Enabled, TA = +25°C)
100
AVDD = 3.3 V
PGA Enabled
TA = +85°C
Absolute Input Current (nA)
Absolute Input Current (nA)
20
1
C030
Figure 14. ABSOLUTE INPUT CURRENT vs
ABSOLUTE INPUT VOLTAGE
(PGA Enabled, TA = –40°C)
0
-10
-20
-30
-40
50
AIN0
AIN1
AIN2
AIN3
AVDD = 3.3 V
PGA Enabled
TA = +125°C
0
-50
-100
-150
-200
-50
-250
0.5
1
1.5
2
2.5
3
Absolute Input Voltage VAINx (V)
0.5
20
2
2.5
3
C033
Figure 17. ABSOLUTE INPUT CURRENT vs
ABSOLUTE INPUT VOLTAGE
(PGA Enabled, TA = +125°C)
40
Ta = -40ƒC
Ta = +25ƒC
Ta = +85ƒC
Ta = +125ƒC
1.5
Absolute Input Voltage VAINx (V)
AVDD = 3.3 V
PGA Enabled
AIN0:AIN1
Differential Input Current (nA)
40
1
C032
Figure 16. ABSOLUTE INPUT CURRENT vs
ABSOLUTE INPUT VOLTAGE
(PGA Enabled, TA = +85°C)
Differential Input Current (nA)
AVDD = 3.3 V
PGA Enabled
TA = +25°C
0
-20
-40
-60
Ta = -40ƒC
Ta = +25ƒC
Ta = +85ƒC
Ta = +125ƒC
20
AVDD = 3.3 V
PGA Enabled
AIN3:AIN2
0
-20
-40
-60
-2
-1.5
-1
-0.5
0
0.5
1
1.5
Differential Input Voltage VIN (V)
2
-2
Figure 18. DIFFERENTIAL INPUT CURRENT vs
DIFFERENTIAL INPUT VOLTAGE
(PGA Enabled, AIN0:AIN1)
-1.5
-1
-0.5
0
0.5
1
1.5
Differential Input Voltage VIN (V)
C038
2
C023
Figure 19. DIFFERENTIAL INPUT CURRENT vs
DIFFERENTIAL INPUT VOLTAGE
(PGA Enabled, AIN3:AIN2)
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TYPICAL CHARACTERISTICS (continued)
At TA = +25°C, AVDD = 3.3 V, and AVSS = 0 V using external VREF = 2.5 V, unless otherwise noted.
15
AIN0
AIN1
AIN2
AIN3
10
AVDD = 3.3 V
PGA Disabled
TA = -40°C
Absolute Input Current (nA)
Absolute Input Current (nA)
15
5
0
-5
-10
-15
AIN0
AIN1
AIN2
AIN3
10
5
0
-5
-10
-15
0.5
1
1.5
2
2.5
3
Absolute Input Voltage VAINx (V)
0.5
10
AIN0
AIN1
AIN2
AIN3
2
2.5
3
C035
Figure 21. ABSOLUTE INPUT CURRENT vs
ABSOLUTE INPUT VOLTAGE
(PGA Disabled, TA = +25°C)
100
AVDD = 3.3 V
PGA Disabled
TA = +85°C
0
-10
-20
-30
-40
50
AIN0
AIN1
AIN2
AIN3
AVDD = 3.3 V
PGA Disabled
TA = +125°C
0
-50
-100
-150
-200
-50
-250
0.5
1
1.5
2
2.5
3
Absolute Input Voltage VAINx (V)
0.5
20
2
2.5
3
C037
Figure 23. ABSOLUTE INPUT CURRENT vs
ABSOLUTE INPUT VOLTAGE
(PGA Disabled, TA = +125°C)
40
Ta = -40ƒC
Ta = +25ƒC
Ta = +85ƒC
Ta = +125ƒC
1.5
Absolute Input Voltage VAINx (V)
AVDD = 3.3 V
PGA Disabled
AIN0:AIN1
Differential Input Current (nA)
40
1
C036
Figure 22. ABSOLUTE INPUT CURRENT vs
ABSOLUTE INPUT VOLTAGE
(PGA Disabled, TA = +85°C)
Differential Input Current (nA)
1.5
Absolute Input Voltage VAINx (V)
Absolute Input Current (nA)
Absolute Input Current (nA)
20
1
C034
Figure 20. ABSOLUTE INPUT CURRENT vs
ABSOLUTE INPUT VOLTAGE
(PGA Disabled, TA = –40°C)
0
-20
-40
-60
Ta = -40ƒC
Ta = +25ƒC
Ta = +85ƒC
Ta = +125ƒC
20
AVDD = 3.3 V
PGA Disabled
AIN3:AIN2
0
-20
-40
-60
-2
-1.5
-1
-0.5
0
0.5
1
1.5
Differential Input Voltage VIN (V)
2
-2
-1.5
-1
-0.5
0
0.5
1
1.5
Differential Input Voltage VIN (V)
C040
Figure 24. DIFFERENTIAL INPUT CURRENT vs
DIFFERENTIAL INPUT VOLTAGE
(PGA Disabled, AIN0:AIN1)
12
AVDD = 3.3 V
PGA Disabled
TA = +25°C
2
C041
Figure 25. DIFFERENTIAL INPUT CURRENT vs
DIFFERENTIAL INPUT VOLTAGE
(PGA Disabled, AIN3:AIN2)
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TYPICAL CHARACTERISTICS (continued)
At TA = +25°C, AVDD = 3.3 V, and AVSS = 0 V using external VREF = 2.5 V, unless otherwise noted.
6
6
IDAC = 1000 µA
IDAC = 500 µA
IDAC = 500 µA
4
Absolute IDAC Error (%)
4
IDAC Error (%)
IDAC = 100 µA
2
0
-2
-4
2
0
-2
-4
-6
-6
0.5
0.6
0.7
0.8
0.9
1
Compliance Voltage (V)
-40
20
IDAC = 500 µA
0.5
IDAC = 100 µA
60
80
100
120
C005
600
IDAC = 1000 µA
0.75
40
Figure 27. IDAC ACCURACY vs TEMPERATURE
500
400
0.25
IAVDD (µA)
IDAC Matching Error (%)
0
Temperature (ƒC)
Figure 26. IDAC ACCURACY vs COMPLIANCE VOLTAGE
1
-20
C006
0
-0.25
300
200
-0.5
PGA = 64, 128
AVDD = 3.3 V
Internal Reference
Normal Mode
100
-0.75
-1
PGA = 1...16
PGA Disabled
0
-40
-20
0
20
40
60
80
100
120
Temperature (ƒC)
-40
-20
0
20
40
60
80
100
120
Temperature (ƒC)
C007
Figure 28. IDAC MATCHING vs TEMPERATURE
C011
Figure 29. IAVDD vs TEMPERATURE
(Normal Mode)
150
1000
125
800
IAVDD (µA)
IAVDD (µA)
100
600
400
75
50
PGA = 64, 128
200
AVDD = 3.3 V
Internal Reference
Turbo Mode
PGA Disabled
0
-40
-20
0
PGA = 64, 128
20
40
60
80
100
Temperature (ƒC)
AVDD = 3.3 V
Internal Reference
Duty-Cycle Mode
25
PGA = 1...16
PGA = 1...16
PGA Disabled
0
120
-40
C012
Figure 30. IAVDD vs TEMPERATURE
(Turbo Mode)
-20
0
20
40
60
80
100
Temperature (ƒC)
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C013
Figure 31. IAVDD vs TEMPERATURE
(Duty-Cycle Mode)
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TYPICAL CHARACTERISTICS (continued)
600
120
500
100
400
80
IDVDD (µA)
IAVDD (µA)
At TA = +25°C, AVDD = 3.3 V, and AVSS = 0 V using external VREF = 2.5 V, unless otherwise noted.
300
200
60
40
Turbo Mode
PGA = 64, 128
100
20
PGA = 1...16
Normal Mode
Internal Reference
Normal Mode
PGA Disabled
0
Duty-Cycle Mode
0
2.5
3
3.5
4
4.5
5
5.5
AVDD (V)
2.5
3
1
0.75
Temperature Error (%)
100
IDVDD (µA)
80
60
40
Turbo Mode
Normal Mode
0
20
40
60
5.5
C010
80
Mean
Mean - 61
0.5
0.25
0
-0.25
-0.5
-0.75
Duty-Cycle Mode
-20
5
Mean + 61
DVDD = 3.3 V
0
4.5
Figure 33. IDVDD vs DVDD
120
20
4
DVDD (V)
C004
Figure 32. IAVDD vs AVDD
-40
3.5
100
-1
120
Temperature (ƒC)
-40
-20
0
20
40
60
Temperature (ƒC)
C014
Figure 34. IDVDD vs TEMPERATURE
80
100
120
C015
Figure 35. INTERNAL TEMPERATURE SENSOR ACCURACY
vs TEMPERATURE
6
5
RON (
4
3
2
AVDD = 2.3 V
1
AVDD = 3.3 V
AVDD = 5.0 V
0
-40
-20
0
20
40
60
Temperature (ƒC)
80
100
120
C001
Figure 36. LOW-SIDE POWER SWITCH RON vs TEMPERATURE
14
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NOISE PERFORMANCE
As is the case with any delta-sigma (ΔΣ) ADC, noise performance can be optimized by adjusting the output data
rate. When reducing the data rate, the input-referred noise drops correspondingly because more samples of the
internal modulator are averaged to yield one conversion result. Increasing the gain also reduces the inputreferred noise, which is particularly useful when measuring low-level signals. Table 1 to Table 4 summarize the
device noise performance. Data are representative of typical noise performance at TA = +25°C with the internal
2.048-V reference. Data shown are the result of averaging readings from a single device over a time period of
approximately 0.75 seconds and are measured with the inputs internally shorted together.
Table 1 and Table 3 list the input-referred noise in units of μVRMS for the conditions shown. Note that µVPP values
are shown in parenthesis. Table 2 and Table 4 list the corresponding data in effective number of bits (ENOB)
calculated from μVRMS values using Equation 1. Note that noise-free bits calculated from peak-to-peak noise
values are shown in parenthesis.
The input-referred noise (Table 1 and Table 3) only changes marginally when using an external low-noise
reference, such as the REF5020. To calculate ENOB values when using a reference voltage other than 2.048 V,
use Equation 1 and Equation 2:
ENOB = ln (Full-Scale Range / Noise) / ln(2)
Full-Scale Range = 2 × VREF / PGA
(1)
(2)
Table 1. Noise in μVRMS (μVPP)
at AVDD = 3.3 V, AVSS = 0 V, and Internal Reference = 2.048 V
DATA
RATE
(SPS)
GAIN (PGA ENABLED)
1
20
45
90
2
4
8
16
32
64
128
3.71 (13.67)
1.54 (5.37)
1.15 (4.15)
0.80 (3.36)
0.35 (1.16)
0.23 (0.73)
0.10 (0.35)
0.09 (0.41)
7.36 (29.54)
2.93 (13.06)
1.71 (9.28)
0.88 (4.06)
0.50 (2.26)
0.29 (1.49)
0.19 (0.82)
0.12 (0.51)
10.55 (47.36)
4.50 (20.75)
2.43 (11.35)
1.51 (6.65)
0.65 (3.62)
0.42 (2.14)
0.27 (1.22)
0.18 (0.85)
175
11.90 (63.72)
6.45 (34.06)
3.26 (17.76)
1.82 (11.20)
1.01 (5.13)
0.57 (3.09)
0.34 (2.14)
0.26 (1.60)
330
19.19 (106.93)
9.38 (50.78)
4.25 (26.25)
2.68 (14.13)
1.45 (7.52)
0.79 (4.66)
0.50 (2.69)
0.34 (1.99)
600
24.78 (151.61)
13.35 (72.27)
6.68 (39.43)
3.66 (19.26)
2.10 (12.77)
1.14 (6.87)
0.70 (4.76)
0.55 (3.34)
1000
37.53 (227.29)
18.87 (122.68)
9.53 (58.53)
5.37 (31.52)
2.95 (18.08)
1.65 (10.71)
1.03 (6.52)
0.70 (4.01)
2000
36.23 (265.14)
18.24 (127.32)
9.24 (65.43)
5.49 (37.02)
2.89 (18.89)
1.77 (12.00)
1.13 (7.60)
0.82 (5.81)
Table 2. ENOB from RMS Noise (Peak-to-Peak Noise)
at AVDD = 3.3 V, AVSS = 0 V, and Internal Reference = 2.048 V
DATA
RATE
(SPS)
GAIN (PGA ENABLED)
1
2
4
8
16
32
64
128
20
20.08 (18.19)
20.34 (18.54)
19.76 (17.91)
19.28 (17.22)
19.48 (17.75)
19.10 (17.42)
19.33 (17.49)
18.49 (16.26)
45
19.09 (17.08)
19.42 (17.26)
19.19 (16.75)
19.15 (16.94)
18.95 (16.79)
18.74 (16.39)
18.38 (16.25)
18.00 (15.49)
90
18.57 (16.40)
18.80 (16.59)
18.68 (16.46)
18.37 (16.23)
18.60 (16.11)
18.20 (15.87)
17.87 (15.67)
17.44 (15.20)
175
18.39 (15.97)
18.28 (15.88)
18.26 (15.82)
18.10 (15.48)
17.96 (15.61)
17.78 (15.34)
17.53 (14.87)
16.91 (14.29)
330
17.70 (15.23)
17.74 (15.30)
17.88 (15.25)
17.54 (15.15)
17.43 (15.05)
17.30 (14.74)
16.96 (14.54)
16.50 (13.97)
600
17.33 (14.72)
17.23 (14.79)
17.23 (14.66)
17.09 (14.70)
16.89 (14.29)
16.77 (14.18)
16.48 (13.72)
15.83 (13.23)
1000
16.74 (14.14)
16.73 (14.03)
16.71 (14.09)
16.54 (13.99)
16.41 (13.79)
16.25 (13.54)
15.92 (13.26)
15.49 (12.96)
2000
16.79 (13.92)
16.78 (13.97)
16.76 (13.93)
16.51 (13.76)
16.44 (13.73)
16.14 (13.38)
15.79 (13.04)
15.25 (12.43)
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Table 3. Noise in μVRMS (μVPP) with PGA Disabled
at AVDD = 3.3 V, AVSS = 0 V, and Internal Reference = 2.048 V
GAIN (PGA DISABLED)
DATA RATE
(SPS)
1
2
4
20
3.89 (13.43)
1.85 (6.84)
1.26 (3.91)
45
6.97 (31.98)
2.94 (12.94)
1.41 (5.62)
90
8.50 (42.48)
4.49 (18.92)
2.07 (9.95)
175
12.99 (65.92)
6.24 (35.40)
3.04 (18.92)
330
18.18 (94.24)
8.12 (50.17)
4.71 (28.75)
600
25.29 (138.67)
12.77 (78.13)
6.27 (39.79)
1000
38.04 (260.50)
18.40 (120.97)
9.48 (63.72)
2000
36.11 (250.98)
17.30 (131.35)
8.77 (68.18)
Table 4. ENOB from RMS Noise (Peak-to-Peak Noise) with PGA Disabled
at AVDD = 3.3 V, AVSS = 0 V, and Internal Reference = 2.048 V
16
GAIN (PGA DISABLED)
DATA RATE
(SPS)
1
2
4
20
20.01 (18.22)
20.08 (18.19)
19.63 (18.00)
45
19.61 (16.97)
19.41 (17.27)
19.47 (17.48)
90
18.88 (16.56)
18.80 (16.72)
18.91 (16.65)
175
18.27 (15.92)
18.32 (15.82)
18.36 (15.72)
330
17.78 (15.41)
17.94 (15.32)
17.73 (15.12)
600
17.31 (14.85)
17.29 (14.68)
17.32 (14.65)
1000
16.72 (13.94)
16.76 (14.05)
16.72 (13.97)
2000
16.79 (13.99)
16.85 (13.93)
16.83 (13.87)
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OVERVIEW
The ADS1220 is a small, low-power, 24-bit, highly-integrated, ΔΣ analog-to-digital converter (ADC). The device is
easy to configure and design into a wide variety of applications and allows precise measurements to be obtained
with little effort.
In addition to the ΔΣ ADC core and single-cycle settling digital filter, the ADS1220 offers a low-noise, high input
impedance, programmable gain amplifier (PGA), an internal voltage reference, a clock oscillator, and an SPIcompatible interface. The device also integrates a highly linear and accurate temperature sensor as well as two
matched programmable current sources (IDACs) for sensor excitation. All of these features are intended to
reduce the required external circuitry in typical sensor applications and improve overall system performance. An
additional low-side power switch eases the design of low-power bridge sensor applications. Figure 37 shows the
ADS1220 functional block diagram.
REFP0
AVDD
REFN0
DVDD
10 A to
1.5 mA
Internal
Reference
AIN0/REFP1
AIN1
Device
24-bit
ûADC
Digital Filter
and
SPI
Interface
Low Drift
Oscillator
Precision
Temp Sensor
CLK
DGND
AINP
Mux
AIN2
Reference
Mux
PGA
AINN
AIN3/REFN1
AVSS
CS
SCLK
DIN
DOUT/DRDY
DRDY
Figure 37. Functional Block Diagram
The ADS1220 ADC measures a differential signal, VIN, which is the difference of AINP and AINN. The converter
core consists of a differential, switched-capacitor ΔΣ modulator followed by a digital filter. The digital filter
receives a high-speed bitstream from the modulator and outputs a code proportional to the input voltage. This
architecture results in a very strong attenuation in any common-mode signals.
The device has two available conversion modes: single-shot and continuous conversion mode. In single-shot
mode, the ADC performs one conversion of the input signal upon request and stores the value to an internal data
buffer. The device then enters a low-power state to save power. Single-shot mode is intended to provide
significant power savings in systems that require only periodic conversions or when there are long idle periods
between conversions. In continuous conversion mode, the ADC automatically begins a conversion of the input
signal as soon as the previous conversion is completed. New data are available at the programmed data rate.
Data can be read at any time without concern of data corruption and always reflect the most recently completed
conversion.
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MULTIPLEXER
The device contains a very flexible input multiplexer, as shown in Figure 38. Either four single-ended signals, two
differential signals, or a combination of two single-ended signals and one differential signal can be measured.
The multiplexer is configured by four bits (MUX[3:0]) in the configuration register. When single-ended signals are
measured, the ADC negative input is internally connected to AVSS by a switch within the multiplexer. For
system-monitoring purposes, the analog supply (AVDD – AVSS) / 4 or the currently-selected external reference
(REFPx – REFNx) / 4 can be selected as inputs to the ADC. The multiplexer also offers the possibility to route
any of the two programmable current sources to any analog input (AINx) or to any dedicated reference pin
(REFP0, REFN0).
System Monitors
(REFPx ± REFNx)/4
(AVDD ± AVSS)/4
AVDD
AVDD
IDAC1
AVDD
AVSS
AVDD
AVSS
IDAC2
(AVDD + AVSS)/2
AIN0/REFP1
AVDD
AIN1
Burnout Current Source (10 µA)
AVDD
AVSS
AVDD
AVSS
AIN2
AINP
PGA
To ADC
AINN
AIN3/REFN1
AVDD
AVSS
AVDD
AVSS
Burnout Current Source (10 µA)
REFP0
AVSS
AVSS
REFN0
Figure 38. Analog Input Multiplexer
Electrostatic discharge (ESD) diodes to AVDD and AVSS protect the inputs. To prevent the ESD diodes from
turning on, the absolute voltage on any input must stay within the range of Equation 3:
AVSS – 0.3 V < AINx < AVDD + 0.3 V
(3)
If the voltages on the input pins have any potential to violate these conditions, external Schottky clamp diodes or
series resistors may be required to limit the input current to safe values (see the Absolute Maximum Ratings
table). Although the analog inputs can support signals marginally above supply, under no circumstances should
any analog or digital input or output be driven to greater than 5.5 V with respect to the GND pin.
Overdriving an unused input on the device may affect conversions taking place on other input pins. If any
overdrive on unused inputs is possible, TI recommends clamping the signal with external Schottky diodes.
18
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LOW-NOISE PGA
The device features a low-noise, low-drift, high input impedance, programmable gain amplifier (PGA). The PGA
can be set to gains of 1, 2, 4, 8, 16, 32, 64, or 128. Three bits (GAIN[2:0]) in the configuration register are used
to configure the gain. A simplified diagram of the PGA is shown in Figure 39. The PGA consists of two chopperstabilized amplifiers (A1 and A2) and a resistor feedback network that sets the PGA gain. The PGA input is
equipped with an electromagnetic interference (EMI) filter.
200 W
AINP
25 pF
A1
R
ADC
C
R
A2
200 W
AINN
25 pF
Figure 39. Simplified Diagram of the PGA
The differential full-scale (FS) input voltage range of the PGA is defined by the gain setting and the reference
voltage used, as shown in Equation 4:
FS = ±VREF / PGA
(4)
Table 5 shows the corresponding full-scale ranges when using the internal 2.048-V reference.
Table 5. PGA Full-Scale Range
GAIN SETTING
FS
1
±2.048 V
2
±1.024 V
4
±0.512 V
8
±0.256 V
16
±0.128 V
32
±0.064 V
64
±0.032 V
128
±0.016 V
Note that as with any PGA, the input voltage must remain within a specified common-mode input voltage range.
The common-mode input voltage (VCM) must stay within the minimum and maximum limits given by Equation 5
and Equation 6:
VIN ´ PGA
AVDD - AVSS
VCM (MIN) ³ AVSS +
VCM (MIN) ³ AVSS + 0.2 V +
and
4
2
(5)
VIN ´ PGA
VCM (MAX) £ AVDD - 0.2 V 2
(6)
where:
•
•
•
VCM = (AINP + AINN) / 2,
PGA = PGA gain, and
VIN = the maximum differential input voltage (AINP – AINN) in the application, which is limited to
[(AINP – AINN) ≤ ±VREF / PGA].
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Figure 40 and Figure 41 show a graphical representation of the common-mode voltage limits for AVDD = 3.3 V,
PGA = 1 and PGA = 16, respectively.
3.3
3.3
AVDD = 3.3 V
PGA = 1
AVDD = 3.3 V
PGA = 16
2.75
2.2
VCM Range (V)
VCM Range (V)
2.75
1.65
3.3 V / 4
1.1
0.55
2.2
1.65
3.3 V / 4
1.1
0.55
0
0
0
0.5
1
1.5
2
2.5
VIN (V)
3
0
Figure 40. Common-Mode Voltage Limits
(AVDD = 3.3 V, PGA = 1)
0.03
0.06
0.09
0.12
0.15
0.18
VIN (V)
C009
C008
Figure 41. Common-Mode Voltage Limits
(AVDD = 3.3 V, PGA = 16)
The following paragraphs explain how to apply Equation 5 and Equation 6 to a hypothetical application. The
setup for this example is AVDD = 3.3 V, AVSS = 0 V, and PGA = 16, using an external reference VREF= 2.5 V.
The maximum differential input voltage VIN = (AINP – AINN) that can be applied is then limited to the full-scale
range of FS = ±2.5 V / 16 = ±0.156 V. Equation 5 and Equation 6 then yield an allowed VCM range of 1.45 V ≤
VCM ≤ 1.85 V.
However, the sensor signal that is connected to the inputs in this example application does not make use of the
entire full-scale range but is limited to VIN = ±0.1 V. Accordingly, this reduced input signal relaxes the VCM
restriction to 1.0 V ≤ VCM ≤ 2.3 V.
In the case of a fully-differential sensor signal, each input (AINP, AINN) can swing up to ±50 mV around the
center voltage (AINP + AINN) / 2, which must remain between the common-mode voltage limits of 1.0 V and
2.3 V. The output of a symmetrical wheatstone bridge is an example of a fully-differential signal.
In contrast, the signal of an RTD is of a pseudo-differential nature (depending on the circuit implementation),
where the negative input is held at a constant voltage other than 0 V. When a pseudo-differential signal must be
measured, the negative input must be biased at a voltage between 1.0 V and 2.25 V. The positive input can then
swing up to 100 mV above the negative input.
Figure 42 and Figure 43 illustrate both fully-differential and pseudo-differential cases for this specific example,
respectively.
AINP
AINP
100 mV
100 mV
1.0 V
1.0 V
AINN
AINN
0V
0V
Figure 42. Fully-Differential Input Signal
20
Figure 43. Pseudo-Differential Input Signal
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BYPASSING THE PGA
At gains of 1, 2, and 4, the ADS1220 can be configured to disable and bypass the low-noise PGA. Disabling the
PGA lowers the overall power consumption and also removes the restrictions of Equation 5 and Equation 6 for
the common-mode input voltage range, VCM. The usable absolute and common-mode input voltage range is
(AVSS – 0.1 V ≤ VCM ≤ AVDD + 0.1 V) when the PGA is disabled. In order to measure single-ended signals that
are referenced to AVSS (VINP = VIN, VINN = AVSS), the PGA must be turned off.
NOTE
When measuring single-ended inputs, the negative range of the output codes is not used.
These codes are for measuring negative differential signals, such as (AINP – AINN) < 0 V.
Consequently, one bit of resolution is lost because only half of the full-scale range is used.
When the PGA is disabled by setting the PGA_BYPASS bit in the configuration register, the device uses a
buffered switched-capacitor stage to obtain gains 1, 2, and 4. An internal buffer in front of the switched-capacitor
stage ensures that the impact on the input loading as a result of the capacitors charging and discharging is
minimal. Refer to Figure 20 to Figure 25 for the typical values of absolute (current flowing into or out of each
input) and differential (difference in absolute current between positive and negative input) input currents when the
PGA is disabled.
For signal sources with high output impedance, external buffering may still be necessary. Note that active buffers
introduce noise and also introduce offset and gain errors. All of these factors should be considered in highaccuracy applications.
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MODULATOR
A ΔΣ modulator is used in the ADS1220 to convert the analog input voltage into a pulse code modulated (PCM)
data stream. The modulator runs at a modulator clock frequency of fMOD = fCLK / 16 in normal and duty-cycle
mode and fMOD = fCLK / 8 in turbo mode, where fCLK is either provided by the internal oscillator or the external
clock source. Table 6 shows the modulator frequency for each mode using either the internal oscillator or an
external clock of 4.096 MHz.
Table 6. Modulator Clock Frequency for Different
Operating Modes using the Internal Oscillator
OPERATING MODE
fMOD
Duty-cycle mode
256 kHz
Normal mode
256 kHz
Turbo mode
512 kHz
DIGITAL FILTER
0
0
-40
-40
Magnitude (dB)
Magnitude (dB)
The device uses a linear-phase finite impulse response (FIR) digital filter that performs both filtering and
decimation of the digital data stream coming from the modulator. The digital filter is automatically adjusted for the
different data rates and always settles within a single cycle. Only at data rates of 5 SPS and 20 SPS can the filter
be configured to reject 50-Hz or 60-Hz line frequencies or to simultaneously reject 50 Hz and 60 Hz. Two bits
(50/60[1:0]) in the configuration register are used to configure the filter accordingly. The frequency responses of
the digital filter are shown in Figure 44 to Figure 57 for different output data rates using the internal oscillator.
-80
-120
-160
-200
0
20
40
60
80 100 120 140
Frequency (Hz)
160
180
200
46
47
48
C006
Figure 44. Filter Response
(Data Rate = 20 SPS, 50-Hz Rejection Only)
49
50
51
Frequency (Hz)
52
53
54
C004
Figure 45. Detailed View of Filter Response
(Data Rate = 20 SPS, 50-Hz Rejection Only)
0
0
-40
-40
Magnitude (dB)
Magnitude (dB)
-120
-160
-200
-80
-120
-160
-80
-120
-160
-200
-200
0
20
40
60
80 100 120 140
Frequency (Hz)
160
180
200
56
C010
Figure 46. Filter Response
(Data Rate = 20 SPS, 60-Hz Rejection Only)
22
-80
57
58
59
60
61
Frequency (Hz)
62
63
64
C008
Figure 47. Detailed View of Filter Response
(Data Rate = 20 SPS, 60-Hz Rejection Only)
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0
0
-40
-40
Magnitude (dB)
Magnitude (dB)
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-80
-120
-160
-120
-160
-200
-200
0
20
40
60
80 100 120 140
Frequency (Hz)
160
180
200
46
50
52
0
-20
-20
Magnitude (dB)
0
-40
54
56
58
Frequency (Hz)
60
62
64
C001
Figure 49. Detailed View of Filter Response
(Data Rate = 20 SPS, Simultaneous 50- and 60-Hz
Rejection)
-60
-40
-60
-80
-80
0
20
40
60
80
100
120
140
160
Frequency (Hz)
180
200
0
20
40
60
-20
-20
Magnitude (dB)
0
-60
100
120
140
160
180
200
C015
Figure 51. Filter Response
(Data Rate = 45 SPS)
0
-40
80
Frequency (Hz)
C016
Figure 50. Filter Response
(Data Rate = 20 SPS, No 50- or 60-Hz Rejection)
Magnitude (dB)
48
C002
Figure 48. Filter Response
(Data Rate = 20 SPS, Simultaneous 50- and 60-Hz
Rejection)
Magnitude (dB)
-80
-40
-60
-80
-80
0
100
200
300
400
500
600
700
Frequency (Hz)
800
900 1000
0
C014
Figure 52. Filter Response
(Data Rate = 90 SPS)
100
200
300
400
500
600
700
800
900 1000
Frequency (Hz)
C013
Figure 53. Filter Response
(Data Rate = 175 SPS)
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0
0
-20
-20
Magnitude (dB)
Magnitude (dB)
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-40
-60
-40
-60
-80
-80
0
200
400
600
800 1000 1200 1400 1600 1800 2000
Frequency (Hz)
0
1500
2000
2500
3000
3500
4000
C008
Figure 55. Filter Response
(Data Rate = 600 SPS)
0
0
-20
-20
Magnitude (dB)
Magnitude (dB)
1000
Frequency (Hz)
Figure 54. Filter Response
(Data Rate = 330 SPS)
-40
-60
-40
-60
-80
-80
0
1
2
3
4
5
6
7
Frequency (kHz)
8
9
10
0
C010
Figure 56. Filter Response
(Data Rate = 1 kSPS)
24
500
C012
1
2
3
4
5
6
7
Frequency (kHz)
8
9
10
C009
Figure 57. Filter Response
(Data Rate = 2 kSPS)
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OUTPUT DATA RATE
Table 7, Table 8, and Table 9 show the actual conversion times for each data rate setting. The values provided
are in terms of tCLK cycles using an external clock with a clock frequency of fCLK = 4.096 MHz.
Single-shot mode data rates are timed from the last SCLK falling edge of the START/SYNC command to the
DRDY falling edge and rounded to the next tCLK. In case the internal oscillator is used, an additional oscillator
wake-up time of up to 50 µs (normal mode, duty-cycle mode) or 25 µs (turbo mode), respectively, for each
conversion in single-shot mode must be added. The internal oscillator starts to power up at the first SCLK rising
edge. Depending on the SCLK frequency, the oscillator cannot be ensured to be fully powered up at the end of
the START/SYNC command. A conversion therefore only starts after the internal oscillator is fully powered up.
Continuous conversion data rates are timed from one DRDY falling edge the next DRDY falling edge. The first
conversion starts 210 × tCLK (normal mode, duty-cycle mode) or 114 × tCLK (turbo mode), respectively, after the
last SCLK falling edge of the START/SYNC command.
Table 7. Normal Mode
ACTUAL CONVERSION TIME (tCLK)
NOMINAL DATA RATE
(SPS)
–3-dB BANDWIDTH
(Hz)
SINGLE-SHOT MODE
CONTINUOUS CONVERSION MODE
20
13.1
204850
204768
45
20.0
91218
91120
90
39.6
46226
46128
175
77.8
23762
23664
330
150.1
12562
12464
600
279.0
6994
6896
1000
483.8
4242
4144
Table 8. Duty-Cycle Mode
ACTUAL CONVERSION TIME (tCLK)
NOMINAL DATA RATE
(SPS)
–3-dB BANDWIDTH
(Hz)
SINGLE-SHOT MODE
CONTINUOUS CONVERSION MODE
5
13.1
204850
823120
11.25
20.0
91218
364560
22.5
39.6
46226
184592
44
77.8
23762
94736
82.5
150.1
12562
49936
150
279.0
6994
27664
250
483.8
4242
16656
Table 9. Turbo Mode
ACTUAL CONVERSION TIME (tCLK)
NOMINAL DATA RATE
(SPS)
–3-dB BANDWIDTH
(Hz)
SINGLE-SHOT MODE
CONTINUOUS CONVERSION MODE
40
26.2
102434
102384
90
39.9
45618
45560
180
79.2
23122
23064
350
155.6
11890
11832
660
300.3
6290
6232
1200
558.1
3506
3448
2000
967.6
2130
2072
Note that even though the data rate at the 20-SPS setting is not exactly 20 SPS, this discrepancy does not effect
the 50-Hz or 60-Hz rejection. To achieve the specified 50-Hz and 60-Hz rejection, the external clock frequency
must only be ensured to be exactly 4.096 MHz.
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ALIASING
As with any sampled system, aliasing can occur if proper antialias filtering is not in place. Aliasing describes the
effect that frequency components in the input signal that are higher than half the sampling frequency of the ADC
(also known as the Nyquist frequency) are folded back and show up in the actual frequency band of interest
below half the sampling frequency. Note that inside a ΔΣ ADC, the input signal is sampled at the modulator
frequency fMOD and not at the output data rate. The filter response of the digital filter repeats at multiples of the
sampling frequency (fMOD), as shown in Figure 58. Signals or noise up to a frequency where the filter response
repeats are attenuated by the digital filter. However, any frequency components present in the input signal
around the modulator frequency or multiples thereof are not attenuated and thus alias back into the band of
interest, unless attenuated by an external analog filter. Some signals are inherently bandlimited; for example, the
output of a thermocouple has a limited rate of change. Nevertheless, these signals can contain noise and
interference components at higher frequencies, which can fold back into the frequency band of interest. A simple
RC filter is (in most cases) sufficient to reject these high-frequency components. When designing an input filter
circuit, be sure to take into account the interaction between the filter network and the input impedance of the
ADS1220.
Magnitude
Sensor
Signal
Unwanted
Signals
Unwanted
Signals
Output
Data Rate
fMOD/2
fMOD
Frequency
fMOD
Frequency
fMOD
Frequency
Magnitude
Digital Filter
Aliasing of
Unwanted Signals
Output
Data Rate
fMOD/2
Magnitude
Anti-Aliasing Filter
Roll-Off
Output
Data Rate
fMOD/2
Figure 58. Effect of Aliasing
26
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VOLTAGE REFERENCE
The ADS1220 offers an integrated low-drift, 2.048-V reference. For applications that require a different reference
voltage value or a ratiometric measurement approach, the device offers two differential reference inputs (REFPx
and REFNx). In addition, the analog supply (AVDD) can be used as a reference. The differential reference inputs
allow freedom in the reference common-mode voltage. REFP0 and REFN0 are dedicated reference inputs
whereas REFP1 and REFN1 are shared with inputs AIN0 and AIN3, respectively. The reference inputs are
internally buffered to increase input impedance. Therefore, additional reference buffers are usually not required
when using an external reference and the reference inputs do not load any external circuitry when used in
ratiometric applications. The reference source is selected by two bits (VREF[1:0]) in the configuration register. By
default, the internal reference is selected.
CLOCK SOURCE
The device system clock can either be provided by the internal low-drift oscillator or by an external clock source
on the CLK input. Connect the CLK pin to DGND before power-up or reset to activate the internal oscillator.
Connecting an external clock to the CLK pin at any time deactivates the internal oscillator after two rising edges
on the CLK pin are detected. The device then operates on the external clock. After the ADS1220 switches to the
external clock, the device cannot be switched back to the internal oscillator without cycling the power supplies or
sending a RESET command.
EXCITATION CURRENT SOURCES
The ADS1220 provides two matched programmable excitation current sources (IDACs) for RTD applications.
The output current of the current sources can be programmed to 10 μA, 50 μA, 100 μA, 250 μA, 500 μA,
1000 μA, or 1500 μA using the respective bits (IDAC[2:0]) in the configuration register. Each current source can
be connected to any of the analog inputs (AINx) as well as to any of the dedicated reference inputs (REFP0 and
REFN0). Both current sources can also be connected to the same pin. Routing of the IDACs is configured by bits
(I1MUX[2:0], I2MUX[2:0]) in the configuration register. Care should be taken not to exceed the compliance
voltage of the IDACs. In other words, the voltage on the pin where the IDAC is routed to should be limited to
≤ (AVDD – 0.9 V), otherwise the specified accuracy of the IDAC current is not met. For three-wire RTD
applications, the matched current sources can be used to cancel the errors caused by sensor lead resistance.
The IDACs require up to 200 µs to start up after the IDAC current is programmed to the respective value using
bits IDAC[2:0]. If configuration register 2 and 3 are not written during the same WREG command, TI
recommends to first set the IDAC current to the respective value using bits IDAC[2:0] and thereafter select the
routing for each IDAC (I1MUX[2:0], I2MUX[2:0]).
In single-shot mode, the IDACs remain active between any two conversions if the IDAC[2:0] bits are set to a
value other than 000. However, the IDACs are powered down whenever the POWERDOWN command is issued.
SENSOR DETECTION
To help detect a possible sensor malfunction, the device provides internal 10-µA, burn-out current sources.
When enabled by setting the respective bit (BCS) in the configuration register, one current source sources
current to the positive analog input (AINP) currently selected and the other current source sinks current form the
selected negative analog input (AINN).
In case of an open circuit in the sensor, these burn-out current sources pull the positive input towards AVDD and
the negative input towards AVSS, resulting in a full-scale reading. A full-scale reading may also indicate that the
sensor is overloaded or that the reference voltage is absent. A near-zero reading may indicate a shorted sensor.
However, because the absolute value of the burn-out current sources typically varies by ±10% and the internal
multiplexer adds a small series resistance, distinguishing a shorted sensor condition from a normal reading can
be difficult, especially if an RC filter is used at the inputs. In other words, even if the sensor fails short, the
voltage drop across the external filter resistance and the residual resistance of the multiplexer causes the output
to read a value higher than zero.
If a higher precision current source is required for sensor short detection, TI recommends using the excitation
current sources (IDACs). Keep in mind that ADC readings of a functional sensor may be corrupted when the
burn-out current sources are enabled.
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LOW-SIDE POWER SWITCH
A low-side power switch with low on-resistance connected between the analog input AIN3/REFN1 and AVSS is
integrated in the ADS1220 as well. This power switch can be used to reduce system power consumption in
bridge sensor applications by powering down the bridge circuit between conversions. When the respective bit
(PSW) in the configuration register is set, the switch automatically closes during conversions and opens when
the device is in power-down mode. By default, the switch is always open.
SYSTEM MONITOR
The device provides some means for monitoring the AVDD analog power supply and the external voltage
reference. To select any monitoring voltages, the internal multiplexer (MUX[3:0]) must be configured accordingly
in the configuration register. Note that the system monitor function only provides a coarse result and is not meant
to be a precision measurement.
When measuring the analog power supply (MUX[3:0] = 1101), the resulting conversion is approximately (AVDD –
AVSS) / 4. The device uses the internal reference for the measurement regardless of what reference source is
selected in the configuration register (VREF[1:0]).
When monitoring one of the two possible external reference voltage sources (MUX[3:0] = 1100), the result is
approximately (REFPx – REFNx) / 4. REFPx and REFNx denote the external reference input pair selected in the
configuration register (VREF[1:0]). The ADS1220 automatically uses the internal reference for the measurement.
OFFSET CALIBRATION
The internal multiplexer offers the option to short both PGA inputs (AINP and AINN) to mid-supply (AVDD +
AVSS) / 2. This option can be used to calibrate the device offset voltage by storing the result of the shorted input
voltage reading in a microcontroller and consequently subtracting the result from each following reading. TI
recommends taking multiple readings with the inputs shorted and averaging the result to reduce the effect of
noise.
POWER SUPPLIES
The device has two power supplies: analog (AVDD, AVSS) and digital (DVDD, DGND). The analog power supply
can be bipolar (for example, AVDD = +2.5 V, AVSS = –2.5 V) or single supply (for example, AVDD = +3.3 V,
AVSS = 0 V) and is independent of the digital power supply. The digital supply range sets the digital I/O levels.
The power supplies can be sequenced in any order but in no case should any of the analog or digital inputs
exceed the respective analog or digital power-supply voltage limits.
28
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TEMPERATURE SENSOR
The temperature measurement mode of the device is configured as a 14-bit result when enabled by the TS bit in
the configuration register. Data are output starting with the most significant byte (MSB). When reading the three
data bytes, the first 14 bits are used to indicate the temperature measurement result. The last 10 bits are random
data and must be ignored. That is, the 14-bit temperature result is left-justified within the 24-bit conversion result.
One 14-bit LSB equals 0.03125°C. Negative numbers are represented in binary twos complement format.
Table 10. 14-Bit Temperature Data Format
TEMPERATURE (°C)
DIGITAL OUTPUT (BINARY)
HEX
128
01 0000 0000 0000
1000
127.96875
00 1111 1111 1111
0FFF
100
00 1100 1000 0000
0C80
80
00 1010 0000 0000
0A00
75
00 1001 0110 0000
0960
50
00 0110 0100 0000
0640
25
00 0011 0010 0000
0320
0.25
00 0000 0000 1000
0008
0
00 0000 0000 0000
0000
–0.25
11 1111 1111 1000
3FF8
–25
11 1100 1110 0000
3CE0
–55
11 1001 0010 0000
3920
Converting from Temperature to Digital Codes
For Positive Temperatures (for example, +50°C):
Twos complement is not performed on positive numbers. Therefore, simply convert the number to binary code in
a 14-bit, left-justified format with the MSB = 0 to denote the positive sign.
Example: (+50°C) / (0.03125°C per count) = 1600 = 0640h = 00 0110 0100 0000
For Negative Temperatures (for example, –25°C):
Generate the twos complement of a negative number by complementing the absolute binary number and adding
'1'. Then, denote the negative sign with the MSB = 1.
Example: (|–25°C|) / (0.03125°C per count) = 800 = 0320h = 00 0011 0010 0000
Twos complement format: 11 1100 1101 1111 + 1 = 11 1100 1110 0000
Converting from Digital Codes to Temperature
To convert from digital codes to temperature, first check whether the MSB is a '0' or a '1'. If the MSB is a '0',
simply multiply the decimal code by 0.03125°C to obtain the result. If the MSB = 1, subtract '1' from the result
and complement all bits. Then, multiply the result by –0.03125°C.
Example: The ADS1220 reads back 0960h: 0960h has an MSB = 0.
(0960h) × (0.03125°C) = (2400) × (0.03125°C) = +75°C
Example: The ADS1220 reads back 3CE0h: 3CE0h has an MSB = 1.
Complement the result: 3CE0h → 0320h (0320h) × (–0.03125°C) = (800) × (–0.03125°C) = –25°C
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RESET AND POWER-UP
When the device powers up, a reset is performed. As part of the reset process, the device sets all bits in the
configuration registers to the respective default settings. By default, the device is set to single-shot mode. After
power-up, the device performs a single conversion using the default register settings and then enters a lowpower state. The power-up behavior is intended to prevent systems with tight power-supply requirements from
encountering a current surge during power-up.
CONVERSION MODES
The device can be operated in one of two conversion modes that can be selected by the CM bit in the
configuration register. These conversion modes are single-shot or continuous conversion mode.
Single-Shot Mode
In single-shot mode, the device only performs a conversion when a START/SYNC command is issued. The
device consequently performs one single conversion and returns to a low-power state afterwards. The internal
oscillator and all analog circuitry (except for the excitation current sources) are turned off while the device waits
in this low-power state until the next conversion is started. In addition, every write access to any configuration
register also starts a new conversion. Writing to any configuration register while a conversion is ongoing
functions as a new START/SYNC command that stops the current conversion and restarts a single new
conversion. Because the ADS1220 digital filter settles within a single cycle, each conversion is fully settled
assuming the analog input signal is settled to its final value before the conversion starts.
Continuous Conversion Mode
In continuous conversion mode, the device continuously performs conversions. When a conversion completes,
the device places the result in the output buffer and immediately begins another conversion.
In order to start continuous conversion mode, the CM bit must be set to '1' followed by a START/SYNC
command. The first conversion starts 210 × tCLK (normal mode, duty-cycle mode) or 114 × tCLK (turbo mode),
respectively, after the last SCLK falling edge of the START/SYNC command. Writing to any configuration register
while the START/SYNC command is not issued starts a single conversion, whereas a write access to the
configuration register during an ongoing conversion restarts the current conversion. TI recommends to always
send a START/SYNC command immediately after the CM bit is set to '1'.
OPERATING MODES
In addition to the different conversion modes, the ADS1220 can also be operated in different operating modes
that can be selected to trade off power consumption, noise performance, and output data rate.
Normal Mode
Normal mode is the default mode the device operates in. In this mode, the internal modulator of the ΔΣ ADC
runs at a modulator clock frequency of fMOD = fCLK / 16, where the system clock (fCLK) is either provided by the
internal oscillator or the external clock source. The modulator frequency using the internal oscillator is 256 kHz.
Normal mode offers output data rate options ranging from 20 SPS to 1 kSPS with the internal oscillator. The data
rate is selected by bits (DR[2:0]) in the configuration register. In case an external clock source with a clock
frequency other than 4.096 MHz is used, the data rates scale accordingly. For example, using an external clock
with fCLK = 2.048 MHz yields data rates ranging from 10 SPS to 500 SPS.
Duty-Cycle Mode
The noise performance of a ΔΣ ADC generally improves when lowering the output data rate because more
samples of the internal modulator can be averaged to yield one conversion result. In applications where power
consumption is critical, the improved noise performance at low data rates may not be required. For these
applications, the device supports an internal duty cycling that can yield significant power savings by periodically
entering a low-power state between conversions. In principle, the device runs in normal mode with a duty cycle
of 25%. This functionality means the device performs one conversion in the same manner as when running in
normal mode but then automatically enters a low power-state for three consecutive conversion cycles. The noise
performance in duty-cycle mode is therefore comparable to the noise performance in normal mode at four times
the data rate. Data rates in duty-cycle mode range from 5 SPS to 250 SPS with the internal oscillator.
30
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Turbo Mode
Applications that require higher data rates up to 2 kSPS can operate the device in turbo mode. In this mode, the
internal modulator runs at a higher frequency of fMOD = fCLK / 8. fMOD = 512 kHz when the internal oscillator or an
external 4.096-MHz clock is used. Note that the device power consumption does increase because the
modulator runs at a higher frequency.
Power-Down Mode
When the POWERDOWN command is issued, the device enters power-down mode after completing the current
conversion. In this mode, all analog circuitry is powered down and the device typically only uses 400 nA of
current. During this time, the device holds the configuration register settings and responds to commands, but
does not perform any data conversion.
Issuing a START/SYNC command wakes up the device and either starts a single conversion or starts continuous
conversion mode depending on the conversion mode selected by the CM bit. Writing to any configuration register
bit wakes up the device as well, but only starts a single conversion regardless of what conversion mode (CM) the
device is set to. TI recommends to always send a START/SYNC command immediately after writing to any of the
configuration registers.
SERIAL INTERFACE
The SPI-compatible serial interface of the device is used to read conversion data, read and write the device
configuration registers, and control device operation. The interface consists of five control lines (CS, SCLK, DIN,
DOUT/DRDY, and DRDY) but can be used with four or even three control signals (SCLK, DIN, and
DOUT/DRDY) as well. In the latter case, CS may be tied low if the serial bus is not shared with any other device.
The dedicated data-ready signal (DRDY) can be configured to be shared with DOUT/DRDY.
CHIP SELECT (CS)
Chip select (CS) is an active-low input that selects the device for SPI communication. This feature is useful when
multiple devices share the same serial bus. CS must remain low for the duration of the serial communication.
When CS is taken high, the serial interface is reset, SCLK is ignored, and DOUT/DRDY enters a high-impedance
state; as such, DOUT/DRDY cannot indicate when data are ready. In situations where multiple devices are
present on the bus, the dedicated DRDY pin can provide an uninterrupted monitor of the result status. New data
can be transferred at anytime without concern of data corruption. When a transmission starts, the current result is
loaded into the output shift register and does not change until the communication is complete. This
implementation avoids any possibility of data corruption. If the serial bus is not shared with another peripheral,
CS may be tied low.
SERIAL CLOCK (SCLK)
The serial clock (SCLK) features a Schmitt-triggered input and is used to clock data into and out of the device on
the DIN and DOUT/DRDY pins, respectively. Even though the input has hysteresis, TI recommends keeping
SCLK as clean as possible to prevent glitches from accidentally shifting the data. If a complete command is not
sent within 13955 × tMOD (normal mode, duty-cycle mode) or 27910 × tMOD (turbo mode), respectively, the serial
interface resets and the next SCLK pulse starts a new communication cycle. This timeout feature can be used to
recover communication when a serial interface transmission is interrupted. When the serial interface is idle, hold
SCLK low.
DATA READY (DRDY)
DRDY indicates when a new conversion result is ready for retrieval. When DRDY falls low, new conversion data
are ready. DRDY always transitions high on the next SCLK rising edge. When no data are read during
continuous conversion mode, DRDY remains low but pulses high 2 × tMOD before the next DRDY falling edge.
The DRDY pin is always actively driven even when CS is high.
DATA INPUT (DIN)
The data input pin (DIN) is used along with SCLK to send data (commands and register data) to the device. The
device latches data on DIN on the SCLK falling edge. The device never drives the DIN pin.
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DATA OUTPUT AND DATA READY (DOUT/DRDY)
DOUT/DRDY has a dual output function. This pin is used with SCLK to read conversion and register data from
the device but can, in addition, be configured as a data-ready indicator. Data on DOUT/DRDY are shifted out on
the SCLK rising edge. DOUT/DRDY goes to a high-impedance state when CS is high.
Setting the DRDYM bit in the configuration register high also configures DOUT/DRDY as a data-ready indicator.
DOUT/DRDY then transitions low at the same time the DRDY pin goes low to indicate new conversion data are
available. Both signals can be used to detect if new data are ready. However, because DOUT/DRDY is disabled
when CS is high, only the dedicated DRDY pin can be used in case multiple devices on the bus must be
monitored for end of conversion.
DATA FORMAT
The device provides 24 bits of data in binary twos complement format. The positive full-scale input produces an
output code of 7FFFFFh and the negative full-scale input produces an output code of 800000h. The output clips
at these codes for signals that exceed full-scale (FS). Table 11 summarizes the ideal output codes for different
input signals.
Table 11. Ideal Output Code versus Input Signal
INPUT SIGNAL, VIN
(AINP – AINN)
IDEAL OUTPUT CODE (1)
≥ +FS (223 – 1) / 223
7FFFFFh
23
000001h
0
0
–FS / 223
FFFFFFh
≤ –FS
800000h
+FS / 2
(1)
Excludes the effects of noise, INL, offset, and gain errors.
Mapping of the analog input signal to the output codes is illustrated in Figure 59.
7FFFFFh
000001h
000000h
FFFFFFh
¼
Output Code
¼
7FFFFEh
800001h
800000h
¼
-FS
2
23
-FS
2
0
¼
FS
Input Voltage (AINP - AINN)
-1
23
2
23
FS
2
-1
23
Figure 59. Code Transition Diagram
32
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COMMANDS
The device offers six different commands to control device operation. Four commands are stand-alone
instructions (RESET, START/SYNC, POWERDOWN, and RDATA). The commands to read (RREG) and write
(WREG) configuration register data from and to the device require additional information as part of the
instruction.
Operands:
rr = Configuration register (00 to 11)
nn = Number of bytes – 1 (00 to 11)
x = Don't care
WREG (0100 rrnn)
Writes the number of bytes specified by nn (number of bytes to be written – 1) to the device configuration
register, starting at register address rr. The command is completed after nn + 1 bytes are clocked in after the
WREG command byte. The configuration registers are updated on the last SCLK falling edge. For example, the
command to write two bytes (nn = 01) starting at configuration register 0 (rr = 00) is 0100 0001.
RREG (0010 rrnn)
Reads the number of bytes specified by nn (number of bytes to be read – 1) from the device configuration
register, starting at register address rr. The command is completed after nn + 1 bytes are clocked out after the
RREG command byte. For example, the command to read three bytes (nn = 10) starting at configuration register
1 (rr = 01) is 0010 0110.
RESET (0000 011x)
Resets the device to the default values.
START/SYNC (0000 100x)
In single-shot mode, the START/SYNC command is used to start a single conversion or when sent during an
ongoing conversion, to reset the digital filter, and to restart a single new conversion. When the device is set to
continuous conversion mode, the START/SYNC command must be issued to start converting. Sending the
START/SYNC command while converting in continuous conversion mode resets the digital filter and starts
converting from there.
POWERDOWN (0000 001x)
Places the device into power-down mode. This command shuts down all internal analog components, opens the
low-side switch, turns off both IDACs, but holds all register values. As soon as a START/SYNC command is
issued, all analog components return to their previous states.
RDATA (0001 xxxx)
Loads the output shift register with the most recent conversion result. This command can be used when
DOUT/DRDY or DRDY are not monitored to indicate that a new conversion result is available. If a conversion
finishes in the middle of the RDATA command byte, the more reliable result (either the old result or the new one)
is loaded into the output shift register. The state of the DRDY pin signals whether the old or the new result is
loaded. If the old result is loaded, DRDY stays low, indicating that the new result has not been read out. The new
conversion result loads when DRDY is high.
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SENDING COMMANDS
The device serial interface is capable of full-duplex operation, which means commands are decoded at the same
time that conversion data are read. Commands may be sent on any 8-bit data boundary during a data read
operation. When a RREG or RDATA command is recognized, the current data read operation is overridden and
the data is corrupted, unless the command is sent together with the last byte of the data read operation. The
device starts to output the requested data on DOUT/DRDY at the first SCLK rising edge after the command byte.
To read data without interruption, keep DIN low.
A WREG command can be sent without corrupting an ongoing read operation. Figure 60 shows an example for
sending a WREG command to write two configuration registers while reading conversion data in continuous
conversion mode. Note that after the command is clocked in (after the 32nd SCLK falling edge), the device
resets the digital filter and starts converting using the new register settings. The WREG command can be sent on
any of the 8-bit boundaries. That means on the first, ninth, 17th or 25th SCLK rising edge in Figure 60.
Hi-Z
DATA MSB
17
DATA
25
DATA LSB
§
DRDY
9
§ §
DOUT/DRDY
1
§ § § § §
SCLK
§
§
CS
Next Data Ready
2· tMOD
REG_DATA
REG_DATA
§
WREG
§
DIN
Figure 60. Example of a WREG Command
READING DATA
Output pins DRDY and DOUT/DRDY (if configured in the respective DRDYM configuration register bit) transition
low when new data are ready for retrieval. The conversion data are written to an internal data buffer. Data can be
read directly from this buffer on DOUT/DRDY when DRDY falls low. A command does not have to be sent. Data
are shifted out on the SCLK rising edges, MSB first, and consist of three bytes of data.
Figure 61 to Figure 63 show the timing diagrams for continuous conversion mode and single-shot mode.
Hi-Z
DRDY
9
DATA MSB
17
DATA
§ § § §
DOUT/DRDY
1
§ § § § §
SCLK
§
§
CS
DATA LSB
Next Data Ready
2· tMOD
§
§
DIN
Figure 61. Continuous Conversion Mode (DRDYM = 0)
DRDY
Hi-Z
9
DATA MSB
17
DATA
§ § § §
DOUT/DRDY
1
§ § § § §
SCLK
§
§
CS
DATA LSB
Next Data Ready
2· tMOD
§
§
DIN
Figure 62. Continuous Conversion Mode (DRDYM = 1)
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Hi-Z
DRDY
START/SYNC
9
DATA MSB
17
DATA
DATA LSB
Next Data Ready
§
§
DIN
1
§ § § §
DOUT/DRDY
1
§ § § § §
SCLK
§
§
CS
Figure 63. Single-Shot Mode (DRDYM = 0)
Data can also by read at any time without necessarily synchronizing to the DRDY signal using the RDATA
command. When an RDATA command is issued, the conversion result currently stored in the data buffer can be
shifted out on DOUT/DRDY on the following SCLK rising edge. Data can be read continuously with the RDATA
command as an alternative to monitoring DRDY or DOUT/DRDY. The DRDY pin must then be polled after the
LSB is clocked out to determine if a new conversion result is loaded. If a new conversion completes during the
read operation but data from the previous conversion is read, then DRDY is low. Otherwise, if the most recent
result is read, DRDY is high. Figure 64 and Figure 65 illustrate the behavior for both cases.
Hi-Z
DRDY
9
DATA MSB
17
DATA
DATA LSB
Next Data Ready
§
RDATA
§
DIN
1
§ § §
DOUT/DRDY
1
§ § § § §
SCLK
§
§
CS
Figure 64. State of DRDY when a New Conversion Finishes During an RDATA Command
DRDY
9
DATA MSB
17
DATA
RDATA
DATA LSB
Next Data Ready
§
§
DIN
Hi-Z
1
§ § § §
DOUT/DRDY
1
§ § § § §
SCLK
§
§
CS
Figure 65. State of DRDY when the Most Recent Conversion Result is Read During an RDATA Command
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CONFIGURATION REGISTERS
The device has four 8-bit configuration registers that are accessible via the SPI port. The configuration registers
control how the device operates and can be changed at any time without causing data corruption. After power-up
and reset, all registers are set to the default values (which are all '0'). Table 12 shows the register map of the
configuration register.
Table 12. Configuration Register Map (Read/Write)
REGISTER
(Hex)
BIT 7
BIT 6
00h
BIT 5
BIT 4
BIT 3
MUX[3:0]
01h
VREF[1:0]
03h
BIT 1
GAIN[2:0]
DR[2:0]
02h
BIT 2
MODE[1:0]
50/60 [1:0]
I1MUX[2:0]
PSW
I2MUX[2:0]
00h
Configuration Register 0
Bits [7:4]
MUX[3:0]: Input multiplexer configuration
CM
BIT 0
PGA_BYPASS
TS
BCS
IDAC[2:0]
DRDYM
RESERVED
These bits configure the input multiplexer. No effect when in temperature sensor mode.
For settings where AINN = AVSS, the PGA must be disabled (PGA_BYPASS = 1) and only gains 1, 2, and 4 can be
used.
0000 :
0001 :
0010 :
0011 :
0100 :
0101 :
0110 :
0111 :
Bits [3:1]
AINP
AINP
AINP
AINP
AINP
AINP
AINP
AINP
= AIN0, AINN = AIN1 (default)
= AIN0, AINN = AIN2
= AIN0, AINN = AIN3
= AIN1, AINN = AIN2
= AIN1, AINN = AIN3
= AIN2, AINN = AIN3
= AIN1, AINN = AIN0
= AIN3, AINN = AIN2
1000 :
1001 :
1010 :
1011 :
1100 :
1101 :
1110 :
1111 :
AINP = AIN0, AINN = AVSS
AINP = AIN1, AINN = AVSS
AINP = AIN2, AINN = AVSS
AINP = AIN3, AINN = AVSS
(REFPx – REFNx) / 4 monitor (PGA bypassed)
(AVDD – AVSS) / 4 monitor (PGA bypassed)
AINP and AINN shorted to (AVDD + AVSS) / 2
Not used
GAIN[2:0]: Gain configuration
These bits configure the device gain.
Gains 1, 2, and 4 can be used without the PGA. In this case, gain is obtained by a switched-capacitor structure.
The gain setting has no effect when in temperature sensor mode.
000 :
001 :
010 :
011 :
100 :
101 :
110 :
111 :
Bit 0
Gain
Gain
Gain
Gain
Gain
Gain
Gain
Gain
= 1 (default)
=2
=4
=8
= 16
= 32
= 64
= 128
PGA_BYPASS: Disables internal low-noise PGA
Disabling the PGA reduces overall power consumption and allows the common-mode voltage range (VCM) to include
AVSS and AVDD.
The PGA can only be disabled for gains 1, 2, and 4.
The PGA is always enabled for gain settings 8…128, regardless of the PGA_BYPASS setting.
0 : PGA enabled (default)
1 : PGA disabled and bypassed
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01h
Configuration Register 1
Bits [7:5]
DR[2:0]: Data rate
These bits control the data rate setting depending on the selected operating mode.
Normal mode
000 : 20 SPS (default)
001 : 45 SPS
010 : 90 SPS
011 : 175 SPS
100 : 330 SPS
101 : 600 SPS
110 : 1000 SPS
111 : Not used
Bits [4:3]
Duty-cycle mode
000 : 5 SPS
001 : 11.25 SPS
010 : 22.5 SPS
011 : 44 SPS
100 : 82.5 SPS
101 : 150 SPS
110 : 250 SPS
111 : Not used
Turbo mode
000 : 40 SPS
001 : 90 SPS
010 : 180 SPS
011 : 350 SPS
100 : 660 SPS
101 : 1200 SPS
110 : 2000 SPS
111 : Not used
MODE[1:0]: Operating mode
This bit controls the operating mode the device operates in.
00
01
10
11
Bit 2
:
:
:
:
Normal mode (256-kHz modulator clock) (default)
Duty-cycle mode (internal duty cycle of 1:4)
Turbo mode (512-kHz modulator clock)
Not used
CM: Conversion mode
This bit sets the conversion mode for the device.
0 : Single-shot mode (default)
1 : Continuous conversion mode
Bit 1
TS: Temperature sensor mode
This bit enables the internal temperature sensor and puts the device in temperature sensor mode.
0 : Disables temperature sensor (default)
1 : Enables temperature sensor
Bit 0
BCS: Burn-out current sources
This bit controls the 10-µA, burn-out current sources to detect wire breaks and shorts in the sensor.
0 : Current sources off (default)
1 : Current sources on
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02h
Configuration Register 2
Bits [7:6]
VREF[1:0]: Voltage reference selection
These bits select the voltage reference that is used for the conversion.
00
01
10
11
Bits [5:4]
:
:
:
:
Internal 2.048-V reference selected (default)
External reference selected using dedicated REFP0 and REFN0 inputs
External reference selected using AIN0/REFP1 and AIN3/REFN1 inputs
Analog supply AVDD used as reference
50/60[1:0]: FIR filter configuration
Configures the filter coefficients for the internal FIR filter.
Only affects 20-SPS setting in normal mode and 5-SPS setting in duty-cycle mode.
00
01
10
11
Bit [3]
:
:
:
:
No 50-Hz or 60-Hz rejection (default)
Simultaneous 50-Hz and 60-Hz rejection
50-Hz rejection only
60-Hz rejection only
PSW: Low-side power switch configuration
When enabled, the low-side switch connected to AIN3/REFN1 automatically opens when the device is in power-down
mode. When the device is converting, the switch closes.
0 : Switch is always open (default)
1 : Switch closes during conversions
Bits [2:0]
IDAC[2:0]: IDAC current setting
These bits set the current for both IDAC1 and IDAC2 excitation current sources.
000 :
001 :
010 :
011 :
100 :
101 :
110 :
111 :
38
Off (default)
10 µA
50 µA
100 µA
250 µA
500 µA
1000 µA
1500 µA
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03h
Configuration Register 3
Bits [7:5]
I1MUX[2:0]: IDAC1 routing configuration
Selects the channel where IDAC1 is routed to.
000 :
001 :
010 :
011 :
100 :
101 :
110 :
111 :
Bits [4:2]
IDAC1 disabled (default)
IDAC1 connected to AIN0/REFP1
IDAC1 connected to AIN1
IDAC1 connected to AIN2
IDAC1 connected to AIN3/REFN1
IDAC1 connected to REFP0
IDAC1 connected to REFN0
Not used
I2MUX[2:0]: IDAC2 routing configuration
Selects the channel where IDAC2 is routed to.
000 :
001 :
010 :
011 :
100 :
101 :
110 :
111 :
Bit 1
IDAC2 disabled (default)
IDAC2 connected to AIN0/REFP1
IDAC2 connected to AIN1
IDAC2 connected to AIN2
IDAC2 connected to AIN3/REFN1
IDAC2 connected to REFP0
IDAC2 connected to REFN0
Not used
DRDYM: DRDY mode
Controls the behavior of the DOUT/DRDY pin when new data are ready.
0 : Only the dedicated DRDY pin is used to indicate when data are ready (default)
1 : Data ready is indicated simultaneously on DOUT/DRDY and DRDY
Bit 0
Reserved
Always write '0'
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APPLICATION INFORMATION
The following sections give example circuits and suggestions for using the ADS1220 in various situations.
BASIC CONNECTIONS AND LAYOUT CONSIDERATIONS
For many applications, connecting the ADS1220 is simple. Figure 66 shows the principle power-supply and
interface connections for the ADS1220.
GPIO/IRQ
DVSS
DVDD
DIN
0.1 PF
47 O
47 O
47 O
GPIO
SCLK
DOUT
Microcontroller with SPI Interface
16
15
14
13
CS
SCLK
DIN
DOUT/DRDY
3.3 V
1 CLK
2 DGND
DRDY 12
DVDD 11
Device
3 AVSS
3.3 V
3.3 V
0.1 PF
AVDD 10
REFN0
REFP0
AIN1
AIN0/REFP1 9
AIN2
4 AIN3/REFN1
5
6
7
8
0.1 PF
Figure 66. Power-Supply and Interface Connections
Most microcontroller SPI peripherals can operate with the ADS1220. The interface operates in SPI mode 1
where CPOL = 0 and CPHA = 1. In SPI mode 1, SCLK idles low and data are launched or changed only on
SCLK rising edges; data are latched or read by the master and slave on SCLK falling edges. Details of the SPI
communication protocol employed by the ADS1220 can be found in the SPI Timing Characteristics. TI
recommends to place 47-Ω resistors in series with all digital input pins (CS, SCLK, and DIN). This resistance
smooths sharp transitions, suppresses overshoot, and offers some overvoltage protection. Care must be taken to
still meet all SPI timing requirements because the additional resistors interact with the bus capacitances present
on the digital signal lines.
Good power-supply decoupling is important to achieve optimum performance. Both AVDD and DVDD should be
decoupled with at least a 0.1-μF bypass capacitor each. The bypass capacitors should be placed as close to the
power-supply pins as possible with a low impedance connection. For very sensitive systems, or systems in harsh
noise environments, avoiding the use of vias for connecting the bypass capacitor may offer superior bypass and
noise immunity.
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TI recommends employing best design practices when laying out a printed circuit board (PCB) for both analog
and digital components. This recommendation generally means that the layout should separate analog
components [such as ADCs, amplifiers, references, digital-to-analog converters (DACs), and analog MUXs] from
digital components [such as microcontrollers, complex programmable logic devices (CPLDs), field-programmable
gate arrays (FPGAs), radio frequency (RF) transceivers, universal serial bus (USB) transceivers, and switching
regulators]. An example of good component placement is shown in Figure 67. While Figure 67 provides a good
example of component placement, the best placement for each application is unique to the geometries,
components, and PCB fabrication capabilities employed. That is, there is no single layout that is perfect for every
design and careful consideration must always be used when designing with any analog components.
Microcontroller
Device
Ground fill or
Ground plane
Optional: Split
Ground Cut
Signal
Conditioning
(RC filters
and
amplifiers)
Ground fill or
Ground plane
Optional: Split
Ground Cut
Ground fill or
Ground plane
Supply
Generation
Interface
Tranceiver
Connector
or Antenna
Ground fill or
Ground plane
Figure 67. System Component Placement
The use of split analog and digital ground planes is not necessary for improved noise performance (although for
thermal isolation this option is a worthwhile consideration). However, the use of a solid ground plane or ground
fill in PCB areas with no components is essential for optimum performance. If the system being used employs a
split digital and analog ground plane, TI generally recommends that the ground planes be connected together as
close to the ADS1220 as possible.
TI also strongly recommends that digital components, especially RF portions, be kept as far as practically
possible from analog circuitry in a given system. Additionally, minimize the distance that digital control traces run
through analog areas and avoid placing these traces near sensitive analog components. Digital return currents
usually flow through a ground path that is as close to the digital path as possible. If a solid ground connection to
a plane is not available, these currents may find paths back to the source that interfere with analog performance.
The implications that layout has on the temperature sensing functions are much more significant than for ADC
functions.
CONNECTING MULTIPLE DEVICES
When connecting multiple ADS1220 devices to a single SPI bus, SCLK, DIN, and DOUT/DRDY can be safely
shared by using a dedicated chip-select (CS) line for each SPI-enabled device. When CS transitions high for the
respective ADS1220, DOUT/DRDY enters a tri-state mode. Therefore, DOUT/DRDY cannot be used to indicate
when new data are available if CS is high, regardless if bit DRDYM in the configuration register is set to '0' or '1'.
Only the dedicated DRDY pin indicates that new data are available, because the DRDY pin is actively driven
even when CS is high.
In some cases, however, the DRDY pin cannot be interfaced to the microcontroller, perhaps because of
insufficient GPIO channels on the microcontroller or because the serial interface must be galvanically isolated
and thus the amount of channels has to be limited. Therefore, in order to evaluate when a new conversion of one
of the devices is ready, the microcontroller can periodically drop CS to the respective ADS1220. When CS goes
low, the DOUT/DRDY pin immediately drives either high or low, provided that bit DRDYM is configured to '1'. If
the DOUT/DRDY line drives low on a low CS, new data are currently available for clocking out. If the
DOUT/DRDY line drives high, no new data are available. Alternatively, valid data can be retrieved from the
ADS1220 at any time without concern of data corruption by using the RDATA command.
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THERMOCOUPLE MEASUREMENT
Figure 68 shows the basic connections of a thermocouple measurement system, using the internal high-precision
temperature sensor for cold-junction compensation. Apart from the thermocouple itself, the only external circuitry
required are two biasing resistors, a simple low-pass, antialiasing filter, and the power-supply decoupling
capacitors.
3.3V
3.3V
0.1 PF
3.3V
0.1 PF
REFP0
10 A to
1.5 mA
CCM1
RB1
RF1
DVDD
AVDD
Internal
Reference
AIN0
REFN0
Reference
Mux
Device
24-bit
ûADC
Digital Filter
and
SPI
Interface
Low Drift
Oscillator
Precision
Temp Sensor
CDIF
RF2
Thermocouple
RB2
AIN1
Mux
CCM2
PGA
AIN2
AIN3
CLK
AVSS
CS
SCLK
DIN
DOUT/DRDY
DRDY
DGND
Figure 68. Thermocouple Measurement
The biasing resistors RB1 and RB2 are used to set the common-mode voltage of the thermocouple to within the
specified common-mode voltage range of the PGA (in this example, to mid-supply AVDD / 2). In case the
application requires the thermocouple to be biased to GND, a bipolar supply (for example, AVSS = –2.5 V and
AVDD = +2.5 V) must be used for the ADS1220 to meet the common-mode voltage requirement. When choosing
the values of the biasing resistors, care must be taken so that the biasing current does not degrade
measurement accuracy. The biasing current flows through the thermocouple where it can cause self-heating and
additional voltage drops in the thermocouple leads.
In addition to biasing the thermocouple, RB1 and RB2 are also useful to detect an open thermocouple lead. When
one of the thermocouple leads fails open, the biasing resistors pull the analog inputs AIN0 and AIN1 to AVDD
and AVSS, respectively. The ADC consequently reads a full-scale value, which is outside the normal
measurement range of the thermocouple voltage, to indicate this failure condition.
While the digital filter of the ADS1220 attenuates high-frequency components of noise, TI generally recommends
providing a first-order, passive RC filter at the inputs to further improve performance. The differential RC filter
formed by RF1, RF2 and the differential capacitor CDIF offers a cutoff frequency of fC = 1 / (2π × RF1 × CDIF). Two
common-mode filter capacitors CM1 and CM2 are also added to offer attenuation of high-frequency common-mode
noise components. Because mismatches in the common-mode capacitors cause differential noise, TI
recommends that the differential capacitor CDIF be at least an order of magnitude (10x) larger than the commonmode capacitors CM1 and CM2.
The filter resistors RF1 and RF2 also serve as current-limiting resistors. These resistors limit the current into the
analog inputs (AIN0 and AIN1) of the ADS1220 to safe levels, should an overvoltage on the inputs occur. TI
recommends limiting the filter resistor values to below 1 kΩ. Larger filter resistor values can lead to additional
offset errors because of the voltage drops across them caused by the differential input currents of the ADS1220.
The ADS1220 integrates a high-precision temperature sensor that can be used to measure the temperature of
the cold junction. To measure the internal temperature of the ADS1220, the device must be set to internal
temperature sensor mode by setting bit TS to '1' in the configuration register. For best performance, careful
board layout is critical to achieve good thermal conductivity between the cold junction and the ADS1220
package.
42
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However, the ADS1220 does not perform automatic cold-junction compensation of the thermocouple. This
compensation must be done in the microcontroller that interfaces to the ADS1220. The microcontroller requests
one or multiple readings of the thermocouple voltage from the ADS1220 and then sets the device to internal
temperature sensor mode (TS = 1) to acquire the temperature of the cold junction. The calculations to
compensate for the cold-junction temperature must be implemented on the microcontroller.
In some applications, the integrated temperature sensor cannot be used (for example, if the accuracy is not high
enough or if the ADS1220 cannot be placed close enough to the cold junction). The additional analog input
channels of the ADS1220 can be used in this case to measure the cold-junction temperature using a thermistor
or RTD.
RTD MEASUREMENT
The ADS1220 integrates all necessary features (such as dual-matched programmable current sources, buffered
reference inputs, PGA, and so forth) to ease the implementation of ratiometric 2-, 3-, and 4-wire RTD
measurements. Figure 69 shows a typical implementation of a ratiometric 3-wire RTD measurement using the
excitation current sources integrated in the ADS1220 to excite the RTD as well as to implement automatic RTD
lead-resistance compensation.
RFEF
3.3V
RF3
CCM3
0.1 PF
10 A to
1.5 mA
RLEAD3
RLEAD2
RF2
RLEAD1
RF1
CCM2
AVDD
CDIF2
REFP0
Internal
Reference
AIN0
3.3V
RF4
CCM4
0.1 PF
REFN0
DVDD
Reference
Mux
Device
24-bit
ûADC
Digital Filter
and
SPI
Interface
Low Drift
Oscillator
Precision
Temp Sensor
CDIF1
3-wire RTD
AIN1
CCM1
Mux
PGA
IDAC1
IDAC2
AVSS
CLK
CS
SCLK
DIN
DOUT/DRDY
DRDY
DGND
Figure 69. 3-Wire RTD Measurement
The circuit in Figure 69 employs a ratiometric measurement approach. In other words, the sensor signal (that is
the voltage across the RTD in this case) and the reference voltage for the ADC are derived from the same
excitation source. Therefore, errors resulting from temperature drift or noise cancel out because these errors are
common to both the sensor signal and the reference.
In order to implement a ratiometric 3-wire RTD measurement using the ADS1220, IDAC1 is routed to one of the
excitation leads of the RTD while IDAC2 is routed to the second excitation lead. Both currents have the same
value, which is programmable by bits IDAC[2:0] in the configuration register. The design of the ADS1220 ensures
that both IDAC values are closely matched, even across temperature. The sum of both currents flows through a
low-drift reference resistor, RREF. The voltage, VREF, generated across the reference resistor is as shown in
Equation 7. Because IDAC1 = IDAC2, Equation 8 is then used as the ADC reference voltage.
VREF = (IDAC1 + IDAC2) × RREF
VREF = 2 × IDAC1 × RREF
(7)
(8)
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Equation 9 assumes for the moment that the individual lead resistance values of the RTD (RLEADx) are zero. Only
IDAC1 excites the RTD to produce a voltage VRTD, which is proportional to the temperature dependable RTD
value and the IDAC1 value.
VRTD = RRTD (Temperature) × IIDAC1
(9)
The ADS1220 internally amplifies the voltage across the RTD using the PGA and compares the resulting voltage
against the reference voltage to produce a digital output code, which is proportional to Equation 10 to
Equation 12:
Code ∝ VRTD × PGA / VREF
Code ∝ [RRTD (Temperature) × IIDAC1 × PGA] / [2 × IDAC1 × RREF]
Code ∝ [RRTD (Temperature) × PGA] / [2 × RREF]
(10)
(11)
(12)
As can be seen from Equation 12, the output code only depends on the value of the RTD, the PGA gain and the
reference resistor (RREF), but not on the IDAC1 value. The absolute accuracy and temperature drift of the
excitation current therefore does not matter. However, because the value of the reference resistor directly
impacts the measurement result, choosing a reference resistor with a very low temperature coefficient is
important to limit errors introduced by the temperature drift of RREF.
The second IDAC2 is used to compensate for errors introduced by the voltage drop across the lead resistance of
the RTD. All three leads of a 3-wire RTD typically have the same length and, thus, the same lead resistance.
Also, IDAC1 and IDAC2 have the same value. Consequently, the differential voltage (VIN) across the ADC inputs,
AIN0 and AIN1, is as shown in Equation 13:
VIN = VAIN0 – VAIN1 = IIDAC1 × (RRTD + RLEAD1) – IIDAC2 × RLEAD2
(13)
When RLEAD1 = RLEAD2 and IIDAC1 = IIDAC2, Equation 13 reduces to Equation 14:
VIN = IIDAC1 × RRTD
(14)
In other words, the measurement error resulting from the voltage drop across the RTD lead resistance is
compensated, as long as the lead resistance values and the IDAC values are well matched.
A first-order differential and common-mode RC filter (RF1, RF2, CDIF1, CCM1, CCM2) is placed on the ADC inputs,
as well as on the reference inputs (RF3, RF4, CDIF2, CCM3, CCM4). The same guidelines for designing the input filter
apply as described in the Thermocouple Measurement section. For best performance, TI recommends to match
the corner frequencies of the input and reference filter. More detailed information on matching the input and
reference filter can be found in application report RTD Ratiometric Measurements and Filtering Using the
ADS1148 and ADS1248 (SBAA201).
The reference resistor RREF not only serves to generate the reference voltage for the ADS1220, but also sets the
common-mode voltage of the RTD to within the specified common-mode voltage range of the PGA. In other
words, the voltage across the reference resistor must meet Equation 5.
When designing the circuit, care should also be taken to meet the compliance voltage requirement of the IDACs.
The IDACs require a minimum headroom of (AVDD – 0.9 V) in order to operate accurately. This requirement
means that Equation 15 must be met at all times.
AVSS + IIDAC1 × (RLEAD1 + RRTD) + (IIDAC1 + IIDAC2) × (RLEAD3 + RREF) ≤ AVDD – 0.9 V
(15)
The ADS1220 also offers the possibility to route the IDACs to the same inputs used for measurement. In case
the filter resistor values RF1 and RF2 are small enough and well matched, IDAC1 can be routed to AIN1 and
IDAC2 to AIN0, respectively, in Figure 69. In this manner, even two 3-wire RTDs sharing the same reference
resistor can be measured with a single ADS1220.
Implementing a 2- or 4-wire RTD measurement is very similar to the 3-wire RTD measurement shown in
Figure 69 except that only one IDAC is required.
44
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HIGH-LEVEL CODE EXAMPLE
The following list shows a high-level code sequence with steps necessary to set up the ADS1220 and the
microcontroller interfacing to it, in order to take subsequent readings from the ADS1220 in continuous conversion
mode. The dedicated DRDY pin is used to indicated availability of new conversion data. The default configuration
register settings are changed to PGA = 16, continuous conversion mode, and simultaneous 50-Hz and 60-Hz
rejection.
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
Power-up
Delay
Configure the SPI interface of the microcontroller to SPI mode 1
If the CS pin is not tied low permanently, configure the microcontroller GPIO connected to CS as an output
Configure the microcontroller GPIO connected to the DRDY pin as an interrupt input
Set CS to the ADS1220 low
Delay
Send the RESET command (06h) to make sure the ADS1220 is properly reset after power-up
Write the respective register configuration using the WREG command (43h, 08h, 04h, 10h, and 00h)
Delay
Read back all configuration registers using the RREG command (23h) to make sure the correct values are
written
Delay
Send the START/SYNC command (08h) to start converting in continuous conversion mode
Delay
Clear CS to high (resets the serial interface)
Loop
{
Wait for DRDY to transition low
Take CS low
Delay
Send 24 SCLK rising edges to read out conversion data on DOUT
Delay
Clear CS to high
}
•
•
•
•
•
Take CS low
Delay
Send the POWERDOWN command (02h) to stop conversions and put the ADS1220 in power-down mode
Delay
Clear CS to high
TI recommends running an offset calibration before performing any measurements or when changing the gain of
the PGA. The internal offset of the ADS1220 can, for example, be measured by shorting the inputs to mid-supply
(MUX[3:1] = 1110). The microcontroller then takes multiple readings from the ADS1220 with the inputs shorted
and stores the average value in the microcontroller memory. When measuring the sensor signal, the
microcontroller then subtracts the stored offset value from each ADS1220 reading to get an offset compensated
result.
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ADS1220
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www.ti.com
REVISION HISTORY
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Original (May 2013) to Revision A
•
46
Page
Changed document status to Mixed Status; pre-RTM changes made throughout ............................................................... 1
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PACKAGE OPTION ADDENDUM
www.ti.com
5-Feb-2014
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
Lead/Ball Finish
MSL Peak Temp
(2)
(6)
(3)
Op Temp (°C)
Device Marking
(4/5)
ADS1220IPW
ACTIVE
TSSOP
PW
16
90
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
-40 to 125
ADS1220
ADS1220IPWR
ACTIVE
TSSOP
PW
16
2500
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
-40 to 125
ADS1220
ADS1220IRVAR
PREVIEW
VQFN
RVA
16
3000
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
-40 to 125
1220
ADS1220IRVAT
PREVIEW
VQFN
RVA
16
250
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
-40 to 125
1220
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3)
MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4)
There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5)
Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6)
Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish
value exceeds the maximum column width.
Addendum-Page 1
Samples
PACKAGE OPTION ADDENDUM
www.ti.com
5-Feb-2014
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
30-Jul-2013
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
ADS1220IPWR
Package Package Pins
Type Drawing
TSSOP
PW
16
SPQ
Reel
Reel
A0
Diameter Width (mm)
(mm) W1 (mm)
2500
330.0
12.4
Pack Materials-Page 1
6.9
B0
(mm)
K0
(mm)
P1
(mm)
5.6
1.6
8.0
W
Pin1
(mm) Quadrant
12.0
Q1
PACKAGE MATERIALS INFORMATION
www.ti.com
30-Jul-2013
*All dimensions are nominal
Device
Package Type
Package Drawing
Pins
SPQ
Length (mm)
Width (mm)
Height (mm)
ADS1220IPWR
TSSOP
PW
16
2500
367.0
367.0
35.0
Pack Materials-Page 2
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