Micrel MIC2164-2YMM Constant frequency, synchronous buck controllers featuring adaptive on-time control Datasheet

MIC2164/-2/-3
Constant Frequency, Synchronous
Buck Controllers Featuring
Adaptive On-Time Control
Hyper Speed Control™ Family
General Description
Features
The Micrel MIC2164/-2/-3 are constant frequency,
synchronous buck controllers featuring adaptive on-time
control. The MIC2164/-2/-3 are the first products in the new
TM
Hyper Speed Control family of buck controllers introduced
by Micrel.
The MIC2164/-2/-3 controllers operate over an input supply
range of 3V to 28V, and are independent of the IC supply
voltage. The devices are capable of supplying 25A output
current. While the MIC2164 operates at 300kHz, the
MIC2164-2 operates at 600kHz, and the MIC2164-3 operates
at 1MHz.
TM
A unique Hyper Speed Control architecture allows for ultrafast transient response while reducing the output capacitance
and also makes High VIN/Low VOUT operation possible. The
MIC2164/-2/-3 controllers utilizes an architecture which is
adaptive Ton ripple controlled. A UVLO feature is provided to
ensure proper operation under power-sag conditions to
prevent the external power MOSFET from overheating. A soft
start feature is provided to reduce the inrush current.
Foldback current limit and “hiccup” mode short-circuit
protection ensure FET and load protection.
The MIC2164/-2/-3 controllers are available in a 10-pin
MSOP (MAX1954A-compatible) package with a junction
operating range from –40°C to +125°C.
All support documentation can be found on Micrel’s web site
at: www.micrel.com.
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TM
Hyper Speed Control architecture enables
- High delta V operation (VIN=28V and VOUT=0.8V)
- Smaller output capacitors than competitors
3V to 28V input voltage
Stable with zero-ESR output capacitor
25A output current capability
300kHz/600kHz/1MHz switching frequency
Adaptive on-time mode control
Output down to 0.8V with ±1% FB accuracy
Up to 95% efficiency
Foldback current limit and “hiccup” mode short-circuit
protection
6ms Internal soft start
Thermal shutdown
Pre-bias output safe
–40°C to +125°C junction temperature range
Available in 10-pin MSOP package
Applications
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Set-top box, gateways and routers
Printers, scanners, graphic cards and video cards
Telecommunication, PCs and servers
Microprocessor core supply
Low-voltage distributed power
________________________________________________________________________________________________________________________
Typical Application
MIC2164
12V to 3.3V Efficiency
100
95
EFFICIENCY (%)
90
85
80
75
70
65
60
VIN=5V
55
50
MIC2164/-2/-3 Synchronous Controllers Featuring Adaptive On-Time Control
0
4
8
12
16
20
OUTPUT CURRENT (A)
MLF and MicroLeadFrame are registered trademarks of Amkor Technology, Inc.
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
September 2009
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Ordering Information
Part Number
Voltage
Switching Frequency
Junction Temp. Range
Package
Lead Finish
MIC2164YMM
Adj.
300kHz
–40° to +125°C
10-pin MSOP
Pb-Free
MIC2164-2YMM
Adj.
600kHz
–40° to +125°C
10-pin MSOP
Pb-Free
MIC2164-3YMM
Adj.
1MHz
–40° to +125°C
10-pin MSOP
Pb-Free
Pin Configuration
10-Pin MSOP (MM)
Pin Description
Pin
Number
Pin Name
1
HSD
High-Side N-MOSFET Drain Connection (input): Power to the drain of the external high-side N-channel
MOSFET. The HSD operating voltage range is from 3V to 28V. Input capacitors between HSD and the
power ground (PGND) are required.
2
EN
Enable (input): A logic level control of the output. The EN pin is CMOS-compatible. Logic high or floating
= enable, logic low = shutdown. In the off state, supply current of the device is greatly reduced (typically
0.8mA).
3
FB
Feedback (input): Input to the transconductance amplifier of the control loop. The FB pin is regulated to
0.8V. A resistor divider connecting the feedback to the output is used to adjust the desired output
voltage.
4
GND
5
IN
Input Voltage (input): Power to the internal reference and control sections of the MIC2164/-2/-3. The IN
operating voltage range is from 3V to 5.5V. A 1µF and 0.1µF ceramic capacitors from IN to GND are
recommended for clean operation.
6
DL
Low-Side Drive (output): High-current driver output for external low-side MOSFET. The DL driving
voltage swings from ground-to-IN.
7
PGND
8
DH
High-Side Drive (output): High-current driver output for external high-side MOSFET. The DH driving
voltage is floating on the switch node voltage (LX). It swings from ground to VIN minus the diode drop.
Adding a small resistor between DH pin and the gate of the high-side N-channel MOSFETs can slow
down the turn-on and turn-off time of the MOSFETs.
9
LX
Switch Node and Current Sense input: High current output driver return. The LX pin connects directly to
the switch node. Due to the high speed switching on this pin, the LX pin should be routed away from
sensitive nodes. LX pin also senses the current by monitoring the voltage across the low-side MOSFET
during OFF time. In order to sense the current accurately, connect the low-side MOSFET drain to LX
using a Kelvin connection.
10
BST
September 2009
Pin Function
Signal ground. GND is the ground path for the device input voltage VIN and the control circuitry. The loop
for the signal ground should be separate from the power ground (PGND) loop.
Power Ground. PGND is the ground path for the MIC2164/-2/-3 buck converter power stage. The PGND
pin connects to the sources of low-side N-Channel MOSFETs, the negative terminals of input capacitors,
and the negative terminals of output capacitors. The loop for the power ground should be as small as
possible and separate from the Signal ground (GND) loop.
Boost (output): Bootstrapped voltage to the high-side N-channel MOSFET driver. A Schottky diode is
connected between the IN pin and the BST pin. A boost capacitor of 0.1μF is connected between the
BST pin and the LX pin. Adding a small resistor in series with the boost capacitor can slow down the
turn-on time of high-side N-Channel MOSFETs.
2
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Absolute Maximum Ratings(1)
Operating Ratings(2)
IN, FB, EN to GND ........................................... -0.3V to +6V
BST to LX ......................................................... -0.3V to +6V
BST to GND ................................................... -0.3V to +37V
DH to LX.............................................-0.3V to (VBST + 0.3V)
DL, COMP to GND ...............................-0.3V to (VIN + 0.3V)
HSD to GND..................................................... -0.3V to 31V
PGND to GND ............................................... -0.3V to +0.3V
Junction Temperature .............................................. +150°C
Storage Temperature (TS)..........................-65°C to +150°C
Lead Temperature (soldering, 10sec) ........................ 260°C
Input Voltage (VIN) .......................................... 3.0V to 5.5V
Supply Voltage (VHSD) ....................................... 3.0V to 28V
Operating Temperature Range ..................-40°C to +125°C
Junction Temperature (TJ) .........................-40°C to +125°C
Junction Thermal Resistance
MSOP (θJA) ............................................130.5°C/W
Continuous Power Dissipation (TA = 70°C)..............421mW
(derate 5.6mW/°C above 70°C)
Electrical Characteristics(4)
VBST - VLX = 5V; TA = 25°C, unless noted. Bold values indicate -40°C ≤ TJ ≤ +125°C.
Parameter
Condition
Min
Typ
Max
Units
3.0
5.5
V
3.0
28
V
General
Operating Input Voltage (VIN)
(5)
HSD Voltage Range (VHSD)
Quiescent Supply Current
Standby Supply Current
(6)
(VFB = 1.5V, output switching but excluding
external MOSFET gate current)
1.4
3.0
mA
VIN = VBST = 5.5V, VHSD = 28, LX = unconnected,
EN = GND
0.8
2
mA
2.7
3
V
Under-voltage Lockout Trip Level
2.4
UVLO Hysteresis
50
mV
DC-DC Controller
Output-Voltage Adjust Range
(7)
(VOUT)
V
0.8
Error Amplifier
FB Regulation Voltage
0°C ≤ TJ ≤ 85°C
-1
FB Regulation Voltage
-40°C ≤ TJ ≤ 125°C
-2
FB Input Leakage Current
Current-Limit Threshold
1
%
2
%
5
500
nA
VFB = 0.8V
103
130
162
mV
VFB = 0V
19
48
77
mV
Soft-Start
Soft-start Period
6
ms
Notes:
1. Exceeding the absolute maximum rating may damage the device.
2. The device is not guaranteed to function outside its operating rating.
3. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5k in series with 100pF.
4. Specification for packaged product only.
5. The application is fully functional at low IN (supply of the control section) if the external MOSFETs have enough low voltage VTH.
6. The current will come only from the internal 100kΩ pull-up resistor sitting on the EN Input and tied to IN.
7. The maximum VOUT value is limited by the Fixed TON estimator which obtains VOUT a divided by 6 value (1/6).
September 2009
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M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Parameter
Condition
Min
Typ
Max
Units
MIC2164
0.225
0.3
0.375
MHz
MIC2164-2
0.45
0.6
0.75
MHz
MIC2164-3
0.75
1
1.25
MHz
Oscillator
Switching Frequency
Maximum Duty Cycle
(8)
(9)
Minimum Duty Cycle
MIC2164
87
%
MIC2164-2
74
%
MIC2164-3
66
%
Measured at DH
0
%
FET Drives
DH, DL Output Low Voltage
ISINK = 10mA
DH, DL Output High Voltage
ISOURCE = 10mA
0.1
VIN-0.1V
or
VBST-0.1V
DH On-Resistance, High State
V
V
2.1
3.3
Ω
DH On-Resistance, Low State
1.8
3.3
Ω
DL On-Resistance, High State
1.8
3.3
Ω
DL On-Resistance, Low State
1.2
2.3
Ω
LX Leakage Current
VLX = 28V, VIN = 5.5V,VBST = 33.5V
50
µA
HSD Leakage Current
VLX = 28V, VIN = 5.5V,VBST = 33.5V
20
µA
Thermal Protection
Over-temperature Shutdown
155
°C
Over-temperature Shutdown
Hysteresis
10
°C
0.8
V
Shutdown Control
En Logic Level Low
3V < VIN <5.5V
En Logic Level High
3V < VIN <5.5V
0.4
0.9
En Pull-up Current
50
1.2
V
µA
Note:
8. Measured in test mode.
9. Measured at DH. The maximum duty cycle is limited by the fixed mandatory off time Toff of typical 363ns.
September 2009
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Micrel, Inc.
MIC2164/-2/-3
Typical Characteristics
MIC2164
12V to 1.5V Efficiency
100
95
90
90
90
80
75
70
65
60
85
80
75
70
65
60
VIN=5V
55
EFFICIENCY (%)
95
85
75
70
65
60
VIN=5V
4
8
12
16
20
50
0
4
8
12
16
20
0
MIC2164-3
12V to 1.5V Efficiency
100
90
80
75
70
65
EFFICIENCY (%)
90
EFFICIENCY (%)
95
90
85
85
80
75
70
65
60
VIN=5V
55
50
6
9
12
15
75
70
65
60
2
4
6
8
10
0
Feedback Voltage
vs. Input Voltage
0.84
0.81
0.80
0.79
0.78
VHSD=12V
VIN=5V
FEEDBACK VOLTAGE (V)
0.85
0.84
FEEDBACK VOLTAGE (V)
0.85
0.82
0.83
0.82
0.81
0.80
0.79
0.78
0.77
VHSD=12V
0.76
4
8
12
16
3.5
Feedback Voltage
vs. Temperature
0.82
0.81
0.80
0.79
0.78
4
4.5
5
5.5
3
680
0.796
0.794
VIN=5V
0.792
0.790
-40
-20
0
20
40
60
80
TEMPERATURE (°C)
100
120
330
320
310
300
290
280
VHSD=12V
VIN=5V
270
260
250
23
28
660
640
620
600
580
560
540
VHSD=12V
VIN=5V
520
500
0
4
8
12
OUTPUT CURRENT (A)
September 2009
SWITCHING FREQUENCY (kHz)
700
0.798
18
MIC2164-2
Switching Frequency vs. Load
340
0.800
13
MIC2164
Switching Frequency vs. Load
350
0.802
8
HSD VOLTAGE (V)
0.808
0.804
VIN=5V
0.77
0.810
SWITCHING FREQUENCY (kHz)
FEEDBACK VOLTAGE (V)
0.83
INPUT VOLTAGE (V)
OUTPUT CURRENT (A)
0.806
10
0.75
3
20
8
0.76
0.75
0.75
6
Feedback Voltage
vs. HSD Voltage
0.84
0
4
OUTPUT CURRENT (A)
0.85
0.76
2
OUTPUT CURRENT (A)
Feedback Voltage vs. Load
0.77
VIN=5V
55
50
0
OUTPUT CURRENT (A)
0.83
15
80
50
3
12
85
VIN=5V
55
9
MIC2164-3
12V to 3.3V Efficiency
100
95
0
6
OUTPUT CURRENT (A)
95
60
3
OUTPUT CURRENT (A)
MIC2164-2
12V to 3.3V Efficiency
100
VIN=5V
55
OUTPUT CURRENT (A)
EFFICIENCY (%)
80
50
0
FEEDBACK VOLTAGE (V)
85
55
50
MIC2164-2
12V to 1.5V Efficiency
100
95
EFFICIENCY (%)
EFFICIENCY (%)
100
MIC2164
12V to 3.3V Efficiency
5
16
20
0
3
6
9
12
15
OUTPUT CURRENT (A)
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Typical Characteristics (continued)
MIC2164
Switching Frequency vs. VIN
350
1120
340
1090
1060
1030
1000
970
940
VHSD=12V
VIN=5V
910
880
850
MIC2164-2
Switching Frequency vs. VIN
700
SWITCHING FREQUENCY (kHz)
1150
SWITCHING FREQUENCY (kHz)
SWITCHING FREQUENCY (kHz)
MIC2164-3
Switching Frequency vs. Load
330
320
310
300
290
280
VHSD=12V
270
260
250
0
2
4
6
8
10
3
3.5
OUTPUT CURRENT (A)
4
4.5
5
5.5
680
660
640
620
600
580
560
VHSD=12V
540
520
500
INPUT VOLTAGE (V)
3
3.5
4
4.5
5
5.5
INPUT VOLTAGE (V)
MIC2164
Switching Frequency vs. VHSD
MIC2164-3
Switching Frequency vs. VIN
1120
1090
1060
1030
1000
970
VHSD=12V
910
880
700
340
330
VIN=5V
320
310
300
290
280
270
260
250
850
3
3.5
4
4.5
5
3
5.5
8
13
MIC2164-3
Switching Frequency vs. VHSD
23
660
620
600
580
560
540
520
500
28
3
700
340
680
VIN=5V
1030
1000
970
940
910
880
850
SWITCHING FREQUENCY (kHz)
350
1090
330
320
310
300
290
280
270
VIN=5V
260
250
13
18
23
28
MIC2164-3 Switching
Frequency vs. Temperature
-20
1120
135
CURRENT LIMIT THRESHOLD
(mV)
1150
150
1030
1000
970
940
910
VIN=5V
0
20
40
60
80
September 2009
80
100
620
600
580
560
540
VIN=5V
520
120
-40
-20
0
20
100
120
40
60
80
100
120
TEMPERATURE (°C)
Current Limit Threshold vs.
Feedback Voltage Percentage
Current Limit Threshold
vs. Temperature
150
75
60
45
30
0
TEMPERATURE (°C)
60
90
850
0
40
105
15
-20
20
120
880
-40
28
640
TEMPERATURE (°C)
1060
23
500
-40
HSD VOLTAGE (V)
1090
18
660
CURRENT LIMIT THRESHOLD
(mV)
8
13
MIC2164-2 Switching
Frequency vs. Temperature
1120
3
8
HSD VOLTAGE (V)
1150
1060
VIN=5V
VOUT=2.5V
640
MIC2164 Switching Frequency
vs. Temperature
SWITCHING FREQUENCY (kHz)
SWITCHING FREQUENCY (kHz)
18
680
HSD VOLTAGE (V)
INPUT VOLTAGE (V)
SWITCHING FREQUENCY (kHz)
SWITCHING FREQUENCY (kHz)
SWITCHING FREQUENCY (kHz)
SWITCHING FREQUENCY (kHz)
1150
940
MIC2164-2
Switching Frequency vs. VHSD
350
VIN=5V
135
120
105
90
VFB=0.8V
75
VFB=0V
60
45
30
15
0
0
10
20
30
40
50
60
70
80
90
Feedback Voltage Percentage (%)
6
100
-40
-20
0
20
40
60
80
100
120
TEMPERATURE (°C)
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Typical Characteristics (continued)
Quiescent Supply Current
vs. Input Voltage
2
QUIESCENT SUPPLY
CURRENT (mA)
1.8
1.6
1.4
1.2
1
0.8
0.6
0.4
0.2
0
3
3.5
4
4.5
5
5.5
INPUT VOLTAGE (V)
September 2009
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M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Functional Characteristics
MIC2164 Switching Waveforms (Light Load)
IL
(5A/div)
DH
(20V/div)
LX
(10V/div)
DL
(5V/div)
Vhsd = 12V Vin = 5V Vout = 3.3V L = 1.5H Iout = 1A
Time 2μs/div
September 2009
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Micrel, Inc.
MIC2164/-2/-3
Functional Characteristics (continue)
September 2009
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M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Functional Characteristics (continue)
September 2009
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M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Functional Diagram
Figure 1. MIC2164/-2/-3 Block Diagram
September 2009
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Micrel, Inc.
MIC2164/-2/-3
The estimated-ON-time method results in a constant
switching frequency in MIC2164/-2/-3. The actual ON
time is varied with the different rising and falling time of
the external MOSFETs. Therefore, the type of the
external MOSFETs, the output load current, and the
control circuitry power supply VIN will modify the actual
ON time and the switching frequency. Also, the minimum
Ton results in a lower switching frequency in the high
VHSD and low VOUT applications, such as 24V to 1.0V
MIC2164-3 application. The minimum Ton measured on
the MIC2164 evaluation board is about 138ns. During
the load transient, the switching frequency is changed
due to the varying OFF time.
To illustrate the control loop, the steady-state scenario
and the load transient scenario are analyzed. For easy
analysis, the gain of the gm amplifier is assumed to be 1.
With this assumption, the inverting input of the error
comparator is the same as the FB voltage. Figure 2
shows the MIC2164/-2/-3 control loop timing during the
steady-state. During the steady-state, the gm amplifier
senses the FB voltage ripple, which is proportional to the
output voltage ripple and the inductor current ripple, to
trigger the ON-time period. The ON time is
predetermined by the estimation. The ending of OFF
time is controlled by the FB voltage. At the valley of the
FB voltage ripple, which is below than VREF, OFF period
ends and the next ON-time period is triggered through
the control logic circuitry.
Functional Description
The MIC2164/-2/-3 is a adaptive on-time synchronous
buck controller family built for low cost and high
performance. They are designed for wide input voltage
range from 3V to 28V and for high output power buck
converters. An estimated-ON-time method is applied in
MIC2164/-2/-3 to obtain a constant switching frequency
and to simplify the control compensation. The overcurrent protection is implemented without the use of an
external sense resistor. It includes an internal soft-start
function which reduces the power supply input surge
current at start-up by controlling the output voltage rise
time.
Theory of Operation
The MIC2164/-2/-3 is a adaptive on-time buck controller
family. Figure 1 illustrates the block diagram for the
control loop. The output voltage variation will be sensed
by the MIC2164/-2/-3 feedback pin FB via the voltage
divider R1 and R2, and compared to a 0.8V reference
voltage VREF at the error comparator through a low gain
transconductance (gm) amplifier, which improves the
MIC2164/-2/-3 converter output voltage regulation. If the
FB voltage decreases and the output of the gm amplifier
is below 0.8V, The error comparator will trigger the
control logic and generate an ON-time period, in which
DH pin is logic high and DL pin is logic low. The ON-time
period length is predetermined by the “FIXED TON
ESTIMATION” circuitry:
TON(estimat ed) =
VOUT
VHSD ⋅ fsw
(1)
where VOUT is the output voltage, VHSD is the power
stage input voltage, and fSW is the switching frequency
(300kHz for MIC2164, 600kHz for MIC2164-2, and
1MHz for MIC2164-3).
After ON-time period, the MIC2164/-2/-3 goes into the
OFF-time period, in which DH pin is logic low and DL pin
is logic high. The OFF-time period length is depending
on the FB voltage in most cases. When the FB voltage
decreases and the output of the gm amplifier is below
0.8V, the ON-time period is trigger and the OFF-time
period ends. If the OFF-time period decided by the FB
voltage is less than the minimum OFF time TOFF(min),
which is about 363ns typical, the MIC2164/-2/-3 control
logic will apply the TOFF(min) instead. TOFF(min) is required
by the BST charging. The maximum duty cycle is
obtained from the 363ns TOFF(min):
Dmax =
TS − TOFF(min)
TS
= 1−
Figure 2. MIC2164/-2/-3 Control Loop Timing
Figure 3 shows the load transient scenario of the
MIC2164/-2/-3 converter. The output voltage drops due
to the sudden load increasing, which would cause the
FB voltage to be less than VREF. This will cause the error
comparator to trigger ON-time period. At the end of the
ON-time period, a minimum OFF time TOFF(min) is
generated to charge BST since the FB voltage is still
below the VREF. Then, the next ON-time period is
triggered due to the low FB voltage. Therefore, the
switching frequency changes during the load transient.
363ns
TS
where Ts = 1/fSW. It is not recommended to use
MIC2164/-2/-3 with a OFF time close to TOFF(min) at the
steady state.
September 2009
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Micrel, Inc.
MIC2164/-2/-3
With the varying duty cycle and switching frequency, the
output recovery time is fast and the output voltage
deviation is small in MIC2164/-2/-3 converter.
circuitry is disabled to reduce the current consumption.
VIN should be powered up no earlier than VHSD to make
the soft-start function behavior correctly.
Current Limit
The MIC2164/-2/-3 uses the RDS(ON) of the low-side
power MOSFET to sense over-current conditions. The
lower-side MOSFET is used because it displays much
lower parasitic oscillations during switching then the
high-side MOSFET. Using the low-side MOSFET RDS(ON)
as a current sense is an excellent method for circuit
protection. This method will avoid adding cost, board
space and power losses taken by discrete current sense
resistors.
In each switching cycle of the MIC2164/-2/-3 converter,
the inductor current is sensed by monitoring the low-side
MOSFET in the OFF period. The sensed voltage is
compared with a current-limit threshold voltage VCL after
a blanking time of 150ns. If the sensed voltage is over
VCL, which is 130mV typical at 0.8V feedback voltage,
the MIC2164/-2/-3 turns off the high-side MOSFET and a
soft-start sequence is trigged. This mode of operation is
called the “hiccup mode” and its purpose is to protect the
down stream load in case of a hard short. The current
limit threshold VCL has a fold back characteristics related
to the FB voltage. Please refer to the “Typical
Characteristics” for the curve of VCL vs. FB voltage. The
circuit in Figure 4 illustrates the MIC2164/-2/-3 current
limiting circuit.
Figure 3. MIC2164/-2/-3 Load-Transient Response
Unlike the current-mode control, MIC2164/-2/-3 uses the
output voltage ripple, which is proportional to the
inductor current ripple if the ESR of the output capacitor
is large enough, to trigger an ON-time period. The
predetermined ON time makes MIC2164/-2/-3 control
loop has the advantage as the adaptive on-time mode
control. Therefore, the slope compensation, which is
necessary for the current-mode control, is not required in
the MIC2164/-2/-3 family.
The MIC2164/-2/-3 family has its own stability concern:
the FB voltage ripple should be in phase with the
inductor current ripple and large enough to be sensed by
the gm amplifier and the error comparator. The
recommended minimum FB voltage ripple is 20mV. If a
low ESR output capacitor is selected, the FB voltage
ripple may be too small to be sensed by the gm amplifier
and the error comparator. Also, the output voltage ripple
and the FB voltage ripple are not in phase with the
inductor current ripple if the ESR of the output capacitor
is very low. Therefore, the ripple injection is required for
a low ESR output capacitor. Please refer to “Ripple
Injection” subsection in “Application Information” for
more details about the ripple injection.
Figure 4. MIC2164/-2/-3 Current Limiting Circuit
Using the typical VCL value of 130mV, the current limit
value is roughly estimated as:
Soft-Start
Soft-start reduces the power supply input surge current
at startup by controlling the output voltage rise time. The
input surge appears while the output capacitor is
charged up. A slower output rise time will draw a lower
input surge current.
MIC2164/-2/-3 implements an internal digital soft-start by
making the 0.8V reference voltage VREF ramp from 0 to
100% in about 6ms with a 9.7mV step. Therefore, the
output voltage is controlled to increase slowly by a staircase VREF ramp. Once the soft-start ends, the related
September 2009
ICL ≈
130mV
R DS(ON)
For designs where the current ripple is significant
compared to the load current IOUT, or for low duty cycle
operation, calculating the current limit ICL should take
into account that one is sensing the peak inductor
current and that there is a blanking delay of
approximately 150ns.
13
M9999-090409-B
Micrel, Inc.
ICL
MIC2164/-2/-3
V
* TDLY ΔIL(pp)
130mV
=
+ OUT
−
R DS(ON)
L
2
ΔIL(pp) =
VOUT ⋅ (1 − D)
f SW ⋅L
MOSFET Gate Drive
The MIC2164/-2/-3 high-side drive circuit is designed to
switch an N-Channel MOSFET. The Block Diagram of
Figure 1 shows a bootstrap circuit, consisting of D1 (a
Schottky diode is recommended) and CBST. This circuit
supplies energy to the high-side drive circuit. Capacitor
CBST is charged while the low-side MOSFET is on and
the voltage on the LX pin is approximately 0V. When the
high-side MOSFET driver is turned on, energy from CBST
is used to turn the MOSFET on. As the high-side
MOSFET turns on, the voltage on the LX pin increases
to approximately VHSD. Diode D1 is reversed biased
and CBST floats high while continuing to keep the highside MOSFET on. The bias current of the high-side
driver is less than 10mA so a 0.1μF to 1μF is sufficient to
hold the gate voltage with minimal droop for the power
stroke (high-side switching) cycle, i.e. ΔBST = 10mA x
3.33μs/0.1μF = 333mV for MIC2164. When the low-side
MOSFET is turned back on, CBST is recharged through
D1. A small resistor RG, which is in series with CBST, can
slow down the turn-on time of the high-side N-channel
MOSFET.
The drive voltage is derived from the supply voltage VIN.
The nominal low-side gate drive voltage is VIN and the
nominal high-side gate drive voltage is approximately
VIN – VDIODE, where VDIODE is the voltage drop
across D1. An approximate 30ns delay between the
high-side and low-side driver transitions is used to
prevent current from simultaneously flowing unimpeded
through both MOSFETs.
(2)
(3)
where
VOUT = The output voltage
TDLY = Current limit blanking time, 150ns typical
ΔIL(pp) = Inductor current ripple peak-to-peak value
D = Duty Cycle
fSW = Switching frequency
The MOSFET RDS(ON) varies 30% to 40% with
temperature; therefore, it is recommended to add a 50%
margin to ICL in the above equation to avoid false current
limiting due to increased MOSFET junction temperature
rise. It is also recommended to connect LX pin directly to
the drain of the low-side MOSFET to accurately sense
the MOSFETs RDS(ON).
September 2009
14
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
For the low-side MOSFET:
Application Information
IG[low - side] (avg) = C ISS × VGS × f SW
MOSFET Selection
The MIC2164/-2/-3 controller works from power stage
input voltages of 3V to 28V and has an external 3V to
5.5V VIN to provide power to turn the external N-Channel
power MOSFETs for the high- and low-side switches.
For applications where VIN < 5V, it is necessary that the
power MOSFETs used are sub-logic level and are in full
conduction mode for VGS of 2.5V. For applications when
VIN > 5V; logic-level MOSFETs, whose operation is
specified at VGS = 4.5V must be used.
There are different criteria for choosing the high-side and
low-side MOSFETs. These differences are more
significant at lower duty cycles such as 12V to 1.8V
conversion. In such an application, the high-side
MOSFET is required to switch as quickly as possible to
minimize transition losses, whereas the low-side
MOSFET can switch slower, but must handle larger
RMS currents. When the duty cycle approaches 50%,
the current carrying capability of the high-side MOSFET
starts to become critical.
It is important to note that the on-resistance of a
MOSFET increases with increasing temperature. A 75°C
rise in junction temperature will increase the channel
resistance of the MOSFET by 50% to 75% of the
resistance specified at 25°C. This change in resistance
must be accounted for when calculating MOSFET power
dissipation and in calculating the value of current limit.
Total gate charge is the charge required to turn the
MOSFET on and off under specified operating conditions
(VDS and VGS). The gate charge is supplied by the
MIC2164/-2/-3 gate-drive circuit. At 300kHz switching
frequency and above, the gate charge can be a
significant source of power dissipation in the MIC2164/2/-3. At low output load, this power dissipation is
noticeable as a reduction in efficiency. The average
current required to drive the high-side MOSFET is:
IG[high-side] (avg) = Q G × f SW
Since the current from the gate drive comes from the
VIN, the power dissipated in the MIC2164/-2/-3 due to
gate drive is:
PGATEDRIVE = VIN .(IG[high- side] (avg) + IG[low -side] (avg)) (6)
A convenient figure of merit for switching MOSFETs is
the on resistance times the total gate charge RDS(ON) ×
QG. Lower numbers translate into higher efficiency. Low
gate-charge logic-level MOSFETs are a good choice for
use with the MIC2164/-2/-3. Also, the RDS(ON) of the lowside MOSFET will determine the current limit value.
Please refer to “Current Limit” subsection is “Functional
Description” for more details.
Parameters that are important to MOSFET switch
selection are:
•
Voltage rating
•
On-resistance
•
Total gate charge
The voltage ratings for the high-side and low-side
MOSFETs are essentially equal to the power stage input
voltage VHSD. A safety factor of 20% should be added to
the VDS(max) of the MOSFETs to account for voltage
spikes due to circuit parasitic elements.
The power dissipated in the MOSFETs is the sum of the
conduction losses during the on-time (PCONDUCTION) and
the switching losses during the period of time when the
MOSFETs turn on and off (PAC).
PSW = PCONDUCTION + PAC
(7)
2
PCONDUCTION = I SW(RMS) * R DS(ON)
(8)
PAC = PAC(off ) + PAC(on)
(9)
where:
RDS(ON) = on-resistance of the MOSFET switch
D = Duty Cycle = VOUT / VHSD
Making the assumption that the turn-on and turn-off
transition times are equal; the transition times can be
approximated by:
(4)
where:
IG[high-side](avg) = Average high-side MOSFET gate
current
QG = Total gate charge for the high-side MOSFET taken
from the manufacturer’s data sheet for VGS = VIN.
fSW = Switching Frequency
The low-side MOSFET is turned on and off at VDS = 0
because an internal body diode or external freewheeling
diode is conducting during this time. The switching loss
for the low-side MOSFET is usually negligible. Also, the
gate-drive current for the low-side MOSFET is more
accurately calculated using CISS at VDS = 0 instead of
gate charge.
September 2009
(5)
tT =
C ISS × VIN + C OSS × VHSD
IG
(10)
where:
CISS and COSS are measured at VDS = 0
IG = gate-drive current
The total high-side MOSFET switching loss is:
PAC = (VHSD + VD ) × IPK × t T × f SW
15
(11)
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Lower cost iron powder cores may be used but the
increase in core loss will reduce the efficiency of the
power supply. This is especially noticeable at low output
power. The winding resistance decreases efficiency at
the higher output current levels. The winding resistance
must be minimized although this usually comes at the
expense of a larger inductor. The power dissipated in the
inductor is equal to the sum of the core and copper
losses. At higher output loads, the core losses are
usually insignificant and can be ignored. At lower output
currents, the core losses can be a significant contributor.
Core loss information is usually available from the
magnetics vendor. Copper loss in the inductor is
calculated by the equation below:
2
PINDUCTORCu=IL(RMS) × RWINDING (16)
The resistance of the copper wire, RWINDING, increases
with the temperature. The value of the winding
resistance used should be at the operating temperature.
RWINDING = RWINDING(20°c) × (1 + 0.0042 × (TH – T20°C)) (17)
where:
TH = temperature of wire under full load
T20°C = ambient temperature
RWINDING(20°C) = room temperature winding resistance
(usually specified by the manufacturer)
where:
tT = Switching transition time
VD = Body diode drop (0.5v)
fSW = Switching Frequency
The high-side MOSFET switching losses increase with
the switching frequency and the input voltage VHSD. The
low-side MOSFET switching losses are negligible and
can be ignored for these calculations.
Inductor Selection
Values for inductance, peak, and RMS currents are
required to select the output inductor. The input and
output voltages and the inductance value determine the
peak-to-peak inductor ripple current. Generally, higher
inductance values are used with higher input voltages.
Larger peak-to-peak ripple currents will increase the
power dissipation in the inductor and MOSFETs. Larger
output ripple currents will also require more output
capacitance to smooth out the larger ripple current.
Smaller peak-to-peak ripple currents require a larger
inductance value and therefore a larger and more
expensive inductor. A good compromise between size,
loss and cost is to set the inductor ripple current to be
equal to 20% of the maximum output current. The
inductance value is calculated by the equation below.
L=
(
VOUT × VHSD(max) − VOUT
)
Output Capacitor Selection
The type of the output capacitor is usually determined by
its ESR (equivalent series resistance). Voltage and RMS
current capability are two other important factors for
selecting the output capacitor. Recommended capacitors
are tantalum, low-ESR aluminum electrolytic, OS-CON
and POSCAPS. The output capacitor’s ESR is usually
the main cause of the output ripple. The output capacitor
ESR also affects the control loop from a stability point of
view. The maximum value of ESR is calculated:
(12)
VHSD(max) × f SW × 20% × IOUT(max)
where:
fSW = switching frequency
20% = ratio of AC ripple current to DC output current
VHSD(max) = maximum power stage input voltage
The peak-to-peak inductor current ripple is:
ΔIL(PP ) =
VOUT × ( VHSD(max) − VOUT )
VHSD(max) × f SW × L
ESR COUT ≤
(13)
The peak inductor current is equal to the average output
current plus one half of the peak-to-peak inductor current
ripple.
IL(PK) = IOUT(max) + 0.5 × ΔIL(PP)
(14)
2
ΔIL(PP)
12
2
ΔIL(PP)
⎞
⎛
2
⎟ + ΔIL(PP) ⋅ ESR C
ΔVOUT(pp) = ⎜⎜
OUT
⎟
C
f
8
⋅
⋅
⎠
⎝ OUT SW
(19)
2
(15)
Maximizing efficiency requires the proper selection of
core material and minimizing the winding resistance. The
high frequency operation of the MIC2164/-2/-3 requires
the use of ferrite materials for all but the most cost
sensitive applications.
September 2009
(18)
ΔIL(PP)
where:
ΔVOUT(pp) = peak-to-peak output voltage ripple
ΔIL(PP) = peak-to-peak inductor current ripple
The total output ripple is a combination of the ESR and
output capacitance. The total ripple is calculated below:
The RMS inductor current is used to calculate the I R
losses in the inductor.
IL(RMS) = IOUT(max)2 +
ΔVOUT(pp)
16
(
)
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
output voltage, as shown in Figure 5.
Where:
D = duty cycle
COUT = output capacitance value
fSW = switching frequency
As described in the “Theory of Operation” subsection in
“Functional Description”, MIC2164/-2/-3 requires at least
20mV peak-to-peak ripple at the FB pin to make the gm
amplifier and the error comparator to behavior properly.
Also, the output voltage ripple should be in phase with
the inductor current. Therefore, the output voltage ripple
caused by the output capacitor COUT should be much
smaller than the ripple caused by the output capacitor
ESR. If low ESR capacitors are selected as the output
capacitors, such as ceramic capacitors, a ripple injection
method is applied to provide the enough FB voltage
ripples. Please refer to the “Ripple Injection” subsection
for more details.
The voltage rating of the capacitor should be twice the
output voltage for a tantalum and 20% greater for
aluminum electrolytic or OS-CON. The output capacitor
RMS current is calculated below:
ICOUT (RMS) =
ΔIL(PP)
Figure 5. Voltage-Divider Configuration
The output voltage is determined by the equation:
R1
)
(25)
R2
where VREF = 0.8V. A typical value of R1 can be between
3kΩ and 10kΩ. If R1 is too large, it may allow noise to be
introduced into the voltage feedback loop. If R1 is too
small in value, it will decrease the efficiency of the power
supply, especially at light loads. Once R1 is selected, R2
can be calculated using:
VOUT = VREF ⋅ (1 +
(20)
12
R2 =
The power dissipated in the output capacitor is:
2
PDISS(COUT ) = I COUT (RMS) ⋅ ESR COUT
(21)
ID(avg) = I OUT ⋅ 2 ⋅ 30ns ⋅ f SW
The power dissipated in the input capacitor is:
2
PDISS(CIN) = ICIN(RMS) ×ESRCIN
VDIODE(rrm) = VHSD
The power dissipated by the Schottky diode is:
PDIODE = ID(avg) × VF
(22)
(28)
where, VF = forward voltage at the peak diode current.
The external Schottky diode is not necessary for the
circuit operation since the low-side MOSFET contains a
parasitic body diode. The external diode will improve
efficiency and decrease the high frequency noise. If the
MOSFET body diode is used, it must be rated to handle
the peak and average current. The body diode has a
relatively slow reverse recovery time and a relatively
high forward voltage drop. The power lost in the diode is
proportional to the forward voltage drop of the diode. As
the high-side MOSFET starts to turn on, the body diode
(23)
(24)
Voltage Setting Components
The MIC2164/-2/-3 requires two resistors to set the
September 2009
(27)
The reverse voltage requirement of the diode is:
The input capacitor must be rated for the input current
ripple. The RMS value of input capacitor current is
determined at the maximum output current. Assuming
the peak-to-peak inductor current ripple is low:
ICIN(RMS) ≈ IOUT(max) × D × (1 − D )
(26)
External Schottky Diode (Optional)
An external freewheeling diode, which is not necessary,
can be used to keep the inductor current flow continuous
while both MOSFETs are turned off. This dead time
prevents current from flowing unimpeded through both
MOSFETs and is typically 30ns. The diode conducts
twice during each switching cycle. Although the average
current through this diode is small, the diode must be
able to handle the peak current.
Input Capacitor Selection
The input capacitor for the power stage input VHSD
should be selected for ripple current rating and voltage
rating. Tantalum input capacitors may fail when
subjected to high inrush currents, caused by turning the
input supply on. A tantalum input capacitor’s voltage
rating should be at least two times the maximum input
voltage to maximize reliability. Aluminum electrolytic,
OS-CON, and multilayer polymer film capacitors can
handle the higher inrush currents without voltage derating. The input voltage ripple will primarily depend
upon the input capacitor’s ESR. The peak input current
is equal to the peak inductor current, so:
ΔVIN = IL(PK ) × ESR CIN
VREF ⋅ R1
VOUT − VREF
17
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
becomes a short circuit for the reverse recovery period,
dissipating additional power. The diode recovery and the
circuit inductance will cause ringing during the high-side
MOSFET turn-on.
An external Schottky diode conducts at a lower forward
voltage preventing the body diode in the MOSFET from
turning on. The lower forward voltage drop dissipates
less power than the body diode. The lack of a reverse
recovery mechanism in a Schottky diode causes less
ringing and less power loss. Depending upon the circuit
components and operating conditions, an external
Schottky diode will give a 1/2% to 1% improvement in
efficiency.
Figure 6a. Enough Ripple at FB
Ripple Injection
The minimum FB voltage ripple requested by the
MIC2164/-2/-3 gm amplifier and error comparator is
20mV. However, the output voltage ripple is generally
designed as 1% to 2% of the output voltage. For a low
output voltage, such as 1V output, the output voltage
ripple is only 10mV to 20mV, and the FB voltage ripple is
less than 20mV. If the FB voltage ripple is so small that
the gm amplifier and error comparator could not sense it,
the MIC2164/-2/-3 will lose control and the output
voltage is not regulated. In order to have some amount
of FB voltage ripple, the ripple injection method is
applied for low output voltage ripple applications.
The applications are divided into three situations
according to the amount of the FB voltage ripple:
1) Enough ripple at the FB voltage due to the large ESR
of the output capacitors.
As shown in Figure 6a, the converter is stable without
any adding in this situation. The FB voltage ripple is:
Figure 6b. Inadequate Ripple at FB
Figure 6c. Invisible Ripple at FB
In this situation, the output voltage ripple is less than
20mV. Therefore, additional ripple is injected into the FB
pin from the switching node LX via a resistor Rinj and a
capacitor Cinj, as shown in Figure 6c. The injected ripple
is:
1
ΔVFB(pp) = VHSD × K div × D × (1- D) ×
(31)
f SW × τ
R2
⋅ ESR COUT ⋅ ΔIL (pp)
(29)
R1 + R2
is the peak-to-peak value of the inductor
ΔVFB(pp) =
where ΔIL(pp)
current ripple.
2) Inadequate ripple at the FB voltage due to the small
ESR of the output capacitors.
The output voltage ripple is fed into the FB pin through a
feedforward capacitor Cff in this situation, as shown in
Figure 6b. The typical Cff value is between 1nF to
100nF. With the feedforward capacitor, the FB voltage
ripple is very close to the output voltage ripple:
ΔVFB(pp) ≈ ESR ⋅ ΔIL (pp)
K div =
(32)
where
VHSD = Power stage input voltage at HSD pin
D = Duty Cycle
fSW = switching frequency
τ = (R1// R2 // Rinj) ⋅ Cff
(30)
In the formula (31) and (32), it is assumed that the time
constant associated with Cff must be much greater than
the switching period:
3) Invisible ripple at the FB voltage due to the very low
ESR of the output capacitors.
September 2009
R1//R2
Rinj + R1//R2
18
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Step 2. Select Rinj according to the expected feedback
voltage ripple. According to the equation (32),
1
T
= << 1
fsw × τ τ
K div =
If the voltage divider resistors R1 and R2 are in the kΩ
range, a Cff of 1nF to 100nF can easily satisfy the large
time constant consumption. Also, a 100nF injection
capacitor Cinj is used in order to be considered as short
for a wide range of the frequencies.
The process of sizing the ripple injection resistor and
capacitors is:
Step 1. Select Cff to feed all output ripples into the
feedback pin and make sure the large time constant
assumption is satisfied. Typical choice of Cff is 1nF to
100nF if R1 and R2 are in kΩ range.
September 2009
ΔVFB(pp )
VHSD
⋅
f SW ⋅ τ
D ⋅ (1 − D)
(33)
Then the value of Rinj is obtained as:
R inj = (R1 // R2) ⋅ (
1
K div
− 1)
(34)
Step 3. Select Cinj as 100nF, which could be considered
as short for a wide range of the frequencies.
19
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
capacitor is very critical. Connections must be made
with wide trace.
PCB Layout Guideline
Warning!!! To minimize EMI and output noise, follow
these layout recommendations.
PCB Layout is critical to achieve reliable, stable and
efficient performance. A ground plane is required to
control EMI and minimize the inductance in power,
signal and return paths.
The following guidelines should be followed to insure
proper operation of the MIC2164/-2/-3 converter.
Inductor
IC
•
Keep the inductor connection to the switch node
(LX) short.
•
Do not route any digital lines underneath or close to
the inductor.
•
Keep the switch node (LX) away from the feedback
(FB) pin.
•
The LX pin should be connected directly to the drain
of the low-side MOSFET to accurate sense the
voltage across the low-side MOSFET.
To minimize noise, place a ground plane underneath
the inductor.
•
Place the IC and MOSFETs close to the point of
load (POL).
•
Use fat traces to route the input and output power
lines.
•
•
Signal and power grounds should be kept separate
and connected at only one location.
Output Capacitor
•
Use a wide trace to connect the output capacitor
ground terminal to the input capacitor ground
terminal.
•
Phase margin will change as the output capacitor
value and ESR changes. Contact the factory if the
output capacitor is different from what is shown in
the BOM.
•
The feedback trace should be separate from the
power trace and connected as close as possible to
the output capacitor. Sensing a long high current
load trace can degrade the DC load regulation.
Input Capacitor
•
Place the HSD input capacitor next.
•
Place the HSD input capacitors on the same side of
the board and as close to the MOSFETs as
possible.
•
Keep both the HSD and PGND connections short.
•
Place several vias to the ground plane close to the
HSD input capacitor ground terminal.
•
Use either X7R or X5R dielectric input capacitors.
Do not use Y5V or Z5U type capacitors.
•
Do not replace the ceramic input capacitor with any
other type of capacitor. Any type of capacitor can be
placed in parallel with the input capacitor.
•
If a Tantalum input capacitor is placed in parallel
with the input capacitor, it must be recommended for
switching regulator applications and the operating
voltage must be derated by 50%.
•
Schottky Diode (Optional)
In “Hot-Plug” applications, a Tantalum or Electrolytic
bypass capacitor must be used to limit the overvoltage spike seen on the input supply with power is
suddenly applied.
•
An additional Tantalum or Electrolytic bypass input
capacitor of 22uF or higher is required at the input
power connection.
•
The 1µF and 0.1µF capacitors, which connect to the
VIN terminal, must be located right at the IC. The VIN
terminal is very noise sensitive and placement of the
September 2009
•
Place the Schottky diode on the same side of the
board as the MOSFETs and HSD input capacitor.
•
The connection from the Schottky diode’s Anode to
the input capacitors ground terminal must be as
short as possible.
•
The diode’s Cathode connection to the switch node
(LX) must be keep as short as possible.
RC Snubber
•
20
Place the RC snubber on the same side of the board
and as close to the MOSFETs as possible.
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Evaluation Board Schematics
Figure 7. Schematic of MIC2164 20A Evaluation Board
September 2009
21
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Bill of Materials
Item
C1, C8, C17, C19
Part Number
06035C104KAT
0805ZD225MAT
C2
GRM216R61A225ME24D
C2012X5R1A225K/0.85
C3
C4, C5, C6
C9
C10
C11
C12
222215095001
1210YD226MAT
GRM32ER61C226ME20L
0805ZD105KAT
GRM219R61A105MA01D
06035C223KAT
Manufacturer
AVX
MuRata
TDK
(4)
Vishay
AVX
MuRata
AVX
MuRata
AVX
Sanyo
12106D107MAT
AVX
muRata
06035C102KAT
AVX
D1
SD103BWS
FDS6699S
Q2, Q3
FDS7766S
CMPDM7002A
2N7002E-T1-E3
2.2µF Ceramic Capacitor, X5R, Size 0805, 10V
1
220µF Aluminum Capacitor, SMD, 35V
1
22µF Ceramic Capacitor, X5R, Size 1210, 16V
3
1µF Ceramic Capacitor, X5R, Size 0805, 10V
1
22nF Ceramic Capacitor, X7R, 0603, 50V
(5)
C15
Q1
4
22nF Ceramic Capacitor, X7R, Size 0603, 50V
16ME1000WGL
TDK
Q5
(2)
MuRata
CDEP147NP-1R5M
0.1µF Ceramic Capacitor, X7R, Size 0603, 50V
(3)
C3225X5R0J107M
L1
Qty
AVX
GRM188R71H223MA01D
GRM32ER60J107ME20L
Description
(1)
Vishay
Sumida
1
1000µF Aluminum Capacitor, 16V
1
100µF Ceramic Capacitor, X5R, Size 1210, 6.3V
1
1nF Ceramic Capacitor, X7R, 0603, 50V
1
Small Signal Schottky Diode
1
(6)
1.5µH Inductor, 27.2A Saturation Current
1
(7)
30V N-Channel MOSFET 4.5mΩ Rds(on) @ 4.5V
1
30V N-Channel MOSFET 6.5mΩ Rds(on) @ 4.5V
2
Signal MOSFET, 60V
1
2.21Ω Resistor, Size 0603, 1%
1
Fairchild
Fairchild
(8)
Central Semiconductor
Vishay
CRCW06032R21FKEY3
R2, R7
CRCW06030000FKEY3
Vishay-Dale
0Ω Resistor, Size 0603, 1%
2
R3
CRCW06034992FKEY3
Vishay-Dale
49.9kΩ Resistor, Size 0603, 1%
1
R5
CRCW06031R21FKEY3
Vishay-Dale
1.21Ω Resistor, Size 0603, 1%
1
R6, R9
CRCW06031002FKEY3
Vishay-Dale
10kΩ Resistor, Size 0603, 1%
2
R12
CRCW06034751FKEY3
Vishay-Dale
4.75kΩ Resistor, Size 0603, 1%
1
R13
CRCW06038061FKEY3
Vishay-Dale
8.06kΩ Resistor, Size 0603, 1%
1
R14
CRCW06034022FKEY3
Vishay-Dale
40.2kΩ Resistor, Size 0603, 1%
1
R15
CRCW06033241FKEY3
Vishay-Dale
3.24kΩ Resistor, Size 0603, 1%
1
U1
MIC2164YMM
Micrel. Inc.
300kHz Buck Controller
1
U2
MIC5233-5.0YM5
LDO
1
September 2009
Vishay-Dale
(4)
R1
(9)
Micrel. Inc.
22
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Notes:
1.
AVX: www.avx.com
2.
MuRata: www.murata.com
3.
TDK: www.tdk.com
4.
Vishay: www.vishay.com
5.
Sanyo: www.sanyo.com
6.
Sumida: www.sumida.com
7.
Fairchild: www.fairchildsemi.com
8.
Central Semiconductor: www.centralsemi.com
9.
Micrel, Inc: www.micrel.com
September 2009
23
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
PCB Layout
Figure 8. MIC2164 20A Evaluation Board Top Layer
Figure 9. MIC2164 20A Evaluation Board Bottom Layer
September 2009
24
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Figure 10. MIC2164 20A Evaluation Board Mid-Layer 1
Figure 11. MIC2164 20A Evaluation Board Mid-Layer 2
September 2009
25
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Application Schematics and Bill of Materials
Figure 12. MIC2164 12V to 3.3V @ 20A Buck Converter
Bill of Materials (MIC2164 12V to 3.3V @ 20A)
Item
Part Number
C1, C8, C17, C19
06035C104KAT
C2
0805ZD225MAT
C3
222215095001
Manufacturer
(1)
AVX
AVX
(2)
Vishay
Description
Qty
0.1µF Ceramic Capacitor, X7R, Size 0603, 50V
4
2.2µF Ceramic Capacitor, X5R, Size 0805, 10V
1
220µF Aluminum Capacitor, SMD, 35V
1
C4, C5, C6
1210YD226MAT
AVX
22µF Ceramic Capacitor, X5R, Size 1210, 16V
3
C9
0805ZD105KAT
AVX
1µF Ceramic Capacitor, X5R, Size 0805, 10V
1
C10
06035C223KAT
AVX
22nF Ceramic Capacitor, X7R, Size 0603, 50V
1
1000µF Aluminum Capacitor, 16V
1
(3)
C11
16ME1000WGL
Sanyo
C12
12106D107MAT
AVX
100µF Ceramic Capacitor, X5R, Size 1210, 6.3V
1
C15
06035C102KAT
AVX
1nF Ceramic Capacitor, X7R, Size 0603, 50V
1
D1
SD103BWS
L1
Q1
September 2009
CDEP147NP-1R5M
FDS6699S
Vishay
Sumida
Small Signal Schottky Diode
1
(4)
1.5µH Inductor, 27.2A Saturation Current
1
(5)
30V N-Channel MOSFET 4.5mΩ Rds(on) @ 4.5V
1
Fairchild
26
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Bill of Materials (MIC2164 12V to 3.3V @ 20A)
Item
Part Number
Manufacturer
Qty
Q2, Q3
FDS7766S
30V N-Channel MOSFET 6.5mΩ Rds(on) @ 4.5V
2
R1
CRCW06032R21FKEY3
Vishay Dale
2.21Ω Resistor, Size 0603, 1%
1
R5
CRCW06031R21FKEY3
Vishay Dale
1.21Ω Resistor, Size 0603, 1%
1
R6, R9
CRCW06031002FKEY3
Vishay Dale
10k Resistor, Size 0603, 1%
2
R15
CRCW06033241FKEY3
Vishay Dale
3.24k Resistor, Size 0603 1%
1
300kHz Buck Controller
1
LDO
1
U1
MIC2164YMM
U2
MIC5233-5.0YM5
Fairchild
Description
(6)
Micrel. Inc.
Micrel. Inc.
Notes:
1.
AVX: www.avx.com
2.
Vishay: www.vishay.com
3.
Sanyo: www.sanyo.com
4.
Sumida: www.sumida.com
5.
6.
Fairchild: www.fairchildsemi.com
Micrel, Inc: www.micrel.com.
September 2009
27
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Figure 13. MIC2164 12V to 1.8V @ 10A Buck Converter
Bill of Materials (MIC2164 12V to 1.8V @ 10A)
Item
Part Number
C1, C8, C17, C19
06035C104KAT
C2
0805ZD225MAT
Manufacturer
(1)
AVX
AVX
(2)
Qty
0.1µF Ceramic Capacitor, X7R, Size 0603, 50V
4
2.2µF Ceramic Capacitor, X5R, Size 0805, 10V
1
220µF Aluminum Capacitor, SMD, 35V
1
C3
222215095001
C4, C5
1210YD106MAT
AVX
10µF Ceramic Capacitor, X5R, Size 1210, 16V
2
C9
0805ZD105KAT
AVX
1µF Ceramic Capacitor, X5R, Size 0805, 10V
1
C10
06035C223KAT
AVX
22nF Ceramic Capacitor, X7R, Size 0603, 50V
1
560µF OSCON Capacitor, 6.3V
1
100µF Ceramic Capacitor, X5R, Size 1210, 6.3V
1
C11
6SEPC560MX
C12
12106D107MAT
C15
06035C102KAT
D1
SD103BWS
L1
CDEP105-2R0MC-32
Q1, Q2
September 2009
FDS7764A
Vishay
Description
Sanyo
(3)
AVX
AVX
1nF Ceramic Capacitor, X7R, Size 0603, 50V
1
Small Signal Schottky Diode
1
(4)
2.0µH Inductor, 15.8A Saturation Current
1
(5)
30V N-Channel MOSFET 7.5mΩ Rds(on) @ 4.5V
2
Vishay
Sumida
Fairchild
28
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Bill of Materials (MIC2164 12V to 1.8V @ 10A)
Item
Part Number
Manufacturer
Description
Qty
R1
CRCW06032R21FKEY3
Vishay Dale
2.21Ω Resistor, Size 0603, 1%
1
R5
CRCW06031R21FKEY3
Vishay Dale
1.21Ω Resistor, Size 0603, 1%
1
R6, R9
CRCW06031002FKEY3
Vishay Dale
10k Resistor, Size 0603, 1%
2
R15
CRCW06038061FKEY3
Vishay Dale
8.06k Resistor, Size 0603, 1%
1
300kHz Buck Controller
1
LDO
1
U1
MIC2164YMM
U2
MIC5233-5.0YM5
(6)
Micrel. Inc.
Micrel. Inc.
Notes:
1.
AVX: www.avx.com
2.
Vishay: www.vishay.com
3.
Sanyo: www.sanyo.com
4.
Sumida: www.sumida.com
5.
6.
Fairchild: www.fairchildsemi.com
Micrel, Inc: www.micrel.com.
September 2009
29
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Figure 14. MIC2164 12V to 1.0V @ 5A Buck Converter
Bill of Materials (MIC2164 12V to 1.0V @ 5A)
Item
Part Number
C1, C8, C17, C19
06035C104KAT
C2
0805ZD225MAT
Manufacturer
(1)
AVX
AVX
(2)
Vishay
Description
Qty
0.1µF Ceramic Capacitor, X7R, Size 0603, 50V
4
2.2µF Ceramic Capacitor, X5R, Size 0805, 10V
1
220µF Aluminum Capacitor, SMD, 35V
1
C3
222215095001
C4
1210YD106MAT
AVX
10µF Ceramic Capacitor, X5R, Size 1210, 16V
1
C9
0805ZD105KAT
AVX
1µF Ceramic Capacitor, X5R, Size 0805, 10V
1
C10
06035C223KAT
AVX
22nF Ceramic Capacitor, X7R, Size 0603, 50V
1
C11, C12, C13
12106D107MAT
AVX
100µF Ceramic Capacitor, X5R, Size 1210, 6.3V
3
C15
06035C102KAT
D1
SD103BWS
AVX
Vishay
(4)
L1
CDRH104RNP-3R8
Sumida
Q1
FDS6910
Fairchild
R1
CRCW06032R21FKEY3
R5
CRCW06031R21FKEY3
September 2009
(5)
Vishay Dale
(2)
Vishay Dale
30
1nF Ceramic Capacitor, X7R, Size 0603, 50V
1
Small Signal Schottky Diode
1
3.8µH Inductor, 6A Saturation Current
1
Dual 30V N-Channel MOSFET 17mΩ Rds(on) @
4.5V
1
2.21Ω Resistor, Size 0603, 1%
1
1.21Ω Resistor, Size 0603, 1%
1
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Bill of Materials (MIC2164 12V to 1.0V @ 5A)
Item
Part Number
R6, R9
CRCW06031002FKEY3
Vishay Dale
10k Resistor, Size 0603, 1%
2
R15
CRCW06034022FKEY3
Vishay Dale
40.2k Resistor, Size 0603, 1%
1
300kHz Buck Controller
1
LDO
1
U1
MIC2164YMM
U2
MIC5233-5.0YM5
Manufacturer
(6)
Micrel. Inc.
Micrel. Inc.
Description
Qty
Notes:
1.
AVX: www.avx.com
2.
Vishay: www.vishay.com
3.
Sanyo: www.sanyo.com
4.
Sumida: www.sumida.com
5.
Fairchild: www.fairchildsemi.com
6.
Micrel, Inc: www.micrel.com
September 2009
31
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Figure 15. MIC2164-2 12V to 3.3V @ 15A Buck Converter
Bill of Materials (MIC2164-2 12V to 3.3V @ 15A)
Item
Part Number
C1, C8, C17, C19
06035C104KAT
C2
0805ZD225MAT
Manufacturer
(1)
AVX
AVX
(2)
Vishay
Description
Qty
0.1µF Ceramic Capacitor, X7R, Size 0603, 50V
4
2.2µF Ceramic Capacitor, X5R, Size 0805, 10V
1
220µF Aluminum Capacitor, SMD, 35V
1
C3
222215095001
C4, C5
1210YD226MAT
AVX
22µF Ceramic Capacitor, X5R, Size 1210, 16V
2
C9
0805ZD105KAT
AVX
1µF Ceramic Capacitor, X5R, Size 0805, 10V
1
C10
06035C472KAT
AVX
4.7nF Ceramic Capacitor, X7R, Size 0603, 50V
1
(3)
C11
16ME1000WGL
Sanyo
1000µF Aluminum Capacitor, 16V
1
C12
12106D107MAT
AVX
100µF Ceramic Capacitor, X5R, Size 1210, 6.3V
1
C15
06035C102KAT
AVX
1nF Ceramic Capacitor, X7R, Size 0603, 50V
1
D1
SD103BWS
Small Signal Schottky Diode
1
L1
HCP1305-1R0
1.0µH Inductor, 29A DC Current
1
Q1
FDS8672S
30V N-Channel MOSFET 7.0mΩ Rds(on) @ 4.5V
1
September 2009
Vishay
Cooper
(4)
Bussmann
Fairchild
(5)
32
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Bill of Materials (MIC2164-2 12V to 3.3V @ 15A)
Item
Part Number
Manufacturer
Qty
Q2, Q3
FDS8874
30V N-Channel MOSFET 7.0mΩ Rds(on) @ 4.5V
2
R1
CRCW06032R21FKEY3
Vishay Dale
2.21Ω Resistor, Size 0603, 1%
1
R5
CRCW06031R21FKEY3
Vishay Dale
1.21Ω Resistor, Size 0603, 1%
1
R6
CRCW06031002FKEY3
Vishay Dale
10k Resistor, Size 0603, 1%
1
R6
CRCW06034021FKEY3
Vishay Dale
4.02k Resistor, Size 0603, 1%
1
R15
CRCW06033241FKEY3
Vishay Dale
3.24k Resistor, Size 0603 1%
1
600kHz Buck Controller
1
LDO
1
U1
MIC2164-2YMM
U2
MIC5233-5.0YM5
Fairchild
Description
(6)
Micrel. Inc.
Micrel. Inc.
Notes:
1.
AVX: www.avx.com
2.
Vishay: www.vishay.com
3.
Sanyo: www.sanyo.com
4.
Cooper Bussmann: www.cooperbussmann.com
5.
6.
Fairchild: www.fairchildsemi.com
Micrel, Inc: www.micrel.com.
September 2009
33
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Figure 16. MIC2164-3 12V to 1.8V @ 10A Buck Converter
Bill of Materials (MIC2164-3 12V to 1.8V @ 10A)
Item
Part Number
C1, C8, C17, C19
06035C104KAT
C2
0805ZD225MAT
C3
222215095001
C4
1210YD106MAT
C9
C10
Manufacturer
(1)
AVX
Description
Qty
0.1µF Ceramic Capacitor, X7R, Size 0603, 50V
AVX
4
2.2µF Ceramic Capacitor, X5R, Size 0805, 10V
1
220µF Aluminum Capacitor, SMD, 35V
1
AVX
10µF Ceramic Capacitor, X5R, Size 1210, 16V
1
0805ZD105KAT
AVX
1µF Ceramic Capacitor, X5R, Size 0805, 10V
1
06035C222KAT
AVX
2.2nF Ceramic Capacitor, X7R, Size 0603, 50V
1
560µF OSCON Capacitor, 6.3V
1
(2)
Vishay
C11
6SEPC560MX
C12
12106D107MAT
AVX
100µF Ceramic Capacitor, X5R, Size 1210, 6.3V
1
C15
06035C102KAT
AVX
1nF Ceramic Capacitor, X7R, Size 0603, 50V
1
D1
SD103BWS
Small Signal Schottky Diode
1
L1
HCF1305-1R0
1.0µH Inductor, 20A Saturation Current
1
Q1, Q2
FDS8672S
30V N-Channel MOSFET 7.0mΩ Rds(on) @ 4.5V
2
R1
CRCW06032R21FKEY3
2.21Ω Resistor, Size 0603, 1%
1
September 2009
Sanyo
(3)
Vishay
Cooper
(4)
Bussmann
Fairchild
(5)
Vishay Dale
34
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Bill of Materials (MIC2164-3 12V to 1.8V @ 10A)
Item
Part Number
R5
CRCW06031R21FKEY3
Vishay Dale
1.21Ω Resistor, Size 0603, 1%
1
R6
CRCW06031002FKEY3
Vishay Dale
10k Resistor, Size 0603, 1%
1
R9
CRCW06032001FKEY3
Vishay Dale
2k Resistor, Size 0603, 1%
1
R15
CRCW06038061FKEY3
Vishay Dale
8.06k Resistor, Size 0603, 1%
1
1MHz Buck Controller
1
LDO
1
U1
MIC2164-3YMM
U2
MIC5233-5.0YM5
Manufacturer
(6)
Micrel. Inc.
Micrel. Inc.
Description
Qty
Notes:
1.
AVX: www.avx.com
2.
Vishay: www.vishay.com
3.
Sanyo: www.sanyo.com
4.
Cooper: www.cooperbussmann.com
5.
6.
Fairchild: www.fairchildsemi.com
Micrel, Inc: www.micrel.com.
September 2009
35
M9999-090409-B
Micrel, Inc.
MIC2164/-2/-3
Package Information
10-Pin MSOP (MM)
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its
use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product
can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant
into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A
Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully
indemnify Micrel for any damages resulting from such use or sale.
© 2009 Micrel, Incorporated.
September 2009
36
M9999-090409-B