Intersil ISL6312CRZ-T Four-phase buck pwm controller with integrated mosfet drivers for intel vr10,vr11, and amd application Datasheet

ISL6312
®
Data Sheet
February 25, 2010
Four-Phase Buck PWM Controller with
Integrated MOSFET Drivers for Intel VR10,
VR11, and AMD Applications
The ISL6312 four-phase PWM control IC provides a
precision voltage regulation system for advanced
microprocessors. The integration of power MOSFET drivers
into the controller IC marks a departure from the separate
PWM controller and driver configuration of previous
multiphase product families. By reducing the number of
external parts, this integration is optimized for a cost and
space saving power management solution.
One outstanding feature of this controller IC is its
multi-processor compatibility, allowing it to work with both Intel
and AMD microprocessors. Included are programmable VID
codes for Intel VR10, VR11, as well as AMD DAC tables. A
unity gain, differential amplifier is provided for remote voltage
sensing, compensating for any potential difference between
remote and local grounds. The output voltage can also be
positively or negatively offset through the use of a single
external resistor.
The ISL6312 also includes advanced control loop features
for optimal transient response to load apply and removal.
One of these features is highly accurate, fully differential,
continuous DCR current sensing for load line programming
and channel current balance. Active Pulse Positioning (APP)
modulation is another unique feature, allowing for quicker
initial response to high di/dt load transients.
This controller also allows the user the flexibility to choose
between PHASE detect or LGATE detect adaptive dead time
schemes. This ability allows the ISL6312 to be used in a
multitude of applications where either scheme is required.
Protection features of this controller IC include a set of
sophisticated overvoltage, undervoltage, and overcurrent
protection. Furthermore, the ISL6312 includes protection
against an open circuit on the remote sensing inputs.
Combined, these features provide advanced protection for the
microprocessor and power system.
FN9289.5
Features
• Integrated Multiphase Power Conversion
- 2- or 3-Phase Operation with Internal Drivers
- 4-Phase Operation with External PWM Driver Signal
• Precision Core Voltage Regulation
- Differential Remote Voltage Sensing
- ±0.5% System Accuracy Over-Temperature
- Adjustable Reference-Voltage Offset
• Optimal Transient Response
- Active Pulse Positioning (APP) Modulation
- Adaptive Phase Alignment (APA)
• Fully Differential, Continuous DCR Current Sensing
- Accurate Load Line Programming
- Precision Channel Current Balancing
• User Selectable Adaptive Dead Time Scheme
- PHASE Detect or LGATE Detect for Application Flexibility
• Variable Gate Drive Bias: 5V to 12V
• Multi-Processor Compatible
- Intel VR10 and VR11 Modes of Operation
- AMD Mode of Operation
• Microprocessor Voltage Identification Inputs
- 8-bit DAC
- Selectable between Intel’s Extended VR10, VR11, AMD
5-bit, and AMD 6-bit DAC Tables
- Dynamic VID Technology
• Overcurrent Protection
• Multi-Tiered Overvoltage Protection
• Digital Soft-Start
• Selectable Operation Frequency up to 1.5MHz Per Phase
• Pb-Free (RoHS Compliant)
Ordering Information
PART NUMBER
(Note 1)
PART
MARKING
TEMP.
(°C)
PACKAGE
(Pb-Free)
PKG.
DWG. #
ISL6312CRZ
ISL6312 CRZ 0 to +70 48 Ld 7x7 QFN
L48.7x7
ISL6312CRZ-T
(Note 2)
ISL6312 CRZ 0 to +70 48 Ld 7x7 QFN
L48.7x7
ISL6312CRZ-TK ISL6312 CRZ 0 to +70 48 Ld 7x7 QFN
(Note 2)
L48.7x7
ISL6312IRZ
ISL6312 IRZ -40 to +85 48 Ld 7x7 QFN
L48.7x7
ISL6312IRZ-T
(Note 2)
ISL6312 IRZ -40 to +85 48 Ld 7x7 QFN
L48.7x7
NOTES:
1. These Intersil Pb-free plastic packaged products employ special Pbfree material sets, molding compounds/die attach materials, and
100% matte tin plate plus anneal (e3 termination finish, which is
RoHS compliant and compatible with both SnPb and Pb-free
soldering operations). Intersil Pb-free products are MSL classified at
Pb-free peak reflow temperatures that meet or exceed the Pb-free
requirements of IPC/JEDEC J STD-020.
2. Please refer to TB347 for details on reel specifications.
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright Intersil Americas Inc. 2006, 2007, 2010. All Rights Reserved
All other trademarks mentioned are the property of their respective owners.
ISL6312
Pinout
VID5
VID6
VID7
FS
ISEN3+
ISEN3-
PVCC3
LGATE3
BOOT3
UGATE3
PHASE3
PGOOD
ISL6312
(48 LD QFN)
TOP VIEW
48
47
46
45
44
43
42
41
40
39
38
37
VID4
1
36 EN
VID3
2
35 ISEN1+
VID2
3
34 ISEN1-
VID1
4
33 PHASE1
VID0
5
32 UGATE1
VRSEL
6
31 BOOT1
49
GND
DRSEL
7
30 LGATE1
OVPSEL
8
29 PVCC1_2
SS
9
28 LGATE2
16
17
18
19
20
21
22
23
24
ISEN2-
ISEN4+
ISEN4-
EN_PH4
PWM4
15
ISEN2+
14
VSEN
13
VDIFF
25 PHASE2
RGND
26 UGATE2
IDROOP
REF 11
OFS 12
FB
27 BOOT2
COMP
VCC 10
ISL6312 Integrated Driver Block Diagram
PVCC
DRSEL
BOOT
UGATE
20kΩ
PWM
GATE
CONTROL
LOGIC
SOFT-START
SHOOTTHROUGH
PROTECTION
PHASE
10kΩ
AND
FAULT LOGIC
LGATE
2
FN9289.5
February 25, 2010
ISL6312
Block Diagram
PGOOD
EN
SS
OPEN SENSE
LINE PREVENTION
0.85V
VSEN
x1
VCC
POWER-ON
RESET
RGND
VDIFF
PVCC1_2
SOFT-START
AND
UNDERVOLTAGE
DETECTION
LOGIC
FAULT LOGIC
BOOT1
MOSFET
DRIVER
OVERVOLTAGE
DETECTION
LOGIC
UGATE1
PHASE1
0.2V
LGATE1
LOAD APPLY
TRANSIENT
ENHANCEMENT
OVPSEL
DRSEL
CLOCK AND
MODULATOR
WAVEFORM
GENERATOR
VRSEL
FS
MODE/DAC
SELECT
BOOT2
MOSFET
DRIVER
PWM1
∑
VID7
VID6
VID4
VID3
PHASE2
LGATE2
OC
VID5
UGATE2
PWM2
DYNAMIC
VID
D/A
∑
VID2
I_TRIP
PWM3
VID1
∑
VID0
PH4 POR/
DETECT
PVCC3
PWM4
REF
E/A
EN_PH4
CHANNEL
DETECT
∑
FB
BOOT3
COMP
OFS
MOSFET
DRIVER
OFFSET
CHANNEL
CURRENT
BALANCE
I_AVG
1
N
UGATE3
PHASE3
LGATE3
I_AVG
IDROOP
∑
PWM4
SIGNAL
LOGIC
CH1
CURRENT
SENSE
CH2
CURRENT
SENSE
CH3
CURRENT
SENSE
CH4
CURRENT
SENSE
ISEN1- ISEN1+ ISEN2- ISEN2+ ISEN3- ISEN3+ ISEN4- ISEN4+
3
PWM4
GND
FN9289.5
February 25, 2010
ISL6312
Typical Application - ISL6312 (4-Phase)
+12V
FB
IDROOP
VDIFF
COMP
VSEN
BOOT1
RGND
UGATE1
+5V
PHASE1
VCC
LGATE1
ISEN1ISEN1+
OFS
FS
+12V
REF
PVCC1_2
BOOT2
SS
UGATE2
PHASE2
LGATE2
LOAD
OVPSEL
ISL6312
ISEN2ISEN2+
+12V
VID7
VID6
VID5
VID4
VID3
VID2
VID1
VID0
PVCC3
BOOT3
UGATE3
PHASE3
VRSEL
PGOOD
LGATE3
+12V
ISEN3ISEN3+
+12V
+12V
EN
EN_PH4
DRSEL
BOOT
VCC UGATE
PVCC
PHASE
ISL6612
LGATE
PWM4
GND
PWM
GND
ISEN4ISEN4+
4
FN9289.5
February 25, 2010
ISL6312
Typical Application - ISL6312 with NTC Thermal Compensation (4-Phase)
+12V
FB IDROOP
COMP
VSEN
VDIFF
BOOT1
RGND
NTC
UGATE1
PLACE IN
CLOSE
PROXIMITY
+5V
PHASE1
VCC
LGATE1
ISEN1ISEN1+
OFS
FS
+12V
REF
PVCC1_2
BOOT2
SS
UGATE2
PHASE2
LGATE2
LOAD
OVPSEL
ISL6312
ISEN2ISEN2+
+12V
VID7
VID6
VID5
VID4
VID3
VID2
VID1
VID0
PVCC3
BOOT3
UGATE3
PHASE3
VRSEL
PGOOD
LGATE3
+12V
ISEN3ISEN3+
+12V
+12V
EN
EN_PH4
DRSEL
BOOT
VCC UGATE
PVCC
PHASE
ISL6612
LGATE
PWM4
PWM
GND
GND
ISEN4ISEN4+
5
FN9289.5
February 25, 2010
ISL6312
Absolute Maximum Ratings
Thermal Information
Supply Voltage, VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +6V
Supply Voltage, PVCC . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +15V
BOOT Voltage, VBOOT . . . . . . . . . . . . . . GND - 0.3V to GND + 36V
BOOT to PHASE Voltage, VBOOT-PHASE . . . . . . -0.3V to 15V (DC)
-0.3V to 16V (<10ns, 10µJ)
PHASE Voltage, VPHASE . . . . . . . GND - 0.3V to 15V (PVCC = 12)
GND - 8V (<400ns, 20µJ) to 24V (<200ns, VBOOT - PHASE = 12V)
UGATE Voltage, VUGATE. . . . . . . . VPHASE - 0.3V to VBOOT + 0.3V
VPHASE - 3.5V (<100ns Pulse Width, 2µJ) to VBOOT + 0.3V
LGATE Voltage, VLGATE . . . . . . . . . . . GND - 0.3V to PVCC + 0.3V
GND - 5V (<100ns Pulse Width, 2µJ) to PVCC + 0.3V
Input, Output, or I/O Voltage . . . . . . . . . GND - 0.3V to VCC + 0.3V
ESD Classification . . . . . . . . . . . . . . . . . . . . . . . Class I JEDEC STD
Thermal Resistance
θJA (°C/W)
θJC (°C/W)
QFN Package (Notes 3, 4) . . . . . . . . . .
32
3.5
Maximum Junction Temperature . . . . . . . . . . . . . . . . . . . . . . +150°C
Maximum Storage Temperature Range . . . . . . . . . .-65°C to +150°C
Pb-free reflow profile . . . . . . . . . . . . . . . . . . . . . . . . . .see link below
http://www.intersil.com/pbfree/Pb-FreeReflow.asp
Recommended Operating Conditions
VCC Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .+5V ±5%
PVCC Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . +5V to 12V ±5%
Ambient Temperature (ISL6312CRZ) . . . . . . . . . . . . . 0°C to +70°C
Ambient Temperature (ISL6312IRZ) . . . . . . . . . . . . .-40°C to +85°C
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product reliability and
result in failures not covered by warranty.
NOTES:
3. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See
Tech Brief TB379.
4. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside.
Electrical Specifications
Recommended Operating Conditions, Unless Otherwise Specified.
PARAMETER
TEST CONDITIONS
MIN
(Note 6)
TYP
MAX
(Note 6) UNITS
BIAS SUPPLIES
Input Bias Supply Current
IVCC; EN = high
15
20
25
mA
Gate Drive Bias Current - PVCC1_2 Pin
IPVCC1_2; EN = high
2
4.3
6
mA
Gate Drive Bias Current - PVCC3 Pin
IPVCC3; EN = high
1
2.1
3
mA
VCC POR (Power-On Reset) Threshold
VCC rising
4.25
4.38
4.50
V
VCC falling
3.75
3.88
4.00
V
PVCC rising
4.25
4.38
4.50
V
PVCC falling
3.60
3.88
4.00
V
Oscillator Frequency Accuracy, fSW
RT = 100kΩ (± 0.1%)
225
250
275
kHz
Adjustment Range of Switching Frequency
(Note 5)
0.08
-
1.0
MHz
Oscillator Ramp Amplitude, VPP
(Note 5)
-
1.50
-
V
EN Rising Threshold
-
0.85
-
V
EN Hysteresis
-
110
-
mV
EN_PH4 Rising Threshold
1.160
1.210
1.250
V
EN_PH4 Falling Threshold
1.00
1.06
1.10
V
0.1
0.2
0.3
V
System Accuracy (1.000V - 1.600V)
-0.5
-
0.5
%
System Accuracy (0.600V - 1.000V)
-1.0
-
1.0
%
System Accuracy (0.375V - 0.600V)
-2.0
-
2.0
%
DAC Input Low Voltage (VR10, VR11)
-
-
0.4
V
DAC Input High Voltage (VR10, VR11)
0.8
-
-
V
PVCC POR (Power-On Reset) Threshold
PWM MODULATOR
CONTROL THRESHOLDS
COMP Shutdown Threshold
COMP falling
REFERENCE AND DAC
6
FN9289.5
February 25, 2010
ISL6312
Electrical Specifications
Recommended Operating Conditions, Unless Otherwise Specified. (Continued)
MIN
(Note 6)
TYP
DAC Input Low Voltage (AMD)
-
-
0.6
V
DAC Input High Voltage (AMD)
1.0
-
-
V
PARAMETER
TEST CONDITIONS
MAX
(Note 6) UNITS
PIN-ADJUSTABLE OFFSET
OFS Sink Current Accuracy (Negative Offset)
ROFS = 10kΩ from OFS to GND
37.0
40.0
43.0
μA
OFS Source Current Accuracy (Positive Offset)
ROFS = 30kΩ from OFS to VCC
50.5
53.5
56.5
μA
ERROR AMPLIFIER
DC Gain
RL = 10k to ground, (Note 5)
-
96
-
dB
Gain-Bandwidth Product
CL = 100pF, RL = 10k to ground, (Note 5)
-
20
-
MHz
Slew Rate
CL = 100pF, Load = ±400mA, (Note 5)
-
8
-
V/μs
Maximum Output Voltage
Load = 1mA
3.90
4.20
-
V
Minimum Output Voltage
Load = -1mA
-
1.30
1.5
V
VR10/VR11, RS = 100kΩ
-
1.563
-
mV/µs
SOFT-START RAMP
Soft-Start Ramp Rate
AMD
Adjustment Range of Soft-Start Ramp Rate (Note 5)
2.063
mV/µs
0.625
-
6.25
mV/µs
PWM OUTPUT
PWM Output Voltage LOW Threshold
Iload = ±500µA
-
-
0.5
V
PWM Output Voltage HIGH Threshold
Iload = ±500µA
4.5
-
-
V
Current Sense Resistance, RISEN
T = +25°C
297
300
303
Ω
Sensed Current Tolerance
ISEN1+ = ISEN2+ = ISEN3+ = ISEN4+ = 80μA
76
80
84
μA
Normal operation
110
125
140
μA
Dynamic VID change
143
163
183
μA
Normal operation
150
177
204
μA
209.4
238
266.6
μA
CURRENT SENSING
OVERCURRENT PROTECTION
Overcurrent Trip Level - Average Channel
Overcurrent Trip Level - Individual Channel
Dynamic VID change (Note NOTES:)
PROTECTION
Undervoltage Threshold
VSEN falling
55
60
65
%VID
Undervoltage Hysteresis
VSEN rising
-
10
-
%VID
Overvoltage Threshold During Soft-Start
VR10/VR11
1.24
1.28
1.32
V
AMD
2.13
2.20
2.27
V
VR10/VR11, OVPSEL tied to ground, VSEN rising
VDAC +
150mV
VDAC +
175mV
VDAC +
200mV
V
AMD, OVPSEL tied to ground, VSEN rising
VDAC +
225mV
VDAC +
250mV
VDAC +
275mV
V
Overvoltage Threshold (Alternate)
OVPSEL tied to +5V, VSEN rising
VDAC +
325mV
VDAC +
350mV
VDAC +
375mV
V
Overvoltage Hysteresis
VSEN falling
-
100
-
mV
tRUGATE; VPVCC = 12V, 3nF load, 10% to 90%
-
26
-
ns
Overvoltage Threshold (Default)
SWITCHING TIME (Note 5)
UGATE Rise Time
7
FN9289.5
February 25, 2010
ISL6312
Electrical Specifications
Recommended Operating Conditions, Unless Otherwise Specified. (Continued)
PARAMETER
TEST CONDITIONS
MIN
(Note 6)
TYP
MAX
(Note 6) UNITS
LGATE Rise Time
tRLGATE; VPVCC = 12V, 3nF load, 10% to 90%
-
18
-
ns
UGATE Fall Time
tFUGATE; VPVCC = 12V, 3nF load, 90% to 10%
-
18
-
ns
LGATE Fall Time
tFLGATE; VPVCC = 12V, 3nF load, 90% to 10%
-
12
-
ns
UGATE Turn-On Non-Overlap
tPDHUGATE; VPVCC = 12V, 3nF load, adaptive
-
10
-
ns
LGATE Turn-On Non-Overlap
tPDHLGATE; VPVCC = 12V, 3nF load, adaptive
-
10
-
ns
GATE DRIVE RESISTANCE (Note 5)
Upper Drive Source Resistance
VPVCC = 12V, 15mA source current
1.25
2.0
3.0
Ω
Upper Drive Sink Resistance
VPVCC = 12V, 15mA sink current
0.9
1.65
3.0
Ω
Lower Drive Source Resistance
VPVCC = 12V, 15mA source current
0.85
1.25
2.2
Ω
Lower Drive Sink Resistance
VPVCC = 12V, 15mA sink current
0.60
0.80
1.35
Ω
Thermal Shutdown Setpoint
-
160
-
°C
Thermal Recovery Setpoint
-
100
-
°C
OVER TEMPERATURE SHUTDOWN (Note 5)
NOTES:
5. Limits established by characterization and are not production tested.
6. Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established by characterization
and are not production tested.
Timing Diagram
tPDHUGATE
tRUGATE
tFUGATE
UGATE
LGATE
tFLGATE
tRLGATE
tPDHLGATE
8
FN9289.5
February 25, 2010
ISL6312
Functional Pin Descriptions
VCC
VCC is the bias supply for the ICs small-signal circuitry.
Connect this pin to a +5V supply and decouple using a
quality 0.1µF ceramic capacitor.
PVCC1_2 and PVCC3
These pins are the power supply pins for the corresponding
channel MOSFET drive, and can be connected to any
voltage from +5V to +12V depending on the desired
MOSFET gate-drive level. Decouple these pins with a quality
1.0µF ceramic capacitor.
Leaving PVCC3 unconnected or grounded programs the
controller for 2-phase operation.
GND
GND is the bias and reference ground for the IC.
EN
This pin is a threshold-sensitive (approximately 0.85V) enable
input for the controller. Held low, this pin disables controller
operation. Pulled high, the pin enables the controller for
operation.
FS
A resistor, placed from FS to ground, sets the switching
frequency of the controller.
VID0, VID1, VID2, VID3, VID4, VID5, VID6, and VID7
These are the inputs for the internal DAC that provides the
reference voltage for output regulation. These pins respond to
TTL logic thresholds. These pins are internally pulled high, to
approximately 1.2V, by 40µA internal current sources for Intel
modes of operation, and pulled low by 20µA internal current
sources for AMD modes of operation. The internal pull-up
current decreases to 0 as the VID voltage approaches the
internal pull-up voltage. All VID pins are compatible with
external pull-up voltages not exceeding the IC’s bias voltage
(VCC).
VRSEL
The state of this pin selects which of the available DAC tables
will be used to decode the VID inputs and puts the controller
into the corresponding mode of operation. For VR10 mode of
operation, VRSEL should be less then 0.6V. The VR11 mode of
operation can be selected by setting VRSEL between 0.6V and
3.0V, and AMD compliance is selected if this pin is between
3.0V and VCC. The VRSEL pin has an internal 40µA pull-up
current to approximately 2V.
VSEN and RGND
VSEN and RGND are inputs to the precision differential
remote-sense amplifier and should be connected to the sense
pins of the remote load.
9
VDIFF
VDIFF is the output of the differential remote-sense amplifier.
The voltage on this pin is equal to the difference between
VSEN and RGND.
FB and COMP
These pins are the internal error amplifier inverting input and
output respectively. FB, VDIFF, and COMP are tied together
through external R-C networks to compensate the regulator.
IDROOP
The IDROOP pin is the average channel-current sense
output. Connecting this pin through a tuned parallel R-C
network to FB allows the converter to incorporate output
voltage droop proportional to the output current. If voltage
droop is not desired leave this pin unconnected.
REF
The REF input pin is the positive input of the error amplifier. It
is internally connected to the DAC output through a 1kΩ
resistor. A capacitor is used between the REF pin and ground
to smooth the voltage transition during Dynamic VID
operations.
OFS
The OFS pin provides a means to program a DC current for
generating an offset voltage across the resistor between FB
and VDIFF. The offset current is generated via an external
resistor and precision internal voltage references. The polarity
of the offset is selected by connecting the resistor to GND or
VCC. For no offset, the OFS pin should be left unconnected.
ISEN1-, ISEN1+, ISEN2-, ISEN2+, ISEN3-, ISEN3+,
ISEN4-, and ISEN4+
These pins are used for differentially sensing the
corresponding channel output currents. The sensed currents
are used for channel balancing, protection, and load line
regulation.
Connect ISEN1-, ISEN2-, ISEN3-, and ISEN4- to the node
between the RC sense elements surrounding the inductor of
their respective channel. Tie the ISEN+ pins to the VCORE
side of their corresponding channel’s sense capacitor.
UGATE1, UGATE2, and UGATE3
Connect these pins to the corresponding upper MOSFET
gates. These pins are used to control the upper MOSFETs
and are monitored for shoot-through prevention purposes.
BOOT1, BOOT2, and BOOT3
These pins provide the bias voltage for the corresponding
upper MOSFET drives. Connect these pins to appropriatelychosen external bootstrap capacitors. Internal bootstrap
diodes connected to the PVCC pins provide the necessary
bootstrap charge.
FN9289.5
February 25, 2010
ISL6312
PHASE1, PHASE2, and PHASE3
Connect these pins to the sources of the corresponding
upper MOSFETs. These pins are the return path for the
upper MOSFET drives.
LGATE1, LGATE2, and LGATE3
These pins are used to control the lower MOSFETs. Connect
these pins to the corresponding lower MOSFETs’ gates.
PWM4
Pulse-width modulation output. Connect this pin to the PWM
input pin of an Intersil driver IC if 4-phase operation is
desired.
Operation
Multiphase Power Conversion
Microprocessor load current profiles have changed to the
point that using single-phase regulators is no longer a viable
solution. Designing a regulator that is cost-effective,
thermally sound, and efficient has become a challenge that
only multiphase converters can accomplish. The ISL6322
controller helps simplify implementation by integrating vital
functions and requiring minimal external components. The
“Block Diagram” on page 3 provides a top level view of
multiphase power conversion using the ISL6322 controller.
EN_PH4
This pin has two functions. First, a resistor divider connected
to this pin will provide a POR power-up synch between the
on-chip and external driver. The resistor divider should be
designed so that when the POR-trip point of the external
driver is reached the voltage on this pin should be 1.21V.
IL1 + IL2 + IL3, 7A/DIV
IL3, 7A/DIV
PWM3, 5V/DIV
IL2, 7A/DIV
The second function of this pin is disabling PWM4 for
3-phase operation. This can be accomplished by connecting
this pin to a +5V supply.
PWM2, 5V/DIV
IL1, 7A/DIV
SS
A resistor, placed from SS to ground, will set the soft-start
ramp slope for the Intel DAC modes of operation. Refer to
Equations 18 and 19 for proper resistor calculation.
For AMD modes of operation, the soft-start ramp frequency
is preset, so this pin can be left unconnected.
OVPSEL
This pin selects the OVP trip point during normal operation.
Leaving it unconnected or tieing it to ground selects the
default setting of VDAC+175mV for Intel Modes of operation
and VDAC+250mV for AMD modes of operation. Connecting
this pin to VCC will select an OVP trip setting of VID+350mV
for all modes of operation.
DRSEL
This pin selects the adaptive dead time scheme the internal
drivers will use. If driving MOSFETs, tie this pin to ground to
select the PHASE detect scheme or to a +5V supply through
a 50kΩ resistor to select the LGATE detect scheme.
PGOOD
During normal operation PGOOD indicates whether the
output voltage is within specified overvoltage and
undervoltage limits. If the output voltage exceeds these limits
or a reset event occurs (such as an overcurrent event),
PGOOD is pulled low. PGOOD is always low prior to the end
of soft-start.
10
PWM1, 5V/DIV
1µs/DIV
FIGURE 1. PWM AND INDUCTOR-CURRENT WAVEFORMS
FOR 3-PHASE CONVERTER
Interleaving
The switching of each channel in a multiphase converter is
timed to be symmetrically out of phase with each of the other
channels. In a 3-phase converter, each channel switches 1/3
cycle after the previous channel and 1/3 cycle before the
following channel. As a result, the three-phase converter has a
combined ripple frequency three times greater than the ripple
frequency of any one phase. In addition, the peak-to-peak
amplitude of the combined inductor currents is reduced in
proportion to the number of phases (Equations 1 and 2).
Increased ripple frequency and lower ripple amplitude mean
that the designer can use less per-channel inductance and
lower total output capacitance for any performance
specification.
Figure 1 illustrates the multiplicative effect on output ripple
frequency. The three channel currents (IL1, IL2, and IL3)
combine to form the AC ripple current and the DC load
current. The ripple component has three times the ripple
frequency of each individual channel current. Each PWM
pulse is terminated 1/3 of a cycle after the PWM pulse of the
previous phase. The peak-to-peak current for each phase is
about 7A, and the DC components of the inductor currents
combine to feed the load.
FN9289.5
February 25, 2010
ISL6312
To understand the reduction of ripple current amplitude in the
multiphase circuit, examine the equation representing an
individual channel peak-to-peak inductor current.
( V IN – V OUT ) ⋅ V OUT
I ( P – P ) = --------------------------------------------------------L ⋅ fS ⋅ V
(EQ. 1)
IN
In Equation 1, VIN and VOUT are the input and output
voltages respectively, L is the single-channel inductor value,
and fS is the switching frequency.
The output capacitors conduct the ripple component of the
inductor current. In the case of multiphase converters, the
capacitor current is the sum of the ripple currents from each
of the individual channels. Compare Equation 1 to the
expression for the peak-to-peak current after the summation
of N symmetrically phase-shifted inductor currents in
Equation 2. Peak-to-peak ripple current decreases by an
amount proportional to the number of channels. Output
voltage ripple is a function of capacitance, capacitor
equivalent series resistance (ESR), and inductor ripple
current. Reducing the inductor ripple current allows the
designer to use fewer or less costly output capacitors.
( V IN – N ⋅ V OUT ) ⋅ V OUT
I C ( P – P ) = ------------------------------------------------------------------L ⋅ fS ⋅ V
(EQ. 2)
IN
Another benefit of interleaving is to reduce input ripple
current. Input capacitance is determined in part by the
maximum input ripple current. Multiphase topologies can
improve overall system cost and size by lowering input ripple
current and allowing the designer to reduce the cost of input
capacitance. The example in Figure 2 illustrates input
currents from a three-phase converter combining to reduce
the total input ripple current.
The converter depicted in Figure 2 delivers 1.5V to a 36A load
from a 12V input. The RMS input capacitor current is 5.9A.
Compare this to a single-phase converter also stepping down
12V to 1.5V at 36A. The single-phase converter has 11.9A
RMS input capacitor current. The single-phase converter
must use an input capacitor bank with twice the RMS current
capacity as the equivalent three-phase converter.
INPUT-CAPACITOR CURRENT, 10A/DIV
CHANNEL 3
INPUT CURRENT
10A/DIV
CHANNEL 2
INPUT CURRENT
10A/DIV
CHANNEL 1
INPUT CURRENT
10A/DIV
1µs/DIV
FIGURE 2. CHANNEL INPUT CURRENTS AND INPUTCAPACITOR RMS CURRENT FOR 3-PHASE
CONVERTER
11
Active Pulse Positioning (APP) Modulated PWM
Operation
The ISL6312 uses a proprietary Active Pulse Positioning
(APP) modulation scheme to control the internal PWM
signals that command each channel’s driver to turn their
upper and lower MOSFETs on and off. The time interval in
which a PWM signal can occur is generated by an internal
clock, whose cycle time is the inverse of the switching
frequency set by the resistor between the FS pin and
ground. The advantage of Intersil’s proprietary Active Pulse
Positioning (APP) modulator is that the PWM signal has the
ability to turn on at any point during this PWM time interval,
and turn off immediately after the PWM signal has
transitioned high. This is important because is allows the
controller to quickly respond to output voltage drops
associated with current load spikes, while avoiding the ring
back affects associated with other modulation schemes.
The PWM output state is driven by the position of the error
amplifier output signal, VCOMP, minus the current correction
signal relative to the proprietary modulator ramp waveform
as illustrated in Figure 3. At the beginning of each PWM time
interval, this modified VCOMP signal is compared to the
internal modulator waveform. As long as the modified
VCOMP voltage is lower then the modulator waveform
voltage, the PWM signal is commanded low. The internal
MOSFET driver detects the low state of the PWM signal and
turns off the upper MOSFET and turns on the lower
synchronous MOSFET. When the modified VCOMP voltage
crosses the modulator ramp, the PWM output transitions
high, turning off the synchronous MOSFET and turning on
the upper MOSFET. The PWM signal will remain high until
the modified VCOMP voltage crosses the modulator ramp
again. When this occurs the PWM signal will transition low
again.
During each PWM time interval the PWM signal can only
transition high once. Once PWM transitions high it can not
transition high again until the beginning of the next PWM
time interval. This prevents the occurrence of double PWM
pulses occurring during a single period.
To further improve the transient response, ISL6312 also
implements Intersil’s proprietary Adaptive Phase Alignment
(APA) technique, which turns on all phases together under
transient events with large step current. With both APP and
APA control, ISL6312 can achieve excellent transient
performance and reduce the demand on the output
capacitors.
Channel-Current Balance
One important benefit of multiphase operation is the thermal
advantage gained by distributing the dissipated heat over
multiple devices and greater area. By doing this the designer
avoids the complexity of driving parallel MOSFETs and the
expense of using expensive heat sinks and exotic magnetic
materials.
FN9289.5
February 25, 2010
ISL6312
Continuous Current Sampling
In order to realize proper current-balance, the currents in
each channel are sensed continuously every switching
cycle. During this time the current-sense amplifier uses the
ISEN inputs to reproduce a signal proportional to the
inductor current, IL. This sensed current, ISEN, is simply a
scaled version of the inductor current.
The ISL6312 supports inductor DCR current sensing to
continuously sense each channel’s current for channelcurrent balance. The internal circuitry, shown in Figure 5
represents channel n of an N-channel converter. This
circuitry is repeated for each channel in the converter, but
may not be active depending on how many channels are
operating.
VIN
IL
UGATE(n)
L
MOSFET
DRIVER
LGATE(n)
VL(s)
f(s)
R1
I4
IER
IAVG
-
Σ
÷N
DCR
+
I3
VOUT
INDUCTOR
+
FILTER
-
TO GATE
CONTROL
LOGIC
+
+
MODULATOR
RAMP
WAVEFORM
-
PWM1
COUT
VC(s)
-
+
VCOMP
-
In order to realize the thermal advantage, it is important that
each channel in a multiphase converter be controlled to
carry equal amounts of current at any load level. To achieve
this, the currents through each channel must be sampled
every switching cycle. The sampled currents, In, from each
active channel are summed together and divided by the
number of active channels. The resulting cycle average
current, IAVG, provides a measure of the total load-current
demand on the converter during each switching cycle.
Channel-current balance is achieved by comparing the
sampled current of each channel to the cycle average
current, and making the proper adjustment to each channel
pulse width based on the error. Intersil’s patented currentbalance method is illustrated in Figure 3, with error
correction for Channel 1 represented. In the figure, the cycle
average current, IAVG, is compared with the Channel 1
sample, I1 , to create an error signal IER.
C1
R2*
ISL6312 INTERNAL CIRCUIT
I2
I1
In
NOTE: Channel 3 and 4 are optional.
SAMPLE
FIGURE 3. CHANNEL-1 PWM FUNCTION AND CURRENTBALANCE ADJUSTMENT
-
The filtered error signal modifies the pulse width
commanded by VCOMP to correct any unbalance and force
IER toward zero. The same method for error signal
correction is applied to each active channel.
VC(s)
RISEN
ISEN
-
+
+
ISEN-(n)
ISEN+(n)
*R2 is OPTIONAL
FIGURE 5. INDUCTOR DCR CURRENT SENSING
CONFIGURATION
PWM
SWITCHING PERIOD
IL
Inductor windings have a characteristic distributed
resistance or DCR (Direct Current Resistance). For
simplicity, the inductor DCR is considered as a separate
lumped quantity, as shown in Figure 5. The channel current
IL, flowing through the inductor, passes through the DCR.
Equation 3 shows the s-domain equivalent voltage, VL,
across the inductor.
V L ( s ) = I L ⋅ ( s ⋅ L + DCR )
(EQ. 3)
ISEN
TIME
A simple R-C network across the inductor (R1 and C)
extracts the DCR voltage, as shown in Figure 5. The voltage
across the sense capacitor, VC, can be shown to be
proportional to the channel current IL, shown in Equation 4.
FIGURE 4. CONTINUOUS CURRENT SAMPLING
12
FN9289.5
February 25, 2010
ISL6312
s⋅L
⎛ ------------+ 1⎞
⎝ DCR
⎠
------------------------------------- ⋅ DCR ⋅ I L
VC ( s ) =
( s ⋅ R1 ⋅ C + 1 )
(EQ. 4)
In some cases it may be necessary to use a resistor divider
R-C network to sense the current through the inductor. This
can be accomplished by placing a second resistor, R2,
across the sense capacitor. In these cases the voltage
across the sense capacitor, VC, becomes proportional to the
channel current IL, and the resistor divider ratio, K.
s⋅L
⎛ ------------+ 1⎞
⎝ DCR
⎠
------------------------------------------------------- ⋅ K ⋅ DCR ⋅ I L
VC ( s ) =
(
⋅
R
)
R
⎛
⎞
1
2
⎜ s ⋅ ------------------------ ⋅ C + 1⎟
R1 + R2
⎝
⎠
(EQ. 5)
TABLE 1. ISL6312 DAC SELECT TABLE
DAC VERSION
VRSEL PIN
VID7 PIN
VR10(Extended)
VRSEL < 0.6V
-
VR11
0.8V < VRSEL < 3.0V
-
AMD 5-Bit
3.0V < VRSEL < VCC
low
AMD 6-Bit
3.0V < VRSEL < VCC
high
TABLE 2. VR10 (EXTENDED) VOLTAGE IDENTIFICATION
CODES
VID4
VID3
VID2
VID1
VID0
VID5
VID6
VDAC
0
1
0
1
0
1
1
1.60000
0
1
0
1
0
1
0
1.59375
0
1
0
1
1
0
1
1.58750
0
1
0
1
1
0
0
1.58125
0
1
0
1
1
1
1
1.57500
0
1
0
1
1
1
0
1.56875
0
1
1
0
0
0
1
1.56250
0
1
1
0
0
0
0
1.55625
0
1
1
0
0
1
1
1.55000
0
1
1
0
0
1
0
1.54375
0
1
1
0
1
0
1
1.53750
0
1
1
0
1
0
0
1.53125
0
1
1
0
1
1
1
1.52500
0
1
1
0
1
1
0
1.51875
0
1
1
1
0
0
1
1.51250
Output Voltage Setting
0
1
1
1
0
0
0
1.50625
The ISL6312 uses a digital to analog converter (DAC) to
generate a reference voltage based on the logic signals at
the VID pins. The DAC decodes the logic signals into one of
the discrete voltages shown in Tables 2, 3, 4 and 5. In Intel
modes of operation, each VID pin is pulled up to an internal
1.2V voltage by a weak current source (40µA), which
decreases to 0A as the voltage at the VID pin varies from 0
to the internal 1.2V pull-up voltage. In AMD modes of
operation the VID pins are pulled low by a weak 20µA
current source. External pull-up resistors or active-high
output stages can augment the pull-up current sources, up to
a voltage of 5V.
0
1
1
1
0
1
1
1.50000
0
1
1
1
0
1
0
1.49375
0
1
1
1
1
0
1
1.48750
0
1
1
1
1
0
0
1.48125
0
1
1
1
1
1
1
1.47500
0
1
1
1
1
1
0
1.46875
1
0
0
0
0
0
1
1.46250
1
0
0
0
0
0
0
1.45625
1
0
0
0
0
1
1
1.45000
1
0
0
0
0
1
0
1.44375
1
0
0
0
1
0
1
1.43750
1
0
0
0
1
0
0
1.43125
1
0
0
0
1
1
1
1.42500
1
0
0
0
1
1
0
1.41875
1
0
0
1
0
0
1
1.41250
1
0
0
1
0
0
0
1.40625
R2
K = --------------------R2 + R1
(EQ. 6)
If the R-C network components are selected such that the
RC time constant matches the inductor L/DCR time
constant, then VC is equal to the voltage drop across the
DCR multiplied by the ratio of the resistor divider, K. If a
resistor divider is not being used, the value for K is 1.
The capacitor voltage VC, is then replicated across the
sense resistor RISEN. The current through RISEN is
proportional to the inductor current. Equation 7 shows that
the proportion between the channel current and the sensed
current (ISEN) is driven by the value of the sense resistor,
the resistor divider ratio, and the DCR of the inductor.
DCR
I SEN = K ⋅ I L ⋅ -----------------R ISEN
(EQ. 7)
The ISL6312 accommodates four different DAC ranges: Intel
VR10 (Extended), Intel VR11, AMD K8/K9 5-bit, and AMD
6-bit. The state of the VRSEL and VID7 pins decide which
DAC version is active. Refer to Table 1 for a description of
how to select the desired DAC version.
13
FN9289.5
February 25, 2010
ISL6312
TABLE 2. VR10 (EXTENDED) VOLTAGE IDENTIFICATION
CODES (Continued)
TABLE 2. VR10 (EXTENDED) VOLTAGE IDENTIFICATION
CODES (Continued)
VID4
VID3
VID2
VID1
VID0
VID5
VID6
VDAC
VID4
VID3
VID2
VID1
VID0
VID5
VID6
VDAC
1
0
0
1
0
1
1
1.40000
1
1
1
0
0
0
0
1.15625
1
0
0
1
0
1
0
1.39375
1
1
1
0
0
1
1
1.15000
1
0
0
1
1
0
1
1.38750
1
1
1
0
0
1
0
1.14375
1
0
0
1
1
0
0
1.38125
1
1
1
0
1
0
1
1.13750
1
0
0
1
1
1
1
1.37500
1
1
1
0
1
0
0
1.13125
1
0
0
1
1
1
0
1.36875
1
1
1
0
1
1
1
1.12500
1
0
1
0
0
0
1
1.36250
1
1
1
0
1
1
0
1.11875
1
0
1
0
0
0
0
1.35625
1
1
1
1
0
0
1
1.11250
1
0
1
0
0
1
1
1.35000
1
1
1
1
0
0
0
1.10625
1
0
1
0
0
1
0
1.34375
1
1
1
1
0
1
1
1.10000
1
0
1
0
1
0
1
1.33750
1
1
1
1
0
1
0
1.09375
1
0
1
0
1
0
0
1.33125
1
1
1
1
1
0
1
OFF
1
0
1
0
1
1
1
1.32500
1
1
1
1
1
0
0
OFF
1
0
1
0
1
1
0
1.31875
1
1
1
1
1
1
1
OFF
1
0
1
1
0
0
1
1.31250
1
1
1
1
1
1
0
OFF
1
0
1
1
0
0
0
1.30625
0
0
0
0
0
0
1
1.08750
1
0
1
1
0
1
1
1.30000
0
0
0
0
0
0
0
1.08125
1
0
1
1
0
1
0
1.29375
0
0
0
0
0
1
1
1.07500
1
0
1
1
1
0
1
1.28750
0
0
0
0
0
1
0
1.06875
1
0
1
1
1
0
0
1.28125
0
0
0
0
1
0
1
1.06250
1
0
1
1
1
1
1
1.27500
0
0
0
0
1
0
0
1.05625
1
0
1
1
1
1
0
1.26875
0
0
0
0
1
1
1
1.05000
1
1
0
0
0
0
1
1.26250
0
0
0
0
1
1
0
1.04375
1
1
0
0
0
0
0
1.25625
0
0
0
1
0
0
1
1.03750
1
1
0
0
0
1
1
1.25000
0
0
0
1
0
0
0
1.03125
1
1
0
0
0
1
0
1.24375
0
0
0
1
0
1
1
1.02500
1
1
0
0
1
0
1
1.23750
0
0
0
1
0
1
0
1.01875
1
1
0
0
1
0
0
1.23125
0
0
0
1
1
0
1
1.01250
1
1
0
0
1
1
1
1.22500
0
0
0
1
1
0
0
1.00625
1
1
0
0
1
1
0
1.21875
0
0
0
1
1
1
1
1.00000
1
1
0
1
0
0
1
1.21250
0
0
0
1
1
1
0
0.99375
1
1
0
1
0
0
0
1.20625
0
0
1
0
0
0
1
0.98750
1
1
0
1
0
1
1
1.20000
0
0
1
0
0
0
0
0.98125
1
1
0
1
0
1
0
1.19375
0
0
1
0
0
1
1
0.97500
1
1
0
1
1
0
1
1.18750
0
0
1
0
0
1
0
0.96875
1
1
0
1
1
0
0
1.18125
0
0
1
0
1
0
1
0.96250
1
1
0
1
1
1
1
1.17500
0
0
1
0
1
0
0
0.95625
1
1
0
1
1
1
0
1.16875
0
0
1
0
1
1
1
0.95000
1
1
1
0
0
0
1
1.16250
0
0
1
0
1
1
0
0.94375
14
FN9289.5
February 25, 2010
ISL6312
TABLE 2. VR10 (EXTENDED) VOLTAGE IDENTIFICATION
CODES (Continued)
VID4
VID3
VID2
VID1
VID0
VID5
VID6
VDAC
0
0
1
1
0
0
1
0.93750
0
0
1
1
0
0
0
0.93125
0
0
1
1
0
1
1
0.92500
0
0
1
1
0
1
0
0.91875
0
0
1
1
1
0
1
0.91250
0
0
1
1
1
0
0
0.90625
0
0
1
1
1
1
1
0.90000
0
0
1
1
1
1
0
0.89375
0
1
0
0
0
0
1
0.88750
0
1
0
0
0
0
0
0.88125
0
1
0
0
0
1
1
0.87500
0
1
0
0
0
1
0
0.86875
0
1
0
0
1
0
1
0.86250
0
1
0
0
1
0
0
0.85625
0
1
0
0
1
1
1
0.85000
0
1
0
0
1
1
0
0.84375
0
1
0
1
0
0
1
0.83750
0
1
0
1
0
0
0
0.83125
TABLE 3. VR11 VOLTAGE IDENTIFICATION CODES
TABLE 3. VR11 VOLTAGE IDENTIFICATION CODES (Continued)
VID7
VID6
VID5
VID4
VID3
VID2
VID1
VID0
VDAC
0
0
0
1
0
0
1
1
1.49375
0
0
0
1
0
1
0
0
1.48750
0
0
0
1
0
1
0
1
1.48125
0
0
0
1
0
1
1
0
1.47500
0
0
0
1
0
1
1
1
1.46875
0
0
0
1
1
0
0
0
1.46250
0
0
0
1
1
0
0
1
1.45625
0
0
0
1
1
0
1
0
1.45000
0
0
0
1
1
0
1
1
1.44375
0
0
0
1
1
1
0
0
1.43750
0
0
0
1
1
1
0
1
1.43125
0
0
0
1
1
1
1
0
1.42500
0
0
0
1
1
1
1
1
1.41875
0
0
1
0
0
0
0
0
1.41250
0
0
1
0
0
0
0
1
1.40625
0
0
1
0
0
0
1
0
1.40000
0
0
1
0
0
0
1
1
1.39375
0
0
1
0
0
1
0
0
1.38750
0
0
1
0
0
1
0
1
1.38125
0
0
1
0
0
1
1
0
1.37500
VID7
VID6
VID5
VID4
VID3
VID2
VID1
VID0
VDAC
0
0
1
0
0
1
1
1
1.36875
0
0
0
0
0
0
0
0
OFF
0
0
1
0
1
0
0
0
1.36250
0
0
0
0
0
0
0
1
OFF
0
0
1
0
1
0
0
1
1.35625
0
0
0
0
0
0
1
0
1.60000
0
0
1
0
1
0
1
0
1.35000
0
0
0
0
0
0
1
1
1.59375
0
0
1
0
1
0
1
1
1.34375
0
0
0
0
0
1
0
0
1.58750
0
0
1
0
1
1
0
0
1.33750
0
0
0
0
0
1
0
1
1.58125
0
0
1
0
1
1
0
1
1.33125
0
0
0
0
0
1
1
0
1.57500
0
0
1
0
1
1
1
0
1.32500
0
0
0
0
0
1
1
1
1.56875
0
0
1
0
1
1
1
1
1.31875
0
0
0
0
1
0
0
0
1.56250
0
0
1
1
0
0
0
0
1.31250
0
0
0
0
1
0
0
1
1.55625
0
0
1
1
0
0
0
1
1.30625
0
0
0
0
1
0
1
0
1.55000
0
0
1
1
0
0
1
0
1.30000
0
0
0
0
1
0
1
1
1.54375
0
0
1
1
0
0
1
1
1.29375
0
0
0
0
1
1
0
0
1.53750
0
0
1
1
0
1
0
0
1.28750
0
0
0
0
1
1
0
1
1.53125
0
0
1
1
0
1
0
1
1.28125
0
0
0
0
1
1
1
0
1.52500
0
0
1
1
0
1
1
0
1.27500
0
0
0
0
1
1
1
1
1.51875
0
0
1
1
0
1
1
1
1.26875
0
0
0
1
0
0
0
0
1.51250
0
0
1
1
1
0
0
0
1.26250
0
0
0
1
0
0
0
1
1.50625
0
0
1
1
1
0
0
1
1.25625
0
0
0
1
0
0
1
0
1.50000
0
0
1
1
1
0
1
0
1.25000
15
FN9289.5
February 25, 2010
ISL6312
TABLE 3. VR11 VOLTAGE IDENTIFICATION CODES (Continued)
TABLE 3. VR11 VOLTAGE IDENTIFICATION CODES (Continued)
VID7
VID6
VID5
VID4
VID3
VID2
VID1
VID0
VDAC
VID7
VID6
VID5
VID4
VID3
VID2
VID1
VID0
VDAC
0
0
1
1
1
0
1
1
1.24375
0
1
1
0
0
0
1
1
0.99375
0
0
1
1
1
1
0
0
1.23750
0
1
1
0
0
1
0
0
0.98750
0
0
1
1
1
1
0
1
1.23125
0
1
1
0
0
1
0
1
0.98125
0
0
1
1
1
1
1
0
1.22500
0
1
1
0
0
1
1
0
0.97500
0
0
1
1
1
1
1
1
1.21875
0
1
1
0
0
1
1
1
0.96875
0
1
0
0
0
0
0
0
1.21250
0
1
1
0
1
0
0
0
0.96250
0
1
0
0
0
0
0
1
1.20625
0
1
1
0
1
0
0
1
0.95625
0
1
0
0
0
0
1
0
1.20000
0
1
1
0
1
0
1
0
0.95000
0
1
0
0
0
0
1
1
1.19375
0
1
1
0
1
0
1
1
0.94375
0
1
0
0
0
1
0
0
1.18750
0
1
1
0
1
1
0
0
0.93750
0
1
0
0
0
1
0
1
1.18125
0
1
1
0
1
1
0
1
0.93125
0
1
0
0
0
1
1
0
1.17500
0
1
1
0
1
1
1
0
0.92500
0
1
0
0
0
1
1
1
1.16875
0
1
1
0
1
1
1
1
0.91875
0
1
0
0
1
0
0
0
1.16250
0
1
1
1
0
0
0
0
0.91250
0
1
0
0
1
0
0
1
1.15625
0
1
1
1
0
0
0
1
0.90625
0
1
0
0
1
0
1
0
1.15000
0
1
1
1
0
0
1
0
0.90000
0
1
0
0
1
0
1
1
1.14375
0
1
1
1
0
0
1
1
0.89375
0
1
0
0
1
1
0
0
1.13750
0
1
1
1
0
1
0
0
0.88750
0
1
0
0
1
1
0
1
1.13125
0
1
1
1
0
1
0
1
0.88125
0
1
0
0
1
1
1
0
1.12500
0
1
1
1
0
1
1
0
0.87500
0
1
0
0
1
1
1
1
1.11875
0
1
1
1
0
1
1
1
0.86875
0
1
0
1
0
0
0
0
1.11250
0
1
1
1
1
0
0
0
0.86250
0
1
0
1
0
0
0
1
1.10625
0
1
1
1
1
0
0
1
0.85625
0
1
0
1
0
0
1
0
1.10000
0
1
1
1
1
0
1
0
0.85000
0
1
0
1
0
0
1
1
1.09375
0
1
1
1
1
0
1
1
0.84375
0
1
0
1
0
1
0
0
1.08750
0
1
1
1
1
1
0
0
0.83750
0
1
0
1
0
1
0
1
1.08125
0
1
1
1
1
1
0
1
0.83125
0
1
0
1
0
1
1
0
1.07500
0
1
1
1
1
1
1
0
0.82500
0
1
0
1
0
1
1
1
1.06875
0
1
1
1
1
1
1
1
0.81875
0
1
0
1
1
0
0
0
1.06250
1
0
0
0
0
0
0
0
0.81250
0
1
0
1
1
0
0
1
1.05625
1
0
0
0
0
0
0
1
0.80625
0
1
0
1
1
0
1
0
1.05000
1
0
0
0
0
0
1
0
0.80000
0
1
0
1
1
0
1
1
1.04375
1
0
0
0
0
0
1
1
0.79375
0
1
0
1
1
1
0
0
1.03750
1
0
0
0
0
1
0
0
0.78750
0
1
0
1
1
1
0
1
1.03125
1
0
0
0
0
1
0
1
0.78125
0
1
0
1
1
1
1
0
1.02500
1
0
0
0
0
1
1
0
0.77500
0
1
0
1
1
1
1
1
1.01875
1
0
0
0
0
1
1
1
0.76875
0
1
1
0
0
0
0
0
1.01250
1
0
0
0
1
0
0
0
0.76250
0
1
1
0
0
0
0
1
1.00625
1
0
0
0
1
0
0
1
0.75625
0
1
1
0
0
0
1
0
1.00000
1
0
0
0
1
0
1
0
0.75000
16
FN9289.5
February 25, 2010
ISL6312
TABLE 3. VR11 VOLTAGE IDENTIFICATION CODES (Continued)
TABLE 3. VR11 VOLTAGE IDENTIFICATION CODES (Continued)
VID7
VID6
VID5
VID4
VID3
VID2
VID1
VID0
VDAC
VID7
VID6
VID5
VID4
VID3
VID2
VID1
VID0
1
0
0
0
1
0
1
1
0.74375
1
1
1
1
1
1
1
0
OFF
1
0
0
0
1
1
0
0
0.73750
1
1
1
1
1
1
1
1
OFF
1
0
0
0
1
1
0
1
0.73125
1
0
0
0
1
1
1
0
0.72500
1
0
0
0
1
1
1
1
0.71875
1
0
0
1
0
0
0
0
0.71250
1
0
0
1
0
0
0
1
0.70625
1
0
0
1
0
0
1
0
0.70000
1
0
0
1
0
0
1
1
0.69375
1
0
0
1
0
1
0
0
0.68750
1
0
0
1
0
1
0
1
0.68125
1
0
0
1
0
1
1
0
0.67500
1
0
0
1
0
1
1
1
0.66875
1
0
0
1
1
0
0
0
0.66250
1
0
0
1
1
0
0
1
0.65625
1
0
0
1
1
0
1
0
0.65000
1
0
0
1
1
0
1
1
0.64375
1
0
0
1
1
1
0
0
0.63750
1
0
0
1
1
1
0
1
0.63125
1
0
0
1
1
1
1
0
0.62500
1
0
0
1
1
1
1
1
0.61875
1
0
1
0
0
0
0
0
0.61250
1
0
1
0
0
0
0
1
0.60625
1
0
1
0
0
0
1
0
0.60000
1
0
1
0
0
0
1
1
0.59375
1
0
1
0
0
1
0
0
0.58750
1
0
1
0
0
1
0
1
0.58125
1
0
1
0
0
1
1
0
0.57500
1
0
1
0
0
1
1
1
0.56875
1
0
1
0
1
0
0
0
0.56250
1
0
1
0
1
0
0
1
0.55625
1
0
1
0
1
0
1
0
0.55000
1
0
1
0
1
0
1
1
0.54375
1
0
1
0
1
1
0
0
0.53750
1
0
1
0
1
1
0
1
0.53125
1
0
1
0
1
1
1
0
0.52500
1
0
1
0
1
1
1
1
0.51875
1
0
1
1
0
0
0
0
0.51250
1
0
1
1
0
0
0
1
0.50625
1
0
1
1
0
0
1
0
0.50000
17
VDAC
TABLE 4. AMD 5-BIT VOLTAGE IDENTIFICATION CODES
VID4
VID3
VID2
VID1
VID0
VDAC
1
1
1
1
1
Off
1
1
1
1
0
0.800
1
1
1
0
1
0.825
1
1
1
0
0
0.850
1
1
0
1
1
0.875
1
1
0
1
0
0.900
1
1
0
0
1
0.925
1
1
0
0
0
0.950
1
0
1
1
1
0.975
1
0
1
1
0
1.000
1
0
1
0
1
1.025
1
0
1
0
0
1.050
1
0
0
1
1
1.075
1
0
0
1
0
1.100
1
0
0
0
1
1.125
1
0
0
0
0
1.150
0
1
1
1
1
1.175
0
1
1
1
0
1.200
0
1
1
0
1
1.225
0
1
1
0
0
1.250
0
1
0
1
1
1.275
0
1
0
1
0
1.300
0
1
0
0
1
1.325
0
1
0
0
0
1.350
0
0
1
1
1
1.375
0
0
1
1
0
1.400
0
0
1
0
1
1.425
0
0
1
0
0
1.450
0
0
0
1
1
1.475
0
0
0
1
0
1.500
0
0
0
0
1
1.525
0
0
0
0
0
1.550
FN9289.5
February 25, 2010
ISL6312
TABLE 5. AMD 6-BIT VOLTAGE IDENTIFICATION CODES
TABLE 5. AMD 6-BIT VOLTAGE IDENTIFICATION CODES
(Continued)
VID5
VID4
VID3
VID2
VID1
VID0
VDAC
VID5
VID4
VID3
VID2
VID1
VID0
VDAC
0
0
0
0
0
0
1.5500
1
0
0
1
1
1
0.6750
0
0
0
0
0
1
1.5250
1
0
1
0
0
0
0.6625
0
0
0
0
1
0
1.5000
1
0
1
0
0
1
0.6500
0
0
0
0
1
1
1.4750
1
0
1
0
1
0
0.6375
0
0
0
1
0
0
1.4500
1
0
1
0
1
1
0.6250
0
0
0
1
0
1
1.4250
1
0
1
1
0
0
0.6125
0
0
0
1
1
0
1.4000
1
0
1
1
0
1
0.6000
0
0
0
1
1
1
1.3750
1
0
1
1
1
0
0.5875
0
0
1
0
0
0
1.3500
1
0
1
1
1
1
0.5750
0
0
1
0
0
1
1.3250
1
1
0
0
0
0
0.5625
0
0
1
0
1
0
1.3000
1
1
0
0
0
1
0.5500
0
0
1
0
1
1
1.2750
1
1
0
0
1
0
0.5375
0
0
1
1
0
0
1.2500
1
1
0
0
1
1
0.5250
0
0
1
1
0
1
1.2250
1
1
0
1
0
0
0.5125
0
0
1
1
1
0
1.2000
1
1
0
1
0
1
0.5000
0
0
1
1
1
1
1.1750
1
1
0
1
1
0
0.4875
0
1
0
0
0
0
1.1500
1
1
0
1
1
1
0.4750
0
1
0
0
0
1
1.1250
1
1
1
0
0
0
0.4625
0
1
0
0
1
0
1.1000
1
1
1
0
0
1
0.4500
0
1
0
0
1
1
1.0750
1
1
1
0
1
0
0.4375
0
1
0
1
0
0
1.0500
1
1
1
0
1
1
0.4250
0
1
0
1
0
1
1.0250
1
1
1
1
0
0
0.4125
0
1
0
1
1
0
1.0000
1
1
1
1
0
1
0.4000
0
1
0
1
1
1
0.9750
1
1
1
1
1
0
0.3875
0
1
1
0
0
0
0.9500
1
1
1
1
1
1
0.3750
0
1
1
0
0
1
0.9250
0
1
1
0
1
0
0.9000
0
1
1
0
1
1
0.8750
0
1
1
1
0
0
0.8500
0
1
1
1
0
1
0.8250
0
1
1
1
1
0
0.8000
0
1
1
1
1
1
0.7750
1
0
0
0
0
0
0.7625
1
0
0
0
0
1
0.7500
1
0
0
0
1
0
0.7375
1
0
0
0
1
1
0.7250
1
0
0
1
0
0
0.7125
1
0
0
1
0
1
0.7000
1
0
0
1
1
0
0.6875
18
Voltage Regulation
The integrating compensation network shown in Figure 6
insures that the steady-state error in the output voltage is
limited only to the error in the reference voltage (output of
the DAC) and offset errors in the OFS current source,
remote-sense and error amplifiers. Intersil specifies the
guaranteed tolerance of the ISL6312 to include the
combined tolerances of each of these elements.
The output of the error amplifier, VCOMP, is compared to the
triangle waveform to generate the PWM signals. The PWM
signals control the timing of the Internal MOSFET drivers
and regulate the converter output so that the voltage at FB is
equal to the voltage at REF. This will regulate the output
voltage to be equal to Equation 8. The internal and external
circuitry that controls voltage regulation is illustrated in
Figure 6.
FN9289.5
February 25, 2010
ISL6312
V OUT = V REF – V OFS – V DROOP
(EQ. 8)
The ISL6312 incorporates an internal differential
remote-sense amplifier in the feedback path. The amplifier
removes the voltage error encountered when measuring the
output voltage relative to the controller ground reference
point resulting in a more accurate means of sensing output
voltage. Connect the microprocessor sense pins to the
non-inverting input, VSEN, and inverting input, RGND, of the
remote-sense amplifier. The remote-sense output, VDIFF, is
connected to the inverting input of the error amplifier through
an external resistor.
EXTERNAL CIRCUIT
CC
⎛ I OUT DCR
⎞
V OUT = V REF – V OFS – ⎜ ------------- ⋅ ------------------ ⋅ R FB⎟
R ISEN
⎝ N
⎠
REF
1k
ERROR
AMPLIFIER
FB
+
-
VCOMP
IOFS
IAVG
IDROOP
RFB
+
(VDROOP + VOFS)
-
VOUT+
VOUT-
V DROOP = I AVG ⋅ R FB
(EQ. 9)
The regulated output voltage is reduced by the droop voltage
VDROOP. The output voltage as a function of load current is
derived by combining Equations 7, 8, and 9.
VID DAC
CREF
As shown in Figure 6, a current proportional to the average
current of all active channels, IAVG, flows from FB through a
load-line regulation resistor RFB. The resulting voltage drop
across RFB is proportional to the output current, effectively
creating an output voltage droop with a steady-state value
defined as:
ISL6312 INTERNAL CIRCUIT
COMP
RC
negative spike can be sustained without crossing the lower
limit. By adding a well controlled output impedance, the
output voltage under load can effectively be level shifted
down so that a larger positive spike can be sustained without
crossing the upper specification limit.
(EQ. 10)
In Equation 10, VREF is the reference voltage, VOFS is the
programmed offset voltage, IOUT is the total output current
of the converter, RISEN is the internal sense resistor
connected to the ISEN+ pin, RFB is the feedback resistor, N
is the active channel number, and DCR is the Inductor DCR
value.
VDIFF
Therefore the equivalent loadline impedance, i.e. droop
impedance, is equal to Equation 11:
VSEN
R FB DCR
R LL = ------------ ⋅ -----------------N
R ISEN
(EQ. 11)
+
-
RGND
DIFFERENTIAL
REMOTE-SENSE
AMPLIFIER
FIGURE 6. OUTPUT VOLTAGE AND LOAD-LINE
REGULATION WITH OFFSET ADJUSTMENT
Load-Line (Droop) Regulation
Some microprocessor manufacturers require a
precisely-controlled output resistance. This dependence of
output voltage on load current is often termed “droop” or
“load line” regulation. By adding a well controlled output
impedance, the output voltage can effectively be level shifted
in a direction which works to achieve the load-line regulation
required by these manufacturers.
Output-Voltage Offset Programming
The ISL6312 allows the designer to accurately adjust the
offset voltage by connecting a resistor, ROFS, from the OFS
pin to VCC or GND. When ROFS is connected between OFS
and VCC, the voltage across it is regulated to 1.6V. This
causes a proportional current (IOFS) to flow into the FB pin.
If ROFS is connected to ground, the voltage across it is
regulated to 0.4V, and IOFS flows out of the FB pin. The
offset current flowing through the resistor between VDIFF
and FB will generate the desired offset voltage which is
equal to the product (IOFS x RFB). These functions are
shown in Figures 7 and 8.
Once the desired output offset voltage has been determined,
use the following formulas to set ROFS:
In other cases, the designer may determine that a more
cost-effective solution can be achieved by adding droop.
Droop can help to reduce the output-voltage spike that
results from fast load-current demand changes.
For Negative Offset (connect ROFS to GND):
The magnitude of the spike is dictated by the ESR and ESL
of the output capacitors selected. By positioning the no-load
voltage level near the upper specification limit, a larger
For Positive Offset (connect ROFS to VCC):
19
0.4 ⋅ R FB
R OFS = -------------------------V OFFSET
1.6 ⋅ R FB
R OFS = -------------------------V OFFSET
(EQ. 12)
(EQ. 13)
FN9289.5
February 25, 2010
ISL6312
a new code is established and it remains stable for 3
consecutive readings (1ms to 1.33ms), the ISL6312
recognizes the new code and changes the internal DAC
reference directly to the new level. The Intel processor
controls the VID transitions and is responsible for
incrementing or decrementing one VID step at a time. In
VR10 and VR11 settings, the ISL6312 will immediately
change the internal DAC reference to the new requested
value as soon as the request is validated, which means the
fastest recommended rate at which a bit change can occur is
once every 2ms. In cases where the reference step is too
large, the sudden change can trigger overcurrent or
overvoltage events.
FB
VOFS
+
RFB
E/A
IOFS
REF
1:1
CURRENT
MIRROR
VDIFF
IOFS
VCC
1.6V
+
ROFS
OFS
ISL6312
VCC
FIGURE 7. POSITIVE OFFSET OUTPUT VOLTAGE
PROGRAMMING
Assuming the microprocessor controls the VID change at 1
bit every TVID, the relationship between CREF and TVID is
given by Equation 14.
FB
+
VOFS
-
E/A
RFB
In order to ensure the smooth transition of output voltage
during a VR10 or VR11 VID change, a VID step change
smoothing network is required. This network is composed of
an internal 1kΩ resistor between the DAC and the REF pin,
and the external capacitor CREF, between the REF pin and
ground. The selection of CREF is based on the time duration
for 1 bit VID change and the allowable delay time.
C REF = 0.001 ( S ) ⋅ T VID
IOFS
As an example, for a VID step change rate of 5ms per bit,
the value of CREF is 5600pF based on Equation 14.
REF
VDIFF
(EQ. 14)
VCC
AMD DYNAMIC VID TRANSITIONS
1:1
CURRENT
MIRROR
IOFS
+
0.4V
OFS
ROFS
ISL6312
GND
GND
FIGURE 8. NEGATIVE OFFSET OUTPUT VOLTAGE
PROGRAMMING
Dynamic VID
Modern microprocessors need to make changes to their core
voltage as part of normal operation. They direct the ISL6312 to
do this by making changes to the VID inputs. The ISL6312 is
required to monitor the DAC inputs and respond to on-the-fly
VID changes in a controlled manner, supervising a safe output
voltage transition without discontinuity or disruption. The DAC
mode the ISL6312 is operating in determines how the controller
responds to a dynamic VID change.
INTEL DYNAMIC VID TRANSITIONS
When in Intel VR10 or VR11 mode the ISL6312 checks the
VID inputs on the positive edge of an internal 3MHz clock. If
20
When running in AMD 5-bit or 6-bit modes of operation, the
ISL6312 responds differently to a dynamic VID change then
when in Intel VR10 or VR11 mode. In the AMD modes the
ISL6312 still checks the VID inputs on the positive edge of
an internal 3MHz clock. In these modes the VID code can be
changed by more than a 1-bit step at a time. If a new code is
established and it remains stable for 3 consecutive readings
(1ms to 1.33ms), the ISL6312 recognizes the change and
begins slewing the DAC in 6.25mV steps at a stepping
frequency of 330kHz until the VID and DAC are equal. Thus,
the total time required for a VID change, tDVID, is dependent
only on the size of the VID change (DVVID).
The time required for a ISL6312-based converter in AMD
5-bit DAC configuration to make a 1.1V to 1.5V reference
voltage change is about 194ms, as calculated using
Equation 15.
ΔV VID
1
t DVID = -------------------------- ⋅ ⎛ ---------------------⎞
3 ⎝ 0.00625⎠
330 × 10
(EQ. 15)
In order to ensure the smooth transition of output voltage
during an AMD VID change, a VID step change smoothing
network is required. This network is composed of an internal
1kΩ resistor between the DAC and the REF pin, and the
external capacitor CREF, between the REF pin and ground.
For AMD VID transitions CREF should be a 1000pF
capacitor.
FN9289.5
February 25, 2010
ISL6312
The ISL6312 integrated drivers incorporate two different
adaptive deadtime control techniques, which the user can
choose between. Both of these control techniques help to
minimize deadtime, resulting in high efficiency from the reduced
freewheeling time of the lower MOSFET body-diode
conduction, and both help to prevent the upper and lower
MOSFETs from conducting simultaneously. This is
accomplished by ensuring either rising gate turns on its
MOSFET with minimum and sufficient delay after the other has
turned off.
The difference between the two adaptive deadtime control
techniques is the method in which they detect that the lower
MOSFET has transitioned off in order to turn on the upper
MOSFET. The state of the DRSEL pin chooses which of the
two control techniques is active. By tying the DRSEL pin
directly to ground, the PHASE Detect Scheme is chosen,
which monitors the voltage on the PHASE pin to determine if
the lower MOSFET has transitioned off or not. Tying the
DRSEL pin to VCC though a 50kΩ resistor selects the
LGATE Detect Scheme, which monitors the voltage on the
LGATE pin to determine if the lower MOSFET has turned off
or not. For both schemes, the method for determining
whether the upper MOSFET has transitioned off in order to
signal to turn on the lower MOSFET is the same.
Once the PHASE is high, the advanced adaptive
shoot-through circuitry monitors the PHASE and UGATE
voltages during a PWM falling edge and the subsequent
UGATE turn-off. If either the UGATE falls to less than 1.75V
above the PHASE or the PHASE falls to less than +0.8V, the
LGATE is released to turn on.
Internal Bootstrap Device
All three integrated drivers feature an internal bootstrap
schottky diode. Simply adding an external capacitor across
the BOOT and PHASE pins completes the bootstrap circuit.
The bootstrap function is also designed to prevent the
bootstrap capacitor from overcharging due to the large
negative swing at the PHASE node. This reduces voltage
stress on the boot to phase pins.
1.6
1.4
1.2
CBOOT_CAP (µF)
User Selectable Adaptive Deadtime Control
Techniques
1.0
0.8
0.6
QGATE = 100nC
0.4
50nC
PHASE DETECT
0.2
If the DRSEL pin is tied directly to ground, the PHASE Detect
adaptive deadtime control technique is selected. For the
PHASE detect scheme, during turn-off of the lower MOSFET,
the PHASE voltage is monitored until it reaches a -0.3V/+0.8V
(forward/reverse inductor current). At this time the UGATE is
released to rise. An auto-zero comparator is used to correct the
rDS(ON) drop in the phase voltage preventing false detection of
the -0.3V phase level during rDS(ON) conduction period. In the
case of zero current, the UGATE is released after 35ns delay of
the LGATE dropping below 0.5V. When LGATE first begins to
transition low, this quick transition can disturb the PHASE node
and cause a false trip, so there is 20ns of blanking time once
LGATE falls until PHASE is monitored.
0.0
0.0
Once the PHASE is high, the advanced adaptive
shoot-through circuitry monitors the PHASE and UGATE
voltages during a PWM falling edge and the subsequent
UGATE turn-off. If either the UGATE falls to less than 1.75V
above the PHASE or the PHASE falls to less than +0.8V, the
LGATE is released to turn-on.
20nC
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.9
1.0
ΔVBOOT_CAP (V)
FIGURE 9. BOOTSTRAP CAPACITANCE vs BOOT RIPPLE
VOLTAGE
The bootstrap capacitor must have a maximum voltage
rating above PVCC + 4V and its capacitance value can be
chosen from Equation 16: where QG1 is the amount of gate
charge per upper MOSFET at VGS1 gate-source voltage and
NQ1 is the number of control MOSFETs. The ΔVBOOT_CAP
term is defined as the allowable droop in the rail of the upper
gate drive.
Q GATE
C BOOT_CAP ≥ -------------------------------------ΔV BOOT_CAP
(EQ. 16)
Q G1 ⋅ PVCC
Q GATE = ---------------------------------- ⋅ N Q1
V GS1
LGATE DETECT
Gate Drive Voltage Versatility
If the DRSEL pin is tied to VCC through a 50kΩ resistor, the
LGATE Detect adaptive deadtime control technique is selected.
For the LGATE detect scheme, during turn-off of the lower
MOSFET, the LGATE voltage is monitored until it reaches
1.75V. At this time the UGATE is released to rise.
The ISL6312 provides the user flexibility in choosing the
gate drive voltage for efficiency optimization. The controller
ties the upper and lower drive rails together. Simply applying
a voltage from 5V up to 12V on PVCC sets both gate drive
rail voltages simultaneously.
21
FN9289.5
February 25, 2010
ISL6312
Initialization
Prior to initialization, proper conditions must exist on the EN,
VCC, PVCC and the VID pins. When the conditions are met,
the controller begins soft-start. Once the output voltage is
within the proper window of operation, the controller asserts
PGOOD.
ISL6312 INTERNAL CIRCUIT
EXTERNAL CIRCUIT
VCC
PVCC1
+12V
POR
CIRCUIT
ENABLE
COMPARATOR
10.7kΩ
EN
+
-
1.40kΩ
0.85V
In order for the ISL6312 to begin operation, PVCC1 is the
only pin that is required to have a voltage applied that
exceeds POR. However, for 2 or 3-phase operation
PVCC2 and PVCC3 must also exceed the POR
threshold. Hysteresis between the rising and falling
thresholds assure that once enabled, the ISL6312 will not
inadvertently turn off unless the PVCC bias voltage drops
substantially (see “Electrical Specifications” on page 6).
For Intel VR10, VR11 and AMD 6-bit modes of operation
these are the only conditions that must be met for the
controller to immediately begin the soft-start sequence. If
running in AMD 5-bit mode of operation there is one more
condition that must be met:
5. The VID code must not be 11111 in AMD 5-bit mode. This
code signals the controller that no load is present. The
controller will not allow soft-start to begin if this VID code
is present on the VID pins.
Once all of these conditions are met the controller will begin
the soft-start sequence and will ramp the output voltage up
to the user designated level.
Intel Soft-Start
+
SOFT-START
AND
FAULT LOGIC
EN_PH4
-
1.21V
FIGURE 10. POWER SEQUENCING USING THRESHOLDSENSITIVE ENABLE (EN) FUNCTION
Enable and Disable
While in shutdown mode, the PWM outputs are held in a
high-impedance state to assure the drivers remain off. The
following input conditions must be met, for both Intel and
AMD modes of operation, before the ISL6312 is released
from shutdown mode to begin the soft-start start-up
sequence:
1. The bias voltage applied at VCC must reach the internal
power-on reset (POR) rising threshold. Once this
threshold is reached, proper operation of all aspects of
the ISL6312 is guaranteed. Hysteresis between the rising
and falling thresholds assure that once enabled, the
ISL6312 will not inadvertently turn off unless the bias
voltage drops substantially (see “Electrical
Specifications” on page 6).
The soft-start function allows the converter to bring up the
output voltage in a controlled fashion, resulting in a linear
ramp-up. The soft-start sequence for the Intel modes of
operation is slightly different then the AMD soft-start
sequence.
For the Intel VR10 and VR11 modes of operation, the
soft-start sequence if composed of four periods, as shown in
Figure 11. Once the ISL6312 is released from shutdown and
soft-start begins (as described in “Enable and Disable” on
page 22), the controller will have fixed delay period TD1.
After this delay period, the VR will begin first soft-start ramp
until the output voltage reaches 1.1V VBOOT voltage. Then,
the controller will regulate the VR voltage at 1.1V for another
fixed period TD3. At the end of TD3 period, ISL6312 will
read the VID signals. If the VID code is valid, ISL6312 will
initiate the second soft-start ramp until the output voltage
reaches the VID voltage plus/minus any offset or droop
voltage.
The soft-start time is the sum of the 4 periods as shown in
Equation 17.
T SS = TD1 + TD2 + TD3 + TD4
(EQ. 17)
2. The voltage on EN must be above 0.85V. The EN input
allows for power sequencing between the controller bias
voltage and another voltage rail. The enable comparator
holds the ISL6312 in shutdown until the voltage at EN
rises above 0.85V. The enable comparator has 110mV of
hysteresis to prevent bounce.
3. The voltage on the EN_PH4 pin must be above 1.21V.
The EN_PH4 input allows for power sequencing between
the controller and the external driver.
4. The driver bias voltage applied at the PVCC pins must
reach the internal power-on reset (POR) rising threshold.
22
FN9289.5
February 25, 2010
ISL6312
.
V VID
1
TDB = -------------------------- ⋅ ⎛ ---------------------⎞
3 ⎝ 0.00625⎠
330 × 10
VOUT, 500mV/DIV
TD1
TD2
(EQ. 20)
After the DAC voltage reaches the final VID setting, PGOOD
will be set to high with the fixed delay TDC. The typical value
for TDC can range between 1.5ms and 3.0ms.
TD3 TD4
TD5
VOUT, 500mV/DIV
EN_VTT
PGOOD
TDB
TDA
TDC
500µs/DIV
FIGURE 11. SOFT-START WAVEFORMS
EN_VTT
TD1 is a fixed delay with the typical value as 1.40ms. TD3 is
determined by the fixed 85µs plus the time to obtain valid
VID voltage. If the VID is valid before the output reaches the
1.1V, the minimum time to validate the VID input is 500ns.
Therefore the minimum TD3 is about 86µs.
During TD2 and TD4, ISL6312 digitally controls the DAC
voltage change at 6.25mV per step. The time for each step is
determined by the frequency of the soft-start oscillator which
is defined by the resistor RSS from SS pin to GND. The
second soft-start ramp time TD2 and TD4 can be calculated
based on Equations 18 and 19:
1.1 ⋅ R SS
TD2 = ------------------------ ( μs )
6.25 ⋅ 25
(EQ. 18)
( V VID – 1.1 ) ⋅ R SS
TD4 = ---------------------------------------------------- ( μs )
6.25 ⋅ 25
(EQ. 19)
PGOOD
500µs/DIV
FIGURE 12. SOFT-START WAVEFORMS
Pre-Biased Soft-Start
The ISL6312 also has the ability to start up into a
pre-charged output, without causing any unnecessary
disturbance. The FB pin is monitored during soft-start, and
should it be higher than the equivalent internal ramping
reference voltage, the output drives hold both MOSFETs off.
OUTPUT PRECHARGED
ABOVE DAC LEVEL
For example, when VID is set to 1.5V and the RSS is set at
100kΩ, the first soft-start ramp time TD2 will be 704µs and
the second soft-start ramp time TD4 will be 256µs.
OUTPUT PRECHARGED
BELOW DAC LEVEL
NOTE: If the SS pin is grounded, the soft-start ramp in TD2
and TD4 will be defaulted to a 6.25mV step frequency of
330kHz.
GND>
After the DAC voltage reaches the final VID setting, PGOOD
will be set to high with the fixed delay TD5. The typical value
for TD5 is 440µs.
GND>
VOUT (0.5V/DIV)
EN (5V/DIV)
T1 T2
AMD Soft-Start
For the AMD 5-bit and 6-bit modes of operation, the
soft-start sequence is composed of three periods, as shown
in Figure 12. At the beginning of soft-start, the VID code is
immediately obtained from the VID pins, followed by a fixed
delay period TDA. After this delay period the ISL6312 will
begin ramping the output voltage to the desired DAC level at
a fixed rate of 6.25mV per step, with a stepping frequency of
330kHz. The amount of time required to ramp the output
voltage to the final DAC voltage is referred to as TDB, and
can be calculated as shown in Equation 20:
23
T3
FIGURE 13. SOFT-START WAVEFORMS FOR ISL6312-BASED
MULTIPHASE CONVERTER
Once the internal ramping reference exceeds the FB pin
potential, the output drives are enabled, allowing the output
to ramp from the pre-charged level to the final level dictated
by the DAC setting. Should the output be pre-charged to a
level exceeding the DAC setting, the output drives are
enabled at the end of the soft-start period, leading to an
abrupt correction in the output voltage down to the DAC-set
level.
FN9289.5
February 25, 2010
ISL6312
Fault Monitoring and Protection
Overvoltage Protection
The ISL6312 actively monitors output voltage and current to
detect fault conditions. Fault monitors trigger protective
measures to prevent damage to a microprocessor load. One
common power good indicator is provided for linking to
external system monitors. The schematic in Figure 14
outlines the interaction between the fault monitors and the
power good signal.
The ISL6312 constantly monitors the sensed output voltage
on the VDIFF pin to detect if an overvoltage event occurs.
When the output voltage rises above the OVP trip level
actions are taken by the ISL6312 to protect the
microprocessor load. The overvoltage protection trip level
changes depending on what mode of operation the controller
is in and what state the OVPSEL and VRSEL pins are in.
Tables 6 and 7 list what the OVP trip levels are under all
conditions.
-
170µA
OCL
+
I1
REPEAT FOR
EACH CHANNEL
VDAC
VRSEL
-
+175mV,
+250mV,
+350mV
125µA
OCP
+
OVPSEL
IAVG
SOFT-START, FAULT
AND CONTROL LOGIC
VOVP
At the inception of an overvoltage event, LGATE1, LGATE2
and LGATE3 are commanded high, PWM4 is commanded
low, and the PGOOD signal is driven low. This turns on the
all of the lower MOSFETs and pulls the output voltage below
a level that might cause damage to the load. The LGATE
outputs remain high and PWM4 remains low until VDIFF falls
100mV below the OVP threshold that tripped the overvoltage
protection circuitry. The ISL6312 will continue to protect the
load in this fashion as long as the overvoltage condition
recurs. Once an overvoltage condition ends the ISL6312
latches off, and must be reset by toggling EN, or through
POR, before a soft-start can be reinitiated.
TABLE 6. INTEL VR10 AND VR11 OVP THRESHOLDS
MODE OF
OPERATION
VSEN
+
+
OV
PGOOD
x1
-
OVPSEL PIN OPEN
OR TIED TO GND
OVPSEL PIN TIED
TO VCC
Soft-Start
(TD1 and TD2)
1.280V and
VDAC + 175mV
(higher of the two)
1.280V and
VDAC + 350mV
(higher of the two)
Soft-Start
(TD3 and TD4)
VDAC + 175mV
VDAC + 350mV
Normal Operation
VDAC + 175mV
VDAC + 350mV
-
RGND
+
UV
VDIFF
0.60 x DAC
ISL6312 INTERNAL CIRCUITRY
TABLE 7. AMD OVP THRESHOLDS
FIGURE 14. POWER GOOD AND PROTECTION CIRCUITRY
Power Good Signal
The power good pin (PGOOD) is an open-drain logic output
that signals whether or not the ISL6312 is regulating the
output voltage within the proper levels, and whether any fault
conditions exist. This pin should be tied to a +5V source
through a resistor.
During shutdown and soft-start PGOOD pulls low and
releases high after a successful soft-start and the output
voltage is operating between the undervoltage and
overvoltage limits. PGOOD transitions low when an
undervoltage, overvoltage, or overcurrent condition is
detected or when the controller is disabled by a reset from
EN, EN_PH4, POR, or one of the no-CPU VID codes. In the
event of an overvoltage or overcurrent condition, the
controller latches off and PGOOD will not return high until
after a successful soft-start. In the case of an undervoltage
event, PGOOD will return high when the output voltage
returns to within the undervoltage.
24
MODE OF
OPERATION
Soft-Start
Normal Operation
OVPSEL PIN OPEN
OR TIED TO GND
OVPSEL PIN TIED
TO VCC
2.200V and
VDAC + 250mV
(higher of the two)
2.200V and
VDAC + 350mV
(higher of the two)
VDAC + 250mV
VDAC + 350mV
One exception that overrides the overvoltage protection
circuitry is a dynamic VID transition in AMD modes of
operation. If a new VID code is detected during normal
operation, the OVP protection circuitry is disabled from the
beginning of the dynamic VID transition, until 50µs after the
internal DAC reaches the final VID setting. This is the only
time during operation of the ISL6312 that the OVP circuitry is
not active.
Pre-POR Overvoltage Protection
Prior to PVCC and VCC exceeding their POR levels, the
ISL6312 is designed to protect the load from any overvoltage
events that may occur. This is accomplished by means of an
internal 10kΩ resistor tied from PHASE to LGATE, which
FN9289.5
February 25, 2010
ISL6312
turns on the lower MOSFET to control the output voltage
until the overvoltage event ceases or the input power supply
cuts off. For complete protection, the low side MOSFET
should have a gate threshold well below the maximum
voltage rating of the load/microprocessor.
In the event that during normal operation the PVCC or VCC
voltage falls back below the POR threshold, the pre-POR
overvoltage protection circuitry reactivates to protect from
any more pre-POR overvoltage events.
Undervoltage Detection
The undervoltage threshold is set at 60% of the VID code.
When the output voltage (VSEN-RGND) is below the
undervoltage threshold, PGOOD gets pulled low. No other
action is taken by the controller. PGOOD will return high if
the output voltage rises above 70% of the VID code.
Open Sense Line Prevention
In the case that either of the remote sense lines, VSEN or
GND, become open, the ISL6312 is designed to prevent the
controller from regulating. This is accomplished by means of
a small 5µA pull-up current on VSEN, and a pull-down
current on RGND. If the sense lines are opened at any time,
the voltage difference between VSEN and RGND will
increase until an overvoltage event occurs, at which point
overvoltage protection activates and the controller stops
regulating. The ISL6312 will be latched off and cannot be
restarted until the controller is reset.
–6
⎛ 125 ⋅ 10 ⋅ R ISEN ⋅ N⎞ ⎛ R 1 + R 2⎞
I OCP = ⎜ ----------------------------------------------------------⎟ ⋅ ⎜ ---------------------⎟
DCR
⎝
⎠ ⎝ R2 ⎠
(EQ. 22)
I OCP > I OCP, min
The overcurrent trip level of the ISL6312 cannot be set any
lower then the IOCP,min level calculated above. If an
overcurrent trip level lower then IOCP,min is desired, then the
ISL6312A should be used in the place of the ISL6312.
At the beginning of overcurrent shutdown, the controller sets
all of the UGATE and LGATE signals low, puts PWM4 in a
high-impedance state, and forces PGOOD low. This turns off
all of the upper and lower MOSFETs. The system remains in
this state for fixed period of 12ms. If the controller is still
enabled at the end of this wait period, it will attempt a softstart. If the fault remains, the trip-retry cycles will continue
indefinitely until either the controller is disabled or the fault is
cleared. Note that the energy delivered during trip-retry
cycling is much less than during full-load operation, so there
is no thermal hazard.
OUTPUT CURRENT, 50A/DIV
0A
Overcurrent Protection
The ISL6312 takes advantage of the proportionality between
the load current and the average current, IAVG, to detect an
overcurrent condition. See “Continuous Current Sampling”
on page 12 for more detail on how the average current is
measured. The average current is continually compared with
a constant 125µA OCP reference current as shown in
Figure 14. Once the average current exceeds the OCP
reference current, a comparator triggers the converter to
begin overcurrent protection procedures.
This method for detecting overcurrent events limits the
minimum overcurrent trip threshold because of the fact the
ISL6312 uses set internal RISEN current sense resistors.
The minimum overcurrent trip threshold is dictated by the
DCR of the inductors and the number of active channels. To
calculate the minimum overcurrent trip level, IOCP,min, use
Equation 21, where N is the number of active channels, DCR
is the individual inductor’s DCR, and RISEN is the 300Ω
internal current sense resistor.
–6
125 ⋅ 10 ⋅ R ISEN ⋅ N
I OCP, min = --------------------------------------------------------DCR
(EQ. 21)
OUTPUT VOLTAGE,
500mV/DIV
0V
3ms/DIV
FIGURE 15. OVERCURRENT BEHAVIOR IN HICCUP MODE
Individual Channel Overcurrent Limiting
The ISL6312 has the ability to limit the current in each
individual channel without shutting down the entire regulator.
This is accomplished by continuously comparing the sensed
currents of each channel with a constant 170μA OCL
reference current as shown in Figure 14. If a channel’s
individual sensed current exceeds this OCL limit, the UGATE
signal of that channel is immediately forced low, and the
LGATE signal is forced high. This turns off the upper
MOSFET(s), turns on the lower MOSFET(s), and stops the
rise of current in that channel, forcing the current in the
channel to decrease. That channel’s UGATE signal will not
be able to return high until the sensed channel current falls
back below the 170µA reference.
If the desired overcurrent trip level is greater then the
minimum overcurrent trip level, IOCP,min, then the resistor
divider R-C circuit around the inductor shown in Figure 5
should be used to set the desired trip level.
25
FN9289.5
February 25, 2010
ISL6312
General Design Guide
This design guide is intended to provide a high-level
explanation of the steps necessary to create a multiphase
power converter. It is assumed that the reader is familiar with
many of the basic skills and techniques referenced below. In
addition to this guide, Intersil provides complete reference
designs that include schematics, bills of materials, and example
board layouts for all common microprocessor applications.
Power Stages
The first step in designing a multiphase converter is to
determine the number of phases. This determination
depends heavily on the cost analysis which in turn depends
on system constraints that differ from one design to the next.
Principally, the designer will be concerned with whether
components can be mounted on both sides of the circuit
board, whether through-hole components are permitted, the
total board space available for power-supply circuitry, and
the maximum amount of load current. Generally speaking,
the most economical solutions are those in which each
phase handles between 25A and 30A. All surface-mount
designs will tend toward the lower end of this current range.
If through-hole MOSFETs and inductors can be used, higher
per-phase currents are possible. In cases where board
space is the limiting constraint, current can be pushed as
high as 40A per phase, but these designs require heat sinks
and forced air to cool the MOSFETs, inductors and heatdissipating surfaces.
P
LOW, 2
= V
D ( ON )
⋅f
S
⎛I
⎞
⎛I
⎞
⎟
M I PP⎟ ⋅ t
M I--------⋅ ⎜ -----+ ⎜ -----+ ---------– PP-⎟ ⋅ t d2
2 ⎠
⎝N
2 ⎠ d1 ⎜⎝ N
(EQ. 24)
The total maximum power dissipated in each lower MOSFET
is approximated by the summation of PLOW,1 and PLOW,2.
UPPER MOSFET POWER CALCULATION
In addition to rDS(ON) losses, a large portion of the
upper-MOSFET losses are due to currents conducted
across the input voltage (VIN) during switching. Since a
substantially higher portion of the upper-MOSFET losses are
dependent on switching frequency, the power calculation is
more complex. Upper MOSFET losses can be divided into
separate components involving the upper-MOSFET
switching times, the lower-MOSFET body-diode
reverse-recovery charge, Qrr, and the upper MOSFET
rDS(ON) conduction loss.
When the upper MOSFET turns off, the lower MOSFET does
not conduct any portion of the inductor current until the
voltage at the phase node falls below ground. Once the
lower MOSFET begins conducting, the current in the upper
MOSFET falls to zero as the current in the lower MOSFET
ramps up to assume the full inductor current. In Equation 25,
the required time for this commutation is t1 and the
approximated associated power loss is PUP,1.
I M I PP⎞ ⎛ t 1 ⎞
P UP,1 ≈ V IN ⋅ ⎛ ----- ⋅ ⎜ ---- ⎟ ⋅ f
⎝ N- + -------2 ⎠ ⎝ 2⎠ S
(EQ. 25)
MOSFETS
The choice of MOSFETs depends on the current each
MOSFET will be required to conduct, the switching frequency,
the capability of the MOSFETs to dissipate heat, and the
availability and nature of heat sinking and air flow.
LOWER MOSFET POWER CALCULATION
The calculation for power loss in the lower MOSFET is
simple, since virtually all of the loss in the lower MOSFET is
due to current conducted through the channel resistance
(rDS(ON)). In Equation 23, IM is the maximum continuous
output current, IPP is the peak-to-peak inductor current (see
Equation 1), and d is the duty cycle (VOUT/VIN).
I L, 2PP ⋅ ( 1 – d )
⎛ I M⎞ 2
P LOW, 1 = r DS ( ON ) ⋅ ⎜ -----⎟ ⋅ ( 1 – d ) + ------------------------------------12
⎝ N⎠
(EQ. 23)
An additional term can be added to the lower-MOSFET loss
equation to account for additional loss accrued during the
dead time when inductor current is flowing through the
lower-MOSFET body diode. This term is dependent on the
diode forward voltage at IM, VD(ON), the switching
frequency, fS, and the length of dead times, td1 and td2, at
the beginning and the end of the lower-MOSFET conduction
interval respectively.
26
At turn on, the upper MOSFET begins to conduct and this
transition occurs over a time t2. In Equation 26, the
approximate power loss is PUP,2.
⎛ I M I PP⎞ ⎛ t 2 ⎞
P UP, 2 ≈ V IN ⋅ ⎜ ----- – ---------⎟ ⋅ ⎜ ---- ⎟ ⋅ f S
2 ⎠ ⎝ 2⎠
⎝N
(EQ. 26)
A third component involves the lower MOSFET reverserecovery charge, Qrr. Since the inductor current has fully
commutated to the upper MOSFET before the lowerMOSFET body diode can recover all of Qrr, it is conducted
through the upper MOSFET across VIN. The power
dissipated as a result is PUP,3.
P UP,3 = V IN ⋅ Q rr ⋅ f S
(EQ. 27)
Finally, the resistive part of the upper MOSFET is given in
Equation 28 as PUP,4.
2
⎛ I M⎞
I PP2
P UP,4 ≈ r DS ( ON ) ⋅ d ⋅ ⎜ -----⎟ + ---------12
⎝ N⎠
(EQ. 28)
FN9289.5
February 25, 2010
ISL6312
The total power dissipated by the upper MOSFET at full load
can now be approximated as the summation of the results
from Equations 25, 26, 27 and 28. Since the power
equations depend on MOSFET parameters, choosing the
correct MOSFETs can be an iterative process involving
repetitive solutions to the loss equations for different
MOSFETs and different switching frequencies.
PVCC
BOOT
D
CGD
RHI1
G
UGATE
RLO1
RG1
CDS
RGI1
Package Power Dissipation
CGS
When choosing MOSFETs it is important to consider the
amount of power being dissipated in the integrated drivers
located in the controller. Since there are a total of three
drivers in the controller package, the total power dissipated
by all three drivers must be less than the maximum
allowable power dissipation for the QFN package.
Calculating the power dissipation in the drivers for a desired
application is critical to ensure safe operation. Exceeding the
maximum allowable power dissipation level will push the IC
beyond the maximum recommended operating junction
temperature of +125°C. The maximum allowable IC power
dissipation for the 7x7 QFN package is approximately 3.5W
at room temperature. See “Layout Considerations” on
page 32 for thermal transfer improvement suggestions.
Q1
S
PHASE
FIGURE 16. TYPICAL UPPER-GATE DRIVE TURN-ON PATH
PVCC
D
CGD
RHI2
LGATE
RLO2
G
RG2
CDS
RGI2
CGS
Q2
S
When designing the ISL6312 into an application, it is
recommended that the following calculation is used to
ensure safe operation at the desired frequency for the
selected MOSFETs. The total gate drive power losses,
PQg_TOT, due to the gate charge of MOSFETs and the
integrated driver’s internal circuitry and their corresponding
average driver current can be estimated with Equations 29
and 30, respectively.
P Qg_TOT = P Qg_Q1 + P Qg_Q2 + I Q ⋅ VCC
(EQ. 29)
3
P Qg_Q1 = --- ⋅ Q G1 ⋅ PVCC ⋅ F SW ⋅ N Q1 ⋅ N PHASE
2
P Qg_Q2 = Q G2 ⋅ PVCC ⋅ F SW ⋅ N Q2 ⋅ N PHASE
(EQ. 30)
3
I DR = ⎛ --- ⋅ Q G1 ⋅ N
+ Q G2 ⋅ N Q2⎞ ⋅ N PHASE ⋅ F SW + I Q
⎝2
⎠
Q1
In Equations 29 and 30, PQg_Q1 is the total upper gate drive
power loss and PQg_Q2 is the total lower gate drive power
loss; the gate charge (QG1 and QG2) is defined at the
particular gate to source drive voltage PVCC in the
corresponding MOSFET data sheet; IQ is the driver total
quiescent current with no load at both drive outputs; NQ1
and NQ2 are the number of upper and lower MOSFETs per
phase, respectively; NPHASE is the number of active
phases. The IQ*VCC product is the quiescent power of the
controller without capacitive load and is typically 75mW at
300kHz.
27
FIGURE 17. TYPICAL LOWER-GATE DRIVE TURN-ON PATH
The total gate drive power losses are dissipated among the
resistive components along the transition path and in the
bootstrap diode. The portion of the total power dissipated in
the controller itself is the power dissipated in the upper drive
path resistance, PDR_UP, the lower drive path resistance,
PDR_UP, and in the boot strap diode, PBOOT. The rest of the
power will be dissipated by the external gate resistors (RG1
and RG2) and the internal gate resistors (RGI1 and RGI2) of
the MOSFETs. Figures 16 and 17 show the typical upper
and lower gate drives turn-on transition path. The total power
dissipation in the controller itself, PDR, can be roughly
estimated as:
P DR = P DR_UP + P DR_LOW + P BOOT + ( I Q ⋅ VCC )
(EQ. 31)
P Qg_Q1
P BOOT = --------------------3
R LO1
R HI1
⎛
⎞ P Qg_Q1
P DR_UP = ⎜ -------------------------------------- + ----------------------------------------⎟ ⋅ --------------------R
+
R
R
+
R
3
⎝ HI1
EXT1
LO1
EXT1⎠
R HI2
R LO2
⎛
⎞ P Qg_Q2
P DR_LOW = ⎜ -------------------------------------- + ----------------------------------------⎟ ⋅ --------------------2
⎝ R HI2 + R EXT2 R LO2 + R EXT2⎠
R GI1
R EXT1 = R G1 + ------------N
Q1
R GI2
R EXT2 = R G2 + ------------N
Q2
FN9289.5
February 25, 2010
ISL6312
Inductor DCR Current Sensing Component
Selection
The ISL6312 senses each individual channel’s inductor
current by detecting the voltage across the output inductor
DCR of that channel (as described in the “Continuous
Current Sampling” on page 12). As Figure 18 illustrates, an
R-C network is required to accurately sense the inductor
DCR voltage and convert this information into a current,
which is proportional to the total output current. The time
constant of this R-C network must match the time constant
of the inductor L/DCR.
VIN
I
L
UGATE(n)
L
MOSFET
LGATE(n)
+
VL(s)
R1
COUT
VC(s)
-
+
INDUCTOR
VOUT
-
DRIVER
DCR
C1
L
R 1 = ------------------------DCR ⋅ C 1
I OCP = I OCP, min
If the desired overcurrent trip level, IOCP, is greater then the
minimum overcurrent trip level, IOCP,min, then a resistor
divider R-C circuit should be used to set the desired trip
level. Follow the steps below to choose the component
values for the resistor divider R-C current sensing
network:
1. Choose an arbitrary value for C1. The recommended
value is 0.1μF.
2. Plug the inductor L and DCR component values, the
value for C1 chosen in step 1, the number of active
channels N, and the desired overcurrent protection level
IOCP into Equations 34 and 35 to calculate the values for
R1 and R2.
I OCP > I OCP, min
L ⋅ I OCP
R 2 = --------------------------------------------------------------------------------C 1 ⋅ ( I OCP ⋅ DCR – 0.0375 ⋅ N )
In
SAMPLE
-
VC(s)
RISEN
-
+
+
ISEN-(n)
ISEN+(n)
*R2 is OPTIONAL
ISEN
(EQ. 34)
(EQ. 35)
Due to errors in the inductance or DCR it may be necessary
to adjust the value of R1 and R2 to match the time constants
correctly. The effects of time constant mismatch can be seen
in the form of droop overshoot or undershoot during the
initial load transient spike, as shown in Figure 19. Follow the
steps below to ensure the R-C and inductor L/DCR time
constants are matched accurately.
1. Capture a transient event with the oscilloscope set to
about L/DCR/2 (sec/div). For example, with L = 1μH and
DCR = 1mΩ, set the oscilloscope to 500μs/div.
FIGURE 18. DCR SENSING CONFIGURATION
The R-C network across the inductor also sets the
overcurrent trip threshold for the regulator. Before the R-C
components can be selected, the desired overcurrent
protection level should be chosen. The minimum overcurrent
trip threshold the controller can support is dictated by the
DCR of the inductors and the number of active channels. To
calculate the minimum overcurrent trip level, IOCP,min, use
Equation 32, where N is the number of active channels, and
DCR is the individual inductor’s DCR.
0.0375 ⋅ N
I OCP, min = --------------------------DCR
(EQ. 33)
3. Resistor R2 should be left unpopulated.
L ⋅ I OCP
R 1 = -------------------------------------C 1 ⋅ 0.0375 ⋅ N
R2*
ISL6312 INTERNAL CIRCUIT
2. Plug the inductor L and DCR component values, and the
value for C1 chosen in step 1, into Equation 33 to
calculate the value for R1.
2. Record ΔV1 and ΔV2 as shown in Figure 19.
ΔV2
ΔV1
VOUT
(EQ. 32)
The overcurrent trip level of the ISL6312 cannot be set any
lower then the IOCP,min level calculated above. If the
minimum overcurrent trip level is desired, follow the
steps below to choose the component values for the
R-C current sensing network:
1. Choose an arbitrary value for C1. The recommended
value is 0.1µF.
28
ITRAN
ΔI
FIGURE 19. TIME CONSTANT MISMATCH BEHAVIOR
FN9289.5
February 25, 2010
ISL6312
3. Select new values, R1,NEW and R2,NEW, for the time
constant resistors based on the original values, R1,OLD
and R2,OLD, using Equations 36 and 37.
ΔV 1
R 1, NEW = R 1, OLD ⋅ ---------ΔV
(EQ. 36)
ΔV 1
R 2, NEW = R 2, OLD ⋅ ---------ΔV
(EQ. 37)
2
2
COMPENSATION WITH LOAD-LINE REGULATION
The load-line regulated converter behaves in a similar
manner to a peak current mode controller because the two
poles at the output filter L-C resonant frequency split with the
introduction of current information into the control loop. The
final location of these poles is determined by the system
function, the gain of the current signal, and the value of the
compensation components, RC and CC.
C2 (OPTIONAL)
4. Replace R1 and R2 with the new values and check to see
that the error is corrected. Repeat the procedure if
necessary.
RC
Loadline Regulation Resistor
If loadline regulation is desired, the IDROOP pin should be
shorted to the FB pin in order for the internal average
sense current to flow out across the loadline regulation
resistor, labeled RFB in Figure 6. This resistor’s value sets
the desired loadline required for the application. The
desired loadline, RLL, can be calculated by Equation 38
where VDROOP is the desired droop voltage at the full load
current IFL.
V DROOP
R LL = -----------------------I FL
CC
COMP
FB
ISL6312
IDROOP
RFB
VDIFF
FIGURE 20. COMPENSATION CONFIGURATION FOR
LOAD-LINE REGULATED ISL6312 CIRCUIT
(EQ. 38)
Based on the desired loadline, the loadline regulation
resistor, RFB, can be calculated from Equation 39 or
Equation 40, depending on the R-C current sense circuitry
being employed. If a basic R-C sense circuit consisting of C1
and R1 is being used, use Equation 39. If a resistor divider
R-C sense circuit consisting of R1, R2, and C1 is being
used, use Equation 40.
R LL ⋅ N ⋅ 300
R FB = --------------------------------DCR
(EQ. 39)
R LL ⋅ N ⋅ 300 ⋅ ( R 1 + R 2 )
R FB = ---------------------------------------------------------------DCR ⋅ R 2
(EQ. 40)
In Equations 39 and 40, RLL is the loadline resistance; N is
the number of active channels; DCR is the DCR of the
individual output inductors; and R1 and R2 are the current
sense R-C resistors.
If no loadline regulation is required, the IDROOP pin should
be left open and not connected to anything. To choose the
value for RFB in this situation, please refer to the
“COMPENSATION WITHOUT LOAD-LINE REGULATION”
on page 30.
Compensation
The two opposing goals of compensating the voltage
regulator are stability and speed. Depending on whether the
regulator employs the optional load-line regulation as
described in Load-Line Regulation, there are two distinct
methods for achieving these goals.
29
Since the system poles and zero are affected by the values
of the components that are meant to compensate them, the
solution to the system equation becomes fairly complicated.
Fortunately, there is a simple approximation that comes very
close to an optimal solution. Treating the system as though it
were a voltage-mode regulator, by compensating the L-C
poles and the ESR zero of the voltage mode approximation,
yields a solution that is always stable with very close to ideal
transient performance.
Select a target bandwidth for the compensated system, f0.
The target bandwidth must be large enough to assure
adequate transient performance, but smaller than 1/3 of the
per-channel switching frequency. The values of the
compensation components depend on the relationships of f0
to the L-C pole frequency and the ESR zero frequency. For
each of the following three, there is a separate set of
equations for the compensation components.
In Equation 41, L is the per-channel filter inductance divided
by the number of active channels; C is the sum total of all
output capacitors; ESR is the equivalent series resistance of
the bulk output filter capacitance; and VPP is the
peak-to-peak sawtooth signal amplitude as described in the
“Electrical Specifications” on page 6.
Once selected, the compensation values in Equation 41
assure a stable converter with reasonable transient
performance. In most cases, transient performance can be
improved by making adjustments to RC. Slowly increase the
value of RC while observing the transient performance on an
oscilloscope until no further improvement is noted. Normally,
CC will not need adjustment. Keep the value of CC from
Equation 41 unless some performance issue is noted.
FN9289.5
February 25, 2010
ISL6312
.
C2
Case 1:
1
-------------------------------- > f 0
2⋅π⋅ L⋅C
RC
2 ⋅ π ⋅ f 0 ⋅ V pp ⋅ L ⋅ C
R C = R FB ⋅ -------------------------------------------------------0.66 ⋅ V IN
0.66 ⋅ V IN
C C = --------------------------------------------------2 ⋅ π ⋅ V PP ⋅ R FB ⋅ f 0
FB
ISL6312
RFB
1
1
-------------------------------- ≤ f 0 < -----------------------------------2 ⋅ π ⋅ C ⋅ ESR
2⋅π⋅ L⋅C
V PP ⋅ ( 2 ⋅ π ) 2 ⋅ f 02 ⋅ L ⋅ C
R C = R FB ⋅ ----------------------------------------------------------------0.66 ⋅ V
(EQ. 41)
0.66 ⋅ V IN
C C = ------------------------------------------------------------------------------------2
2
( 2 ⋅ π ) ⋅ f 0 ⋅ V PP ⋅ R FB ⋅ L ⋅ C
1
f 0 > ------------------------------------2 ⋅ π ⋅ C ⋅ ESR
2 ⋅ π ⋅ f 0 ⋅ V pp ⋅ L
R C = R FB ⋅ --------------------------------------------0.66 ⋅ V IN ⋅ ESR
0.66 ⋅ V IN ⋅ ESR ⋅ C
C C = ---------------------------------------------------------------2 ⋅ π ⋅ V PP ⋅ R FB ⋅ f 0 ⋅ L
The optional capacitor C2, is sometimes needed to bypass
noise away from the PWM comparator (see Figure 20). Keep
a position available for C2, and be prepared to install a
high-frequency capacitor of between 22pF and 150pF in
case any leading edge jitter problem is noted.
COMPENSATION WITHOUT LOAD-LINE REGULATION
The non load-line regulated converter is accurately modeled
as a voltage-mode regulator with two poles at the L-C
resonant frequency and a zero at the ESR frequency. A
type III controller, as shown in Figure 20, provides the
necessary compensation.
The first step is to choose the desired bandwidth, f0, of the
compensated system. Choose a frequency high enough to
assure adequate transient performance but not higher than
1/3 of the switching frequency. The type-III compensator has
an extra high-frequency pole, fHF. This pole can be used for
added noise rejection or to assure adequate attenuation at
the error-amplifier high-order pole and zero frequencies. A
good general rule is to choose fHF = 10f0, but it can be
higher if desired. Choosing fHF to be lower than 10f0 can
cause problems with too much phase shift below the system
bandwidth.
30
IDROOP
VDIFF
IN
Case 3:
COMP
C1
R1
Case 2:
CC
FIGURE 21. COMPENSATION CIRCUIT WITHOUT LOAD-LINE
REGULATION
In the solutions to the compensation equations, there is a
single degree of freedom. For the solutions presented in
Equation 42, RFB is selected arbitrarily. The remaining
compensation components are then selected according to
Equation 42.
In Equation 42, L is the per-channel filter inductance divided
by the number of active channels; C is the sum total of all
output capacitors; ESR is the equivalent-series resistance of
the bulk output-filter capacitance; and VPP is the peak-topeak sawtooth signal amplitude as described in “Electrical
Specifications” on page 6.
C ⋅ ESR
R 1 = R FB ⋅ -------------------------------------------L ⋅ C – C ⋅ ESR
L ⋅ C – C ⋅ ESR
C 1 = -------------------------------------------R FB
0.75 ⋅ V IN
C 2 = -----------------------------------------------------------------------------------------------------------2
( 2 ⋅ π ) ⋅ f 0 ⋅ f HF ⋅ ( L ⋅ C ) ⋅ R FB ⋅ V ( P – P )
2
V PP ⋅ ⎛ 2π⎞ ⋅ f 0 ⋅ f HF ⋅ L ⋅ C ⋅ R FB
⎝ ⎠
R C = ---------------------------------------------------------------------------------------0.75 ⋅ V ⋅ ( 2 ⋅ π ⋅ f HF ⋅ L ⋅ C – 1 )
IN
0.75 ⋅ V IN ⋅ ( 2 ⋅ π ⋅ f HF ⋅ L ⋅ C – 1 )
C C = -----------------------------------------------------------------------------------------------------------( 2 ⋅ π ) 2 ⋅ f 0 ⋅ f HF ⋅ ( L ⋅ C ) ⋅ R FB ⋅ V ( P – P )
(EQ. 42)
Output Filter Design
The output inductors and the output capacitor bank together
to form a low-pass filter responsible for smoothing the
pulsating voltage at the phase nodes. The output filter also
must provide the transient energy until the regulator can
respond. Because it has a low bandwidth compared to the
switching frequency, the output filter limits the system
transient response. The output capacitors must supply or
sink load current while the current in the output inductors
increases or decreases to meet the demand.
FN9289.5
February 25, 2010
ISL6312
At the beginning of the load transient, the output capacitors
supply all of the transient current. The output voltage will
initially deviate by an amount approximated by the voltage
drop across the ESL. As the load current increases, the
voltage drop across the ESR increases linearly until the load
current reaches its final value. The capacitors selected must
have sufficiently low ESL and ESR so that the total
output-voltage deviation is less than the allowable maximum.
Neglecting the contribution of inductor current and regulator
response, the output voltage initially deviates by an amount:
di
ΔV ≈ ESL ⋅ ----- + ESR ⋅ ΔI
dt
(EQ. 43)
The filter capacitor must have sufficiently low ESL and ESR
so that ΔV < ΔVMAX.
Most capacitor solutions rely on a mixture of high frequency
capacitors with relatively low capacitance in combination
with bulk capacitors having high capacitance but limited
high-frequency performance. Minimizing the ESL of the
high-frequency capacitors allows them to support the output
voltage as the current increases. Minimizing the ESR of the
bulk capacitors allows them to supply the increased current
with less output voltage deviation.
The ESR of the bulk capacitors also creates the majority of
the output-voltage ripple. As the bulk capacitors sink and
source the inductor AC ripple current (see “Interleaving” on
page 10 and Equation 2), a voltage develops across the bulk
capacitor ESR equal to IC,PP (ESR). Thus, once the output
capacitors are selected, the maximum allowable ripple
voltage, VPP(MAX), determines the lower limit on the
inductance.
⎛V – N ⋅ V
⎞
OUT⎠ ⋅ V OUT
⎝ IN
L ≥ ESR ⋅ -------------------------------------------------------------------f S ⋅ V IN ⋅ V PP( MAX )
output-voltage deviation than the leading edge. Equation 46
addresses the leading edge. Normally, the trailing edge
dictates the selection of L because duty cycles are usually
less than 50%. Nevertheless, both inequalities should be
evaluated, and L should be selected based on the lower of
the two results. In each equation, L is the per-channel
inductance, C is the total output capacitance, and N is the
number of active channels.
2 ⋅ N ⋅ C ⋅ VO
L ≤ --------------------------------- ⋅ ΔV MAX – ( ΔI ⋅ ESR )
( ΔI ) 2
(EQ. 45)
1.25 ⋅ N ⋅ C- ⋅ ΔV
⎛
⎞
L ≤ ---------------------------MAX – ( ΔI ⋅ ESR ) ⋅ ⎝ V IN – V O⎠
( ΔI ) 2
(EQ. 46)
Switching Frequency
There are a number of variables to consider when choosing
the switching frequency, as there are considerable effects on
the upper MOSFET loss calculation. These effects are
outlined in “MOSFETs” on page 26, and they establish the
upper limit for the switching frequency. The lower limit is
established by the requirement for fast transient response
and small output-voltage ripple as outlined in
“COMPENSATION WITHOUT LOAD-LINE REGULATION”
on page 30. Choose the lowest switching frequency that
allows the regulator to meet the transient-response
requirements.
Switching frequency is determined by the selection of the
frequency-setting resistor, RT. Figure 22 and Equation 47
are provided to assist in selecting the correct value for RT.
R T = 10
[10.61 – ( 1.035 ⋅ log ( f S ) ) ]
(EQ. 47)
1000
RT (kΩ)
In high-speed converters, the output capacitor bank is usually
the most costly (and often the largest) part of the circuit.
Output filter design begins with minimizing the cost of this part
of the circuit. The critical load parameters in choosing the
output capacitors are the maximum size of the load step, ΔI,
the load-current slew rate, di/dt, and the maximum allowable
output-voltage deviation under transient loading, ΔVMAX.
Capacitors are characterized according to their capacitance,
ESR, and ESL (equivalent series inductance).
100
(EQ. 44)
10
10
100
1k
10k
SWITCHING FREQUENCY (Hz)
Since the capacitors are supplying a decreasing portion of
the load current while the regulator recovers from the
transient, the capacitor voltage becomes slightly depleted.
The output inductors must be capable of assuming the entire
load current before the output voltage decreases more than
ΔVMAX. This places an upper limit on inductance.
Equation 45 gives the upper limit on L for the cases when
the trailing edge of the current transient causes a greater
31
FIGURE 22. RT vs SWITCHING FREQUENCY
Input Capacitor Selection
The input capacitors are responsible for sourcing the AC
component of the input current flowing into the upper
MOSFETs. Their RMS current capacity must be sufficient to
handle the AC component of the current drawn by the upper
MOSFETs which is related to duty cycle and the number of
active phases.
FN9289.5
February 25, 2010
ISL6312
IL(P-P) = 0
IL(P-P) = 0.25 IO
0.3
IL(P-P) = 0.5 IO
IL(P-P) = 0.75 IO
INPUT-CAPACITOR CURRENT (IRMS/IO)
INPUT-CAPACITOR CURRENT (IRMS/IO)
0.3
0.2
0.1
0
0
0.2
0.4
0.6
0.8
1.0
DUTY CYCLE (VO/VIN)
For a four-phase design, use Figure 23 to determine the
input-capacitor RMS current requirement set by the duty
cycle, maximum sustained output current (IO), and the ratio
of the peak-to-peak inductor current (IL(P-P)) to IO. Select a
bulk capacitor with a ripple current rating which will minimize
the total number of input capacitors required to support the
RMS current calculated.
The voltage rating of the capacitors should also be at least
1.25x greater than the maximum input voltage. Figures 24 and
25 provide the same input RMS current information for threephase and two-phase designs respectively. Use the same
approach for selecting the bulk capacitor type and number.
INPUT-CAPACITOR CURRENT (IRMS/IO)
IL(P-P) = 0
IL(P-P) = 0.5 IO
IL(P-P) = 0.25 IO
IL(P-P) = 0.75 IO
0.2
IL(P-P) = 0
IL(P-P) = 0.5 IO
IL(P-P) = 0.75 IO
0
0
0.2
0.4
0.6
0.8
1.0
FIGURE 25. NORMALIZED INPUT-CAPACITOR RMS
CURRENT FOR 2-PHASE CONVERTER
Layout Considerations
MOSFETs switch very fast and efficiently. The speed with
which the current transitions from one device to another
causes voltage spikes across the interconnecting
impedances and parasitic circuit elements. These voltage
spikes can degrade efficiency, radiate noise into the circuit
and lead to device overvoltage stress. Careful component
selection, layout, and placement minimizes these voltage
spikes. Consider, as an example, the turnoff transition of the
upper PWM MOSFET. Prior to turnoff, the upper MOSFET
was carrying channel current. During the turnoff, current
stops flowing in the upper MOSFET and is picked up by the
lower MOSFET. Any inductance in the switched current path
generates a large voltage spike during the switching interval.
Careful component selection, tight layout of the critical
components, and short, wide circuit traces minimize the
magnitude of voltage spikes.
There are two sets of critical components in a DC/DC
converter using a ISL6312 controller. The power
components are the most critical because they switch large
amounts of energy. Next are small signal components that
connect to sensitive nodes or supply critical bypassing
current and signal coupling.
0.1
0
0.1
DUTY CYCLE (VIN/VO)
FIGURE 23. NORMALIZED INPUT-CAPACITOR RMS CURRENT
vs DUTY CYCLE FOR 4-PHASE CONVERTER
0.3
0.2
0
0.2
0.4
0.6
0.8
1.0
DUTY CYCLE (VIN/VO)
FIGURE 24. NORMALIZED INPUT-CAPACITOR RMS
CURRENT FOR 3-PHASE CONVERTER
Low capacitance, high-frequency ceramic capacitors are
needed in addition to the input bulk capacitors to suppress
leading and falling edge voltage spikes. The spikes result from
the high current slew rate produced by the upper MOSFET turn
on and off. Select low ESL ceramic capacitors and place one as
close as possible to each upper MOSFET drain to minimize
board parasitics and maximize suppression.
32
The power components should be placed first, which include
the MOSFETs, input and output capacitors, and the inductors. It
is important to have a symmetrical layout for each power train,
preferably with the controller located equidistant from each.
Symmetrical layout allows heat to be dissipated equally
across all power trains. Equidistant placement of the controller
to the first three power trains it controls through the integrated
drivers helps keep the gate drive traces equally short,
resulting in equal trace impedances and similar drive
capability of all sets of MOSFETs.
When placing the MOSFETs try to keep the source of the
upper FETs and the drain of the lower FETs as close as
FN9289.5
February 25, 2010
ISL6312
thermally possible. Input Bulk capacitors should be placed
close to the drain of the upper FETs and the source of the lower
FETs. Locate the output inductors and output capacitors
between the MOSFETs and the load. The high-frequency input
and output decoupling capacitors (ceramic) should be placed
as close as practicable to the decoupling target, making use of
the shortest connection paths to any internal planes, such as
vias to GND next or on the capacitor solder pad.
The critical small components include the bypass capacitors
for VCC and PVCC, and many of the components
surrounding the controller including the feedback network
and current sense components. Locate the VCC/PVCC
bypass capacitors as close to the ISL6312 as possible. It is
especially important to locate the components associated
with the feedback circuit close to their respective controller
pins, since they belong to a high-impedance circuit loop,
sensitive to EMI pick-up.
A multi-layer printed circuit board is recommended. Figure 26
shows the connections of the critical components for the
converter. Note that capacitors CxxIN and CxxOUT could each
represent numerous physical capacitors. Dedicate one solid
layer, usually the one underneath the component side of the
board, for a ground plane and make all critical component
ground connections with vias to this layer. Dedicate another
solid layer as a power plane and break this plane into smaller
islands of common voltage levels. Keep the metal runs from the
PHASE terminal to output inductors short. The power plane
should support the input power and output power nodes. Use
copper filled polygons on the top and bottom circuit layers for
the phase nodes. Use the remaining printed circuit layers for
small signal wiring.
Routing UGATE, LGATE, and PHASE Traces
Great attention should be paid to routing the UGATE, LGATE,
and PHASE traces since they drive the power train MOSFETs
using short, high current pulses. It is important to size them as
large and as short as possible to reduce their overall
impedance and inductance. They should be sized to carry at
least one ampere of current (0.02” to 0.05”). Going between
layers with vias should also be avoided, but if so, use two vias
for interconnection when possible.
Extra care should be given to the LGATE traces in particular
since keeping their impedance and inductance low helps to
significantly reduce the possibility of shoot-through. It is also
important to route each channels UGATE and PHASE traces
in as close proximity as possible to reduce their inductances.
Current Sense Component Placement and Trace
Routing
One of the most critical aspects of the ISL6312 regulator
layout is the placement of the inductor DCR current sense
components and traces. The R-C current sense components
must be placed as close to their respective ISEN+ and
ISEN- pins on the ISL6312 as possible.
The sense traces that connect the R-C sense components to
each side of the output inductors should be routed on the
bottom of the board, away from the noisy switching
components located on the top of the board. These traces
should be routed side by side, and they should be very thin
traces. It’s important to route these traces as far away from
any other noisy traces or planes as possible. These traces
should pick up as little noise as possible.
Thermal Management
For maximum thermal performance in high current, high
switching frequency applications, connecting the thermal
GND pad of the ISL6312 to the ground plane with multiple
vias is recommended. This heat spreading allows the part to
achieve its full thermal potential. It is also recommended
that the controller be placed in a direct path of airflow if
possible to help thermally manage the part.
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems.
Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and
reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result
from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com
33
FN9289.5
February 25, 2010
ISL6312
C2
RFB
LOCATE CLOSE TO IC
(MINIMIZE CONNECTION PATH)
C1
KEY
HEAVY TRACE ON CIRCUIT PLANE LAYER
ISLAND ON POWER PLANE LAYER
+12V
ISLAND ON CIRCUIT PLANE LAYER
R1
FB
IDROOP
VDIFF
VIA CONNECTION TO GROUND PLANE
COMP
CBIN1
CBOOT1
VSEN
LOCATE NEAR SWITCHING TRANSISTORS;
(MINIMIZE CONNECTION PATH)
BOOT1
RGND
UGATE1
+5V
PHASE1
VCC
(CF1)
ROFS
OFS
C1
R1
C1
ISEN1ISEN1+
FS
+12V
REF
RT
R1
LGATE1
PVCC1_2
CREF
CBIN2
(CF2)
CBOOT2
BOOT2
SS
UGATE2
RSS
PHASE2
(CHFOUT)
CBOUT
LGATE2
OVPSEL
ISL6312
LOAD
ISEN2ISEN2+
+12V
VID7
VID6
VID5
PVCC3
CBIN3
(CF2)
VID4
VID3
LOCATE NEAR LOAD;
(MINIMIZE CONNECTION
PATH)
CBOOT3
BOOT3
VID2
UGATE3
VID1
VID0
PHASE3
VRSEL
R1
PGOOD
C1
LGATE3
+12V
ISEN3ISEN3+
REN1
+12V
+12V
EN
CBIN4
REN2
EN_PH4
DRSEL
BOOT
VCC UGATE
PVCC
PHASE
ISL6612
RDR
LGATE
PWM4
GND
PWM
R1
C1
GND
ISEN4ISEN4+
FIGURE 26. PRINTED CIRCUIT BOARD POWER PLANES AND ISLANDS
34
FN9289.5
February 25, 2010
ISL6312
Package Outline Drawing
L48.7x7
48 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
Rev 4, 10/06
4X 5.5
7.00
A
44X 0.50
B
37
6
PIN 1
INDEX AREA
6
PIN #1 INDEX AREA
48
1
7.00
36
4. 30 ± 0 . 15
12
25
(4X)
0.15
13
24
0.10 M C A B
48X 0 . 40± 0 . 1
TOP VIEW
4 0.23 +0.07 / -0.05
BOTTOM VIEW
SEE DETAIL "X"
( 6 . 80 TYP )
(
0.10 C
BASE PLANE
0 . 90 ± 0 . 1
4 . 30 )
C
SEATING PLANE
0.08 C
SIDE VIEW
( 44X 0 . 5 )
C
0 . 2 REF
5
( 48X 0 . 23 )
( 48X 0 . 60 )
0 . 00 MIN.
0 . 05 MAX.
TYPICAL RECOMMENDED LAND PATTERN
DETAIL "X"
NOTES:
1. Dimensions are in millimeters.
Dimensions in ( ) for Reference Only.
2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994.
3. Unless otherwise specified, tolerance : Decimal ± 0.05
4. Dimension b applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
5. Tiebar shown (if present) is a non-functional feature.
6. The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 indentifier may be
either a mold or mark feature.
35
FN9289.5
February 25, 2010