LMX2470
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LMX2470 2.6 GHz Delta-Sigma Fractional-N PLL with 800 MHz Integer-N PLL
Check for Samples: LMX2470
FEATURES
1
•
2
•
•
•
•
•
•
Low in-band phase noise and low fractional
spurs
12 bit or 22 bit selectable fractional modulus
Up to 4th order programmable delta-sigma
modulator
Enhanced Anti-Cycle Slip Fastlock Circuitry
– Fastlock
– Cycle slip reduction
– Integrated timeout counters
Digital lock detect output
Prescalers allow wide range of N values
– RF PLL: 16/17/20/21
– IF PLL: 8/9 or 16/17
Crystal Reference Frequency up to 110 MHz
•
•
•
•
On-chip crystal reference frequency doubler.
Phase Comparison Frequency up to 30 MHz
Hardware and software power-down control
Ultra low consumption: ICC = 4.1 mA (typical)
APPLICATIONS
•
•
•
•
Cellular Phones and Base Stations
– CDMA, WCDMA, GSM/GPRS, TDMA, EDGE,
PDC
Applications requiring fine frequency
resolution
Satellite and Cable TV Tuners
WLAN Standards
DESCRIPTION
The LMX2470 is a low power, high performance delta-sigma fractional-N PLL with an auxiliary integer-N PLL.
The device is fabricated using National Semiconductor’s advanced BiCMOS process.
With delta-sigma architecture, fractional spur compensation is achieved with noise shaping capability of the deltasigma modulator and the inherent low pass filtering of the PLL loop filter. Fractional spurs at lower frequencies
are pushed to higher frequencies outside the loop bandwidth. Unlike analog compensation, the digital feedback
techniques used in the LMX2470 are highly resistant to changes in temperature and variations in wafer
processing. With delta-sigma architecture, the ability to push close in spur and phase noise energy to higher
frequencies is a direct function of the modulator order. The higher the order, the more this energy can be spread
to higher frequencies. The LMX2470 has a programmable modulator up to order four, which allows the designer
to select the optimum modulator order to fit the phase noise, spur, and lock time requirements of the system.
Programming is fast and simple. Serial data is transferred into the LMX2470 via a three line MICROWIRE
interface (Data, Clock, Load Enable). Nominal supply voltage is 2.5 V. The LMX2470 features a typical current
consumption of 4.1 mA at 2.5 V. The LMX2470 is available in a 24 lead 3.5 X 4.5 X 0.6 mm package.
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2004, Texas Instruments Incorporated
LMX2470
SNAS195A – MAY 2004 – REVISED MAY 2004
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Functional Block Diagram
Connection Diagram
Figure 1. 24-Pin CSP (SLE) Package
Pin Functions
Pin Descriptions
2
Pin #
Pin Name
I/O
1
CPoutRF
O
RF charge pump output.
Pin Description
2
GND
-
Ground
3
GND
-
RF Ground
4
GND
-
Ground for RF PLL digital circuitry.
5
FinRF
I
RF prescaler input. Small signal input from the VCO.
6
FinRF*
I
RF prescaler complimentary input. For single-ended operation, a bypass capacitor should be placed
as close as possible to this pin and be connected directly to the ground plane.
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Pin Descriptions (continued)
Pin #
Pin Name
7
VccRF
I/O
Pin Description
8
EN
I
Chip enable input. High impedance CMOS input. When EN is high, the chip is powered up, otherwise
it is powered down.
RF PLL power supply voltage input. Must be equal to VccIF . May range from 2.25V to 2.75V.
Bypass capacitors should be placed as close as possible to this pin and be connected directly to the
ground plane.
9
ENOSC
I
This pin should be grounded for normal operation.
10
CLK
I
MICROWIRE Clock. High impedance CMOS Clock input. Data for the various counters is clocked
into the 24 bit shift register on the rising edge.
11
DATA
I
MICROWIRE Data. High impedance binary serial data input.
12
LE
13
Ftest/LD
O
Test frequency output / Lock Detect
14
VddIF
-
Digital power supply for IF PLL
15
VccIF
-
IF power supply voltage input. Must be equal to VccRF. Input may range from 2.25 V to 2.75 V.
Bypass capacitors should be placed as close as possible to this pin and be connected directly to the
ground plane.
16
GND
-
Ground for RF PLL digital circuitry.
17
FinIF
I
IF prescaler input. Small signal input from the VCO.
MICROWIRE Load Enable. High impedance CMOS input. Data stored in the shift registers is loaded
into the internal latches when LE goes HIGH
18
GND
-
Digital ground for IF PLL
19
CPoutIF
O
IF PLL charge pump output
20
FLoutIF
O
IF Fastlock Output. Also functions as Programmable TRI-STATE CMOS output.
21
OSCout*
I/O
Complementary reference input or oscillator output.
22
OSCin
I
Reference input
23
VddRF
-
Digital power supply for RF PLL
24
FLoutRF
O
RF Fastlock Output. Also functions as Programmable TRI-STATE CMOS output.
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
Absolute Maximum Ratings
(1) (2)
Parameter
Power Supply Voltage
Symbol
Value
Min
Typ
Max
Units
VCC
-0.3
3.0
V
VDD
VCC
VCC
V
Vi
-0.3
VCC + 0.3
V
Storage Temperature Range
Ts
-65
+150
°C
Lead Temperature (Solder 4 sec.)
TL
+260
°C
Voltage on any pin with GND =VSS = 0V
(1)
(2)
“Absolute Maximum Ratings” indicate limits beyond which damage to the device may occur. "Recommended Operating Conditions"
indicate conditions for which the device is intended to be functional, but do not guarantee specific performance limits. For guaranteed
specifications and test conditions, see the Electrical Characteristics. The guaranteed specifications apply only for the test conditions
listed. Note also that these maximum ratings imply that the voltage at all the power supply pins of VccRF, VccIF, VddRF, and VddIF are
the same. VCCwill be used to refer to the voltage at these pins.
This Device is a high performance RF integrated circuit with an ESD rating < 2 kV and is ESD sensitive. Handling and assembly of this
device should only be done at ESD-free workstations.
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Recommended Operating Conditions
Parameter
Power Supply Voltage
Symbol
(1)
Operating Temperature
(1)
4
Value
Min
Typ
Max
Units
VCC
2.25
2.75
V
VDD
VCC
VCC
V
TA
-40
+85
°C
“Absolute Maximum Ratings” indicate limits beyond which damage to the device may occur. "Recommended Operating Conditions"
indicate conditions for which the device is intended to be functional, but do not guarantee specific performance limits. For guaranteed
specifications and test conditions, see the Electrical Characteristics. The guaranteed specifications apply only for the test conditions
listed. Note also that these maximum ratings imply that the voltage at all the power supply pins of VccRF, VccIF, VddRF, and VddIF are
the same. VCCwill be used to refer to the voltage at these pins.
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Electrical Characteristics
(VCC = 2.5V; -40°C ≤ TA ≤ +85°C unless otherwise specified)
Symbol
Parameter
Conditions
Value
Min
Typ
Max
Units
Icc PARAMETERS
ICCRF
ICCIF
ICCTOTAL
ICCPD
Power Supply Current, RF
Synthesizer
IF PLL OFF
RF PLL ON
Charge Pump TRI-STATE
OSC=0
2.7
3.9
mA
Power Supply Current, IF
Synthesizer
IF PLL ON
RF PLL OFF
Charge Pump TRI-STATE
OSC=0
1.4
2.3
mA
Power Supply Current, Entire
Synthesizer
IF PLL ON
RF PLL ON
Charge Pump TRI-STATE
OSC=0
4.1
6.0
mA
Power Down Current
EN = ENOSC = 0V
CLK, DATA, LE = 0V
1
10
µA
RF SYNTHESIZER PARAMETERS
fFinRF
Operating Frequency
500
2600
MHz
pFinRF
Input Sensitivity
-15
0
dBm
fCOMP
Phase Detector Frequency
30
MHz
ICPoutRFSRCE
RF Charge Pump Source
Current
RF_CPG = 0
VCPoutRF=VCC /2
100
µA
RF_CPG = 1
VCPoutRF=VCC/2
200
µA
...
ICPoutRFSINK
RF Charge Pump Sink Current
...
µA
RF_CPG = 15
VCPoutRF=VCC/2
1600
µA
RF_CPG = 0
VCPoutRF=VCC/2
-100
µA
RF_CPG = 1
VCPoutRF=VCC/2
-200
µA
...
µA
-1600
µA
...
RF_CPG = 15
VCPoutRF=VCC/2
ICPoutRFTRI
RF Charge Pump TRI-STATE
Current Magnitude
0.4 ≤ VCPoutRF ≤ VCC -0.4
2
10
nA
ICPoutRF%MIS
RF CP Sink vs. CP Source
Mismatch
VCPoutRF = VCC/2
TA = 25°C
3
10
%
ICPoutRF%V
RF CP Current vs. CP Voltage
0.4 ≤ VCPoutRF ≤ VCC -0.4
TA = 25°C
5
15
%
ICPoutRF%TEMP
RF CP Current vs. Temperature
VCPoutRF = VCC/2
8
%
IF SYNTHESIZER PARAMETERS
fFinIF
Operating Frequency
75
800
MHz
pFinIF
IF Input Sensitivity
-15
0
dBm
fCOMP
Phase Detector Frequency
10
MHz
ICPoutIFSRCE
IF Charge Pump Source Current
ICPoutIFSINK
IF Charge Pump Sink Current
IF_CPG = 0
VCPoutIF = VCC /2
1
mA
IF_CPG = 1
VCPoutIF = VCC/2
4
mA
IF_CPG = 0
VCPoutIF = VCC/2
-1
mA
IF_CPG = 1
VCPoutIF = VCC/2
-4
mA
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Electrical Characteristics (continued)
(VCC = 2.5V; -40°C ≤ TA ≤ +85°C unless otherwise specified)
Symbol
Parameter
Conditions
Value
Min
Typ
Max
10
ICPoutIFTRI
IF Charge Pump TRI-STATE
Current Magnitude
0.4 ≤ VCPoutIF ≤ VCC RF -0.4
2
ICPoutIF%MIS
IF CP Sink vs. CP Source
Mismatch
VCPoutIF = VCC/2
TA = 25°C
3
ICPoutIF%V
IF CP Current vs. CP Voltage
0.4 ≤ VCPoutIF ≤ VCC -0.4
TA = 25°C
8
ICPoutIF%TEMP
IF CP Current vs. Temperature
VCPoutIF = VCC/2
8
Units
nA
%
15
%
%
OSCILLATOR PARAMETERS
fOSCin
Oscillator Operating Frequency
vOSCin
Oscillator Input Sensitivity
IOSCin
Oscillator Input Current
OSC2X = 0
5
110
MHz
OSC2X = 1
5
20
MHz
0.5
VCC
V
-100
100
µA
1.6
VCC
V
DIGITAL INTERFACE (DATA, CLK, LE, EN, ENRF, Ftest/LD, FLoutRF, FLoutIF)
VIH
High-Level Input Voltage
VIL
Low-Level Input Voltage
IIH
High-Level Input Current
VIH = VCC
IIL
Low-Level Input Current
VIL = 0 V
VOH
High-Level Output Voltage
IOH = -500 µA
VOL
Low-Level Output Voltage
IOL = 500 µA
0.4
V
-1.0
1.0
µA
-1.0
1.0
µA
VCC-0.4
V
0.4
V
MICROWIRE INTERFACE TIMING
TCS
Data to Clock Set Up Time
See Microwire Input Timing
50
ns
TCH
Data to Clock Hold Time
See Microwire Input Timing
10
ns
TCWH
Clock Pulse Width High
See Microwire Input Timing
50
ns
TCWL
Clock Pulse Width Low
See Microwire Input Timing
50
ns
TES
Clock to Load Enable Set Up
Time
See Microwire Input Timing
50
ns
TEW
Load Enable Pulse Width
See Microwire Input Timing
50
ns
RF Synthesizer Normalized
Phase Noise Contribution (1)
RF_CPG = 0
-200
dBc/Hz
RF_CPG = 3
-206
dBc/Hz
RF_CPG = 7
-208
dBc/Hz
RF_CPG = 15
-210
dBc/Hz
Applies to both low and high current
modes
-209
dBc/Hz
PHASE NOISE
LF1HzRF
LF1HzIF
(1)
6
IF Synthesizer Normalized
Phase Noise Contribution (1)
Normalized Phase Noise Contribution is defined as: LN(f) = L(f) – 20log(N) – 10log(fCOMP) where L(f) is defined as the single side band
phase noise measured at an offset frequency, f, in a 1 Hz Bandwidth. The offset frequency, f, must be chosen sufficiently smaller than
the PLL loop bandwidth, yet large enough to avoid substantial phase noise contribution from the reference source. The offset chosen
was 4 KHz.
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MICROWIRE INPUT TIMING DIAGRAM
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: Sensitivity
8
RF N Counter Sensitivity
TA = 25°C
RF N Counter Sensitivity
VCC = 2.5 V
IF N Counter Sensitivity
TA = 25°C
IF N Counter Sensitivity
VCC = 2.5 V
OSCin Counter Sensitivity
OSC=0
TA = 25°C
OSCin Counter Sensitivity
OSC=0
VCC= 2.5 V
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: FinRF Input Impedance
FinRF Input Impedance
Frequency (MHz)
Real (Ohms)
Imaginary (Ohms)
500
214
-255
600
175
-245
700
144
-230
800
118
-216
900
98
-203
1000
80
-189
1100
69
-177
1200
57
-165
1300
48
-153
1400
39
-141
1500
34
-130
1600
28
-119
1700
24
-110
1800
20
-101
1900
17
-94
2000
14
-87
2100
13
-82
2200
11
-77
2300
10
-72
2400
8
-67
2500
7
-62
2600
7
-56
2700
7
-53
2800
7
-46
2900
7
-41
3000
7
-39
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: FinIF Input Impedance
FinIF Input Impedance
10
Freqeuncy (MHz)
Real (Ohms)
Imaginary (Ohms)
50
580
-313
100
500
-273
150
445
-250
200
410
-256
250
378
-259
300
349
-264
350
322
-267
400
297
-270
450
274
-271
500
253
-272
550
232
-272
600
214
-269
650
198
-265
700
184
-262
750
170
-259
800
157
-256
900
151
-253
1000
143
-250
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: OSCin Input Impedance
OSCin Input Impedance
Frequency (MHz)
Real (Ohms)
Imaginary (Ohms)
Magnitude (Ohms)
50
2200
-4700
5189
10
710
-2700
2792
20
229
-1500
1517
30
133
-988
997
40
93
-752
758
50
74
-606
611
60
62
-505
509
70
53
-435
438
80
49
-382
385
90
45
-341
344
100
42
-309
312
110
40
-282
285
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: Currents
Total Current Consumption
OSC=0
Powerdown Current
EN = LOW
RF Charge Pump Current
VCC = 2.5 Volts
IF Charge Pump Current
VCC = 2.5 Volts
Charge Pump Leakage
RF PLL
Charge Pump Leakage
IF PLL
Typical performance characteristics do not imply any sort of
guarantee. Guaranteed specifications are in the electrical
characteristics section.
12
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Bench Test Setups
Charge Pump Current Measurement Procedure
The above block diagram shows the test procedure for testing the RF and IF charge pumps. These tests include
absolute current level, mismatch, and leakage. In order to measure the charge pump currents, a signal is applied
to the high frequency input pins. The reason for this is to guarantee that the phase detector gets enough
transitions in order to be able to change states. If no signal is applied, it is possible that the charge pump current
reading will be low due to the fact that the duty cycle is not 100%. The OSCin Pin is tied to the supply. The
charge pump currents can be measured by simply programming the phase detector to the necessary polarity.
For instance, in order to measure the RF charge pump current, a 10 MHz signal is applied to the FinRF pin. The
source current can be measured by setting the RF PLL phase detector to a positive polarity, and the sink current
can be measured by setting the phase detector to a negative polarity. The IF PLL currents can be measured in a
similar way. Note that the magnitude of the RF and IF PLL charge pump currents are also controlled by the
RF_CPG and IF_CPG bits. Once the charge pump currents are known, the mismatch can be calculated as well.
In order to measure leakage currents, the charge pump current is set to a TRI-STATE mode by enabling the
counter reset bits. This is RF_RST for the RF PLL and IF_RST for the IF PLL. The table below shows a
summary of the various charge pump tests.
Current Test
RF_CPG
RF_CPP
RF_CPT
IF_CPG
IF_CPP
IF_CPT
RF Source
0 to 15
0
0
X
X
X
RF Sink
0 to 15
1
0
X
X
X
RF TRI-STATE
X
X
1
X
X
X
IF Source
X
X
X
0 to 1
0
0
IF Sink
X
X
X
0 to 1
1
0
IF TRI-STATE
X
X
X
X
X
1
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Frequency Input Pin
DC Blocking
Capacitor
Corresponding
Counter
Default Counter
Value
MUX Value
OSCin
1000 pF
FinRF
100 pF
FinIF
100 pF
OSC
RF_R / 2
50
14
0
RF_N / 2
500
15
X
IF_N / 2
500
13
X
Sensitivity Measurement Procedure
Sensitivity is defined as the power level limits beyond which the output of the counter being tested is off by 1 Hz
or more of its expected value. It is typically measured over frequency, voltage, and temperature. In order to test
sensitivity, the MUX[3:0] word is programmed to the appropriate value. The counter value is then programmed to
a fixed value and a frequency counter is set to monitor the frequency of this pin. The expected frequency at the
Ftest/LD pin should be the signal generator frequency divided by twice the corresponding counter value. The
factor of two comes in because the LMX2470 has a flip-flop which divides this frequency by two to make the duty
cycle 50% in order to make it easier to read with the frequency counter. The frequency counter input impedance
should be set to high impedance. In order to perform the measurement, the temperature, frequency, and voltage
is set to a fixed value and the power level of the signal is varied. Note that the power level at the part is assumed
to be 4 dB less than the signal generator power level. This accounts for 1 dB for cable losses and 3 dB for the
pad. The power level range where the frequency is correct at the Ftest/LD pin to within 1 Hz accuracy is
recorded for the sensitivity limits. The temperature, frequency, and voltage can be varied in order to produce a
family of sensitivity curves. Since this is an open-loop test, the charge pump is set to TRI-STATE and the unused
side of the PLL (RF or IF) is powered down when not being tested. For this part, there are actually four frequency
input pins, although there is only one frequency test pin (Ftest/LD). The conditions specific to each pin are show
above.
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Input Impedance Measurement Procedure
The above block diagram shows the test procedure measuring the input impedance for the LMX2470. This
applies to the FinRF, FinIF, and OSCin pins. The basic test procedure is to calibrate the network analyzer,
ensure that the part is powered up, and then measure the input impedance. The network analyzer can be
calibrated by using either calibration standards or by soldering resistors directly to the evaluation board. An open
can be implemented by putting no resistor, a short can be implemented by using a 0 ohm resistor, and a short
can be implemented by using two 100 ohm resistors in parallel. Note that no DC blocking capacitor is used for
this test procedure. This is done with the PLL removed from the PCB. This requires the use of a clamp down
fixture that may not always be generally available. If no clamp down fixture is available, then this procedure can
be done by calibrating up to the point where the DC blocking capacitor usually is, and then adding a 0 ohm
resistor back for the actual measurement. Once that the network analyzer is calibrated, it is necessary to ensure
that the PLL is powered up. This can be done by toggling the power down bits (RF_PD and IF_PD) and
observing that the current consumption indeed increases when the bit is disabled. Sometimes it may be
necessary to apply a signal to the OSCin pin in order to program the part. If this is necessary, disconnect the
signal once it is established that the part is powered up. It is useful to know the input impedance of the PLL for
the purposes of debugging RF problems and designing matching networks. Another use of knowing this
parameter is make the trace width on the PCB such that the input impedance of this trace matches the real part
of the input impedance of the PLL frequency of operation. In general, it is good practice to keep trace lengths
short and make designs that are relatively resistant to variations in the input impedance of the PLL.
Functional Description
GENERAL
The basic phase-lock-loop (PLL) configuration consists of a high-stability crystal reference oscillator, a frequency
synthesizer such as the National Semiconductor LMX2470, a voltage controlled oscillator (VCO), and a passive
loop filter. The frequency synthesizer includes a phase detector, current mode charge pump, as well as
programmable reference [R] and feedback [N] frequency dividers. The VCO frequency is established by dividing
the crystal reference signal down via the R counter to obtain a frequency that sets the comparison frequency.
This comparison frequency, fCOMP, is input of a phase/frequency detector and compared with another signal, fN,
the feedback signal, which was obtained by dividing the VCO frequency down by way of the N counter and
fractional circuitry. The phase/frequency detector's current source outputs a charge into the loop filter, which is
then converted into the VCO's control voltage. The function of the phase/frequency comparator is to adjust the
voltage presented to the VCO until the frequency and phase of the feedback signal match that of the reference
signal. When this ‘phase-locked’ condition exists, the VCO frequency will be N+F times that of the comparison
frequency, where N is the integer component of the divide ratio and F is the fractional component. Fractional
synthesis allows the phase detector frequency to be increased while maintaining the same frequency step size
for channel selection. The division value N is thereby reduced giving a lower phase noise referred to the phase
detector input, and the comparison frequency is increased allowing faster switching times.
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PHASE DETECTOR OPERATING FREQUENCY
The maximum phase detector operating frequency for the LMX2470 is 30 MHz. However, this is not possible in
all circumstances due to illegal divide ratios of the N counter. The crystal reference frequency also limits the
phase detector frequency. There are trade-offs in choosing what phase detector frequency to operate at. If this
frequency is run higher, then phase noise will be lower, but lock time may be increased due to cycle slipping.
After this phase detector frequency gets sufficiently high, then there are diminishing returns for phase noise, and
raising the charge pump current has a greater impact on phase noise. This phase detector frequency also has an
impact on fractional spurs. In general, the spur performance is better at higher phase detector frequencies,
although this is application specific. The current consumption may also slightly increase with higher phase
detector frequencies.
OSCILLATOR
The LMX2470 provides maximum flexibility for choosing an oscillator reference. One possible method is to use a
single-ended TCXO to drive the OSCin pin. The part can also be configured to be driven differentially using the
OSCin and OSCout* pins. Note that the OSCin and OSCout* pins can not be used as an inverter for a crystal.
Selection between these two modes does have a noticeable impact on phase noise and sub-fractional spurs.
Regardless of which mode is used, the performance is generally best for higher oscillator power levels.
POWER DOWN AND POWER UP MODES
The power down state of the LMX2470 is controlled by many factors. The one factor that overrides all other
factors is the EN pin. If this pin is low, this guarantees the part will be powered down. Asserting a high logic level
on EN is necessary to power up the chip, however, there are other bits in the programming registers that can
override this and put the PLL back in a power down state. Provided that the voltage on the EN pin is high,
programming the RF_PD and IF_PD bits to zero guarantees that the part will be powered up. Programming
either one of these bits to one will power down the appropriate section of the synthesizer, provided that the
ATPU[1:0] ( Auto Power Up ) bits do not override this.
There are many different ways to power down this chip and many different things that can be powered down.
This section discusses how to power down the PLLs on the chip. There are two terms that need to be defined
first: synchronous power down and asynchronous power down. In the case of synchronous power down, the PLL
chip powers down after the charge pump turns off. This is best to prevent unwanted frequency glitches upon
power up. However, in certain cases where the charge pump is stuck on, such as the case when there is no
VCO signal applied, this type of power down will not reliably work and asynchronous power down is necessary.
In the case of asynchronous power down, the PLL powers down regardless of the status of the charge pump.
There are 4 factors that affect the power down state of the chip: the EN pin, the power down bit, the TRI-STATE
bit, and writing to the RF N counter with the RF_ATPU[1:0] bits enabled
16
EN Pin
ATPU[1:0] Bits Enabled
+
RF N Counter Written To
RF_PD Bit
RF_CPT Bit
PLL State
Low
X
X
X
Asynchronous Power
Down
High
Yes
X
X
PLL is active with charge
pump in the active state.
High
No
0
0
PLL is active with charge
pump in the active state.
High
No
0
1
PLL is active, but charge
pump is TRI-STATE.
High
No
1
0
Synchronous Power Down
High
No
1
1
Asynchronous Power
Down
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DIGITAL LOCK DETECT OPERATION
The RF PLL digital lock detect circuitry compares the difference between the phase of the inputs of the phase
detector to a RC generated delay of 10 nS. To enter the locked state (Lock = HIGH) the phase error must be
less than the 10nS RC delay for 5 consecutive reference cycles. Once in lock (Lock = HIGH), the RC delay is
changed to approximately 20nS. To exit the locked state (Lock = LOW), the phase error must become greater
than the 20nS RC delay. When the PLL is in the power down mode, Lock is forced LOW. For the RF PLL, the
digital lock detect circuitry does not function reliably for comparison frequencies above 20 MHz.
The IF PLL digital lock detect circuitry works in a very similar way as the RF PLL digitial lock circuitry, except that
it uses a delay of less than 15 nS for 5 reference cycles to determine a locked condition and a delay of greater
than 30 nS to determine the IF PLL is unlocked. Note that if the MUX[3:0] word is set such as to view lock detect
for both PLLs, an unlocked (LOW) condition is shown whenever either one of the PLLs is determined to be out of
lock. A flow chart of the IF digital lock detect circuitry is shown below.
PCB LAYOUT CONSIDERATIONS
Power Supply Pins For these pins, it is recommended that these be filtered by taking a series 18 ohm resistor
and then placing two capacitors shunt to ground, thus creating a low pass filter. Although it makes sense to use
large capacitor values in theory, the ESR ( Equivalent Series Resistance ) is greater for larger capacitors. For
optimal filtering minimize the sum of the ESR and theoretical impedance of the capacitor. It is therefore
recommended to provide two capacitors of very different sizes for the best filtering. 0.1 µF and 100 pF are typical
values. The charge pump supply pins in particular are vuvulnerablenerable to power supply noise.
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High Frequency Input Pins, FinRF and FinIF The signal path from the VCO to the PLL is the most sensitive
and challenging for board layout. It is generally recommended that the VCO output go through a resistive pad
and then through a DC blocking capacitor before it gets to these high frequency input pins. If the trace length is
sufficiently short ( < 1/10th of a wavelength ), then the pad may not be necessary, but a series resistor of about
39 ohms is still recommended to isolate the PLL from the VCO. The DC blocking capacitor should be chosen at
least to be 100 pF. It may turn out that the frequency in this trace is above the self-resonant frequency of the
capacitor, but since the input impedance of the PLL tends to be capacitive, it actually be a benefit to exceed the
self-resonant frequency. The pad and the DC blocking capacitor should be placed as close to the PLL as
possible
Complimentary High Frequency Pin, FinRF* These inputs may be used to drive the PLL differentially, but it is
very common to drive the PLL in a single ended fashion. A shunt capacitor should be placed at the FinRF* pin.
The value of this capacitor should be chosen such that the impedance, including the ESR of the capacitor, is as
close to an AC short as possible at the operating frequency of the PLL. 100 pF is a typical value.
FASTLOCK AND CYCLE SLIP REDUCTION
The LMX2470 has enhanced features for Fastlock and cycle slip operation. The next several sections discuss
the the benefits of using both of these features. There are four possible combinations that are possible, and
these are shown in the table below:
Keep Comparison
Frequency the Same
Charge Pump Current
Increase Charge Pump Current
Keep Charge Pump Current the Same
Decrease Charge Pump Current
Decrease Comparison
Frequency (CSR)
(RF Side Only)
Classical Fastlock
Allows the loop bandwidth to be increased.
This has a frequency glitch caused by
switching the charge pump currents, but
there is no frequency glitch caused by
switching from fractional to integer mode
CSR/Fastlock Combination
Engaging the CSR does decrease the loop
bandwidth during frequency acquisition, but
may be necessary to reduce cycle slipping.
By also increasing the charge pump current,
this can compensate for the reduce loop
bandwidth due to the CSR
Operation with No Fastlock
This mode represents using no Fastlock
CSR Only
This mode is not generally recommended,
but may reduce cycle slipping in some
applications. Although the theoretical lock
time is decreased, due to the decreased loop
bandwidth during Fastlock, cycle slips can be
reduced or eliminated.
It never makes sense to use a lower charge
pump current during Fastlock than in the
steady state.
Note that if the charge pump current and cycle slip reduction circuitry are engaged in the same proportion, then it
is not necessary to switch in a Fastlock resistor and the loop filter will be optimized for both normal mode and
Fastlock mode. For third and fourth order filters which have problems with cycle slipping, this may prove to be
the optimal choice of settings.
Determining the Loop Gain Multiplier, K
The loop bandwidth multiplier, K, is needed in order to determine the theoretical impact of fastlock/CSR on the
loop bandwidth and also which resistor should be switched in parallel with the loop filter resistor R2. K = K_K ·
K_Fcomp where K is the loop gain multiplier K_K is the ratio of the Fastlock charge pump current to the steady
state charge pump current. Note that this should always be greater than or equal to one. K_Fcomp is the ratio of
the Fastlock comparison frequency to the steady state comparison frequency. If this ratio is less than one, this
implies that the CSR is being used.
Determining the Theoretical Lock Time Improvement and Fastlock Resistor, R2’
When using fastlock, it is necessary to switch in a resistor R2’, in parallel with R2 in order to keep the loop filter
optimized and maintain the same phase margin. After the PLL has achieved a frequency that is sufficiently close
to the desired frequency, the resistor R2’ is disengaged and the charge pump current is and comparison
frequency are returned to normal. Of special concern is the glitch that is caused when the resistor R2’ is
disengaged. This glitch can take up a significant portion of the lock time. The LMX2470 has enhanced switching
circuitry to minimize this glitch and therefore improve the lock time.
18
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The change in loop bandwidth is dependent upon the loop gain multiplier, K, as determined in section 4. The
theoretical improvement in lock time is given below, but the actual improvement will be less than this due to the
glitch that is caused by disengaging Fastlock. The theoretical improvement is given to show an upper bound on
what improvement is possible with Fastlock. In the case that K < 1, this implies the CSR is being engaged and
that the theoretical lock time will be degraded. However, since this mode reduces or eliminates cycle slipping, the
actual lock time may be better in cases where the loop bandwidth is small relative to the comparison frequency.
Realize that the theoretical lock time multiplier does not account for the fastlock/CSR disengagement glitch,
which is most severe for larger values of K.
Loop Gain
Multiplier, K
Loop Bandwidth
Multiplier
R2’ Value
Lock Time
Multiplier
1:8*
0.35
open
× 2.828
1:4*
0.50
open
× 2.000
1:2*
0.71
open
× 1.414
4:1
2.00
R2/1.00
× 0.500
8:1
2.83
R2/1.83
× 0.354
16:1
4.00
R2/3.00
× 0.250
32:1
5.66
R2/4.65
× 0.177
* These modes of operation are generally not recommended
Using Fastlock and Cycle Slip Reduction (CSR) to Avoid Cycle Slipping
In the case that the comparison frequency is very large ( i.e. 100 X ) of the loop bandwidth, cycle slipping may
occur when an instantaneous phase error is presented to the phase detector. This can be reduced by increasing
the loop bandwidth during frequency acquisition, decreasing the comparison frequency during frequency
acquisition, or some combination of the these. If increasing the loop bandwidth during frequency acquisition is
not sufficient to reduce cycle slipping, the LMX2470 also has a routine to decrease the comparison frequency.
RF PLL Fastlock Reference Table and Example
The table below shows most of the trade-offs involved in choosing a steady-state charge pump current
(RF_CPG), the Fastlock charge pump current (RF_CPF), and the Cycle Slip Reduction Factor (CSR).
Parameter
Advantages to Choosing Smaller
RF_CPG
1. Allows capacitors in loop filter to be smaller values
making it easier to find physically smaller components and
components with better dielectric properties.
Advantages to Choosing Larger
Phase noise, especially within the loop bandwidth of the
system
will be slightly worse for lower charge pump currents.
2. Allows a larger loop bandwidth multiplier for fastlock, or
a higher cycle slip reduction factor.
RF_CPF
CSR
The only reason not to always choose this to 1600 µA is to This allows the maximum possible benefit for fastlock.
make it such that no resistor is required for fastlock. For
3rd and 4th order filters, it is not possible to keep the filter
perfectly optimized by simply switching in a resistor for
fastlock.
Do not choose this any larger than necessary to eliminate
cycle slipping. Keeping this small allows a larger loop
bandwidth multiplier for fastlock.
This will eliminate cycle slips better.
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The above table shows various combinations for using RF_CPG, RF_CPF, and CSR. Although this table does
not show all possible combinations, it does show all the modes that give the best possible performance. To use
this table, choose a CSR factor on the horizontal axis, then a fastlock loop bandwidth multiplier on the vertical
axis, and the table will show all possible combinations of steady state current, Fastlock current, and what resistor
value (R2’) to use during Fastlock. In order to better illustrate the cycle slipping and Fastlock circuitry, consider
the following example:
Crystal Reference
10 MHz
Comparison Frequency
10 MHz × 2 = 20 MHz (OSC2X = 1)
Output Frequency
1930 – 1990 MHz
PLL Loop Bandwidth
10 KHz
Loop Filter Order
4th ( i.e. 7 components )
The comparison frequency is 20 MHz and the loop bandwidth is 10 KHz. 20 MHz is a good comparison
freqeuncy to use because it yields the best phase noise performance. This ratio of the comparison frequency to
the loop bandwidth is 2000, so cycle slipping will occur and degrade the lock time, unless something is done to
prevent it. Because the filter is fourth order, it would be difficult to keep the loop filter optimized if the loop gain
multiplier, K was not one. For this reason, choosing a loop gain multiplier of one makes sense. One solution is to
set the steady state current to be 100 µA, and the fastlock current to be 1600 µA. The CSR factor could be set to
1/16 and reduce this ratio to 2000/16 = 125. However, using 100 µA charge pump current has phase noise that
is significantly worse than the higher charge pump current modes. A better solution would be to use 200 µA
current and 1600 µA X2 ( using PDCP = X2 Fastlock ), since the 200 µA mode will have better phase noise.
Depending on how important phase noise is, it could make sense to use a higher steady state current. Using 800
µA steady state current provides much better phase noise than 200 uA ( about 5 dB ), but then the cycle slip
reduction factor would need to be reduced to 4. In general, it is good practice to use the PDCP = X2 fastlock
mode whenever cycle slip reduction is used, so that the best phase noise can be achieved. If the ¼ CSR factor
is used, then the ratio of comparison frequency to loop bandwidth in fastlock is reduced to 250. There may be
some cycle slipping, but the phase noise benefit of using the higher charge pump current may be worth it. If
phase noise is even more important, it might even make sense to have a steady state current of 1600 µA and
use a CSR factor of ½ and the PDCP mode of X2 Fastlock. Another consideration is that the comparison
frequency could be lowered in the steady state mode to reduce cycle slipping. This sacrifices phase noise for
lock time. In general, using Fastlock and CSR is not the same for every application. There is a trade-off of lock
time vs. phase noise. It might be tempting to try to achieve the best Fastlock benefit by using a K value of 32.
Even if the loop filter could be kept well optimized in Fastlock, this hypothetical design would probably switch
very fast when the Fastlock was engaged, but then when Fastlock is disengaged, a large frequency glitch would
appear, and the majority of the lock time would consist of waiting for this glitch to settle out. Although this would
definitely improve the lock time, even accounting for the glitch, the same result could probably be obtained by
using a lower K value, like 8, and having better phase noise instead.
Capacitor Dielectric Considerations for Lock Time
The LMX2470 has a high fractional modulus and high charge pump gain for the lowest possible phase noise.
One consideration is that the reduced N value and higher charge pump may cause the capacitors in the loop
filter to become larger in value. For larger capacitor values, it is common to have a trade-off between capacitor
dielectric quality and physical size. Using film capacitors or NP0/CG0 capacitors yields the best possible lock
times, where as using X7R or Z5R capacitors can increase lock time by 0 – 500%. However, it is a general
tendency that designs that use a higher compare frequency tend to be less sensitive to the effects of capacitor
dielectrics. Although the use of lesser quality dielectric capacitors may be unavoidable in many circumstances,
allowing a larger footprint for the loop filter capacitors, using a lower charge pump current, and reducing the
fractional modulus are all ways to reduce capacitor values. Capacitor dielectrics have very little impact on phase
noise and spurs.
FRACTIONAL SPUR AND PHASE NOISE CONTROLS FOR THE LMX2470
The LMX2470 has several bits that have a large impact on fractional spurs. These bits also have a lesser effect
on phase noise. The control words in question are CPUD[2:0], FM[1:0], and DITH[1:0]. It is difficult to predict
which settings will be optimal for a particular application without testing them, but the general recipe for using
these bits can be seen.
20
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A good algorithm is to start with a 3rd order fractional modulator (FM=3) and dithering disabled. Then depending
on whether phase noise, fractional spurs, or sub-fractional spurs are most important, optimize the settings.
Integer spurs and fractional spurs are nothing new, but sub-fractional spurs are something unique to delta-sigma
PLLs. These are spurs that occur at a fraction of the frequency of where a fractional spur would appear.
First adjust the delta-sigma modulator order. Often increasing from a 2nd to a 3rd order modulator provides a
large benefit in spur levels. Increasing from a 3rd to a 4th order modulator usually provides some benefit, but it is
usually on the order of a few dB. The modulator order by far has the greatest impact on the main fractional
spurs. If the loop bandwidth is very wide, or the loop filter order is not high enough, higher order modulators will
introduce a lot of sub-fractional spurs. The second order modulator usually does not have these sub-fractional
spurs. The third order modulator will introduce them at ½ of the frequency where one would expect to see a
traditional fractional spur, thus the name "sub-fractional spur". The fourth order modulator will introduce these
spurs at ½ and ¼ of where a traditional fractional spur would be. If the benefit of using a higher order modulator
seems significant enough, it may make sense to try to compensate for them using the other two test bits, or
designing a higher order loop filter. Be aware that the impact of the modulator order on the spurs may not be
consistent across tuning voltage. When the charge pump mismatch is not so bad, the lower order modulators
may seem to outperform the higher order modulators, but when the worst case fractional spurs are considered
over the whole range, often the higher order modulator performs better.
Second, adjust with the CPUD[2:0] bits. Setting this bit to maximum tends to reduce the sub-fractional spurs the
most, however, it may degrade phase noise by up to 1 dB.
Third, experiment with the dithering. When dithering is enabled, it may increase phase noise by up to 2 dB.
However, enabling dithering may also reduce the sub-fractional spurs. Also, sometimes both the fractional spurs
and the sub-fractional spurs can be unpredictable with dithering disabled. This is because the delta-sigma
sequence is periodic, but the starting point changes. Dithering takes these problems away. When the fractional
numerator is 0, enabling dithering typically hurts spur performance, because it is trying to correct for spur that are
not there.
Fourth, consider experimenting with the loop filter order and comparison frequency. In general, higher order loop
filters are always better, but they require more components. Often, the best spur performance is at higher
comparison frequencies as well. The reason why this is the last step is not because it has the least impact, but
because it takes more labor to do this than to change the FM[1:0], CPUD[2:0], and DITH[1:0] bits.
Although general trends do exist, the optimal settings for test bits may depend on the comparison frequency and
loop filter. Also the output frequency in important. In particular, the charge pump tuning voltage is relevant. The
recommended way to do this is to test the spur levels at the low, middle, and high range of the VCO, and use the
worst case over these three frequencies as a metric for performance. Also, it is important to be aware that all the
rules stated above have counterexamples and exceptions. However, more often than not, these rules apply.
Programming Description
GENERAL PROGRAMMING INFORMATION
The descriptions below describe the 24-bit data registers loaded through the MICROWIRE Interface. These data
registers are used to program the R counter, the N counter, and the internal mode control latches. The data
format of a typical 24-bit data register is shown below. The control bits CTL [3:0] decode the register address. On
the rising edge of LE, data stored in the shift register is loaded into one of the appropriate latches (selected by
address bits). Data is shifted in MSB first. Note that it is best to program the N counter last, since doing so
initializes the digital lock detector and Fastlock circuitry. Note that initialize means it resets the counters, but it
does NOT program values into these registers. Upon a cold power-up, it is necessary to program all the
registers. The exception is when 22-bit is not being used. In this case, it is not necessary to program the R7
register.
MSB
LSB
DATA [21:0]
23
CTL [3:0]
4 3
2
1
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Register Location Truth Table
The control bits CTL [2:0] decode the internal register address. The table below shows how the control bits are
mapped to the target control register.
C3
C2
C1
C0
DATA Location
x
x
x
0
R0
0
0
0
1
R1
0
0
1
1
R2
0
1
0
1
R3
0
1
1
1
R4
1
0
0
1
R5
1
0
1
1
R6
1
1
0
1
R7
1
1
1
1
R8
Control Register Content Map
Because the LMX2470 registers are complicated, they are organized into two groups, basic and advanced. The
first four registers are basic registers that contain critical information necessary for the PLL to achieve lock. The
last 5 registers are for features that optimize spur, phase noise, and lock time performance. The next page
shows these registers.
Quick Start Register Map
Although it is highly recommended that the user eventually take advantage of all the modes of the LMX2470, the quick start register map is
shown in order for the user to get the part up and running quickly using only those bits critical for basic functionality. The following default
conditions for this programming state are a third order delta-sigma modulator in 22-bit mode with no dithering and no Fastlock.
RE
GIS
TE
R
23
22
21
20
19
18
17
16
15
14
13
12
11
10
9
8
7
6
5
4
DATA[19:0] ( Except for the RF_N Register, which is [22:0] )
R0
RF_N[10:0]
3
2
1
0
C3
C2
C1
C0
RF_FN[11:0]
RF_R[5:0]
0
R1
RF
_P
D
1
RF_FD[11:0]
R2
IF_
PD
IF_
P
R3
0
R4
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
R5
0
0
0
1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
R6
0
0
0
0
0
0
0
0
0
0
1
1
0
0
0
1
0
0
R7
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
R8
0
0
0
0
1
1
0
0
0
0
0
0
1
0
0
0
8
IF_
CP
G
0
0
0
1
0
0
1
1
0
1
0
1
0
0
1
1
1
0
1
0
0
1
0
0
1
0
1
1
0
0
0
1
1
0
1
1
1
1
0
1
1
1
1
7
6
5
4
3
2
1
0
C3
C2
C1
C0
IF_N[16:0]
RF_CPG[3:0]
IF_R[14:0]
Complete Register Map
The complete register map shows all the functionality of all registers, including the last five.
RE
GIS
TE
R
23
22
21
20
19
18
17
16
15
14
13
12
11
10
9
DATA[19:0] ( Except for the RF_N Register, which is [22:0] )
R0
R1
22
RF_N[10:0]
RF
_P
D
1
RF_R[5:0]
RF_FN[11:0]
RF_FD[11:0]
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R2
IF_
PD
R3
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IF_
P
0
R4
IF_
CP
G
IF_N[17:0]
RF_CPG[3:0]
CSR[1:0]
IF_R[14:0]
RF_CPF[3:0]
RF_TOC[13:0]
R5
0
0
0
1
0
0
0
0
R6
0
0
0
0
0
RF
_C
PT
RF
_C
PP
IF_
CP
T
R7
IF_TOC[11:0]
IF_
CP
P
FD
M
FM[1:0]
ATPU[1:
0]
RF_FD2[9:0]
R8
0
0
0
0
OS
C2
X
OS
C
MUX[3:0]
RF_FN2[9:0]
DITH[1:0]
0
0
0
0
0
0
PDCP[1:
0]
0
0
CPUD[2:0]
0
0
0
1
1
0
1
0
1
0
1
1
1
1
0
0
1
1
0
1
1
1
1
0
1
1
1
1
1
R0 REGISTER
Note that this register has only one control bit. The reason for this is that it enables the N counter value to be
changed with a single write statement to the PLL.
RE
GIS
TE
R
23
22
21
20
19
18
17
16
15
14
13
12
11
10
9
8
7
6
5
4
DATA[19:0] ( Except for the RF_N Register, which is [22:0] )
R0
RF_N[10:0]
3
2
1
0
C3
C2
C1
C0
FN[11:0]
0
RF_FN[11:0] -- Fractional Numerator for RF PLL
Refer to section 2.8.1 for a more detailed description of this control word.
RF_N[10:0] -- RF N Counter Value
The RF N counter contains a 16/17/20/21 prescaler. Because there is only one selection of prescaler, the value
that is programmed is simply the N counter value converted into binary form. However, because this counter
does have a prescaler, there are limitations on the divider values.
RF_N[10:0]
RF_N
RF_C
RF_B
≤65
N values less than or equal to 65 are prohibited.
66
Possible only with a second order delta-sigma engine
67
Possible with a second or third order delta-sigma engine.
RF_A
68
0
0
0
0
1
0
0
0
1
0
0
69
0
0
0
0
1
0
0
0
1
0
1
...
.
.
.
.
.
.
.
.
.
.
.
2040
1
1
1
1
1
1
1
1
0
0
0
20412044
Possible with a second or third order delta-sigma engine.
20452046
Possible only with a second order delta-sigma engine.
>2046
N values above 2046 are prohibited.
R1 REGISTER
RE
GIS
TE
R
23
22
21
20
19
18
17
16
15
14
13
12
11
10
9
DATA[19:0] ( Except for the RF_N Register, which is [22:0] )
8
7
6
5
4
3
2
1
0
C3
C2
C1
C0
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R1
RF
_P
D
1
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RF_R[5:0]
RF_FD[11:0]
0
0
0
1
RF_FD[11:0] -- RF PLL Fractional Denominator
The function of these bits are described in section 2.8.2.
RF_R [5:0] -- RF R Divider Value
The RF R Counter value is determined by this control word. Note that this counter does allow values down to
one.
R Value
RF_R[5:0]
1
0
0
0
0
0
...
.
.
.
.
.
1
.
63
1
1
1
1
1
1
RF_PD -- RF Power Down Control Bit
When this bit is set to 0, the RF PLL operates normally. When it is set to one, the RF PLL is powered down and
the RF Charge pump is set to a TRI-STATE mode. Because the EN pin and ATPU[1:0] word also controls power
down functions, there may be some conflicts. The order of precedence is as follows. First, if the EN pin is LOW,
then the PLL will be powered down. Provided this is not the case, the PLL will be powered up if the ATPU[1:0]
word says to do so, regardless of the state of the RF_PD bit. After the EN pin and the ATPU[1:0] word are
considered, then the RF_PD bit then takes control of the power down function for the RF PLL.
R2 REGISTER
RE
GIS
TE
R
23
22
21
R2
IF_
PD
IF_
P
IF_
CP
G
20
19
18
17
16
15
14
13
12
11
10
9
8
7
6
5
4
DATA[19:0] ( Except for the RF_N Register, which is [22:0] )
IF_N[17:0]
3
2
1
0
C3
C2
C1
C0
0
0
1
1
IF_N[16:0] -- IF N Divider Value
The IF N divider is a classical dual modulus prescaler with a selectable 8/9 or 16/17 modulus. The IF_N value is
determined by the IF_A , IF_B, and IF_P values. Note that the IF_P word can assume a value of 8 or 16. The
RF_A and RF_B counter values can be determined in accordance with the following equations.
B = N div P
A = N mod P
B≥A is required in order to have a legal N divider ratio
Here the div operator is defined as the division of two numbers with the remainder disregarded and the mod
operator is defined as the remainder as a result of this division. For the purposes of programming, it turns out
that the register value is just the binary representation of the N value, with the exception that the 4th LSB is not
used and must be programmed to 0 when the 8/9 prescaler is used.
IF_N Programming with the 8/9 Prescaler
IF_N[16:0]
N
Value
IF_B
IF_A
<24
N Values Below 24 are prohibited since IF_B≥3 is required.
24-55
Legal divide ratios in this range are: 24-27, 32-36, 40-45, 48-54
56
24
0
0
0
0
0
0
0
0
0
0
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1
1
1
0
0
0
0
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SNAS195A – MAY 2004 – REVISED MAY 2004
...
.
.
.
.
.
.
.
.
.
.
.
.
.
0
.
.
.
6553
5
1
1
1
1
1
1
1
1
1
1
1
1
1
0
1
1
1
0
0
RF_N Programming with 16/17 Prescaler
IF_N[16:0]
N
Value
IF_B
IF_A
≤47
N values less than or equal to 47 are prohibited because IF_B≥3 is required.
48239
Legal divide ratios in this range are: 48-51, 64-68, 80-85, 96-102
240
0
...
.
0
1310
71
0
0
0
0
0
0
0
0
1
1
1
0
0
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
IF_CPG -- IF Charge Pump Gain
This bit determines the magnitude of the IF charge pump current
IF_CPG
IF Charge Pump Current (mA)
0
Low (1 mA)
1
High (4 mA)
IF_P -- IF Prescaler Value
This bit selects which prescaler will be used for the IF N counter.
IF_P
IF Prescaler Value
0
8 (8/9 Prescaler)
1
16 (16/17 Prescaler)
IF_PD -- IF Power Down Bit
When this bit is set to 0, the IF PLL operates normally. When it is set to 1, the IF PLL powers down and the
output of the IF PLL charge pump is set to a TRI-STATE mode. If the IF_CPT bit is set to 0, then the power
down state is synchronous and will not occur until the charge pump is off. If the IF_CPT bit is set to 1, then the
power down will occur immediately regardless of the state of the IF PLL charge pump.
R3 REGISTER
RE
GIS
TE
R
23
R3
0
22
21
20
19
18
17
16
15
14
13
12
11
10
9
8
7
6
5
4
DATA[19:0] ( Except for the RF_N Register, which is [22:0] )
RF_CPG[3:0]
IF_R[14:0]
3
2
1
0
C3
C2
C1
C0
0
1
0
1
IF_R[14:0] -- IF R Divider Value
For the IF R divider, the R value is determined by the IF_R[14:0] bits in the R3 register. The minimum value for
IF_R is 3.
R
Value
IF_R[14:0]
3
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
...
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
32767
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
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RF_CPG -- RF PLL Charge Pump Gain
This is used to control the magnitude of the RF PLL charge pump in steady state operation
RF_CPG[3:0]
RF Charge Pump Current (µA)
0
100
1
200
2
300
3
400
4
500
5
600
6
700
7
800
8
900
9
1000
10
1100
11
1200
12
1300
13
1400
14
1500
15
1600
R4 REGISTER
This register controls the conditions for the RF PLL in Fastlock.
RE
GIS
TE
R
23
22
R4
CSR[1:0]
21
20
19
18
17
16
15
14
13
12
11
10
9
8
7
DATA[19:0] ( Except for the RF_N Register, which is [22:0] )
RF_CPF[3:0]
RF_TOC[13:0]
6
5
4
3
2
1
0
C3
C2
C1
C0
0
1
1
1
RF_TOC -- RF Time Out Counter and Control for FLoutRF Pin
The RF_TOC[13:0] word controls the operation of the RF Fastlock circuitry as well as the function of the
FLoutRF output pin. When this word is set to a value between 0 and 3, the RF Fastlock circuitry is disabled and
the FLoutRF pin operates as a general purpose CMOS TRI-STATE I/O. When RF_TOC is set to a value
between 4 and 16383, the RF Fastlock mode is enabled and the FLoutRF pin is utilized as the RF Fastlock
output pin. The value programmed into the RF_TOC[13:0] word represents four times the number of phase
detector comparison cycles the RF synthesizer will spend in the Fastlock state.
26
RF_TOC[13:0]
Fastlock Mode
Fastlock Period [CP events]
FLoutRF Pin Functionality
0
Disabled
N/A
High Impedance
1
Manual
N/A
Logic “0” State.
Forces all fastlock conditions
2
Disabled
N/A
Logic “0” State
3
Disabled
N/A
Logic “1” State
4
Enabled
4X2 = 8
Fastlock
5
Enabled
5X2 = 10
Fastlock
…
Enabled
…
Fastlock
16383
Enabled
16383X2=32766
Fastlock
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RF_CPF -- RF PLL Fastlock Charge Pump Current
Specify the charge pump current for the Fastlock operation mode for the RF PLL. Note that the Fastlock charge
pump current, steady state current, and CSR control are all interrelated. Refer to section 4.0 for more details.
RF_CPF [3:0]
Fastlock Charge Pump Current (µA)
0000
100
0001
200
0010
300
0011
400
0100
500
0101
600
0110
700
0111
800
1000
900
1001
1000
1010
1100
1011
1200
1100
1300
1101
1400
1110
1500
1111
1600
RF_CSR[1:0] -- RF Cycle Slip Reduction
CSR controls the operation of the Cycle Slip Reduction Circuit. This circuit can be used to reduce the occurrence
of phase detector cycle slips. Note that the Fastlock charge pump current, steady state current, and CSR control
are all interrelated. The table below gives some rough guidelines. In the table below, fCOMP is the comparison
frequency, and BW is the loop bandwidth of the PLL system. The rough guideline gives an idea of when it makes
sense to use this cycle slip reduction based on the steady-state conditions of the PLL system.
CSR[1:0]
CSR State
Sample Rate Reduction Factor
Rough Guideline
0
Disabled
1
fCOMP < 100 × BW
1
Enabled
1/2
100 × BW < fCOMP < 200 × BW
2
Enabled
1/4
200 × BW < fCOMP < 400 × BW
3
Enabled
1/16
fCOMP > 400 × BW
R5 REGISTER
RE
GIS
TE
R
23
22
21
20
19
R5
0
0
0
0
0
18
17
16
15
14
13
12
11
10
9
8
7
6
DATA[19:0] ( Except for the RF_N Register, which is [22:0] )
0
0
0
IF_TOC[11:0]
5
4
3
2
1
0
C3
C2
C1
C0
1
0
0
1
IF_TOC[11:0] IF Timeout Counter for Fastlock
The IF_TOC word controls the operation of the IF Fastlock circuitry as well as the function of the FLoutIF output
pin. When IF_TOC is set to a value between 0 and 3, the IF Fastlock circuitry is disabled and the FLoutIF pin
operates as a general purpose CMOS TRI-STATE output. When IF_TOC is set to a value between 4 and 4095,
the IF Fastlock mode is enabled and FLoutIF is utilized as the IF Fastlock output pin. The value programmed into
IF_TOC represents the number of phase comparison cycles that the IF synthesizer will spend in the Fastlock
state.
IF_TOC[11:0]
Fastlock Mode
Fastlock Period [Charge Pump
Cycles]
FLoutIF Pin Functionality
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0
Disabled
N/A
High Impedance
1
Manual
N/A
Logic “0” State
Forces IF charge pump current to
4 mA
2
Disabled
N/A
Logic “0” State
3
Disabled
N/A
Logic “1” State
4
Enabled
5
Fastlock
…
Enabled
…
Fastlock
4095
Enabled
4095
Fastlock
R6 REGISTER
RE
GIS
TE
R
23
22
21
20
19
18
17
16
15
14
13
12
11
10
9
8
7
6
5
DATA[19:0] ( Except for the RF_N Register, which is [22:0] )
R6
0
0
0
0
0
RF
_
CP
T
RF
_
CP
P
IF_
CP
T
IF_
CP
P
FD
M
FM[1:0]
ATPU
[1:0]
OS
C
2X
OS
C
MUX
[3:0]
4
3
2
1
0
C3
C2
C1
C0
1
0
1
1
1 MUX[3:0] Frequency Out & Lock Detect MUX
These bits determine the output state of the Ftest/LD pin.
MUX[3:0]
28
Output Type
Output Description
0
0
0
0
High Impedance
Disabled
0
0
0
1
Push-Pull
General purpose
output, Logical “High”
State
0
0
1
0
Push-Pull
General purpose
output, Logical “Low”
State
0
0
1
1
Push-Pull
RF & IF Digital Lock
Detect
0
1
0
0
Push-Pull
RF Digital Lock
Detect
0
1
0
1
Push-Pull
IF Digital Lock Detect
0
1
1
0
Open Drain
RF & IF Analog Lock
Detect
0
1
1
1
Open Drain
RF Analog Lock
Detect
1
0
0
0
Open Drain
IF Analog Lock
Detect
1
0
0
1
Push-Pull
RF & IF Analog Lock
Detect
1
0
1
0
Push-Pull
RF Analog Lock
Detect
1
0
1
1
Push-Pull
IF Analog Lock
Detect
1
1
0
0
Push-Pull
IF R Divider divided
by 2
1
1
0
1
Push-Pull
IF N Divider divided
by 2
1
1
1
0
Push-Pull
RF R Divider divided
by 2
1
1
1
1
Push-Pull
RF N Divider divided
by 2
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OSC -- Differential Oscillator Mode Enable
This bit selects between single-ended and differential mode for the OSCin and OSCout* pins. When this bit is set
to 0, the RF R and IF R counters are driven in a single-ended fashion through the OSCin pin. Note that the
OSCin and OSCout* pin can not be used to drive a crystal. When this bit is set to 1, the OSCin and OSCout*
pins are used to drive these R counters differentially. In some cases, spur performance may be better when this
is set to differential mode, even if the R counters are being driven in a single-ended fashion. Current
consumption in differential mode is slightly higher than when in single-ended mode.
OSC2X -- Oscillator Doubler Enable
When this bit is set to 0, the oscillator doubler is disabled TCXO frequency presented to the IF R counter is
unaffected. Phase noise added by the doulber is negligible.
ATPU -- PLL Automatic Power Up
This word enables the PLLs to be automatically powered up when their respective registers are written to. Note
that since the IF Powerdown bit is in the IF register, there is no need to have an ATPU function activated by the
R2 word.
ATPU
RF PLL
IF PLL
0
No auto power up
No auto power up
1
Powers up when R0 is written to
No auto power up
2
Powers up when R0 is written to
Powers up when R0 is written to
3
Reserved
FM[1:0] -- Fractional Mode
Determines the order of the delta-sigma modulator. Higher order delta-sigma modulators reduce the spur levels
closer to the carrier by pushing this noise to higher frequency offsets from the carrier. In general, the order of the
loop filter should be at least one greater than the order of the delta-sigma modulator in order to allow for
sufficient roll-off.
FM
Function
0
Fractional PLL mode with a 4th order delta-sigma modulator
1
Disable the delta-sigma modulator. Recommended for test use only.
2
Fractional PLL mode with a 2nd order delta-sigma modulator
3
Fractional PLL mode with a 3rd order delta-sigma modulator
FDM -- Fractional Denominator Mode
When this bit is set to 0, the part operates with a 12- bit fractional denominator. For most applications, 12-bit
mode should be adequate, but for those applications requiring ultra fine tuning resolution, there is 22-bit mode.
Note that the PLL may consume slightly more current when it is in 22-bit mode.
FDM
Bits for Fractional
Denominator/Numerator
Maximum Size of Fractional
Denominator/Numerator
0
12-bit
4095
1
22-bit
4194303
IF_CPP -- IF PLL Charge Pump Polarity
When this bit is set to 1, the phase detector polarity for the IF PLL charge pump is positive. Otherwise set this bit
to 0 for a negative phase detector polarity
IF_CPT -- IF PLL Charge Pump TRI-STATE Mode
This bit enables the user to put the charge pump in a TRI-STATE ( high impedance ) condition. Note that if there
is a conflict, the ATPU bit overrides this bit.
RF_CPT
Charge Pump State
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0
ACTIVE
1
TRI-STATE
RF_CPP -- RF PLL Charge Pump Polarity
For a positive phase detector polarity, which is normally the case, set this bit to 1. Otherwise set this bit to 0 for a
negative phase detector polarity.
RF_CPT -- RF PLL Charge Pump TRI-STATE Mode
This bit enables the user to put the charge pump in a TRI-STATE ( high impedance) condition. Note that if there
is a conflict, the ATPU bit overrides this bit.
RF_CPT
Charge Pump State
0
Active
1
TRI-STATE
R7 REGISTER
RE
GIS
TE
R
23
22
21
20
19
18
17
16
15
14
13
12
11
10
9
8
7
6
5
4
DATA[19:0]
R7
RF_FD2[9:0]
3
2
1
0
C3
C2
C1
C0
1
1
0
1
RF_FN2[9:0]
Fractional Numerator Determination { RF_FN2[9:0], RF_FN[11:0], FDM }
In the case that the FDM bit is 0, then the part operates in 12-bit fractional mode, and the RF_FN2 bits become
don’t care bits. When the FDM is set to 1, the part operates in 22-bit mode and the fractional numerator is
expanded from 12 to 22-bits.
Fra
ctio
nal
RF_FN2[9:0]
Nu
mer
ator
( These bits only apply in 22- bit mode)
RF_FN[11:0]
0
1
In 12- bit mode, these are don’t care.
In 22- bit mode, for N <4096,
these bits should be all set to 0.
...
409
5
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
.
.
.
.
.
.
.
.
.
.
.
.
1
1
1
1
1
1
1
1
1
1
1
1
409
6
0
0
0
0
0
0
0
0
0
1
0
0
0
0
0
0
0
0
0
0
0
0
...
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
419
430
3
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
Fractional Denominator Determination { RF_FD2[9:0], RF_FD[11:0], FDM }
In the case that the FDM bit is 0, then the part is operates in the 12-bit fractional mode, and the RF_FD2 bits
become don’t care bits. When the FDM is set to 1, the part operates in 22-bit mode and the fractional
denominator is expanded from 12 to 22-bits.
Fra
ctio
nal
RF_FD2[9:0]
Den
omi
nat
or
( These bits only apply in 22- bit mode)
30
RF_FD[11:0]
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0
In 12- bit mode, these are don’t care.
In 22- bit mode, for N <4096,
these bits should be all set to 0.
1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
...
.
.
.
.
.
.
.
.
.
.
.
.
409
5
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
409
6
0
0
0
0
0
0
0
0
0
1
...
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
.
419
430
3
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
14
13
12
11
10
R8 REGISTER
23
22
21
20
19
18
17
16
15
9
8
7
6
5
4
DATA[19:0]
R8
0
0
0
0
DITH
[1:0]
0
0
0
0
0
0
PDCP
[1:0]
0
0
CPUD
[2:0]
0
3
2
1
0
C3
C2
C1
C0
1
1
1
1
The R8 Register controls some additional bits that may be useful in optimizing phase noise, lock time, and spurs.
CPUD[2:0] -- Charge Pump User Definition
This bit allows the user to choose from several different modes in the charge pump. The charge pump current is
unaffected, but the fractional spurs and phase noise are impacted by a few dB. In some designs, particularly if
the loop bandwidth is wide and a 4th order delta-sigma engine is used, small spurs may appear at a fraction of
where the first fractional spur should appear. In other designs, these sub-fractional spurs are not present. The
user needs to use this adjustment to make these sub-fractional spurs go away, while still getting the best phase
noise possible.
CPUD
Mode Name
Phase Noise
Sub-Fractional Spurs
0
Reserved
N/A
N/A
1
Reserved
N/A
N/A
2
Minimum
Best
Worst
3
Maximum
Worst
Best
4
Reserved
N/A
N/A
5
Reserved
N/A
N/A
6
Reserved
N/A
N/A
7
Nominal
Medium
Medium
PDCP[1:0] -- Power Drive for Charge Pump
If this bit is enabled, the Fastlock current can be doubled during Fastlock. The charge pump current in steady
state is unaffected. States 0 and 1 should never be used.
PDCP
Fastlock Charge Pump Current
0
Reserved
1
Reserved
2
Double Fastlock Current
3
Disabled
DITH[1:0] -- Dithering Control
Dithering is a technique used to spread out the spur energy. Enabling dithering can reduce the main fractional
spurs, but can also give rise to a family of smaller spurs. Whether dithering helps or hurts is application specific.
Enabling the dithering may also increase the phase noise. In most cases where the fractional numerator is zero,
dithering usually degrades performance.
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Determining tends to be most beneficial in applications where there is insufficient filtering of the spurs. This often
occurs when the loop bandwidth is very wide or a higher order delta-sigma modulator is used. Dithering tends
not to impact the main fractional spurs much, but has a much larger inpact on the sub-fractional spurs. If it is
decided that dithering will be used, best results will be obtained when the fractional denominator is at least 1000.
32
DITH
Dithering Mode Used
0
Dithering Enabled
1
Reserved
2
Reserved
3
Dithering Disabled
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PACKAGE OPTION ADDENDUM
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17-Nov-2012
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package Qty
Drawing
Eco Plan
Lead/Ball Finish
(2)
MSL Peak Temp
Samples
(3)
(Requires Login)
LMX2470SLEX
ACTIVE
ULGA
NPF
24
2500
Green (RoHS
& no Sb/Br)
NIAU
Level-1-260C-UNLIM
LMX2470SLEX/NOPB
ACTIVE
ULGA
NPF
24
2500
Green (RoHS
& no Sb/Br)
NIAU
Level-1-260C-UNLIM
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
17-Nov-2012
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
Diameter Width (mm)
(mm) W1 (mm)
B0
(mm)
K0
(mm)
P1
(mm)
W
Pin1
(mm) Quadrant
LMX2470SLEX
ULGA
NPF
24
2500
330.0
12.4
3.8
4.8
1.3
8.0
12.0
Q1
LMX2470SLEX/NOPB
ULGA
NPF
24
2500
330.0
12.4
3.8
4.8
1.3
8.0
12.0
Q1
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
17-Nov-2012
*All dimensions are nominal
Device
Package Type
Package Drawing
Pins
SPQ
Length (mm)
Width (mm)
Height (mm)
LMX2470SLEX
ULGA
NPF
24
2500
349.0
337.0
45.0
LMX2470SLEX/NOPB
ULGA
NPF
24
2500
349.0
337.0
45.0
Pack Materials-Page 2
MECHANICAL DATA
NPF0024A
www.ti.com
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