LINER LT8641 65v, 3.5a synchronous step-down silent switcher with 2.5î¼a quiescent current Datasheet

LT8641
65V, 3.5A Synchronous Step-Down
Silent Switcher with 2.5µA Quiescent Current
FEATURES
DESCRIPTION
Silent Switcher® Architecture
nn Ultralow EMI Emissions
nn Spread Spectrum Frequency Modulation
nn High Efficiency at High Frequency
nn Up to 95% Efficiency at 1MHz, 12V to 5V
IN
OUT
nn Up to 94% Efficiency at 2MHz, 12V to 5V
IN
OUT
nn Wide Input Voltage Range: 3V to 65V
nn 3.5A Maximum Continuous Output,
5A Peak Transient Output
nn Ultralow Quiescent Current Burst Mode® Operation
nn 2.5μA I Regulating 12V to 3.3V
Q
IN
OUT
nn Output Ripple < 10mV
P-P
nn Fast Minimum Switch On-Time: 35ns
nn Low Dropout Under All Conditions: 130mV at 1A
nn Safely Tolerates Inductor Saturation in Overload
nn Adjustable and Synchronizable: 200kHz to 3MHz
nn Peak Current Mode Operation
nn Output Soft-Start and Tracking
nn Small 18-Lead 3mm × 4mm QFN
The LT®8641 step-down regulator features Silent Switcher
architecture designed to minimize EMI emissions while
delivering high efficiency at frequencies up to 3MHz. Assembled in a 3mm × 4mm QFN, the monolithic construction with integrated power switches and inclusion of all
necessary circuitry yields a solution with a minimal PCB
footprint. An ultralow 2.5µA quiescent current—with the
output in full regulation—enables applications requiring
highest efficiency at very small load currents. Transient
response remains excellent and output voltage ripple is
below 10mVP-P at any load, from zero to full current.
nn
APPLICATIONS
Automotive and Industrial Supplies
General Purpose Step-Down
nn GSM Power Supplies
nn
nn
The LT8641 allows high VIN to low VOUT conversion at
high frequency with a fast minimum top switch on-time of
35ns. Operation is safe in overload even with a saturated
inductor.
Essential features are included and easy to use: An
open-drain PG pin signals when the output is in regulation. The SYNC/MODE pin selects between Burst Mode,
pulse-skipping, or spread spectrum mode, and also allows synchronization to an external clock. Soft-start and
tracking functionality is accessed via the TR/SS pin. An
accurate enable threshold can be set using the EN/UV pin
and a resistor at the RT pin programs switch frequency.
All registered trademarks and trademarks are the property of their respective owners. Protected
by U.S. patents, including 8823345.
TYPICAL APPLICATION
12VIN to 5VOUT Efficiency
5V 3.5A Step-Down Converter
4.7µF
1µF
EN/UV
GND1
PG
10nF
SYNC/MODE
TR/SS
BST
fSW = 1MHz
1µF
0.1µF 4.7µH
4.7pF
INTVCC
RT
VOUT
5V
3.5A
SW
BIAS
1µF
41.2k
95
2.275
90
VIN2
GND2
LT8641
2.600
1M
47µF
FB
GND
191k
85
1.625
80
1.300
0.975
75
70
POWER LOSS
65
60
0.5
8641 TA01a
1.950
EFFICIENCY
1
POWER LOSS (W)
VIN1
EFFICIENCY (%)
VIN
5.5V TO 65V
100
0.650
1MHz, L = 3.3μH
2MHz, L = 2.2μH 0.325
3MHz, L = 1.5μH
0
1.5
2
2.5
3
3.5
LOAD CURRENT (A)
8641 TA01b
Rev A
Document Feedback
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1
LT8641
PIN CONFIGURATION
VIN, EN/UV.................................................................65V
PG..............................................................................42V
BIAS...........................................................................25V
FB, TR/SS ...................................................................4V
SYNC/MODE Voltage ..................................................6V
Operating Junction Temperature Range (Note 2)
LT8641E.............................................. –40°C to 125°C
LT8641I............................................... –40°C to 125°C
LT8641H............................................. –40°C to 150°C
Storage Temperature Range................... –65°C to 150°C
SYNC/MODE
17
PG
20 19 18
FB
GND
TOP VIEW
16 TR/SS
BIAS 1
INTVCC 2
15 RT
BST 3
GND1 6
21
SW
14 EN/UV
13 VIN2
11 GND2
7
8
9
10
GND2
22
SW
SW
VIN1 4
SW
(Note 1)
GND1
ABSOLUTE MAXIMUM RATINGS
UDC PACKAGE
18-LEAD (3mm × 4mm) PLASTIC QFN
θJA = 40°C/W, θJC(PAD) = 12°C/W (NOTE 3)
EXPOSED PAD (PINS 21, 22) ARE SW, SHOULD BE SOLDERED TO PCB
NOTE: PINS 5 AND 12 ARE REMOVED. CONFIGURATION DOES NOT MATCH
JEDEC 20-LEAD PACKAGE OUTLINE
ORDER INFORMATION
http://www.linear.com/product/LT8641#orderinfo
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT8641EUDC#PBF
LT8641EUDC#TRPBF
LGSN
18-Lead (3mm × 4mm) Plastic QFN
–40°C to 125°C
LT8641IUDC#PBF
LT8641IUDC#TRPBF
LGSN
18-Lead (3mm × 4mm) Plastic QFN
–40°C to 125°C
LT8641HUDC#PBF
LT8641HUDC#TRPBF
LGSN
18-Lead (3mm × 4mm) Plastic QFN
–40°C to 150°C
Consult ADI Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/. Some packages are available in 500 unit reels through
designated sales channels with #TRMPBF suffix.
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C.
PARAMETER
CONDITIONS
MIN
Minimum Input Voltage
VIN Quiescent Current
TYP
MAX
l
2.6
3.0
V
l
0.75
0.75
3
10
µA
µA
l
1.7
1.7
4
10
µA
µA
0.3
0.5
mA
17
200
50
350
µA
µA
0.81
0.81
0.816
0.822
V
V
0.004
0.03
%/V
VEN/UV = 0V
VEN/UV = 2V, Not Switching, VSYNC = 0V
VEN/UV = 2V, Not Switching, VSYNC = 2V
VIN Current in Regulation
VOUT = 0.8V, VIN = 6V, Output Load = 100µA
VOUT = 0.8V, VIN = 6V, Output Load = 1mA
l
l
Feedback Reference Voltage
VIN = 6V, ILOAD = 0.5A
VIN = 6V, ILOAD = 0.5A
l
VIN = 4.0V to 42V, ILOAD = 0.5A
l
Feedback Voltage Line Regulation
2
0.804
0.79
UNITS
Rev A
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LT8641
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C.
PARAMETER
CONDITIONS
MIN
Feedback Pin Input Current
VFB = 1V
–20
BIAS Pin Current Consumption
VBIAS = 3.3V, ILOAD = 1A, 2MHz
Minimum On-Time
ILOAD = 1.5A, SYNC = 0V
ILOAD = 1.5A, SYNC = 2V
RT = 221k, ILOAD = 1A
RT = 60.4k, ILOAD = 1A
RT = 18.2k, ILOAD = 1A
Top Power NMOS On-Resistance
ISW = 1A
MAX
20
9
l
l
Minimum Off-Time
Oscillator Frequency
TYP
l
l
l
180
665
1.85
l
Bottom Power NMOS On-Resistance
VINTVCC = 3.4V, ISW = 1A
Bottom Power NMOS Current Limit
VINTVCC = 3.4V
SW Leakage Current
VIN = 42V, VSW = 0V, 42V
mA
50
50
ns
ns
80
110
ns
210
700
2.00
240
735
2.15
kHz
kHz
MHz
EN/UV Pin Threshold
EN/UV Rising
6.2
8.2
4.8
5.8
mΩ
9.9
55
–15
l
0.95
EN/UV Pin Hysteresis
nA
35
35
105
Top Power NMOS Current Limit
UNITS
1.01
mΩ
7.25
A
15
µA
1.07
45
–20
A
V
mV
EN/UV Pin Current
VEN/UV = 2V
20
nA
PG Upper Threshold Offset from VFB
VFB Falling
l
5
7.5
10.25
%
PG Lower Threshold Offset from VFB
VFB Rising
l
–5.25
–8
–10.75
%
40
nA
750
2000
Ω
0.9
1.2
2.6
1.4
2.9
V
V
V
PG Hysteresis
0.4
PG Leakage
VPG = 3.3V
PG Pull-Down Resistance
VPG = 0.1V
SYNC/MODE Threshold
SYNC/MODE DC and Clock Low Level Voltage
SYNC/MODE Clock High Level Voltage
SYNC/MODE DC High Level Voltage
Spread Spectrum Modulation
Frequency Range
–40
l
0.7
2.3
RT = 60.4k, VSYNC = 3.3V
Spread Spectrum Modulation Frequency VSYNC = 3.3V
TR/SS Source Current
TR/SS Pull-Down Resistance
l
Fault Condition, TR/SS = 0.1V
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LT8641E is guaranteed to meet performance specifications
from 0°C to 125°C junction temperature. Specifications over the –40°C
to 125°C operating junction temperature range are assured by design,
characterization, and correlation with statistical process controls. The
LT8641I is guaranteed over the full –40°C to 125°C operating junction
temperature range. The LT8641H is guaranteed over the full –40°C to
150°C operating junction temperature range. High junction temperatures
degrade operating lifetimes. Operating lifetime is derated at junction
temperatures greater than 125°C.
1.2
%
22
%
2.5
kHz
1.9
220
2.6
µA
Ω
The junction temperature (TJ, in °C) is calculated from the ambient
temperature (TA in °C) and power dissipation (PD, in Watts) according to
the formula:
TJ = TA + (PD • θJA)
where θJA (in °C/W) is the package thermal impedance.
Note 3: θ values determined per JEDEC 51-7, 51-12. See Applications
Information section for information on improving the thermal resistance
and for actual temperature measurements of a demo board in typical
operating conditions.
Note 4: This IC includes overtemperature protection that is intended to
protect the device during overload conditions. Junction temperature will
exceed 150°C when overtemperature protection is active. Continuous
operation above the specified maximum operating junction temperature
will reduce lifetime.
Rev A
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3
LT8641
TYPICAL PERFORMANCE CHARACTERISTICS
12VIN to 5VOUT Efficiency
vs Frequency
12VIN to 3.3VOUT Efficiency
vs Frequency
2.3
95
1.9
80
1.3
1.0
75
POWER LOSS
65
60
0.5
1
L = WE–LHMI7050 0.7
1MHz, L = 3.3μH
2MHz, L = 2.2μH 0.3
3MHz, L = 1.5μH
0
1.5
2
2.5
3
3.5
LOAD CURRENT (A)
1.2
80
0.9
75
POWER LOSS
70
L = WE–LHMI7050 0.6
1MHz, L = 2.2µH
2MHz, L = 1.5µH 0.3
3MHz, L = 1µH
0
1.5
2
2.5
3
3.5
LOAD CURRENT (A)
65
60
0.5
1
8641 G01
Efficiency at 3.3V OUT
EFFICIENCY
fSW = 1MHz
L = WE–LHMI7050, 2.2µH 2.3
90
2.0
85
1.8
80
1.5
75
1.3
70
1.0
65
0.8
60
55
50
POWER LOSS
0
0.5
100
2.5
1
1.5
2
2.5
LOAD CURRENT (A)
POWER LOSS (W)
EFFICIENCY (%)
95
2.0
85
1.8
80
1.5
fSW = 1MHz
75 L = WE–LHMI7050, 4.7µH
1.0
65
0.8
VIN = 12V
VIN = 24V 0.5
VIN = 36V 0.3
VIN = 48V
POWER LOSS
0
0.5
1
1.5
2
2.5
3
3.5
LOAD CURRENT (A)
60
55
50
0
8641 G03
Efficiency at 5V OUT
100
90
90
80
80
70
60
50
VIN = 12V
0.5
VIN = 24V
VIN = 36V 0.3
VIN = 48V
0
3
3.5
40
fSW = 1MHz
L = WE–LHMI7050, 4.7µH
30
20
0.01
0.1
8641 G04
VIN = 12V
VIN = 24V
VIN = 36V
VIN = 48V
1
10
100
LOAD CURRENT (mA)
Efficiency at 3.3V OUT
70
60
50
40
30
20
0.01
1000
fSW = 1MHz
L = WE–LHMI7050, 4.7µH
0.1
1
10
100
LOAD CURRENT (mA)
8641 G05
96
88
VIN = 24V
86
82
80
0.5
VIN = 24V
85
80
75
VOUT = 3.3V
ILOAD = 1.5A
L = WE–LHMI7050, 4.7µH
1.0
1.5
2.0
2.5
SWITCHING FREQUENCY (MHz)
VOUT = 5V
ILOAD = 10mA
L = WE–LHMI7050
70
3.0
8641 G07
4
817
65
1
2
3
4
5
6
7
8
INDUCTOR VALUE (µH)
9
10
8641 G08
REFERENCE VOLTAGE (mV)
90
84
VIN = 12V
90
EFFICIENCY (%)
EFFICIENCY (%)
92
819
95
VIN = 12V
1000
Reference
Reference Voltage
Voltage
100
94
VIN = 12V
V = 24V
VIN = 36V
VIN = 48V
8641 G06
Burst Mode Operation Efficiency
vs Inductor Value
Efficiency vs Frequency
1.3
70
8641 G02
EFFICIENCY (%)
100
1.5
2.3
90
EFFICIENCY (%)
70
EFFICIENCY
85
2.5
EFFICIENCY
POWER LOSS (W)
1.6
1.8
POWER LOSS (W)
85
90
EFFICIENCY (%)
EFFICIENCY
Efficiency at 5V OUT
95
2.1
95
POWER LOSS (W)
EFFICIENCY (%)
90
100
2.4
100
EFFICIENCY (%)
2.6
100
815
813
811
809
807
805
803
801
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
8641 G09
Rev A
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LT8641
TYPICAL PERFORMANCE CHARACTERISTICS
EN
EN Pin
Pin Thresholds
Thresholds
1.02
0.99
0.98
EN FALLING
0
0.05
0
–0.05
0
0.5
1
1.5
2
2.5
LOAD CURRENT (A)
3
–0.15
2.0
15
25
35
45
INPUT VOLTAGE (V)
55
65
8641 G12
Top
Top FET
FET Current
Current Limit
Limit
10
9
CURRENT LIMIT (A)
2.5
5
8.0
CURRENT LIMIT (A)
7.5
7.0
6.5
8
5% DC
7
6.0
0
10
20
30
40
50
INPUT VOLTAGE (V)
60
5.5
0.1
0.3
0.5
DUTY CYCLE
0.7
6
–50 –25
0.9
8641 G14
8641 G13
Switch Drop
500
SWITCH CURRENT = 1A
43
40
400
SWITCH DROP (mV)
150
TOP SWITCH
100
50
350
300
250
TOP SWITCH
200
150
100
BOTTOM SWITCH
0
25 50 75 100 125 150
TEMPERATURE (°C)
0
0
0.5
1
1.5
2
2.5
SWITCH CURRENT (A)
3
3.5
8641 G17
8641 G16
37
34
31
28
BOTTOM SWITCH
50
0
–50 –25
VSYNC = 0
VSYNC = FLOAT
450
200
25 50 75 100 125 150
TEMPERATURE (°C)
Minimum On-Time
Switch Drop
250
0
8641 G15
MINIMUM ON-TIME (ns)
INPUT CURRENT (µA)
3.0
1.5
SWITCH DROP (mV)
3.5
8.5
VOUT = 3.3V
L = 4.7µH
IN-REGULATION
3.5
1.0
–0.05
Top FET Current Limit vs Duty Cycle
No-Load Supply Current
4.0
0
8641 G11
8641 G10
4.5
0.05
–0.10
–0.15
25 50 75 100 125 150
TEMPERATURE (°C)
VOUT = 5V
ILOAD = 1A
0.10
–0.10
0.96
0.95
–50 –25
VOUT = 5V
VIN = 12V
CHANGE IN VOUT (%)
CHANGE IN VOUT (%)
EN THRESHOLD (V)
1.00
0.97
0.15
0.10
EN RISING
1.01
Line Regulation
Load Regulation
0.15
1.03
25
–50
ILOAD = 2A
–25
0
25
50
75
TEMPERATURE (°C)
100
125
8641 G18
Rev A
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5
LT8641
TYPICAL PERFORMANCE CHARACTERISTICS
Dropout Voltage
VIN = 5V
VOUT SET TO REGULATE AT 5V
L = WE–LHMI7050, 1µH
400
300
200
100
0
720
710
700
690
680
0
0.5
1
1.5
2
2.5
LOAD CURRENT (A)
3
660
–50 –25
3.5
0
400
200
0
25 50 75 100 125 150
TEMPERATURE (°C)
60
40
20
200
400
600
LOAD CURRENT (mA)
600
800
Soft-Start Tracking
1.0
0.8
FB VOLTAGE (V)
SWITCHING FREQUENCY (kHz)
80
0
8641 G21
VOUT = 3.3V
VIN = 12V
VSYNC = 0V
RT = 60.4k
700
100
0
600
Frequency Foldback
800
FRONT PAGE APPLICATION
VOUT = 5V
fSW = 1MHz
120
800
8641 G20
Minimum Load to Full Frequency
(Pulse-Skipping Mode)
140
FRONT PAGE APPLICATION
VIN = 12V
VOUT = 5V
1000
670
8641 G19
LOAD CURRENT (mA)
Burst Frequency
1200
RT = 60.4k
730
SWITCHING FREQUENCY (kHz)
500
DROPOUT VOLTAGE (mV)
Switching
SwitchingFrequency
Frequency
740
SWITCHING FREQUENCY (kHz)
600
500
400
300
200
0.6
0.4
0.2
100
5
15
25
35
45
INPUT VOLTAGE (V)
55
0
65
0
0.2
8641 G22
0.4
0.6
FB VOLTAGE (V)
0.8
0
1
0
0.2
0.4
0.6
0.8
TR/SS VOLTAGE (V)
1.0
1.2
8641 G24
8641 G23
Soft-StartCurrent
Current
Soft–Start
VSS = 0.5V
2.0
1.9
1.8
1.7
1.6
1.5
1.4
–50 –25
0
–6.0
25 50 75 100 125 150
TEMPERATURE (°C)
9.5
9.0
8.5
FB RISING
8.0
7.5
FB FALLING
7.0
6.5
6.0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
8641 G25
6
PG THRESHOLD OFFSET FROM VREF (%)
SS PIN CURRENT (µA)
2.1
PG
PG Low
Low Thresholds
Thresholds
PG High Thresholds
10.0
PG THRESHOLD OFFSET FROM VREF (%)
2.2
8641 G26
–6.5
–7.0
–7.5
FB RISING
–8.0
–8.5
FB FALLING
–9.0
–9.5
–10.0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
8641 G27
Rev A
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LT8641
TYPICAL PERFORMANCE CHARACTERISTICS
RT Programmed Switching
Frequency
VMinimum
IN UVLO Input Voltage
250
Bias Pin Current
3.2
7.5
3.0
7.0
VBIAS = 5V
VOUT = 5V
ILOAD = 1A
fSW = 1MHz
175
INPUT VOLTAGE (V)
RT PIN RESISTOR (kΩ)
200
150
125
100
75
50
BIAS PIN CURRENT (mA)
225
2.8
2.6
2.4
0
0.2
0.6
1.4 1.8 2.2 2.6
1
SWITCHING FREQUENCY (MHz)
2.0
–50 –25
3
6.0
5.5
5.0
2.2
25
6.5
0
4.5
25 50 75 100 125 150
TEMPERATURE (°C)
5
15
25
35
45
INPUT VOLTAGE (V)
55
65
8641 G30
8641 G29
8641 G28
10
5
0
0.2
0.6 1.0 1.4 1.8 2.2 2.6
SWITCHING FREQUENCY (MHz)
3.0
50
40
20
10
0
500ns/DIV
FRONT PAGE APPLICATION
12VIN TO 5VOUT AT 1A
8641 G34
70
60
50
40
30
20
10
0
0.5
1
1.5
2
2.5
LOAD CURRENT (A)
3
3.5
0
0
0.2
0.4
0.6
DUTY CYCLE OF 5A LOAD
Switching Waveforms, Burst
Mode Operation
Switching Waveforms
IL
500mA/DIV
IL
1A/DIV
VSW
5V/DIV
VSW
20V/DIV
10µs/DIV
FRONT PAGE APPLICATION
12VIN TO 5VOUT AT 10mA
VSYNC = 0V
0.8
8641 G33
8641 G32
Switching Waveforms, Full
Frequency Continuous Operation
VSW
5V/DIV
DC2373A DEMO BOARD
VIN = 12V
VOUT = 5V
fSW = 2MHz
STANDBY LOAD = 0.25A
1kHz PULSED LOAD = 5A
80
30
8641 G31
IL
1A/DIV
DC2373A DEMO BOARD
VIN = 12V, fSW = 1MHz
VIN = 24V, fSW = 1MHz
VIN = 12V, fSW = 2MHz
VIN = 24V, fSW = 2MHz
CASE TEMPERATURE RISE (°C)
15
90
60
VBIAS = 5V
VOUT = 5V
VIN = 12V
ILOAD = 1A
CASE TEMPERATURE RISE (°C)
BIAS PIN CURRENT (mA)
20
Case Temperature Rise vs 5A
Pulsed Load
Case
Case Temperature
Temperature Rise
Rise
Bias Pin Current
8641 G35
500ns/DIV
FRONT PAGE APPLICATION
48VIN TO 5VOUT AT 1A
8641 G36
Rev A
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7
LT8641
TYPICAL PERFORMANCE CHARACTERISTICS
Transient Response; Load Current
Stepped from 300mA (Burst Mode
Operation) to 1.3A
Transient Response; Load Current
Stepped from 1A to 2A
ILOAD
1A/DIV
ILOAD
1A/DIV
VOUT
100mA/DIV
VOUT
200mA/DIV
8641 G37
50µs/DIV
50µs/DIV
FRONT PAGE APPLICATION
300mA (Burst Mode OPERATION) TO
1.3A TRANSIENT
12VIN, 5VOUT
COUT = 47µF
FRONT PAGE APPLICATION
1A TO 2A TRANSIENT
12VIN, 5VOUT
COUT = 47µF
Start-Up Dropout Performance
Start-Up Dropout Performance
VIN
VIN
2V/DIV
VIN
VIN
2V/DIV
VOUT
VOUT
2V/DIV
8641 G38
VOUT
VOUT
2V/DIV
100ms/DIV
2.5Ω LOAD
(2A IN REGULATION)
100ms/DIV
20Ω LOAD
(250mA IN REGULATION)
8641 G39
8641 G40
Conducted EMI Performance
60
50
AMPLITUDE (dBµV)
40
30
20
10
0
–10
–20
SPREAD SPECTRUM MODE
FIXED FREQUENCY MODE
–30
–40
0
3
6
9
12
15
18
21
FREQUENCY (MHz)
24
27
30
8641 G41
DC2373A DEMO BOARD
(WITH EMI FILTER INSTALLED)
14V INPUT TO 5V OUTPUT AT 3.5A, fSW = 2MHz
8
Rev A
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LT8641
TYPICAL PERFORMANCE CHARACTERISTICS
Radiated EMI Performance
(CISPR25 Radiated Emission Test with Class 5 Peak Limits)
50
VERTICAL POLARIZATION
PEAK DETECTOR
45
AMPLITUDE (dBµV/m)
40
35
30
25
20
15
10
5
CLASS 5 PEAK LIMIT
FIXED FREQUENCY MODE
SPREAD SPECTRUM MODE
0
–5
0
100
200
300
400
500
600
700
800
900
1000
FREQUENCY (MHz)
50
HORIZONTAL POLARIZATION
PEAK DETECTOR
45
AMPLITUDE (dBµV/m)
40
35
30
25
20
15
10
5
CLASS 5 PEAK LIMIT
FIXED FREQUENCY MODE
SPREAD SPECTRUM MODE
0
-5
0
100
200
300
400
500
600
700
800
900
1000
FREQUENCY (MHz)
DC2373A DEMO BOARD
(WITH EMI FILTER INSTALLED)
14V INPUT TO 5V OUTPUT AT 3.5A, fSW = 2MHz
8641 G42
Rev A
For more information www.analog.com
9
LT8641
PIN FUNCTIONS
BIAS (Pin 1): The internal regulator will draw current
from BIAS instead of VIN when BIAS is tied to a voltage
higher than 3.1V. For output voltages of 3.3V to 25V this
pin should be tied to VOUT. If this pin is tied to a supply
other than VOUT use a 1µF local bypass capacitor on this
pin. If no supply is available, tie to GND.
GND2 (10, 11): Power Switch Ground. These pins are the
return path of the internal bottom side power switch and
must be tied together. Place the negative terminal of the
input capacitor as close to the GND2 pins as possible. Also
be sure to tie GND2 to the ground plane. See the Applications Information section for sample layout.
INTVCC (Pin 2): Internal 3.4V Regulator Bypass Pin. The
internal power drivers and control circuits are powered from
this voltage. INTVCC maximum output current is 20mA.
Do not load the INTVCC pin with external circuitry. INTVCC
current will be supplied from BIAS if BIAS > 3.1V, otherwise
current will be drawn from VIN. Voltage on INTVCC will
vary between 2.8V and 3.4V when BIAS is between 3.0V
and 3.6V. Decouple this pin to power ground with at least
a 1μF low ESR ceramic capacitor placed close to the IC.
VIN2 (Pin 13): The LT8641 requires two 1µF small input
bypass capacitors. One 1µF capacitor should be placed
between VIN1 and GND1. A second 1µF capacitor should
be placed between VIN2 and GND2. These capacitors must
be placed as close as possible to the LT8641. A third larger
capacitor of 2.2µF or more should be placed close to the
LT8641 with the positive terminal connected to VIN1 and
VIN2, and the negative terminal connected to ground. See
the Applications Information section for sample layout.
BST (Pin 3): This pin is used to provide a drive voltage,
higher than the input voltage, to the topside power switch.
Place a 0.1µF boost capacitor as close as possible to the IC.
EN/UV (Pin 14): The LT8641 is shut down when this pin
is low and active when this pin is high. The hysteretic
threshold voltage is 1.00V going up and 0.96V going
down. Tie to VIN if the shutdown feature is not used. An
external resistor divider from VIN can be used to program
a VIN threshold below which the LT8641 will shut down.
VIN1 (Pin 4): The LT8641 requires two 1µF small input
bypass capacitors. One 1µF capacitor should be placed
between VIN1 and GND1. A second 1µF capacitor should
be placed between VIN2 and GND2. These capacitors must
be placed as close as possible to the LT8641. A third larger
capacitor of 2.2µF or more should be placed close to the
LT8641 with the positive terminal connected to VIN1 and
VIN2, and the negative terminal connected to ground. See
applications section for sample layout.
GND1 (6, 7): Power Switch Ground. These pins are the
return path of the internal bottom side power switch and
must be tied together. Place the negative terminal of the
input capacitor as close to the GND1 pins as possible. Also
be sure to tie GND1 to the ground plane. See the Applications Information section for sample layout.
SW (Pins 8, 9): The SW pins are the outputs of the internal
power switches. Tie these pins together and connect them
to the inductor and boost capacitor. This node should be
kept small on the PCB for good performance and low EMI.
10
RT (Pin 15): A resistor is tied between RT and ground to
set the switching frequency.
TR/SS (Pin 16): Output Tracking and Soft-Start Pin. This
pin allows user control of output voltage ramp rate during
start-up. A TR/SS voltage below 0.8V forces the LT8641 to
regulate the FB pin to equal the TR/SS pin voltage. When
TR/SS is above 0.8V, the tracking function is disabled
and the internal reference resumes control of the error
amplifier. An internal 1.9μA pull-up current from INTVCC
on this pin allows a capacitor to program output voltage
slew rate. This pin is pulled to ground with an internal 200Ω
MOSFET during shutdown and fault conditions; use a series
resistor if driving from a low impedance output. This pin
may be left floating if the tracking function is not needed.
Rev A
For more information www.analog.com
LT8641
PIN FUNCTIONS
SYNC/MODE (Pin 17): This pin programs four different
operating modes: 1) Burst Mode. Tie this pin to ground
for Burst Mode operation at low output loads—this will
result in ultralow quiescent current. 2) Pulse-skipping
mode. Float this pin for pulse-skipping mode. This mode
offers full frequency operation down to low output loads
before pulse skipping occurs. When floating, pin leakage
currents should be <1µA. 3) Spread spectrum mode. Tie
this pin high to INTVCC (~3.4V) or an external supply of
3V to 4V. for pulse-skipping mode with spread spectrum
modulation. 4) Synchronization mode. Drive this pin with
a clock source to synchronize to an external frequency.
During synchronization the part will operate in pulseskipping mode.
PG (Pin 19): The PG pin is the open-drain output of an
internal comparator. PG remains low until the FB pin is
within ±8% of the final regulation voltage, and there are
no fault conditions. PG is valid when VIN is above 3.4V,
regardless of EN/UV pin state.
FB (Pin 20): The LT8641 regulates the FB pin to 0.8V.
Connect the feedback resistor divider tap to this pin. Also,
connect a phase lead capacitor between FB and VOUT.
Typically, this capacitor is 4.7pF to 22pF.
SW (Exposed Pad Pins 21, 22): The exposed pads should
be connected and soldered to the SW trace for good thermal
performance. If necessary due to manufacturing limitations Pins 21 and 22 may be left disconnected, however
thermal performance will be degraded.
GND (Pins 18): LT8641 Ground Pin. Connect this pin to
system ground and to the ground plane.
BLOCK DIAGRAM
VIN
4
CIN3
VIN2
VIN1
CIN1
R3
OPT
14
R4
OPT
19
EN/UV
1V
PG
+
–
SHDN
±8%
C1
R2
R1
20
CSS
OPT
16
RT
15
17
FB
TR/SS
INTVCC
OSCILLATOR
200kHz TO 3MHz
ERROR
AMP
VC
BST
BURST
DETECT
SHDN
THERMAL SHDN
INTVCC UVLO
VIN UVLO
1.9µA
BIAS
3.4V
REG
SLOPE COMP
+
+
–
VOUT
CIN2
–
+
INTERNAL 0.8V REF
13
INTVCC
SHDN
THERMAL SHDN
VIN UVLO
SWITCH
LOGIC
AND
ANTISHOOT
THROUGH
1
2
CVCC
3
CBST
M1
L
SW
8, 9, 21, 22
VOUT
COUT
M2
GND1
6, 7
600k
GND2
10, 11
RT
60k
SYNC/MODE
GND
18
8641 BD
Rev A
For more information www.analog.com
11
LT8641
OPERATION
The LT8641 is a monolithic, constant frequency, current
mode step-down DC/DC converter. An oscillator, with
frequency set using a resistor on the RT pin, turns on
the internal top power switch at the beginning of each
clock cycle. Current in the inductor then increases until
the top switch current comparator trips and turns off the
top power switch. The peak inductor current at which
the top switch turns off is controlled by the voltage on
the internal VC node. The error amplifier servos the VC
node by comparing the voltage on the VFB pin with an
internal 0.8V reference. When the load current increases
it causes a reduction in the feedback voltage relative to
the reference leading the error amplifier to raise the VC
voltage until the average inductor current matches the new
load current. When the top power switch turns off, the
synchronous power switch turns on until the next clock
cycle begins or inductor current falls to zero. If overload
conditions result in more than 5.5A flowing through the
bottom switch, the next clock cycle will be delayed until
switch current returns to a safe level.
If the EN/UV pin is low, the LT8641 is shut down and
draws 1µA from the input. When the EN/UV pin is above
1V, the switching regulator will become active.
To optimize efficiency at light loads, the LT8641 operates
in Burst Mode operation in light load situations. Between
bursts, all circuitry associated with controlling the output
switch is shut down, reducing the input supply current to
1.7μA. In a typical application, 2.5μA will be consumed
from the input supply when regulating with no load. The
SYNC/MODE pin is tied low to use Burst Mode operation
and can be floated to use pulse-skipping mode. If a clock is
applied to the SYNC/MODE pin the part will synchronize to
12
an external clock frequency and operate in pulse-skipping
mode. While in pulse-skipping mode the oscillator operates
continuously and positive SW transitions are aligned to
the clock. During light loads, switch pulses are skipped to
regulate the output and the quiescent current will be several
hundred µA. The SYNC/MODE pin may be tied high for
pulse-skipping mode with spread spectrum modulation.
To improve EMI the LT8641 can operate in spread spectrum mode. This feature varies the clock with a triangular frequency modulation of +20%. For example, if the
LT8641’s frequency is programmed to switch at 2MHz,
spread spectrum mode will modulate the oscillator between
2MHz and 2.4MHz.
To improve efficiency across all loads, supply current to
internal circuitry can be sourced from the BIAS pin when
biased at 3.3V or above. Else, the internal circuitry will draw
current from VIN. The BIAS pin should be connected to
VOUT if the LT8641 output is programmed at 3.3V to 25V.
Comparators monitoring the FB pin voltage will pull the
PG pin low if the output voltage varies more than ±8%
(typical) from the set point, or if a fault condition is present.
The oscillator reduces the LT8641’s operating frequency
when the voltage at the FB pin is low. This frequency
foldback helps to control the inductor current when the
output voltage is lower than the programmed value which
occurs during start-up or overcurrent conditions. When a
clock is applied to the SYNC/MODE pin, the SYNC/MODE
pin is floated, or held DC high, the frequency foldback is
disabled and the switching frequency will slow down only
during overcurrent conditions.
Rev A
For more information www.analog.com
LT8641
APPLICATIONS INFORMATION
Low EMI PCB Layout
should be as small as possible by placing the capacitors
adjacent to the VIN1/2 and GND1/2 pins. Capacitors with
small case size such as 0603 are optimal due to lowest
parasitic inductance.
The LT8641 is specifically designed to minimize EMI emissions and also to maximize efficiency when switching at
high frequencies. For optimal performance the LT8641
requires the use of multiple VIN bypass capacitors.
The input capacitors, along with the inductor and output
capacitors, should be placed on the same side of the
circuit board, and their connections should be made on
that layer. Place a local, unbroken ground plane under the
application circuit on the layer closest to the surface layer.
The SW and BOOST nodes should be as small as possible.
Finally, keep the FB and RT nodes small so that the ground
traces will shield them from the SW and BOOST nodes.
The exposed pad on the bottom of the package should be
soldered to SW to reduce thermal resistance to ambient. To
keep thermal resistance low, extend the ground plane from
GND1 and GND2 as much as possible, and add thermal
vias to additional ground planes within the circuit board
and on the bottom side.
Two small 1µF capacitors should be placed as close as
possible to the LT8641: One capacitor should be tied to
VIN1/GND1; a second capacitor should be tied to VIN2/
GND2. A third capacitor with a larger value, 2.2µF or
higher, should be placed near VIN1 or VIN2.
See Figure 1 for a recommended PCB layout.
For more detail and PCB design files refer to the Demo
Board guide for the LT8641.
Note that large, switched currents flow in the LT8641
VIN1, VIN2, GND1, and GND2 pins and the input capacitors
(CIN1, CIN2). The loops formed by the input capacitors
GROUND PLANE
ON LAYER 2
C1
R1
RPG
V
R2
CVCC
CSS
V
V
1
V
16
17
20
V
V
22
CIN1
6
21
7
RT
10
11
CIN2
CIN3
CBST
L
GROUND VIA
VIN VIA
VOUT VIA
COUT
V OTHER SIGNAL VIAS
8641 F01
Figure 1. Recommended PCB Layout for the LT8641
Rev A
For more information www.analog.com
13
LT8641
APPLICATIONS INFORMATION
Achieving Ultralow Quiescent Current
Burst Frequency
As the output load decreases, the frequency of single current pulses decreases (see Figure 2a) and the percentage
of time the LT8641 is in sleep mode increases, resulting in
much higher light load efficiency than for typical converters. By maximizing the time between pulses, the converter
quiescent current approaches 2.5µA for a typical application
when there is no output load. Therefore, to optimize the
quiescent current performance at light loads, the current
in the feedback resistor divider must be minimized as it
appears to the output as load current.
In order to achieve higher light load efficiency, more energy
must be delivered to the output during the single small
pulses in Burst Mode operation such that the LT8641 can
stay in sleep mode longer between each pulse. This can be
achieved by using a larger value inductor (i.e., 4.7µH), and
should be considered independent of switching frequency
when choosing an inductor. For example, while a lower
inductor value would typically be used for a high switching frequency application, if high light load efficiency is
desired, a higher inductor value should be chosen. See
curve in Typical Performance Characteristics.
While in Burst Mode operation the current limit of the top
switch is approximately 950mA (as shown in Figure 3),
resulting in low output voltage ripple. Increasing the output
capacitance will decrease output ripple proportionally. As
load ramps upward from zero the switching frequency
will increase but only up to the switching frequency programmed by the resistor at the RT pin as shown in Figure
2a. The output load at which the LT8641 reaches the
programmed frequency varies based on input voltage,
output voltage, and inductor choice.
SWITCHING FREQUENCY (kHz)
FRONT PAGE APPLICATION
VIN = 12V
VOUT = 5V
1000
800
600
400
200
0
0
200
400
600
LOAD CURRENT (mA)
800
8641 F02a
(2a)
Minimum Load to Full Frequency
(Pulse-Skipping Mode)
140
FRONT PAGE APPLICATION
VOUT = 5V
fSW = 1MHz
120
LOAD CURRENT (mA)
To enhance efficiency at light loads, the LT8641 operates
in low ripple Burst Mode operation, which keeps the output capacitor charged to the desired output voltage while
minimizing the input quiescent current and minimizing
output voltage ripple. In Burst Mode operation the LT8641
delivers single small pulses of current to the output capacitor followed by sleep periods where the output power is
supplied by the output capacitor. While in sleep mode the
LT8641 consumes 1.7μA.
1200
100
80
60
40
20
0
5
15
25
35
45
INPUT VOLTAGE (V)
55
65
8641 F02b
(2b)
Figure 2. SW Frequency vs Load Information in Burst Mode
Operation (2a) and Pulse-Skipping Mode (2b)
Switching Waveforms, Burst
Mode Operation
IL
500mA/DIV
VSW
5V/DIV
10µs/DIV
8641 F03
FRONT PAGE APPLICATION
12VIN TO 5VOUT AT 10mA
VSYNC = 0V
Figure 3. Burst Mode Operation
14
Rev A
For more information www.analog.com
LT8641
APPLICATIONS INFORMATION
For some applications it is desirable for the LT8641 to operate in pulse-skipping mode, offering two major differences
from Burst Mode operation. First is the clock stays awake
at all times and all switching cycles are aligned to the clock.
In this mode much of the internal circuitry is awake at all
times, increasing quiescent current to several hundred µA.
Second is that full switching frequency is reached at lower
output load than in Burst Mode operation (see Figure 2b).
To enable pulse-skipping mode, float the SYNC/MODE pin.
When a clock is applied to the SYNC/MODE pin the LT8641
will also operate in pulse-skipping mode.
FB Resistor Network
The output voltage is programmed with a resistor divider
between the output and the FB pin. Choose the resistor
values according to:
⎛ V
⎞
R1= R2 ⎜ OUT – 1⎟
⎝ 0.81V
⎠
(1)
Reference designators refer to the Block Diagram. 1%
resistors are recommended to maintain output voltage
accuracy.
When using large FB resistors, a 4.7pF to 22pF phase-lead
capacitor should be connected from VOUT to FB.
Setting the Switching Frequency
The LT8641 uses a constant frequency PWM architecture
that can be programmed to switch from 200kHz to 3MHz
by using a resistor tied from the RT pin to ground. A table
showing the necessarºy RT value for a desired switching
frequency is in Table 1.
The RT resistor required for a desired switching frequency
can be calculated using:
RT =
46.5
fSW
Table 1. SW Frequency vs RT Value
fSW (MHz)
RT (kΩ)
0.2
232
0.3
150
0.4
110
0.5
88.7
0.6
71.5
0.7
60.4
0.8
52.3
1.0
41.2
1.2
33.2
1.4
28.0
1.6
23.7
1.8
20.5
2.0
18.2
2.2
15.8
3.0
10.7
(2)
where 1.7µA is the quiescent current of the LT8641 and
the second term is the current in the feedback divider
reflected to the input of the buck operating at its light
load efficiency n. For a 3.3V application with R1 = 1M and
R2 = 324k, the feedback divider draws 2.5µA. With VIN =
12V and n = 85%, this adds 0.8µA to the 1.7µA quiescent
current resulting in 2.5µA no-load current from the 12V
supply. Note that this equation implies that the no-load
current is a function of VIN; this is plotted in the Typical
Performance Characteristics section.
(3)
where RT is in kΩ and fSW is the desired switching frequency in MHz.
If low input quiescent current and good light-load efficiency
are desired, use large resistor values for the FB resistor
divider. The current flowing in the divider acts as a load
current, and will increase the no-load input current to the
converter, which is approximately:
⎛ V
⎞ ⎛ V ⎞ ⎛ 1⎞
IQ = 1.7µA + ⎜ OUT ⎟ ⎜ OUT ⎟ ⎜ ⎟
⎝ R1+ R2 ⎠ ⎝ VIN ⎠ ⎝ n ⎠
– 5.2
Operating Frequency Selection and Trade-Offs
Selection of the operating frequency is a trade-off between
efficiency, component size, and input voltage range. The
advantage of high frequency operation is that smaller inductor and capacitor values may be used. The disadvantages
are lower efficiency and a smaller input voltage range.
Rev A
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15
LT8641
APPLICATIONS INFORMATION
The highest switching frequency (fSW(MAX)) for a given
application can be calculated as follows:
fSW (M AX ) =
VOUT + VSW (BOT )
t ON(M IN ) ( VIN – VSW ( TOP ) + VSW (BOT ) )
(4)
where VIN is the typical input voltage, VOUT is the output
voltage, VSW(TOP) and VSW(BOT) are the internal switch
drops (~0.3V, ~0.15V, respectively at maximum load)
and tON(MIN) is the minimum top switch on-time (see the
Electrical Characteristics). This equation shows that a
slower switching frequency is necessary to accommodate
a high VIN/VOUT ratio.
For transient operation, VIN may go as high as the absolute maximum rating of 65V regardless of the RT value,
however the LT8641 will reduce switching frequency as
necessary to maintain control of inductor current to assure safe operation.
The LT8641 is capable of a maximum duty cycle of approximately 99%, and the VIN-to-VOUT dropout is limited
by the RDS(ON) of the top switch. In this mode the LT8641
skips switch cycles, resulting in a lower switching frequency
than programmed by RT.
For applications that cannot allow deviation from the programmed switching frequency at low VIN/VOUT ratios use
the following formula to set switching frequency:
VIN(M IN ) =
VOUT + VSW (BOT )
1– fSW • t OFF (M IN )
– VSW (BOT ) + VSW ( TOP )
(5)
where VIN(MIN) is the minimum input voltage without
skipped cycles, VOUT is the output voltage, VSW(TOP) and
VSW(BOT) are the internal switch drops (~0.3V, ~0.15V,
respectively at maximum load), fSW is the switching frequency (set by RT), and tOFF(MIN) is the minimum switch
off-time. Note that higher switching frequency will increase
the minimum input voltage below which cycles will be
dropped to achieve higher duty cycle.
16
Inductor Selection and Maximum Output Current
The LT8641 is designed to minimize solution size by
allowing the inductor to be chosen based on the output
load requirements of the application. During overload or
short-circuit conditions the LT8641 safely tolerates operation with a saturated inductor through the use of a high
speed peak-current mode architecture.
A good first choice for the inductor value is:
L=
VOUT + VSW (BOT )
fSW
(6)
where fSW is the switching frequency in MHz, VOUT is
the output voltage, VSW(BOT) is the bottom switch drop
(~0.15V) and L is the inductor value in μH.
To avoid overheating and poor efficiency, an inductor must
be chosen with an RMS current rating that is greater than
the maximum expected output load of the application. In
addition, the saturation current (typically labeled ISAT)
rating of the inductor must be higher than the load current
plus 1/2 of in inductor ripple current:
1
IL (PEAK ) = ILOAD(M AX ) ΔIL
2
(7)
where ∆IL is the inductor ripple current as calculated in
Equation 9 and ILOAD(MAX) is the maximum output load
for a given application.
As a quick example, an application requiring 2A output
should use an inductor with an RMS rating of greater than
2A and an ISAT of greater than 3A. During long duration
overload or short-circuit conditions, the inductor RMS
rating requirement is greater to avoid overheating of the
inductor. To keep the efficiency high, the series resistance
(DCR) should be less than 0.04Ω, and the core material
should be intended for high frequency applications.
Rev A
For more information www.analog.com
LT8641
APPLICATIONS INFORMATION
(8)
inductor provides a higher maximum load current and
reduces the output voltage ripple. For applications requiring smaller load currents, the value of the inductor may
be lower and the LT8641 may operate with higher ripple
current. This allows use of a physically smaller inductor,
or one with a lower DCR resulting in higher efficiency. Be
aware that low inductance may result in discontinuous
mode operation, which further reduces maximum load
current.
The peak-to-peak ripple current in the inductor can be
calculated as follows:
For more information about maximum output current
and discontinuous operation, see Linear Technology’s
Application Note 44.
The LT8641 limits the peak switch current in order to protect
the switches and the system from overload faults. The top
switch current limit (ILIM) is 8.2A at low duty cycles and
decreases linearly to 6.4A at DC = 0.8. The inductor value
must then be sufficient to supply the desired maximum
output current (IOUT(MAX)), which is a function of the switch
current limit (ILIM) and the ripple current.
IOUT (M AX ) = ILIM –
ΔIL =
VOUT
L • fSW
ΔIL
2
⎛
V
• ⎜⎜ 1– OUT
⎝ VIN(M AX )
⎞
⎟⎟
⎠
9)
where fSW is the switching frequency of the LT8641, and
L is the value of the inductor. Therefore, the maximum
output current that the LT8641 will deliver depends on
the switch current limit, the inductor value, and the input
and output voltages. The inductor value may have to be
increased if the inductor ripple current does not allow
sufficient maximum output current (IOUT(MAX)) given the
switching frequency, and maximum input voltage used in
the desired application.
In order to achieve higher light load efficiency, more energy
must be delivered to the output during the single small
pulses in Burst Mode operation such that the LT8641 can
stay in sleep mode longer between each pulse. This can be
achieved by using a larger value inductor (i.e., 4.7µH), and
should be considered independent of switching frequency
when choosing an inductor. For example, while a lower
inductor value would typically be used for a high switching frequency application, if high light load efficiency is
desired, a higher inductor value should be chosen. See
curve in Typical Performance Characteristics.
The optimum inductor for a given application may differ
from the one indicated by this design guide. A larger value
Finally, for duty cycles greater than 50% (VOUT/VIN > 0.5),
a minimum inductance is required to avoid sub-harmonic
oscillation. See Application Note 19.
Input Capacitors
The VIN of the LT8641 should be bypassed with at least
three ceramic capacitors for best performance. Two small
ceramic capacitors of 1µF should be placed close to the
part; one at the VIN1/GND1 pins and a second at VIN2/GND2
pins. These capacitors should be 0402 or 0603 in size. For
automotive applications requiring 2 series input capacitors, two small 0402 or 0603 may be placed at each side
of the LT8641 near the VIN1/GND1 and VIN2/GND2 pins.
A third, larger ceramic capacitor of 2.2µF or larger should
be placed close to VIN1 or VIN2. See Low EMI PCB Layout
section for more detail. X7R or X5R capacitors are recommended for best performance across temperature and
input voltage variations.
Note that larger input capacitance is required when a lower
switching frequency is used. If the input power source has
high impedance, or there is significant inductance due to
long wires or cables, additional bulk capacitance may be
necessary. This can be provided with a low performance
electrolytic capacitor.
Rev A
For more information www.analog.com
17
LT8641
APPLICATIONS INFORMATION
A ceramic input capacitor combined with trace or cable
inductance forms a high quality (under damped) tank circuit. If the LT8641 circuit is plugged into a live supply, the
input voltage can ring to twice its nominal value, possibly
exceeding the LT8641’s voltage rating. This situation is
easily avoided (see Linear Technology Application Note 88).
Output Capacitor and Output Ripple
The output capacitor has two essential functions. Along
with the inductor, it filters the square wave generated
by the LT8641 to produce the DC output. In this role it
determines the output ripple, thus low impedance at the
switching frequency is important. The second function
is to store energy in order to satisfy transient loads and
stabilize the LT8641’s control loop. Ceramic capacitors
have very low equivalent series resistance (ESR) and
provide the best ripple performance. For good starting
values, see the Typical Applications section.
Use X5R or X7R types. This choice will provide low output
ripple and good transient response. Transient performance
can be improved with a higher value output capacitor and
the addition of a feedforward capacitor placed between
VOUT and FB. Increasing the output capacitance will also
decrease the output voltage ripple. A lower value of output
capacitor can be used to save space and cost but transient
performance will suffer and may cause loop instability. See
the Typical Applications in this data sheet for suggested
capacitor values.
When choosing a capacitor, special attention should be
given to the data sheet to calculate the effective capacitance
under the relevant operating conditions of voltage bias and
temperature. A physically larger capacitor or one with a
higher voltage rating may be required.
Ceramic Capacitors
Ceramic capacitors are small, robust and have very low
ESR. However, ceramic capacitors can cause problems
when used with the LT8641 due to their piezoelectric nature.
18
When in Burst Mode operation, the LT8641’s switching
frequency depends on the load current, and at very light
loads the LT8641 can excite the ceramic capacitor at audio
frequencies, generating audible noise. Since the LT8641
operates at a lower current limit during Burst Mode operation, the noise is typically very quiet to a casual ear. If
this is unacceptable, use a high performance tantalum or
electrolytic capacitor at the output. Low noise ceramic
capacitors are also available.
A final precaution regarding ceramic capacitors concerns
the maximum input voltage rating of the LT8641. As
previously mentioned, a ceramic input capacitor combined
with trace or cable inductance forms a high quality (underdamped) tank circuit. If the LT8641 circuit is plugged
into a live supply, the input voltage can ring to twice its
nominal value, possibly exceeding the LT8641’s rating.
This situation is easily avoided (see Linear Technology
Application Note 88).
Enable Pin
The LT8641 is in shutdown when the EN pin is low and
active when the pin is high. The rising threshold of the EN
comparator is 1.01V, with 45mV of hysteresis. The EN pin
can be tied to VIN if the shutdown feature is not used, or
tied to a logic level if shutdown control is required.
Adding a resistor divider from VIN to EN programs the
LT8641 to regulate the output only when VIN is above a
desired voltage (see the Block Diagram). Typically, this
threshold, VIN(EN), is used in situations where the input
supply is current limited, or has a relatively high source
resistance. A switching regulator draws constant power
from the source, so source current increases as source
voltage drops. This looks like a negative resistance load
to the source and can cause the source to current limit or
latch low under low source voltage conditions. The VIN(EN)
threshold prevents the regulator from operating at source
voltages where the problems might occur. This threshold
Rev A
For more information www.analog.com
LT8641
APPLICATIONS INFORMATION
can be adjusted by setting the values R3 and R4 such that
they satisfy the following equation:
⎛ R3 ⎞
VIN(EN ) = ⎜
+ 1⎟ • 1.01V
⎝ R4 ⎠
(10)
where the LT8641 will remain off until VIN is above VIN(EN).
Due to the comparator’s hysteresis, switching will not stop
until the input falls slightly below VIN(EN).
When operating in Burst Mode operation for light load
currents, the current through the VIN(EN) resistor network
can easily be greater than the supply current consumed
by the LT8641. Therefore, the VIN(EN) resistors should be
large to minimize their effect on efficiency at low loads.
INTVCC Regulator
An internal low dropout (LDO) regulator produces the 3.4V
supply from VIN that powers the drivers and the internal
bias circuitry. The INTVCC can supply enough current for
the LT8641’s circuitry and must be bypassed to ground
with a minimum of 1μF ceramic capacitor. Good bypassing
is necessary to supply the high transient currents required
by the power MOSFET gate drivers. To improve efficiency
the internal LDO can also draw current from the BIAS
pin when the BIAS pin is at 3.1V or higher. Typically the
BIAS pin can be tied to the output of the LT8641, or can
be tied to an external supply of 3.3V or above. If BIAS is
connected to a supply other than VOUT, be sure to bypass
with a local ceramic capacitor. If the BIAS pin is below
3.0V, the internal LDO will consume current from VIN.
Applications with high input voltage and high switching
frequency where the internal LDO pulls current from VIN
will increase die temperature because of the higher power
dissipation across the LDO. Do not connect an external
load to the INTVCC pin.
Output Voltage Tracking and Soft-Start
The LT8641 allows the user to program its output voltage
ramp rate by means of the TR/SS pin. An internal 1.9μA
pulls up the TR/SS pin to INTVCC. Putting an external
capacitor on TR/SS enables soft starting the output to prevent current surge on the input supply. During the soft-start
ramp the output voltage will proportionally track the TR/SS
pin voltage. For output tracking applications, TR/SS can
be externally driven by another voltage source. From 0V
to 0.8V, the TR/SS voltage will override the internal 0.8V
reference input to the error amplifier, thus regulating the
FB pin voltage to that of TR/SS pin. When TR/SS is above
0.8V, tracking is disabled and the feedback voltage will
regulate to the internal reference voltage. The TR/SS pin
may be left floating if the function is not needed.
An active pull-down circuit is connected to the TR/SS pin
which will discharge the external soft-start capacitor in
the case of fault conditions and restart the ramp when the
faults are cleared. Fault conditions that clear the soft-start
capacitor are the EN/UV pin transitioning low, VIN voltage
falling too low, or thermal shutdown.
Output Power Good
When the LT8641’s output voltage is within the ±8%
window of the regulation point, the output voltage is
considered good and the open-drain PG pin goes high
impedance and is typically pulled high with an external
resistor. Otherwise, the internal pull-down device will pull
the PG pin low. To prevent glitching both the upper and
lower thresholds include 0.4% of hysteresis.
The PG pin is also actively pulled low during several fault
conditions: EN/UV pin is below 1V, INTVCC has fallen too
low, VIN is too low, or thermal shutdown.
Synchronization and Spread Spectrum
To select low ripple Burst Mode operation, tie the SYNC pin
below 0.4V (this can be ground or a logic low output). To
synchronize the LT8641 oscillator to an external frequency
connect a square wave (with 20% to 80% duty cycle) to
the SYNC pin. The square wave amplitude should have valleys that are below 0.4V and peaks above 1.5V (up to 6V).
Rev A
For more information www.analog.com
19
LT8641
APPLICATIONS INFORMATION
The LT8641 will not enter Burst Mode operation at low
output loads while synchronized to an external clock, but
instead will pulse skip to maintain regulation. The LT8641
may be synchronized over a 200kHz to 3MHz range. The
RT resistor should be chosen to set the LT8641 switching
frequency equal to or below the lowest synchronization
input. For example, if the synchronization signal will be
500kHz and higher, the RT should be selected for 500kHz.
The slope compensation is set by the RT value, while the
minimum slope compensation required to avoid subharmonic oscillations is established by the inductor size,
input voltage, and output voltage. Since the synchronization frequency will not change the slopes of the inductor
current waveform, if the inductor is large enough to avoid
subharmonic oscillations at the frequency set by RT, then
the slope compensation will be sufficient for all synchronization frequencies.
For some applications it is desirable for the LT8641 to
operate in pulse-skipping mode, offering two major differences from Burst Mode operation. First is the clock stays
awake at all times and all switching cycles are aligned
to the clock. Second is that full switching frequency is
reached at lower output load than in Burst Mode operation.
These two differences come at the expense of increased
quiescent current. To enable pulse-skipping mode, the
SYNC pin is floated. Leakage current on this pin should
be <1µA. See Block Diagram for internal pull-up and pulldown resistance.
The LT8641 features spread spectrum operation to further
reduce EMI emissions. To enable spread spectrum operation, the SYNC/MODE pin should be tied high either to
INTVCC (~3.4V) or an external supply of 3V to 4V. In this
mode, triangular frequency modulation is used to vary the
switching frequency between the value programmed by RT
to approximately 20% higher than that value. The modulation frequency is approximately 3kHz. For example, when
the LT8641 is programmed to 2MHz, the frequency will
vary from 2MHz to 2.4MHz at a 3kHz rate. When spread
20
spectrum operation is selected, Burst Mode operation is
disabled, and the part will run in pulse-skipping mode.
The LT8641 does not operate in forced continuous mode
regardless of SYNC signal.
Shorted and Reversed Input Protection
The LT8641 will tolerate a shorted output. Several features
are used for protection during output short-circuit and
brownout conditions. The first is the switching frequency
will be folded back while the output is lower than the set
point to maintain inductor current control. Second, the
bottom switch current is monitored such that if inductor
current is beyond safe levels switching of the top switch
will be delayed until such time as the inductor current
falls to safe levels.
Frequency foldback behavior depends on the state of the
SYNC pin: If the SYNC pin is low the switching frequency
will slow while the output voltage is lower than the programmed level. If the SYNC pin is connected to a clock
source, floated, or tied high, the LT8641 will stay at the
programmed frequency without foldback and only slow
switching if the inductor current exceeds safe levels.
There is another situation to consider in systems where the
output will be held high when the input to the LT8641 is
absent. This may occur in battery charging applications or
in battery-backup systems where a battery or some other
supply is diode ORed with the LT8641’s output. If the VIN
pin is allowed to float and the EN pin is held high (either by
a logic signal or because it is tied to VIN), then the LT8641’s
internal circuitry will pull its quiescent current through its
SW pin. This is acceptable if the system can tolerate several
μA in this state. If the EN pin is grounded the SW pin current
will drop to near 1µA. However, if the VIN pin is grounded
while the output is held high, regardless of EN, parasitic
body diodes inside the LT8641 can pull current from the
output through the SW pin and the VIN pin, which may
damage the IC. Figure 4 shows a connection of the VIN and
Rev A
For more information www.analog.com
LT8641
APPLICATIONS INFORMATION
Case Temperature Rise
D1
DC2373A DEMO BOARD
VIN = 12V, fSW = 1MHz
VIN = 24V, fSW = 1MHz
VIN = 12V, fSW = 2MHz
VIN = 24V, fSW = 2MHz
VIN
LT8641
EN/UV
GND
8641 F04
Figure 4. Reverse VIN Protection
EN/UV pins that will allow the LT8641 to run only when
the input voltage is present and that protects against a
shorted or reversed input.
CASE TEMPERATURE RISE (°C)
VIN
60
50
40
30
20
10
0
Temperature rise of the LT8641 is worst when operating
at high load, high VIN, and high switching frequency. If
the case temperature is too high for a given application,
then either VIN, switching frequency, or load current can
be decreased to reduce the temperature to an acceptable
level. Figure 5 shows examples of how case temperature
rise can be managed by reducing VIN, switching frequency,
or load.
1
1.5
2
2.5
LOAD CURRENT (A)
3
3.5
Figure 5. Case Temperature Rise
The LT8641’s internal power switches are capable of safely
delivering up to 5A of peak output current. However, due
to thermal limits, the package can only handle 5A loads
for short periods of time. This time is determined by how
quickly the case temperature approaches the maximum
junction rating. Figure 6 shows an example of how case
temperature rise changes with the duty cycle of a 1kHz
pulsed 5A load.
The LT8641’s top switch current limit decreases with
higher duty cycle operation for slope compensation. This
also limits the peak output current the LT8641 can deliver
for a given application. See curve in Typical Performance
Characteristics.
Pulsed Load
90
DC2373A DEMO BOARD
VIN = 12V
VOUT = 5V
fSW = 2MHz
STANDBY LOAD = 0.25A
1kHz PULSED LOAD = 5A
80
CASE TEMPERATURE RISE (°C)
The internal overtemperature protection monitors the
junction temperature of the LT8641. If the junction temperature reaches approximately 160°C, the LT8641 will
stop switching and indicate a fault condition until the
temperature drops about 1°C cooler.
0.5
8641 F05
Thermal Considerations and Peak Output Current
For higher ambient temperatures, care should be taken in
the layout of the PCB to ensure good heat sinking of the
LT8641. The ground pins on the bottom of the package
should be soldered to a ground plane. This ground should
be tied to large copper layers below with thermal vias;
these layers will spread heat dissipated by the LT8641.
Placing additional vias can reduce thermal resistance
further. The maximum load current should be derated
as the ambient temperature approaches the maximum
junction rating. Power dissipation within the LT8641 can
be estimated by calculating the total power loss from
an efficiency measurement and subtracting the inductor
loss. The die temperature is calculated by multiplying the
LT8641 power dissipation by the thermal resistance from
junction to ambient.
0
70
60
50
40
30
20
10
0
0
0.2
0.4
0.6
DUTY CYCLE OF 5A LOAD
0.8
8641 F06
Figure 6. Case Temperature Rise vs 5A Pulsed Load
Rev A
For more information www.analog.com
21
LT8641
TYPICAL APPLICATIONS
5V 3.5A Step-Down Converter
VIN
5.5V TO 65V
4.7µF
EN/UV
VIN1
1µF
0603
GND1
PG
10nF
VIN2
GND2
LT8641
SYNC/MODE
TR/SS
BST
0.1µF 4.7µH
VOUT
5V
3.5A
SW
BIAS
1µF
4.7pF
1M
FB
INTVCC
41.2k
1µF
0603
RT
191k
GND
fSW = 1MHz
L: VISHAY IHLP2525EZ-01
47µF
1210
X5R/X7R
8641 TA08
3.3V, 3.5A Step-Down Converter
VIN
3.8V TO 65V
4.7µF
EN/UV
VIN1
1µF
0603
GND1
PG
10nF
VIN2
GND2
LT8641
SYNC/MODE
TR/SS
BST
0.1µF 2.2µH
VOUT
3.3V
3.5A
SW
BIAS
1µF
4.7pF
1M
FB
INTVCC
41.2k
1µF
0603
RT
324k
GND
fSW = 1MHz
L: VISHAY IHLP2525EZ-01
47µF
1210
X5R/X7R
8641 TA05
Ultralow EMI 5V, 3.5A Step-Down Converter
VIN
5.5V TO 65V
FB1
BEAD
L2
0.22µH
4.7µF
1210
4.7µF
1210
4.7µF
1206
1µF
0603
VIN1
EN/UV
GND1
PG
10nF
VIN2
GND2
LT8641
SYNC/MODE
TR/SS
BST
L
0.1µF 2.2µH
4.7pF
INTVCC
RT
1M
FB
GND
fSW = 2MHz
FB1 BEAD: MPZ2012S300A
L: IHLP2525CZ-01
L2: IHLP1212BZ-11
22
VOUT
5V
3.5A
SW
BIAS
1µF
18.2k
1µF
0603
191k
47µF
1210
X5R/X7R
8641 TA02
Rev A
For more information www.analog.com
LT8641
TYPICAL APPLICATIONS
2MHz 5V, 3.5A Step-Down Converter
VIN
5.5V TO 65V
4.7µF
VIN1
1µF
0603
EN/UV
GND1
PG
10nF
VIN2
GND2
LT8641
SYNC/MODE
TR/SS
BST
0.1µF 2.2µH
VOUT
5V
3.5A
SW
BIAS
1µF
4.7pF
1M
FB
INTVCC
18.2k
1µF
0603
RT
191k
GND
fSW = 2MHz
L: VISHAY IHLP2525CZ-01
47µF
1210
X5R/X7R
8641 TA03
2MHz 3.3V, 3.5A Step-Down Converter
VIN
3.8V TO 65V
4.7µF
VIN1
1µF
0603
EN/UV
GND1
PG
10nF
VIN2
GND2
LT8641
SYNC/MODE
TR/SS
BST
0.1µF 1.5µH
VOUT
3.3V
3.5A
SW
BIAS
1µF
4.7pF
1M
FB
INTVCC
18.2k
1µF
0603
RT
324k
GND
fSW = 2MHz
L: VISHAY IHLP2525CZ-01
47µF
1210
X5R/X7R
8641 TA06
12V, 3.5A Step-Down Converter
VIN
12.5V TO 65V
4.7µF
VIN1
1µF
0603
EN/UV
GND1
PG
10nF
VIN2
GND2
LT8641
SYNC/MODE
TR/SS
BST
0.1µF 6.8µH
4.7pF
INTVCC
RT
VOUT
12V
3.5A
SW
BIAS
1µF
41.2k
1µF
0603
1M
FB
GND
fSW = 1MHz
L: VISHAY IHLP2525EZ-01
71.5k
47µF
1210
X5R/X7R
8641 TA04
Rev A
For more information www.analog.com
23
LT8641
PACKAGE DESCRIPTION
Please refer to http://www.linear.com/product/LT8641#packaging for the most recent package drawings.
UDC Package
Variation: UDC20(18)
20(18)-Lead Plastic QFN (3mm × 4mm)
(Reference LTC DWG # 05-08-1956 Rev C)
Exposed Pad Variation AA
0.055 BSC
0.70 ±0.05
3.50 ±0.05
2.10 ±0.05
1.50 REF
0.770
BSC
0.220 ±0.05
0.356 ±0.05
0.400 ±0.05
PACKAGE OUTLINE
0.25 ±0.05
0.50 BSC
2.50 REF
3.10 ±0.05
4.50 ±0.05
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED
3.00 ±0.10
0.75 ±0.05
1.50 REF
PIN 1 ID
0.12 × 45°
0.40 ±0.10
1
PIN 1
TOP MARK
(NOTE 5)
4.00 ±0.10
0.220 ±0.05
2.127 ±0.10
2
2.50 REF
0.770
BSC
0.356 ±0.05
0.400 ±0.05
(UDC20(18)) QFN 1116 REV C
0.200 REF
0.00 – 0.05
R = 0.110
TYP
0.25 ±0.05
0.50 BSC
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING IS NOT A JEDEC PACKAGE OUTLINE
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
24
Rev A
For more information www.analog.com
LT8641
REVISION HISTORY
REV
DATE
DESCRIPTION
A
05/18
Clarified benefits
Clarified Power Loss scale
Added LT8641HUDC
Added new Note 3, Note 3 became Note 4
Clarified SYNC/MODE (Pin 17) description
Clarified Block Diagram
Clarified Applications Synchronization and Spread Spectrum section
Clarified Figure 4
Clarified Applications section
Clarified output capacitor in Typical Applications
Clarified Package page moves to page 24
PAGE NUMBER
1
1
2
3
11
11
20
21
21
22, 23
24
Rev A
Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog
Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications
subject to change without notice. No license For
is granted
implication or
otherwise under any patent or patent rights of Analog Devices.
more by
information
www.analog.com
25
LT8641
TYPICAL APPLICATIONS
2MHz 1.8V, 3.5A Step-Down Converter
VIN
3V TO 22V
(65V TRANSIENT)
4.7µF
VIN1
1µF
0603
EN/UV
GND1
PG
10nF
VIN2
GND2
LT8641
SYNC/MODE
TR/SS
BST
SW
BIAS
1µF
18.2k
INTVCC
RT
FB
1µF
0603
1µH
0.1µF
EXTERNAL
1µF SOURCE >3.1V
OR GND
VOUT
1.8V
3.5A
10pF
1M
825k
GND
fSW = 2MHz
L: VISHAY IHLP2525CZ-01
100µF
1210
X5R/X7R
8641 TA07
RELATED PARTS
PART
DESCRIPTION
COMMENTS
LT8640/
LT8640-1
42V, 5A, 96% Efficiency, 3MHz Synchronous MicroPower Step-Down
DC/DC Converter with IQ = 2.5µA
VIN(MIN) = 3.4V, VIN(MAX) = 42V, VOUT(MIN) = 0.97V, IQ = 2.5μA,
ISD < 1μA, 3mm × 4mm QFN-18
LT8609/
LT8609A
42V, 2A, 94% Efficiency, 2.2MHz Synchronous MicroPower Step-Down
DC/DC Converter with IQ = 2.5µA
VIN(MIN) = 3V, VIN(MAX) = 42V, VOUT(MIN) = 0.8V, IQ = 2.5μA,
ISD < 1μA, MSOP-10E
LT8610A/
LT8610AB
42V, 3.5A, 96% Efficiency, 2.2MHz Synchronous MicroPower Step-Down
DC/DC Converter with IQ = 2.5µA
VIN(MIN) = 3.4V, VIN(MAX) = 42V, VOUT(MIN) = 0.97V, IQ = 2.5μA,
ISD < 1μA, MSOP-16E
LT8610AC
42V, 3.5A, 96% Efficiency, 2.2MHz Synchronous MicroPower Step-Down
DC/DC Converter with IQ = 2.5µA
VIN(MIN) = 3V, VIN(MAX) = 42V, VOUT(MIN) = 0.8V, IQ = 2.5μA,
ISD < 1μA, MSOP-16E
LT8610
42V, 2.5A, 96% Efficiency, 2.2MHz Synchronous MicroPower Step-Down
DC/DC Converter with IQ = 2.5µA
VIN(MIN) = 3.4V, VIN(MAX) = 42V, VOUT(MIN) = 0.97V, IQ = 2.5μA,
ISD < 1μA, MSOP-16E
LT8611
42V, 2.5A, 96% Efficiency, 2.2MHz Synchronous MicroPower Step-Down
DC/DC Converter with IQ = 2.5µA and Input/Output Current
Limit/Monitor
VIN(MIN) = 3.4V, VIN(MAX) = 42V, VOUT(MIN) = 0.97V, IQ = 2.5μA,
ISD < 1μA, 3mm × 5mm QFN-24
LT8616
42V, Dual 2.5A + 1.5A, 95% Efficiency, 2.2MHz Synchronous MicroPower VIN(MIN) = 3.4V, VIN(MAX) = 42V, VOUT(MIN) = 0.8V, IQ = 5μA,
Step-Down DC/DC Converter with IQ = 5µA
ISD < 1μA, TSSOP-28E, 3mm × 6mm QFN-28
LT8620
65V, 2.5A, 94% Efficiency, 2.2MHz Synchronous MicroPower Step-Down
DC/DC Converter with IQ = 2.5µA
LT8614
42V, 4A, 96% Efficiency, 2.2MHz Synchronous Silent Switcher Step-Down VIN(MIN) = 3.4V, VIN(MAX) = 42V, VOUT(MIN) = 0.97V, IQ = 2.5μA,
DC/DC Converter with IQ = 2.5µA
ISD < 1μA, 3mm × 4mm QFN18
LT8612
42V, 6A, 96% Efficiency, 2.2MHz Synchronous MicroPower Step-Down
DC/DC Converter with IQ = 2.5µA
VIN(MIN) = 3.4V, VIN(MAX) = 42V, VOUT(MIN) = 0.97V, IQ = 3.0μA,
ISD < 1μA, 3mm × 6mm QFN-28
LT8613
42V, 6A, 96% Efficiency, 2.2MHz Synchronous MicroPower Step-Down
DC/DC Converter with Current Limiting
VIN(MIN) = 3.4V, VIN(MAX) = 42V, VOUT(MIN) = 0.97V, IQ = 3.0μA,
ISD < 1μA, 3mm × 6mm QFN-28
LT8602
42V, Quad Output (2.5A + 1.5A + 1.5A + 1.5A) 95% Efficiency, 2.2MHz
Synchronous MicroPower Step-Down DC/DC Converter with IQ = 25µA
VIN(MIN) = 3V, VIN(MAX) = 42V, VOUT(MIN) = 0.8V, IQ = 2.5μA,
ISD < 1μA, 6mm × 6mm QFN-40
26
VIN(MIN) = 3.4V, VIN(MAX) = 65V, VOUT(MIN) = 0.97V, IQ = 2.5μA,
ISD < 1μA, MSOP-16E, 3mm × 5mm QFN-24
Rev A
D16866-0-5/18(A)
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