AD ADP1110AR-5 Micropower, step-up/step-down switching regulator; adjustable and fixed 3.3 v, 5 v, 12 v Datasheet

a
Micropower, Step-Up/Step-Down Switching
Regulator; Adjustable and Fixed 3.3 V, 5 V, 12 V
ADP1110
FEATURES
Operates at Supply Voltages From 1.0 V to 30 V
Step-Up or Step-Down Mode
Minimal External Components Required
Low-Battery Detector
User-Adjustable Current Limiting
Fixed or Adjustable Output Voltage Versions
8-Pin DIP or SO-8 Package
APPLICATIONS
Cellular Telephones
Single-Cell to 5 V Converters
Laptop and Palmtop Computers
Pagers
Cameras
Battery Backup Supplies
Portable Instruments
Laser Diode Drivers
Hand-Held Inventory Computers
FUNCTIONAL BLOCK DIAGRAMS
SET
ADP1110
A2
A0
VIN
GAIN BLOCK/
ERROR AMP
ILIM
SW1
220mV
REFERENCE
A1
OSCILLATOR
COMPARATOR
R1
Q1
DRIVER
R2
300kΩ
GND
SW2
SENSE
ADP1110 Block Diagram—Fixed Output Version
SET
ADP1110
A2
A0
VIN
GAIN BLOCK/
ERROR AMP
GENERAL DESCRIPTION
The ADP1110 is part of a family of step-up/step-down switching regulators that operate from an input voltage supply as little
as 1.0 V. This very low input voltage allows the ADP1110 to be
used in applications that use a single cell as the primary power
source.
The ADP1110 can be configured to operate in either step-up or
step-down mode, but for input voltages greater than 3 V, the
ADP1111 would be a more effective solution.
ILIM
SW1
220mV
REFERENCE
A1
OSCILLATOR
COMPARATOR
GND
FB
Q1
DRIVER
SW2
An auxiliary gain amplifier can serve as a low battery detector or
as a linear regulator.
ADP1110 Block Diagram—Adjustable Output Version
The 70 kHz frequency operation also allows for the use of
surface-mount external capacitors and inductors.
The quiescent current of 300 µA makes the ADP1110 useful in
remote or battery powered applications.
Battery protection circuitry limits the effect of reverse current to
safe levels at reverse voltages up to 1.6 V.
REV. 0
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 617/329-4700
World Wide Web Site: http://www.analog.com
Fax: 617/326-8703
© Analog Devices, Inc., 1996
ADP1110–SPECIFICATIONS (08C to +708C, V = 1.5 V unless otherwise noted)
IN
Parameter
Conditions
VS
QUIESCENT CURRENT
Switch Off
IQ
INPUT VOLTAGE
Step-Up Mode
Step-Down Mode
VIN
COMPARATOR TRIP POINT VOLTAGE
ADP11101
2
Min
Typ
Max
µA
300
1.15
Units
12.6
30
V
V
210
220
230
mV
3.13
4.75
11.4
3.30
5.00
12.00
3.47
5.25
12.6
V
V
V
OUTPUT SENSE VOLTAGE
ADP1110-3.3
ADP1110-52
ADP1110-122
COMPARATOR HYSTERESIS
ADP1110
4
8
mV
OUTPUT HYSTERESIS
ADP1110-3.3
ADP1110-5
ADP1110-12
66
90
200
130
180
400
mV
mV
mV
VOUT
OSCILLATOR FREQUENCY
DUTY CYCLE
Full Load (VFB < VREF)
SWITCH ON TIME
fOSC
52
70
90
kHz
DC
62
69
78
%
tON
7.5
10
12.5
µs
FEEDBACK PIN BIAS CURRENT
ADP1110 VFB = 0 V
IFB
150
240
nA
SET PIN BIAS CURRENT
VSET = VREF
ISET
300
500
nA
A0 OUTPUT LOW
IAO = 300 µA
VSET = 150 mV
VAO
0.15
0.4
V
REFERENCE LINE REGULATION
1.0 V ≤ VIN ≤ 1.5 V
1.5 V ≤ VIN ≤ 12 V
0.35
0.05
0.1
%/V
%/V
500
600
650
750
1000
mV
mV
mV
mV
mV
SWITCH SATURATION VOLTAGE
STEP-UP MODE
A2 ERROR AMP GAIN
VIN = 1.5 V, ISW = 400 mA, +25°C
TMIN to TMAX
VIN = 1.5 V, ISW = 500 mA, +25°C
TMIN to TMAX
VIN = 5 V, ISW = 1 A, +25°C
VCESAT
RL = 100 kΩ3
AV
4
300
400
700
IREV
1000
5000
V/V
750
mA
–0.3
%/°C
REVERSE BATTERY CURRENT
TA = +25°C
CURRENT LIMIT TEMPERATURE
COEFFICIENT
VIN, TA = +25°C
SWITCH OFF LEAKAGE CURRENT
Measured at SW1 Pin,
TA = +25°C
ILEAK
1
10
µA
MAXIMUM EXCURSION BELOW GND
ISW1 ≤ 10 µA, Switch Off
TA = +25°C
VSW2
–400
–350
mV
NOTES
1
This specification guarantees that both the high and low trip point of the comparator fall within the 210 mV to 230 mV range.
2
This specification guarantees that the output voltage of the fixed versions will always fall within the specified range. The waveform at the sense pin will exhibit a sawtooth shape due to the comparator hysteresis.
3
100 kΩ resistor connected between a 5 V source and the AO pin.
4
The ADP1110 is guaranteed to withstand continuous application of +1.6 V applied to the GND and SW2 pins while V IN, ILIM, and SW1 pins are grounded.
5
All limits at temperature extremes are guaranteed via correlation using standard statistical quality control methods.
Specifications subject to change without notice.
–2–
REV. 0
ADP1110
ABSOLUTE MAXIMUM RATINGS
PIN CONFIGURATIONS
Input Supply Voltage, Step-Up Mode . . . . . . . . . . . . . . . 15 V
Input Supply Voltage, Step-Down Mode . . . . . . . . . . . . . 36 V
SW1 Pin Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 50 V
SW2 Pin Voltage . . . . . . . . . . . . . . . . . . . . . . . . . –0.5 V to VIN
Feedback Pin Voltage (ADP1110) . . . . . . . . . . . . . . . . . . 5.5 V
Switch Current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.5 A
Maximum Power Dissipation . . . . . . . . . . . . . . . . . . 500 mW
Operating Temperature Range . . . . . . . . . . . . . 0°C to +70°C
Storage Temperature Range . . . . . . . . . . . . . –65°C to 150°C
Lead Temperature (Soldering, 10 sec) . . . . . . . . . . . . . 300°C
8-Lead Plastic DIP
(N-8)
ILIM 1
VIN
2
ADP1110
8 FB (SENSE)*
ILIM
1
7 SET
VIN
2
ADP1110
SW1 3
TOP VIEW
(Not to Scale)
TOP VIEW
SW1 3 (Not to Scale) 6
TJMAX =
A0
5 GND
SW2 4
90o,
θJA =
130oC/W
*FIXED VERSIONS
TYPICAL APPLICATION
8-Lead SOIC
(SO-8)
8 FB (SENSE)*
7 SET
6 A0
5 GND
SW2 4
TJMAX = 90o, θJA = 150oC/W
*FIXED VERSIONS
47µH
5V
PIN DESCRIPTION
1
ILIM
1.5V
AA CELL*
2
VIN
SW1 3
ADP1110-5
Mnemonic
Function
ILIM
For normal conditions this pin is connected to
VIN. When lower current is required, a resistor
should be connected between ILIM and VIN.
Limiting the switch current to 400 mA is
achieved by connecting a 220 Ω resistor.
Input Voltage.
Collector Node of Power Transistor. For stepdown configuration, connect to VIN. For stepup configuration, connect to an inductor/diode.
Emitter Node of Power Transistor. For stepdown configuration, connect to inductor/diode.
For step-up configuration, connect to ground.
Do not allow this pin to go more than a diode
drop below ground.
Ground.
Auxiliary Gain (GB) Output. The open collector can sink 300 µA. It can be left open if unused.
Gain Amplifier Input. The amplifier has positive input connected to SET pin and negative
input connected to 220 mV reference. It can be
left open if unused.
On the ADP1110 (adjustable) version this pin
is connected to the comparator input. On the
ADP1110-3.3, ADP1110-5 and ADP1110-12,
the pin goes directly to the internal application
resistor that set output voltage.
SENSE 8
GND
5
SW2
4
15µF
TANTALUM
VIN
SW1
OPERATES WITH CELL VOLTAGE ≥1.0V
*ADD 10µF DECOUPLING CAPACITOR IF BATTERY IS
*MORE THAN 2' AWAY FROM ADP1110.
Figure 1. 1.5 V to 5 V Converter
SW2
ORDERING GUIDE
Model
Output Voltage
Package
ADP1110AN
ADP1110AR
ADP1110AN-3.3
ADP1110AR-3.3
ADP1110AN-5
ADP1110AR-5
ADP1110AN-12
ADP1110AR-12
ADJ
ADJ
3.3 V
3.3 V
5V
5V
12 V
12 V
N-8
SO-8
N-8
SO-8
N-8
SO-8
N-8
SO-8
GND
AO
SET
FB/SENSE
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the ADP1110 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.
REV. 0
–3–
WARNING!
ESD SENSITIVE DEVICE
ADP1110-Typical Characteristics
76
74
OSCILLATOR FREQUENCY – kHz
SATURATION VOLTAGE – V
1.4
VIN = +2V
1.2
VIN = +1.5V
1
VIN = +1.2V
0.8
VIN = +3V
0.6
VIN = +1V
VIN = +5V
0.4
0.2
0.2
0.4
0.5
0.6
0.8
1
ISWITCH CURRENT – A
1.2
1.25
68
66
64
60
1.4
2
4
6
8
10
12 15
18
INPUT VOLTAGE – V
21
24
27
30
Figure 5. Oscillator Frequency vs. Input Voltage
Figure 2. Saturation Voltage vs. ISWITCH Current in Step-Up
Mode
2
1.9
1.8
1.7
1.6
1.5
SWITCH CURRENT – A
VIN = +12V
1.4
ON VOLTAGE – V
OSCILLATOR FREQUENCY
70
62
0
0.1
1.2
1
0.8
0.6
STEP-DOWN WITH
V = +12V
1.3
1.1
0.9
0.7
0.5
0.4
0.3
0.2
0.1
0
0.1
0.2
0.4
0.6
ISWITCH CURRENT – A
0.8
1
0.9
10
100
1000
RLIM – Ω
Figure 3. Switch ON Voltage vs. ISWITCH Current In StepDown Mode
Figure 6. Maximum Switch Current vs. RLIM
1.5
1800
1600
1.3
1400
SWITCH CURRENT – A
QUIESCENT CURRENT – µA
72
1200
QUIESCENT CURRENT
1000
800
600
1.1
STEP-UP MODE
WITH V ≤ +5V
0.9
0.7
0.5
400
0.3
200
0.1
0
1
3
6
9
12
15
18
21
24
27
1
30
INPUT VOLTAGE – V
Figure 4. Quiescent Current vs. Input Voltage
10
RLIM – Ω
100
1000
Figure 7. Maximum Switch Current vs. RLIM
–4–
REV. 0
78
0.54
76
0.53
74
0.52
72
VCE(SAT) – V
OSCILLATOR FREQUENCY – kHz
ADP1110
OSCILLATOR FREQUENCY
70
0.51
VIN = +1.5 @ ISWITCH = +0.5A
0.5
68
0.49
66
0.48
64
0
25
TEMPERATURE – 8C
0.47
70
Figure 8. Oscillator Frequency vs. Temperature
0
350
8.9
300
QUIESCENT CURRENT
QUIESCENT CURRENT – µA
ON TIME – µs
8.8
8.7
8.6
SWITCH ON TIME
8.5
8.4
250
200
150
100
50
8.3
0
25
TEMPERATURE – 8C
0
70
0
Figure 9. Switch ON Time vs. Temperature
25
TEMPERATURE – 8C
70
Figure 12. Quiescent Current vs. Temperature
66
160
65
155
150
BIAS CURRENT – nA
64
DUTY CYCLE – %
70
Figure 11. Switch ON Voltage Step-Down vs. Temperature
9.0
8.2
25
TEMPERATURE – 8C
63
DUTY CYCLE
62
61
145
140
BIAS CURRENT
135
130
60
59
125
0
25
TEMPERATURE – 8C
120
70
0
Figure 10. Duty Cycle vs. Temperature
REV. 0
25
TEMPERATURE – 8C
70
Figure 13. FB Pin Bias Current vs. Temperature
–5–
ADP1110
400
220
219
350
BIAS CURRENT
218
217
250
VREF – mV
BIAS CURRENT – nA
300
200
REFERENCE VOLTAGE
216
215
150
214
100
213
50
212
211
0
0
25
TEMPERATURE – 8C
0
70
25
TEMPERATURE – 8C
70
Figure 15. Reference Voltage vs. Temperature
Figure 14. Set Pin Bias Current vs. Temperature
THEORY OF OPERATION
The ADP1110 is a flexible, low-power, switch-mode power
supply (SMPS) controller. The regulated output voltage can be
greater than the input voltage (boost or step-up mode) or less
than the input (buck or step-down mode). This device uses a
gated-oscillator technique to provide very high performance with
low quiescent current.
A functional block diagram of the ADP1110 is shown on the
first page. The internal 220 mV reference is connected to one
input of the comparator, while the other input is externally
connected (via the FB pin) to a feedback network connected to
the regulated output. When the voltage at the FB pin falls below
220 mV, the 70 kHz oscillator turns on. A driver amplifier provides
base drive to the internal power switch, and the switching action
raises the output voltage. When the voltage at the FB pin exceeds
220 mV, the oscillator is shut off. While the oscillator is off, the
ADP1110 quiescent current is only 300 µA. The comparator
includes a small amount of hysteresis, which ensures loop
stability without requiring external components for frequency
compensation.
both step-up and step-down modes of operation. For the stepup mode, the emitter (Pin SW2) is connected to GND and the
collector (Pin SW1) drives the inductor. For step-down mode,
the emitter drives the inductor while the collector is connected
to VIN.
The output voltage of the ADP1110 is set with two external
resistors. Three fixed-voltage models are also available:
ADP1110–3.3 (+3.3 V), ADP1110–5 (+5 V) and ADP1110-12
(+12 V). The fixed-voltage models are identical to the
ADP1110 except that laser-trimmed voltage-setting resistors are
included on the chip. Only three external components are
required to form a +3.3 V, +5 V or +12 V converter. On the
fixed-voltage models of the ADP1110, simply connect the
SENSE pin (Pin 8) directly to the output voltage.
COMPONENT SELECTION
General Notes on Inductor Selection
The maximum current in the internal power switch can be set
by connecting a resistor between VIN and the ILIM pin. When the
maximum current is exceeded, the switch is turned OFF. The
current limit circuitry has a time delay of about 800 ns. If an
external resistor is not used, connect ILIM to VIN. Further information on ILIM is included in the “Applications” section of this data
sheet.
The ADP1110 internal oscillator provides 10 µs ON and 5 µs
OFF times, which is ideal for applications where the ratio between
VIN and VOUT is roughly a factor of three (such as generating +5 V
from a single 1.5 V cell). Wider range conversions, as well as
step-down converters, can also be accomplished with a slight
loss in the maximum output power that can be obtained.
An uncommitted gain block on the ADP1110 can be connected
as a low–battery detector. The inverting input of the gain block
is internally connected to the 220 mV reference. The noninverting
input is available at the SET pin. A resistor divider, connected
between VIN and GND with the junction connected to the SET
pin, causes the AO output to go LOW when the low battery set
point is exceeded. The AO output is an open collector NPN
transistor that can sink 300 µA.
When the ADP1110 internal power switch turns on, current
begins to flow in the inductor. Energy is stored in the inductor
core while the switch is on, and this stored energy is then
transferred to the load when the switch turns off. Because both
the collector and the emitter of the switch transistor are
accessible on the ADP1110, the output voltage can be higher,
lower, or of opposite polarity than the input voltage.
To specify an inductor for the ADP1110, the proper values of
inductance, saturation current, and DC resistance must be
determined. This process is not difficult, and specific equations
for each circuit configuration are provided in this data sheet. In
general terms, however, the inductance value must be low
enough to store the required amount of energy (when both
input voltage and switch ON time are at a minimum) but high
enough that the inductor will not saturate when both VIN and
switch ON time are at their maximum values. The inductor
must also store enough energy to supply the load without
saturating. Finally, the dc resistance of the inductor should be
low so that excessive power will not be wasted by heating the
windings. For most ADP1110 applications, an inductor of
15 µH to 100 µH with a saturation current rating of 300 mA to
1A and dc resistance <0.4 Ω is suitable. Ferrite-core inductors
that meet these specifications are available in small, surfacemount packages.
To minimize Electro-Magnetic Interference (EMI), a toroid or
pot-core type inductor is recommended. Rod-core inductors are
a lower-cost alternative if EMI is not a problem.
The ADP1110 provides external connections for both the
collector and emitter of its internal power switch, which permits
–6–
REV. 0
ADP1110
CALCULATING THE INDUCTOR VALUE
Selecting the proper inductor value is a simple three-step
process:
1. Define the operating parameters: minimum input voltage,
maximum input voltage, output voltage and output current.
2. Select the appropriate conversion topology (step-up, stepdown, or inverting).
3. Calculate the inductor value, using the equations in the
following sections.
INDUCTOR SELECTION–STEP-UP CONVERTER
In a step-up or boost converter (Figure 19), the inductor must
store enough power to make up the difference between the input
voltage and the output voltage. The power that must be stored
is calculated from the equation:
(
)
P L = V OUT +V D −V IN (MIN ) • ( IOUT )
(Equation 1)
where VD is the diode forward voltage (≈ 0.5 V for a 1N5818
Schottky). Because energy is only stored in the inductor while
the ADP1110 switch is ON, the energy stored in the inductor
on each switching cycle must be must be equal to or greater
than:
PL
(Equation 2)
f OSC
in order for the ADP1110 to regulate the output voltage.
When the internal power switch turns ON, current flow in the
inductor increases at the rate of:
I L (t) =
–R't 
V IN 
1− e L 

R' 

(Equation 3)
where L is in Henrys and R' is the sum of the switch equivalent
resistance (typically 0.8 Ω at +25°C) and the dc resistance of
the inductor. If the voltage drop across the switch is small
compared to VIN, a simpler equation can be used:
I L (t) =
Assuming a peak current of 1 A as a starting point, (Equation 4)
can be rearranged to recommend an inductor value:
L=
Substituting a standard inductor value of 47 µH with 0.2 Ω dc
resistance will produce a peak switch current of:
I PEAK =
–1.0 Ω •10 µs 
4.5V 
1− e 47 µH  = 862 mA

1.0 Ω 

Once the peak current is known, the inductor energy can be
calculated from Equation 5:
EL =
1
(47 µH ) • (862 mA)2 =17.5 µJ
2
Since the inductor energy of 17.5 µJ is greater than the PL/fOSC
requirement of 13.7 µJ, the 47 µH inductor will work in this
application. By substituting other inductor values into the same
equations, the optimum inductor value can be determined.
When selecting an inductor, the peak current must not exceed
the maximum switch current of 1.5 A.
The peak current must be evaluated for both minimum and
maximum values of input voltage. If the switch current is high
when VIN is at its minimum, the 1.5 A limit may be exceeded at the
maximum value of VIN. In this case, the ADP1110’s current limit
feature can be used to limit switch current. Simply select a resistor
(using Figure 7) that will limit the maximum switch current to the
IPEAK value calculated for the minimum value of VIN. This will
improve efficiency by producing a constant IPEAK as VIN increases.
See the “Limiting the Switch Current” section of this data sheet for
more information.
Note that the switch current limit feature does not protect the
circuit if the output is shorted to ground. In this case, current is
only limited by the dc resistance of the inductor and the forward
voltage of the diode.
INDUCTOR SELECTION–STEP-DOWN CONVERTER
V IN
t
L
(Equation 4)
Replacing ‘t’ in the above equation with the ON time of the
ADP1110 (10 µs, typical) will define the peak current for a
given inductor value and input voltage. At this point, the
inductor energy can be calculated as follows:
1
E L = L • I 2 PEAK
2
(Equation 5)
As previously mentioned, EL must be greater than PL/fOSC so
that the ADP1110 can deliver the necessary power to the load.
For best efficiency, peak current should be limited to 1 A or
less. Higher switch currents will reduce efficiency because of
increased saturation voltage in the switch. High peak current also
increases output ripple. As a general rule, keep peak current as low
as possible to minimize losses in the switch, inductor and diode.
The step-down mode of operation is shown in Figure 20.
Unlike the step-up mode, the ADP1110’s power switch does not
saturate when operating in the step-down mode; therefore,
switch current should be limited to 800 mA in this mode. If the
input voltage will vary over a wide range, the ILIM pin can be
used to limit the maximum switch current. Higher switch
current is possible by adding an external switching transistor as
shown in Figure 22.
The first step in selecting the step-down inductor is to calculate
the peak switch current as follows:
IPEAK =
VSW = voltage drop across the switch
VD = diode drop (0.5 V for a 1N5818)
IOUT = output current
P L = (12 V + 0.5 V − 4.5 V ) •120 mA = 960 mW
VOUT = the output voltage
On each switching cycle, the inductor must supply:
PL
f OSC
2 IOUT  V OUT +VD 
DC V IN –V SW +V D 
where: DC = duty cycle (0.69 for the ADP1110)
In practice, the inductor value is easily selected using the equations
above. For example, consider a supply that will generate 12 V at
120 mA from a 4.5 V to 8 V source. The inductor power required
is from Equation 1:
REV. 0
V IN
4.5V
t=
10 µs = 45 µH
I L(MAX )
1A
VIN = the minimum input voltage
960 mW
=
=13.7 µJ
70 kHz
–7–
(Equation 6)
ADP1110
For example, assume that a –5 V output at 75 mA is to be
generated from a +4.5 V to +5.5 V source. The power in the
inductor is calculated from Equation 8:
As previously mentioned, the switch voltage is higher in stepdown mode than in step-up mode. VSW is a function of switch
current and is therefore a function of VIN, L, time and VOUT.
For most applications, a VSW value of 1.5 V is recommended.
P L = (|– 5V|+ 0.5V ) • (75 mA) = 413 mW
The inductor value can now be calculated:
–V SW –V OUT
V
L = IN(MIN )
• tON
I PEAK
During each switching cycle, the inductor must supply the
following energy:
(Equation 7)
PL
413 mW
=
= 5.9 µJ
f OSC 70 kHz
where: tON = Switch ON time (10 µs)
Using a standard inductor value of 56 µH with 0.2 Ω dc
resistance will produce a peak switch current of:
If the input voltage will vary (such as an application that must
operate from a 9 V, 12 V or 15 V source), an RLIM resistor
should be selected from Figure 6. The RLIM resistor will keep
switch current constant as the input voltage rises. Note that
there are separate RLIM values for step-up and step-down modes
of operation.
I PEAK =
EL =
The input voltage only varies between 4.5 V and 5.5 V in this
example. Therefore, the peak current will not change enough to
require an RLIM resistor and the ILIM pin can be connected
directly to VIN. Care should be taken, of course, to ensure that
the peak current does not exceed 800 mA.
Then, the peak current can be inserted into Equation 7 to
calculate the inductor value:
9 –1.5 – 5
•10 µs = 50 µs
498 mA
CAPACITOR SELECTION
Since 50 µH is not a standard value, the next lower standard
value of 47 µH would be specified.
For optimum performance, the ADP1110’s output capacitor
must be selected carefully. Choosing an inappropriate capacitor
can result in low efficiency and/or high output ripple.
To avoid exceeding the maximum switch current when the
input voltage is at +18 V, an RLIM resistor should be specified.
Using the step-down curve of Figure 6, a value of 560 Ω will
limit the switch current to 500 mA.
Ordinary aluminum electrolytic capacitors are inexpensive but
often have poor Equivalent Series Resistance (ESR) and
Equivalent Series Inductance (ESL). Low ESR aluminum
capacitors, specifically designed for switch mode converter
applications, are also available, and these are a better choice
than general purpose devices. Even better performance can be
achieved with tantalum capacitors, although their cost is higher.
Very low values of ESR can be achieved by using OS-CON
capacitors (Sanyo Corporation, San Diego, CA). These devices
are fairly small, available with tape-and-reel packaging and have
very low ESR.
INDUCTOR SELECTION—POSITIVE-TO-NEGATIVE
CONVERTER
The configuration for a positive-to-negative converter using the
ADP1110 is shown in Figure 23. As with the step-up converter,
all of the output power for the inverting circuit must be supplied
by the inductor. The required inductor power is derived from
the formula:
PL =
(V
OUT
) (
+ V D • I OUT
)
(Equation 8)
The effects of capacitor selection on output ripple are demonstrated in Figures 16, 17 and 18. These figures show the output
of the same ADP1110 converter, that was evaluated with three
different output capacitors. In each case, the peak switch
current is 500 mA, and the capacitor value is 100 µF. Figure 16
shows a Panasonic HF-series 16-volt radial cap. When the
switch turns off, the output voltage jumps by about 90 mV and
then decays as the inductor discharges into the capacitor. The
rise in voltage indicates an ESR of about 0.18 Ω. In Figure 17,
the aluminum electrolytic has been replaced by a Sprague 293D
series, a 6 V tantalum device. In this case the output jumps
The ADP1110 power switch does not saturate in positive-tonegative mode. The voltage drop across the switch can be
modeled as a 0.75 V base-emitter diode in series with a 0.65 Ω
resistor. When the switch turns on, inductor current will rise at
a rate determined by:
I L (t) =
–R't 
VL 
1− e L 

R' 

1
(56 µH ) • (621 mA)2 =10.8 µJ
2
Since the inductor energy of 10.8 µJ is greater than the PL/fOSC
requirement of 5.9 µJ, the 56 µH inductor will work in this
application.
2•250 mA  5+ 0.5 
= 498 mA
0.69  9 −1.5+ 0.5
L=
–0.85 Ω •10 µs 

56 µH
1− e
 = 621 mA


Once the peak current is known, the inductor energy can be
calculated from Equation 9:
For example, assume that +5 V at 250 mA is required from a
+9 V to +18 V source. Deriving the peak current from Equation
6 yields:
I PEAK =
4.5V – 0.75V
0.65 Ω + 0.2 Ω
(Equation 9)
where: R' = 0.65 Ω + RL(DC)
VL = VIN – 0.75 V
–8–
REV. 0
ADP1110
about 30 mV, which indicates an ESR of 0.06 Ω. Figure 18
shows an OS-CON 16–volt capacitor in the same circuit, and
ESR is only 0.02 Ω.
50mV
100
90
important in low-current applications, where the leakage can be
a significant percentage of the total quiescent current.
For most circuits, the 1N5818 is a suitable companion to the
ADP1110. This diode has a VF of 0.5 V at 1 A, 4 µA to 10 µA
leakage, and fast turn-on and turn-off times. A surface mount
version, the MBRS130LT3, is also available.
For switch currents of 100 mA or less, a Shottky diode such as
the BAT85 provides a VF of 0.8 V at 100 mA and leakage less
than 1 µA. A similar device, the BAT54, is available in a SOT23
package. Even lower leakage, in the 1 nA to 5 nA range, can be
obtained with a 1N4148 signal diode.
10
0%
1µs
Figure 16. Aluminum Electrolytic
50mV
100
90
General purpose rectifiers, such as the 1N4001, are not suitable
for ADP1110 circuits. These devices, which have turn-on times
of 10 µs or more, are too slow for switching power supply
applications. Using such a diode “just to get started” will result
in wasted time and effort. Even if an ADP1110 circuit appears
to function with a 1N4001, the resulting performance will not
be indicative of the circuit performance when the correct diode
is used.
CIRCUIT OPERATION, STEP-UP (BOOST) MODE
10
0%
1µs
Figure 17. Tantalum Electrolytic
50mV
100
90
In boost mode, the ADP1110 produces an output voltage that is
higher than the input voltage. For example, +5 V can be derived
from one alkaline cell (+1.5 V), or +12 V can be generated from
a +5 V logic power supply.
Figure 19 shows an ADP1110 configured for step-up operation.
The collector of the internal power switch is connected to the
output side of the inductor, while the emitter is connected to
GND. When the switch turns on, pin SW1 is pulled near
ground. This action forces a voltage across L1 equal to VIN –
VCE(SAT), and current begins to flow through L1. This current
reaches a final value (ignoring second-order effects) of:
I PEAK ≅
V IN –V CE(SAT )
•10 µs
L
where 10 µs is the ADP1110 switch’s “on” time.
10
0%
L1
D1
VIN
1µs
VOUT
R3*
1
Figure 18. OS-CON Capacitor
ILIM
If low output ripple is important, the user should consider the
ADP3000. Because this device switches at 400 kHz, lower peak
current can be used. Also, the higher switching frequency
simplifies the design of the output filter. Consult the ADP3000
data sheet for additional details.
REV. 0
R1
3
ADP1110
FB
GND
SW2
5
4
C1
8
R2
*OPTIONAL
DIODE SELECTION
In specifying a diode, consideration must be given to speed,
forward voltage drop and reverse leakage current. When the
ADP1110 switch turns off, the diode must turn on rapidly if
high efficiency is to be maintained. Shottky rectifiers, as well as
fast signal diodes such as the 1N4148, are appropriate. The
forward voltage of the diode represents power that is not
delivered to the load, so VF must also be minimized. Again,
Schottky diodes are recommended. Leakage current is especially
2
VIN
SW1
Figure 19. Step-Up Mode Operation
When the switch turns off, the magnetic field collapses. The
polarity across the inductor changes, current begins to flow
through D1 into the load, and the output voltage is driven above
the input voltage.
The output voltage is fed back to the ADP1110 via resistors R1
and R2. When the voltage at pin FB falls below 220 mV, SW1
–9–
ADP1110
base junction when the switch is off. To protect the switch, the
output voltage should be limited to 6.2 V or less. If a higher
output voltage is required, a Schottky diode should be placed in
series with SW2, as shown in Figure 21.
turns “on” again, and the cycle repeats. The output voltage is
therefore set by the formula:
 R1
V OUT = 220 mV • 1+ 
 R2
INPUT
The circuit of Figure 19 shows a direct current path from VIN to
VOUT, via the inductor and D1. Therefore, the boost converter
is not protected if the output is short circuited to ground.
CINPUT
RLIM
1
2
The ADP1110’s step-down mode is used to produce an output
voltage that is lower than the input voltage. For example, the
output of four NiCd cells (+4.8 V) can be converted to a +3 V
logic supply.
A typical configuration for step-down operation of the ADP1110 is
shown in Figure 20. In this case, the collector of the internal
power switch is connected to VIN and the emitter drives the
inductor. When the switch turns on, SW2 is pulled up towards
VIN. This forces a voltage across L1 equal to VIN – VCE – VOUT
and causes current to flow in L1. This current reaches a final
value of:
V IN −V CE −V OUT
•10 µs
L
where 10 µs is the ADP1110 switch’s “on” time.
VIN
C2
ILIM
2
3
VIN SW1
FB 8
ADP1110
L1
6
7
D1
1N5818
NC
NC
VOUT
SW2 4
AO SET GND
5
C1
OUTPUT
R1
AO
6
SET GND
7
5
NC
NC
FB 8
D1
1N5818
R2
CL
Figure 21. Step-Down Mode, VOUT > 6.2 V
If the input voltage to the ADP1110 varies over a wide range, a
current limiting resistor at Pin 1 may be required. If a particular
circuit requires high peak inductor current with minimum input
supply voltage, the peak current may exceed the switch maximum rating and/or saturate the inductor when the supply
voltage is at the maximum value. See the “Limiting the Switch
Current” section of this data sheet for specific recommendations.
INCREASING OUTPUT CURRENT IN THE STEP-DOWN
REGULATOR
RLIM
100Ω
1
L1
ADP1110
CIRCUIT OPERATION, STEP-DOWN (BUCK) MODE
IPEAK ≅
3
VIN SW1
SW2 4
ILIM
R1
R2
Figure 20. Step-Down Mode Operation
When the switch turns off, the magnetic field collapses. The
polarity across the inductor changes, and the switch side of the
inductor is driven below ground. Schottky diode D1 then turns
on, and current flows into the load. Notice that the Absolute
Maximum Rating for the ADP1110’s SW2 pin is 0.5 V below
ground. To avoid exceeding this limit, D1 must be a Schottky
diode. Using a silicon diode in this application will generate
forward voltages above 0.5 V that will cause potentially
damaging power dissipation within the ADP1110.
Unlike the boost configuration, the ADP1110’s internal power
switch is not saturated when operating in step-down mode. A
conservative value for the voltage across the switch in step-down
mode is 1.5 V. This results in high power dissipation within the
ADP1110 when high peak current is required. To increase the
output current, an external PNP switch can be added (Figure
22). In this circuit, the ADP1110 provides base drive to Q1
through R3, while R4 ensures that Q1 turns off rapidly. Because
the ADP1110’s internal current limiting function will not work
in this circuit, R5 is provided for this purpose. With the value
shown, R5 limits current to 2 A. In addition to reducing power
dissipation on the ADP1110, this circuit also reduces the switch
voltage. When selecting an inductor value for the circuit of
Figure 22, the switch voltage can be calculated from the
formula:
V SW =V R5 +V Q1(SAT ) ≅ 0.6V + 0.4V ≅1V
0.3Ω
INPUT
The output voltage of the buck regulator is fed back to the
ADP1110’s FB pin by resistors R1 and R2. When the voltage at
pin FB falls below 220 mV, the internal power switch turns
“on” again and the cycle repeats. The output voltage is set by
the formula:
CINPUT
R5
RLIM
R4
220Ω
MJE210
1
2
ILIM
VIN
R3
330Ω
L1
OUTPUT
SW1 3
ADP1110
R1
FB 8
AO SET GND SW2

R1
V OUT = 220 mV • 1+ 
 R2
When operating the ADP1110 in step-down mode, the output
voltage is impressed across the internal power switch’s emitter-
6
7
NC
NC
5
4
D1
1N5821
R2
CL
Figure 22. High Current Step-Down Operation
–10–
REV. 0
ADP1110
POSITIVE-TO-NEGATIVE CONVERSION
LIMITING THE SWITCH CURRENT
The ADP1110 can convert a positive input voltage to a negative
output voltage as shown in Figure 23. This circuit is essentially
identical to the step-down application of Figure 19, except that
the “output” side of the inductor is connected to power ground.
When the ADP1110’s internal power switch turns off, current
flowing in the inductor forces the output (–VOUT) to a negative
potential. The ADP1110 will continue to turn the switch on
until its FB pin is 220 mV above its GND pin, so the output
voltage is determined by the formula:
The ADP1110’s RLIM pin permits the switch current to be
limited with a single resistor. This current limiting action occurs
on a pulse by pulse basis. This feature allows the input voltage
to vary over a wide range without saturating the inductor or
exceeding the maximum switch rating. For example, a particular
design may require peak switch current of 800 mA with a 2.0 V
input. If VIN rises to 4 V, however, the switch current will
exceed 1.6 A. The ADP1110 limits switch current to 1.5 A and
thereby protects the switch, but the output ripple will increase.
Selecting the proper resistor will limit the switch current to
800 mA, even if VIN increases. The relationship between RLIM
and maximum switch current is shown in Figure 6.
 R1
V OUT = 220 mV • 1+ 
 R2
The ILIM feature is also valuable for controlling inductor current
when the ADP1110 goes into continuous-conduction mode.
This occurs in the step-up mode when the following condition is
met:
INPUT
RLIM
1
CINPUT
2
3
ILIM
VIN
L1
SW1
SW2 4
OUTPUT
ADP1110
AO
SET
6
7
NC
NC
V OUT +VD IODE 
1
 V –V
 < 1– DC


IN
SW
R1
FB 8
GND
D1
1N5818
5
CL
R2
NEGATIVE
OUTPUT
Figure 23. A Positive-to-Negative Converter
The design criteria for the step-down application also apply to
the positive-to-negative converter. The output voltage should be
limited to |6.2 V| unless a diode is inserted in series with the
SW2 pin (see Figure 21.) Also, D1 must again be a Schottky
diode to prevent excessive power dissipation in the ADP1110.
where DC is the ADP1110’s duty cycle. When this relationship
exists, the inductor current does not go all the way to zero
during the time that the switch is OFF. When the switch turns
on for the next cycle, the inductor current begins to ramp up
from the residual level. If the switch ON time remains constant,
the inductor current will increase to a high level (see Figure 25).
This increases output ripple and can require a larger inductor
and capacitor. By controlling switch current with the ILIM
resistor, output ripple current can be maintained at the design
values. Figure 26 illustrates the action of the ILIM circuit.
NEGATIVE-TO-POSITIVE CONVERSION
The circuit of Figure 24 converts a negative input voltage to a
positive output voltage. Operation of this circuit configuration is
similar to the step-up topology of Figure 19, except the current
through feedback resistor R1 is level-shifted below ground by a
PNP transistor. The voltage across R1 is VOUT – VBEQ1. However,
diode D2 level-shifts the base of Q1 about 0.6 V below ground
thereby cancelling the VBE of Q1. The addition of D2 also reduces
the circuit’s output voltage sensitivity to temperature, which otherwise would be dominated by the –2 mV VBE contribution of Q1.
The output voltage for this circuit is determined by the formula:
100
90
200mA/div.
10
0%
10mV
 R1
V OUT = 220 mV •  
 R2
Figure 25. ILIM Operation—IL Characteristic
Unlike the positive step-up converter, the negative-to-positive
converter’s output voltage can be either higher or lower than the
input voltage.
L1
2
ILIM
VIN
CINPUT
SW1 3
NEGATIVE
INPUT
NC
NC
CL
5
4
POSITIVE
OUTPUT
D2
1N4148
10
0%
10mV
Figure 26. ILIM Operation—IL Characteristic
R2
Figure 24. A Negative-to-Positive Converter
REV. 0
10µs
10K
FB 8
AO SET GND SW2
7
90
200mA/div.
Q1
2N3906
ADP1110
6
100
D1
R1
RLIM
1
10µs
–11–
ADP1110
The internal structure of the ILIM circuit is shown in Figure 27.
Q1 is the ADP1110’s internal power switch, that is paralleled by
sense transistor Q2. The relative sizes of Q1 and Q2 are scaled
so that IQ2 is 0.5% of IQ1. Current flows to Q2 through an
internal 80 Ω resistor and through the RLIM resistor. These two
resistors parallel the base-emitter junction of the oscillatordisable transistor, Q3. When the voltage across R1 and RLIM
exceeds 0.6 V, Q3 turns on and terminates the output pulse. If
only the 80 Ω internal resistor is used (i.e., the ILIM pin is
connected directly to VIN), the maximum switch current will be
1.5 A. Figure 6 gives RLIM values for lower current-limit values.
ILIM
VIN
R1
80Ω
(INTERNAL)
Q3
ADP1110
IQ1
DRIVER
72kHz
OSC
RL
ADP1110
220V
VREF
R1
VBAT
AO
SET
R2
33kΩ
GND
RHYS
Figure 28. Setting the Low Battery Detector Trip Point
RLIM
(EXTERNAL)
VIN
VLOGIC
SW1
200
Q2
Q1
POWER
SWITCH
The circuit of Figure 28 may produce multiple pulses when
approaching the trip point due to noise coupled into the SET
input. To prevent multiple interrupts to the digital logic,
hysteresis can be added to the circuit. Resistor RHYS, with a
value of 1 MΩ to 10 MΩ, provides the hysteresis. The addition
of RHYS will change the trip point slightly, so the new value for
R1 will be:
SW2
Figure 27. ADP1110 Current Limit Operation
The delay through the current limiting circuit is approximately
800 ns. If the switch ON time is reduced to less than 3 µs,
accuracy of the current trip-point is reduced. Attempting to
program a switch ON time of 800 ns or less will produce
spurious responses in the switch ON time; however, the
ADP1110 will still provide a properly-regulated output voltage.
PROGRAMMING THE GAIN BLOCK
The gain block of the ADP1110 can be used as a low-battery
detector, error amplifier or linear post regulator. The gain block
consists of an op amp with PNP inputs and an open-collector
NPN output. The inverting input is internally connected to the
ADP1110’s 220 mV reference, while the noninverting input is
available at the SET pin. The NPN output transistor will sink
about 300 µA.
Figure 28 shows the gain block configured as a low-battery
monitor. Resistors R1 and R2 should be set to high values to
reduce quiescent current, but not so high that bias current in
the SET input causes large errors. A value of 33 kΩ for R2 is a
good compromise. The value for R1 is then calculated from the
formula:
R1=
V LOBATT – 220 mV
220 mV V L – 220 mV 
–

R2
 RL + RHYS 
where VL is the logic power supply voltage, RL is the pull-up
resistor, and RHYS creates the hysteresis.
The gain block can also be used as a control element to reduce
output ripple. The ADP3000 is normally recommended for lowripple applications, but its minimum input voltage is 2 V. The
gain-block technique using the ADP1110 can be useful for stepup converters operating down to 1 V.
A step-up converter using this technique is shown in Figure 29.
This configuration uses the gain block to sense the output
voltage and control the comparator. The result is that the
comparator hysteresis is reduced by the open loop gain of the
gain block. Output ripple can be reduced to only a few millivolts
with this technique, versus a typical value of 90 mV for a +5 V
converter using just the comparator. For best results, a large
output capacitor (1000 µF or more) should be specified. This
technique can also be used for step-down or inverting applications, but the ADP3000 is usually a more appropriate choice.
See the ADP3000 data sheet for further details.
INPUT
CINPUT
10µF
– 220 mV
V
R1= LOBATT
220 mV
R2
270kΩ
1
2
ILIM
VIN
L1
D1
15µH
CTX15-4
1N5818
OUTPUT
R1
300kΩ
SW1 3
ADP1110
SET 7
AO FB GND SW2
where VLOBATT is the desired low battery trip point. Since the
gain block output is an open-collector NPN, a pull-up resistor
should be connected to the positive logic power supply.
6
8
5
4
R2
13.8kΩ
CL
1000µF
Figure 29. Using the Gain Block to Reduce Output Ripple
–12–
REV. 0
ADP1110
1.5 V to 65 V Dual-Output Step-Up Converter
APPLICATION CIRCUITS
All-Surface-Mount, Single-Cell to 5 V Converter
This is a very simple, compact, low-part-count circuit that takes
a single alkaline 1.5 V cell input and produces a 5 V output.
The output current should be kept to 10 mA or less to conserve
battery life.
This circuit works from a single 1.5 V cell and provides simultaneous outputs of +5 V and –5 V. The accuracy of the negative
output suffers slightly because of the extra diode drop of around
0.4 V.
L16
8µH
L1
D1
1N5817
ONE
ALKALINE
CELL
1.5V
1
2
ILIM
VIN
CTX68-4
+5V
10mA
50µH
CTX50-4
ONE
ALKALINE
CELL
1.5V
6
NC
NC
Similar to the previous circuit, this circuit takes a 3-volt input
and provides a 5 V output at 40 mA. As in the single-cell version,
the circuit is compact and uses only four external components.
L1
D1
1N5817
50µH
CTX50-4
220Ω
1
2
ILIM
VIN
+5V
40mA
SW1 3
SENSE 8
AO SET GND SW2
7
NC
NC
NC
NC
5
4
Figure 33. 1.5 V to ± 5 V Dual-Output Step-Up
Converter
5
All-Surface-Mount Flash Memory VPP Generator
Figure 34 shows a circuit that can generate the programming
voltage, VPP to program flash memory. The key components
are the MOSFET and the bipolar transistor. These two devices
form a switch that, when ON, allows the ADP1110 to power-up
and function as a step-up converter. The output is +12 V at 120
mA. When the MOSFET switch is OFF, the output of the
circuit drops to just under +5 V thereby disabling the programming capability.
Care should be taken so there is no short-circuit-current limiting
in the circuit in either operating mode.
ADP1110-5
6
7
NOTE: ALL DIODES 1N5818
All-Surface-Mount, 3 V to 5 V Step-Up Converter
3V
6
4.7µF
CL
15µF
4
Figure 30. All-Surface-Mount, Single-Cell to 5 V Converter
TWO
ALKALINE
CELLS
+5V
3mA
–5V
3mA
4.7µF
SENSE 8
AO SET GND SW2
SENSE 8
AO SET GND SW2
5
VIN SW1 3
ADP1110-5
SW1 3
ADP1110-5
7
4.7µF
2
1
ILIM
CL
10µF
4
L1
50µH
+5V
MMBT4403
CTX50-4
D1
1N5818
VPP
+12V
120mA
10kΩ
Figure 31. All-Surface-Mount, 3 V to 5 V Step-Up
Converter
All-Surface-Mount, 9 V to 5 V Step-Down Converter
2
1
ILIM
VIN
3
SW1
SW2 4
L1
50µH
CTX50-4
ADP1110-5
7
5
D1
1N5817
+5V
40mA
CL
10µF
NC NC
Figure 32. All-Surface-Mount, 9 V to 5 V Step-Down
Converter
REV. 0
SENSE 8
AO SET GND SW2
6
7
NC
NC
5
4
CL
10µF
Figure 34. All Surface-Mount Flash Memory VPP Generator
SENSE 8
AO SET GND
6
SW1 3
ADP1110-12
MMBF170
LOGIC1 = PROGRAM
LOGIC0 = SHUTDOWN
9V
BATTERY
2
VIN
1kΩ
Featuring the same low parts count of the step-up design, this
circuit is the complement to the preceding one. The 220 Ω
resistor programs the current limit to around 600 mA.
RLIM
220Ω
1
ILIM
–13–
ADP1110
1.5 V to +5 V, +10 V Dual Output Step-Up Converter
1.5 V-Powered Laser Diode Driver
The circuit of Figure 35 illustrates a way to get outputs of
+10 V and +5 V from the same converter. The main 5 V output
is derived from the feedback provided by the 487 kΩ and 11 kΩ
resistors. Capacitor C1 should be a multilayer ceramic variety
for best performance, but a good quality tantalum capacitor will
also give good performance at lower cost.
Figure 36 shows a circuit suitable for driving many laser diodes
that incorporate a photodiode to monitor the laser diode
current, this circuit makes use of the gain block and currentlimit functions to provide a feedback system based on the
average laser diode current. This current must be controlled
very closely or permanent damage to the laser diode is likely to
be the result.
L1
50µH
220Ω
+10V
3mA
2
1
ONE
ALKALINE
CELL
1.5V
ILIM
D2
VIN SW1 3
487kΩ
D3
ADP1110
FB 8
AO SET GND SW2
6
7
NC
NC
5
To ensure that the laser is operating at the proper power level,
the actual optical power from the laser should be monitored
with a calibrated photodiode or optical power meter. In
addition, the actual diode current should also be monitored, and
R1 can be adjusted to give the correct output power.
D1
CTX50-4
+5V
3mA
NOTES
1. All inductors referenced are Coiltronics CTX-series except
where noted.
4
4.7µF
11kΩ
4.7µF
2. If the source of power is more than an inch or so from the
converter, the input to the converter should be bypassed with
approximately 10 µF of capacitance. This capacitor should be
a good quality tantalum or aluminum electrolytic.
NOTE: ALL DIODES 1N5818
Figure 35. 1.5 V to +5 V, +10 V Dual Output Step-Up
Converter
TOSHIBA
TOLD-9321
0.022µF
5.1kΩ
2N3906
1kΩ
1N4148
1µF
220Ω
1
2
ILIM
VIN
100µF
OS-Con
MJE210
10Ω
1N5818
SW1 3
2Ω
ADP1110
1.5V
AO
FB
6
8
SET 7
GND SW2
5
4
1kΩ
2.2µH
Figure 36. 1.5 V-Powered Laser Diode Driver
–14–
REV. 0
ADP1110
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
8-Lead Plastic DIP
(N-8)
0.430 (10.92)
0.348 (8.84)
8
5
0.280 (7.11)
0.240 (6.10)
1
4
0.060 (1.52)
0.015 (0.38)
PIN 1
0.210 (5.33)
MAX
0.325 (8.25)
0.300 (7.62)
0.195 (4.95)
0.115 (2.93)
0.130
(3.30)
MIN
0.160 (4.06)
0.115 (2.93)
0.022 (0.558) 0.100 0.070 (1.77)
0.014 (0.356) (2.54) 0.045 (1.15)
BSC
0.015 (0.381)
0.008 (0.204)
SEATING
PLANE
8-Lead SOIC
(SO-8)
0.1968 (5.00)
0.1890 (4.80)
0.1574 (4.00)
0.1497 (3.80)
PIN 1
0.0098 (0.25)
0.0040 (0.10)
SEATING
PLANE
REV. 0
8
5
1
4
0.2440 (6.20)
0.2284 (5.80)
0.0688 (1.75)
0.0532 (1.35)
0.0500 0.0192 (0.49)
(1.27) 0.0138 (0.35)
BSC
0.0196 (0.50)
x 45°
0.0099 (0.25)
0.0098 (0.25)
0.0075 (0.19)
–15–
8°
0°
0.0500 (1.27)
0.0160 (0.41)
–16–
PRINTED IN U.S.A.
C2212–12–10/96
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