TI OPA2690 Dual, wideband, high output current, operational amplifier with current limit Datasheet

OPA2613
SBOS249D − JUNE 2003− REVISED APRIL 2004
Dual, Wideband, High Output Current,
Operational Amplifier with Current Limit
FEATURES
D
D
D
D
D
D
D
APPLICATIONS
D
D
D
D
D
D
D
LOW INPUT NOISE VOLTAGE: 1.8nV/√Hz
HIGH UNITY-GAIN BANDWIDTH: 230MHz
HIGH GAIN BANDWIDTH PRODUCT: 125MHz
HIGH OUTPUT CURRENT: 350mA
LOW INPUT OFFSET VOLTAGE: ±0.2mV
FLEXIBLE SUPPLY RANGE:
Single +5V to +12V Operation
Dual ±2.5V to ±6V Operation
LOW SUPPLY CURRENT: 6.0mA/ch
xDSL DIFFERENTIAL LINE DRIVERS
16-BIT ADC DRIVER
LOW NOISE PLL INTEGRATORS
TRANSIMPEDANCE AMPLIFIERS
PRECISION BASEBAND I/Q AMPLIFIERS
ACTIVE FILTERS
TS613 IMPROVED REPLACEMENT
OPA2613 RELATED PRODUCTS
DESCRIPTION
FEATURES
The OPA2613 offers very low 1.8nV√Hz input noise in a
wideband,
unity-gain
stable,
voltage-feedback
architecture. Intended for xDSL driver applications, the
OPA2613 also supports this low input noise with
exceptionally low harmonic distortion, particularly in
differential configurations. Adequate output current is
provided to drive the potentially heavy load of a
twisted-pair line. Harmonic distortion for a 2VPP differential
output operating from +5V to +12V supplies is ≤ −95dBc
through 1MHz input frequencies. Operating on a low
6.0mA/ch supply current, the OPA2613 can satisfy most
xDSL driver requirements over a wide range of possible
supply voltagefrom a single +5 condition, to ±5V, on up
to a single +12V design.
General-purpose applications on a single +5V supply will
benefit from the high input and output voltage swing
available on this reduced supply voltage. Low-cost
precision integrators for PLLs will also benefit from the low
voltage noise and offset voltage. Baseband I/Q receiver
channels can achieve almost perfect channel match with
noise and distortion to support signals through 5MHz with
> 14-bit dynamic range.
Very high line power requirements can be supported using
the thermally-enhanced heat slug package. Soldered into
a standard printed circuit board, this heat slug reduces the
thermal impedance junction-to-ambient to < 50°C/W.
SINGLES
DUALS
TRIPLES
High Gain Stable

OPA2614

High Slew Rate VFB
OPA690
OPA2690
OPA3690
R/R Input/Output VFB
OPA353
OPA2353

Current-Feedback
OPA691
OPA2691
OPA3691
Current-Feedback

OPA2677

OPA2613
RO
n:1
xDSL D river
RO
500Ω
1kΩ
500Ω
OP A2822
500Ω
1kΩ
xDSL Receiver
OP A2822
500Ω
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments
semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
Copyright  2003-2004, Texas Instruments Incorporated
! ! www.ti.com
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SBOS249D − JUNE 2003− REVISED APRIL 2004
ELECTROSTATIC
DISCHARGE SENSITIVITY
ABSOLUTE MAXIMUM RATINGS(1)
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±6.5V
Internal Power Dissipation . . . . . . . . . See Thermal Characteristics
Differential Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±1.2V
Input Voltage Range . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±VS
Storage Temperature Range . . . . . . . . . . . . . . . . . . −40°C to +125°C
Lead Temperature (SO-8, PSO-8) . . . . . . . . . . . . . . . . . . . . . . +260°C
Junction Temperature (TJ) . . . . . . . . . . . . . . . . . . . . . . . . . . . +150°C
ESD Rating (Human Body Model) . . . . . . . . . . . . . . . . . . . . 2000V
(Machine Model) . . . . . . . . . . . . . . . . . . . . . . . . . . 200V
(Charge Device Model) . . . . . . . . . . . . . . . . . . . 1500V
This integrated circuit can be damaged by ESD. Texas Instruments
recommends that all integrated circuits be handled with appropriate
precautions. Failure to observe proper handling and installation
procedures can cause damage.
ESD damage can range from subtle performance degradation to
complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could
cause the device not to meet its published specifications.
(1) Stresses above these ratings may cause permanent damage.
Exposure to absolute maximum conditions for extended periods
may degrade device reliability. These are stress ratings only, and
functional operation of the device at these or any other conditions
beyond those specified is not supported.
ORDERING INFORMATION
PRODUCT
PACKAGE−LEAD
PACKAGE
DESIGNATOR(1)
SPECIFIED
TEMPERATURE
RANGE
PACKAGE
MARKING
ORDERING
NUMBER
TRANSPORT
MEDIA, QUANTITY
OPA2613
SO-8
D
−40°C to +85°C
OPA2613
OPA2613ID
OPA2613IDR
Rails, 100
Tape and Reel, 2500
OPA2613
PSO-8
DTJ
−40°C to +85°C
OPA2613H
OPA2613IDTJ
OPA2613IDTJR
Rails, 100
Tape and Reel, 2500
″
″
″
″
″
″
″
″
″
″
(1) For the most current specification and package information, refer to our web site at www.ti.com.
PIN CONFIGURATION
SO, PSO
Top View
OPA2613
2
Out A
1
8
+VS
−In A
2
7
Out B
+In A
3
6
−In B
−VS
4
5
+In B
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SBOS249D − JUNE 2003− REVISED APRIL 2004
ELECTRICAL CHARACTERISTICS: VS = ±6V
Boldface limits are tested at +25°C.
RF = 402Ω, RL = 100Ω, and G = +2, unless otherwise noted. See Figure 1 for AC performance only.
OPA2613ID, OPA2613IDTJ
TYP
PARAMETER
AC Performance (see Figure 1)
Small-Signal Bandwidth
Gain-Bandwidth Product
Bandwidth for 0.1dB Gain Flatness
Peaking at a Gain of +1
Large-Signal Bandwidth
Slew Rate
Rise-and-Fall Time
Settling Time to 0.02%
0.1%
Harmonic Distortion
2nd-Harmonic
3rd-Harmonic
Input Voltage Noise
Input Current Noise
Differential Gain
Differential Phase
Channel-to-Channel Crosstalk
DC Performance(4)
Open-Loop Gain (AOL)
Input Offset Voltage
Average Offset Voltage Drift
Input Bias Current
Average Bias Current Drift (Magnitude)
Input Offset Current
Average Offset Bias Current Drift
UNITS
MIN/
MAX
TEST
LEVEL
(3)
MHz
MHz
MHz
MHz
MHz
dB
MHz
V/µs
ns
ns
ns
typ
min
min
min
typ
typ
typ
min
typ
typ
typ
C
B
B
B
C
C
C
B
C
C
C
−60
−87
−77
−89
2.3
2.4
dBc
dBc
dBc
dBc
nV/√Hz
pA/√Hz
%
deg
dBc
max
max
max
max
max
max
typ
typ
typ
B
B
B
B
B
B
C
C
C
92
±1.15
±3.3
−13
−30
±520
±5
91
±1.2
±3.3
−14.5
−35
±750
±7
dB
mV
µV/°C
µA
nA/°C
nA
nA/°C
min
max
max
max
max
max
max
A
A
B
A
B
A
B
+25°C(1)
0°C to
+70°C(2)
−40°C to
+85°C(2)
80
10
95
75
9
80
70
9
75
56
4.8
68
51
51
5.4
71
53
50
5.5
72
54
−70
−95
−84
−97
1.8
1.7
0.02
0.03
−80
−63
−90
−80
−92
2.0
2.1
−61
−88
−78
−90
2.1
2.2
97
±0.2
92
±1.0
−6
−12
±50
±300
TEST CONDITIONS
+25°C
G = +1, VO = 0.1VPP, RF = 0Ω
G = +2, VO = 0.1VPP
G = +10, VO = 0.1VPP
G ≥ 20
G = +2, VO < 0.1VPP
VO < 0.1VPP
G = +2, VO = 2VPP
G = +2, 4V step
G = +2, VO = 0.2V Step
G = +2, VO = 2V Step
G = +2, VO = 2V Step
G = +2, f = 1MHz, VO = 2VPP
RL = 20Ω
RL ≥ 500Ω
RL = 20Ω
RL ≥ 500Ω
f > 10kHz
f > 10kHz
G = +2, PAL, VO = 1.4VP, RL = 150Ω
G = +2, PAL, VO = 1.4VP, RL = 150Ω
f = 1MHz, Input Referred
230
110
13
125
5
1
22
70
3.6
55
40
VO = 0V, RL = 100Ω
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
MIN/MAX OVER TEMPERATURE
Input
Common-Mode Input Range (CMIR)(5)
Common-Mode Rejection Ratio (CMRR)
Input Impedance
Differential-Mode
Common-Mode
±4.7
±4.5
±4.5
±4.4
V
min
A
VCM = ±1V
100
88
87
86
dB
min
A
VCM = 0
VCM = 0
18 0.6
7 1
kΩ pF
MΩ pF
typ
typ
C
C
No Load
100Ω
VO = 0, Linear Operation
VO = 0, Linear Operation
Output Shorted to Ground
G = +2, f = 100kHz
±5.0
±4.9
+350
−350
500
0.01
V
V
mA
mA
mA
Ω
min
min
min
min
typ
typ
A
A
A
A
C
C
Output
Output Voltage Swing
Current Output, Sourcing
Current Output, Sinking
Short-Circuit Current
Closed-Loop Output Impedance
±4.8
±4.7
+280
−280
±4.8
±4.7
+240
−240
±4.7
±4.6
+220
−220
(1) Junction temperature = ambient for +25°C tested specifications.
(2) Junction temperature = ambient at low temperature limit; junction temperature = ambient +23°C at high temperature limit for over temperature
tested specifications.
(3) Test levels: (A) 100% tested at +25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and
simulation. (C) Typical value only for information.
(4) Current is considered positive-out-of-node. VCM is the input common-mode voltage.
(5) Tested < 3dB below minimum CMRR specification at ± CMIR limits.
(6) Heat slug soldered to heat spreading plane. This plane should be electrically floating or at VS− voltage.
3
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SBOS249D − JUNE 2003− REVISED APRIL 2004
ELECTRICAL CHARACTERISTICS: VS = ±6V (continued)
Boldface limits are tested at +25°C.
RF = 402Ω, RL = 100Ω, and G = +2, unless otherwise noted. See Figure 1 for AC performance only.
OPA2613ID, OPA2613IDTJ
TYP
PARAMETER
Power Supply
Specified Operating Voltage
Maximum Operating Voltage Range
Maximum Quiescent Current
Minimum Quiescent Current
Power-Supply Rejection Ratio (−PSRR)
TEST CONDITIONS
MIN/MAX OVER TEMPERATURE
+25°C(1)
0°C to
+70°C(2)
−40°C to
+85°C(2)
UNITS
MIN/
MAX
TEST
LEVEL
(3)
±6.3
12.4
11.6
90
±6.3
12.8
11.2
88
±6.3
13
11
87
V
V
mA
mA
dB
typ
max
max
min
min
C
A
A
A
A
−40 to
+85
°C
typ
C
125
50(6)
°C/W
°C/W
typ
typ
C
C
+25°C
±6
VS = ±6V, both channels
VS = ±6V, both channels
Input Referred
12
12
95
Thermal Characteristics
Specified Operating Range D Package
Thermal Resistance, qJA
D
SO-8
DTJ
PSO-8
Junction-to-Ambient
(1) Junction temperature = ambient for +25°C tested specifications.
(2) Junction temperature = ambient at low temperature limit; junction temperature = ambient +23°C at high temperature limit for over temperature
tested specifications.
(3) Test levels: (A) 100% tested at +25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and
simulation. (C) Typical value only for information.
(4) Current is considered positive-out-of-node. VCM is the input common-mode voltage.
(5) Tested < 3dB below minimum CMRR specification at ± CMIR limits.
(6) Heat slug soldered to heat spreading plane. This plane should be electrically floating or at VS− voltage.
4
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SBOS249D − JUNE 2003− REVISED APRIL 2004
ELECTRICAL CHARACTERISTICS: VS = +5V
Boldface limits are tested at +25°C.
RF = 402Ω, RL = 100Ω, and G = +2, unless otherwise noted. See Figure 3 for AC performance only.
OPA2613ID, OPA2613IDTJ
TYP
MIN/MAX OVER TEMPERATURE
UNITS
MIN/
MAX
TEST
LEVEL
(3)
MHz
MHz
MHz
MHz
MHz
dB
MHz
V/µs
ns
ns
ns
typ
min
min
min
typ
typ
typ
min
typ
typ
typ
C
B
B
B
C
C
C
B
B
B
B
−57
−76
−75
−86
2.4
2.4
dBc
dBc
dBc
dBc
nV/√Hz
pA/√Hz
dBc
max
max
max
max
max
max
typ
B
B
B
B
B
B
C
89
±1.15
±3.3
−12
−35
±520
±5
88
±1.2
±3.3
−13.5
−35
±750
±7
dB
mV
µV/°C
µA
nA/°C
nA
nA/°C
min
max
max
max
max
max
max
A
A
B
A
B
A
B
1.4
1.4
1.5
V
max
A
3.6
3.6
3.5
V
min
A
85
84
83
dB
min
kΩ pF
MΩ pF
typ
typ
A
A
C
C
3.75
3.7
V
V
min
min
A
A
1.25
1.3
V
V
mA
mA
mA
Ω
min
min
typ
typ
typ
typ
A
A
C
C
C
C
+25°C(1)
0°C to
+70°C(2)
−40°C to
+85°C(2)
75
10
93
69
8
78
68
8
76
47
5.0
78
62
46
5.6
80
64
46
5.7
81
64
−67
−82
−84
−94
1.9
1.7
−80
−60
−79
−78
−89
2.1
2.1
−58
−77
−76
−87
2.2
2.2
95
±0.2
91
±1.0
−6
−11
±50
±300
Least Positive Input Voltage
1.2
Most Positive Input Voltage
3.8
VCM = ±1V
95
VCM = 0
VCM = 0
15 1
5 1.3
Most Positive Output Voltage
No Load
100Ω Load to 2.5V
4.0
3.95
3.85
3.8
3.8
3.75
Least Positive Output Voltage
No Load
100Ω Load to 2.5V
VO = 0, Linear Operation
VO = 0, Linear Operation
Output Shorted to Mid-Supply
G = +2, f = 100kHz
1.0
1.05
+300
−300
±400
0.01
1.15
1.20
1.2
1.25
PARAMETER
AC Performance (see Figure 3)
Small-Signal Bandwidth
Gain-Bandwidth Product
Bandwidth for 0.1dB Gain Flatness
Peaking at a Gain of +1
Large-Signal Bandwidth
Slew Rate
Rise-and-Fall Time
Settling Time to 0.02%
0.1%
Harmonic Distortion
2nd-Harmonic
3rd-Harmonic
Input Voltage Noise
Input Current Noise
Channel-to-Channel Crosstalk
DC Performance(4)
Open-Loop Gain (AOL)
Input Offset Voltage
Average Offset Voltage Drift
Input Bias Current
Average Bias Current Drift (Magnitude)
Input Offset Current
Average Offset Bias Current Drift
TEST CONDITIONS
+25°C
G = +1, VO = 0.1VPP, RF = 0Ω
G = +2, VO = 0.1VPP
G = +10, VO = 0.1VPP
G ≥ 20
G = +2, VO < 0.1VPP
VO < 0.1VPP
G = +2, VO = 2VPP
G = +2, 2V step
G = +2, VO = 0.2V Step
G = +2, VO = 2V Step
G = +2, VO = 2V Step
G = +2, f = 1MHz, VO = 2VPP
RL = 20Ω to VS/2
RL ≥ 500Ω to VS/2
RL = 20Ω to VS/2
RL ≥ 500Ω to VS/2
f > 10kHz
f > 10kHz
f = 1MHz, Input Referred
230
105
12
118
5
2.6
21
60
3.8
63
52
VO = 0V, RL = 100Ω
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
VCM = 0V
Input
Common-Mode Rejection Ratio (CMRR)
Input Impedance
Differential-Mode
Common-Mode
Output
Current Output, Sourcing
Current Output, Sinking
Short-Circuit Current
Closed-Loop Output Impedance
(1) Junction temperature = ambient for +25°C tested specifications.
(2) Junction temperature = ambient at low temperature limit; junction temperature = ambient +23°C at high temperature limit for over temperature
tested specifications.
(3) Test levels: (A) 100% tested at +25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and
simulation. (C) Typical value only for information.
(4) Current is considered positive-out-of-node. VCM is the input common-mode voltage.
(5) Tested < 3dB below minimum CMRR specification at ± CMIR limits.
(6) Heat slug soldered to heat spreading plane. This plane should be electrically floating or at VS− voltage.
5
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SBOS249D − JUNE 2003− REVISED APRIL 2004
ELECTRICAL CHARACTERISTICS: VS = +5V (continued)
Boldface limits are tested at +25°C.
RF = 402Ω, RL = 100Ω, and G = +2, unless otherwise noted. See Figure 3 for AC performance only.
OPA2613ID, OPA2613IDTJ
TYP
PARAMETER
TEST CONDITIONS
MIN/MAX OVER TEMPERATURE
UNITS
MIN/
MAX
TEST
LEVEL
(3)
V
V
mA
mA
dB
typ
max
max
min
typ
C
A
A
A
C
−40 to
+85
°C
typ
C
125
50(6)
°C/W
°C/W
typ
typ
C
C
+25°C
+25°C(1)
0°C to
+70°C(2)
−40°C to
+85°C(2)
12.6
11.0
9.4
12.6
11.3
9.4
12.6
11.5
9.1
Power Supply
Specified Operating Voltage
Maximum Operating Voltage Range
Maximum Quiescent Current
Minimum Quiescent Current
Power-Supply Rejection Ratio (−PSRR)
5
VS = ±6V, both channels
VS = ±6V, both channels
Input Referred
10.5
10.5
95
Thermal Characteristics
Specified Operating Range D Package
Thermal Resistance, qJA
D
SO-8
DTJ
PSO-8
Junction-to-Ambient
(1) Junction temperature = ambient for +25°C tested specifications.
(2) Junction temperature = ambient at low temperature limit; junction temperature = ambient +23°C at high temperature limit for over temperature
tested specifications.
(3) Test levels: (A) 100% tested at +25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and
simulation. (C) Typical value only for information.
(4) Current is considered positive-out-of-node. VCM is the input common-mode voltage.
(5) Tested < 3dB below minimum CMRR specification at ± CMIR limits.
(6) Heat slug soldered to heat spreading plane. This plane should be electrically floating or at VS− voltage.
6
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TYPICAL CHARACTERISTICS: VS = ±6V
At TA = +25°C, G = +2, RF = 402Ω, and RL = 100Ω, unless otherwise noted.
INVERTING SMALL−SIGNAL
FREQUENCY RESPONSE
NONINVERTING SMALL−SIGNAL
FREQUENCY RESPONSE
6
6
VO = 100mVPP
G = +1
3
3
Normalized Gain (dB)
0
−3
−6
−9
G = +8
−12
G = +4
−15
0
G = −1
−3
−6
G = −4
−9
G = −2
−12
G = −8
−15
See Figure 1
See Figure 2
−18
−18
1
10
100
1
500
10
100
Frequency (MHz)
Frequency (MHz)
NONINVERTING LARGE−SIGNAL
FREQUENCY RESPONSE
INVERTING LARGE−SIGNAL
FREQUENCY RESPONSE
12
6
V O = 100mVP P
VO = 100mV PP
9
3
0
6
3
Gain (dB)
Gain (dB)
V O = 500mVP P
V O = 2V PP
0
VO = 1VP P
−3
V O = 5V PP
−6
VO = 500mV PP
−3
VO = 2V PP
−6
VO = 1V PP
−9
V O = 5V PP
−12
−9
−15
See Figure 1
−12
See Figure 2
−18
1
10
100
500
1
10
Frequency (MHz)
Left Scale
200mVPP
Small Signal
−1
Right Scale
0.1
0
−0.1
−2
−0.2
2
Output Voltage (1V/div)
0.2
Output Voltage (100mV/div)
Output Voltage (1V/div)
G = +2V/V
Large Signal
0
3
0.3
1
500
INVERTING PULSE RESPONSE
NONINVERTING PULSE RESPONSE
Left Scale 4V
PP
100
Frequency (MHz)
3
2
500
4VPP
200mVPP
Small Signal
−1
0.3
0.2
Large Signal
1
0
G = −1V/V
Right Scale
0.1
0
−0.1
−2
−0.2
Output Voltage (100mV/div)
Normalized Gain (dB)
G = +2
See Figure 2
See Figure 1
−3
−0.3
Time (50ns/div)
−3
−0.3
Time (50ns/div)
7
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TYPICAL CHARACTERISTICS: VS = ±6V (continued)
At TA = +25°C, G = +2, RF = 402Ω, and RL = 100Ω, unless otherwise noted.
HARMONIC DISTORTION vs FREQUENCY
G = +2
RL = 100Ω
−70
2nd−Harmonic
−80
3rd−Harmonic
−90
Single Channel (see Figure 1)
−100
0.1
1
HARMONIC DISTORTION vs OUTPUT VOLTAGE
−60
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
−60
G = +2
f = 1MHz
RL = 100Ω
−70
2nd−Harmonic
−80
−90
Single Channel (see Figure 1)
−100
0.1
10
1
HARMONIC DISTORTION vs NONINVERTING GAIN
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
2nd−Harmonic
−80
3rd−Harmonic
−90
VO = 2VPP
f = 1MHz
R L = 100Ω
−70
2nd−Harmonic
−80
3rd−Harmonic
−90
Single Channel (see Figure 2)
Single Channel (see Figure 1)
−100
HARMONIC DISTORTION vs INVERTING GAIN
−60
VO = 2VPP
f = 1MHz
RL = 100Ω
−70
−100
1
10
1
10
Gain Magnitude (V/V)
Gain Magnitude (V/V)
HARMONIC DISTORTION vs LOAD RESISTANCE
−60
Harmonic Distortion (dBc)
VO = 2VPP
f = 1MHz
−70
2nd−Harmonic
−80
3rd−Harmonic
−90
Single Channel (see Figure 1)
−100
10
8
10
Output Voltage (VPP)
Frequency (MHz)
−60
3rd−Harmonic
100
Load Resistance (Ω)
1000
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TYPICAL CHARACTERISTICS: VS = ±6V (continued)
At TA = +25°C, G = +2, RF = 402Ω, and RL = 100Ω, unless otherwise noted.
MAXIMUM OUTPUT SWING
vs LOAD RESISTANCE
OUTPUT VOLTAGE AND CURRENT LIMITATIONS
5
6
5
4
3
4
RL = 100Ω
3
2
2
1
0
−1
VO (V)
Output Voltage (V)
6
−2
−3
RL = 50Ω
1
0
−1
RL = 25Ω
−2
−3
−4
−5
−4
−5
See Figure 1
−6
10
100
−6
−400
1000
1W Internal Power
Single Channel
−300
−200
−100
0
100
200
300
400
IO (mA)
Load Resistance (Ω)
INPUT VOLTAGE AND CURRENT NOISE DENSITY
CHANNEL−TO−CHANNEL CROSSTALK
−30
10
Voltage Noise 1.8nV/√Hz
Crosstalk, Input Referred (dB)
Voltage Noise (nV/√Hz)
Current Noise (pA/√Hz)
Input Referred
−40
−50
−60
−70
−80
Current Noise 1.7pA/√Hz
−90
1
102
103
104
105
106
107
1
10
RECOMMENDED RS vs CAPACITIVE LOAD
FREQUENCY RESPONSE vs CAPACITIVE LOAD
Normalized Gain to Capacitive Load (dB)
30
RS(Ω)
20
10
0
1
10
100
Capacitive Load (pF)
100
Frequency (MHz)
Frequency (Hz)
1000
3
CL = 22pF
0
CL = 10pF
CL = 100pF
−3
CL = 47pF
−6
−9
1/2
OPA2613
−12
402Ω
−15
402Ω
RS
CL
1kΩ
1kΩ is optional.
−18
1
10
100
500
Frequency (MHz)
9
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TYPICAL CHARACTERISTICS: VS = ±6V (continued)
At TA = +25°C, G = +2, RF = 402Ω, and RL = 100Ω, unless otherwise noted.
OPEN−LOOP GAIN AND PHASE
CMRR AND PSRR vs FREQUENCY
20 log (AOL)
−PSRR
60
40
80
20
1k
10k
100k
1M
10M
60
−90
40
−120
20
−150
0
−180
−20
100
100M
−210
1k
10k
100k
CLOSED−LOOP OUTPUT IMPEDANCE
vs FREQUENCY
10M
100M
1G
VIDEO DIFFERENTIAL GAIN/DIFFERENTIAL PHASE
100
0.14
0.28
G = +2
VS = ±5V
0.12
10
Differential Gain (%)
1
0.1
0.01
dφ, Negative Video
0.24
0.10
0.20
dG , Positive Video
0.08
0.16
0.06
0.12
0.04
0.08
dφ, Positive Video
0.001
0.02
0.0001
0.04
dG, Negative Video
0
10k
100k
1M
10M
100M
1
2
3
4
5
Frequency (Hz)
4
2
2
1
0
0
−2
−1
−4
−2
−6
−8
−10
−3
G = +2
RL = 100Ω
See Figure 1
−4
−5
Time (100ns/div)
Output Voltage (2V/div)
Output
4
Input
6
3
Input Voltage (1V/div)
6
8
0
9
10
INVERTING OVERDRIVE RECOVERY
8
5
Input
8
7
Video Loads
NONINVERTING OVERDRIVE RECOVERY
10
6
G = −1
RL = 100Ω
8
6
4
4
2
2
0
0
−2
−2
−4
−4
−6
−8
−6
Output
See Figure 2
Time (100ns/div)
−8
Input Voltage (2V/div)
Output Impedance Magnitude (Ω)
1M
Frequency (Hz)
Frequency (Hz)
Output Voltage (2V/div)
−60
∠ AOL
Differential Phase (_)
100
Open−Loop Phase (_ )
CMRR
0
10
−30
100
100
80
0
120
+PSRR
Open−Loop Gain (dB)
Common−Mode Rejection Ratio (dB)
Power−Supply Rejection Ratio (dB)
120
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TYPICAL CHARACTERISTICS: VS = ±6V (continued)
At TA = +25°C, G = +2, RF = 402Ω, and RL = 100Ω, unless otherwise noted.
SUPPLY AND OUTPUT CURRENT vs TEMPERATURE
TYPICAL DC DRIFT OVER TEMPERATURE
0
0
Input Offset Voltage (VIO)
−5
−0.5
Inverting Bias Current (IB)
−25
0
25
50
75
100
−10
125
290
12.2
Left Scale
12.1
280
Supply Current
Right Scale
270
12.0
260
11.9
250
−50
−25
0
Ambient Temperature (_C)
25
50
75
100
Supply Current (0.1mA/div)
(10 Times Input Offset Current) 10 x IOS
Sourcing and Sinking Current
Output Current (10mA/div)
5
0.5
11.8
125
Ambient Temperature (_ C)
COMMON−MODE INPUT RANGE AND OUTPUT SWING
vs SUPPLY VOLTAGE
6
RL = 100Ω
5
Voltage Range (V)
Input Offset Voltage (mV)
Input Bias and Offset Current (µA)
10
1
−1
−50
12.3
300
−V Input Voltage
4
3
−Output Voltage
2
+V Input Voltage
1
+Output Voltage
0
2.5
3.0
3.5
4.0
4.5
5.0
5.5
6
Supply Voltage (±V)
11
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TYPICAL CHARACTERISTICS: VS = ±6V, Differential Configuration
At TA = +25°C, Differential Gain = 4, RF = 402Ω, and RL = 70Ω, unless otherwise noted. See Figure 5 for AC performance only.
DIFFERENTIAL SMALL−SIGNAL
FREQUENCY RESPONSE
DIFFERENTIAL LARGE−SIGNAL
FREQUENCY RESPONSE
15
3
R L = 70Ω
R L = 70Ω
G D = +4
GD = +1
0
12
GD = 2
V O = 0.2V PP
Gain (dB)
Normalized Gain (dB)
VO = 200mV PP
−3
−6
9
VO = 1VP P
6
V O = 2VP P
3
G D = +4
See Figure 5
0
10
1
100
200
1
10
Frequency (MHz)
DIFFERENTIAL DISTORTION vs LOAD RESISTANCE
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
DIFFERENTIAL DISTORTION vs FREQUENCY
−55
GD = +4
f = 1MHz
VO = 2VPP
3rd−Harmonic
−90
2nd−Harmonic
−95
GD = 4
RL = 70Ω
VO = 2VPP
−65
3rd−Harmonic
−75
−85
2nd−Harmonic
−95
−105
See Figure 5
See Figure 5
10
−115
100
1k
0.1
1
Resistance (Ω)
Frequency (MHz)
DIFFERENTIAL DISTORTION
vs OUTPUT VOLTAGE
−70
GD = 4
RL = 70Ω
f = 1MHz
Harmonic Distortion (dBc)
−75
−80
3rd−Harmonic
−85
−90
−95
2nd−Harmonic
−100
See Figure 5
−105
0.1
1
Output Voltage Swing (VPP)
12
100
Frequency (MHz)
−85
−100
V O = 5V PP
See Figure 5
GD = +8
−9
10
20
10
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TYPICAL CHARACTERISTICS: VS = +5V
At TA = +25°C, G = +2, RF = 402Ω, and RL = 100Ω, unless otherwise noted.
INVERTING SMALL−SIGNAL
FREQUENCY RESPONSE
NONINVERTING SMALL−SIGNAL
FREQUENCY RESPONSE
VO = 100mVPP
RL = 100Ω to VS/2
3
G = +1
VO = 100mVPP
RL = 100Ω to VS/2
Normalized Gain (dB)
0
G = +2
−3
−6
G = +8
See Figure 3
−9
1
G = −1
0
G = −2
−3
G = −4
−6
G = −8
G = +4
See Figure 4
−9
10
100
1
500
10
100
Frequency (MHz)
Frequency (MHz)
NONINVERTING LARGE−SIGNAL
FREQUENCY RESPONSE
INVERTING LARGE−SIGNAL
FREQUENCY RESPONSE
9
3
G = +2
G = −1
R L = 100Ω to VS /2
VO = 0.1V PP
R L = 100Ω to VS /2
6
VO = 0.1V P P
0
VO = 0.5V PP
VO = 0.5V P P
Gain (dB)
Gain (dB)
3
0
VO = 1V PP
−3
1
−3
−6
V O = 1V PP
−9
V O = 2VP P
See Figure 3
−6
10
100
300
1
10
2VPP
Large Signal
3.3
2.9
2.5
200mVPP
Small Signal
4.5
2.9
4.1
2.8
3.7
2.7
Right Scale
2.6
2.5
2.1
2.4
1.7
2.3
1.3
0.9
0.5
See Figure 3
Time (50ns/div)
Output Voltage (mV)
Output Voltage (1V/div)
Left Scale
300
INVERTING PULSE RESPONSE
3.0
Output Voltage (mV)
G = +2V/V
RL = 100Ωto VS/2
4.1
100
Frequency (MHz)
NONINVERTING PULSE RESPONSE
4.5
VO = 2VP P
See Figure 4
−12
Frequency (MHz)
3.7
300
Left Scale
2.5
2VPP
200mVPP
Small Signal
3.0
2.9
2.8
Large Signal
3.3
2.9
G = −1V/V
RL = 100Ωto VS/2
2.7
Right Scale
2.6
2.5
2.1
2.4
1.7
2.3
2.2
1.3
2.2
2.1
0.9
2.0
0.5
Output Voltage (mV)
Normalized Gain (dB)
3
2.1
See Figure 4
2.0
Time (50ns/div)
13
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SBOS249D − JUNE 2003− REVISED APRIL 2004
TYPICAL CHARACTERISTICS: VS = +5V (continued)
At TA = +25°C, G = +2, RF = 402Ω, and RL = 100Ω, unless otherwise noted.
HARMONIC DISTORTION vs FREQUENCY
VO = 2VPP
G = +2
RL = 100Ω to VS/2
−70
f = 1MHz
RL = 100Ω to VS/2
2nd−Harmonic
−80
3rd−Harmonic
−90
Single Channel
(see Figure 3)
−100
0.1
HARMONIC DISTORTION vs OUTPUT VOLTAGE
−70
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
−60
1
−80
2nd−Harmonic
−90
3rd−Harmonic
−100
−110
0.1
10
1
HARMONIC DISTORTION vs INVERTING GAIN
HARMONIC DISTORTION vs NONINVERTING GAIN
−50
VO = 2VPP
f = 1MHz
RL = 100Ωto VS/2
−60
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
−50
2nd−Harmonic
−70
−80
3rd−Harmonic
−90
Single Channel (see Figure 3)
−100
1
VO = 2VPP
f = 1MHz
RL = 100Ω to VS/2
−60
2nd−Harmonic
−70
−80
3rd−Harmonic
−90
Single Channel (see Figure 4)
−100
1
10
10
Gain Magnitude (V/V)
Gain Magnitude (V/V)
HARMONIC DISTORTION vs LOAD RESISTANCE
−60
VO = 2VPP
f = 1MHz
G = +2
RL to VS/2
Harmonic Distortion (dBc)
2nd−Harmonic
−70
−80
3rd−Harmonic
−90
Single Channel (see Figure 3)
−100
10
100
Load Resistance (Ω)
14
5
Output Voltage (VPP)
Frequency (MHz)
1000
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TYPICAL CHARACTERISTICS: VS = +5V, Differential Configuration
At TA = +25°C, GD = 4, RF = 402Ω, and RL = 70Ω, unless otherwise noted.
+5V
DIFFERENTIAL SMALL−SIGNAL
FREQUENCY RESPONSE
806Ω
3
RL = 70Ω
806Ω
Normalized Gain (dB)
1/2
OPA2613
RF
402Ω
0.01µF
RG
VI
RL
RF
402Ω
0.01µF
VI
0.01µF
806Ω
GD = 1
0
GD = 2
−3
−6
GD = 4
1/2
OPA2613
806Ω
GD = 8
−9
2RF
GD = 1 +
RG
1
10
100
200
Frequency (MHz)
DIFFERENTIAL LARGE−SIGNAL
FREQUENCY RESPONSE
DIFFERENTIAL DISTORTION vs LOAD RESISTANCE
−85
15
RL = 70Ω
GD = 4
Harmonic Distortion (dBc)
12
Gain (dB)
VO = 0.2VPP
9
VO = 1VPP
6
VO = 2VPP
3
VO = 5VPP
−90
2nd−Harmonic
−95
−100
0
1
10
100
10
200
100
DIFFERENTIAL DISTORTION vs FREQUENCY
DIFFERENTIAL DISTORTION vs OUTPUT VOLTAGE
−85
GD = 4V/V
RL = 70Ω
VO = 2VPP
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
−85
1k
Resistance (Ω)
Frequency (MHz)
−75
GD = +4
RL = 70Ω
VO = 2VPP
f = 1MHz
3rd−Harmonic
3rd−Harmonic
−95
2nd−Harmonic
−105
GD = 4V/V
R L = 70Ω
f = 1MHz
3rd−Harmonic
2nd−Harmonic
−90
−95
−115
0.1
1
Frequency (MHz)
2
0.1
1
10
Output Voltage Swing (VPP )
15
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APPLICATION INFORMATION
WIDEBAND VOLTAGE-FEEDBACK OPERATION
The OPA2613 gives the exceptional AC performance of a
wideband voltage-feedback op amp with a highly linear,
high-power output stage. Requiring only 6mA/ch
quiescent current, the OPA2613 swings to within 1.0V of
either supply rail and delivers in excess of 280mA at room
temperature. This low-output headroom requirement,
along with supply voltage independent biasing, gives
remarkable single (+5V) supply operation. The OPA2613
delivers greater than 20MHz bandwidth driving a 2VPP
output into 100Ω on a single +5V supply. Previous boosted
output stage amplifiers typically suffer from very poor
crossover distortion as the output current goes through
zero. The OPA2613 achieves exceptional power gain with
much better linearity. Figure 1 shows the DC-coupled,
gain of +2, dual power-supply circuit configuration used as
the basis of the ±6V Electrical and Typical Characteristics.
For test purposes, the input impedance is set to 50Ω with
a resistor to ground; and the output impedance is set to
50Ω with a series output resistor. Voltage swings reported
in the electrical characteristics are taken directly at the
input and output pins, whereas load powers (dBm) are
defined at a matched 50Ω load. For the circuit of Figure 1,
the total effective load is 100Ω || 804Ω = 89Ω.
0.1µF
+6V
+VS
6.8µF
+
50Ω Source
VI
50Ω
VO
1/2
OPA2613
50Ω
50Ω Load
RF
402Ω
RG
402Ω
6.8µF
0.1µF
+
−VS
−6V
Figure 1. DC-Coupled, G = +2, Bipolar Supply,
Specification and Test Circuit
16
Figure 2 shows the DC-coupled, bipolar supply circuit
configuration used as the basis for the Inverting Gain
−1V/V Typical Characteristics. Key design considerations
of the inverting configuration are developed in the Inverting
Amplifier Operation section.
+5V
208Ω
50Ω
Source
RF
402Ω
Power−Supply
decoupling
not shown.
1/2
OPA2613
VO
50Ω
50ΩLoad
−5V R
F
402Ω
VI
RM
57.6Ω
Figure 2. DC-Coupled, G = −1, Bipolar Supply,
Specification and Test Circuit
Figure 3 shows the AC-coupled, gain of +2, single-supply
circuit configuration used as the basis of the +5V Electrical
and Typical Characteristics. Though not a “rail-to-rail”
design, the OPA2613 requires minimal input and output
voltage headroom compared to other very wideband
voltage-feedback op amps. It will deliver a 2.6VPP output
swing on a single +5V supply with greater than 20MHz
bandwidth. The key requirement of broadband singlesupply operation is to maintain input and output signal
swings within the usable voltage ranges at both the input
and the output. The circuit of Figure 3 establishes an input
midpoint bias using a simple resistive divider from the +5V
supply (two 806Ω resistors). The input signal is then
AC-coupled into this midpoint voltage bias. The input
voltage can swing to within 1.4V of either supply pin, giving
a 2.2VPP input signal range centered between the supply
pins. The input impedance matching resistor (57.6Ω) used
for testing is adjusted to give a 50Ω input match when the
parallel combination of the biasing divider network is
included. The gain resistor (RG) is AC-coupled, giving the
circuit a DC gain of +1which puts the input DC bias
voltage (2.5V) on the output as well. Again, on a single +5V
supply, the output voltage can swing to within 1.1V of either
supply pin while delivering more than 100mA output
current. A demanding 100Ω load to a midpoint bias is used
in this characterization circuit. The new output stage used
in the OPA2613 can deliver large bipolar output currents
into this midpoint load with minimal crossover distortion,
as shown by the +5V supply, harmonic distortion plots.
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the OPA2613. Each has its advantages and disadvantages. Figure 5 shows a basic starting point for
noninverting input differential I/O applications.
+5V
+VS
+
0.1µF
6.8µF
806Ω
+VCC
0.1µF
VI
57.6Ω
806Ω
1/2
OPA2613
VO
100Ω
1/2
VS /2
O P A2613
RF
402Ω
RF
402Ω
RG
402Ω
VI
RG
268Ω
0.1µF
Figure 3. AC-Coupled, G = +2, Single-Supply,
Specification and Test Circuit
The last configuration used as the basis of the +5V
Electrical and Typical Characteristics is shown in Figure 4.
Design considerations for this inverting, bipolar supply
configuration are covered either in single-supply
configuration (as shown in Figure 3) or in the Inverting
Amplifier Operation section.
+5V
0.1µF
806Ω
0.1µF
RG
0.1µF 402Ω
806Ω
1/2
OPA2613
Power−Supply
decoupling not
shown.
VO
+
6.8µF
RF
402Ω
1/2
O P A2613
−VCC
Figure 5. Noninverting Differential I/O Amplifier
This approach provides for a source termination
impedance that is independent of the signal gain. For
instance, simple differential filters may be included in the
signal path right up to the noninverting inputs without
interacting with the gain setting. The differential signal gain
for the circuit of Figure 5 is:
AD + 1 ) 2
100Ω
VS /2
RF
402Ω
VI
RM
57.6Ω
Figure 4. AC-Coupled, G = −1, Single-Supply,
Specification and Test Circuit
DIFFERENTIAL INTERFACE APPLICATIONS
Dual op amps are particularly suitable to differential input
to differential output applications. Typically, these fall into
either Analog-to-Digital Converter (ADC) input interface or
line driver applications. Two basic approaches to
differential I/O are noninverting or inverting configurations.
Since the output is differential, the signal polarity is
somewhat meaningless—the noninverting and inverting
terminology applies here to where the input is brought into
VO
RF
RG
(1)
Since the OPA2613 is a voltage-feedback (VFB) amplifier,
its bandwidth is principally controlled by the noise gain.
The equivalent noise gain for Figure 5 is:
1)2
402W + 4VńV
268W
(2)
Various combinations of single-supply or AC-coupled gain
can also be delivered using the basic circuit of Figure 5.
Common-mode bias voltages on the two noninverting
inputs pass on to the output with a gain of 1 since an equal
DC voltage at each inverting node creates no current
through RG. This circuit does show a common-mode gain
of 1 from input to output. The source connection should
either remove this common-mode signal if undesired
(using an input transformer can provide this function), or
the common-mode voltage at the inputs can be used to set
the output common-mode bias. If the low common-mode
rejection of this circuit is a problem, the output interface
may also be used to reject that common-mode. For
instance, most modern differential input ADCs reject
common-mode signals very well, while a line driver
application through a transformer will also remove the
common-mode signal through to the line.
17
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SINGLE-SUPPLY ADSL UPSTREAM DRIVER
Figure 6 shows an example of a single-supply ADSL
upstream driver. The dual OPA2613 is configured as a
differential gain stage to provide signal drive to the primary
of the transformer (here, a step-up transformer with a turns
ratio of 1:2). The main advantage of this configuration is
the cancellation of all even harmonic distortion products.
Another important advantage for ADSL is that each
amplifier needs only to swing half of the total output
required driving the load.
receiver. The value of these resistors (RM) is a function of
the line impedance and the transformer turns ratio (n),
given by the following equation:
Z LINE
2n2
RM +
(4)
LINE DRIVER HEADROOM MODEL
The first step in a transformer-coupled, twisted-pair driver
design is to compute the peak-to-peak output voltage from
the target specifications. This is done using the following
equations:
2
P L + 10
+12V
1/2
OPA2613
AFE
2VPP
Max
Assumed
+6.3V
0.1µF
IP = 150mA
RM
12.5Ω
1:n
RF
1kΩ
1kΩ
1kΩ
RG
308Ω
V RMS +
15VPP
RM
12.5Ω
1/2
OPA2613
V LPP + 2
IP = 150mA
Figure 6. Single-Supply ADSL Upstream Driver
The analog front-end (AFE) signal is AC-coupled to the
driver, and the noninverting input of each amplifier is
biased slightly above the mid-supply voltage (+6.3V in this
case). In addition to providing the proper biasing to the
amplifier, this approach also provides a high-pass filtering
with a corner frequency, set here at 1.6kHz. As the
upstream signal bandwidth starts at 26kHz, this high-pass
filter does not generate any problems and has the
advantage of filtering out unwanted lower frequencies.
The input signal is amplified with a gain set by the following
equation:
2
RF
RG
PL
10 10
V RMS + CF
(6)
V RMS (7)
CF
VRMS
(8)
The two back-termination resistors (12.5Ω each) added at
each input of the transformer make the impedance of the
modem match the impedance of the phone line, and also
provide a means of detecting the received signal for the
CF
Ǹ(1mW)
RL
PL
10 10 (9)
This VLPP is usually computed for a nominal line
impedance and may be taken as a fixed design target.
The next step for the driver is to compute the individual
amplifier output voltage and currents as a function of VPP
on the line and transformer turns ratio. As the turns ratio
changes, the minimum allowed supply voltage changes
along with it. The peak current in the amplifier output is
given by:
"I P + 1
2
2
V LPP
n
1
4R M
(10)
With VLPP as defined in Equation 8, and RM as defined in
Equation 4 and shown in Figure 7.
RM
(3)
With RF = 1kΩ and RG = 308Ω, the gain for this differential
amplifier is 7.5. This gain boosts the AFE signal, assumed
to be a maximum of 2VPP, to a maximum of 15VPP.
18
RL
with VLPP: peak-to-peak voltage at the load.
Consolidating Equations 4 through 7 allows expressing
the required peak-to-peak voltage at the load as a function
of the crest factor, the load impedance, and the power at
the load. Thus,
V LPP + 2
GD + 1 )
(5)
with VP peak voltage at the load and CF Crest Factor.
100Ω
1µF
Ǹ(1mW)
V P + Crest Factor
ZLINE
RF
1kΩ
20Ω
VRMS
(1mW) RL
With PL power and VRMS voltage at the load, and RL load
impedance, this gives the following:
20Ω
0.1µF
log
Vpp =
2VLpp
n
VLpp
n
1:n
RL
VLpp
RM
Figure 7. Driver Peak Output Voltage
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With the previous information available, it is now possible
to select a supply voltage and the turns ratio desired for the
transformer as well as calculate the headroom for the
OPA2613.
The model (shown in Figure 8) can be described with the
following set of equations:
OPA2613 holds a relatively constant quiescent current
versus supply voltage—giving a power contribution that is
simply the quiescent current times the supply voltage used
(the supply voltage will be greater than the solution given
in Equation 12). The total output stage power may be
computed with reference to Figure 9.
1. First, as available output swing:
V PP + VCC * (V1 ) V2) * I P
+VCC
(R 1 ) R 2) (11)
IAVG =
IP
CF
2. Or as required supply voltage:
V CC + VPP ) (V1 ) V2) ) I P
(R 1 ) R 2) (12)
RT
The minimum supply voltage for a power and load
requirement is given by Equation 11.
+VCC
Figure 9. Output Stage Power Model
R1
V1
VO
IP
V2
R2
Figure 8. Line Driver Headroom Model
V1, V2, R1, and R2 are given in Table 1 for both +12V and
+5V operation.
Table 1. Line Driver Headroom Model Values
V1
R1
V2
R2
+5V
1.0V
2Ω
1.0V
5.5Ω
+12V
1.0V
2Ω
1.0V
5.5Ω
TOTAL DRIVER POWER FOR xDSL
APPLICATIONS
The total internal power dissipation for the OPA2613 in an
xDSL line driver application will be the sum of the
quiescent power and the output stage power. The
The two output stages used to drive the load of Figure 7
can be seen as an H-Bridge in Figure 9. The average
current drawn from the supply into this H-Bridge and load
will be the peak current in the load given by Equation 10
divided by the crest factor (CF) for the xDSL modulation.
This total power from the supply is then reduced by the
power in RT to leave the power dissipated internal to the
drivers in the four output stage transistors. That power is
simply the target line power used in Equation 5 plus the
power lost in the matching elements (RM). In the examples
here, a perfect match is targeted giving the same power in
the matching elements as in the load. The output stage
power is then set by Equation 13.
P OUT +
IP
CF
V CC * 2P L
(13)
The total amplifier power is then:
P TOT + I q
VCC )
IP
CF
V CC * 2P L
(14)
For the ADSL CPE upstream driver design of Figure 6, the
peak current is 150mA for a signal that requires a crest
factor of 5.33 with a target line power of 13dBm into 100Ω
(20mW). With a typical quiescent current of 12mA and a
nominal supply voltage of +12V, the total internal power
dissipation for the solution of Figure 6 will be:
PTOT + 12mA(12V) ) 150mA (12V) * 2(20mW) + 400mW
5.33
(15)
19
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DESIGN-IN TOOLS
+6V
Power−supply
decoupling not
shown.
DEMONSTRATION BOARDS
A PC board is available to assist in the initial evaluation of
circuit performance using the OPA2613 in its two package
styles. It is available, free, as an unpopulated PC board
delivered with descriptive documentation. The summary
information for this unit is shown in Table 2.
50Ω Load
1/2
OPA2613
50Ω
Source
Check the TI web site (www.ti.com) to request this board.
RG
200Ω
VO
50Ω
RF
402Ω
VI
Table 2. Demonstration Board Ordering
Information
PRODUCT
PACKAGE
DEMO BOARD
NUMBER
ORDERING
NUMBER
OPA2613ID
SO-8
DEM-OPA268XU
SBOU003
MACROMODELS AND APPLICATIONS
SUPPORT
Computer simulation of circuit performance using SPICE
is often useful when analyzing the performance of analog
circuits and systems. This is particularly true for video and
RF amplifier circuits where parasitic capacitance and
inductance can have a major effect on circuit performance.
A SPICE model for the OPA2613 is available through the
TI web site (www.ti.com). This model does a good job of
predicting small-signal AC and transient performance
under a wide variety of operating conditions, but does not
do as well in predicting the harmonic distortion or video
dG/dP characteristics. This model does not attempt to
distinguish between the package types in small-signal AC
performance, nor does it attempt to simulate channel-tochannel coupling.
INVERTING AMPLIFIER OPERATION
As the OPA2613 is a general-purpose, wideband
voltage-feedback op amp, most of the familiar op amp
application circuits are available to the designer.
Wideband inverting operation is particularly suited to the
OPA2613. Figure 10 shows a typical inverting
configuration where the I/O impedances and signal gain
from Figure 1 are retained in an inverting circuit
configuration.
20
VO
RM
66.7Ω
VI
=−
RF
RG
= −2
− 6V
Figure 10. Inverting Gain of −1 with Impedance
Matching
In the inverting configuration, two key design
considerations must be noted. The first is that the gain
resistor (RG) becomes part of the input impedance. If input
impedance matching is desired (which is beneficial
whenever the signal is coupled through a cable, twistedpair, long PC board trace, or other transmission line
conductor), it is normally necessary to add an additional
matching resistor to ground. RG, by itself, is not normally
set to the required input impedance since its value, along
with the desired gain, will determine an RF, which may be
non-optimal from a frequency response standpoint. The
total input impedance for the source becomes the parallel
combination of RG and RM.
The second major consideration, touched on in the
previous paragraph, is that the signal source impedance
becomes part of the noise gain equation and has an effect
on the bandwidth. In the example of Figure 10, the RM
value combines in parallel with the external 50Ω source
impedance, yielding an effective driving impedance of
50Ω || 66.7Ω = 28.6Ω. This impedance is added in series
with RG for calculating the noise gainwhich gives
NG = 2.76. Note that the noninverting input in this bipolar
supply inverting application is connected to ground
through a 146Ω resistor. It is often suggested that an
additional resistor be connected to ground on the
noninverting input to achieve bias current error
cancellation at the output.
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OUTPUT CURRENT AND VOLTAGE
The OPA2613 provides output voltage and current
capabilities that are unsurpassed in a low-cost dual
monolithic op amp. Under no-load conditions at 25°C, the
output voltage typically swings closer than 1V to either
supply rail; tested at +25°C, swing limit is within 1.1V of
either rail. Into a 12Ω load (the minimum tested load), it
delivers more than ±280mA continuous output current.
The specifications described previously, though familiar in
the industry, consider voltage and current limits separately.
In many applications, it is the voltage times current (or V-I
product) that is more relevant to circuit operation. Refer to
the Output Voltage and Current Limitations plot in the
Typical Characteristics. The X and Y axes of this graph
show the zero-voltage output current limit and the
zero-current output voltage limit, respectively. The four
quadrants give a more detailed view of the OPA2613
output drive capabilities, noting that the graph is bounded
by a safe operating area of 1W maximum internal power
dissipation (in this case, for one channel only).
Superimposing resistor load lines onto the plot shows that
the OPA2613 can drive +4.8 and −4.1 into 25Ω without
exceeding the output capabilities or the 1W dissipation
limit. A 100Ω load line (the standard test circuit load)
shows the full ±4.9V output swing capability, as shown in
the Electrical Characteristics tables. The minimum
specified output voltage and current over temperature are
set by worst-case simulations at the cold temperature
extreme. Only at cold startup will the output current and
voltage decrease to the numbers shown in the Electrical
Characteristics tables. As the output transistors deliver
power, the junction temperatures increase, decreasing the
VBEs (increasing the available output voltage swing), and
increasing the current gains (increasing the available
output current). In steady-state operation, the available
output voltage and current will always be greater than that
shown in the over-temperature specifications, since the
output stage junction temperatures will be higher than the
minimum specified operating ambient.
DRIVING CAPACITIVE LOADS
One of the most demanding and yet very common load
conditions for an op amp is capacitive loading. Often, the
capacitive load is the input of an ADCincluding
additional external capacitance that may be recommended to improve the ADC linearity. A high-speed, high
open-loop gain amplifier like the OPA2613 can be very
susceptible to decreased stability and closed-loop
response peaking when a capacitive load is placed directly
on the output pin. When the amplifier open-loop output
resistance is considered, this capacitive load introduces
an additional pole in the signal path that can decrease the
phase margin. Several external solutions to this problem
have been suggested.
When the primary considerations are frequency response
flatness, pulse response fidelity, and/or distortion, the
simplest and most effective solution is to isolate the
capacitive load from the feedback loop by inserting a
series isolation resistor between the amplifier output and
the capacitive load. This does not eliminate the pole from
the loop response, but rather shifts it and adds a zero at a
higher frequency. The additional zero acts to cancel the
phase lag from the capacitive load pole, thus increasing
the phase margin and improving stability. The Typical
Characteristics show the Recommended RS vs Capacitive
Load and the resulting frequency response at the load.
Parasitic capacitive loads greater than 2pF can begin to
degrade the performance of the OPA2613. Long PC board
traces, unmatched cables, and connections to multiple
devices can easily cause this value to be exceeded.
Always consider this effect carefully, and add the
recommended series resistor as close as possible to the
OPA2613 output pin (see the Board Layout Guidelines
section).
The very high output current and unity gain stability for the
OPA2613 can be used to drive large capacitive loads with
moderate slew rates. An example is shown in Figure 11
where a 5000pF load cap is driven with a 1MHz square
wave to give a ±5V swing. The supplies were slightly
increased to give more headroom for the charging current
through the 2Ω isolation resistor.
+6.2V
VI
±2.5V
1MHz Square
Wave
Input
1/2
OPA2613
Supply decoupling
not shown.
2Ω
VO
5000pF
402Ω
− 6.2V
402Ω
Figure 11. Large Capacitive Load Driver
21
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Figure 12 shows a comparison of 2 • Input voltage to the
capacitor voltage. The transition time is set by the 70V/µs
slew rate for the OPA2613. For this controlled dV/dT, the
charging current into the 5000pF load will be given by:
Slew Rate = IP/C
Solving for IP gives:
I P + 5000pF
70Vńms + 350mA peak current (16)
Input and Output Voltage
6
5
Capacitor Voltage
4
3
2X Input Voltage
2
NOISE PERFORMANCE
Wideband voltage-feedback op amps generally have a
lower output noise than comparable current-feedback op
amps. The OPA2613 offers an excellent balance between
voltage and current noise terms to achieve low output
noise. The input voltage noise (1.8nV/√Hz) is lower than
most unity-gain stable, wideband voltage-feedback op
amps. The op amp input voltage noise and the two input
current noise terms combine to give low output noise
under a wide variety of operating conditions. Figure 13
shows the op amp noise analysis model with all the noise
terms included. In this model, all noise terms are taken to
be noise voltage or current density terms in either nV/√Hz
or pA/√Hz.
1
0
−1
70V/µs Slew Rate
ENI
−2
−3
−4
1/2
OPA2613
RS
−5
−6
Time (100ns/div)
ERS
Figure 12. Large-Signal Capacitive Load Drive
At these larger capacitive loads, very low series R will
maintain stabilitybut some R is always required.
The OPA2613 provides good distortion performance into
a 100Ω load on ±6V supplies. Generally, until the
fundamental signal reaches high frequency or power
levels, the 2nd-harmonic dominates the distortion with a
negligible 3rd-harmonic component. Focusing then on the
2nd-harmonic, increasing the load impedance improves
distortion directly. Remember that the total load includes
the feedback networkin the noninverting configuration
(see Figure 1), this is the sum of RF + RG, whereas in the
inverting configuration, it is just RF. Also, providing an
additional supply decoupling capacitor (0.01µF) between
the supply pins (for bipolar operation) improves the
2nd-order distortion slightly (3dB to 6dB).
In most op amps, increasing the output voltage swing
increases harmonic distortion directly. The Typical
Characteristics show the 2nd-harmonic increasing at a
little less than the expected 2x rate whereas the
3rd-harmonic increases at a little less than the expected 3x
rate. Where the test power doubles, the difference
between it and the 2nd-harmonic decreases less than the
expected 6dB, whereas the difference between it and the
3rd-harmonic decreases by less than the expected 12dB.
Operating differentially will suppress the 2nd-order
harmonics below the 3rd.
Operating as a differential I/O stage will also suppress the
2nd-harmonic distortion.
22
RF
√4kTRS
√4kTRF
IBI
RG
4kT
RG
DISTORTION PERFORMANCE
EO
IBN
4kT = 1.6E −20J
at 290_K
Figure 13. Op Amp Noise Analysis Model
The total output spot noise voltage can be computed as the
square root of the sum of all squared output noise voltage
contributors. Equation 17 shows the general form for the
output noise voltage using the terms given in Figure 13.
EO +
Ǹǒ
E NI ) ǒI BN
RSǓ ) 4kTRS
2
2
Ǔ
NG 2 ) ǒI BI
2
RFǓ ) 4kTRFNG
(17)
Dividing this expression by the noise gain (NG = (1 + RF/RG))
gives the equivalent input-referred spot noise voltage at the
noninverting input, as shown in Equation 18.
EN +
Ǹ
ǒ
2
E NI ) I BN
R
Ǔ
S
2
) 4kTR )
S
ǒ
I BI
Ǔ
RF
NG
2
)
4kTR F
NG
(18)
Evaluating these two equations for the OPA2613 circuit
and component values (see Figure 1) gives a total output
spot noise voltage of 6.34nV/√Hz and a total equivalent
input spot noise voltage of 3.2nV/√Hz. This total input
referred spot noise voltage is higher than the 1.8nV/√Hz
specification for the op amp voltage noise alone. This
reflects the noise added to the output by the inverting
current noise times the feedback resistor.
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DIFFERENTIAL NOISE PERFORMANCE
As the OPA2613 is used as a differential driver in xDSL
applications, it is important to analyze the noise in such a
configuration. Figure 14 shows the op amp noise model for
the differential configuration.
IN
Evaluating these equations for the OPA2613 ADSL circuit
and component values of Figure 6 gives a total output spot
noise voltage of 23.3nV/√Hz and a total equivalent input
spot noise voltage of 3.2nV/√Hz.
In order to minimize the output noise due to the
noninverting input bias current noise, it is recommended to
keep the noninverting source impedance as low as
possible.
DC ACCURACY AND OFFSET CONTROL
Driver
EN
RS
IN
ERS
√4kTRF
RF
√4kTRS
RG
EO2
√4kTRG
√4kTRF
RF
IN
EN
RS
IN
ERS
√4kTRS
Figure 14. Differential Op Amp Noise Analysis
Model
The OPA2613 can provide excellent DC signal accuracy
due to its high open-loop gain, high common-mode
rejection, high power-supply rejection, and low input offset
voltage and bias current offset errors. To take full
advantage of the low input offset voltage (±1.0mV
maximum at 25°C), careful attention to input bias current
cancellation is also required. The high-speed input stage
for the OPA2613 has relatively high input bias current (6µA
typical into the pins) but with a very close match between
the two input currents, typically 50nA input offset current.
The total output offset voltage may be reduced
considerably by matching the source impedances looking
out of the two inputs. For example, one way to add bias
current cancellation to the circuit of Figure 1 would be to
insert a 175Ω series resistor into the noninverting input
from the 50Ω terminating resistor. If the 50Ω source
resistor is DC-coupled, this will increase the source
impedance for the noninverting input bias current to 200Ω.
Since this is now equal to the impedance looking out of the
inverting input (RF || RG), the circuit will cancel the bias
current effects, leaving only the offset current times the
feedback resistor as a residual DC error term at the output.
Evaluating the configuration of Figure 1 adding a 175Ω in
series with the noninverting input pin, using worst-case
+25°C input offset voltage and the two input bias currents,
gives a worst-case output offset range equal to:
VOFF = ± (NG × VOS(MAX)) ± (IOS × RF)
As a reminder, the differential gain is expressed as:
GD + 1 )
2
where NG = noninverting signal gain
= ± (2 × 1.0mV) ± (402Ω × 300nA)
RF
RG
(19)
VOFF = ±2.12mV
The output noise can be expressed as shown below:
(20)
e
O
+
Ǹ
2
G
2
D
ǒ
ǒ
e 2) i
N
N
R
S
Ǔ
2
) 4kTR
S
Ǔ
2
) 2ǒi R Ǔ ) 2ǒ4kTR G Ǔ
I F
F D
Dividing this expression by the differential noise gain
(G D = (1 + 2R F /R G )) gives the equivalent input referred
spot noise voltage at the noninverting input, as shown in
Equation 21.
ei +
Ǹ
2
(21)
ǒ
eN 2 ) ǒi N
Ǔ ǒ Ǔ ǒ
R SǓ ) 4kTR S ) 2
2
i IR F
GD
2
)2
= ±2.0mV ± 0.12mV
Ǔ
4kTR F
GD
THERMAL ANALYSIS
Due to the high output power capability of the OPA2613,
heat-sinking or forced airflow may be required under
extreme operating conditions. Maximum desired junction
temperature sets the maximum allowed internal power
dissipation as described below. In no case should the
maximum junction temperature be allowed to exceed
150°C. Operating junction temperature (TJ) is given by TA
+ PD × qJA. The total internal power dissipation (PD) is the
sum of quiescent power (PDQ) and additional power
dissipation in the output stage (PDL) to deliver load power.
Quiescent power is the specified no-load supply current
times the total supply voltage across the part. PDL
23
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depends on the required output signal and load, but for a
grounded resistive load, PDL is at a maximum when the
output is fixed at a voltage equal to 1/2 of either supply
voltage (for equal bipolar supplies). Under this condition,
PDL = VS2/(4 × RL) where RL includes feedback network
loading. Note that it is the power in the output stage and not
into the load that determines internal power dissipation. As
a worst-case example, compute the maximum TJ using an
OPA2613 SO-8 in the circuit of Figure 1 operating at the
maximum specified ambient temperature of +85°C with
both outputs driving a grounded 20Ω load to +3.0V.
PD = 12V × 13.0mA + 2 × [62/ (4 × (20Ω  804Ω))] = 1. 08W
Maximum TJ = +85°C + (1.08W × 125°C/W) = 220°C
This absolute worst-case condition exceeds specified
maximum junction temperature. This extreme case is not
normally encountered. Where high internal power dissipation is anticipated, consider the thermal slug package
version. Under the same worst case conditions the
junction temperature will drop to 139°C with the 50°C/W
thermal impedance available using the PSO-8 package.
BOARD LAYOUT GUIDELINES
Achieving optimum performance with a high-frequency
amplifier like the OPA2613 requires careful attention to
board layout parasitic and external component types.
Recommendations that optimize performance include:
a) Minimize parasitic capacitance to any AC ground for
all of the signal I/O pins. Parasitic capacitance on the
output and inverting input pins can cause instability; on the
noninverting input, it can react with the source impedance
to cause unintentional band limiting. To reduce unwanted
capacitance, a window around the signal I/O pins should
be opened in all of the ground and power planes around
those pins. Otherwise, ground and power planes should
be unbroken elsewhere on the board.
b) Minimize the distance (< 0.25″) from the power-supply
pins to high-frequency 0.1µF decoupling capacitors. At the
device pins, the ground and power plane layout should not
be in close proximity to the signal I/O pins. Avoid narrow
power and ground traces to minimize inductance between
the pins and the decoupling capacitors. The power-supply
connections (on pins 4 and 7) should always be decoupled
with these capacitors. An optional supply decoupling
capacitor across the two power supplies (for bipolar
operation) improves 2nd-harmonic distortion performance.
Larger (2.2µF to 6.8µF) decoupling capacitors, effective at
a lower frequency, should also be used on the main supply
pins. These can be placed somewhat farther from the
device and may be shared among several devices in the
same area of the PC board.
24
c) Careful selection and placement of external
components preserve the high-frequency performance
of the OPA2613. Resistors should be of a very low
reactance type. Surface-mount resistors work best and
allow a tighter overall layout. Metal film and carbon
composition axially leaded resistors can also provide good
high-frequency performance. Again, keep the leads and
PC board trace length as short as possible. Never use
wire-wound type resistors in a high-frequency application.
Although the output pin and inverting input pin are the most
sensitive to parasitic capacitance, always position the
feedback and series output resistor, if any, as close as
possible to the output pin. Other network components,
such as noninverting input termination resistors, should
also be placed close to the package. Where double-side
component mounting is allowed, place the feedback
resistor directly under the package on the other side of the
board between the output and inverting input pins. The
402Ω feedback resistor used in the Typical Characteristics
at a gain of +2 on ±6V supplies is a good starting point for
design.
d) Connections to other wideband devices on the board
may be made with short direct traces or through onboard
transmission lines. For short connections, consider the
trace and the input to the next device as a lumped
capacitive load. Relatively wide traces (50mils to 100mils)
should be used, preferably with ground and power planes
opened up around them. Estimate the total capacitive load
and set RS from the plot of Recommended RS vs
Capacitive Load. Low parasitic capacitive loads (< 5pF)
may not need an RS because the OPA2613 is nominally
compensated to operate with a 2pF parasitic load. If a long
trace is required, and the 6dB signal loss intrinsic to a
doubly-terminated transmission line is acceptable,
implement a matched impedance transmission line using
microstrip or stripline techniques (consult an ECL design
handbook for microstrip and stripline layout techniques). A
50Ω environment is normally not necessary on board; in
fact, a higher impedance environment improves distortion
(see the distortion versus load plots). With a characteristic
board trace impedance defined based on board material
and trace dimensions, a matching series resistor into the
trace from the output of the OPA2613 is used, as well as
a terminating shunt resistor at the input of the destination
device. Remember also that the terminating impedance is
the parallel combination of the shunt resistor and the input
impedance of the destination device.
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This total effective impedance should be set to match the
trace impedance. The high output voltage and current
capability of the OPA2613 allows multiple destination
devices to be handled as separate transmission lines,
each with their own series and shunt terminations. If the
6dB attenuation of a doubly-terminated transmission line
is unacceptable, a long trace can be series-terminated at
the source end only. Treat the trace as a capacitive load in
this case and set the series resistor value as shown in the
plot of RS vs Capacitive Load. However, this does not
preserve signal integrity as well as a doubly-terminated
line. If the input impedance of the destination device is low,
there is some signal attenuation due to the voltage divider
formed by the series output into the terminating
impedance.
e) Socketing a high-speed part like the OPA2613 is not
recommended. The additional lead length and pin-to-pin
capacitance introduced by the socket can create an
extremely troublesome parasitic network, which can make
it almost impossible to achieve a smooth, stable frequency
response. Best results are obtained by soldering the
OPA2613 onto the board.
f) Use the −VS plane to conduct heat out of the PSO-8
power package (OPA2613H). This package attaches the
die directly to a metal slug in the bottom, which should be
soldered to the board. This slug needs to be connected
electrically to the same voltage plane as the most negative
supply applied to the OPA2613 (in Figure 6, this would be
ground), which must have a minimum area of 2″ x 2″
(50mm x 50mm) to produce the qJA values in the
specifications table.
INPUT AND ESD PROTECTION
The OPA2613 is built using a high-speed complementary
bipolar process. The internal junction breakdown voltages
are relatively low for these very small geometry devices
and are reflected in the absolute maximum ratings table.
All device pins have limited ESD protection using internal
diodes to the power supplies, as shown in Figure 15.
These diodes provide moderate protection to input
overdrive voltages above the supplies as well. The
protection diodes can typically support 30mA continuous
current. Where higher currents are possible (for example,
in systems with ±15V supply parts driving into the
OPA2613), current-limiting series resistors should be
added into the two inputs. Keep these resistor values as
low as possible, because high values degrade both noise
performance and frequency response.
+VCC
External
Pin
Internal
Circuitry
−VCC
Figure 15. Internal ESD Protection
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IMPORTANT NOTICE
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Following are URLs where you can obtain information on other Texas Instruments products and application
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Products
Applications
Amplifiers
amplifier.ti.com
Audio
www.ti.com/audio
Data Converters
dataconverter.ti.com
Automotive
www.ti.com/automotive
DSP
dsp.ti.com
Broadband
www.ti.com/broadband
Interface
interface.ti.com
Digital Control
www.ti.com/digitalcontrol
Logic
logic.ti.com
Military
www.ti.com/military
Power Mgmt
power.ti.com
Optical Networking
www.ti.com/opticalnetwork
Microcontrollers
microcontroller.ti.com
Security
www.ti.com/security
Telephony
www.ti.com/telephony
Video & Imaging
www.ti.com/video
Wireless
www.ti.com/wireless
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