Maxim MAX1802EHJ Digital camera step-down power supply Datasheet

19-1850; Rev 0; 10/00
Digital Camera Step-Down
Power Supply
The main step-down DC-DC controller accepts inputs
from 2.5V to 11V and regulates a resistor-adjustable output from 2.7V to 5.5V. It uses a synchronous rectifier to
regulate the output with up to 94% efficiency. An
adjustable operating frequency (up to 1MHz) facilitates
designs for optimum size, cost, and efficiency.
The core step-down DC-DC converter accepts inputs
from 2.7V to 5.5V and regulates a resistor-adjustable
output from 1.25V to 5.5V. It delivers 500mA with up to
94% efficiency.
The three auxiliary step-up controllers can be used to
power the digital camera’s CCD, LCD, and backlight.
The MAX1802 also features expandability by supplying
power, an oscillator signal, and a reference to the
MAX1801, a low-cost slave DC-DC controller that supports step-up, single-ended primary inductance converter (SEPIC), and fly-back configurations.
The MAX1802 is available in a space-saving 32-pin
TQFP package (5mm x 5mm body), and the MAX1801
is available in an 8-pin SOT-23 package. An evaluation
kit (MAX1802EVKIT) featuring both devices is available
to expedite designs.
________________________Applications
Digital Still Cameras
Digital Video Cameras
Hand-Held Devices
Internet Access Tablets
PDAs
Features
♦ 2.5V to 11V Input Voltage Range
♦ Main DC-DC Controller
94% Efficiency
+2.7V to +5.5V Adjustable Output Voltage
Up to 100% Duty Cycle
Independent Shutdown
♦ Core DC-DC Converter
94% Efficiency
Up to 500mA Load Efficiency
Output Voltage Adjustable Down to 1.25V
Independent Shutdown
♦ Three Auxiliary DC-DC Controllers
Adjustable Maximum Duty Cycle
Independent Shutdown
♦ Power, Oscillator, and Reference Outputs to Drive
External Slave Controllers (MAX1801)
♦ Up to 1MHz Switching Frequency
♦ 3µA Supply Current in Shutdown Mode
♦ Internal Soft-Start
♦ Overload Protection for All DC-DC Converters
♦ Compact 32-Pin TQFP Package
Ordering Information
PART
MAX1802EHJ
TEMP. RANGE
PIN-PACKAGE
-40°C to +85°C
32 TQFP
Note: Refer to the separate data sheet for MAX1801EKA in an 8pin SOT.
Typical Operating Circuit
DVD Players
MAIN
INPUT
2.5V TO 11V
Pin Configuration appears at end of data sheet.
CORE
CCD
MAX1802
MASTER
CCFL
TFT
OSC
POWER
MAX1801
SLAVE
REF
MOTOR
________________________________________________________________ Maxim Integrated Products
1
For price, delivery, and to place orders, please contact Maxim Distribution at 1-888-629-4642,
or visit Maxim’s website at www.maxim-ic.com.
MAX1802
General Description
The MAX1802 provides a complete power-supply solution for digital still cameras and video cameras by integrating two high-efficiency step-down DC-DC converters
and three auxiliary step-up controllers. This complete
solution is targeted for applications that use either three
to four alkaline cells or two lithium-ion (Li+) cells.
MAX1802
Digital Camera Step-Down
Power Supply
ABSOLUTE MAXIMUM RATINGS
VDDM, VH, ONM to GND .......................................-0.3V to +12V
PGNDM, PGND to GND ........................................-0.3V to +0.3V
VH to VDDM .............................................................-6V to +0.3V
VL to VDDM ............................................................-12V to +0.3V
VL, ONC, ON1, FB_, DCON_ to GND ......................-0.3V to +6V
VDDC, REF, OSC, COMP_ to GND ..............-0.3V to (VL + 0.3V)
DHM, DLM to PGNDM............................-0.3V to (VDDM + 0.3V)
LXM to PGNDM ......................................-0.6V to (VDDM + 0.6V)
DL1, DL2, DL3, LXC to PGND ................-0.3V to (VDDC + 0.3V)
Continuous Power Dissipation (TA = +70°C)
32-Pin TQFP (derate 11.1mW/°C above +70°C)........889mW
Operating Temperature Range ...........................-40°C to +85°C
Junction Temperature ......................................................+150°C
Storage Temperature Range. ............................-65°C to +150°C
Lead Temperature (soldering, 10s) .................................+300°C
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(Circuit of Figure 1, VVDDM = 6V, VVDDC = 3V, PGNDM = PGND = GND, DCON1 = REF, VONM = 3V, VONC = VON1 = VDCON2 =
VDCON3 = 0, TA = 0°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C.)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
11
V
3
20
µA
VFBM = 1.5V, VVDDC = 0
370
600
VFBM = 1.5V, VVDDC = 3V
35
55
Main DC-DC Converter
Supply Current (from VDDC)
VFBM = 1.5V, VVDDC = 3V
270
450
µA
Main plus Core Supply Current
(from VDDC)
VFBM = VFBC = 1.5V, VONC = 3V
410
700
µA
Main plus Auxiliary 1
Supply Current (from VDDC)
VFBM = VFB1 = 1.5V, VON1 = 3V
470
750
µA
Main plus Auxiliary 2
Supply Current (from VDDC)
VFBM = VFB2 = 1.5V, VDCON2 = 3V
470
750
µA
Main plus Auxiliary 3
Supply Current (from VDDC)
VFBM = VFB3 = 1.5V, VDCON3 = 3V
470
750
µA
Total Supply Current
(from VDDC)
VFBM = VFBC = VFB1 = VFB2 = VFB3 = 1.5V,
VONC = VON1 = VDCON2 = VDCON3 = 3V
960
1700
µA
3.00
3.12
V
3
%
GENERAL
Input Voltage Range
VIN
2.5
SUPPLY CURRENT
Shutdown Supply Current
(from VDDM and VDDC)
Main DC-DC Converter
Supply Current (from VDDM)
VONM = 0
µA
VL REGULATOR
VL Output Voltage
6V < VVDDM < 11V, 0.1mA < ILOAD < 10mA
VL Supply Rejection
3.5V < VVDDM < 11V, VVDDC = 0
2.83
VL Undervoltage Lockout
Threshold
VL rising, 40mV hysteresis
2.25
2.40
2.50
V
VL Switchover Voltage to
VDDC
VL rising, 100mV hysteresis
2.3
2.4
2.5
V
7
Ω
VL to VDDC Switch Resistance
2
_______________________________________________________________________________________
Digital Camera Step-Down
Power Supply
(Circuit of Figure 1, VVDDM = 6V, VVDDC = 3V, PGNDM = PGND = GND, DCON1 = REF, VONM = 3V, VONC = VON1 = VDCON2 =
VDCON3 = 0, TA = 0°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C.)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
1.235
UNITS
REFERENCE
Reference Output Voltage
1.248
1.260
V
REF Load Regulation
VREF
IREF = 20µA
10µA < IREF < 200µA
5
9
mV
REF Line Rejection
2.7V < VOUT < 5.5V
1
5
mV
REF Undervoltage Lockout
Threshold
REF rising, 20mV hysteresis
0.9
1
1.1
V
1.225
1.250
1.275
V
0.2
100
nA
30
100
OSCILLATOR
OSC Discharge Trip Level
OSC rising
OSC Input Bias Current
VOSC = 1.1V
OSC Discharge Resistance
VOSC = 1.5V
OSC Discharge Pulse Width
100
Ω
ns
LOGIC INPUTS (ONM, ONC, ON1)
Input Low Level
VIL
Input High Level
VIH
0.4
ONM
1.8
ONC, ON1
1.6
ONM: VIN = 0 or 11V;
ONC, ON1: VIN = 0 or 5V
Input Leakage Current
V
V
0.01
1
µA
5.5
V
MAIN DC-DC CONVERTER
Main Output Voltage Adjust
Range
VOUT
VOSC = 0.625V, measured between VDDM
and LXM
Main Idle Mode™ Threshold
Main Current-Sense Amplifier
Voltage Gain
AVCSM
Main N Channel Turn-Off
Threshold
Main Slope Compensation
Gain
2.7
8
20
32
mV
Measured between VDDM and LXM
8.4
9.3
10.2
V/V
Measured between LXM and PGNDM
-26
-17
-8
mV
0.16
0.20
0.24
V/V
Unity gain configuration, FBM = COMPM
1.233
1.248
1.263
V
Unity gain configuration, FBM = COMPM,
-5µA < ILOAD < 5µA
70
100
160
µS
5
100
nA
AVSWM
MAIN ERROR AMPLIFIER
FBM Regulation Voltage
FBM to COMPM
Transconductance
GEA
FBM Input Leakage Current
VFBM = 1.35V
COMPM Minimum Output
Voltage
VFBM = 1.35V, COMPM open
0.3
VFBM = 1.15V, COMPM open
2.00
COMPM Maximum Output
Voltage
VCOMPM(MAX)
V
2.14
2.27
V
Idle Mode is a trademark of Maxim Integrated Products.
_______________________________________________________________________________________
3
MAX1802
ELECTRICAL CHARACTERISTICS (continued)
MAX1802
Digital Camera Step-Down
Power Supply
ELECTRICAL CHARACTERISTICS (continued)
(Circuit of Figure 1, VVDDM = 6V, VVDDC = 3V, PGNDM = PGND = GND, DCON1 = REF, VONM = 3V, VONC = VON1 = VDCON2 =
VDCON3 = 0, TA = 0°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C.)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
MAIN SOFT-START
Soft-Start Interval
OSC falling edge
OSC
cycles
1024
MAIN DRIVERS (DHM, DLM)
Output Low Voltage
ISINK = 10mA
Output High Voltage
ISOURCE = 10mA
Driver Resistance
IDHM = 10mA, IDLM = 10mA
Drive Current
Sourcing or sinking,
VDHM or VVL = VVDDM / 2
0.11
VVDDM 0.11
V
V
4
11
400
Ω
mA
CORE DC-DC CONVERTER (VONC = 3V)
Core Output Voltage Adjust
Range
VOUT
Core Idle Mode Threshold
Core Current-Sense Amplifier
Transresistance
Core Slope Compensation Gain
1.25
VOSC = 0.625V
5.5
V
70
190
320
mA
RCSC
0.7
1.0
1.3
V/A
AVSWC
0.16
0.20
0.24
V/V
Unity gain configuration, FBC = COMPC
1.233
1.248
1.263
V
Unity gain configuration, FBC = COMPC,
-5µA < ILOAD < 5µA
70
100
160
µS
5
100
nA
CORE ERROR AMPLIFIER (VONC = 3V)
FBC Regulation Voltage
FBC to COMPC
Transconductance
GEA
FBC Input Leakage Current
VFBC = 1.35V
COMPC Minimum Output
Voltage
VFBC = 1.35V, COMPC open
0.3
VFBC = 1.15V, COMPC open
2.00
COMPC Maximum Output
Voltage
VCOMPM(MAX)
V
2.14
2.27
V
CORE SOFT-START (VONC = 3V)
Soft-Start Interval
OSC
cycles
1024
CORE POWER SWITCHES (VONC = 3V)
LXC Leakage Current
Switch On-Resistance
P-Channel Current Limit
N-Channel Turn-Off Current
4
VLXC = 0, 5.5V
0.01
20
RDSN
N-channel, ILXC = 0.75A
150
350
RDSP
P-channel, ILXC = 0.75A
180
400
VOSC = 0.625V
0.75
18
100
_______________________________________________________________________________________
µA
mΩ
A
180
mA
Digital Camera Step-Down
Power Supply
(Circuit of Figure 1, VVDDM = 6V, VVDDC = 3V, PGNDM = PGND = GND, DCON1 = REF, VONM = 3V, VONC = VON1 = VDCON2 =
VDCON3 = 0, TA = 0°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C.)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
V
AUXILIARY DC-DC CONTROLLERS 1, 2, 3 (VON1 = VCON_ = 3V)
INTERNAL CLOCK
OSC Clock Low Trip Level
OSC Clock High Trip Level
OSC falling edge
0.2
0.25
0.3
VDCON _ = 0.625V
0.575
0.625
0.675
VDCON _ = 1.25V to VVL
1.00
1.05
1.10
Maximum Duty Cycle
Adjustment Range
40
Maximum Duty Cycle
VDCON _ = 0.625V
Default Maximum Duty Cycle
VDCON _ = 1.25V to VVL
DCON_ Input Leakage Current
VDCON _ = 0V to 3V
DCON_ Input Sleep-Mode
Threshold
VDCON _ rising, 50mV hysteresis
90
43
V
%
%
76
%
0.01
1
µA
0.35
0.4
0.45
V
Unity gain configuration, FB_ = COMP_
1.233
1.248
1.263
V
Unity gain configuration, FB_ = COMP_,
-5µA < ILOAD < 5µA
70
100
160
µs
5
100
nA
4
11
AUXILIARY ERROR AMPLIFIER
FB_ Regulation Voltage
FB_ to COMP_
Transconductance
GEA
FB_ Input Leakage Current
VFB_ = 1.35V
AUXILIARY DRIVERS (DL1, DL2, DL3)
DL_ Driver Resistance
Output high or low
DL_ Drive Current
Sourcing or sinking, VDL_ = VVDDC / 2
Ω
400
mA
1024
OSC
cycles
1024
OSC
cycles
AUXILIARY SOFT-START
Soft-Start Interval
AUXILIARY SHORT-CIRCUIT PROTECTION
Fault Interval
_______________________________________________________________________________________
5
MAX1802
ELECTRICAL CHARACTERISTICS (continued)
MAX1802
Digital Camera Step-Down
Power Supply
ELECTRICAL CHARACTERISTICS
(Circuit of Figure 1, VVDDM = 6V, VVDDC = 3V, PGNDM = PGND = GND, DCON1 = REF, VONM = 3V, VONC = VON1 = VDCON2 =
VDCON3 = 0, TA = -40°C to +85°C, unless otherwise noted.) (Note 1)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
11
V
VONM = 0
20
µA
VFBM = 1.5V, VVDDC = 0
600
VFBM = 1.5V, VVDDC = 3V
55
Main DC-DC Converter
Supply Current (from VDDC)
VFBM = 1.5V, VVDDC = 3V
450
µA
Main plus Core Supply Current
(from VDDC)
VFBM = VFBC = 1.5V, VONC = 3V
700
µA
Main plus Auxiliary 1 Supply
Current (from VDDC)
VFBM = VFB1 = 1.5V, VON1 = VDCON1 = 3V
750
µA
Main plus Auxiliary 2 Supply
Current (from VDDC)
VFBM = VFB2 = 1.5V, VDCON2 = 3V
750
µA
Main plus Auxiliary 3 Supply
Current (from VDDC)
VFBM = VFB3 = 1.5V, VDCON3 = 3V
750
µA
Total Supply Current
(from VDDC)
VFBM = VFBC = VFB1 = VFB2 = VFB3 = 1.5V,
VONC = VON1 = VDCON1 = VDCON2 =
VDCON3 = 3V
1700
µA
3.12
V
3
%
V
GENERAL
Input Voltage Range
VIN
2.5
SUPPLY CURRENT
Shutdown Supply Current
(from VDDM and VDDC)
Main DC-DC Converter
Supply Current (from VDDM)
µA
VL REGULATOR
VL Output Voltage
6V < VVDDM < 11V,
0.1mA < ILOAD < 10mA
VL Supply Rejection
3.5V < VVDDM < 11V, VVDDC = 0
VL Undervoltage Lockout
Threshold
VL rising, 40mV hysteresis
2.25
2.50
VL Switchover Voltage to VDDC
VL rising, 100mV hysteresis
2.3
2.5
V
7
Ω
2.83
VL to VDDC Switch Resistance
REFERENCE
Reference Output Voltage
1.262
V
REF Load Regulation
VREF
IREF = 20µA
10µA < IREF < 200µA
1.230
9
mV
REF Line Rejection
2.7V < VOUT < 5.5V
5
mV
REF Undervoltage Lockout
Threshold
REF rising, 20mV hysteresis
0.9
1.1
V
1.225
OSCILLATOR
OSC Discharge Trip Level
OSC rising
1.275
V
OSC Input Bias Current
VOSC = 1.1V
100
nA
OSC Discharge Resistance
VOSC = 1.5V
100
Ω
6
_______________________________________________________________________________________
Digital Camera Step-Down
Power Supply
(Circuit of Figure 1, VVDDM = 6V, VVDDC = 3V, PGNDM = PGND = GND, DCON1 = REF, VONM = 3V, VONC = VON1 = VDCON2 =
VDCON3 = 0, TA = -40°C to +85°C, unless otherwise noted.) (Note 1)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
0.4
V
LOGIC INPUTS (ONM, ONC, ON1)
Input Low Level
VIL
Input High Level
VIH
ONM
1.8
ONC, ON1
1.6
ONM: VIN = 0 or 11V;
ONC, ON1: VIN = 0 or 5V
Input Leakage Current
V
1
µA
2.7
5.5
V
VOSC = 0.625V, measured between
VDDM and LXM
2
35
mV
Measured between VDDM and LXM
8.4
10.2
V/V
MAIN DC-DC CONVERTER
Main Output Voltage Adjust Range
VOUT
Main Idle Mode Threshold
Main Current-Sense Amplifier
Voltage Gain
AVCSM
Main Zero-Crossing Threshold
Main Slope Compensation Gain
Measured between LXM and PGNDM
AVSWM
-20
-8
mV
0.16
0.24
V/V
1.230
1.265
V
70
160
µS
100
nA
MAIN ERROR AMPLIFIER
FBM Regulation Voltage
FBM to COMPM
Transconductance
Unity gain configuration, FBM = COMPM
GEA
Unity gain configuration, FBM = COMPM,
-5µA < ILOAD < 5µA
FBM Input Leakage Current
VFBM = 1.35V
COMPM Minimum Output
Voltage
VFBM = 1.35V, COMPM open
0.3
COMPM Maximum Output
Voltage
VCOMPM(MAX) VFBM = 1.15V, COMPM open
2.00
V
2.27
V
0.11
V
MAIN DRIVERS (DHM, DLM)
Output Low Voltage
ISINK = 10mA
Output High Voltage
ISOURCE = 10mA
Driver Resistance
IDHM = 10mA, IDLM = 10mA
VVDDM 0.11
V
11
Ω
1.25
5.5
V
40
360
mA
RCSC
0.7
1.3
V/A
AVSWC
0.16
0.24
V/V
1.230
1.265
V
70
160
µS
CORE DC-DC CONVERTER (VONC = 3V)
Core Output Voltage Adjust
Range
VOUT
Core Idle Mode Threshold
Core Current-Sense Amplifier
Transresistance
Core Slope Compensation Gain
VOSC = 0.625V
CORE ERROR AMPLIFIER (VONC = 3V)
FBC Regulation Voltage
FBC to COMPC
Transconductance
Unity gain configuration, FBC = COMPC
GEA
Unity gain configuration, FBC = COMPC,
-5µA < ILOAD < 5µA
_______________________________________________________________________________________
7
MAX1802
ELECTRICAL CHARACTERISTICS (continued)
MAX1802
Digital Camera Step-Down
Power Supply
ELECTRICAL CHARACTERISTICS (continued)
(Circuit of Figure 1, VVDDM = 6V, VVDDC = 3V, PGNDM = PGND = GND, DCON1 = REF, VONM = 3V, VONC = VON1 = VDCON2 =
VDCON3 = 0, TA = -40°C to +85°C, unless otherwise noted.) (Note 1)
PARAMETER
SYMBOL
CONDITIONS
MIN
FBC Input Leakage Current
VFBC = 1.35V
COMPC Minimum Output
Voltage
VFBC = 1.35V, COMPC open
0.3
VFBC = 1.15V, COMPC open
2.00
COMPC Maximum Output
Voltage
VCOMPC(MAX)
TYP
MAX
UNITS
100
nA
V
2.27
V
µA
CORE POWER SWITCHES (VONC = 3V)
LXC Leakage Current
Switch On-Resistance
VLXC = 0, 5.5V
20
RDSN
N-channel, ILXC = 0.75A
350
RDSP
P-channel, ILXC = 0.75A
400
N-Channel Turn-Off Current
5
mΩ
190
mA
AUXILIARY DC-DC CONTROLLERS 1, 2, 3 (VON1 = VDCON_= 3V)
INTERNAL CLOCK
OSC Clock Low Trip Level
OSC Clock High Trip Level
OSC falling edge
0.2
0.3
V
VDCON_ = 0.625V
0.575
0.675
V
VDCON_ = 1.25V to VVL
1.00
1.10
40
90
%
1
µA
Maximum Duty Cycle
Adjustment Range
DCON_ Input Leakage Current
VDCON_ = 0V to 3V
DCON_ Input Sleep-Mode
Threshold
VDCON_ rising, 50mV hysteresis
0.35
0.45
V
Unity gain configuration, FB_ = COMP_
1.230
1.265
V
Unity gain configuration, FB_ = COMP_,
-5µA < ILOAD < 5µA
70
160
µs
VFB_ = 1.35V
100
nA
Output high or low
11
Ω
AUXILIARY ERROR AMPLIFIER
FB_ Regulation Voltage
FB_ to COMP_
Transconductance
GEA
FB_ Input Leakage Current
AUXILIARY DRIVERS (DL1, DL2, DL3)
DL_ Driver Resistance
Note 1: Specifications to -40°C are guaranteed by design and not production tested.
8
_______________________________________________________________________________________
Digital Camera Step-Down
Power Supply
EFFICIENCY vs. LOAD CURRENT
(MAIN CONVERTER)
EFFICIENCY vs. LOAD CURRENT
(MAIN CONVERTER)
50
40
50
40
20
20
10
100
1000
10,000
MAX1802 toc03
VIN = +5V
40
VOUT = +1.8V
10
0
1
10
100
1000
10,000
LOAD CURRENT (mA)
10
100
LOAD CURRENT (mA)
EFFICIENCY vs. LOAD CURRENT
(CORE CONVERTER)
MAXIMUM DUTY CYCLE vs. VDCON_
DEFAULT MAXIMUM DUTY CYCLE
vs. FREQUENCY
VIN = +5V
60
50
40
30
20
DEFAULT MAXIMUM DUTY CYCLE (%)
70
80
60
40
20
VOUT = +2.5V
10
1
100
MAX1802 toc05
80
100
MAXIMUM DUTY CYCLE (%)
MAX1802 toc04
VIN = +3.3V
0
0
10
100
LOAD CURRENT (mA)
COSC = 470pF
80
60
40
20
0.5
0.6
0.7
0.8
0.9
1.0
1.1
1.2
0
200
VDCON_ (V)
600
800
1000
COSC = 100pF
COSC = 47pF
400
200
SHUTDOWN CURRENT (µA)
COSC = 220pF
MAX1802 toc08
10
MAX1802 toc07
COSC = 470pF
600
400
FREQUENCY (kHz)
SHUTDOWN CURRENT
vs. INPUT VOLTAGE
1000
800
1000
0
0.4
1000
OSCILLATOR FREQUENCY
vs. ROSC
OSCILLATOR FREQUENCY (kHz)
1
VIN = +3.3V
50
LOAD CURRENT (mA)
100
90
VIN = +2.5V
60
20
VOUT = +5V
0
1
70
30
10
0
EFFICIENCY (%)
60
30
VOUT = 3.3V
80
VIN = +11V
70
30
10
90
MAX1802 toc06
VIN = +11V
60
100
EFFICIENCY (%)
VIN = +7.2V
70
VIN = +7.2V
80
EFFICIENCY (%)
EFFICIENCY (%)
80
90
MAX1802 toc02
VIN = +5V
90
100
MAX1802 toc01
100
EFFICIENCY vs. LOAD CURRENT
(CORE CONVERTER)
8
6
4
2
0
0
1
10
100
ROSC (kΩ)
1000
0
2
4
6
8
10
12
INPUT VOLTAGE (V)
_______________________________________________________________________________________
9
MAX1802
Typical Operating Characteristics
(Circuit of Figure 1, VVDDM = 6V, VVDDC = 3.3V, VONM = 3V, VONC = VON1 = VDCON2 = VDCON3 = 0, TA = +25°C, unless otherwise
noted.)
Typical Operating Characteristics (continued)
(Circuit of Figure 1, VVDDM = 6V, VVDDC = 3.3V, VONM = 3V, VONC = VON1 = VDCON2 = VDCON3 = 0, TA = +25°C, unless otherwise
noted.)
REFERENCE VOLTAGE
vs. TEMPERATURE
REFERENCE VOLTAGE
vs. REFERENCE CURRENT
1.255
1.250
1.245
MAX1802 toc10
MAX1802 toc09
1.253
1.252
REFERENCE VOLTAGE (V)
REFERENCE VOLTAGE (V)
1.260
1.251
1.250
1.249
1.248
1.240
-40
1.247
-20
0
20
40
60
80
0
50
100
150
200
TEMPERATURE (°C)
REFERENCE CURRENT (µA)
FB_ TO COMP_ SMALL-SIGNAL
OPEN-LOOP FREQUENCY RESPONSE
MAIN OUTPUT STARTUP RESPONSE
250
MAX1802 toc12
MAX1802 toc11
60
SMALL-SIGNAL RESPONSE (dB)
MAX1802
Digital Camera Step-Down
Power Supply
50
VONM
5V/div
0V
VMAIN
2V/div
40
0V
30
IOUT
200mA/div
20
10
0A
0
1
10
100
1000
1ms/div
10,000
FREQUENCY (kHz)
AUXILIARY CONTROLLER
STARTUP RESPONSE
CORE OUTPUT STARTUP RESPONSE
MAX1802 toc13
MAX1802 toc14
VONC
5V/div
VON_
5V/div
VOUT
2V/div
0V
0V
VCORE
2V/div
0V
IOUT
100mA/div
0V
IOUT
200mA/div
0A
0A
1ms/div
10
1ms/div
______________________________________________________________________________________
Digital Camera Step-Down
Power Supply
MAIN OUTPUT
LOAD-TRANSIENT RESPONSE
STARTUP SEQUENCE
MAX1802 toc16
MAX1802 toc15
VONM
5V/div
VOUT
AC-COUPLED
100mV/div
0V
VMAIN
2V/div
ILOAD
200mA/div
0V
VCORE
2V/div
0A
0A
COUT = 100µF
1ms/div
400µs/div
CORE OUTPUT
LOAD-TRANSIENT RESPONSE
AUXILIARY OUTPUT
LOAD-TRANSIENT RESPONSE
MAX1802 toc18
MAX1802 toc17
VOUT
AC-COUPLED
100mV/div
VOUT
AC-COUPLED
200mV/div
ILOAD
100mA/div
ILOAD
200mA/div
0A
0A
VOUT = 2.5V
400µs/div
500µs/div
MAIN TRANSIENT RESPONSE
SUBJECT TO CORE TRANSIENT
MAX1802 toc19
VOUT (MAIN)
AC-COUPLED
20mV/div
ILOAD (CORE)
100mA/div
0A
VOUT = 2.5V
2.5ms/div
______________________________________________________________________________________
11
MAX1802
Typical Operating Characteristics (continued)
(Circuit of Figure 1, VVDDM = 6V, VVDDC = 3.3V, VONM = 3V, VONC = VON1 = VDCON2 = VDCON3 = 0, TA = +25°C, unless otherwise
noted.)
MAX1802
Digital Camera Step-Down
Power Supply
Pin Description
PIN
NAME
1
FBM
Main DC-DC Converter Feedback Input. Connect a feedback resistive voltage-divider from the output
to FBM to set the main output voltage. Regulation voltage is VREF (1.25V).
2
COMPM
Compensation for Main Controller. Output of main transconductance error amplifier. Connect a series
resistor and capacitor to GND to compensate the main control loop (see Compensation Design).
3
ONM
Main Converter Enable Input. High level turns on the main converter and VL regulator. Connect ONM
to VDDM to automatically start the converter. When the main converter is off, all other outputs are
disabled.
4
VH
Internal Bias Voltage. VH provides bias to the main controller. Bypass VH to VDDM with a 0.1µF or
greater ceramic capacitor.
5
VDDM
Battery Input. VDDM supplies power to the IC and also serves as a high-side current-sense input
for the main DC-DC controller. Connect VDDM as close as possible to the source of the external
P-channel switching MOSFET for the main controller.
6
DHM
7
12
LXM
FUNCTION
External P-Channel MOSFET Gate-Drive Output for Main Controller. DHM swings between VDDM and
PGNDM with 400mA (typ) drive current. Connect DHM to the gate of the external P-channel switching
MOSFET for the main controller.
Main DC-DC Controller Current-Sense Input. Connect LXM to the drains of the external P- and Nchannel switching MOSFETs for the main converter. LXM serves as the current-sense input for both
P- and N-channel switching MOSFETs. Connect LXM as close as possible to the drain of the external
P-channel switching MOSFET for the main controller.
8
DLM
External N-Channel MOSFET Gate-Drive Output for Main Controller. DLM swings between VDDM
and PGNDM with 400mA (typ) drive current. Connect DLM to the gate of the external N-channel
switching MOSFET for the main controller.
9
PGNDM
Power Ground for Main DC-DC Controller. PGNDM also serves as a low-side current-sense input for
the main DC-DC controller. Connect PGNDM as close as possible to the source of the external
N-channel switching MOSFET for the main controller.
10
OSC
Oscillator Control. Connect a timing capacitor from OSC to GND and a timing resistor from OSC to VL
to set the switching frequency between 100kHz and 1MHz (see Setting the Switching Frequency).
11
DCON1
Maximum Duty Cycle Control Input for Auxiliary Controller 1. Connect DCON1 to VL to set the default
maximum duty cycle. Connect a resistive voltage-divider from REF to DCON1 to set the maximum
duty cycle between 40% and 90%. Pull DCON1 below 300mV to turn the controller off.
12
DL1
External MOSFET Gate Drive Output for Auxiliary Controller 1. DL1 swings between VDDC and PGND
with 400mA (typ) drive current. Connect DL1 to the gate of the external switching N-channel MOSFET
for auxiliary controller 1.
13
ON1
Enable Input for Auxiliary Controller 1. Connect ON1 to VL to automatically start auxiliary controller 1.
Compensation for Auxiliary Controller 1. Output of auxiliary controller 1 transconductance error
amplifier. Connect a series resistor and capacitor from COMP1 to GND to compensate the auxiliary
controller 1 control loop (see Compensation Design).
14
COMP1
15
FB1
Feedback Input for Auxiliary Controller 1. Connect a feedback resistive voltage-divider from the
output of auxiliary controller 1 to FB1 to set the output voltage. Regulation voltage is VREF (1.25V).
16
FB2
Feedback Input for Auxiliary Controller 2. Connect a feedback resistive voltage-divider from the
output of auxiliary controller 2 to FB2 to set the output voltage. Regulation voltage is VREF (1.25V).
______________________________________________________________________________________
Digital Camera Step-Down
Power Supply
PIN
17
18
19
20
NAME
FUNCTION
COMP2
Compensation for Auxiliary Controller 2. Output of auxiliary controller 2 transconductance error
amplifier. Connect a series resistor and capacitor from COMP2 to GND to compensate the auxiliary
controller 2 control loop (see Compensation Design).
DCON2
Maximum Duty Cycle Control Input for Auxiliary Controller 2. Connect DCON2 to VL to set the default
maximum duty cycle. Connect a resistive voltage-divider from REF to DCON2 to set the maximum
duty cycle between 40% and 90%. Pull DCON2 below 300mV to turn the controller off.
DL2
External MOSFET Gate Drive Output for Auxiliary Controller 2. DL2 swings between VDDC and PGND
with 400mA (typ) drive current. Connect DL2 to the gate of the external switching N-channel MOSFET
for auxiliary controller 2.
DL3
External MOSFET Gate Drive Output for Auxiliary Controller 3. DL3 swings between VDDC and PGND
with 400mA (typ) drive current. Connect DL3 to the gate of the external switching N-channel MOSFET
for auxiliary controller 3.
Compensation for Auxiliary Controller 3. Output of auxiliary controller 3 transconductance error
amplifier. Connect a series resistor and capacitor from COMP3 to GND to compensate the auxiliary
controller 3 control loop (see Compensation Design).
21
COMP3
22
FB3
23
DCON3
24
ONC
25
PGND
Power Ground. Sources of internal N-channel MOSFET power switches. Connect PGND to GND as
close to the IC as possible.
26
LXC
Core Power Switching Node. Drains of the internal P- and N-channel MOSFET switches for the core
converter.
27
VDDC
Core DC-DC Converter Power Input. VDDC is connected to the source of the internal P-channel
MOSFET power switch for the core converter. VDDC is limited to 5.5V. For battery voltages greater
than 5.5V, connect VDDC to the main output. Bypass VDDC to PGND with a 1µF or greater ceramic
capacitor.
28
VL
Internal Low-Voltage Bypass. The internal circuitry is powered from VL. An internal linear regulator
powers VL from VDDM when VDDC is less than 2.4V. When VDDC is greater than 2.4V, an internal
switch connects VL to VDDC. Bypass VL to GND with a 1.0µF or greater ceramic capacitor.
29
COMPC
30
FBC
Core DC-DC Converter Feedback Input. Connect a feedback resistive voltage-divider from the core
output to FBC to set the output voltage. Regulation voltage is VREF (1.25V).
31
REF
1.25V Reference Output. Bypass REF to GND with a 0.1µF or greater ceramic capacitor.
32
GND
Analog Ground
Feedback Input for Auxiliary Controller 3. Connect a feedback resistive voltage-divider from the
output of auxiliary controller 3 to FB3 to set the output voltage. Regulation voltage is VREF (1.25V).
Maximum Duty Cycle Control Input for Auxiliary Controller 3. Connect DCON3 to VL to set the default
maximum duty cycle. Connect a resistive voltage-divider from REF to DCON3 to set the maximum
duty cycle between 40% and 90%. Pull DCON3 below 300mV to turn the controller off.
Core Converter Enable Input. High level turns on the core converter. Connect ONC to VL to
automatically start the core converter.
Compensation for Core Converter. Output of core transconductance error amplifier. Connect a series
resistor and capacitor to GND to compensate the core control loop (see Compensation Design).
______________________________________________________________________________________
13
MAX1802
Pin Description (continued)
MAX1802
Digital Camera Step-Down
Power Supply
Detailed Description
The MAX1802 typical application circuit is shown in
Figure 1. It features two step-down DC-DC converters
(main and core), three auxiliary step-up DC-DC controllers, and control capability for multiple external
MAX1801 slave DC-DC controllers. Together, these
provide a complete high-efficiency power-supply solution for digital still cameras. Figures 2 and 3 show the
MAX1802 functional block diagrams.
Master-Slave Configuration
The MAX1802 supports MAX1801 “slave” controllers
that obtain input power, a voltage reference, and an
oscillator signal directly from the MAX1802 “master”
DC-DC converter. The master-slave configuration
reduces system cost by eliminating redundant circuitry
and controlling the harmonic content of noise with synchronized converter switching.
Main DC-DC Converter
The MAX1802 main step-down DC-DC converter generates a 2.7V to 5.5V output voltage from a 2.5V to 11V
battery input voltage. When the battery voltage is lower
than the main regulation voltage, the regulator goes
into dropout and the P-channel switch remains on. In
this condition, the output voltage is slightly lower than
the input voltage. The converter drives an external Pchannel MOSFET power switch and an external Nchannel MOSFET synchronous rectifier. The converter
operates in a low-noise, constant-frequency PWM current mode to regulate the voltage across the load.
Switching harmonics generated by fixed-frequency
operation are consistent and easily filtered.
The external P-channel MOSFET switch turns on during
the first part of each cycle, allowing current to ramp up
in the inductor and store energy in a magnetic field
while supplying current to the load. During the second
part of each cycle, the P-channel MOSFET turns off and
the voltage across the inductor reverses, forcing current through the external N-channel synchronous rectifier to the output filter capacitor and load. As the energy
stored in the inductor is depleted, the current ramps
down. The synchronous rectifier turns off when the
inductor current approaches zero or at the beginning of
a new cycle, at which time the P-channel switch turns
on again.
The current-mode PWM converter uses the voltage at
COMPM to program the inductor current and regulate
the output voltage. The converter detects inductor current by sensing the voltage across the source and
14
drain of the external P-channel MOSFET. The MAX1802
main output switches to Idle Mode at light loads to
improve efficiency by leaving the P-channel switch on
until the voltage across the MOSFET reaches the 20mV
Idle Mode threshold. The Idle Mode current is 20mV
divided by the MOSFET on-resistance. By forcing the
inductor current above the Idle Mode threshold, more
energy is supplied to the output capacitor than is
required by the load. The switch and synchronous rectifiers then remain off until the output capacitor discharges to the regulation voltage. This causes the
converter to operate at a lower effective switching frequency at light loads, thus improving efficiency.
An internal comparator turns off the N-channel synchronous rectifier as the inductor current drops near zero,
by measuring the voltage across the MOSFET. If the Nchannel MOSFET on-resistance is low (less than that of
the P-channel switch), it may cause the MOSFET to turn
off prematurely, degrading efficiency. This is especially
critical for high input voltage applications, such as with
2 series Li+ cells. In this case, use an N-channel MOSFET with greater on-resistance than the P-channel
switch, and/or place a Schottky recitifier across the Nchannel MOSFET gate-source.
The voltage at COMPM is typically clamped to
VCOMPM(MAX) = 2.14V, thereby limiting the inductor
current. The peak inductor current (ILIM) and the maximum average output current (I OUT(MAX)) are determined by the following equations:


V
A
VCOMPM(MAX) − VREF 1 + OUT VSWM 
VIN


ILIM =
A VCSM RDSP

 1 −

IOUT(MAX) = ILIM − 



VOUT 
VOUT 

VIN 


2 fOSC L


where AVSWM is the main slope compensation gain
(0.20V/V), AVCSM is the voltage gain of the main current-sense amplifier (9.3V/V), RDSP is the on-resistance
of the external P-channel MOSFET switch, and L is the
inductor value. Note that the current limit increases as
the input/output voltage ratio increases.
______________________________________________________________________________________
+5V
MOTOR DRIVE
OFF
CCM
4.7nF
RCM
33k
100k
464k
4.7µF
6
2
CCC
470pF
RCC
90k
Q1
D5
Q1, Q2, Q3: FDN337N
Q4, Q5: SEE MOSFET SELECTION SECTION
D1, D2, D3, D4: CMSD-4448
D5: MBR0502L
ON
+7V
BACKLIGHT
OSC
MAX1801
IN
1
0.1µH
3
REF
COMP
4
GND DCON
5
8 DL
7
VL
CC1
1000pF
RC1
10k
1µF
10
40.2k
VL
32
FB2
DL2
FB1
DL1
0.1µF
25
PGND
FBC
LXC
VDDC
FBM
PGNDM
DLM
LXM
DHM
VH
VDDM
MAX1802
4
5
GND
21 COMP3
CC3
1000pF
RC3
10k
COMP1
COMPC
COMPM
ONM
ON1
ONC
VL
FB3
DL3
DCON3
DCON2
DCON1
REF
OSC
17 COMP2
14
29
2
3
13
24
28
22
20
23
18
11
COSC
31
100pF
ROSC
CC2
1000pF
RC2
10k
0.1µF
4.7µH
10µF
30
26
27
1
9
8
7
6
16
19
15
12
44.2k
10µH
Q5
Q4
Q3
D6
10µH
Q2
10µF
100k
+15V
LCD BIAS
+12V
+18V
100k
+1.8V
CORE
+3.3V
MAIN
1.34MΩ
100k
-7.5V
1.1MΩ CCD BIAS
165k
1µF
100k
1µF
1µF
100µF
D4
D3
D2
1µF
1µF
D1
MAX1802
INPUT
2.5V TO 11V
Digital Camera Step-Down
Power Supply
Figure 1. Typical Application Circuit
______________________________________________________________________________________
15
MAX1802
Digital Camera Step-Down
Power Supply
OSC
VREF
CLOCK
GENERATOR
100ns
ONE-SHOT
CLK
CLK
REFERENCE
REF
VH
VH
COMPM
VDDM
DHM
MAIN
CURRENT-MODE DC-DC
CONTROLLER
FBM
LXM
DLM
SOFT-START
PGNDM
VREF
VL LDO
ONM
VL
GND
2.4V
CLK
COMPC
VDDC
FBC
CORE
CURRENT MODE
DC-DC
CONTROLLER
VREF
LXC
SOFT-START
PGND
ONC
Figure 2. Simplified Block Diagram, Including Main and Core
Core DC-DC Converter
The MAX1802 core step-down DC-DC converter generates a 1.25V to 5.5V output voltage from the main controller output. The core converter has the same
low-noise, constant-frequency PWM current-mode
architecture as the main controller. However, it uses an
internal P-channel MOSFET power switch and N-channel MOSFET synchronous rectifier to maximize efficiency and reduce circuit size and external component
count. The core converter internally monitors the inductor current for current-mode regulation of the output
voltage, as well as overload protection, automatic Idle
Mode switchover, and turning off the synchronous rectifier when the inductor current approaches zero. By
switching to Idle Mode at light loads and turning the
synchronous rectifier off at zero current, light-load efficiency is improved. The core converter is inactive until
the main output has started.
The voltage at COMPC is typically clamped to
VCOMPC(MAX) = 2.14V, thereby limiting the inductor
current. The peak inductor current limit (ILIM) and the
maximum average output current (I OUT(MAX) ) are
determined by the following equations:
16


V
A
VCOMPC(MAX) − VREF 1 + OUT VSWC 
VIN


ILIM =
RCSC



V
 1 − OUT  VOUT 
VIN 


IOUT(MAX) = ILIM − 

2 fOSC L






where A VSWC is the core slope compensation gain
(0.20V/V), RCSC is the transresistance of the core current-sense amplifier (1V/A), and L is the inductor value.
Note that the current limit increases as the input/output
ratio increases.
Auxiliary DC-DC Controllers
The MAX1802’s three auxiliary controllers operate in a
low-noise, fixed-frequency, PWM mode with output
power limited by the external components. The con-
______________________________________________________________________________________
Digital Camera Step-Down
Power Supply
MAX1802
FB_
COMP_
R
LEVEL
SHIFT
Q
DL_
S
SOFTSTART
REF
DCON_
CLK
OSC
FAULT
PROTECTION
Figure 3. Auxiliary Controller Block Diagram
trollers regulate their output voltages by modulating the
pulse width of the drive signal for an external N-channel
MOSFET switch. The auxiliary controllers are inactive
until the main output has started.
Figure 3 shows a block diagram for a MAX1802 auxiliary PWM controller. The sawtooth oscillator signal at
OSC governs the internal timing. At the beginning of
each cycle, DL_ goes high to turn on the external MOSFET switch. The MOSFET switch turns off when the
internally level-shifted sawtooth rises above COMP_ or
when the maximum duty cycle is exceeded. The switch
remains off until the beginning of the next cycle. An
internal transconductance amplifier establishes an integrated error voltage at COMP_, thereby increasing the
loop gain for improved regulation accuracy.
Power-Up Sequence
The MAX1802 is in the shutdown state with all circuitry
off when the ONM input is low (<1.3V). When ONM
goes high, an internal linear regulator generates 3V at
the VL output from the VDDM input to power internal
circuitry. As VL rises above the 2.4V undervoltage lockout threshold, the internal reference and oscillator
begin to function and the main DC-DC converter
begins soft-start operation. The main DC-DC output
reaches full regulation voltage after 1024 soft-start
oscillator cycles. Once the main DC-DC converter completes soft-start, the core DC-DC converter and the
auxiliary DC-DC controllers are enabled.
As the voltage at VDDC rises above 2.4V, the internal
linear regulator turns off and an internal 3Ω switch connects VL directly to VDDC, which is typically connected
to the output of the main DC-DC converter.
The core DC-DC converter and the auxiliary DC-DC
controllers have independent on-off control and softstart. The main DC-DC converter shuts down with a low
input at ONM. The core DC-DC converter shuts down
with a low input at ONC. Turn auxiliary DC-DC converter 1 off by driving either ON1 or DCON1 to GND. Turn
off auxiliary controller 2 or 3 by driving DCON2 or
DCON3 to GND.
Reference
The MAX1802 has an internal 1.248V, 1% reference.
Connect a 0.1µF bypass capacitor from REF to GND
within 0.2in (5mm) of the REF pin. REF can source up
to 200µA of external load current, and it is enabled
whenever ONM is high and VL is above the undervolt-
______________________________________________________________________________________
17
MAX1802
Digital Camera Step-Down
Power Supply
age lockout threshold. The internal core converter, auxiliary controllers, and MAX1801 slave controllers each
sink up to 30µA REF current during startup. If multiple
MAX1801 controllers are turned on simultaneously,
ensure that the master voltage reference can provide
sufficient current, or buffer the reference with an appropriate unity-gain amplifier.
Oscillator
The oscillator uses a comparator, a 100ns one-shot,
and an internal N-channel MOSFET switch in conjunction with an external timing resistor and capacitor to
generate the oscillator signal at OSC (Figure 4). The
capacitor voltage exponentially approaches VL from
zero with a time constant given by the R OSC C OSC
product when the switch is open, and the comparator
output becomes high when the capacitor voltage
reaches VREF (1.25V). At that time, the one-shot activates the internal MOSFET switch to discharge the
capacitor within a 100ns interval, and the cycle
repeats. Note that the oscillation frequency changes as
VL changes during startup. The oscillation frequency is
constant while the VL voltage is constant.
Maximum Duty Cycle
The MAX1802’s three auxiliary controllers use the sawtooth oscillator signal generated at OSC, the voltage at
DCON_, and an internal comparator to limit their maximum duty cycles (see Setting the Maximum Duty
Cycle). Limiting the duty cycle can prevent saturation in
some magnetic components. A low maximum duty
cycle can also force the converter to operate in discontinuous current mode, simplifying design stability at the
cost of a slight reduction in efficiency.
Soft-Start
All the MAX1802 converters feature a soft-start function
that limits inrush current and prevents excessive battery loading at startup by ramping the output voltage to
the regulation voltage. This is achieved by increasing
the internal reference inputs to the controller transconductance amplifiers from 0 to the 1.25V reference voltage over 1024 oscillator cycles when initial power is
applied or when the controller is enabled.
Overload Protection
The MAX1802’s three auxiliary controllers have fault
protection that prevents damage to transformer-coupled or SEPIC circuits due to an output overload condition. When the output voltage drops out of regulation
for 1024 oscillator clock periods, the auxiliary controller
is disabled to prevent excessive output current. Restart
the controller by cycling the voltage at ON_ or DCON_
to GND and back to the on state. For a step-up appli18
VL
ROSC
OSC
COSC
VREF
(1.25V)
100ns
ONE-SHOT
MAX1802
Figure 4. Oscillator
cation, short-circuit current is not limited, due to the DC
current path through the inductor and output rectifier to
the short circuit. If short-circuit protection is required in
a step-up configuration, use a protection device such
as a fuse to limit short-circuit current.
Design Procedure
Setting the Switching Frequency
Choose a switching frequency to optimize external
component size or circuit efficiency for the particular
MAX1802 application. Switching frequencies between
400kHz and 500kHz offer a good balance between
component size and circuit efficiency. Higher frequencies allow smaller components, and lower frequencies
improve efficiency.
The switching frequency is set with an external timing
resistor (ROSC) and capacitor (COSC). At the beginning
of a cycle, the timing capacitor charges through the
resistor until it reaches VREF. The charge time t1 is:
t1 = -ROSC(COSC+10pF) In [1 - (VREF / VVL)]
Once the voltage at OSC reaches VREF, it discharges
through an internal switch over time t2 = 200ns. The
oscillator frequency is fOSC = 1 / (t1 + t2). Set fOSC in
the range 100kHz ≤ f OSC ≤ 1MHz. Choose C OSC
between 47pF and 470pF. Determine ROSC from the
relation:
______________________________________________________________________________________
Digital Camera Step-Down
Power Supply
See the Typical Operating Characteristics for fOSC vs.
ROSC using different values of COSC. Due to duty cycle
limitation in the main controller, keep fOSC ≤ VMAIN /
(VVDDM(MAX) ✕ 500ns).
Setting the Output Voltages
Set the MAX1802 output voltage of each converter by
connecting a resistive voltage-divider from the output
voltage to the corresponding FB_ input. The FB_ input
bias current is <100nA, so choose RL (the low-side
FB_-to-GND resistor) to be 100kΩ. Choose R H (the
high-side output-to-FB_ resistor) according to the relation:
VOSC (V)
DMAX =
tH
tL + tH
1.25
VDCON_
0.25
0
CLK
V

RH = RL  OUT − 1
1
.
248


tL
tH
Setting the Maximum Duty Cycle
The oscillator signal at OSC and the voltage at DCON_
are used to generate the internal clock signals for the
three MAX1802 auxiliary controllers (CLK in Figure 3).
The internal clock’s falling edge occurs when VOSC
exceeds VDCON_ (set by a resistive divider). The internal clock’s rising edge occurs when VOSC falls below
0.25V (Figure 5).
The adjustable maximum duty cycle range is 40% to
90% (see Maximum Duty Cycle vs. V DCON _ in the
Typical Operating Characteristics). The maximum duty
cycle defaults to 76% at 100kHz if VDCON_ is at or
above the voltage at V REF (1.25V) (see Default
Maximum Duty Cycle vs. Frequency in the Typical
Operating Characteristics). The controller shuts down if
VDCON_ is <0.3V.
Inductor Selection
Main and Core Step-Down Converters
MAX1802 main and core step-down converters offer
best efficiency when the inductor current is continuous.
For most designs, a reasonable inductor value (LIDEAL)
can be derived from the following equation, which sets
continuous peak-to-peak inductor current at 1/3 the DC
inductor current:
 3 (VIN − VDSP ) D (1− D) 
LIDEAL = 

IOUT fOSC


where D, the duty cycle, is given by:
D=
VOUT + VDSN
VIN − VDSP + VDSN
Figure 5. Auxiliary Controller Internal Clock Signal Generation
In these equations, VDSP is the voltage drop across the
P-channel MOSFET switch, and VDSN is the voltage
drop across the N-channel MOSFET synchronous rectifier. Given LIDEAL, the consistent peak-to-peak inductor
current is 0.33 IOUT. The maximum inductor current is
1.17 IOUT.
Inductance values smaller than LIDEAL can be used;
however, the maximum inductor current will rise as L is
reduced, and a larger output capacitance will be
required to maintain the same output ripple. For stable
operation, the minimum inductance is limited by the
internal slope compensation. The minimum inductor
values for main and core are given by:

0.5  VOUT RDSP
LMIN(MAIN) = 1 −
DMAX  0.013 fOSC

and

0.5  VOUT
LMIN(CORE) = 1 −
DMAX  0.13 fOSC

where RDSP is the on-resistance of the P-channel MOSFET switch, and DMAX = VOUT / VIN.
Auxiliary Step-Up Controllers
The three MAX1802 auxiliary step-up controllers offer
best efficiency when the inductor current is continuous.
______________________________________________________________________________________
19
MAX1802
ROSC = (200ns - 1/fOSC) / (COSC + 10pF) ✕
ln (1 - VREF / VVL)
MAX1802
Digital Camera Step-Down
Power Supply
Use discontinuous current when the step-up ratio
(VOUT / VIN) is greater than 1 / (1 - DMAX).
Continuous Inductor Current
A reasonable inductor value (LIDEAL) can be derived
from the following equation, which sets continuous
peak-to-peak inductor current at 1/3 the DC inductor
current:
LIDEAL =
(
)
3 VIN(MAX ) − VDSN D(1− D)
IOUT fOSC
where D, the duty cycle, is given by:
D ≈ 1−
VIN
VOUT + VD
In these equations, VDSN is the voltage drop across the
N-channel MOSFET switch, and VD is the forward voltage drop across the rectifier. Given LIDEAL, the consistent peak-to-peak inductor current is 0.33 IOUT / (1 - D).
The maximum inductor current is 1.17 IOUT / (1 - D).
Inductance values smaller than LIDEAL can be used;
however, the maximum inductor current will rise as L is
reduced, and a larger output capacitance will be
required to maintain the same output ripple.
The inductor current will become discontinuous if IOUT
decreases by more than a factor of six from the value
used to determine LIDEAL.
Discontinuous Inductor Current
In the discontinuous mode, each MAX1802 auxiliary
controller regulates the output voltage by adjusting the
duty cycle to allow adequate power transfer to the load.
To ensure regulation under worst-case load conditions
(maximum IOUT), choose:
L =
VOUT DMAX
2 IOUT fOSC
The peak inductor current is VIN DMAX / (L fOSC).
The inductor’s saturation current rating should meet or
exceed the calculated peak inductor current.
Input and Output Filter Capacitors
The input capacitor (CIN) reduces the current peaks
drawn from the battery or input power source. The
impedance of the input capacitor at the switching frequency should be less than that of the input source so
that high-frequency switching currents do not pass
through the input source.
20
The output capacitor is required to keep the output voltage ripple small and to ensure regulation control-loop
stability. The output capacitor must have low impedance at the switching frequency. Tantalum and ceramic
capacitors are good choices. Tantalum capacitors typically have high capacitance and medium-to-low equivalent series resistance (ESR) so that ESR dominates the
impedance at the switching frequency. In turn, the output ripple is approximately:
VRIPPLE ≈ IL(p-p) ESR
where IL(p-p) is the peak-to-peak inductor current.
Ceramic capacitors typically have lower ESR than tantalum capacitors, but with relatively small capacitance
that dominates the impedance at the switching frequency. In turn, the output ripple is approximately:
VRIPPLE ≈ IL(p-p) ZC
where IL(p-p) is the peak-to-peak inductor current, and
ZC ≈ 1 / (2 π fOSC COUT ).
See the Compensation Design section for a discussion
of the influence of output capacitance and ESR on regulation control-loop stability.
The capacitor voltage rating must exceed the maximum
applied capacitor voltage. For most tantalum capacitors, manufacturers suggest derating the capacitor by
applying no more than 70% of the rated voltage to the
capacitor. Ceramic capacitors are typically used up to
the voltage rating of the capacitor. Consult the manufacturer’s specifications for proper capacitor derating.
MOSFET Selection
The MAX1802 main converter and auxiliary controllers
drive external logic-level P- and/or N-channel MOSFETs
as the circuit switching elements. The key selection
parameters are:
• On-resistance (RDS(ON))
• Maximum drain-to-source voltage (VDS(MAX))
• Total gate charge (Qg)
• Reverse transfer capacitance (CRSS)
Because the main converter’s external MOSFETs are
used for current sense, they directly determine the output current capability and efficiency of the main converter. It is important to select the appropriate external
MOSFETs for the main converter. The P-channel onresistance (RDSP) at minimum input voltage (VVDDM)
must be low enough so that the converter can produce
the desired output current as determined by the
IOUT(MAX) equation in the Main DC-DC Converter section. The N-channel on-resistance (RDSN) determines
______________________________________________________________________________________
Digital Camera Step-Down
Power Supply
For the main converter, the external gate drive swings
between the voltage at VDDM and GND. For the auxiliary controllers, the external gate drive swings between
the voltage at VDDC and GND. Use a MOSFET whose
on-resistance is specified at or below the minimum
gate drive voltage swing, and make sure that the maximum voltage swing does not exceed the maximum
gate-source voltage specification of the MOSFET. The
gate charge, Qg, includes all capacitance associated
with gate charging and helps to predict the transition
time required to drive the MOSFET between on and off
states. The power dissipated in the MOSFET is due to
RDS(ON) and transition losses. The RDS(ON) loss is:
P1 ≈ D IL2 RDS(ON)
where D is the duty cycle, IL is the average inductor
current, and RDS(ON) is the on-resistance of the MOSFET. The transition loss is approximately:
P2 ≈
VSWING IL fOSC t T
3
where VSWING is VOUT for the auxiliary controllers or
VIN(MAX) for the main and core converters, IL is the
average inductor current, fOSC is the converter switching frequency, and tT is the transition time. The transition time is approximately Qg / IG, where Qg is the total
gate charge, and IG is the gate drive current (0.4A typ).
The total power dissipation in the MOSFET is
PMOSFET = P1 + P2.
Diode Selection
The main and core converters use synchronous rectifiers and thus do not require a diode. However, if the
external N-channel synchronous rectifier has low onresistance (less than the P-channel on-resistance), the
high N-channel turn-off current results in lower efficiency. In that case, connect a Schottky diode, rated for
maximum output current, from PGNDM to LXM to
improve efficiency.
The auxiliary controllers require external rectifiers. For
low-output-voltage applications, use a Schottky diode
to rectify the output voltage because of the diode’s low
forward voltage and fast recovery time. Schottky diodes
exhibit significant leakage current at high reverse voltages and high temperatures. Thus, for high-voltage,
high-temperature applications, use ultra-fast junction
rectifiers.
Compensation Design
Each DC-DC converter has an internal transconductance error amplifier whose output is used to compensate the control loop. Typically, a series resistor and
capacitor are inserted from COMP_ to GND to form a
pole-zero pair. The external inductor, the output capacitor, the compensation resistor and capacitor, and for
the main converter, the external P-channel MOSFET,
govern control-loop stability. The inductor and output
capacitor are usually chosen in consideration of performance, size, and cost, but the compensation resistor
and capacitor are chosen to optimize control-loop stability. The component values in the circuit of Figure 1
yield stable operation over a broad range of input/output voltages and converter switching frequencies.
Follow the procedures below for optimal compensation.
In the following descriptions, Bode plots are used to
graphically describe the loop response of the converters over frequency. The Bode plot shows loop gain and
phase vs. frequency. A single pole results in a -20dB
per decade slope and a -90° phase shift, and a single
zero results in a +20dB per decade slope and a +90°
phase shift. The stability of the system can be determined by the phase margin (how far from 0° the loop
phase is when the response drops to 0dB) and gain
margin (how far below 0dB the gain is when the phase
reaches 0°). The system is stable for phase margins
>30°, and a phase margin of 45° is preferred. The gain
margin should be at least 10dB.
Main Converter
The main converter uses current mode to regulate the
output voltage by forcing the required current through
the inductor. Since the P-channel MOSFET operates
with constant drain-source on-resistance (RDSP), the
voltage across the MOSFET is proportional to the
inductor current. The converter current-sense amplifier
measures the “on” MOSFET drain-source voltage to
determine the inductor current for regulation. The gain
through the current-sense amplifier (measured across
the MOSFET) is AVCSM = 9.3V/V. The voltage-divider
attenuates the loop gain by AVDV = VREF / VOUT, and
the gain DC voltage of the error amplifier is AVEA =
2000V/V. The controller forces the peak inductor current (IL) such that:
IL RDSP AVCSM = VOUT AVDV AVEA
or
IL = VOUT AVDV AVEA / (AVCSM RDSP)
______________________________________________________________________________________
21
MAX1802
the N-channel turn-off current (equal to 17mV/RDSN).
Choose RDSN value between RDSP and 3RDSP to keep
the N-channel turn-off current low for optimal efficiency.
If a lower RDSN is used, connect a Schottky diode from
PGNDM to LXM for better efficiency (see Diode
Selection).
MAX1802
Digital Camera Step-Down
Power Supply
and the output voltage is IOUT RLOAD, which is equal to
IL RLOAD. Thus, the total DC loop gain is:
AVDC = RLOAD AVDV AVEA / (AVCSM RDSP)
180°
or
AVDC
AVDC = 215 VREF RLOAD / (VOUT RDS(ON))
PC
PHASE
Because of the current-mode control, there is a single
pole in the loop response due to the output capacitor.
This pole is at the frequency (in Hz):
90°
ZC = PO
GAIN
(dB)
PHASE
MARGIN
PO = 1 / (2π RLOAD COUT)
GAIN
Note that as the load resistance increases, the pole
moves to a lower frequency. However, the DC loop
gain increases by the same amount since they are both
dependent on RLOAD. Thus, the crossover frequency
(frequency at which the loop gain drops to 0dB), which
is the product of the pole and the gain, remains at the
same frequency.
The compensation network creates a pole and zero at
the frequencies (in Hz):
PC = GEA / (4000π CC) = 1 / (4x107 π CC)
and
ZC = 1 / (2π RC CC)
and the ESR of the output filter capacitor causes a zero
in the loop response at the frequency (in Hz):
ZO = 1 / (2π COUT ESR)
The DC gain and the poles and zeros are shown in the
Bode plot of Figure 6.
To achieve a stable circuit with the Bode plot of Figure
6, use the following procedure:
1) Determine the desired crossover frequency, either
1/3 of the zero due to the output capacitor ESR:
fC = Z O / 3 =
1
6 π COUT ESR
or 1/5 of the switching frequency:
f
fC = SW
5
whichever is lower.
2) Determine the pole frequency due to the output
capacitor and the load resistor:
PO =
22
1
PHASE
O
0°
Z0
FREQUENCY
Figure 6. Current-Mode Step-Down Converter Bode Plot
or
PO =
ILOAD(MAX)
2 π VOUT COUT
3) Determine the compensation resistor required to set
the desired crossover frequency:
RC =
20MΩ fC
A VDC PO
or, by simplifying and using the typical VREF = 1.25V:
RC = 468kΩ/V VOUT COUT RDSP fC
4) Determine the compensation capacitor to set the
proper error-amplifier pole and zero determined from
the above equations:
CC =
1
2 π RC PO
Core Converter
Compensating the core converter is similar to the compensation of the main converter described above. The
only difference is that the current is measured internally, and the gain (transresistance) of the current-sense
amplifier is RCSC = 1.0V/A. The DC loop gain is:
AVDC = 2000 VREF RLOAD / VOUT
2 π RLOAD(MIN) COUT
______________________________________________________________________________________
Digital Camera Step-Down
Power Supply
1) Determine the desired crossover frequency, either
1/3 of the zero due to the output capacitor ESR:
fC =
ZO
1
=
3
6 π COUT ESR
or 1/5 of the switching frequency:
f
fC = SW
5
whichever is lower.
1
2 π RLOAD(MIN) COUT
or
PO =
D = (2 L fOSC / RE)1/2
where RE is the equivalent load resistance, or:
RE = VIN2 RLOAD / (VOUT (VOUT - VIN))
The frequency of single pole due to the PWM converter
is:
PO = (2 VOUT - VIN) / (2π (VOUT - VIN) RLOAD COUT)
2) Determine the pole frequency due to the output
capacitor and the load resistor:
PO =
Discontinuous Inductor Current
For discontinuous inductor current, the PWM controller
has a single pole. The pole frequency and DC gain of
the PWM controller are dependent on the operating
duty cycle, which is:
ILOAD(MAX)
2 π VOUT COUT
3) Determine the compensation resistor required to set
the desired crossover frequency:
RC =
20MΩ fC
A VDC PO
or, by simplifying and using the typical VREF = 1.25V:
and the DC gain of the PWM controller is:
AVO = 2 VOUT (VOUT - VIN) RLOAD / ((2 VOUT - VIN) D)
Note that, as in the current-mode, step-down cases
above, as R LOAD is increased, the pole frequency
decreases and the DC gain increases proportionally.
Since the crossover frequency is the product of the
pole frequency and the DC gain, it remains independent of the load.
As in the cases of the main and core converters, the gain
through the voltage-divider is AVDV = VREF / VOUT, and
the DC gain of the error amplifier is AVEA = 2000V/V.
Thus, the DC loop gain is AVDC = AVDV AVEA AVO.
The compensation resistor-capacitor pair at COMP
cause a pole and zero at frequencies (in Hz):
PC = GEA / (4000π CC) = 1 / (4x107 π CC)
ZC = 1 / (2π RC CC)
RC = 50kΩ/V VOUT COUT fC
4) Determine the compensation capacitor to set the
proper error-amplifier pole and zero determined from
the above equations:
CC =
1
2 π RC PO
Auxiliary Controllers
The auxiliary controllers use voltage mode to regulate
their output voltages. The following explains how to
compensate the control system for optimal performance. The compensation differs depending on
whether the inductor current is continuous or discontinuous.
and the ESR of the output filter capacitor causes a zero
in the loop response at the frequency (in Hz): ZO = 1 /
(2π COUT ESR).
The DC gain and the poles and zeros are shown in the
Bode plot of Figure 7. To achieve a stable circuit with
the Bode plot of Figure 7, follow the procedure below:
1) Choose the RC that is equivalent to the inverse of
the transconductance of the error amplifier, 1 / RC =
GEA = 100µs, or RC = 10kΩ. This sets the high-frequency voltage gain of the error amplifier to 0dB.
2) Determine the maximum output pole frequency:
PO(MAX) =
2VOUT − VIN
2 π (VOUT − VIN )RLOAD(MIN) COUT
where RLOAD(MIN) = VOUT / IOUT(MAX).
______________________________________________________________________________________
23
MAX1802
To achieve a stable circuit for the core converter, use
the following procedure:
MAX1802
Digital Camera Step-Down
Power Supply
3) Place the compensation zero at the same frequency
as the maximum output pole frequency (in Hz):
ZC =
1
2VOUT − VIN
=
2πRC CC 2 π (VOUT − VIN )RLOAD(MIN) COUT
180°
80
AVDC
60
PC
PHASE
40
Solving for CC:


VOUT − VIN
CC = COUT VOUT 

 RC IOUT(MAX) (2VOUT − VIN ) 
90°
ZC = PO
GAIN
(dB)
PHASE
20
GAIN
O
Use values of CC <10nF. If the above calculation determines that the capacitor should be >10nF, use CC =
10nF, skip step 4, and go to step 5.
4) Determine the crossover frequency (in Hz):
VREF
fC =
π DCOUT
and to maintain at least 10dB gain margin, make sure
that the crossover frequency is ≤1/3 of the ESR zero
frequency, or 3fC ≤ ZO, or ESR ≤ D / 6 VREF.
If this is not the case, go to step 5 to reduce the erroramplifier high-frequency gain to decrease the
crossover frequency.
5) The high-frequency gain may be reduced, thus
reducing the crossover frequency, as long as the
zero due to the compensation network remains at or
below the crossover frequency. In this case:
ESR ≤
0°
Z0
-20
FREQUENCY
Figure 7. Discontinuous-Current, Voltage-Mode, Step-Up
Controller Bode Plot
increasing the phase margin. If a low-value, low-ESR
output capacitor (such as a ceramic capacitor) is used,
the ESR-related zero occurs at too high a frequency
and does not increase the phase margin. In this case,
use a lower value inductor so that it operates with discontinuous current (see the Discontinuous Inductor
Current section).
For continuous inductor current, the gain of the voltage
divider is AVDV = VREF / VOUT, and the DC gain of the
error amplifier is AVEA = 2000. The gain through the
PWM controller in continuous current is:
A VO =
D
GEA RC 6VREF
VOUT 2
VIN VREF
Thus, the total DC loop gain is: AVDC = 2000 VOUT / VIN.
and
fC =
GEA RC VREF
π DCOUT
1
≥
2π RC CC
Choose COUT, RC, and CC to satisfy both equations
simultaneously.
Continuous Inductor Current
For continuous inductor current, there are two conditions that change, requiring different compensation.
The response of the control loop includes a right-halfplane zero and a complex pole pair due to the inductor
and output capacitor. For stable operation, the controller-loop gain must drop below unity (0dB) at a much
lower frequency than the right-half-plane zero frequency. The zero arising from the ESR of the output capacitor is typically used to compensate the control circuit
by increasing the phase near the crossover frequency,
24
The complex pole pair due to the inductor and output
capacitor occurs at the frequency (in Hz):
PO =
VOUT
2πVIN LCOUT
The pole and zero due to the compensation network at
COMP occur at the frequencies (in Hz):
PC =
1
GEA
=
4000
π
C
(
4 × 107 πCC
C)
ZC =
1
2πRC CC
______________________________________________________________________________________
Digital Camera Step-Down
Power Supply
AVDC
30
180°
PC
GAIN
PHASE
ZC = PO
20
90°
GAIN
(dB)
PHASE
4) Since response is 2nd order (-40dB per decade)
between the complex pole pair and the ESR zero,
determine the desired amplitude at the complex
pole pair to force the crossover frequency equal to
the ESR zero frequency. Thus:
A(PO ) = (Z O / PO ) =
L VIN2
2
COUT ESR2 VOUT 2
10
PHASE
MARGIN
Z0
0
GAIN
MARGIN
0°
ZRHP
-10
FREQUENCY
5) Determine the desired compensation pole. Since
the response between the compensation pole and
the complex pole pair is 1st order (-20dB per
decade), the ratio of the frequencies is equal to the
ratio of the amplitudes at those frequencies. Thus:
PO
A
= DC
PC A(PO )
Figure 8. Continuous-Current, Voltage-Mode, Step-Up
Converter Bode Plot
The frequency (in Hz) of the zero due to the ESR of the
output capacitor is:
ZO =
1
2πCOUT ESR
and the right-half-plane zero frequency (in Hz) is:
ZRHP =
(1-D)2 RLOAD
2πL
Figure 8 shows the Bode plot of the loop gain of this
control circuit.
To configure the compensation network for a stable
control loop, set the crossover frequency at that of the
zero due to the output capacitor ESR. Use the following
procedure:
1) Determine the frequency of the right-half-plane
zero:
ZRHP
2
1-D) RLOAD
(
=
2πL
2) Find the DC loop gain:
A VDC =
2000VOUT
VIN
3) Determine the frequency of the complex pole pair
due to the inductor and output capacitor:
VOUT
fO =
2π VIN LCOUT
Solving this equation for CC:
CC =
VOUT (COUT )
3/2
ESR2
20MΩ VIN (L)
1/ 2
6) Determine RC for the compensation zero frequency
as equal to the complex pole-pair frequency:
ZC = PO.
Solving for RC:
RC =
VIN LCOUT
VOUT CC
Applications Information
Using the MAX1801 with the MAX1802
Step-Down Master
The MAX1801 is a slave DC-DC controller that can be
used with the MAX1802 to generate additional output
voltages. The MAX1801 does not generate its own reference or oscillator. Instead it uses the reference and
oscillator from the MAX1802 step-down master converter controller (Figure 1). MAX1801 controller operation
and design is similar to that of the MAX1802 auxiliary
controllers. For more details, refer to the MAX1801 data
sheet.
Using an Auxiliary Controller in an
SEPIC Configuration
Where the battery voltage may be above or below the
required output voltage, neither a step-up nor a stepdown converter is suitable; instead, use a step-up/stepdown converter. One type of step-up/step-down
______________________________________________________________________________________
25
MAX1802
40
MAX1802
Digital Camera Step-Down
Power Supply
converter is the SEPIC, shown in Figure 9. Inductors L1
and L2 can be separate inductors or can be wound on
a single core and coupled like a transformer. Typically,
using a coupled inductor will improve efficiency since
some power is transferred through the coupling, so less
power passes through the coupling capacitor (C2).
Likewise, C2 should have low ESR to improve efficiency. The ripple current rating must be greater than the
larger of the input and output currents. The MOSFET
(Q1) drain-source voltage rating and the rectifier (D1)
reverse-voltage rating must exceed the sum of the
input and output voltages. Other types of step-up/stepdown circuits are a flyback converter and a step-up
converter followed by a linear regulator.
Using an Auxiliary Controller for a
Multi-Output Flyback Circuit
Some applications require multiple voltages from a single converter that features a flyback transformer.
Figure 10 shows a MAX1802 auxiliary controller in a
two-output flyback configuration. The controller drives
an external MOSFET that switches the transformer primary, and the two secondaries generate the outputs.
Only a single positive output voltage can be regulated
using the feedback resistive voltage-divider, so the
other voltages are set by the turns ratio of the transformer secondaries. The regulation of the other secondary voltages degrades due to transformer leakage
inductance and winding resistance. Voltage regulation
is best when the load current is limited to a small range.
Consult the transformer manufacturer for the proper
design for a given application.
Using a Charge Pump for Negative
Output Voltages
Negative output voltages can be produced without a
transformer using a charge-pump circuit with an auxiliary controller as shown in Figure 11. When MOSFET
Q1 turns off, the voltage at its drain rises to supply current to VOUT+. At the same time, C1 charges to the voltage at VOUT+ through D1. When the MOSFET turns on,
C1 discharges through D3, thereby charging C3 to
VOUT- minus the drop across D3 to create roughly the
same voltage as V OUT+ at V OUT- but with inverted
polarity. If different magnitudes are required for the
positive and negative voltages, a linear regulator can
be used at one of the outputs to achieve the desired
voltage.
INPUT
1 CELL
Li+
L2
L1
MAIN
ON
DCON
EXT
C2
Q1
D1
R1
MAX1802
FB
COMP
R2
RC
GC
Figure 9. Auxiliary Controller, SEPIC Configuration
Conductors carrying discontinuous currents should be
kept as short as possible. Conductors carrying high
currents should be made as wide as possible. A separate low-noise ground plane containing the reference
and signal grounds should only connect to the powerground plane at one point to minimize the effects of
power-ground currents.
Keep the voltage feedback network very close to the
IC, preferably within 0.2in (5mm) of the FB_ pin. Nodes
with high dv/dt (switching nodes) should be kept as
small as possible and should stay away from highimpedance nodes such as FB_ and COMP_.
Refer to the MAX1802EVKIT evaluation kit manual for a
full PC board example.
Chip Information
TRANSISTOR COUNT: 7740
Designing a PC Board
A good PC board layout is important to achieve optimal
performance from the MAX1802. Good design reduces
excessive conducted and/or radiated noise, both of
which are undesirable.
26
OUTPUT
3.3V
______________________________________________________________________________________
Digital Camera Step-Down
Power Supply
MAX1802
+ OUTPUT
INPUT
1 CELL
Li+
D3
INPUT
1 CELL
Li+
VOUTC3
D1
L
Main
C1
MAIN
D22
- OUTPUT
ON
ON
DCON
EXT
EXT
VOUT+
DCON
Q1
Q1
C2
R1
R1
MAX1802
FB
MAX1802
FB
COMP
COMP
R2
R2
RC
RC
GC
GC
Figure 10. Auxiliary Controller, Flyback Configuration
Figure 11. Auxiliary Controller, Charge-Pump Configuration
Pin Configuration
GND
REF
FBC
COMPC
VL
VDDC
LXC
PGND
TOP VIEW
32
31
30
29
28
27
26
25
FBM
1
24 ONC
COMPM
2
23 DCON3
ONM
3
22 FB3
VH
4
VDDM
5
21 COMP3
DHM
6
19 DL2
LXM
7
18 DCON2
DLM
8
17 COMP2
15
16
FB2
DCON1
14
FB1
OSC
12 13
COMP1
11
DL1
10
20 DL3
ON1
9
PGNDM
MAX1802
TQFP
______________________________________________________________________________________
27
Digital Camera Step-Down
Power Supply
32L TQFP, 5x5x01.0.EPS
MAX1802
Package Information
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
28 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600
© 2000 Maxim Integrated Products
Printed USA
is a registered trademark of Maxim Integrated Products.