AD AD5737 Quad-channel, 12-bit, serial input, 4 ma to 20 ma Datasheet

Quad-Channel, 12-Bit, Serial Input, 4 mA to 20 mA
and Voltage Output DAC with Dynamic Power Control
AD5735
Data Sheet
FEATURES
On-chip dynamic power control minimizes package power
dissipation in current mode. This reduced power dissipation
is achieved by regulating the voltage on the output driver from
7.4 V to 29.5 V using a dc-to-dc boost converter optimized for
minimum on-chip power dissipation.
12-bit resolution and monotonicity
Dynamic power control for thermal management
Current and voltage output pins connectable to a single
terminal
Current output ranges: 0 mA to 20 mA, 4 mA to 20 mA,
and 0 mA to 24 mA
±0.1% total unadjusted error (TUE) maximum
Voltage output ranges (with 20% overrange): 0 V to 5 V,
0 V to 10 V, ±5 V, and ±10 V
±0.09% total unadjusted error (TUE) maximum
User-programmable offset and gain
On-chip diagnostics
On-chip reference: ±10 ppm/°C maximum
−40°C to +105°C temperature range
The AD5735 uses a versatile 3-wire serial interface that operates
at clock rates of up to 30 MHz and is compatible with standard
SPI, QSPI™, MICROWIRE®, DSP, and microcontroller interface
standards. The serial interface also features optional CRC-8 packet
error checking, as well as a watchdog timer that monitors activity
on the interface.
PRODUCT HIGHLIGHTS
1.
2.
3.
APPLICATIONS
Dynamic power control for thermal management.
12-bit performance.
Quad channel.
COMPANION PRODUCTS
Process control
Actuator control
PLCs
Product Family: AD5755, AD5755-1, AD5757, AD5737
External References: ADR445, ADR02
Digital Isolators: ADuM1410, ADuM1411
Power: ADP2302, ADP2303
Additional companion products on the AD5735 product page
GENERAL DESCRIPTION
The AD5735 is a quad-channel voltage and current output DAC
that operates with a power supply range from −26.4 V to +33 V.
FUNCTIONAL BLOCK DIAGRAM
AVCC
5.0V
AVSS
–15V
AGND
AVDD
+15V
SWx
DVDD
VBOOST_x
7.4V TO 29.5V
DGND
LDAC
DC-TO-DC
CONVERTER
SCLK
SDIN
SYNC
SDO
CLEAR
DIGITAL
INTERFACE
IOUT_x
+
FAULT
ALERT
GAIN REG A
OFFSET REG A
AD1
AD0
DAC A
CURRENT AND
VOLTAGE
OUTPUT RANGE
SCALING
RSET_x
+VSENSE_x
VOUT_x
–VSENSE_x
DAC CHANNEL A
REFOUT
REFERENCE
REFIN
DAC CHANNEL B
DAC CHANNEL C
DAC CHANNEL D
09961-100
AD5735
NOTES
1. x = A, B, C, OR D.
Figure 1.
Rev. B
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
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Tel: 781.329.4700
www.analog.com
Fax: 781.461.3113 ©2011–2012 Analog Devices, Inc. All rights reserved.
AD5735
Data Sheet
TABLE OF CONTENTS
Features .............................................................................................. 1
Control Registers ........................................................................ 34
Applications ....................................................................................... 1
Readback Operation .................................................................. 37
General Description ......................................................................... 1
Device Features ............................................................................... 39
Product Highlights ........................................................................... 1
Fault Output ................................................................................ 39
Companion Products ....................................................................... 1
Voltage Output Short-Circuit Protection ................................ 39
Functional Block Diagram .............................................................. 1
Digital Offset and Gain Control ............................................... 39
Revision History ............................................................................... 2
Status Readback During a Write .............................................. 39
Detailed Functional Block Diagram .............................................. 3
Asynchronous Clear................................................................... 40
Specifications..................................................................................... 4
Packet Error Checking ............................................................... 40
AC Performance Characteristics ................................................ 7
Watchdog Timer ......................................................................... 40
Timing Characteristics ................................................................ 8
Alert Output ................................................................................ 40
Absolute Maximum Ratings.......................................................... 11
Internal Reference ...................................................................... 40
Thermal Resistance .................................................................... 11
External Current Setting Resistor ............................................ 40
ESD Caution ................................................................................ 11
Digital Slew Rate Control .......................................................... 41
Pin Configuration and Function Descriptions ........................... 12
Dynamic Power Control............................................................ 41
Typical Performance Characteristics ........................................... 15
DC-to-DC Converters ............................................................... 42
Voltage Outputs .......................................................................... 15
AICC Supply Requirements—Static .......................................... 43
Current Outputs ......................................................................... 19
AICC Supply Requirements—Slewing ...................................... 43
DC-to-DC Converter ................................................................. 23
Applications Information .............................................................. 45
Reference ..................................................................................... 24
Voltage and Current Output Pins on the Same Terminal ..... 45
General ......................................................................................... 25
Current Output Mode with Internal RSET ................................ 45
Terminology .................................................................................... 26
Precision Voltage Reference Selection ..................................... 45
Theory of Operation ...................................................................... 28
Driving Inductive Loads ............................................................ 46
DAC Architecture ....................................................................... 28
Transient Voltage Protection .................................................... 46
Power-On State of the AD5735 ................................................ 29
Microprocessor Interfacing ....................................................... 46
Serial Interface ............................................................................ 29
Layout Guidelines....................................................................... 46
Transfer Function ....................................................................... 29
Galvanically Isolated Interface ................................................. 47
Registers ........................................................................................... 30
Outline Dimensions ....................................................................... 48
Enabling the Output ................................................................... 31
Ordering Guide .......................................................................... 48
Data Registers ............................................................................. 32
REVISION HISTORY
5/12—Rev. A to Rev. B
Changes to Figure 2 ........................................................................... 3
Changes to Power-On State of the AD5735 Section .................. 29
Changes to Readback Operation Section .................................... 37
11/11—Rev. 0 to Rev. A
7/11—Revision 0: Initial Version
Added Comments to OUTPUT CHARACTERISTICS and
ACCURACY, CURRENT OUTPUT Parameters in
Table 1 ................................................................................................. 4
Rev. B | Page 2 of 48
Data Sheet
AD5735
DETAILED FUNCTIONAL BLOCK DIAGRAM
AVCC
5.0V
AVSS
–15V
DVDD
DGND
LDAC
CLEAR
SCLK
SDIN
SYNC
SDO
AGND
SWA
VBOOST_A
POWER-ON
RESET
DC-TO-DC
CONVERTER
DYNAMIC
POWER
CONTROL
INPUT
SHIFT
REGISTER
AND
CONTROL
FAULT
STATUS
REGISTER
ALERT
AVDD
+15V
12
DAC
DATA
REG A
+
DAC
INPUT
REG A
12
7.4V TO 29.5V VSEN1
R2
VSEN2
R3
DAC A
GAIN REG A
OFFSET REG A
IOUT_A
RSET_A
R1
WATCHDOG
TIMER
(SPI ACTIVITY)
30kΩ
REFOUT
REFIN
VOUT
RANGE
SCALING
VREF
REFERENCE
BUFFERS
+VSENSE_A
VOUT_A
DAC CHANNEL A
–VSENSE_A
AD1
AD0
AD5735
RSET_B, RSET_C, RSET_D
DAC CHANNEL D
±V SENSE_B, ±VSENSE_C, ±VSENSE_D
VOUT_B, VOUT_C, VOUT_D
SWB, SWC, SWD
Figure 2.
Rev. B | Page 3 of 48
VBOOST_B, VBOOST_C, VBOOST_D
09961-001
IOUT_B, IOUT_C, IOUT_D
DAC CHANNEL B
DAC CHANNEL C
AD5735
Data Sheet
SPECIFICATIONS
AVDD = VBOOST_x = 15 V; AVSS = −15 V; DVDD = 2.7 V to 5.5 V; AVCC = 4.5 V to 5.5 V; dc-to-dc converter disabled; AGND = DGND =
GNDSWx = 0 V; REFIN = 5 V; voltage outputs: RL = 1 kΩ, CL = 220 pF; current outputs: RL = 300 Ω; all specifications TMIN to TMAX,
unless otherwise noted.
Table 1.
Parameter 1
VOLTAGE OUTPUT
Output Voltage Ranges
Resolution
ACCURACY, VOLTAGE OUTPUT
Total Unadjusted Error (TUE)
TUE Long-Term Stability
Relative Accuracy (INL)
Differential Nonlinearity (DNL)
Zero-Scale Error
Zero-Scale TC 2
Bipolar Zero Error
Bipolar Zero TC2
Offset Error
Offset TC2
Gain Error
Gain TC2
Full-Scale Error
Min
0
0
−5
−10
0
0
−6
−12
12
−0.09
−0.13
−0.032
−1
−0.05
−0.08
−0.05
−0.08
−0.065
−0.09
−0.08
−0.15
−0.09
−0.13
Full-Scale TC2
OUTPUT CHARACTERISTICS,
VOLTAGE OUTPUT2
Headroom
Footroom
Output Voltage Drift vs. Time
Short-Circuit Current
Resistive Load
Capacitive Load Stability
DC Output Impedance
DC PSRR
DC Crosstalk
CURRENT OUTPUT
Output Current Ranges
Resolution
Typ
±0.012
±0.05
35
±0.006
±0.004
±0.004
±2
±0.003
±0.03
±2
±0.005
±0.03
±2
±0.004
±0.004
±3
±0.01
±0.05
±2
1
1
20
12/6
1
Max
Unit
5
10
+5
+10
6
12
+6
+12
V
V
V
V
V
V
V
V
Bits
+0.09
+0.13
% FSR
% FSR
ppm FSR
% FSR
LSB
% FSR
% FSR
ppm FSR/°C
% FSR
% FSR
ppm FSR/°C
% FSR
% FSR
ppm FSR/°C
% FSR
% FSR
ppm FSR/°C
% FSR
% FSR
ppm FSR/°C
0 V to 5 V, 0 V to 10 V, ±5 V, ±10 V ranges
On overranges (0 V to 6 V, 0 V to 12 V, ±6 V, ±12 V)
Drift after 1000 hours, TJ = 150°C
V
V
ppm FSR
With respect to VBOOST supply
With respect to the AVSS supply
+0.032
+1
+0.05
+0.08
+0.05
+0.08
+0.065
+0.09
+0.08
+0.15
+0.09
+0.13
2.2
1.4
16/8
10
2
0.06
50
24
0
0
4
12
24
20
20
Rev. B | Page 4 of 48
mA
kΩ
nF
µF
Ω
µV/V
µV
mA
mA
mA
Bits
Test Conditions/Comments
Guaranteed monotonic
0 V to 5 V, 0 V to 10 V ranges
On overranges (0 V to 6 V, 0 V to 12 V)
±5 V, ±10 V ranges
On overranges (±6 V, ±12 V)
0 V to 5 V, 0 V to 10 V, ±5 V, ±10 V ranges
On overranges (0 V to 6 V, 0 V to 12 V, ±6 V, ±12 V)
0 V to 5 V, 0 V to 10 V, ±5 V, ±10 V ranges
On overranges (0 V to 6 V, 0 V to 12 V, ±6 V, ±12 V)
0 V to 5 V, 0 V to 10 V, ±5 V, ±10 V ranges
On overranges (0 V to 6 V, 0 V to 12 V, ±6 V, ±12 V)
Drift after 1000 hours, ¾ scale output, TJ = 150°C,
AVSS = −15 V
Programmable by user; defaults to 16 mA typical
For specified performance
External 220 pF compensation capacitor connected
Data Sheet
Parameter 1
ACCURACY, CURRENT OUTPUT
(EXTERNAL RSET)
Total Unadjusted Error (TUE)
TUE Long-Term Stability
Relative Accuracy (INL)
Differential Nonlinearity (DNL)
Offset Error
Offset Error Drift2
Gain Error
Gain TC2
Full-Scale Error
Full-Scale TC2
DC Crosstalk
ACCURACY, CURRENT OUTPUT
(INTERNAL RSET)
Total Unadjusted Error (TUE) 3, 4
TUE Long-Term Stability
Relative Accuracy (INL)
Differential Nonlinearity (DNL)
Offset Error3, 4
Offset Error Drift2
Gain Error
Gain TC2
Full-Scale Error3, 4
Full-Scale TC2
DC Crosstalk4
OUTPUT CHARACTERISTICS,
CURRENT OUTPUT2
Current Loop Compliance Voltage
AD5735
Min
Typ
Max
Unit
−0.1
±0.019
100
±0.006
+0.1
% FSR
ppm FSR
% FSR
LSB
% FSR
ppm FSR/°C
% FSR
ppm FSR/°C
% FSR
ppm FSR/°C
% FSR
−0.032
−1
−0.1
−0.1
−0.1
−0.14
−0.032
−1
−0.1
−0.12
−0.14
±0.012
±4
±0.004
±3
±0.014
±5
0.0005
±0.022
180
±0.006
±0.017
±6
±0.004
±9
±0.02
±14
−0.011
VBOOST_x −
2.4
+0.032
+1
+0.1
+0.1
+0.1
+0.14
+0.032
+1
+0.1
+0.12
+0.14
VBOOST_x −
2.7
% FSR
ppm FSR
% FSR
LSB
% FSR
ppm FSR/°C
% FSR
ppm FSR/°C
% FSR
ppm FSR/°C
% FSR
Resistive Load
DC Output Impedance
DC PSRR
REFERENCE INPUT/OUTPUT
Reference Input2
Reference Input Voltage
DC Input Impedance
Reference Output
Output Voltage
Reference TC2
Output Noise (0.1 Hz to 10 Hz)2
Noise Spectral Density2
Output Voltage Drift vs. Time2
Capacitive Load2
Load Current
Short-Circuit Current
Line Regulation2
Load Regulation2
Thermal Hysteresis2
100
0.02
Drift after 1000 hours, TJ = 150°C
Guaranteed monotonic
External RSET
Drift after 1000 hours, TJ = 150°C
Guaranteed monotonic
Internal RSET
V
Output Current Drift vs. Time
90
140
Test Conditions/Comments
Assumes ideal resistor, see External Current
Setting Resistor section for more information.
1000
ppm FSR
ppm FSR
Ω
1
MΩ
µA/V
Drift after 1000 hours, ¾ scale output, TJ = 150°C
External RSET
Internal RSET
The dc-to-dc converter has been characterized
with a maximum load of 1 kΩ, chosen such that
compliance is not exceeded; see Figure 51 and
the DC-DC MaxV bits in Table 28
4.95
45
5
150
5.05
V
MΩ
For specified performance
4.995
−10
5
±5
7
100
180
1000
9
10
3
95
160
5
5.005
+10
V
ppm/°C
µV p-p
nV/√Hz
ppm
nF
mA
mA
ppm/V
ppm/mA
ppm
ppm
TA = 25°C
Rev. B | Page 5 of 48
At 10 kHz
Drift after 1000 hours, TJ = 150°C
See Figure 62
See Figure 63
See Figure 62
First temperature cycle
Second temperature cycle
AD5735
Parameter 1
DC-TO-DC CONVERTER
Switch
Switch On Resistance
Switch Leakage Current
Peak Current Limit
Oscillator
Oscillator Frequency
Maximum Duty Cycle
DIGITAL INPUTS2
Input High Voltage, VIH
Input Low Voltage, VIL
Input Current
Pin Capacitance
DIGITAL OUTPUTS2
SDO, ALERT Pins
Output Low Voltage, VOL
Output High Voltage, VOH
High Impedance Leakage Current
High Impedance Output
Capacitance
FAULT Pin
Output Low Voltage, VOL
Data Sheet
Min
Typ
Max
0.425
10
0.8
11.5
13
14.5
AISS
2
0.8
+1
2.6
0.4
0.4
V
V
V
10 kΩ pull-up resistor to DVDD
At 2.5 mA
10 kΩ pull-up resistor to DVDD
33
−10.8
5.5
5.5
10.5
V
V
V
V
mA
7
−8.8
7.5
mA
mA
9.2
11
mA
mA
1
2.7
mA
mA
1
mA
mW
+1
2.5
3.6
9
−26.4
2.7
4.5
8.6
AICC
IBOOST 5
Power Dissipation
Per pin
Per pin
Sinking 200 µA
Sourcing 200 µA
−1.7
DICC
V
V
µA
pF
This oscillator is divided down to provide the
dc-to-dc converter switching frequency
At 410 kHz dc-to-dc switching frequency
JEDEC compliant
V
V
µA
pF
DVDD − 0.5
−1
−11
MHz
%
0.6
Output High Voltage, VOH
POWER REQUIREMENTS
AVDD
AVSS
DVDD
AVCC
AIDD
Test Conditions/Comments
Ω
nA
A
89.6
−1
Unit
173
Voltage output mode on all channels, outputs
unloaded, over supplies
Current output mode on all channels
Voltage output mode on all channels, outputs
unloaded, over supplies
Current output mode on all channels
VIH = DVDD, VIL = DGND, internal oscillator running,
over supplies
Outputs unloaded, over supplies
Per channel, voltage output mode, outputs
unloaded, over supplies
Per channel, current output mode
AVDD = 15 V, AVSS = −15 V, dc-to-dc converter
enabled, current output mode, outputs disabled
Temperature range: −40°C to +105°C; typical at +25°C.
Guaranteed by design and characterization; not production tested.
3
For current outputs with internal RSET, the offset, full-scale, and TUE measurements exclude dc crosstalk. The measurements are made with all four channels enabled
and loaded with the same code.
4
See the Current Output Mode with Internal RSET section for more information about dc crosstalk.
5
Efficiency plots in Figure 53 through Figure 56 include the IBOOST quiescent current.
1
2
Rev. B | Page 6 of 48
Data Sheet
AD5735
AC PERFORMANCE CHARACTERISTICS
AVDD = VBOOST_x = 15 V; AVSS = −15 V; DVDD = 2.7 V to 5.5 V; AVCC = 4.5 V to 5.5 V; dc-to-dc converter disabled; AGND = DGND =
GNDSWx = 0 V; REFIN = 5 V; voltage outputs: RL = 2 kΩ, CL = 220 pF; current outputs: RL = 300 Ω; all specifications TMIN to TMAX,
unless otherwise noted.
Table 2.
Parameter 1
DYNAMIC PERFORMANCE, VOLTAGE
OUTPUT
Output Voltage Settling Time
Min
Typ
Max
11
Test Conditions/Comments
5 V step to ±0.03% FSR, 0 V to 5 V range
10 V step to ±0.03% FSR, 0 V to 10 V range
0 V to 10 V range
Slew Rate
Power-On Glitch Energy
Digital-to-Analog Glitch Energy
Glitch Impulse Peak Amplitude
Digital Feedthrough
DAC-to-DAC Crosstalk
Output Noise (0.1 Hz to 10 Hz
Bandwidth)
Output Noise Spectral Density
1.9
150
6
25
1
2
0.01
µs
µs
V/µs
nV-sec
nV-sec
mV
nV-sec
nV-sec
LSB p-p
150
nV/√Hz
AC PSRR
83
dB
18
DYNAMIC PERFORMANCE, CURRENT
OUTPUT
Output Current Settling Time
Output Noise (0.1 Hz to 10 Hz
Bandwidth)
Output Noise Spectral Density
1
Unit
15
See Test Conditions/Comments
µs
ms
0.01
LSB p-p
0.5
nA/√Hz
Guaranteed by design and characterization; not production tested.
Rev. B | Page 7 of 48
0 V to 10 V range
12-bit LSB, 0 V to 10 V range
Measured at 10 kHz, midscale output, 0 V to
10 V range
200 mV, 50 Hz/60 Hz sine wave superimposed
on power supply voltage
To 0.1% FSR, 0 mA to 24 mA range
For settling times when using the dc-to-dc converter, see Figure 47, Figure 48, and Figure 49
12-bit LSB, 0 mA to 24 mA range
Measured at 10 kHz, midscale output, 0 mA
to 24 mA range
AD5735
Data Sheet
TIMING CHARACTERISTICS
AVDD = VBOOST_x = 15 V; AVSS = −15 V; DVDD = 2.7 V to 5.5 V; AVCC = 4.5 V to 5.5 V; dc-to-dc converter disabled; AGND = DGND =
GNDSWx = 0 V; REFIN = 5 V; voltage outputs: RL = 1 kΩ, CL = 220 pF; current outputs: RL = 300 Ω; all specifications TMIN to TMAX,
unless otherwise noted.
Table 3.
Parameter1, 2, 3
t1
t2
t3
t4
t5
t6
t7
t8
t9
t10
t11
t12
t13
t14
t15
t16
t17
t18
t194
Limit at TMIN, TMAX
33
13
13
13
13
198
5
5
20
Unit
ns min
ns min
ns min
ns min
ns min
ns min
ns min
ns min
µs min
5
10
500
See Table 2
10
5
40
µs min
ns min
ns max
µs max
ns min
µs max
ns max
21
5
500
800
µs min
µs min
ns min
ns min
20
5
µs min
µs min
Description
SCLK cycle time
SCLK high time
SCLK low time
SYNC falling edge to SCLK falling edge setup time
24th/32nd SCLK falling edge to SYNC rising edge (see Figure 76)
SYNC high time
Data setup time
Data hold time
SYNC rising edge to LDAC falling edge (all DACs updated or any channel has
digital slew rate control enabled)
SYNC rising edge to LDAC falling edge (single DAC updated)
LDAC pulse width low
LDAC falling edge to DAC output response time
DAC output settling time
CLEAR high time
CLEAR activation time
SCLK rising edge to SDO valid
SYNC rising edge to DAC output response time (LDAC = 0)
All DACs updated
Single DAC updated
LDAC falling edge to SYNC rising edge
RESET pulse width
SYNC high to next SYNC low (digital slew rate control enabled)
All DACs updated
Single DAC updated
Guaranteed by design and characterization; not production tested.
All input signals are specified with tRISE = tFALL = 5 ns (10% to 90% of DVDD) and timed from a voltage level of 1.2 V.
3
See Figure 3, Figure 4, Figure 5, and Figure 6.
4
This specification applies if LDAC is held low during the write cycle; otherwise, see t9.
1
2
Rev. B | Page 8 of 48
Data Sheet
AD5735
Timing Diagrams
t1
SCLK
1
2
24
t3
t6
t2
t4
t5
SYNC
t8
t7
SDIN
t19
MSB
LSB
t10
t10
t9
LDAC
t17
t12
t11
VOUT_x
LDAC = 0
t12
t16
VOUT_x
t13
CLEAR
t14
VOUT_x
09961-002
t18
RESET
Figure 3. Serial Interface Timing Diagram
SCLK
1
1
24
24
t6
SYNC
MSB
LSB
MSB
LSB
INPUT WORD SPECIFIES
REGISTER TO BE READ
NOP CONDITION
MSB
SDO
LSB
UNDEFINED
t15
Figure 4. Readback Timing Diagram
Rev. B | Page 9 of 48
SELECTED REGISTER DATA
CLOCKED OUT
09961-003
SDIN
AD5735
Data Sheet
LSB
1
MSB
16
2
SCLK
SDO
R/W
DUT_
AD1
DUT_
AD0
SDO DISABLED
X
X
X
D15
D14
D1
D0
SDO_
ENAB
STATUS
STATUS
STATUS
STATUS
Figure 5. Status Readback During Write, Timing Diagram
200µA
TO OUTPUT
PIN
IOL
VOH (MIN) OR
VOL (MAX)
CL
50pF
200µA
IOH
Figure 6. Load Circuit for SDO Timing Diagrams
Rev. B | Page 10 of 48
09961-005
SDIN
09961-004
SYNC
Data Sheet
AD5735
ABSOLUTE MAXIMUM RATINGS
TA = 25°C, unless otherwise noted. Transient currents of up to
100 mA do not cause SCR latch-up.
Table 4.
Parameter
AVDD, VBOOST_x to AGND, DGND
AVSS to AGND, DGND
AVDD to AVSS
AVCC to AGND
DVDD to DGND
Digital Inputs to DGND
Digital Outputs to DGND
REFIN, REFOUT to AGND
VOUT_x to AGND
+VSENSE_x, −VSENSE_x to AGND
IOUT_x to AGND
SWx to AGND
AGND, GNDSWx to DGND
Operating Temperature Range (TA)
Industrial1
Storage Temperature Range
Junction Temperature (TJ max)
Power Dissipation
Lead Temperature
Soldering
1
Rating
−0.3 V to +33 V
+0.3 V to −28 V
−0.3 V to +60 V
−0.3 V to +7 V
−0.3 V to +7 V
−0.3 V to DVDD + 0.3 V or +7 V
(whichever is less)
−0.3 V to DVDD + 0.3 V or +7 V
(whichever is less)
−0.3 V to AVDD + 0.3 V or +7 V
(whichever is less)
AVSS to VBOOST_x or 33 V if using
the dc-to-dc converter
AVSS to VBOOST_x or 33 V if using
the dc-to-dc converter
AVSS to VBOOST_x or 33 V if using
the dc-to-dc converter
−0.3 V to +33 V
−0.3 V to +0.3 V
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
THERMAL RESISTANCE
Junction-to-air thermal resistance (θJA) is specified for a JEDEC
4-layer test board.
Table 5. Thermal Resistance
Package Type
64-Lead LFCSP (CP-64-3)
ESD CAUTION
−40°C to +105°C
−65°C to +150°C
125°C
(TJ max − TA)/θJA
JEDEC industry standard
J-STD-020
Power dissipated on chip must be derated to keep the junction temperature
below 125°C.
Rev. B | Page 11 of 48
θJA
20
Unit
°C/W
AD5735
Data Sheet
64
63
62
61
60
59
58
57
56
55
54
53
52
51
50
49
RSET_C
RSET_D
REFOUT
REFIN
COMPLV_D
–VSENSE_D
+VSENSE_D
COMPDCDC_D
VBOOST_D
VOUT_D
IOUT_D
AVSS
COMPLV_C
–VSENSE_C
+VSENSE_C
VOUT_C
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
PIN 1
INDICATOR
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
16
AD5735
TOP VIEW
(Not to Scale)
48
47
46
45
44
43
42
41
40
39
38
37
36
35
34
33
COMPDCDC_C
IOUT_C
VBOOST_C
AVCC
SWC
GNDSWC
GNDSWD
SWD
AVSS
SWA
GNDSWA
GNDSWB
SWB
AGND
VBOOST_B
IOUT_B
NOTES
1.THE EXPOSED PADDLE SHOULD BE CONNECTED TO THE POTENTIAL OF THE
AVSS PIN, OR, ALTERNATIVELY, IT CAN BE LEFT ELECTRICALLY UNCONNECTED.
IT IS RECOMMENDED THAT THE PADDLE BE THERMALLY CONNECTED TO A
COPPER PLANE FOR ENHANCED THERMAL PERFORMANCE.
09961-006
POC
RESET
AVDD
COMPLV_A
–VSENSE_A
+VSENSE_A
COMPDCDC_A
VBOOST_A
VOUT_A
IOUT_A
AVSS
COMPLV_B
–VSENSE_B
+VSENSE_B
VOUT_B
COMPDCDC_B
17
18
19
20
21
22
23
24
25
26
27
28
29
30
31
32
RSET_B
RSET_A
REFGND
REFGND
AD0
AD1
SYNC
SCLK
SDIN
SDO
DVDD
DGND
LDAC
CLEAR
ALERT
FAULT
Figure 7. Pin Configuration
Table 6. Pin Function Descriptions
Pin No.
1
Mnemonic
RSET_B
2
RSET_A
3
4
5
6
7
REFGND
REFGND
AD0
AD1
SYNC
8
SCLK
9
10
11
12
13
SDIN
SDO
DVDD
DGND
LDAC
14
CLEAR
Description
An external, precision, low drift, 15 kΩ current setting resistor can be connected to this pin to improve the
IOUT_B temperature drift performance. For more information, see the External Current Setting Resistor section.
An external, precision, low drift, 15 kΩ current setting resistor can be connected to this pin to improve the
IOUT_A temperature drift performance. For more information, see the External Current Setting Resistor section.
Ground Reference Point for Internal Reference.
Ground Reference Point for Internal Reference.
Address Decode for the Device Under Test (DUT) on the Board.
Address Decode for the DUT on the Board.
Frame Synchronization Signal for the Serial Interface. Active low input. When SYNC is low, data is clocked
into the input shift register on the falling edge of SCLK.
Serial Clock Input. Data is clocked into the input shift register on the falling edge of SCLK. The serial interface
operates at clock speeds of up to 30 MHz.
Serial Data Input. Data must be valid on the falling edge of SCLK.
Serial Data Output. Used to clock data from the serial register in readback mode (see Figure 4 and Figure 5).
Digital Supply Pin. The voltage range is from 2.7 V to 5.5 V.
Digital Ground.
Load DAC. This active low input is used to update the DAC register and, consequently, the DAC outputs.
When LDAC is tied permanently low, the addressed DAC data register is updated on the rising edge of
SYNC. If LDAC is held high during the write cycle, the DAC input register is updated, but the DAC output
is updated only on the falling edge of LDAC (see Figure 3). Using this mode, all analog outputs can be
updated simultaneously. The LDAC pin must not be left unconnected.
Active High, Edge Sensitive Input. When this pin is asserted, the output current and voltage are set to the
programmed clear code bit setting. Only channels enabled to be cleared are cleared. For more information,
see the Asynchronous Clear section. When CLEAR is active, the DAC output register cannot be written to.
Rev. B | Page 12 of 48
Data Sheet
Pin No.
15
Mnemonic
ALERT
16
FAULT
17
POC
18
19
20
RESET
AVDD
COMPLV_A
21
−VSENSE_A
22
+VSENSE_A
23
COMPDCDC_A
24
VBOOST_A
25
26
27
28
VOUT_A
IOUT_A
AVSS
COMPLV_B
29
−VSENSE_B
30
+VSENSE_B
31
32
VOUT_B
COMPDCDC_B
33
34
IOUT_B
VBOOST_B
35
36
AGND
SWB
37
38
39
GNDSWB
GNDSWA
SWA
40
41
AVSS
SWD
42
43
44
GNDSWD
GNDSWC
SWC
AD5735
Description
Active High Output. This pin is asserted when there is no SPI activity on the interface pins for a preset time.
For more information, see the Alert Output section.
Active Low, Open-Drain Output. This pin is asserted low when any of the following conditions is detected:
open circuit in current mode; short circuit in voltage mode; PEC error; or an overtemperature condition (see
the Fault Output section).
Power-On Condition. This pin determines the power-on condition and is read during power-on and after a
device reset. If POC = 0, the device is powered up with the voltage and current channels in tristate mode. If
POC = 1, the device is powered up with a 30 kΩ pull-down resistor to ground on the voltage output channel,
and the current channel is in tristate mode.
Hardware Reset, Active Low Input.
Positive Analog Supply Pin. The voltage range is from 9 V to 33 V.
Optional Compensation Capacitor Connection for VOUT_A Output Buffer. Connecting a 220 pF capacitor
between this pin and the VOUT_A pin allows the voltage output to drive up to 2 µF. Note that the addition
of this capacitor reduces the bandwidth of the output amplifier, increasing the settling time.
Sense Connection for the Negative Voltage Output Load Connection for VOUT_A. This pin must stay within
±3.0 V of AGND for specified operation.
Sense Connection for the Positive Voltage Output Load Connection for VOUT_A. The difference in voltage
between this pin and the VOUT_A pin is added directly to the headroom requirement.
DC-to-DC Compensation Capacitor. Connect a 10 nF capacitor from this pin to ground. Used to regulate the
feedback loop of the Channel A dc-to-dc converter. Alternatively, if using an external compensation resistor,
place a resistor in series with a capacitor to ground from this pin. For more information, see the DC-to-DC
Converter Compensation Capacitors section and the AICC Supply Requirements—Slewing section.
Supply for Channel A Current Output Stage (see Figure 71). This pin is also the supply for the VOUT_A stage,
which is regulated to 15 V by the dc-to-dc converter. To use the dc-to-dc converter, connect this pin as
shown in Figure 77.
Buffered Analog Output Voltage for DAC Channel A.
Current Output Pin for DAC Channel A.
Negative Analog Supply Pin. The voltage range is from −10.8 V to −26.4 V.
Optional Compensation Capacitor Connection for VOUT_B Output Buffer. Connecting a 220 pF capacitor
between this pin and the VOUT_B pin allows the voltage output to drive up to 2 µF. Note that the addition
of this capacitor reduces the bandwidth of the output amplifier, increasing the settling time.
Sense Connection for the Negative Voltage Output Load Connection for VOUT_B. This pin must stay within
±3.0 V of AGND for specified operation.
Sense Connection for the Positive Voltage Output Load Connection for VOUT_B. The difference in voltage
between this pin and the VOUT_B pin is added directly to the headroom requirement.
Buffered Analog Output Voltage for DAC Channel B.
DC-to-DC Compensation Capacitor. Connect a 10 nF capacitor from this pin to ground. Used to regulate the
feedback loop of the Channel B dc-to-dc converter. Alternatively, if using an external compensation resistor,
place a resistor in series with a capacitor to ground from this pin. For more information, see the DC-to-DC
Converter Compensation Capacitors section and the AICC Supply Requirements—Slewing section.
Current Output Pin for DAC Channel B.
Supply for Channel B Current Output Stage (see Figure 71). This pin is also the supply for the VOUT_B stage,
which is regulated to 15 V by the dc-to-dc converter. To use the dc-to-dc converter, connect this pin as
shown in Figure 77.
Ground Reference Point for Analog Circuitry. This pin must be connected to 0 V.
Switching Output for Channel B DC-to-DC Circuitry. To use the dc-to-dc converter, connect this pin as
shown in Figure 77.
Ground Connection for DC-to-DC Switching Circuit. This pin should always be connected to ground.
Ground Connection for DC-to-DC Switching Circuit. This pin should always be connected to ground.
Switching Output for Channel A DC-to-DC Circuitry. To use the dc-to-dc converter, connect this pin as
shown in Figure 77.
Negative Analog Supply Pin. The voltage range is from −10.8 V to −26.4 V.
Switching Output for Channel D DC-to-DC Circuitry. To use the dc-to-dc converter, connect this pin as
shown in Figure 77.
Ground Connection for DC-to-DC Switching Circuit. This pin should always be connected to ground.
Ground Connection for DC-to-DC Switching Circuit. This pin should always be connected to ground.
Switching Output for Channel C DC-to-DC Circuitry. To use the dc-to-dc converter, connect this pin as
shown in Figure 77.
Rev. B | Page 13 of 48
AD5735
Data Sheet
Pin No.
45
46
Mnemonic
AVCC
VBOOST_C
47
48
IOUT_C
COMPDCDC_C
49
50
VOUT_C
+VSENSE_C
51
−VSENSE_C
52
COMPLV_C
53
54
55
56
AVSS
IOUT_D
VOUT_D
VBOOST_D
57
COMPDCDC_D
58
+VSENSE_D
59
−VSENSE_D
60
COMPLV_D
61
62
REFIN
REFOUT
63
RSET_D
64
RSET_C
EPAD
Description
Supply for DC-to-DC Circuitry. The voltage range is from 4.5 V to 5.5 V.
Supply for Channel C Current Output Stage (see Figure 71). This pin is also the supply for the VOUT_C stage,
which is regulated to 15 V by the dc-to-dc converter. To use the dc-to-dc converter, connect this pin as
shown in Figure 77.
Current Output Pin for DAC Channel C.
DC-to-DC Compensation Capacitor. Connect a 10 nF capacitor from this pin to ground. Used to regulate the
feedback loop of the Channel C dc-to-dc converter. Alternatively, if using an external compensation resistor,
place a resistor in series with a capacitor to ground from this pin. For more information, see the DC-to-DC
Converter Compensation Capacitors section and the AICC Supply Requirements—Slewing section.
Buffered Analog Output Voltage for DAC Channel C.
Sense Connection for the Positive Voltage Output Load Connection for VOUT_C. The difference in voltage
between this pin and the VOUT_C pin is added directly to the headroom requirement.
Sense Connection for the Negative Voltage Output Load Connection for VOUT_C. This pin must stay within
±3.0 V of AGND for specified operation.
Optional Compensation Capacitor Connection for VOUT_C Output Buffer. Connecting a 220 pF capacitor
between this pin and the VOUT_C pin allows the voltage output to drive up to 2 µF. Note that the addition
of this capacitor reduces the bandwidth of the output amplifier, increasing the settling time.
Negative Analog Supply Pin. The voltage range is from −10.8 V to −26.4 V.
Current Output Pin for DAC Channel D.
Buffered Analog Output Voltage for DAC Channel D.
Supply for Channel D Current Output Stage (see Figure 71). This pin is also the supply for the VOUT_D stage,
which is regulated to 15 V by the dc-to-dc converter. To use the dc-to-dc converter, connect this pin as
shown in Figure 77.
DC-to-DC Compensation Capacitor. Connect a 10 nF capacitor from this pin to ground. Used to regulate the
feedback loop of the Channel D dc-to-dc converter. Alternatively, if using an external compensation resistor,
place a resistor in series with a capacitor to ground from this pin. For more information, see the DC-to-DC
Converter Compensation Capacitors section and the AICC Supply Requirements—Slewing section.
Sense Connection for the Positive Voltage Output Load Connection for VOUT_D. The difference in voltage
between this pin and the VOUT_D pin is added directly to the headroom requirement.
Sense Connection for the Negative Voltage Output Load Connection for VOUT_D. This pin must stay within
±3.0 V of AGND for specified operation.
Optional Compensation Capacitor Connection for VOUT_D Output Buffer. Connecting a 220 pF capacitor
between this pin and the VOUT_D pin allows the voltage output to drive up to 2 µF. Note that the addition
of this capacitor reduces the bandwidth of the output amplifier, increasing the settling time.
External Reference Voltage Input.
Internal Reference Voltage Output. It is recommended that a 0.1 µF capacitor be placed between REFOUT
and REFGND.
An external, precision, low drift, 15 kΩ current setting resistor can be connected to this pin to improve the
IOUT_D temperature drift performance. For more information, see the External Current Setting Resistor section.
An external, precision, low drift, 15 kΩ current setting resistor can be connected to this pin to improve the
IOUT_C temperature drift performance. For more information, see the External Current Setting Resistor section.
Exposed Pad. The exposed paddle should be connected to the potential of the AVSS pin, or, alternatively,
it can be left electrically unconnected. It is recommended that the paddle be thermally connected to a
copper plane for enhanced thermal performance.
Rev. B | Page 14 of 48
Data Sheet
AD5735
TYPICAL PERFORMANCE CHARACTERISTICS
VOLTAGE OUTPUTS
0.008
0.008
AVDD = +15V
AVSS = –15V
TA = 25°C
0.006
0.006
0.004
INL ERROR (%FSR)
0.002
0
–0.002
0.002
0
–0.002
AVDD = +15V
AVSS = –15V
OUTPUT UNLOADED
–0.004
±10V RANGE
±12V RANGE
–0.004
–0.006
±10V RANGE
WITH DC-TO-DC CONVERTER
0
1000
2000
3000
4000
CODE
–0.008
–40
09961-208
–0.006
–20
0
20
40
60
80
100
TEMPERATURE (°C)
Figure 8. Integral Nonlinearity Error vs. DAC Code
Figure 11. Integral Nonlinearity Error vs. Temperature
1.0
1.0
AVDD = +15V
AVSS = –15V
TA = 25°C
0.8
0.6
0.6
0.4
0.4
0.2
0
–0.2
±10V RANGE
–0.4
AVDD = +15V
AVSS = –15V
ALL RANGES
0.8
DNL ERROR (LSB)
DNL ERROR (LSB)
+5V RANGE MAX INL
±10V RANGE MAX INL
±12V RANGE MAX INL
+5V RANGE MIN INL
±10V RANGE MIN INL
±12V RANGE MIN INL
09961-211
INL ERROR (%FSR)
0.004
MAX DNL
0.2
0
MIN DNL
–0.2
–0.4
±12V RANGE
–0.6
–0.6
±10V RANGE
WITH DC-TO-DC CONVERTER
–0.8
0
1000
2000
3000
4000
CODE
–1.0
–40
09961-209
–1.0
0
20
40
60
80
100
TEMPERATURE (°C)
Figure 9. Differential Nonlinearity Error vs. DAC Code
Figure 12. Differential Nonlinearity Error vs. Temperature
0.02
0.06
0
–0.01
–0.02
±10V RANGE
±12V RANGE
–0.03
±10V RANGE
WITH DC-TO-DC CONVERTER
–0.04
0
1000
2000
3000
CODE
4000
Figure 10. Total Unadjusted Error vs. DAC Code
0.05
+5V RANGE
±10V RANGE
±12V RANGE
0.04
0.03
AV DD = +15V
AV SS = –15V
OUTPUT UNLOADED
0.02
0.01
0
–0.01
–40
–20
0
20
40
60
80
TEMPERATURE (°C)
Figure 13. Total Unadjusted Error vs. Temperature
Rev. B | Page 15 of 48
100
09961-129
0.01
TOTAL UNADJUSTED ERROR (%FSR)
AVDD = +15V
AVSS = –15V
TA = 25°C
09961–210
TOTAL UNADJUSTED ERROR (%FSR)
–20
09961-212
–0.8
AD5735
Data Sheet
0.09
0.06
0.08
0.07
+5V RANGE
±10V RANGE
±12V RANGE
0.04
GAIN ERROR (%FSR)
FULL-SCALE ERROR (%FSR)
0.05
0.03
AV DD = +15V
AV SS = –15V
OUTPUT UNLOADED
0.02
0.01
0.06
+5V RANGE
±10V RANGE
±12V RANGE
AV DD = +15V
AV SS = –15V
OUTPUT UNLOADED
0.05
0.04
0.03
0.02
0.01
0
0
0
20
40
60
80
100
TEMPERATURE (°C)
–0.02
–40
09961-132
–20
–20
0
20
40
60
80
100
80
100
TEMPERATURE (°C)
Figure 14. Full-Scale Error vs. Temperature
09961-135
–0.01
–0.01
–40
Figure 17. Gain Error vs. Temperature
0.015
0.006
0.010
0.005
+5V RANGE
±10V RANGE
±12V RANGE
–0.005
–0.010
–0.015
AV DD = +15V
AV SS = –15V
OUTPUT UNLOADED
–0.020
–0.025
–0.030
+5V RANGE
0.004
0.003
+6V RANGE
0.002
0.001
AV DD = +15V
AV SS = –15V
OUTPUT UNLOADED
–0.035
–20
0
20
40
60
80
100
TEMPERATURE (°C)
0
–40
09961-133
–0.040
–40
0
20
40
60
TEMPERATURE (°C)
Figure 15. Offset Error vs. Temperature
Figure 18. Zero-Scale Error vs. Temperature
0.010
0.006
MAX INL
±10V RANGE
0.005
0.004
0
–0.010
–0.015
INL ERROR (%FSR)
–0.005
AV DD = +15V
AV SS = –15V
OUTPUT UNLOADED
–0.020
±12V RANGE
–0.025
0.002
0V TO 5V RANGE
TA = 25°C
AVSS = –26.4V FOR AVDD > +26.4V
AVSS = –10.8V FOR AVDD < +10.8V
0
–0.002
–0.030
–0.035
–0.004
MIN INL
–0.045
–40
–20
0
20
40
60
80
TEMPERATURE (°C)
100
Figure 16. Bipolar Zero Error vs. Temperature
–0.006
5
10
15
20
SUPPLY (V)
25
Figure 19. Integral Nonlinearity Error vs. Supply
Rev. B | Page 16 of 48
30
09961-219
–0.040
09961-134
BIPOLAR ZERO ERROR (%FSR)
–20
09961-136
0
ZERO-SCALE ERROR (%FSR)
OFFSET ERROR (%FSR)
0.005
Data Sheet
AD5735
1.0
12
AV DD = +15V
AV SS = –15V
±10V RANGE
TA = 25°C
OUTPUT UNLOADED
0.8
ALL RANGES
TA = 25°C
AVSS = –26.4V FOR AVDD > +26.4V
AVSS = –10.8V FOR AVDD < +10.8V
0.4
0.2
OUTPUT VOLTAGE (V)
DNL ERROR (LSB)
0.6
8
MAX DNL
0
MIN DNL
–0.2
–0.4
–0.6
4
0
–4
–8
5
10
15
20
SUPPLY (V)
25
–12
–5
09961-220
–1.0
30
10
15
Figure 23. Full-Scale Positive Step
12
0.020
0V TO 5V RANGE
TA = 25°C
AV SS = –26.4V FOR AV DD > +26.4V
AV SS = –10.8V FOR AV DD < +10.8V
0.015
0.010
AV DD = +15V
AV SS = –15V
±10V RANGE
TA = 25°C
OUTPUT UNLOADED
8
OUTPUT VOLTAGE (V)
0.005
MAX TUE
0
–0.005
MIN TUE
–0.010
4
0
–4
–0.015
–8
5
10
15
20
25
–12
–5
09961-035
–0.025
30
SUPPLY (V)
10
15
Figure 24. Full-Scale Negative Step
0.0020
15
8mA LIMIT, CODE = 0xFFFF
16mA LIMIT, CODE = 0xFFFF
0x7FFF TO 0x8000
0x8000 TO 0x7FFF
AV DD = +15V
AV SS = –15V
+10V RANGE
TA = 25ºC
10
0.0010
5
VOLTAGE (V)
0.0005
0
–0.0005
0
–5
–10
–0.0010
AV DD = +15V
AV SS = –15V
±10V RANGE
TA = 25°C
–16
–12
–8
–4
0
4
8
12
16
–15
20
OUTPUT CURRENT (mA)
09961-036
–0.0015
–0.0020
–20
5
TIME (µs)
Figure 21. Total Unadjusted Error vs. Supply
0.0015
0
09961-038
–0.020
Figure 22. Source and Sink Capability of the Output Amplifier
–20
0
1
2
3
TIME (µs)
Figure 25. Digital-to-Analog Glitch
Rev. B | Page 17 of 48
4
5
09961-039
TOTAL UNADJUSTED ERROR (%FSR)
5
TIME (µs)
Figure 20. Differential Nonlinearity Error vs. Supply
OUTPUT VOLTAGE DELTA (V)
0
09961-037
–0.8
AD5735
Data Sheet
15
60
AV DD = +15V
AV SS = –15V
±10V RANGE
TA = 25°C
OUTPUT UNLOADED
10
40
20
0
0
–5
–20
–40
–60
POC = 1
POC = 0
–80
AV DD = +15V
AV SS = –15V
±10V RANGE
TA = 25°C
INT_ENABLE = 1
–100
–10
–120
0
1
2
3
4
5
6
7
8
9
10
TIME (s)
–140
09961-040
–15
0
AV DD = +15V
AV SS = –15V
8
10
0
±10V RANGE OUTPUT UNLOADED
TA = 25°C
VOUT_X PSRR (dB)
–20
100
0
–100
–200
AV DD = +15V
VBOOST = +15V
AV SS = –15V
TA = 25°C
–40
–60
–80
–300
0
1
2
3
4
5
6
7
8
9
10
TIME (µs)
09961-041
–100
Figure 27. Peak-to-Peak Noise (100 kHz Bandwidth)
20
15
10
5
0
–5
–10
–25
0
25
50
75
100
TIME (µs)
125
09961-043
AV DD = +15V
AV SS = –15V
TA = 25°C
–20
100
1k
10k
100k
FREQUENCY (Hz)
Figure 30. VOUT_x PSRR vs. Frequency
25
–15
–120
10
Figure 28. Voltage vs. Time on Power-Up
Rev. B | Page 18 of 48
1M
10M
09961-045
VOLTAGE (µV)
6
Figure 29. Voltage vs. Time on Output Enable
200
VOLTAGE (mV)
4
TIME (µs)
Figure 26. Peak-to-Peak Noise (0.1 Hz to 10 Hz Bandwidth)
300
2
09961-044
VOLTAGE (mV)
VOLTAGE (µV)
5
Data Sheet
AD5735
CURRENT OUTPUTS
0.008
0.008
4mA TO 20mA, INTERNAL RSET, WITH DC-TO-DC CONVERTER
4mA TO 20mA, EXTERNAL RSET, WITH DC-TO-DC CONVERTER
4mA TO 20mA, INTERNAL RSET
4mA TO 20mA, EXTERNAL RSET
0.006
0.006
INL ERROR (%FSR)
0.002
0
–0.002
0.002
0
–0.002
4mA TO 20mA RANGE MAX INL
0mA TO 24mA RANGE MAX INL
0mA TO 20mA RANGE MAX INL
4mA TO 20mA RANGE MIN INL
0mA TO 24mA RANGE MIN INL
0mA TO 20mA RANGE MIN INL
AVDD = +15V
AVSS = –15V/0V
–0.004
AVDD = +15V
AVSS = –15V
TA = 25°C
–0.006
–0.006
0
1000
2000
3000
4000
CODE
–0.008
–40
09961-231
–0.004
–20
0
20
40
60
80
100
TEMPERATURE (°C)
Figure 31. Integral Nonlinearity Error vs. DAC Code
09961-234
INL ERROR (%FSR)
0.004
0.004
Figure 34. Integral Nonlinearity Error vs. Temperature, Internal RSET
0.008
1.0
4mA TO 20mA, INTERNAL RSET, WITH DC-TO-DC CONVERTER
4mA TO 20mA, EXTERNAL RSET, WITH DC-TO-DC CONVERTER
4mA TO 20mA, INTERNAL RSET
4mA TO 20mA, EXTERNAL RSET
0.8
0.6
0.006
INL ERROR (%FSR)
DNL ERROR (LSB)
0.004
0.4
0.2
0
–0.2
–0.4
0.002
0
–0.002
4mA TO 20mA RANGE MAX INL
0mA TO 24mA RANGE MAX INL
0mA TO 20mA RANGE MAX INL
4mA TO 20mA RANGE MIN INL
0mA TO 24mA RANGE MIN INL
0mA TO 20mA RANGE MIN INL
AVDD = +15V
AVSS = –15V/0V
–20
60
–0.004
AVDD = +15V
AVSS = –15V
TA = 25°C
–1.0
0
1000
2000
3000
4000
CODE
Figure 32. Differential Nonlinearity Error vs. DAC Code
0.4
DNL ERROR (LSB)
0.6
0.02
AVDD = +15V
AVSS = –15V
TA = 25°C
MIN DNL
–0.2
–0.4
–0.6
–0.8
4mA TO 20mA, EXTERNAL RSET
4mA TO 20mA, EXTERNAL RSET, WITH DC-TO-DC CONVERTER
–0.04
1000
2000
3000
CODE
4000
Figure 33. Total Unadjusted Error vs. DAC Code
MAX DNL
0
–0.02
0
100
0.2
–0.01
–0.03
80
0.8
0.03
0
40
1.0
0.04
0.01
20
TEMPERATURE (°C)
4mA TO 20mA, INTERNAL RSET
4mA TO 20mA, INTERNAL RSET, WITH DC-TO-DC CONVERTER
0.05
0
Figure 35. Integral Nonlinearity Error vs. Temperature, External RSET
09961-233
TOTAL UNADJUSTED ERROR (%FSR)
0.06
–0.008
–40
09961-235
–0.006
09961-232
–0.8
–1.0
–40
AVDD = +15V
AVSS = –15V/0V
ALL RANGES
INTERNAL AND EXTERNAL RSET
–20
0
20
40
60
80
100
TEMPERATURE (°C)
Figure 36. Differential Nonlinearity Error vs. Temperature
Rev. B | Page 19 of 48
09961-236
–0.6
AD5735
Data Sheet
0.008
0.025
MAX INL
0.006
0.015
0.005
0
–0.005
AV DD = +15V
AV SS = –15V
–0.010
0.004
INL ERROR (%FSR)
0.010
4mA TO 20mA RANGE
TA = 25°C
AVSS = –26.4V FOR AVDD > +26.4V
AVSS = –10.8V FOR AVDD < +10.8V
0.002
0
–0.002
–0.015
–0.004
–0.025
–40
–20
0
20
40
60
TEMPERATURE (°C)
80
100
–0.006
MIN INL
5
10
15
20
25
30
SUPPLY (V)
09961-240
4mA TO 20mA RANGE, INTERNAL RSET
4mA TO 20mA RANGE, EXTERNAL RSET
–0.020
09961-155
TOTAL UNADJUSTED ERROR (%FSR)
0.020
Figure 40. Integral Nonlinearity Error vs. Supply, External RSET
Figure 37. Total Unadjusted Error vs. Temperature
0.020
0.008
0.015
0.006
0.010
0
–0.005
AV DD = +15V
AV SS = –15V
–0.004
–20
0
20
40
60
TEMPERATURE (°C)
80
100
–0.006
10
15
20
25
30
Figure 41. Integral Nonlinearity Error vs. Supply, Internal RSET
1.0
0.005
ALL RANGES
TA = 25°C
AVSS = –26.4V FOR AVDD > +26.4V
AVSS = –10.8V FOR AVDD < +10.8V
0.8
0
DNL ERROR (LSB)
0.6
–0.005
–0.010
–0.015
4mA TO 20mA RANGE, INTERNAL RSET
4mA TO 20mA RANGE, EXTERNAL RSET
0
60
20
40
TEMPERATURE (°C)
MAX DNL
0.2
0
MIN DNL
–0.2
–0.6
–0.020
–20
0.4
–0.4
AV DD = +15V
AV SS = –15V
80
–0.8
100
09961-159
GAIN ERROR (%FSR)
5
SUPPLY (V)
Figure 38. Full-Scale Error vs. Temperature
–0.025
–40
MIN INL
09961-241
–0.020
–40
0
–0.002
4mA TO 20mA RANGE, INTERNAL RSET
4mA TO 20mA RANGE, EXTERNAL RSET
–0.015
4mA TO 20mA RANGE
TA = 25°C
AVSS = –26.4V FOR AVDD > +26.4V
AVSS = –10.8V FOR AVDD < +10.8V
0.002
Figure 39. Gain Error vs. Temperature
–1.0
5
10
15
20
25
30
SUPPLY (V)
Figure 42. Differential Nonlinearity Error vs. Supply
Rev. B | Page 20 of 48
09961-242
–0.010
0.004
INL ERROR (%FSR)
0.005
09961-157
FULL-SCALE ERROR (%FSR)
MAX INL
Data Sheet
AD5735
4
MAX TUE
0
2
–0.005
0
–0.015
–0.020
–2
–4
MIN TUE
–6
AV DD = +15V
AV SS = –15V
TA = 25°C
RLOAD = 300Ω
INT_ENABLE = 1
–0.025
–8
–0.030
5
10
15
20
25
–10
09961-060
–0.035
30
SUPPLY (V)
0
1
0.05
0.04
4mA TO 20mA RANGE
TA = 25°C
AV SS = –26.4V FOR AV DD > +26.4V
AV SS = –10.8V FOR AV DD < +10.8V
0.01
0
MIN TUE
–0.01
5
10
15
20
25
09961-061
–0.02
30
SUPPLY (V)
OUTPUT CURRENT (mA) AND VBOOST_x VOLTAGE (V)
TOTAL UNADJUSTED ERROR (%FSR)
MAX TUE
0.02
4
5
6
Figure 46. Current vs. Time on Output Enable
0.07
0.03
3
TIME (µs)
Figure 43. Total Unadjusted Error vs. Supply, External RSET
0.06
2
30
25
20
IOUT_x
VBOOST_x
15
10
0mA TO 24mA RANGE
1kΩ LOAD
fSW = 410kHz
INDUCTOR = 10µH (XAL4040-103)
AV CC = 5V
TA = 25°C
5
0
–0.50 –0.25
0
0.25
0.50
0.75
1.00 1.25
09961-167
–0.010
09961-063
4mA TO 20mA RANGE
TA = 25°C
AV SS = –26.4V FOR AV DD > +26.4V
AV SS = –10.8V FOR AV DD < +10.8V
CURRENT (µA)
TOTAL UNADJUSTED ERROR (%FSR)
0.005
1.50 1.75 2.00
TIME (ms)
Figure 44. Total Unadjusted Error vs. Supply, Internal RSET
Figure 47. Output Current and VBOOST_x Settling Time
with DC-to-DC Converter (See Figure 77)
6
30
AV DD = +15V
AV SS = –15V
TA = 25°C
RLOAD = 300Ω
25
3
2
1
20
TA = –40°C
TA = +25°C
TA = +105°C
15
10
0mA TO 24mA RANGE
1kΩ LOAD
fSW = 410kHz
INDUCTOR = 10µH (XAL4040-103)
AV CC = 5V
5
0
0
5
10
15
TIME (µs)
20
09961-062
CURRENT (µA)
4
Figure 45. Current vs. Time on Power-Up
0
–0.25
0
0.25
0.50
0.75
1.00
TIME (ms)
1.25
1.50
1.75
09961-168
OUTPUT CURRENT (mA)
5
Figure 48. Output Current Settling Time with DC-to-DC Converter
over Temperature (See Figure 77)
Rev. B | Page 21 of 48
AD5735
Data Sheet
8
30
0mA TO 24mA RANGE
1kΩ LOAD
fSW = 410kHz
INDUCTOR = 10µH (XAL4040-103)
TA = 25°C
7
HEADROOM VOLTAGE (V)
20
AVCC = 4.5V
AVCC = 5.0V
AVCC = 5.5V
10
0mA TO 24mA RANGE
1kΩ LOAD
fSW = 410kHz
INDUCTOR = 10µH (XAL4040-103)
TA = 25°C
5
0
–0.25
0
0.25
0.50
0.75
1.00
1.25
1.50
1.75
TIME (ms)
5
4
3
2
1
Figure 49. Output Current Settling Time with DC-to-DC Converter
over AVCC (See Figure 77)
10
6
0
0
10
15
20
OUTPUT CURRENT (mA)
Figure 51. DC-to-DC Converter Headroom vs. Output Current (See Figure 77)
0
20mA OUTPUT
10mA OUTPUT
8
–20
6
IOUT_x PSRR (dB)
4
2
0
–2
–4
AV DD = +15V
VBOOST_x = +15V
AV SS = –15V
TA = 25°C
–40
–60
–80
–6
–10
0
2
4
6
8
10
–100
0mA TO 24mA RANGE
1kΩ LOAD
EXTERNA L RSET
TA = 25°C
12
TIME (µs)
14
–120
10
100
1k
10k
100k
FREQUENCY (Hz)
Figure 50. Output Current, AC-Coupled vs. Time
with DC-to-DC Converter (See Figure 77)
Figure 52. IOUT_x PSRR vs. Frequency
Rev. B | Page 22 of 48
1M
10M
09961-068
AV CC = 5V
fSW = 410kHz
INDUCTOR = 10µH (XAL4040-103)
–8
09961-170
CURRENT (AC-COUPLED) (µA)
5
09961-067
15
09961-169
OUTPUT CURRENT (mA)
25
Data Sheet
AD5735
DC-TO-DC CONVERTER
90
80
AV CC = 4.5V
AV CC = 5V
AV CC = 5.5V
85
20mA OUTPUT
70
75
70
65
0mA TO 24mA RANGE
1kΩ LOAD
EXTERNAL RSET
fSW = 410kHz
INDUCTOR = 10µH (XAL4040-103)
TA = 25°C
55
50
0
4
8
12
16
20
50
40
30
24
OUTPUT CURRENT (mA)
20
–40
09961-016
60
60
0mA TO 24mA RANGE
1kΩ LOAD
EXTERNAL RSET
AV CC = 5V
fSW = 410kHz
INDUCTOR = 10µH (XAL4040-103)
–20
0
20
40
60
80
100
TEMPERATURE (°C)
Figure 53. Efficiency at VBOOST_x vs. Output Current (See Figure 77)
09961-019
IOUT_x EFFICIENCY (%)
VBOOST_x EFFICIENCY (%)
80
Figure 56. Output Efficiency vs. Temperature (See Figure 77)
0.6
90
20mA OUTPUT
85
SWITCH RESISTANCE (Ω)
75
70
0mA TO 24mA RANGE
1kΩ LOAD
EXTERNAL RSET
AV CC = 5V
fSW = 410kHz
INDUCTOR = 10µH (XAL4040-103)
TA = 25°C
60
55
50
–40
–20
0
20
40
60
80
100
Figure 54. Efficiency at VBOOST_x vs. Temperature (See Figure 77)
AV CC = 4.5V
AV CC = 5V
AV CC = 5.5V
50
0mA TO 24mA RANGE
1kΩ LOAD
EXTERNAL RSET
fSW = 410kHz
INDUCTOR = 10µH (XAL4040-103)
TA = 25°C
20
0
4
8
12
16
20
24
OUTPUT CURRENT (mA)
09961-018
IOUT_x EFFICIENCY (%)
60
30
0.2
0
–40
–20
0
20
40
60
80
TEMPERATURE (°C)
Figure 57. Switch Resistance vs. Temperature
80
40
0.3
0.1
TEMPERATURE (°C)
70
0.4
Figure 55. Output Efficiency vs. Output Current (See Figure 77)
Rev. B | Page 23 of 48
100
09961-123
65
09961-017
VBOOST_x EFFICIENCY (%)
0.5
80
AD5735
Data Sheet
REFERENCE
5.0050
16
AV DD
REFOUT
TA = 25°C
12
10
8
6
4
2
0
5.0040
5.0035
5.0030
5.0025
5.0020
5.0015
5.0010
5.0005
0
0.2
0.4
0.6
0.8
1.0
1.2
TIME (ms)
5.0000
–40
09961-010
–2
Figure 58. REFOUT Voltage Turn-On Transient
–20
0
20
40
60
80
100
TEMPERATURE (°C)
09961-163
VOLTAGE (V)
30 DEVICES SHOWN
AV DD = 15V
5.0045
REFERENCE OUTPUT VOLTAGE (V)
14
Figure 61. REFOUT Voltage vs. Temperature (When the AD5735 is soldered
onto a PCB, the reference shifts due to thermal shock on the package. The
average output voltage shift is −4 mV. Measurement of these parts after seven
days shows that the outputs typically shift back 2 mV toward their initial values.
This second shift is due to the relaxation of stress incurred during soldering.)
4
5.002
REFERENCE OUTPUT VOLTAGE (V)
1
0
–1
–3
0
2
4
6
8
10
TIME (s)
09961-011
–2
5.000
4.999
4.998
4.997
4.996
4.995
0
4
6
8
10
LOAD CURRENT (mA)
Figure 59. REFOUT Output Noise (0.1 Hz to 10 Hz Bandwidth)
Figure 62. REFOUT Voltage vs. Load Current
150
5.00000
REFERENCE OUTPUT VOLTAGE (V)
AV DD = 15V
TA = 25°C
100
50
0
–50
–100
–150
0
5
10
15
TIME (ms)
20
09961-012
VOLTAGE (µV)
2
Figure 60. REFOUT Output Noise (100 kHz Bandwidth)
4.99995
TA = 25°C
4.99990
4.99985
4.99980
4.99975
4.99970
4.99965
4.99960
10
15
20
25
AV DD (V)
Figure 63. REFOUT Voltage vs. AVDD
Rev. B | Page 24 of 48
30
09961-015
VOLTAGE (µV)
2
AV DD = 15V
TA = 25°C
5.001
09961-014
AV DD = 15V
TA = 25°C
3
Data Sheet
AD5735
GENERAL
450
13.4
DVDD = 5V
TA = 25°C
400
13.3
350
FREQUENCY (MHz)
13.2
250
200
150
13.1
13.0
12.9
12.8
100
12.7
50
0
1
2
3
4
5
SDIN VOLTAGE (V)
12.6
–40
09961-007
0
DVDD = 5.5V
–20
0
20
40
60
80
09961-020
DICC (µA)
300
100
TEMPERATURE (°C)
Figure 64. DICC vs. Logic Input Voltage
Figure 67. Internal Oscillator Frequency vs. Temperature
10
14.4
8
14.2
6
AIDD
AISS
TA = 25°C
VOUT = 0V
OUTPUT UNLOADED
2
0
14.0
FREQUENCY (MHz)
CURRENT (mA)
4
–2
–4
13.8
13.6
13.4
–6
–8
13.2
–10
15
20
25
30
VOLTAGE (V)
Figure 65. Supply Current (AIDD/AISS) vs. Supply Voltage (AVDD/|AVSS|)
7
5
4
3
2
AIDD
TA = 25°C
IOUT = 0mA
15
20
25
30
VOLTAGE (V)
09961-009
CURRENT (mA)
6
0
10
3.0
3.5
4.0
4.5
5.0
5.5
VOLTAGE (V)
Figure 68. Internal Oscillator Frequency vs. DVDD Supply Voltage
8
1
13.0
2.5
Figure 66. Supply Current (AIDD) vs. Supply Voltage (AVDD)
Rev. B | Page 25 of 48
09961-021
TA = 25°C
09961-008
–12
10
AD5735
Data Sheet
TERMINOLOGY
Relative Accuracy or Integral Nonlinearity (INL)
Relative accuracy, or integral nonlinearity (INL), is a measure
of the maximum deviation from the best fit line through the
DAC transfer function. INL is expressed in percent of full-scale
range (% FSR). Typical INL vs. code plots are shown in Figure 8
and Figure 31.
Differential Nonlinearity (DNL)
Differential nonlinearity (DNL) is the difference between the
measured change and the ideal 1 LSB change between any two
adjacent codes. A specified DNL of ±1 LSB maximum ensures
monotonicity. The AD5735 is guaranteed monotonic by design.
Typical DNL vs. code plots are shown in Figure 9 and Figure 32.
Monotonicity
A DAC is monotonic if the output either increases or remains
constant for increasing digital input code. The AD5735 is
monotonic over its full operating temperature range.
Negative Full-Scale Error or Zero-Scale Error
Negative full-scale error is the error in the DAC output voltage
when 0x0000 (straight binary coding) is loaded to the DAC
register.
Zero-Scale Temperature Coefficient (TC)
Zero-scale TC is a measure of the change in zero-scale error
with a change in temperature. Zero-scale TC is expressed in
ppm FSR/°C.
Bipolar Zero Error
Bipolar zero error is the deviation of the analog output from the
ideal half-scale output of 0 V when the DAC register is loaded
with 0x8000 (straight binary coding).
Bipolar Zero Temperature Coefficient (TC)
Bipolar zero TC is a measure of the change in the bipolar zero
error with a change in temperature. It is expressed in ppm
FSR/°C.
Offset Error
In voltage output mode, offset error is the deviation of the
analog output from the ideal quarter-scale output when the
DAC is configured for a bipolar output range and the DAC
register is loaded with 0x4000 (straight binary coding).
Gain Temperature Coefficient (TC)
Gain TC is a measure of the change in gain error with changes
in temperature and is expressed in ppm FSR/°C.
Full-Scale Error
Full-scale error is a measure of the output error when full-scale
code is loaded to the DAC register. Ideally, the output should be
full-scale − 1 LSB. Full-scale error is expressed in % FSR.
Full-Scale Temperature Coefficient (TC)
Full-scale TC is a measure of the change in full-scale error with
changes in temperature and is expressed in ppm FSR/°C.
Total Unadjusted Error (TUE)
Total unadjusted error (TUE) is a measure of the output error
that includes all the error measurements: INL error, offset error,
gain error, temperature, and time. TUE is expressed in % FSR.
DC Crosstalk
DC crosstalk is the dc change in the output level of one DAC in
response to a change in the output of another DAC. It is measured
with a full-scale output change on one DAC while monitoring
another DAC, which is at midscale.
Current Loop Compliance Voltage
The current loop compliance voltage is the maximum voltage
at the IOUT_x pin for which the output current is equal to the
programmed value.
Voltage Reference Thermal Hysteresis
Voltage reference thermal hysteresis is the difference in output
voltage measured at +25°C compared to the output voltage
measured at +25°C after cycling the temperature from +25°C to
−40°C to +105°C and back to +25°C. The hysteresis is specified
for the first and second temperature cycles and is expressed in ppm.
Output Voltage Settling Time
Output voltage settling time is the amount of time it takes
for the output to settle to a specified level for a full-scale input
change. Plots of settling time are shown in Figure 23, Figure 48,
and Figure 49.
In current output mode, offset error is the deviation of the
analog output from the ideal zero-scale output when all DAC
registers are loaded with 0x0000.
Slew Rate
The slew rate of a device is a limitation in the rate of change of
the output voltage. The output slewing speed of a voltage output
DAC is usually limited by the slew rate of the amplifier used at
its output. Slew rate is measured from 10% to 90% of the output
signal and is given in V/µs.
Offset Error Drift or Offset TC
Offset error drift, or offset TC, is a measure of the change in
offset error with changes in temperature and is expressed in
ppm FSR/°C.
Power-On Glitch Energy
Power-on glitch energy is the impulse injected into the analog
output when the AD5735 is powered on. It is specified as the
area of the glitch in nV-sec (see Figure 28 and Figure 45).
Gain Error
Gain error is a measure of the span error of the DAC. It is the
deviation in slope of the DAC transfer function from the ideal,
expressed in % FSR.
Rev. B | Page 26 of 48
Data Sheet
AD5735
Digital-to-Analog Glitch Energy
Digital-to-analog glitch energy is the impulse injected into
the analog output when the input code in the DAC register
changes state but the output voltage remains constant. It is
normally specified as the area of the glitch in nV-sec and is
measured when the digital input code is changed by 1 LSB at
the major carry transition (~0x7FFF to 0x8000). See Figure 25.
Glitch Impulse Peak Amplitude
Glitch impulse peak amplitude is the peak amplitude of the
impulse injected into the analog output when the input code in
the DAC register changes state. It is specified as the amplitude
of the glitch in mV and is measured when the digital input code
is changed by 1 LSB at the major carry transition (~0x7FFF to
0x8000). See Figure 25.
Digital Feedthrough
Digital feedthrough is a measure of the impulse injected into
the analog output of the DAC from the digital inputs of the
DAC but is measured when the DAC output is not updated. It is
specified in nV-sec and measured with a full-scale code change
on the data bus.
DAC-to-DAC Crosstalk
DAC-to-DAC crosstalk is the glitch impulse transferred to the
output of one DAC due to a digital code change and a subsequent
output change of another DAC. DAC-to-DAC crosstalk includes
both digital and analog crosstalk. It is measured by loading one
DAC with a full-scale code change (all 0s to all 1s and vice versa)
with LDAC low while monitoring the output of another DAC.
The energy of the glitch is expressed in nV-sec.
Reference Temperature Coefficient (TC)
Reference TC is a measure of the change in the reference output
voltage with changes in temperature. It is expressed in ppm/°C.
Line Regulation
Line regulation is the change in the reference output voltage due
to a specified change in supply voltage. It is expressed in ppm/V.
Load Regulation
Load regulation is the change in the reference output voltage due
to a specified change in load current. It is expressed in ppm/mA.
DC-to-DC Converter Headroom
DC-to-DC converter headroom is the difference between the
voltage required at the current output and the voltage supplied
by the dc-to-dc converter (see Figure 51).
Output Efficiency
Output efficiency is defined as the ratio of the power delivered
to a channel’s load and the power delivered to the channel’s
dc-to-dc input. The VBOOST_x quiescent current is considered
part of the dc-to-dc converter’s losses.
I OUT 2 × R LOAD
AVCC × AI CC
Efficiency at VBOOST_x
The efficiency at VBOOST_x is defined as the ratio of the power
delivered to a channel’s VBOOST_x supply and the power delivered
to the channel’s dc-to-dc input. The VBOOST_x quiescent current is
considered part of the dc-to-dc converter’s losses.
Power Supply Rejection Ratio (PSRR)
PSRR indicates how the output of the DAC is affected by changes
in the power supply voltage.
Rev. B | Page 27 of 48
I OUT × V BOOST _ x
AVCC × AI CC
AD5735
Data Sheet
THEORY OF OPERATION
The AD5735 is a quad, precision digital-to-current loop and
voltage output converter designed to meet the requirements of
industrial process control applications. It provides a high precision,
fully integrated, low cost, single-chip solution for generating
current loop and unipolar/bipolar voltage outputs.
VBOOST_x
R2
T2
A2
DAC ARCHITECTURE
The DAC core architecture of the AD5735 consists of two
matched DAC sections. A simplified circuit diagram is shown
in Figure 69. The four MSBs of the 12-bit data-word are decoded
to drive 15 switches, E1 to E15. Each switch connects one of
15 matched resistors either to ground or to the reference buffer
output. The remaining eight bits of the data-word drive Switch S0
to Switch S7 of an 8-bit voltage mode R-2R ladder network.
VOUT
RSET
Figure 71. Voltage-to-Current Conversion Circuitry
Voltage Output Amplifier
The voltage output amplifier is capable of generating both
unipolar and bipolar output voltages. It is capable of driving a
load of 1 kΩ in parallel with 1 µF (with an external compensation capacitor) to AGND. The source and sink capabilities
of the output amplifier are shown in Figure 22. The slew rate is
1.9 V/µs with a full-scale settling time of 18 µs max (10 V step).
If remote sensing of the load is not required, connect +VSENSE_x
directly to VOUT_x, and connect −VSENSE_x directly to AGND.
−VSENSE_x must stay within ±3.0 V of AGND for specified operation. The difference in voltage between +VSENSE_x and VOUT_x
should be added directly to the headroom requirement.
2R
2R
2R
2R
Driving Large Capacitive Loads
S0
S1
S7
E1
E2
E15
The voltage output amplifier is capable of driving capacitive
loads of up to 2 µF with the addition of a 220 pF, nonpolarized
compensation capacitor on each channel. The 220 pF capacitor
is connected between the COMPLV_x pin and the VOUT_x pin.
FOUR MSBs DECODED INTO
15 EQUAL SEGMENTS
Figure 69. DAC Ladder Structure
The voltage output from the DAC core can be
•
Buffered and scaled to output a software selectable
unipolar or bipolar voltage range (see Figure 70)
Converted to a current, which is then mirrored to the
supply rail so that the application sees only a current
source output (see Figure 71)
Care should be taken to choose an appropriate value of compensation capacitor. This capacitor, while allowing the AD5735
to drive higher capacitive loads and reduce overshoot, increases
the settling time of the part and, therefore, affects the bandwidth
of the system. Without the compensation capacitor, capacitive
loads of up to 10 nF can be driven.
Reference Buffers
Both the voltage and current outputs are supplied by VBOOST_x.
The current and voltage are output on separate pins and cannot
be output simultaneously. The current and voltage output pins
of a channel can be tied together (see the Voltage and Current
Output Pins on the Same Terminal section).
The AD5735 can operate with either an external or internal
reference. The reference input requires a 5 V reference for
specified performance. This input voltage is then buffered
before it is applied to the DAC.
+VSENSE_X
RANGE
SCALING
VOUT_X
VOUT_X SHORT FAULT
–VSENSE_X
09961-070
12-BIT
DAC
IOUT_x
2R
8-BIT R-2R LADDER
•
T1
A1
2R
09961-069
2R
12-BIT
DAC
09961-071
The current ranges available are 0 mA to 20 mA, 4 mA to 20 mA,
and 0 mA to 24 mA. The voltage ranges available are 0 V to 5 V,
±5 V, 0 V to 10 V, and ±10 V. The current and voltage outputs
are available on separate pins, and only one output is active at
any one time. The output configuration is user-selectable via the
DAC control register.
On-chip dynamic power control minimizes package power
dissipation in current mode (see the Dynamic Power Control
section).
R3
Figure 70. Voltage Output
Rev. B | Page 28 of 48
Data Sheet
AD5735
POWER-ON STATE OF THE AD5735
On initial power-up of the AD5735, the state of the power-on
reset circuit is dependent on the power-on condition (POC) pin.
•
•
If POC = 0, both the voltage output and current output
channels power up in tristate mode.
If POC = 1, the voltage output channel powers up with
a 30 kΩ pull-down resistor to ground, and the current
output channel powers up in tristate mode.
The output ranges are not enabled, but the default output range
is 0 V to 5 V, and the clear code register is loaded with all 0s.
Therefore, if the user clears the part after power-up, the output
is actively driven to 0 V if the channel has been enabled for clear.
Simultaneous Updating of All DACs
To update all DACs simultaneously, LDAC is held high while
data is clocked into the DAC data register. After LDAC is taken
high, only the first write to the DAC data register of each channel
is valid; subsequent writes to the DAC data register are ignored,
although these subsequent writes are returned if a readback is
initiated. All DAC outputs are updated by taking LDAC low
after SYNC is taken high.
OUTPUT
AMPLIFIERS
VREFIN
After device power on, or a device reset, it is recommended to
wait 100 μs or more before writing to the device to allow time
for internal calibrations to take place.
LDAC
SERIAL INTERFACE
12-BIT
DAC
VOUT_x
DAC
REGISTER
DAC INPUT
REGISTER
The AD5735 is controlled by a versatile 3-wire serial interface
that operates at clock rates of up to 30 MHz and is compatible
with SPI, QSPI, MICROWIRE, and DSP standards. Data coding
is always straight binary.
OFFSET
AND GAIN
CALIBRATION
DAC DATA
REGISTER
SCLK
SYNC
SDIN
The input shift register is 24 bits wide. Data is loaded into the
device MSB first as a 24-bit word under the control of the serial
clock input, SCLK. Data is clocked in on the falling edge of SCLK.
If packet error checking (PEC) is enabled, an additional eight
bits must be written to the AD5735, creating a 32-bit serial
interface (see the Packet Error Checking section).
The DAC outputs can be updated in one of two ways: individual
DAC updating or simultaneous updating of all DACs.
INTERFACE
LOGIC
SDO
09961-072
Input Shift Register
Figure 72. Simplified Serial Interface of the Input Loading Circuitry
for One DAC Channel
TRANSFER FUNCTION
Table 7 shows the input code to ideal output voltage relationship
for the AD5735 for straight binary data coding of the ±10 V
output range.
Table 7. Input Code to Ideal Output Voltage Relationship
Individual DAC Updating
To update an individual DAC, LDAC is held low while data is
clocked into the DAC data register. The addressed DAC output
is updated on the rising edge of SYNC. See Table 3 and Figure 3
for timing information.
Digital Input
Straight Binary Data Coding
MSB
LSB1
1111
1111
1111
XXXX
1111
1111
1110
XXXX
1000
0000
0000
XXXX
0000
0000
0001
XXXX
0000
0000
0000
XXXX
1
X = don’t care.
Rev. B | Page 29 of 48
Analog Output
VOUT
+2 VREF × (2047/2048)
+2 VREF × (2046/2048)
0V
−2 VREF × (2047/2048)
−2 VREF
AD5735
Data Sheet
REGISTERS
Table 8, Table 9, and Table 10 provide an overview of the registers for the AD5735.
Table 8. Data Registers for the AD5735
Register
DAC Data Registers
Gain Registers
Offset Registers
Clear Code Registers
Description
The four DAC data registers (one register per DAC channel) are used to write a DAC code to each DAC
channel. The DAC data bits are D15 to D4.
The four gain registers (one register per DAC channel) are used to program the gain trim on a per-channel
basis. The gain data bits are D15 to D4.
The four offset registers (one register per DAC channel) are used to program the offset trim on a per-channel
basis. The offset data bits are D15 to D4.
The four clear code registers (one register per DAC channel) are used to program the clear code on a perchannel basis. The clear code data bits are D15 to D4.
Table 9. Control Registers for the AD5735
Register
Main Control Register
DAC Control Registers
Software Register
DC-to-DC Control Register
Slew Rate Control Registers
Description
The main control register is used to configure functions for the entire part. These functions include the
following: enabling status readback during a write; enabling the output on all four DAC channels simultaneously; power-on of the dc-to-dc converter on all four DAC channels simultaneously; and enabling and
configuring the watchdog timer. For more information, see the Main Control Register section.
The four DAC control registers (one register per DAC channel) are used to configure the following functions
on a per-channel basis: output range (for example, 4 mA to 20 mA or 0 V to 10 V); selection of the internal
current sense resistor or an external current sense resistor; enabling/disabling the use of a clear code;
enabling/disabling overrange on a voltage channel; enabling/disabling the internal circuitry (dc-to-dc
converter, DAC, and internal amplifiers); power-on/power-off of the dc-to-dc converter; and enabling/
disabling the output channel.
The software register is used to perform a reset, to toggle the user bit in the status register, and, as part of
the watchdog timer feature, to verify correct data communication operation.
The dc-to-dc control register is used to set the control parameters for the dc-to-dc converter: maximum
output voltage, phase, and switching frequency. This register is also used to select the internal compensation resistor or an external compensation resistor for the dc-to-dc converter.
The four slew rate control registers (one register per DAC channel) are used to program the slew rate of
the DAC output.
Table 10. Readback Register for the AD5735
Register
Status Register
Description
The status register contains any fault information, as well as a user toggle bit.
Rev. B | Page 30 of 48
Data Sheet
AD5735
ENABLING THE OUTPUT
Reprogramming the Output Range
To correctly write to and set up the part from a power-on
condition, use the following sequence:
When changing the range of an output, the same sequence
described in the Enabling the Output section should be used.
It is recommended that the range be set to 0 V (zero scale or
midscale) before the output is disabled. Because the dc-to-dc
switching frequency, maximum output voltage, and phase have
already been selected, there is no need to reprogram these values.
Figure 74 provides a flowchart of this sequence.
3.
4.
5.
Perform a hardware or software reset after initial power-on.
Configure the dc-to-dc converter supply block. Set the
dc-to-dc switching frequency, the maximum output voltage
allowed, and the dc-to-dc converter phase between channels.
Configure the DAC control register on a per-channel basis.
Select the output range, and enable the dc-to-dc converter
block (DC_DC bit). Other control bits can also be configured. Set the INT_ENABLE bit, but do not set the OUTEN
(output enable) bit.
Write the required code to the DAC data register. This step
implements a full internal DAC calibration. For reduced
output glitch, allow at least 200 µs before performing Step 5.
Write to the DAC control register again to enable the
output (set the OUTEN bit).
Figure 73 provides a flowchart of this sequence.
CHANNEL OUTPUT IS ENABLED.
STEP 1: WRITE TO CHANNEL’S DAC DATA
REGISTER. SET THE OUTPUT
TO 0V (ZERO OR MIDSCALE).
STEP 2: WRITE TO DAC CONTROL REGISTER.
DISABLE THE OUTPUT (OUTEN = 0) AND
SET THE NEW OUTPUT RANGE. KEEP THE
DC_DC BIT AND THE INT_ENABLE BIT SET.
STEP 3: WRITE VALUE TO THE DAC DATA REGISTER.
POWER ON.
STEP 4: WRITE TO DAC CONTROL REGISTER.
RELOAD SEQUENCE AS IN STEP 2.
SET THE OUTEN BIT TO ENABLE THE
OUTPUT.
STEP 1: PERFORM A SOFTWARE/HARDWARE RESET.
09961-074
1.
2.
Figure 74. Programming Sequence to Change the Output Range
STEP 2: WRITE TO DC-TO-DC CONTROL REGISTER TO
SET DC-TO-DC CLOCK FREQUENCY, PHASE,
AND MAXIMUM VOLTAGE.
STEP 3: WRITE TO DAC CONTROL REGISTER. SELECT
THE DAC CHANNEL AND OUTPUT RANGE.
SET THE DC_DC BIT AND OTHER CONTROL
BITS AS REQUIRED. SET THE INT_ENABLE BIT
BUT DO NOT SET THE OUTEN BIT.
STEP 5: WRITE TO DAC CONTROL REGISTER. RELOAD
SEQUENCE AS IN STEP 3. SET THE OUTEN
BIT TO ENABLE THE OUTPUT.
09961-073
STEP 4: WRITE TO ONE OR MORE DAC DATA REGISTERS.
ALLOW AT LEAST 200µs BETWEEN STEP 3
AND STEP 5 FOR REDUCED OUTPUT GLITCH.
Figure 73. Programming Sequence to Correctly Enable the Output
Rev. B | Page 31 of 48
AD5735
Data Sheet
DATA REGISTERS
DAC Data Register
The input shift register is 24 bits wide. When PEC is enabled,
the input shift register is 32 bits wide, with the last eight bits
corresponding to the PEC code (see the Packet Error Checking
section for more information about PEC). When writing to a
data register, the format shown in Table 11 must be used.
When writing to a DAC data register, Bit D15 to Bit D4 are the
DAC data bits. Table 13 shows the register format, and Table 12
describes the functions of Bit D23 to Bit D16.
Table 11. Input Shift Register for a Write Operation to a Data Register
MSB
D23
R/W
D22
DUT_AD1
D21
DUT_AD0
D20
DREG2
D19
DREG1
D18
DREG0
D17
DAC_AD1
D16
DAC_AD0
LSB
D15 to D0
Data
Table 12. Descriptions of Data Register Bits[D23:D16]
Bit Name
R/W
DUT_AD1, DUT_AD0
DREG2, DREG1, DREG0
DAC_AD1, DAC_AD0
Description
This bit indicates whether the addressed register is written to or read from.
0 = write to the addressed register.
1 = read from the addressed register.
Used in association with the external pins AD1 and AD0, these bits determine which AD5735 device is being
addressed by the system controller.
DUT_AD1
DUT_AD0
Part Addressed
0
0
Pin AD1 = 0, Pin AD0 = 0
0
1
Pin AD1 = 0, Pin AD0 = 1
1
0
Pin AD1 = 1, Pin AD0 = 0
1
1
Pin AD1 = 1, Pin AD0 = 1
These bits select the register to be written to. If a control register is selected (DREG[2:0] = 111), the CREG bits in
the control register select the specific control register to be written to (see Table 20).
DREG2
DREG1
DREG0
Function
0
0
0
Write to DAC data register (one DAC channel)
0
0
1
Reserved
0
1
0
Write to gain register (one DAC channel)
0
1
1
Write to gain registers (all DAC channels)
1
0
0
Write to offset register (one DAC channel)
1
0
1
Write to offset registers (all DAC channels)
1
1
0
Write to clear code register (one DAC channel)
1
1
1
Write to a control register
These bits are used to specify the DAC channel. If a write to the part does not apply to a specific DAC channel,
these bits are don’t care bits.
DAC_AD1
DAC_AD0
DAC Channel
0
0
DAC A
0
1
DAC B
1
0
DAC C
1
1
DAC D
Table 13. Programming the DAC Data Register
D23
R/W
1
D22
DUT_AD1
D21
DUT_AD0
D20
0
D19
0
D18
0
X = don’t care.
Rev. B | Page 32 of 48
D17
DAC_AD1
D16
DAC_AD0
D15 to D4
DAC data
D3 to D0
X1
Data Sheet
AD5735
Gain Register
DREG[2:0] bits to 100 (see Table 16). To write the same offset
code to all four DAC channels at the same time, set the DREG[2:0]
bits to 101. The offset register coding is straight binary, as shown in
Table 17. The default code in the offset register is 0x8000, which
results in zero offset programmed to the output (for more information, see the Digital Offset and Gain Control section).
The 12-bit gain register allows the user to adjust the gain of
each channel in steps of 1 LSB. To write to the gain register of
one DAC channel, set the DREG[2:0] bits to 010 (see Table 14).
To write the same gain code to all four DAC channels at the
same time, set the DREG[2:0] bits to 011. The gain register
coding is straight binary, as shown in Table 15. The default code
in the gain register is 0xFFFF. The maximum recommended
gain trim is approximately 50% of the programmed range to
maintain accuracy (for more information, see the Digital Offset
and Gain Control section).
Clear Code Register
The 12-bit clear code register allows the user to set the clear
value of each channel. To configure a channel to be cleared
when the CLEAR pin is activated, set the CLR_EN bit in the
DAC control register for that channel (see Table 24). To write
to the clear code register, set the DREG[2:0] bits to 110 (see
Table 18). The default clear code is 0x0000 (for more information, see the Asynchronous Clear section).
Offset Register
The 12-bit offset register allows the user to adjust the offset
of each channel by −2048 LSB to +2047 LSB in steps of 1 LSB.
To write to the offset register of one DAC channel, set the
Table 14. Programming the Gain Register
R/W
0
DUT_AD1
DUT_AD0
Device address
DREG2
0
DREG1
1
DREG0
0
DAC_AD1
DAC_AD0
DAC channel address
D15 to D4
Gain adjustment
D3 to D0
1111
Table 15. Gain Register Bit Descriptions
Gain Adjustment
+4096 LSB
+4095 LSB
…
1 LSB
0 LSB
G15
1
1
…
0
0
G14
1
1
…
0
0
G13 to G5
111111111
111111111
…
000000000
000000000
G4
1
0
…
1
0
G3 to G0
1111
1111
1111
1111
1111
Table 16. Programming the Offset Register
R/W
0
DUT_AD1
DUT_AD0
Device address
DREG2
1
DREG1
0
DREG0
0
DAC_AD1
DAC_AD0
DAC channel address
D15 to D4
Offset adjustment
D3 to D0
0000
Table 17. Offset Register Bit Descriptions
Offset Adjustment
+2047 LSB
+2046 LSB
…
No Adjustment (Default)
…
−2047 LSB
−2048 LSB
OF15
1
1
…
1
…
0
0
OF14
1
1
…
0
…
0
0
OF13
1
1
…
0
…
0
0
OF12 to OF5
11111111
11111111
…
00000000
…
00000000
00000000
OF4
1
0
…
0
…
1
0
OF3 to OF0
0000
0000
0000
0000
0000
0000
0000
Table 18. Programming the Clear Code Register
R/W
0
DUT_AD1
DUT_AD0
Device address
DREG2
1
DREG1
1
DREG0
0
Rev. B | Page 33 of 48
DAC_AD1
DAC_AD0
DAC channel address
D15 to D4
Clear code
D3 to D0
0000
AD5735
Data Sheet
CONTROL REGISTERS
Main Control Register
When writing to a control register, the format shown in Table 19
must be used. See Table 12 for information about the configuration of Bit D23 to Bit D16. The control registers are addressed
by setting the DREG[2:0] bits (Bits[D20:D18] in the input shift
register) to 111 and then setting the CREG[2:0] bits to select the
specific control register (see Table 20).
The main control register options are shown in Table 21 and
Table 22. See the Device Features section for more information
about the features controlled by the main control register.
Table 19. Input Shift Register for a Write Operation to a Control Register
MSB
D23
R/W
D22
DUT_AD1
D21
DUT_AD0
D20
1
D19
1
D18
1
D17
DAC_AD1
D16
DAC_AD0
D15
CREG2
D14
CREG1
D13
CREG0
LSB
D12 to D0
Data
Table 20. Control Register Addresses (CREG[2:0] Bits)
CREG2 (D15)
0
0
0
0
1
CREG1 (D14)
0
0
1
1
0
CREG0 (D13)
0
1
0
1
0
Control Register
Slew rate control register (one per channel)
Main control register
DAC control register (one per channel)
DC-to-DC control register
Software register
Table 21. Programming the Main Control Register
D15
0
1
D14
0
D13
1
D12
POC
D11
STATREAD
D10
EWD
D9
WD1
D8
WD0
D7
X1
D6
ShtCctLim
D5
D4
OUTEN_ALL DCDC_ALL
D3 to D0
X1
X = don’t care.
Table 22. Main Control Register Bit Descriptions
Bit Name
POC
STATREAD
EWD
WD1, WD0
ShtCctLim
OUTEN_ALL
DCDC_ALL
Description
The POC bit determines the state of the voltage output channels during normal operation.
POC = 0: the output goes to the value set by the POC hardware pin when the voltage output is not enabled (default).
POC = 1: the output goes to the opposite value of the POC hardware pin when the voltage output is not enabled.
Enable status readback during a write. See the Status Readback During a Write section.
0 = disable status readback (default).
1 = enable status readback.
Enable the watchdog timer. See the Watchdog Timer section.
0 = disable the watchdog timer (default).
1 = enable the watchdog timer.
Timeout select bits. Used to select the timeout period for the watchdog timer.
WD1
WD0
Timeout Period (ms)
0
0
5
0
1
10
1
0
100
1
1
200
Programmable short-circuit limit on the VOUT_x pin in the event of a short-circuit condition.
0 = 16 mA (default).
1 = 8 mA.
Setting this bit to 1 enables the output on all four DACs simultaneously. Do not use the OUTEN_ALL bit when using the
OUTEN bit in the DAC control register.
Setting this bit to 1 powers up the dc-to-dc converter on all four channels simultaneously. To power down the dc-to-dc
converters, all channel outputs must first be disabled. Do not use the DCDC_ALL bit when using the DC_DC bit in the
DAC control register.
Rev. B | Page 34 of 48
Data Sheet
AD5735
DAC Control Register
The DAC control register is used to configure each DAC channel. The DAC control register options are shown in Table 23 and Table 24.
Table 23. Programming the DAC Control Register
D15
0
1
D14
1
D13
0
D12
X1
D11
X1
D10
X1
D9
X1
D8
D7
D6
INT_ENABLE CLR_EN OUTEN
D5
RSET
D4
DC_DC
D3
OVRNG
D2
R2
D1
R1
D0
R0
X = don’t care.
Table 24. DAC Control Register Bit Descriptions
Bit Name
INT_ENABLE
CLR_EN
OUTEN
RSET
DC_DC
OVRNG
R2, R1, R0
Description
Powers up the dc-to-dc converter, DAC, and internal amplifiers for the selected channel. This bit applies to individual
channels only; it does not enable the output. After setting this bit, it is recommended that a >200 µs delay be observed
before enabling the output to reduce the output enable glitch. See Figure 29 and Figure 46 for plots of this glitch.
Per-channel clear enable bit. This bit specifies whether the selected channel is cleared when the CLEAR pin is activated.
0 = channel is not cleared when the part is cleared (default).
1 = channel is cleared when the part is cleared.
Enables or disables the selected output channel.
0 = channel disabled (default).
1 = channel enabled.
Selects the internal current sense resistor or an external current sense resistor for the selected DAC channel.
0 = external resistor selected (default).
1 = internal resistor selected.
Powers up or powers down the dc-to-dc converter on the selected channel. All dc-to-dc converters can be powered up
simultaneously using the DCDC_ALL bit in the main control register. To power down the dc-to-dc converter, the OUTEN
and INT_ENABLE bits must also be set to 0.
0 = dc-to-dc converter is powered down (default).
1 = dc-to-dc converter is powered up.
Enables 20% overrange on the voltage output channel only. No current output overrange is available.
0 = overrange disabled (default).
1 = overrange enabled.
Selects the output range to be enabled.
R2
R1
R0
Output Range Selected
0
0
0
0 V to 5 V voltage range (default)
0
0
1
0 V to 10 V voltage range
0
1
0
±5 V voltage range
0
1
1
±10 V voltage range
1
0
0
4 mA to 20 mA current range
1
0
1
0 mA to 20 mA current range
1
1
0
0 mA to 24 mA current range
Rev. B | Page 35 of 48
AD5735
Data Sheet
Software Register
The software register allows the user to perform a software reset of
the part. This register is also used to set the user toggle bit, D11,
in the status register and as part of the watchdog timer feature
when that feature is enabled.
Bit D12 in the software register can be used to ensure that
communication has not been lost between the MCU and the
AD5735 and that the datapath lines are working properly (that
is, SDIN, SCLK, and SYNC).
When the watchdog timer feature is enabled, the user must write
0x195 to Bits[D11:D0] of the software register within the timeout
period. If this command is not received within the timeout period,
the ALERT pin signals a fault condition. This command is only
required when the watchdog timer feature is enabled.
DC-to-DC Control Register
The dc-to-dc control register allows the user to configure the
dc-to-dc switching frequency and phase, as well as the maximum allowable dc-to-dc output voltage. The dc-to-dc control
register options are shown in Table 27 and Table 28.
Table 25. Programming the Software Register
D15
1
D14
0
D13
0
D12
User program
D11 to D0
Reset code/SPI code
Table 26. Software Register Bit Descriptions
Bit Name
User Program
Reset Code/SPI Code
Description
This bit is mapped to Bit D11 of the status register. When this bit is set to 1, Bit D11 of the status register is set to 1.
When this bit is set to 0, Bit D11 of the status register is also set to 0. This feature can be used to ensure that the SPI
pins are working correctly by writing a known bit value to this register and then reading back Bit D11 from the
status register.
Option
Description
Reset code
Writing 0x555 to Bits[D11:D0] performs a software reset of the AD5735.
SPI code
If the watchdog timer feature is enabled, 0x195 must be written to the software register
(Bits[D11:D0]) within the programmed timeout period (see Table 22).
Table 27. Programming the DC-to-DC Control Register
D15
0
1
D14
1
D13
1
D12 to D7
X1
D6
DC-DC comp
D5 to D4
DC-DC phase
D3 to D2
DC-DC freq
D1 to D0
DC-DC MaxV
X = don’t care.
Table 28. DC-to-DC Control Register Bit Descriptions
Bit Name
DC-DC Comp
DC-DC Phase
DC-DC Freq
DC-DC MaxV
Description
Selects the internal compensation resistor or an external compensation resistor for the dc-to-dc converter. See the
DC-to-DC Converter Compensation Capacitors section and the AICC Supply Requirements—Slewing section.
0 = selects the internal 150 kΩ compensation resistor (default).
1 = bypasses the internal compensation resistor. When this bit is set to 1, an external compensation resistor must
be used; this resistor is placed at the COMPDCDC_x pin in series with the 10 nF dc-to-dc compensation capacitor to
ground. Typically, a resistor of ~50 kΩ is recommended.
User-programmable dc-to-dc converter phase (between channels).
00 = all dc-to-dc converters clock on the same edge (default).
01 = Channel A and Channel B clock on the same edge; Channel C and Channel D clock on the opposite edge.
10 = Channel A and Channel C clock on the same edge; Channel B and Channel D clock on the opposite edge.
11 = Channel A, Channel B, Channel C, and Channel D clock 90° out of phase from each other.
Switching frequency for the dc-to-dc converter; this frequency is divided down from the internal 13 MHz oscillator
(see Figure 67 and Figure 68).
00 = 250 kHz ± 10%.
01 = 410 kHz ± 10% (default).
10 = 650 kHz ± 10%.
Maximum allowed VBOOST_x voltage supplied by the dc-to-dc converter.
00 = 23 V + 1 V/−1.5 V (default).
01 = 24.5 V ± 1 V.
10 = 27 V ± 1 V.
11 = 29.5 V ± 1 V.
Rev. B | Page 36 of 48
Data Sheet
AD5735
Slew Rate Control Register
This register is used to program the slew rate control for the
selected DAC channel. This feature is available on both the
current and voltage outputs. The slew rate control is enabled/
disabled and programmed on a per-channel basis. See Table 29
and the Digital Slew Rate Control section for more information.
READBACK OPERATION
Readback mode is invoked by setting the R/W bit = 1 in the serial
input register write. See Table 30 for the bits associated with a readback operation. The DUT_AD1 and DUT_AD0 bits, in association
with Bits[RD4:RD0], select the register to be read (see Table 31).
The remaining data bits in the write sequence are don’t care bits.
During the next SPI transfer, the data that appears on the SDO
output contains the data from the previously addressed register
(see Figure 4). This second SPI transfer should be either a request
to read another register on a third data transfer or a no
operation command. The no operation command for DUT
Address 00 is 0x1CE000, for other DUT addresses, Bit D22 and
Bit D21 are set accordingly.
Readback Example
To read back the gain register of AD5735 Device 1, Channel A,
implement the following sequence:
1.
2.
Write 0xA80000 to the input register to configure Device
Address 1 for read mode with the gain register of Channel A
selected. The data bits, D15 to D0, are don’t care bits.
Execute another read command or a no operation command (0x3CE000). During this command, the data from
the Channel A gain register is clocked out on the SDO line.
Table 29. Programming the Slew Rate Control Register
D15
0
1
D14
0
D13
0
D12
SREN
D11 to D7
X1
D6 to D3
SR_CLOCK
D2 to D0
SR_STEP
X = don’t care.
Table 30. Input Shift Register for a Read Operation
MSB
D23
R/W
1
D22
DUT_AD1
D21
DUT_AD0
D20
RD4
D19
RD3
D18
RD2
D17
RD1
X = don’t care.
Table 31. Read Addresses (Bits[RD4:RD0])
RD4
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
1
1
1
RD3
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
1
1
1
RD2
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
RD1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
RD0
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
Function
Read DAC A data register
Read DAC B data register
Read DAC C data register
Read DAC D data register
Read DAC A control register
Read DAC B control register
Read DAC C control register
Read DAC D control register
Read DAC A gain register
Read DAC B gain register
Read DAC C gain register
Read DAC D gain register
Read DAC A offset register
Read DAC B offset register
Read DAC C offset register
Read DAC D offset register
Read DAC A clear code register
Read DAC B clear code register
Read DAC C clear code register
Read DAC D clear code register
Read DAC A slew rate control register
Read DAC B slew rate control register
Read DAC C slew rate control register
Read DAC D slew rate control register
Read status register
Read main control register
Read dc-to-dc control register
Rev. B | Page 37 of 48
D16
RD0
LSB
D15 to D0
X1
AD5735
Data Sheet
Status Register
read back on the SDO pin during every write sequence. Alternatively, if the STATREAD bit is not set, the status register can be
read using the normal readback operation (see the Readback
Operation section).
The status register is a read-only register. This register contains
any fault information, as a well as a ramp active bit (Bit D9) and
a user toggle bit (Bit D11). When the STATREAD bit in the
main control register is set, the status register contents can be
Table 32. Decoding the Status Register
MSB
D15
D14
D13
D12
D11
D10
DC-DCD DC-DCC DC-DCB DC-DCA User
PEC
toggle error
D9
Ramp
active
D8
Over
temp
D7
VOUT_D
fault
D6
VOUT_C
fault
D5
VOUT_B
fault
D4
VOUT_A
fault
D3
IOUT_D
fault
D2
IOUT_C
fault
D1
IOUT_B
fault
LSB
D0
IOUT_A
fault
Table 33. Status Register Bit Descriptions
Bit Name
DC-DCD
Description
In current output mode, this bit is set if the dc-to-dc converter on Channel D cannot maintain compliance, for example, if
the dc-to-dc converter is reaching its VMAX voltage; in this case, the IOUT_D fault bit is also set. See the DC-to-DC Converter VMAX
Functionality section for more information about the operation of this bit under this condition.
In voltage output mode, this bit is set if the dc-to-dc converter on Channel D is unable to regulate to 15 V as expected.
When this bit is set, it does not result in the FAULT pin going high.
DC-DCC
In current output mode, this bit is set if the dc-to-dc converter on Channel C cannot maintain compliance, for example, if
the dc-to-dc converter is reaching its VMAX voltage; in this case, the IOUT_C fault bit is also set. See the DC-to-DC Converter VMAX
Functionality section for more information about the operation of this bit under this condition.
In voltage output mode, this bit is set if the dc-to-dc converter on Channel C is unable to regulate to 15 V as expected.
When this bit is set, it does not result in the FAULT pin going high.
DC-DCB
In current output mode, this bit is set if the dc-to-dc converter on Channel B cannot maintain compliance, for example, if
the dc-to-dc converter is reaching its VMAX voltage; in this case, the IOUT_B fault bit is also set. See the DC-to-DC Converter VMAX
Functionality section for more information about the operation of this bit under this condition.
In voltage output mode, this bit is set if the dc-to-dc converter on Channel B is unable to regulate to 15 V as expected.
When this bit is set, it does not result in the FAULT pin going high.
DC-DCA
In current output mode, this bit is set if the dc-to-dc converter on Channel A cannot maintain compliance, for example, if
the dc-to-dc converter is reaching its VMAX voltage; in this case, the IOUT_A fault bit is also set. See the DC-to-DC Converter VMAX
Functionality section for more information about the operation of this bit under this condition.
In voltage output mode, this bit is set if the dc-to-dc converter on Channel A is unable to regulate to 15 V as expected.
When this bit is set, it does not result in the FAULT pin going high.
User Toggle
PEC Error
Ramp Active
Over Temp
VOUT_D Fault
VOUT_C Fault
VOUT_B Fault
VOUT_A Fault
IOUT_D Fault
IOUT_C Fault
IOUT_B Fault
IOUT_A Fault
User toggle bit. This bit is set or cleared via the software register and can be used to verify data communications, if needed.
Denotes a PEC error on the last data-word received over the SPI interface.
This bit is set while any output channel is slewing (digital slew rate control is enabled on at least one channel).
This bit is set if the AD5735 core temperature exceeds approximately 150°C.
This bit is set if a fault is detected on the VOUT_D pin.
This bit is set if a fault is detected on the VOUT_C pin.
This bit is set if a fault is detected on the VOUT_B pin.
This bit is set if a fault is detected on the VOUT_A pin.
This bit is set if a fault is detected on the IOUT_D pin.
This bit is set if a fault is detected on the IOUT_C pin.
This bit is set if a fault is detected on the IOUT_B pin.
This bit is set if a fault is detected on the IOUT_A pin.
Rev. B | Page 38 of 48
Data Sheet
AD5735
DEVICE FEATURES
FAULT OUTPUT
The AD5735 is equipped with a FAULT pin, an active low,
open-drain output that allows several AD5735 devices to be
connected together to one pull-up resistor for global fault
detection. The FAULT pin is forced active by any one of the
following fault conditions:
•
•
•
•
DAC
INPUT
REGISTER
DAC DATA
REGISTER
DAC
09961-075
GAIN (M)
REGISTER
OFFSET (C)
REGISTER
The voltage at IOUT_x attempts to rise above the compliance
range due to an open-loop circuit or insufficient power
supply voltage. The internal circuitry that develops the
fault output avoids using a comparator with windowed
limits because this requires an actual output error before
the FAULT output becomes active. Instead, the signal is
generated when the internal amplifier in the output stage
has less than approximately 1 V of remaining drive
capability. Thus, the FAULT output is activated slightly
before the compliance limit is reached.
A short circuit is detected on a voltage output pin. The
short-circuit current is limited to 16 mA or 8 mA, which
is programmable by the user. If the AD5735 is used in unipolar supply mode, a short-circuit fault may be generated
if the output voltage is below 50 mV.
An interface error is detected due to a PEC failure (see the
Packet Error Checking section).
The core temperature of the AD5735 exceeds approximately 150°C.
The VOUT_x fault, IOUT_x fault, PEC error, and over temp bits
of the status register are used in conjunction with the FAULT
output to inform the user which fault condition caused the
FAULT output to be activated.
VOLTAGE OUTPUT SHORT-CIRCUIT PROTECTION
Under normal operation, the voltage output sinks/sources up
to 12 mA and maintains specified operation. The maximum
output current or short-circuit current is programmable by
the user and can be set to 16 mA or 8 mA. If a short circuit is
detected, the FAULT pin goes low, and the relevant VOUT_x fault
bit is set in the status register (see Table 33).
DIGITAL OFFSET AND GAIN CONTROL
Each DAC channel has a gain (M) register and an offset (C)
register, which allow trimming out of the gain and offset errors
of the entire signal chain. Data from the DAC data register is
operated on by a digital multiplier and adder controlled by the
contents of the gain and offset registers; the calibrated DAC
data is then stored in the DAC input register (see Figure 75).
Figure 75. Digital Offset and Gain Control
Although Figure 75 indicates a multiplier and adder for each
channel, the device has only one multiplier and one adder,
which are shared by all four channels. This design has implications for the update speed when several channels are updated
at once (see Table 3).
When data is written to the gain (M) or offset (C) register, the
output is not automatically updated. Instead, the next write to
the DAC channel uses the new gain and offset values to perform
a new calibration and automatically updates the channel.
The output data from the calibration is routed to the DAC input
register. This data is then loaded to the DAC, as described in the
Serial Interface section. Both the gain register and the offset
register have 12 bits of resolution. The correct order to calibrate
the gain and offset is to first calibrate the gain and then calibrate
the offset.
The value (in decimal) that is written to the DAC input register
can be calculated as follows:
Code DACRegister = D ×
( M + 1)
212
+ C − 211
(1)
where:
D is the code loaded to the DAC data register of the
DAC channel.
M is the code in the gain register (default code = 212 − 1).
C is the code in the offset register (default code = 211).
STATUS READBACK DURING A WRITE
The AD5735 can be configured to read back the contents of
the status register during every write sequence. This feature is
enabled using the STATREAD bit in the main control register.
When this feature is enabled, the user can continuously monitor
the status register and act quickly in the case of a fault.
When status readback during a write is enabled, the contents
of the 16-bit status register (see Table 33) are output on the SDO
pin, as shown in Figure 5.
When the AD5735 is powered up, the status readback during a
write feature is disabled. When this feature is enabled, readback
of registers other than the status register is not available. To read
back any other register, clear the STATREAD bit before following
the readback sequence (see the Readback Operation section).
The STATREAD bit can be set high again after the register read.
Rev. B | Page 39 of 48
AD5735
Data Sheet
ASYNCHRONOUS CLEAR
ignored. If status readback during a write is disabled, the user
can still use the normal readback operation to monitor status
register activity with PEC.
CLEAR is an active high, edge sensitive input that allows the
output to be cleared to a preprogrammed 12-bit code. This code
is user-programmable via a per-channel 12-bit clear code register.
WATCHDOG TIMER
When enabled, an on-chip watchdog timer generates an alert
signal if 0x195 is not written to the software register within the
programmed timeout period. This feature is useful to ensure
that communication has not been lost between the MCU and
the AD5735 and that the datapath lines are working properly
(that is, SDIN, SCLK, and SYNC). If 0x195 is not received by
the software register within the timeout period, the ALERT pin
signals a fault condition. The ALERT pin is active high and can
be connected directly to the CLEAR pin to enable a clear in the
event that communication from the MCU is lost.
For a channel to be cleared, set the CLR_EN bit in the DAC
control register for that channel. If the clear function on a
channel is not enabled, the output remains in its current state,
independent of the level of the CLEAR pin.
When the CLEAR signal returns low, the relevant outputs remain
cleared until a new value is programmed to them.
PACKET ERROR CHECKING
To verify that data has been received correctly in noisy environments, the AD5735 offers the option of packet error checking
based on an 8-bit cyclic redundancy check (CRC-8). The device
controlling the AD5735 should generate an 8-bit frame check
sequence using the following polynomial:
To enable the watchdog timer and set the timeout period (5 ms,
10 ms, 100 ms, or 200 ms), program the main control register
(see Table 21 and Table 22).
C(x) = x8 + x2 + x1 + 1
ALERT OUTPUT
This value is added to the end of the data-word, and 32 bits are
sent to the AD5735 before SYNC goes high. If the AD5735 sees a
32-bit frame, it performs the error check when SYNC goes high.
If the error check is valid, the data is written to the selected register.
If the error check fails, the FAULT pin goes low and the PEC error
bit in the status register is set. After the status register is read,
FAULT returns high (assuming that there are no other faults),
and the PEC error bit is cleared automatically.
The AD5735 is equipped with an ALERT pin. This pin is an
active high CMOS output. The AD5735 also has an internal
watchdog timer. When enabled, the watchdog timer monitors
SPI communications. If 0x195 is not received by the software
register within the timeout period, the ALERT pin is activated.
INTERNAL REFERENCE
The AD5735 contains an integrated 5 V voltage reference with
initial accuracy of ±5 mV maximum and a temperature coefficient
of ±10 ppm/°C maximum. The reference voltage is buffered and
is externally available for use elsewhere within the system.
UPDATE ON SYNC HIGH
SYNC
EXTERNAL CURRENT SETTING RESISTOR
SCLK
MSB
D23
SDIN
RSET is an internal sense resistor that is part of the voltage-tocurrent conversion circuitry (see Figure 71). The stability of the
output current value over temperature is dependent on the stability
of the RSET value. To improve the stability of the output current
over temperature, the internal RSET resistor, R1, can be bypassed
and an external, 15 kΩ, low drift resistor can be connected to
the RSET_x pin of the AD5735. The external resistor is selected
via the DAC control register (see Table 24).
LSB
D0
24-BIT DATA
24-BIT DATA TRANSFER—NO ERROR CHECKING
UPDATE ON SYNC HIGH
ONLY IF ERROR CHECK PASSED
SYNC
SCLK
MSB
D31
FAULT
24-BIT DATA
D7
D0
8-BIT CRC
FAULT PIN GOES LOW
IF ERROR CHECK FAILS
09961-180
SDIN
LSB
D8
32-BIT DATA TRANSFER WITH ERROR CHECKING
Figure 76. PEC Timing
Packet error checking can be used for transmitting and receiving
data packets. If status readback during a write is enabled, the PEC
values returned during the status readback operation should be
Table 1 provides the performance specifications for the AD5735
with both the internal RSET resistor and an external, 15 kΩ RSET
resistor. The use of an external RSET resistor allows for improved
performance over the internal RSET resistor option. The external
RSET resistor specifications assume an ideal resistor; the actual
performance depends on the absolute value and temperature
coefficient of the resistor used. This directly affects the gain error
of the output and, thus, the total unadjusted error. To arrive at
the gain/TUE error of the output with a specific external RSET
resistor, add the absolute error percentage of the RSET resistor
directly to the gain/TUE error of the AD5735 with the external
RSET resistor, as shown in Table 1 (expressed in % FSR).
Rev. B | Page 40 of 48
Data Sheet
AD5735
DIGITAL SLEW RATE CONTROL
The digital slew rate control feature of the AD5735 allows the
user to control the rate at which the output value changes. This
feature is available on both the current and voltage outputs. With
the slew rate control feature disabled, the output value changes
at a rate limited by the output drive circuitry and the attached
load. To reduce the slew rate, the user can enable the digital slew
rate control feature using the SREN bit of the slew rate control
register (see Table 29).
When slew rate control is enabled, the output, instead of slewing
directly between two values, steps digitally at a rate defined by
the SR_CLOCK and SR_STEP parameters. These parameters
are accessible via the slew rate control register (see Table 29).
•
•
SR_CLOCK defines the rate at which the digital slew is
updated; for example, if the selected update rate is 8 kHz,
the output is updated every 125 µs.
SR_STEP defines by how much the output value changes
at each update.
Together, these parameters define the rate of change of the
output value. Table 34 and Table 35 list the range of values for
the SR_CLOCK and SR_STEP parameters, respectively.
Table 34. Slew Rate Update Clock Options
SR_CLOCK
0000
0001
0010
0011
0100
0101
0110
0111
1000
1001
1010
1011
1100
1101
1110
1111
1
Update Clock Frequency1
64 kHz
32 kHz
16 kHz
8 kHz
4 kHz
2 kHz
1 kHz
500 Hz
250 Hz
125 Hz
64 Hz
32 Hz
16 Hz
8 Hz
4 Hz
0.5 Hz
These clock frequencies are divided down from the 13 MHz internal
oscillator (see Table 1, Figure 67, and Figure 68).
The following equation describes the slew rate as a function of
the step size, the update clock frequency, and the LSB size.
Slew Rate =
Output Change
Step Size × Update Clock Frequency × LSB Size
where:
Slew Rate is expressed in seconds.
Output Change is expressed in amperes for IOUT_x or in
volts for VOUT_x.
The update clock frequency for any given value is the same for
all output ranges. The step size, however, varies across output
ranges for a given value of step size because the LSB size is
different for each output range.
When the slew rate control feature is enabled, all output changes
occur at the programmed slew rate (see the DC-to-DC Converter
Settling Time section for more information). For example, if the
CLEAR pin is asserted, the output slews to the clear value at the
programmed slew rate (assuming that the channel is enabled to
be cleared).
If more than one channel is enabled for digital slew rate control,
care must be taken when asserting the CLEAR pin. If a channel
under slew rate control is slewing when the CLEAR pin is asserted,
other channels under slew rate control may change directly to
their clear code not under slew rate control.
DYNAMIC POWER CONTROL
When configured in current output mode, the AD5735 provides
integrated dynamic power control using a dc-to-dc boost converter
circuit. This circuit reduces power consumption compared with
standard designs.
In standard current input module designs, the load resistor
values can range from typically 50 Ω to 750 Ω. Output module
systems must source enough voltage to meet the compliance
voltage requirement across the full range of load resistor values.
For example, in a 4 mA to 20 mA loop when driving 20 mA, a
compliance voltage of >15 V is required. When driving 20 mA
into a 50 Ω load, a compliance voltage of only 1 V is required.
The AD5735 circuitry senses the output voltage and regulates
this voltage to meet the compliance requirements plus a small
headroom voltage. The AD5735 is capable of driving up to
24 mA through a 1 kΩ load.
Table 35. Slew Rate Step Size Options
SR_STEP
000
001
010
011
100
101
110
111
Step Size (LSB)
1
2
4
16
32
64
128
256
Rev. B | Page 41 of 48
AD5735
Data Sheet
DC-TO-DC CONVERTERS
DC-to-DC Converter VMAX Functionality
The AD5735 contains four independent dc-to-dc converters.
These are used to provide dynamic control of the VBOOST_x supply
voltage for each channel (see Figure 71). Figure 77 shows the
discrete components needed for the dc-to-dc circuitry, and the
following sections describe component selection and operation
of this circuitry.
The maximum VBOOST_x voltage is set in the dc-to-dc control
register (23 V, 24.5 V, 27 V, or 29.5 V; see Table 28). When the
maximum voltage is reached, the dc-to-dc converter is disabled,
and the VBOOST_x voltage is allowed to decay by ~0.4 V. After the
VBOOST_x voltage decays by ~0.4 V, the dc-to-dc converter is
reenabled, and the voltage ramps up again to VMAX, if still
required. This operation is shown in Figure 78.
10Ω
VBOOST_x
CFILTER
0.1µF
SWx
29.6
Table 36. Recommended Components for a DC-to-DC Converter
Component
XAL4040-103
GRM32ER71H475KA88L
PMEG3010BEA
Value
10 µH
4.7 µF
0.285 VF
0mA TO 24mA RANGE, 24mA OUTPUT
OUTPUT UNLOADED
29.4
Figure 77. DC-to-DC Circuit
Symbol
LDCDC
CDCDC
DDCDC
VMAX
DC-DCx BIT
29.5
Manufacturer
Coilcraft®
Murata
NXP
When a channel current output is enabled, the converter regulates
the VBOOST_x supply to 7.4 V (±5%) or (IOUT × RLOAD + Headroom),
whichever is greater (see Figure 51 for a plot of headroom
supplied vs. output current). In voltage output mode with the
output disabled, the converter regulates the VBOOST_x supply to
15 V (±5%). In current output mode with the output disabled,
the converter regulates the VBOOST_x supply to 7.4 V (±5%).
Within a channel, the VOUT_x and IOUT_x stages share a common
VBOOST_x supply; therefore, the outputs of the IOUT_x and VOUT_x
stages can be tied together (see the Voltage and Current Output
Pins on the Same Terminal section).
DC-to-DC Converter Settling Time
In current output mode, the settling time for a step greater than
~1 V (IOUT × RLOAD) is dominated by the settling time of the dc-todc converter. The exception to this is when the required voltage at
the IOUT_x pin plus the compliance voltage is below 7.4 V (±5%).
Figure 47 shows a typical plot of the output settling time. This
plot is for a 1 kΩ load. The settling time for smaller loads is faster.
The settling time for current steps less than 24 mA is also faster.
29.1
DC-DC MaxV BITS = 29.5V
DC-DCx BIT = 1
29.0
fSW = 410kHz
28.9
TA = 25°C
28.7
DC-DCx BIT = 0
28.6
0
DC-to-DC Converter Operation
DC-to-DC Converter Output Voltage
29.2
28.8
It is recommended that a 10 Ω, 100 nF low-pass RC filter be
placed after CDCDC. This filter consumes a small amount of power
but reduces the amount of ripple on the VBOOST_x supply.
The on-board dc-to-dc converters use a constant frequency, peak
current mode control scheme to step up an AVCC input of 4.5 V
to 5.5 V to drive the AD5735 output channel. These converters
are designed to operate in discontinuous conduction mode with
a duty cycle of <90% typical. Discontinuous conduction mode
refers to a mode of operation where the inductor current goes
to zero for an appreciable percentage of the switching cycle. The
dc-to-dc converters are nonsynchronous; that is, they require an
external Schottky diode.
29.3
0.5
1.0
1.5
2.0
2.5
TIME (ms)
3.0
3.5
4.0
09961-183
CDCDC
4.7µF
RFILTER
VBOOST_x VOLTAGE (mV)
CIN
≥10µF
DDCDC
10µH
09961-077
LDCDC
AV CC
Figure 78. Operation on Reaching VMAX
As shown in Figure 78, the DC-DCx bit in the status register
is asserted when the AD5735 ramps up to the VMAX value but
is deasserted when the voltage decays to VMAX − ~0.4 V.
DC-to-DC Converter On-Board Switch
The AD5735 contains a 0.425 Ω internal switch. The switch
current is monitored on a pulse-by-pulse basis and is limited
to 0.8 A peak current.
DC-to-DC Converter Switching Frequency and Phase
The AD5735 dc-to-dc converter switching frequency can be
selected from the dc-to-dc control register (see Table 28). The
phasing of the channels can also be adjusted so that the dc-to-dc
converters can clock on different edges. For typical applications,
a 410 kHz frequency is recommended. At light loads (low output
current and small load resistor), the dc-to-dc converter enters a
pulse-skipping mode to minimize switching power dissipation.
DC-to-DC Converter Inductor Selection
For typical 4 mA to 20 mA applications, a 10 µH inductor (such
as the XAL4040-103 from Coilcraft), combined with a switching
frequency of 410 kHz, allows up to 24 mA to be driven into a
load resistance of up to 1 kΩ with an AVCC supply of 4.5 V to
5.5 V. It is important to ensure that the inductor can handle the
peak current without saturating, especially at the maximum
ambient temperature. If the inductor enters saturation mode,
efficiency decreases. The inductance value also drops during
saturation and may result in the dc-to-dc converter circuit not
being able to supply the required output power.
Rev. B | Page 42 of 48
Data Sheet
AD5735
DC-to-DC Converter External Schottky Diode Selection
AICC SUPPLY REQUIREMENTS—STATIC
The AD5735 requires an external Schottky diode for correct
operation. Ensure that the Schottky diode is rated to handle the
maximum reverse breakdown voltage expected in operation
and that the maximum junction temperature of the diode is not
exceeded. The average current of the diode is approximately
equal to the ILOAD current. Diodes with larger forward voltage
drops result in a decrease in efficiency.
The dc-to-dc converter is designed to supply a VBOOST_x voltage of
VBOOST_x = IOUT × RLOAD + Headroom
See Figure 51 for a plot of headroom supplied vs. output
current. Therefore, for a fixed load and output voltage, the
output current of the dc-to-dc converter can be calculated
by the following formula:
DC-to-DC Converter Compensation Capacitors
The output capacitor affects the ripple voltage of the dc-to-dc
converter and indirectly limits the maximum slew rate at which
the channel output current can rise. The ripple voltage is caused
by a combination of the capacitance and the equivalent series
resistance (ESR) of the capacitor. For typical applications, a
ceramic capacitor of 4.7 µF is recommended. Larger capacitors
or parallel capacitors improve the ripple at the expense of
reduced slew rate. Larger capacitors also affect the current
requirements of the AVCC supply while slewing (see the AICC
Supply Requirements—Slewing section). The capacitance at
the output of the dc-to-dc converter should be >3 µF under all
operating conditions.
The input capacitor provides much of the dynamic current
required for the dc-to-dc converter and should be a low ESR
component. For the AD5735, a low ESR tantalum or ceramic
capacitor of 10 µF is recommended for typical applications.
Ceramic capacitors must be chosen carefully because they can
exhibit a large sensitivity to dc bias voltages and temperature.
X5R or X7R dielectrics are preferred because these capacitors
remain stable over wider operating voltage and temperature
ranges. Care must be taken if selecting a tantalum capacitor to
ensure a low ESR value.
=
I OUT × V BOOST
(3)
ηVBOOST × AVCC
where:
IOUT is the output current from IOUT_x in amperes.
ηVBOOST is the efficiency at VBOOST_x as a fraction (see Figure 53
and Figure 54).
AICC SUPPLY REQUIREMENTS—SLEWING
The AICC current requirement while slewing is greater than in
static operation because the output power increases to charge
the output capacitance of the dc-to-dc converter. This transient
current can be quite large (see Figure 79), although the methods
described in the Reducing AICC Current Requirements section
can reduce the requirements on the AVCC supply.
If not enough AICC current can be provided, the AVCC voltage
drops. Due to this AVCC drop, the AICC current required for
slewing increases further, causing the voltage at AVCC to drop
further (see Equation 3). In this case, the VBOOST_x voltage and,
therefore, the output voltage, may never reach their intended
values. Because the AVCC voltage is common to all channels, this
voltage drop may also affect other channels.
0.8
30
0.7
25
0.6
0mA TO 24mA RANGE
1kΩ LOAD
fSW = 410kHz
INDUCTOR = 10µH (XAL4040-103)
TA = 25°C
0.5
0.4
20
15
0.3
10
0.2
AICC
IOUT
VBOOST
0.1
Rev. B | Page 43 of 48
5
0
0
0
0.5
1.0
1.5
TIME (ms)
2.0
2.5
Figure 79. AICC Current vs. Time for 24 mA Step Through 1 kΩ Load
with Internal Compensation Resistor
09961-184
DC-to-DC Converter Input and Output Capacitor
Selection
Power Out
Efficiency × AVCC
IOUT_x CURRENT (mA)/ VBOOST_x VOLTAGE (V)
Alternatively, an external compensation resistor can be used in
series with the compensation capacitor by setting the DC-DC
comp bit in the dc-to-dc control register (see Table 28). In this
case, a resistor of ~50 kΩ is recommended. The advantages of this
configuration are described in the AICC Supply Requirements—
Slewing section. For typical applications, a 10 nF dc-to-dc compensation capacitor is recommended.
AI CC =
AICC CURRENT (A)
Because the dc-to-dc converter operates in discontinuous conduction mode, the uncompensated transfer function is essentially a
single-pole transfer function. The pole frequency of the transfer
function is determined by the output capacitance, input and output
voltage, and output load of the dc-to-dc converter. The AD5735
uses an external capacitor in conjunction with an internal 150 kΩ
resistor to compensate the regulator loop.
(2)
AD5735
Data Sheet
Reducing AICC Current Requirements
Using Slew Rate Control
Two main methods can be used to reduce the AICC current
requirements. One method is to add an external compensation
resistor, and the other is to use slew rate control. These methods
can be used together.
Using slew rate control can greatly reduce the current requirements of the AVCC supply, as shown in Figure 82.
0.8
0.5
20
0.4
16
0.3
12
0.2
8
AICC
IOUT
VBOOST
0.1
4
0
0
0
0.5
1.0
1.5
TIME (ms)
0.3
12
0.2
8
0.1
4
2.0
2.5
AICC CURRENT (A)
0.6
28
24
0.5
20
0.4
16
0.3
12
0.2
8
0.1
4
0
0
0.5
1.0
1.5
TIME (ms)
2.0
IOUT_x CURRENT (mA)/V BOOST_x VOLTAGE (V)
0mA TO 24mA RANGE
500Ω LOAD
fSW = 410kHz
INDUCTOR = 10µH (XAL4040-103)
TA = 25°C
0
2.5
2
3
TIME (ms)
4
5
6
When using slew rate control, it is important to remember that
the output cannot slew faster than the dc-to-dc converter. The
dc-to-dc converter slews slowest at higher currents through large
loads (for example, 1 kΩ). The slew rate is also dependent on
the configuration of the dc-to-dc converter. Two examples of
the dc-to-dc converter output slew are shown in Figure 80 and
Figure 81. (VBOOST corresponds to the output voltage of the
dc-to-dc converter.)
09961-186
0.7
1
Figure 82. AICC Current vs. Time for 24 mA Step Through 1 kΩ Load
with Slew Rate Control
32
AICC
IOUT
VBOOST
0
0
Figure 80. AICC Current vs. Time for 24 mA Step Through 1 kΩ Load
with External 51 kΩ Compensation Resistor
0.8
IOUT_x CURRENT (mA)/ VBOOST_x VOLTAGE (V)
16
0
IOUT_x CURRENT (mA)/ VBOOST_x VOLTAGE (V)
24
20
0.4
09961-185
AICC CURRENT (A)
0.6
28
24
AICC
IOUT
VBOOST
0.5
32
0mA TO 24mA RANGE
1kΩ LOAD
fSW = 410kHz
INDUCTOR = 10µH (XAL4040-103)
TA = 25°C
0.7
0.6
AICC CURRENT (A)
A compensation resistor can be placed at the COMPDCDC_x pin
in series with the 10 nF compensation capacitor. A 51 kΩ external compensation resistor is recommended. This compensation
increases the slew time of the current output but reduces the AICC
transient current requirements. Figure 80 shows a plot of AICC
current for a 24 mA step through a 1 kΩ load when using a 51 kΩ
compensation resistor. The compensation resistor reduces the
current requirements through smaller loads even further, as
shown in Figure 81.
28
09961-187
0.7
Adding an External Compensation Resistor
0.8
32
0mA TO 24mA RANGE
1kΩ LOAD
fSW = 410kHz
INDUCTOR = 10µH (XAL4040-103)
TA = 25°C
Figure 81. AICC Current vs. Time for 24 mA Step Through 500 Ω Load
with External 51 kΩ Compensation Resistor
Rev. B | Page 44 of 48
Data Sheet
AD5735
APPLICATIONS INFORMATION
VOLTAGE AND CURRENT OUTPUT PINS ON THE
SAME TERMINAL
When using a channel of the AD5735, the current and voltage
output pins can be connected to two separate terminals or tied
together and connected to a single terminal. The two output
pins can be tied together because only the voltage output or the
current output can be enabled at any one time. When the current
output is enabled, the voltage output is in tristate mode, and when
the voltage output is enabled, the current output is in tristate mode.
When the two output pins are tied together, the POC pin must
be tied low and the POC bit in the main control register set to 0,
or, if the POC pin is tied high, the POC bit in the main control
register must be set to 1 before the current output is enabled.
As shown in the Absolute Maximum Ratings section, the output
tolerances are the same for both the voltage and current output
pins. The +VSENSE_x and −VSENSE_x connections are buffered so
that current leakage into these pins is negligible when the part
is operated in current output mode.
CURRENT OUTPUT MODE WITH INTERNAL RSET
When using the internal RSET resistor in current output mode,
the output is significantly affected by how many other channels
using the internal RSET are enabled and by the dc crosstalk from
these channels. The internal RSET specifications in Table 1 are
for all four channels enabled with the internal RSET selected and
outputting the same code.
For every channel enabled with the internal RSET, the offset error
decreases. For example, with one current output enabled using the
internal RSET, the offset error is 0.075% FSR. This value decreases
proportionally as more current channels are enabled; the offset
error is 0.056% FSR on each of two channels, 0.029% FSR on
each of three channels, and 0.01% FSR on each of four channels.
Similarly, the dc crosstalk when using the internal RSET is proportional to the number of current output channels enabled with the
internal RSET. For example, with the measured channel at 0x8000
and another channel going from zero to full scale, the dc crosstalk
is −0.011% FSR. With two other channels going from zero to full
scale, the dc crosstalk is −0.019% FSR, and with all three other
channels going from zero to full scale, it is −0.025% FSR.
For the full-scale error measurement in Table 1, all channels are
at 0xFFFF. This means that as any channel goes to zero scale, the
full-scale error increases due to the dc crosstalk. For example,
with the measured channel at 0xFFFF and three channels at zero
scale, the full-scale error is 0.025% FSR. Similarly, if only one
channel is enabled in current output mode with the internal RSET,
the full-scale error is 0.025% FSR + 0.075% FSR = 0.1% FSR.
PRECISION VOLTAGE REFERENCE SELECTION
To achieve the optimum performance from the AD5735 over its
full operating temperature range, a precision voltage reference
must be used. Care should be taken with the selection of the
precision voltage reference. The voltage applied to the reference
inputs is used to provide a buffered reference for the DAC cores.
Therefore, any error in the voltage reference is reflected in the
outputs of the AD5735.
Four possible sources of error must be considered when choosing
a voltage reference for high accuracy applications: initial accuracy,
long-term drift, temperature coefficient of the output voltage,
and output voltage noise.
Initial accuracy error on the output voltage of an external reference can lead to a full-scale error in the DAC. Therefore, to
minimize these errors, a reference with a low initial accuracy
error specification is preferred. Choosing a reference with an
output trim adjustment, such as the ADR435, allows a system
designer to trim out system errors by setting the reference
voltage to a voltage other than the nominal. The trim adjustment can be used at any temperature to trim out any error.
Long-term drift is a measure of how much the reference output
voltage drifts over time. A reference with a tight long-term drift
specification ensures that the overall solution remains relatively
stable over its entire lifetime.
The temperature coefficient of the reference output voltage affects
INL, DNL, and TUE. A reference with a tight temperature coefficient specification should be chosen to reduce the dependence
of the DAC output voltage on ambient temperature.
In high accuracy applications, which have a relatively low noise
budget, reference output voltage noise must be considered. Choosing a reference with as low an output noise voltage as practical
for the system resolution required is important. Precision voltage
references such as the ADR435 (XFET® design) produce low output
noise in the 0.1 Hz to 10 Hz bandwidth. However, as the circuit
bandwidth increases, filtering the output of the reference may
be required to minimize the output noise.
Table 37. Recommended Precision Voltage References
Part No.
ADR445
ADR02
ADR435
ADR395
AD586
Initial Accuracy
(mV Maximum)
±2
±3
±2
±5
±2.5
Long-Term Drift
(ppm Typical)
50
50
40
50
15
Rev. B | Page 45 of 48
Temperature Coefficient
(ppm/°C Maximum)
3
3
3
9
10
0.1 Hz to 10 Hz Noise
(µV p-p Typical)
2.25
10
8
8
4
AD5735
Data Sheet
DRIVING INDUCTIVE LOADS
MICROPROCESSOR INTERFACING
When driving inductive or poorly defined loads, a capacitor
may be required between the IOUT_x pin and the AGND pin to
ensure stability. A 0.01 µF capacitor between IOUT_x and AGND
ensures stability of a load of 50 mH. The capacitive component
of the load may cause slower settling, although this may be
masked by the settling time of the AD5735. There is no maximum capacitance limit for the current output of the AD5735.
Microprocessor interfacing to the AD5735 is via a serial bus
that uses a protocol compatible with microcontrollers and DSP
processors. The communication channel is a 3-wire minimum
interface consisting of a clock signal, a data signal, and a latch
signal. The AD5735 requires a 24-bit data-word with data valid
on the falling edge of SCLK.
TRANSIENT VOLTAGE PROTECTION
The AD5735 contains ESD protection diodes that prevent damage from normal handling. The industrial control environment
can, however, subject I/O circuits to much higher transients. To
protect the AD5735 from excessively high voltage transients,
external power diodes and a surge current limiting resistor (RP)
are required, as shown in Figure 83. A typical value for RP is 10 Ω.
The two protection diodes and the resistor (RP) must have appropriate power ratings.
AD5735-to-ADSP-BF527 Interface
The AD5735 can be connected directly to the SPORT interface
of the ADSP-BF527, an Analog Devices, Inc., Blackfin® DSP.
Figure 84 shows how the SPORT interface can be connected
to control the AD5735.
AD5735
RFILTER
CDCDC
4.7µF
10Ω
SPORT_TFS
SYNC
SPORT_TSCLK
SCLK
SPORT_DT0
SDIN
CFILTER
0.1µF
ADSP-BF527
VBOOST_x
IOUT_x
AGND
GPIO0
D2
RLOAD
09961-079
AD5735
D1
RP
Figure 83. Output Transient Voltage Protection
LDAC
09961-080
(FROM
DC-TO-DC
CONVERTER)
The DAC output update is initiated either on the rising edge of
LDAC or, if LDAC is held low, on the rising edge of SYNC. The
contents of the registers can be read using the readback function.
Figure 84. AD5735-to-ADSP-BF527 SPORT Interface
LAYOUT GUIDELINES
Grounding
Further protection can be provided using transient voltage
suppressors (TVSs), also referred to as transorbs. These components are available as unidirectional suppressors, which protect
against positive high voltage transients, and as bidirectional
suppressors, which protect against both positive and negative
high voltage transients. Transient voltage suppressors are available in a wide range of standoff and breakdown voltage ratings.
The TVS should be sized with the lowest breakdown voltage
possible while not conducting in the functional range of the
current output.
It is recommended that all field connected nodes be protected.
The voltage output node can be protected with a similar circuit,
where D2 and the transorb are connected to AVSS. For the voltage output node, the +VSENSE_x pin should also be protected with
a large value series resistance to the transorb, such as 5 kΩ. In
this way, the IOUT_x and VOUT_x pins can also be tied together and
share the same protection circuitry.
In any circuit where accuracy is important, careful consideration of the power supply and ground return layout helps to
ensure the rated performance. The printed circuit board on
which the AD5735 is mounted should be designed so that the
analog and digital sections are separated and confined to
certain areas of the board. If the AD5735 is in a system where
multiple devices require an AGND-to-DGND connection, the
connection should be made at one point only. The star ground
point should be established as close as possible to the device.
The GNDSWx pin and the ground connection for the AVCC
supply are referred to as PGND. PGND should be confined to
certain areas of the board, and the PGND-to-AGND connection
should be made at one point only.
Supply Decoupling
The AD5735 should have ample supply bypassing of 10 µF in
parallel with 0.1 µF on each supply, located as close to the package
as possible, ideally right up against the device. The 10 µF capacitors are the tantalum bead type. The 0.1 µF capacitors should
have low effective series resistance (ESR) and low effective series
inductance (ESL), such as the common ceramic types, which
provide a low impedance path to ground at high frequencies to
handle transient currents due to internal logic switching.
Rev. B | Page 46 of 48
Data Sheet
AD5735
•
The power supply lines of the AD5735 should use as large a trace
as possible to provide low impedance paths and reduce the effects
of glitches on the power supply line. Fast switching signals such
as clocks should be shielded with digital ground to prevent radiating noise to other parts of the board and should never be run
near the reference inputs. A ground line routed between the
SDIN and SCLK traces helps reduce crosstalk between them (not
required on a multilayer board that has a separate ground plane,
but separating the lines helps). It is essential to minimize noise on
the REFIN line because it couples through to the DAC output.
Avoid crossover of digital and analog signals. Traces on opposite sides of the board should run at right angles to each other
to reduce the effects of feedthrough on the board. A microstrip
technique is by far the best method, but it is not always possible
with a double-sided board. In this technique, the component
side of the board is dedicated to ground plane, and signal traces
are placed on the solder side.
DC-to-DC Converters
To achieve high efficiency, good regulation, and stability, a
well-designed printed circuit board layout is required.
Keep high current traces as short and as wide as possible.
The path from CIN through the inductor (LDCDC) to SWx
and PGND should be able to handle a minimum of 1 A.
Place the compensation components as close as possible to
the COMPDCDC_x pin.
Avoid routing high impedance traces near any node
connected to SWx or near the inductor to prevent radiated
noise injection.
•
•
GALVANICALLY ISOLATED INTERFACE
In many process control applications, it is necessary to provide
an isolation barrier between the controller and the unit being
controlled to protect and isolate the controlling circuitry from
any hazardous common-mode voltages that may occur. The
Analog Devices iCoupler® products can provide voltage isolation
in excess of 2.5 kV. The serial loading structure of the AD5735
makes it ideal for isolated interfaces because the number of interface lines is kept to a minimum. Figure 85 shows a 4-channel
isolated interface to the AD5735 using an ADuM1411. For
more information, visit www.analog.com.
MICROCONTROLLER
Follow these guidelines when designing printed circuit boards
(see Figure 77):
•
•
•
Keep the low ESR input capacitor, CIN, close to AVCC and
PGND.
Keep the high current path from CIN through the inductor
(LDCDC) to SWx and PGND as short as possible.
Keep the high current path from CIN through the inductor
(LDCDC), the diode (DDCDC), and the output capacitor
(CDCDC) as short as possible.
Rev. B | Page 47 of 48
ADuM1411
SERIAL CLOCK
OUT
VIA
SERIAL DATA
OUT
VIB
SYNC OUT
CONTROL OUT
VIC
VID
ENCODE
DECODE
ENCODE
DECODE
ENCODE
ENCODE
DECODE
DECODE
VOA
VOB
VOC
VOD
TO SCLK
TO SDIN
TO SYNC
TO LDAC
Figure 85. 4-Channel Isolated Interface to the AD5735
09961-081
Traces
AD5735
Data Sheet
OUTLINE DIMENSIONS
0.60 MAX
9.00
BSC SQ
0.60
MAX
64 1
49
PIN 1
INDICATOR
48
PIN 1
INDICATOR
8.75
BSC SQ
TOP VIEW
0.50
BSC
(BOTTOM VIEW)
0.50
0.40
0.30
SEATING
PLANE
0.25 MIN
7.50
REF
0.80 MAX
0.65 TYP
12° MAX
16
17
33
32
FOR PROPER CONNECTION OF
THE EXPOSED PAD, REFER TO
THE PIN CONFIGURATION AND
FUNCTION DESCRIPTIONS
SECTION OF THIS DATA SHEET.
0.05 MAX
0.02 NOM
0.30
0.23
0.18
0.20 REF
COMPLIANT TO JEDEC STANDARDS MO-220-VMMD-4
080108-C
1.00
0.85
0.80
7.25
7.10 SQ
6.95
EXPOSED PAD
Figure 86. 64-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
9 mm × 9 mm Body, Very Thin Quad
(CP-64-3)
Dimensions shown in millimeters
ORDERING GUIDE
Model 1
AD5735ACPZ
AD5735ACPZ-REEL7
1
Resolution (Bits)
12
12
Temperature Range
−40°C to +105°C
−40°C to +105°C
Z = RoHS Compliant Part.
©2011–2012 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D09961-0-5/12(B)
Rev. B | Page 48 of 48
Package Description
64-Lead LFCSP_VQ
64-Lead LFCSP_VQ
Package Option
CP-64-3
CP-64-3
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