HIP6019B TM Data Sheet October 1998 Advanced Dual PWM and Dual Linear Power Control Features The HIP6019B provides the power control and protection for four output voltages in high-performance microprocessor and computer applications. The IC integrates two PWM controllers, a linear regulator and a linear controller as well as the monitoring and protection functions into a single 28 lead SOIC package. One PWM controller regulates the microprocessor core voltage with a synchronous-rectified buck converter, while the second PWM controller supplies the computer’s 3.3V power with a standard buck converter. The linear controller regulates power for the GTL bus and the linear regulator provides power for the clock driver circuits. The HIP6019B includes an Intel-compatible, TTL 5-input digital-to-analog converter (DAC) that adjusts the core PWM output voltage from 2.1VDC to 3.5VDC in 0.1V increments and from 1.3VDC to 2.05VDC in 0.05V steps. The precision reference and voltage-mode control provide ±1% static regulation. The second PWM controller is user-adjustable for output levels between 3.0V and 3.5V with ±2% accuracy. The adjustable linear regulator uses an internal pass device to provide 2.5V ±2.5%. The adjustable linear controller drives an external N-Channel MOSFET to provide 1.5V ±2.5%. The HIP6019B monitors all the output voltages. A single Power Good signal is issued when the core is within ±10% of the DAC setting and the other levels are above their undervoltage levels. Additional built-in over-voltage protection for the core output uses the lower MOSFET to prevent output voltages above 115% of the DAC setting. The PWM controller’s over-current functions monitor the output current by sensing the voltage drop across the upper MOSFET’s rDS(ON), eliminating the need for a current sensing resistor. Pinout HIP6019B (SOIC) TOP VIEW FN4587 • Provides 4 Regulated Voltages - Microprocessor Core, I/O, Clock Chip and GTL Bus • Drives N-Channel MOSFETs • Operates from +5V and +12V Inputs • Simple Single-Loop Control Designs - Voltage-Mode PWM Control • Fast Transient Response - High-Bandwidth Error Amplifiers - Full 0% to 100% Duty Ratios • Excellent Output Voltage Regulation - Core PWM Output: ±1% Over Temperature - I/O PWM Output: ±2% Over Temperature - Other Outputs: ±2.5% Over Temperature • TTL-Compatible 5-Bit Digital-to-Analog Core Output Voltage Selection - Wide Range . . . . . . . . . . . . . . . . . . . 1.3V DC to 3.5VDC - 0.1V Steps . . . . . . . . . . . . . . . . . . . . 2.1VDC to 3.5VDC - 0.05V Steps . . . . . . . . . . . . . . . . . . 1.3VDC to 2.05VDC • Power-Good Output Voltage Monitor • Microprocessor Core Voltage Protection Against Shorted MOSFET • Over-Voltage and Over-Current Fault Monitors - Does Not Require Extra Current Sensing Element, Uses MOSFET’s rDS(ON) • Small Converter Size - Constant Frequency Operation - 200kHz Free-Running Oscillator; Programmable from 50kHz to 1MHz Applications UGATE2 1 28 VCC PHASE2 2 27 UGATE1 • Full Motherboard Power Regulation for Computers VID4 3 26 PHASE1 • Low-Voltage Distributed Power Supplies VID3 4 25 LGATE1 VID2 5 24 PGND VID1 6 23 OCSET1 VID0 7 22 VSEN1 PGOOD 8 21 FB1 OCSET2 9 20 COMP1 FB2 10 Ordering Information PART NUMBER HIP6019BCB TEMP. (oC) 0 to 70 PACKAGE 28 Ld SOIC PKG. NO. M28.3 19 FB3 COMP2 11 18 GATE3 17 GND SS 12 FAULT/RT 13 16 VOUT4 FB4 14 15 VSEN2 267 CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 321-724-7143 | Intersil (and design) is a trademark of Intersil Americas Inc. Copyright © Intersil Americas Inc. 2002. All Rights Reserved VSEN2 FB2 COMP2 PHASE2 GATE2 FB4 VOUT4 GATE3 2.5V VCC 0.25A VSEN2 + - + - + - + 4.3V - + + - 0.3V - INHIBIT + + - PWM COMP2 PWM2 1.26V + 1.26V GATE CONTROL DRIVE2 - + + - - + - + - + ERROR AMP2 - 268 + FB3 FAULT / RT LUV 4V 11µA FIGURE 1. SS OV VCC SOFTSTART AND FAULT LOGIC OC FAULT 200µA OSCILLATOR OC2 OC4 LINEAR UNDERVOLTAGE OCSET2 DACOUT - + - + - + - + OC1 COMP1 ERROR AMP1 FB1 115% 90% 110% VSEN1 PWM1 DRIVE1 RESET (POR) POWER-ON VCC VID4 VID0 VID2 VID1 VID3 LOWER DRIVE GATE CONTROL INHIBIT TTL D/A CONVERTER (DAC) PWM COMP1 - + - + 200µA OCSET1 VCC VCC GND PGND LGATE1 PHASE1 UGATE1 PGOOD HIP6019B Block Diagram HIP6019B Simplified Power System Diagram +5VIN PWM2 CONTROLLER VOUT1 PWM1 CONTROLLER VOUT2 HIP6019B LINEAR CONTROLLER VOUT3 LINEAR REGULATOR VOUT4 FIGURE 2. Typical Application +12VIN +5VIN CIN VCC OCSET2 OCSET1 POWERGOOD PGOOD VOUT2 Q3 LOUT2 3.0V TO 3.5V COUT2 UGATE2 UGATE1 PHASE2 PHASE1 CR2 Q1 Q2 LGATE1 FB2 VSEN1 HIP6019B COMP2 FB1 COMP1 FAULT / RT Q4 VID0 GATE3 VOUT3 1.5V VID1 FB3 VID2 VID3 COUT3 VID4 VOUT4 2.5V SS VOUT4 CSS FB4 COUT4 GND FIGURE 3. 269 COUT1 CR1 PGND VSEN2 LOUT1 VOUT1 1.3V TO 3.5V HIP6019B Absolute Maximum Ratings Thermal Information Supply Voltage, V CC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .+15V PGOOD, RT/FAULT, and GATE Voltage . . GND - 0.3V to VCC + 0.3V Input, Output or I/O Voltage . . . . . . . . . . . . . . . . . . GND -0.3V to 7V ESD Classification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Class 2 Thermal Resistance (Typical, Note 1) Operating Conditions Supply Voltage, V CC . . . . . . . . . . . . . . . . . . . . . . . . . . . +12V ±10% Ambient Temperature Range. . . . . . . . . . . . . . . . . . . . . 0oC to 70oC Junction Temperature Range . . . . . . . . . . . . . . . . . . . 0oC to 125oC θJA (oC/W) SOIC Package . . . . . . . . . . . . . . . . . . . . . . . . . . . . 60 50 SOIC Package (with 3 in2 of copper) . . . . . . . . . . . Maximum Junction Temperature (Plastic Package) . . . . . . . .150oC Maximum Storage Temperature Range . . . . . . . . . -65oC to 150oC Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . . 300oC (SOIC - Lead Tips Only) CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied. NOTE: 1. θJA is measured with the component mounted on an evaluation PC board in free air. Electrical Specifications Recommended Operating Conditions, Unless Otherwise Noted. Refer to Figures 1, 2 and 3 PARAMETER SYMBOL TEST CONDITIONS MIN TYP MAX UNITS - 10 - mA VCC SUPPLY CURRENT Nominal Supply ICC UGATE1, GATE2, GATE3, LGATE1, and VOUT4 Open POWER-ON RESET Rising VCC Threshold VOCSET = 4.5V 8.6 - 10.4 V Falling VCC Threshold VOCSET = 4.5V 8.2 - 10.2 V - 1.25 - V Rising VOCSET1 Threshold OSCILLATOR Free Running Frequency RT = OPEN 185 200 215 kHz Total Variation 6kΩ < RT to GND < 200kΩ -15 - +15 % - 1.9 - V P-P DAC(VID0-VID4) Input Low Voltage - - 0.8 V DAC(VID0-VID4) Input High Voltage 2.0 - - V DACOUT Voltage Accuracy -1.0 - +1.0 % 1.240 1.265 1.290 V -2.5 - 2.5 % - 75 87 % - 6 - % 180 230 - mA CSS Voltage < 4V 560 700 - mA VSEN3 = GATE3 -2.5 - 2.5 % - 75 87 % - 6 - % ∆VOSC Ramp Amplitude RT = Open REFERENCE AND DAC Reference Voltage (Pin FB2, FB3, and FB4) LINEAR REGULATOR Regulation 10mA < IVOUT4 < 150mA Under-Voltage Level FB4UV FB4 Rising Under-Voltage Hysteresis Over-Current Protection Over-Current Protection During Start-Up LINEAR CONTROLLER Regulation Under-Voltage Level FB3UV Under-Voltage Hysteresis 270 FB3 Rising HIP6019B Electrical Specifications Recommended Operating Conditions, Unless Otherwise Noted. Refer to Figures 1, 2 and 3 (Continued) PARAMETER SYMBOL TEST CONDITIONS MIN TYP MAX UNITS - 88 - dB - 15 - MHz COMP = 10pF - 6 - V/µs PWM CONTROLLER ERROR AMPLIFIERS DC Gain Gain-Bandwidth Product GBWP Slew Rate SR PWM CONTROLLER GATE DRIVERS Drive1 (and 2) Source IUGATE VCC = 12V, VUGATE1 (or VGATE2) = 6V - 1 - A Drive1 (and 2) Sink RUGATE VGATE-PHASE = 1V - 1.7 3.5 Ω Lower Gate Source ILGATE VCC = 12V, VLGATE = 1V - 1 - A Lower Gate Sink RLGATE VGATE = 1V - 1.4 3.0 Ω PROTECTION V OUT1 Over-Voltage Trip VSEN1 Rising 112 115 118 % VOUT2 Over-Voltage Trip VSEN2 Rising 4.1 4.3 4.5 V - 70 - kΩ VSEN2 Input Resistance FAULT Sourcing Current OCSET1(and 2) Current Source IOVP VFAULT/RT = 10.0V 10 14 - mA IOCSET VOCSET = 4.5VDC 170 200 230 µA - 11 - µA - - 1.0 V Soft-Start Current ISS Chip Shutdown Soft-Start Threshold POWER GOOD VOUT1 Upper Threshold VSEN1 Rising 108 - 110 % VOUT1 Under-Voltage VSEN1 Rising 92 - 94 % VOUT1 Hysteresis Upper/Lower Threshold - 2 - % VOUT2 Under-Voltage VSEN2 Rising 2.45 2.55 2.65 V - 100 - mV - - 0.5 V VOUT2 Under-Voltage Hysteresis PGOOD Voltage Low V PGOOD IPGOOD = -4mA Typical Performance Curves 140 120 RT PULLUP TO +12V CGATE = 4800pF 100 I CC (mA) RESISTANCE (kΩ) 1000 CUGATE1 = CUGATE2 = CLGATE1 = CGATE VVCC = 12V, VIN = 5V 100 80 CGATE = 3600pF 60 CGATE = 1500pF 40 10 CGATE = 660pF RT PULLDOWN TO VSS 10 100 SWITCHING FREQUENCY (kHz) FIGURE 4. RT RESISTANCE vs FREQUENCY 271 20 1000 100 200 300 400 500 600 700 800 900 SWITCHING FREQUENCY (kHz) FIGURE 5. BIAS SUPPLY CURRENT vs FREQUENCY 1000 HIP6019B Functional Pin Description VSEN1, VSEN2 (Pins 22 and 15) These pins are connected to the PWM converters’ output voltages. The PGOOD and OVP comparator circuits use these signals to report output voltage status and for overvoltage protection. VSEN2 provides the input power to the integrated linear regulator. OCSET1, OCSET2 (Pins 23 and 9) Connect a resistor (ROCSET) from this pin to the drain of the respective upper MOSFET. ROCSET, an internal 200µA current source (IOCSET), and the upper MOSFET onresistance (rDS(ON)) set the converter over-current (OC) trip point according to the following equation: I OCSET × R OCSET I PEAK = ---------------------------------------------------r D S ( ON ) An over-current trip cycles the soft-start function. Sustaining an over-current for 2 soft-start intervals shuts down the controller. PHASE1, PHASE2 (Pins 26 and 2) Connect the PHASE pins to the respective PWM converter’s upper MOSFET source. These pins are used to monitor the voltage drop across the upper MOSFETs for over-current protection. UGATE1, UGATE2 (Pins 27 and 1) Connect UGATE pins to the respective PWM converter’s upper MOSFET gate. These pins provide the gate drive for the upper MOSFETs. PGND (Pin 24) This is the power ground connection. Tie the synchronous PWM converter’s lower MOSFET source to this pin. LGATE1 (Pin 25) Connect LGATE1 to the synchronous PWM converter’s lower MOSFET gate. This pin provides the gate drive for the lower MOSFET. VCC (Pin 28) Additionally, OCSET1 is an output for the inverted FAULT signal (FAULT). If a fault condition causes FAULT to go high, OCSET1 will be simultaneously pulled to ground though an internal MOS device (typical rDS(ON) = 100Ω). Provide a 12V bias supply for the IC to this pin. This pin also provides the gate bias charge for all the MOSFETs controlled by the IC. SS (Pin 12) This pin provides oscillator switching frequency adjustment. By placing a resistor (RT) from this pin to GND, the nominal 200kHz switching frequency is increased according to the following equation: Connect a capacitor from this pin to ground. This capacitor, along with an internal 11µA current source, sets the softstart interval of the converter. Pulling this pin low (typically below 1.0V) with an open drain signal will shutdown the IC. VID0, VID1, VID2, VID3, VID4 (Pins 7, 6, 5, 4 and 3) VID0-4 are the input pins to the 5-bit DAC. The states of these five pins program the internal voltage reference (DACOUT). The level of DACOUT sets the core converter output voltage (VOUT1). It also sets the core PGOOD and OVP thresholds. COMP1, COMP2, and FB1, FB2 (Pins 20, 11, 21, and 10) COMP1, 2 and FB1, 2 are the available external pins of the PWM error amplifiers. Both the FB pins are the inverting input of the error amplifiers. Similarly, the COMP pins are the error amplifier outputs. These pins are used to compensate the voltage-control feedback loops of the PWM converters. GND (Pin 17) Signal ground for the IC. All voltage levels are measured with respect to this pin. PGOOD (Pin 8) PGOOD is an open collector output used to indicate the status of the PWM converter output voltages. This pin is pulled low when the core output is not within ±10% of the DACOUT reference voltage, or when any of the other outputs are below their under-voltage thresholds. The PGOOD output is open for ‘11111’ VID code. See Table 1. 272 FAULT/RT (Pin 13) 6 5 × 10 Fs ≈ 200 kHz + --------------------R T ( kΩ ) (RT to GND) Conversely, connecting a pull-up resistor (RT) from this pin to VCC reduces the switching frequency according to the following equation: 7 4 × 10 Fs ≈ 200 kHz – --------------------R T ( kΩ ) (RT to 12V) Nominally, this pin voltage is 1.26V, but is pulled to VCC in the event of an over-voltage or over-current condition. GATE3 (Pin 18) Connect this pin to the gate of an external MOSFET. This pin provides the drive for the linear controller’s pass transistor. FB3 (Pin 19) Connect this pin to a resistor divider to set the linear controller output. VOUT4 (Pin 16) Output of the linear regulator. Supplies current up to 230mA. FB4 (Pin 14) Connect this pin to a resistor divider to set the linear regulator output. HIP6019B Description Operation The HIP6019B monitors and precisely controls 4 output voltage levels (Refer to Figures 1, 2, and 3). It is designed for microprocessor computer applications with 5V power and 12V bias input from a PS2 or ATX power supply. The IC has 2 PWM controllers, a linear controller, and a linear regulator. The first PWM controller (PWM1) is designed to regulate the microprocessor core voltage (VOUT1). PWM1 controller drives 2 MOSFETs (Q1 and Q2) in a synchronous-rectified buck converter configuration and regulates the core voltage to a level programmed by the 5-bit digital-to-analog converter (DAC). The second PWM controller (PWM2) is designed to regulate the I/O voltage (VOUT2). PWM2 controller drives a MOSFET (Q3) in a standard buck converter configuration and regulates the I/O voltage to a resistor programmable level between 3.0 and 3.5VDC. An integrated linear regulator supplies the 2.5V clock generator power (VOUT4). The linear controller drives an external MOSFET (Q4) to supply the GTL bus power (VOUT3). increases, the pulse-width on the PHASE pin increases. The interval of increasing pulse-width continues until each output reaches sufficient voltage to transfer control to the input reference clamp. If we consider the 3.3V output (VOUT2) in Figure 6, this time occurs at T2. During the interval between T2 and T3, the error amplifier reference ramps to the final value and the converter regulates the output to a voltage proportional to the SS pin voltage. At T3 the input clamp voltage exceeds the reference voltage and the output voltage is in regulation. PGOOD (2V/DIV) 0V SOFT-START (1V/DIV) VOUT2 ( = 3.3V) 0V Initialization The HIP6019B automatically initializes upon receipt of input power. Special sequencing of the input supplies is not necessary. The Power-On Reset (POR) function continually monitors the input supply voltages. The POR monitors the bias voltage (+12VIN ) at the VCC pin and the 5V input voltage (+5VIN ) at the OCSET1 pin. The normal level on OCSET1 is equal to +5VIN less a fixed voltage drop (see over-current protection). The POR function initiates soft-start operation after both input supply voltages exceed their POR thresholds. VOUT4 ( = 2.5V) VOUT1 (DAC = 2V) OUTPUT VOLTAGES (0.5V/DIV) VOUT3 ( = 1.5V) 0V T0 T1 Soft-Start T2 T3 TIME The POR function initiates the soft-start sequence. Initially, the voltage on the SS pin rapidly increases to approximately 1V (this minimizes the soft-start interval). Then an internal 11µA current source charges an external capacitor (CSS) on the SS pin to 4V. The PWM error amplifier reference inputs (+ terminal) and outputs (COMP1 and COMP2 pins) are clamped to a level proportional to the SS pin voltage. As the SS pin voltage ramps from 1V to 4V, the output clamp allows generation of PHASE pulses of increasing width that charge the output capacitor(s). After this initial stage, the reference input clamp slows the output voltage rate-of-rise and provides a smooth transition to the final set voltage. Additionally, both linear regulator’s reference inputs are clamped to a voltage proportional to the SS pin voltage. This method provides a rapid and controlled output voltage rise. Figure 6 shows the soft-start sequence for the typical application. At T0 the SS voltage rapidly increases to approximately 1V. At T1, the SS pin and error amplifier output voltage reach the valley of the oscillator’s triangle wave. The oscillator’s triangular waveform is compared to the clamped error amplifier output voltage. As the SS pin voltage 273 FIGURE 6. SOFT-START INTERVAL The remaining outputs are also programmed to follow the SS pin voltage. Each linear output (VOUT3 and VOUT4) initially follows the 3.3V output (VOUT2). When each output reaches sufficient voltage the input reference clamp slows the rate of output voltage rise. The PGOOD signal toggles ‘high’ when all output voltage levels have exceeded their under-voltage levels. See the Soft-Start Interval section under Applications Guidelines for a procedure to determine the soft-start interval. Fault Protection All four outputs are monitored and protected against extreme overload. A sustained overload on any linear regulator output or an over-voltage on the PWM outputs disables all converters and drives the FAULT/RT pin to VCC. Figure 7 shows a simplified schematic of the fault logic. An over-voltage detected on either VSEN1 or VSEN2 immediately sets the fault latch. A sequence of three overcurrent fault signals also sets the fault latch. A comparator HIP6019B indicates when CSS is fully charged (UP signal), such that an under-voltage event on either linear output (FB3 or FB4) is ignored until after the soft-start interval (approximately T3 in Figure 6). At start-up, this allows VOUT3 and VOUT4 to slew up over increased time intervals, without generating a fault. Cycling the bias input voltage (+12VIN on the VCC pin) off then on resets the counter and the fault latch. LUV OVER CURRENT LATCH OC1 INHIBIT capacitor (CSS) with a 11µA current sink, and increments the counter. CSS recharges at T2 and initiates a soft-start cycle with the error amplifiers clamped by soft-start. With OUT2 still overloaded, the inductor current increases to trip the overcurrent comparator. Again, this inhibits all outputs, but the soft-start voltage continues increasing to 4V before discharging. The counter increments to 2. The soft-start cycle repeats at T3 and trips the over-current comparator. The SS pin voltage increases to 4V at T4 and the counter increments to 3. This sets the fault latch to disable the converter. The fault is reported on the FAULT/RT pin. S R 0.15V + - COUNTER R SS 4V + - FAULT LATCH VCC POR R FAULT OV FIGURE 7. FAULT LOGIC - SIMPLIFIED SCHEMATIC Over-Voltage Protection During operation, a short on the upper MOSFET (Q1) causes VOUT1 to increase. When the output exceeds the over-voltage threshold of 115% of DACOUT, the overvoltage comparator trips to set the fault latch and turns Q2 on as required in order to regulate VOUT1 to 1.15 x DACOUT. This blows the input fuse and reduces VOUT1. The fault latch raises the FAULT/RT pin close to VCC potential. A separate over-voltage circuit provides protection during the initial application of power. For voltages on the VCC pin below the power-on reset (and above ~4V), V OUT1 is monitored for voltages exceeding 1.26V. Should VSEN1 exceed this level, the lower MOSFET (Q2) is driven on, as needed to regulate V OUT1 to 1.26V. Over-Current Protection All outputs are protected against excessive over-currents. Both PWM controllers use the upper MOSFET’s on-resistance, rDS(ON) to monitor the current for protection against shorted outputs. The linear regulator monitors the current of the integrated power device and signals an overcurrent condition for currents in excess of 230mA. Additionally, both the linear regulator and the linear controller monitor FB3 and FB4 for under-voltage to protect against excessive currents. Figures 8 and 9 illustrate the over-current protection with an overload on OUT2. The overload is applied at T0 and the current increases through the output inductor (LOUT2). At time T1, the OVER-CURRENT2 comparator trips when the voltage across Q3 (ID • rDS(ON)) exceeds the level programmed by ROCSET. This inhibits all outputs, discharges the soft-start 274 FAULT REPORTED 10V 0V COUNT =1 S Q UP INDUCTOR CURRENT SOFT-START OC2 FAULT/RT S Q COUNT =2 COUNT =3 4V 2V 0V OVERLOAD APPLIED 0A T0 T1 T2 T3 T4 TIME FIGURE 8. OVER-CURRENT OPERATION The PWM1 controller and the linear regulator operate in the same way as PWM2 to over-current faults. Additionally, the linear regulator and linear controller monitor the feedback pins for an under-voltage. Should excessive currents cause FB3 or FB4 to fall below the linear under-voltage threshold, the LUV signal sets the over-current latch if CSS is fully charged. Blanking the LUV signal during the CSS charge interval allows the linear outputs to build above the under- HIP6019B voltage threshold during normal start-up. Cycling the bias input power off then on resets the counter and the fault latch. OVER-CURRENT TRIP: VDS > VSET VIN = +5V (I D • rDS(ON) > I OCSET • ROCSET ) OCSET IOCSET 200µA ROCSET VSET + ID Application Guidelines VCC DRIVE OC2 + UGATE + VDS PHASE - OVERCURRENT2 PWM VPHASE = VIN - VDS VOCSET = VIN - VSET GATE CONTROL HIP6019B FIGURE 9. OVER-CURRENT DETECTION Resistors (ROCSET1 and ROCSET2) program the overcurrent trip levels for each PWM converter. As shown in Figure 9, the internal 200µA current sink develops a voltage across ROCSET (VSET) that is referenced to VIN. The DRIVE signal enables the over-current comparator (OVERCURRENT1 or OVER-CURRENT2). When the voltage across the upper MOSFET (VDS) exceeds VSET, the overcurrent comparator trips to set the over-current latch. Both VSET and VDS are referenced to VIN and a small capacitor across ROCSET helps VOCSET track the variations of V IN due to MOSFET switching. The over-current function will trip at a peak inductor current (IPEAK) determined by: I PEAK VID pins can be left open for a logic 1 input, because they are internally pulled up to +5V by a 10µA current source. Changing the VID inputs during operation is not recommended. The sudden change in the resulting reference voltage could toggle the PGOOD signal and exercise the over-voltage protection. The ‘11111’ VID pin combination resulting in an INHIBIT disables the IC and the open-collector at the PGOOD pin. IO CSET × R OCSET = --------------------------------------------------------r DS ( ON ) Soft-Start Interval Initially, the soft-start function clamps the error amplifiers’ output of the PWM converters. After the output voltage increases to approximately 80% of the set value, the reference input of the error amplifier is clamped to a voltage proportional to the SS pin voltage. The resulting output voltage sequence is shown in Figure 6. The soft-start function controls the output voltage rate of rise to limit the current surge at start-up. The soft-start interval is programmed by the soft-start capacitor, CSS. Programming a faster soft-start interval increases the peak surge current. The peak surge current occurs during the initial output voltage rise to 80% of the set value. Shutdown Neither PWM output switches until the soft-start voltage (VSS) exceeds the oscillator’s valley voltage. Additionally, the reference on each linear’s amplifier is clamped to the soft-start voltage. Holding the SS pin low (with an open drain or collector signal) turns off all four regulators. The ‘11111’ VID code resulting in an INHIBIT as shown in Table 1 also shuts down the IC. TABLE 1. VID4 VID3 VID2 VID1 VID0 NOMINAL OUT1 VOLTAGE DACOUT 1. The maximum rDS(ON) at the highest junction temperature. 0 1 1 1 1 1.30 2. The minimum IOCSET from the specification table. 0 1 1 1 0 1.35 0 1 1 0 1 1.40 0 1 1 0 0 1.45 For an equation for the output inductor ripple current see the section under component guidelines titled ‘Output Inductor Selection’. 0 1 0 1 1 1.50 0 1 0 1 0 1.55 0 1 0 0 1 1.60 OUT1 Voltage Program 0 1 0 0 0 1.65 0 0 1 1 1 1.70 0 0 1 1 0 1.75 0 0 1 0 1 1.80 0 0 1 0 0 1.85 0 0 0 1 1 1.90 0 0 0 1 0 1.95 0 0 0 0 1 2.00 0 0 0 0 0 2.05 The OC trip point varies with MOSFET’s temperature. To avoid over-current tripping in the normal operating load range, determine the ROCSET resistor from the equation above with: 3. Determine IPEAK for IPEAK > IOUT(MAX) + (∆I)/2, where ∆I is the output inductor ripple current. The output voltage of the PWM1 converter is programmed to discrete levels between 1.3VDC and 3.5VDC . This output is designed to supply the microprocessor core voltage. The voltage identification (VID) pins program an internal voltage reference (DACOUT) through a TTL-compatible 5-bit digital-toanalog converter. The level of DACOUT also sets the PGOOD and OVP thresholds. Table 1 specifies the DACOUT voltage for the different combinations of connections on the VID pins. The 275 PIN NAME HIP6019B VID2 VID1 VID0 1 1 1 1 INHIBIT 1 1 1 1 0 2.1 1 1 1 0 1 2.2 1 1 1 0 0 2.3 1 1 0 1 1 2.4 1 1 0 1 0 2.5 1 1 0 0 1 2.6 1 1 0 0 0 2.7 1 0 1 1 1 2.8 1 0 1 1 0 2.9 1 0 1 0 1 3.0 1 0 1 0 0 3.1 1 0 0 1 1 3.2 1 0 0 1 0 3.3 1 0 0 0 1 3.4 1 0 0 0 0 3.5 NOTE: 0 = connected to GND or VSS, 1 = open or connected to 5V through pull-up resistors. Layout Considerations MOSFETs switch very fast and efficiently. The speed with which the current transitions from one device to another causes voltage spikes across the interconnecting impedances and parasitic circuit elements. The voltage spikes can degrade efficiency, radiate noise into the circuit, and lead to device over-voltage stress. Careful component layout and printed circuit design minimizes the voltage spikes in the converter. Consider, as an example, the turnoff transition of the upper MOSFET. Prior to turnoff, the upper MOSFET was carrying the full load current. During the turnoff, current stops flowing in the upper MOSFET and is picked up by the lower MOSFET or Schottky diode. Any inductance in the switched current path generates a large voltage spike during the switching interval. Careful component selection, tight layout of the critical components, and short, wide circuit traces minimize the magnitude of voltage spikes. Contact Intersil for evaluation board drawings of the component placement and printed circuit board. +5VIN CIN ROCSET2 Q3 LOUT2 VOUT2 +12V COCSET2 CVCC COCSET1 VCC GND OCSET2 OCSET1 R UGATE2 OCSET1 Q1 UGATE1 LOUT1 PHASE2 VOUT1 PHASE1 COUT2 Q4 VOUT3 HIP6019B LGATE1 GATE3 SS Q2 COUT1 LOAD VID3 1 capacitors between the MOSFETs and the load. Locate the PWM controller close to the MOSFETs. CR1 PGND CSS LOAD VID4 NOMINAL OUT1 VOLTAGE DACOUT LOAD TABLE 1. PIN NAME KEY ISLAND ON POWER PLANE LAYER ISLAND ON CIRCUIT PLANE LAYER VIA CONNECTION TO GROUND PLANE FIGURE 10. PRINTED CIRCUIT BOARD POWER PLANES AND ISLANDS The critical small signal components include the bypass capacitor for VCC and the soft-start capacitor, CSS. Locate these components close to their connecting pins on the control IC. Minimize any leakage current paths from SS node because the internal current source is only 11µA. A multi-layer printed circuit board is recommended. Figure 10 shows the connections of the critical components in the converter. Note that capacitors CIN and COUT could each represent numerous physical capacitors. Dedicate one solid layer for a ground plane and make all critical component ground connections with vias to this layer. Dedicate another solid layer as a power plane and break this plane into smaller islands of common voltage levels. The power plane should support the input power and output power nodes. Use copper filled polygons on the top and bottom circuit layers for the phase nodes. Use the remaining printed circuit layers for small signal wiring. The wiring traces from the control IC to the MOSFET gate and source should be sized to carry 1A currents. The traces for OUT4 need only be sized for 0.2A. Locate COUT4 close to the HIP6019B IC. PWM Controller Feedback Compensation There are two sets of critical components in a DC-DC converter using a HIP6019B controller. The power components are the most critical because they switch large amounts of energy. The critical small signal components connect to sensitive nodes or supply critical bypassing current. Both PWM controllers use voltage-mode control for output regulation. This section highlights the design consideration for a voltage-mode controller. Apply the methods and considerations to both PWM controllers. The power components should be placed first. Locate the input capacitors close to the power switches. Minimize the length of the connections between the input capacitors and the power switches. Locate the output inductor and output Figure 11 highlights the voltage-mode control loop for a synchronous-rectified buck converter. The output voltage is regulated to the reference voltage level. The reference voltage level is the DAC output voltage for PWM1 and is 1.265V for PWM2. The error amplifier output (V E /A) is 276 HIP6019B compared with the oscillator (OSC) triangular wave to provide a pulse-width modulated wave with an amplitude of VIN at the PHASE node. The PWM wave is smoothed by the output filter (L O and C O ). VIN OSC ∆ VOSC PWM COMP DRIVER VOUT ESR (PARASITIC) ZFB VE/A ERROR AMP ZIN + REFERENCE DETAILED FEEDBACK COMPENSATION ZFB C2 C1 VOUT ZIN C3 R2 R3 R1 COMP - FB + 5. Place 2ND Pole at half the switching frequency. Compensation Break Frequency Equations PHASE CO - 4. Place 1ST Pole at the ESR Zero. 7. Estimate Phase Margin - repeat if necessary. LO - 3. Place 2ND Zero at filter’s Double Pole. 6. Check Gain against Error Amplifier’s Open-Loop Gain. DRIVER + 2. Place 1ST Zero below filter’s Double Pole (~75% FLC). HIP6019B 1 F Z1 = ----------------------------------2π × R 2 × C1 1 F P1 = ------------------------------------------------------C1 × C2 2π × R 2 × ---------------------- C1 + C2 1 F Z2 = ------------------------------------------------------2π × ( R1 + R3 ) × C3 1 F P2 = ----------------------------------2π × R 3 × C3 Figure 12 shows an asymptotic plot of the DC-DC converter’s gain vs frequency. The actual modulator gain has a peak due to the high Q factor of the output filter at FLC, which is not shown in Figure 12. Using the above guidelines should yield a compensation gain similar to the curve plotted. The open loop error amplifier gain bounds the compensation gain. Check the compensation gain at FP2 with the capabilities of the error amplifier. The closed loop gain is constructed on the log-log graph of Figure 12 by adding the modulator gain (in dB) to the compensation gain (in dB). This is equivalent to multiplying the modulator transfer function to the compensation transfer function and plotting the gain. REFERENCE 100 FIGURE 11. VOLTAGE-MODE BUCK CONVERTER COMPENSATION DESIGN Modulator Break Frequency Equations 1 F LC = ---------------------------------------2π × L O × C O 1 FESR = ----------------------------------------2π × ESR × CO FP1 FP2 OPEN LOOP ERROR AMP GAIN 60 GAIN (dB) The modulator transfer function is the small-signal transfer function of VOUT/ VE/A. This function is dominated by a DC gain and the output filter, with a double pole break frequency at FLC and a zero at FESR. The DC gain of the modulator is simply the input voltage, VIN , divided by the peak-to-peak oscillator voltage, ∆VOSC . FZ1 FZ2 80 40 20 20LOG (R2/R 1) 0 20LOG (VIN /∆VOSC) MODULATOR GAIN -20 COMPENSATION GAIN CLOSED LOOP GAIN -40 FLC FESR -60 10 100 1K 10K 100K 1M 10M FREQUENCY (Hz) The compensation network consists of the error amplifier internal to the HIP6019B and the impedance networks ZIN and Z FB . The goal of the compensation network is to provide a closed loop transfer function with an acceptable 0dB crossing frequency (f0dB) and adequate phase margin. Phase margin is the difference between the closed loop phase at f0dB and 180 degrees. The equations below relate the compensation network’s poles, zeros and gain to the components (R1, R2 , R3 , C1, C2 , and C3) in Figure 11. Use these guidelines for locating the poles and zeros of the compensation network: 1. Pick Gain (R2/R1) for desired converter bandwidth. 277 FIGURE 12. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN The compensation gain uses external impedance networks ZFB and ZIN to provide a stable, high bandwidth loop. A stable control loop has a 0dB gain crossing with -20dB/decade slope and a phase margin greater than 45 degrees. Include worst case component variations when determining phase margin. Oscillator Synchronization The PWM controllers use a triangle wave for comparison with the error amplifier output to provide a pulse-width modulated HIP6019B Component Selection Guidelines Output Capacitor Selection The output capacitors for each output have unique requirements. In general the output capacitors should be selected to meet the dynamic regulation requirements. Additionally, the PWM converters require an output capacitor to filter the current ripple. The linear regulator is internally compensated and requires an output capacitor that meets the stability requirements. The load transient for the microprocessor core requires high quality capacitors to supply the high slew rate (di/dt) current demands. PWM Output Capacitors Modern microprocessors produce transient load rates above 10A/ns. High frequency capacitors initially supply the transient and slow the current load rate seen by the bulk capacitors. The bulk filter capacitor values are generally determined by the ESR (effective series resistance) and ESL (effective series inductance) parameters rather than actual capacitance. High frequency decoupling capacitors should be placed as close to the power pins of the load as physically possible. Be careful not to add inductance in the circuit board wiring that could cancel the usefulness of these low inductance components. Consult with the manufacturer of the load on specific decoupling requirements. Use only specialized low-ESR capacitors intended for switching regulator applications for the bulk capacitors. The bulk capacitor’s ESR determines the output ripple voltage and the initial voltage drop after a high slew-rate transient. An aluminum electrolytic capacitor’s ESR value is related to the case size with lower ESR available in larger case sizes. However, the equivalent series inductance of these capacitors increases with case size and can reduce the usefulness of the capacitor to high slew-rate transient loading. Unfortunately, ESL is not a specified parameter. Work with your capacitor supplier and measure the capacitor’s impedance with frequency to select suitable components. In most cases, multiple electrolytic capacitors of small case size perform better than a single large case capacitor. For a given transient load magnitude, the output voltage transient response due to the output capacitor characteristics can be approximated by the following equation: dITRAN V TRAN = ESL × --------------------- + ESR × ITR AN dt Linear Output Capacitors 278 The output capacitors for the linear regulator and the linear controller provide dynamic load current. The linear controller uses dominant pole compensation integrated in the error amplifier and is insensitive to output capacitor selection. Capacitor, COUT3 should be selected for transient load regulation. The output capacitor for the linear regulator provides loop stability. The linear regulator (OUT4) requires an output capacitor characteristic shown in Figure 13 The upper line plots the 45 phase margin with 150mA load and the lower line is the 45 phase margin limit with a 10mA load. Select a COUT4 capacitor with characteristic between the two limits. 0.7 0.6 0.5 ESR (Ω) wave. Should the output voltages of the two PWM converters be programmed close to each other, then cross-talk could cause nonuniform PHASE pulse-widths and increased output voltage ripple. The HIP6019B avoids this problem by synchronizing the two converters 180o out-of-phase for DAC settings above, and including 2.5V. This is accomplished by inverting the triangle wave sent to PWM 2. 0.4 LE N AB IO ST RAT E OP 0.3 0.2 0.1 10 100 CAPACITANCE (µF) 1000 FIGURE 13. COUT4 OUTPUT CAPACITOR Output Inductor Selection Each PWM converter requires an output inductor. The output inductor is selected to meet the output voltage ripple requirements and sets the converter’s response time to a load transient. The inductor value determines the converter’s ripple current and the ripple voltage is a function of the ripple current. The ripple voltage and current are approximated by the following equations: V IN – VOUT V OUT ∆I = -------------------------------- × ---------------VIN FS × LO ∆V OU T = ∆I × ESR Increasing the value of inductance reduces the ripple current and voltage. However, the large inductance values reduce the converter’s response time to a load transient. One of the parameters limiting the converter’s response to a load transient is the time required to change the inductor current. Given a sufficiently fast control loop design, the HIP6019B will provide either 0% or 100% duty cycle in response to a load transient. The response time is the time interval required to slew the inductor current from an initial current value to the post-transient current level. During this interval the difference between the inductor current and the transient current level must be supplied by the output capacitors. Minimizing the response time can minimize the output capacitance required. HIP6019B The response time to a transient is different for the application of load and the removal of load. The following equations give the approximate response time interval for application and removal of a transient load: L O × ITRAN t RISE = -------------------------------V IN – V OUT L O × I TRAN t FALL = ------------------------------V OUT where: ITRAN is the transient load current step, tRISE is the response time to the application of load, and tFALL is the response time to the removal of load. With a +5V input source, the worst case response time can be either at the application or removal of load and dependent upon the output voltage setting. Be sure to check both of these equations at the minimum and maximum output levels for the worst case response time. Input Capacitor Selection The important parameters for the bulk input capacitor are the voltage rating and the RMS current rating. For reliable operation, select the bulk capacitor with voltage and current ratings above the maximum input voltage and largest RMS current required by the circuit. The capacitor voltage rating should be at least 1.25 times greater than the maximum input voltage and a voltage rating of 1.5 times is a conservative guideline. Use a mix of input bypass capacitors to control the voltage overshoot across the MOSFETs. Use ceramic capacitance for the high frequency decoupling and bulk capacitors to supply the RMS current. Small ceramic capacitors should be placed very close to the upper MOSFET to suppress the voltage induced in the parasitic circuit impedances. For a through hole design, several electrolytic capacitors (Panasonic HFQ series or Nichicon PL series or Sanyo MVGX or equivalent) may be needed. For surface mount designs, solid tantalum capacitors can be used, but caution must be exercised with regard to the capacitor surge current rating. These capacitors must be capable of handling the surge-current at power-up. The TPS series available from AVX, and the 593D series from Sprague are both surge current tested. MOSFET Selection/Considerations The HIP6019B requires 4 N-Channel power MOSFETs. Two MOSFETs are used in the synchronous-rectified buck topology of PWM1 converter. PWM2 converter uses a MOSFET as the buck switch and the linear controller drives a MOSFET as a pass transistor. These should be selected based upon rDS(ON) , gate supply requirements, and thermal management requirements. losses are distributed between the upper and lower MOSFETs according to duty factor (see the equations below). The conduction losses are the only component of power dissipation for the lower MOSFETs. Only the upper MOSFET has switching losses, since the lower device turns on into near zero voltage. The equations below assume linear voltage-current transitions and do not model power loss due to the reverserecovery of the lower MOSFET’s body diode. The gatecharge losses are proportional to the switching frequency (FS) and are dissipated by the HIP6019B, thus not contributing to the MOSFETs’ temperature rise. However, large gate charge increases the switching interval, tSW which increases the upper MOSFET switching losses. Ensure that both MOSFETs are within their maximum junction temperature at high ambient temperature by calculating the temperature rise according to package thermal resistance specifications. A separate heatsink may be necessary depending upon MOSFET power, package type, ambient temperature and air flow. 2 IO × r DS ( ON ) × V OU T I O × VIN × t SW × F S P UPPER = ------------------------------------------------------------ + ---------------------------------------------------VIN 2 2 I O × r DS ( ON ) × ( V I N – V OU T ) P LOWER = --------------------------------------------------------------------------------VIN The rDS(ON) is different for the two previous equations even if the type device is used for both. This is because the gate drive applied to the upper MOSFET is different than the lower MOSFET. Figure 14 shows the gate drive where the upper gate-to-source voltage is approximately VCC less the input supply. For +5V main power and +12V DC for the bias, the gate-to-source voltage of Q1 is 7V. The lower gate drive voltage is +12VDC . A logic-level MOSFET is a good choice for Q1 and a logic-level MOSFET can be used for Q2 if its absolute gate-to-source voltage rating exceeds the maximum voltage applied to V CC . +5V OR LESS +12V VCC HIP6019B UGATE Q1 PHASE - LGATE NOTE: VGS ≈ VCC -5V Q2 CR1 + PGND NOTE: VGS ≈ VCC GND PWM1 MOSFET Selection and Considerations In high-current PWM applications, the MOSFET power dissipation, package selection and heatsink are the dominant design factors. The power dissipation includes two loss components; conduction loss and switching loss. These 279 FIGURE 14. OUTPUT GATE DRIVERS Rectifier CR1 is a clamp that catches the negative inductor swing during the dead time between the turn off of the HIP6019B lower MOSFET and the turn on of the upper MOSFET. The diode must be a Schottky type to prevent the lossy parasitic MOSFET body diode from conducting. It is acceptable to omit the diode and let the body diode of the lower MOSFET clamp the negative inductor swing, but efficiency might drop one or two percent as a result. The diode's rated reverse breakdown voltage must be greater than twice the maximum input voltage. PWM2 MOSFET and Schottky Selection The power dissipation in PWM2 converter power devices is similar to PWM1 except that the power losses of the lower device are representative of a Schottky diode instead of a MOSFET. The transistor power losses follow the PWM1 upper MOSFET equation, so the selection process should be somewhat similar. The equation below describes the conduction power losses incurred by the Schottky diode. IO × Vf × ( VIN – VOUT ) P SC H = ------------------------------------------------------------VIN As it can be observed, conduction losses in the Schottky diode are proportional with the forward voltage drop (Vf). 280 Linear Controller MOSFET Selection The main criteria for selection of MOSFET for the linear regulator is package selection for efficient removal of heat. The power dissipated in a linear regulator is: PLINEAR = I O × ( VIN – VOUT ) Select a package and heatsink that maintains the junction temperature below the maximum rating while operating at the highest expected ambient temperature. HIP6019B DC-DC Converter Application Circuit Figure 15 shows an application circuit of a power supply for a microprocessor computer system. The power supply provides the microprocessor core voltage (V OUT1), the I/O voltage (V OUT2), the GTL bus voltage (VOUT3) and clock generator voltage (VOUT4) from +5VDC and +12VDC. For detailed information on the circuit, including a Bill-ofMaterials and circuit board description, see Application Note AN9800. Also see Intersil’s web page (http://www.intersil.com). HIP6019B +12VIN L1 F1 +5VIN 1µH 30A C1-4 + 4x1000µF GND C14-15 2x1µF C16 1µF C18 C17 VCC 1000pF R1 28 1.21K OCSET2 1000pF R2 23 1.21K OCSET1 9 8 Q3 HUF76137S3S VOUT2 (3.3V) UGATE2 L2 PHASE2 1 27 2 26 POWERGOOD PGOOD Q1 HUF76139S3S UGATE1 PHASE1 5.2µH + 2.9µH C19-23 5x1000µF R3 4.99K R5 FB2 3.32K 0.68µF R21 C38 LGATE1 25 CR2 MBR2535CTL VSEN2 C37 15 22 10 21 HIP6019B COMP2 C24-36 + 7x1000µF Q2 HUF76139S3S PGND 24 R4 4.99K VSEN1 C40 R8 FB1 2.21K 0.68µF 11 10pF COMP1 20 R7 R6 Q4 HUF75307D3S VOUT3 (GTL = 1.5V) C39 220K 5.11K R11 VOUT4 (2.5V) FB3 7 18 6 5 19 3 VOUT4 10K + 4 R12 10K R13 C47 270µF C42 R10 0.01µF 150K R9 0.1µF GATE3 C43-46 4x1000µF C41 10pF 1.87K + VOUT1 (1.3 TO 3.5V) L3 FB4 R14 10K 16 12 VID0 VID1 VID2 VID3 VID4 VID1 VID2 VID3 VID4 SS 14 17 13 732K VID0 C48 0.039µF GND FAULT/RT FIGURE 15. 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