MPS MPQ4560DN Industrial-grade, 2a, 2mhz, 55v step-down converter available in aec-q100 Datasheet

MPQ4560
Industrial-Grade, 2A, 2MHz, 55V
Step-Down Converter
Available in AEC-Q100
DESCRIPTION
The MPQ4560 is a high-frequency, step-down,
switching regulator with an integrated, highside, high-voltage, power MOSFET. It provides
a 2A output with current mode control for fast
loop response and easy compensation.
FEATURES




The
wide
3.8V-to-55V
input
range
accommodates a variety of step-down
applications, including those in automotive input
environment. A 12µA shutdown mode supply
current allows use in battery-powered
applications.






High-power conversion efficiency over a wide
load range is achieved by scaling down the
switching frequency in light load conditions to
reduce the switching and gate driving losses.

Frequency foldback prevents inductor current
runaway during startup and thermal shutdown
provides reliable, fault tolerant operation.
By switching at 2MHz, the MPQ4560 can
prevent electromagnetic interference problems,
such as those found in AM radio and ADSL
applications.
The MPQ4560 is available in small 3mm x 3mm
QFN10 and SOIC8E packages.
Guaranteed Industrial Automotive
Temperature Range Limits
Wide 3.8V-to-55V Operating Input Range
250mΩ Internal Power MOSFET
Up to 2MHz Programmable Switching
Frequency
140μA Quiescent Current
Ceramic Capacitor Stable
Internal Soft-Start
Up to 95% Efficiency
Output Adjustable from 0.8V to 52V
Available in QFN10 (3mmx3mm) and
SOIC8E Packages
AEC-Q100 Qualified
APPLICATIONS





High-Voltage Power Conversion
Automotive Systems
Industrial Power Systems
Distributed Power Systems
Battery Powered Systems
All MPS parts are lead-free and adhere to the RoHS directive. For MPS green
status, please visit MPS website under Products, Quality Assurance page.
“MPS” and “The Future of Analog IC Technology” are Registered Trademarks of
Monolithic Power Systems, Inc.
TYPICAL APPLICATION
MPQ4560 Rev. 1.1
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MPQ4560 – 2A, 2MHz, 55V, STEP-DOWN CONVERTER
ORDERING INFORMATION
Part Number
Package
Top Marking
MPQ4560DN*
SOIC8E
MP4560DN
MPQ4560DQ**
QFN10 (3×3mm)
T8
MPQ4560DN-AEC1
SOIC8E
MP4560DN
MPQ4560DQ-AEC1
QFN10 (3×3mm)
T8
Junction Temperature (TJ)
–40°C to +125°C
* For Tape & Reel, add suffix –Z (e.g. MPQ4560DN-Z)
For RoHS Compliant Packaging, add suffix –LF, (e.g. MPQ4560DN-LF–Z)
** For Tape & Reel, add suffix –Z (e.g. MPQ4560DQ-Z)
For RoHS Compliant Packaging, add suffix –LF, (e.g. MPQ4560DQ-LF–Z)
PACKAGE REFERENCE
QFN10 (3x3mm)
SOIC8E
ABSOLUTE MAXIMUM RATINGS (1)
Thermal Resistance
Supply Voltage (VIN).................... –0.3V to +60V
Switch Voltage (VSW)......... –0.5V to (VIN + 0.5V)
BST to SW .................................... –0.3V to +5V
All Other Pins ................................ –0.3V to +5V
Continuous Power Dissipation .......(TJ = 25°C)(2)
QFN10 (3×3mm) ........................................2.5W
SOIC8E .....................................................2.5W
Junction Temperature .............................. 150°C
Lead Temperature ................................... 260°C
Storage Temperature .............. –65°C to +150°C
QFN10 (3x3mm) ..................... 50 ...... 12 ... °C/W
SOIC8E .................................. 50 ...... 10 ... °C/W
Recommended Operating Conditions
(3)
Supply Voltage VIN .......................... 3.8V to 55V
Output Voltage VOUT........................ 0.8V to 52V
Maximum Junction Temp. (TJ) .............. +125°C
(4)
θJA
θJC
Notes:
1) Exceeding these ratings may damage the device
2) The maximum allowable power dissipation is a function of the
maximum junction temperature TJ(MAX), the junction-toambient thermal resistance θJA, and the ambient temperature
TA. The maximum allowable continuous power dissipation at
any ambient temperature is calculated by PD(MAX)=(TJ(MAX)TA)/ θJA. Exceeding the maximum allowable power dissipation
will cause excessive die temperature, and the regulator will go
into thermal shutdown. Internal thermal shutdown circuitry
protects the device from permanent damage.
3) The device is not guaranteed to function outside of its
operating conditions.
4) Measured on JESD51-7 4-layer board.
MPQ4560 Rev. 1.1
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MPQ4560 – 2A, 2MHz, 55V, STEP-DOWN CONVERTER
ELECTRICAL CHARACTERISTICS
VIN = 12V, VEN = 2.5V, VCOMP = 1.4V, TJ= –40°C to +125°C, unless otherwise noted. Typical Values
are at TJ=25°C.
Parameter
Symbol
Feedback Voltage
VFB
Feedback Leakage Current
Upper Switch On Resistance
4.5V < VIN <
55V
Min
Typ
Max
TJ=25°C
0.780
0.797
0.820
−40°C ≤ TJ ≤85°C
0.772
0.829
−40°C ≤ TJ ≤125°C
0.766
0.829
IFB
(5)
RDS(ON)
Upper Switch Leakage
ISW
Current Limit
ILIM
COMP to Current Sense
(5)
Transconductance
GCS
Error Amp Voltage Gain
Condition
VBST – VSW = TJ=25°C
5V
175
Duty Cycle ≤ 60%
1.0
250
330
160
VEN = 0V, VSW = 0V
TJ=25°C
0.1
400
3.2
2.2
(6)
V
μA
mΩ
μA
1
2.6
Units
4.5
4.7
A
5.7
A/V
400
V/V
Error Amp Transconductance
ICOMP = ±3µA
120
µA/V
Error Amp Min Source current
VFB = 0.7V
10
µA
Error Amp Min Sink current
VFB = 0.9V
−10
µA
TJ=25°C
VIN UVLO Threshold
2.7
2.4
VIN UVLO Hysteresis
Soft-Start Time
(5)
0V < VFB < 0.8V
Oscillator Frequency
3.0
fSW
RFREQ =
95kΩ
TJ=25°C
3.3
V
3.6
0.35
V
0.19
0.5
ms
0.8
1
0.7
1.2
MHz
1.3
Shutdown Supply Current
IS
VEN < 0.3V
12
20
µA
Quiescent Supply Current
IQ
No load, VFB = 0.9V (no switching)
140
200
µA
Hysteresis = 20°C
150
°C
Thermal Shutdown
(5)
Minimum Off Time
(5)
tOFF
100
ns
Minimum On Time
(5)
tON
100
ns
EN Rising Threshold
TJ=25°C
EN Threshold Hysteresis
1.4
1.55
1.3
1.7
V
1.8
320
mV
Note:
5) Derived from bench characterization. Not tested in production.
6) Guaranteed by design. Not tested in production.
MPQ4560 Rev. 1.1
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MPQ4560 – 2A, 2MHz, 55V, STEP-DOWN CONVERTER
PIN FUNCTIONS
QFN
Pin #
SOIC8
Pin #
1, 2
1
3
2
4
3
5
4
6
5
7
6
8, 9
7
10
8
Name
Description
Switch Node. Output from the high-side switch. A low VF Schottky rectifier to ground
is required. The rectifier must be close to the SW pins to reduce switching spikes.
Enable Input. Pull this pin below the specified threshold to shutdown the chip. Pull it
EN
up above the specified threshold or leaving it floating to enable the chip.
Compensation. Output of the GM error amplifier. Control loop frequency
COMP
compensation is applied to this pin.
Feedback. Input to the error amplifier. Sets the regulator voltage by comparing the
FB
tap of an external resistive divider connected between the output and GND to the
internal +0.8V reference.
GND, Ground. Connect as close as possible to the output capacitor and avoid the highExposed current switch paths. Connect exposed pad to GND plane for optimal thermal
pad
performance.
Switching Frequency Program Input. Connect a resistor from this pin to ground to set
FREQ
the switching frequency.
Input Supply. This supplies power to all the internal control circuitry, both BS
VIN
regulators, and the high-side switch. Place a decoupling capacitor to ground close to
this pin to minimize switching spikes.
SW
BST
Bootstrap. Positive power supply for the internal floating high-side MOSFET driver.
Connect a bypass capacitor between this pin and SW pin.
MPQ4560 Rev. 1.1
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MPQ4560 – 2A, 2MHz, 55V, STEP-DOWN CONVERTER
TYPICAL CHARACTERISTICS
MPQ4560 Rev. 1.1
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MPQ4560 – 2A, 2MHz, 55V, STEP-DOWN CONVERTER
TYPICAL CHARACTERISTICS (continued)
MPQ4560 Rev. 1.1
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MPQ4560 – 2A, 2MHz, 55V, STEP-DOWN CONVERTER
TYPICAL PERFORMANCE CHARACTERISTICS
VIN = 12V, VOUT =3.3V, C1 = 4.7µF, C2 = 22µF, L1 = 10µH and TA = 25°C, unless otherwise noted.
MPQ4560 Rev. 1.1
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MPQ4560 – 2A, 2MHz, 55V, STEP-DOWN CONVERTER
TYPICAL PERFORMANCE CHARACTERISTICS (continued)
VIN = 12V, VOUT =3.3V, C1 = 4.7µF, C2 = 22µF, L1 = 10µH and TA = 25°C, unless otherwise noted.
MPQ4560 Rev. 1.1
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MPQ4560 – 2A, 2MHz, 55V, STEP-DOWN CONVERTER
BLOCK DIAGRAM
Figure 1: Functional Block Diagram
MPQ4560 Rev. 1.1
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MPQ4560 – 2A, 2MHz, 55V, STEP-DOWN CONVERTER
OPERATION
The MPQ4560 is an asynchronous, step-down,
switching regulator with an integrated high-side,
high-voltage,
power
MOSFET
and
a
programmable frequency. It provides a single
highly-efficient solution with current-mode control
for fast loop response and easy compensation. It
features a wide input voltage range, internal softstart control, and precise current limiting. Its very
low operational quiescent current makes it
suitable for battery-powered applications.
PWM Control
The MPQ4560 operates in a fixed-frequency,
peak-current-control mode to regulate the output
voltage at moderate-to-high output current. The
internal clock initiates a PWM cycle. The power
MOSFET turns ON and remains ON until its
current reaches the value set by the COMP
voltage. When the power switch is OFF, it
remains OFF for at least 100ns before the next
cycle starts. If the current in the power MOSFET
does not reach the COMP-set current value
within one PWM period, the power MOSFET
remains ON, saving a turn-off operation.
Enable Control
The MPQ4560 has a dedicated enable control
pin (EN) that can enable or disable the chip when
the input voltage exceeds an upper threshold. Its
falling threshold (turn-off) is 1.2V, and its rising
threshold (turn-on) is 1.5V (300mV higher).
When floating, an internal 1µA current source
pulls EN up to ~3.0V to enable the chip. Pulldown requires a 1µA current.
When EN is pulled below 1.2V, the chip enters its
lowest shutdown current mode. When EN
exceeds 0V but remains lower than its rising
threshold, the chip remains in shutdown mode
but the shutdown current increases slightly.
Under-Voltage Lockout
Under-voltage lockout (UVLO) protects the chip
from operating at insufficient supply voltage. The
UVLO rising threshold is about 3.0V while its
falling threshold is a consistent 2.6V.
Pulse-Skipping Mode
Under light-load condition the switching
frequency stretches the zero-voltage period to
reduce the switching loss and driving loss.
Internal Soft-Start
Soft-start prevents the converter output voltage
from overshooting during startup and short-circuit
recovery. When the chip starts, the internal circuit
generates a soft-start voltage (SS) ramping up
from 0V to 2.6V. When it is less than the VREF,
SS overrides VREF so the error amplifier uses SS
as the reference. When SS exceeds VREF, VREF
regains control.
Error Amplifier
The error amplifier compares the FB pin voltage
(VFB) to the internal reference (VREF) and outputs
a current proportional to the difference. This
output
current
charges
the
external
compensation network to form VCOMP, which
controls the power MOSFET current.
Thermal Shutdown
Thermal shutdown prevents the chip from
operating at exceedingly high temperatures.
When the silicon die temperature exceeds its
upper threshold, the whole chip shuts down.
When the temperature is less than its lower
threshold, the chip is enabled again.
During operation, the minimum VCOMP is clamped
to 0.9V and its maximum is clamped to 2.0V.
COMP is internally pulled down to GND in
shutdown mode. Do not pull VCOMP above 2.6V.
Floating Driver and Bootstrap Charging
An external bootstrap capacitor powers the
floating power MOSFET driver. This floating
driver has its own UVLO protection. This UVLO’s
rising threshold is 2.2V with a hysteresis of
150mV. The driver’s UVLO is soft-start related:
When the bootstrap voltage hits its UVLO
threshold, the soft-start circuit resets. To prevent
noise, there is 20µs delay before the reset action.
When bootstrap UVLO is gone, the reset is off
and then the soft-start process resumes.
Internal Regulator
An internal 2.6V regulator powers most of the
internal circuits. This regulator takes the VIN
input and operates in the full VIN range. When VIN
exceeds 3.0V, the output of the regulator is in full
regulation. When VIN is less than 3.0V, the output
decreases.
The dedicated internal bootstrap regulator
regulates and charges the bootstrap capacitor to
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MPQ4560 – 2A, 2MHz, 55V STEP-DOWN CONVERTER
~5V. When the voltage between the BST and SW
nodes is less than its regulation, a PMOS pass
transistor from VIN to BST turns ON. The
charging current path is from VIN, BST and then
to SW. An external circuit must provide enough
voltage headroom to facilitate charging.
During a short circuit, the VFB voltage is low and
pulls down VSS to ~100mV above VFB. Removing
the short circuit causes the output voltage to
recover with VSS. When VFB is high enough, the
frequency and current limit return to normal
values.
As long as VIN is sufficiently higher than VSW, the
bootstrap capacitor can charge. When the power
MOSFET is ON, VIN≈VSW so the bootstrap
capacitor cannot charge. When the external
diode is ON, the difference between VIN and VSW
is at its largest, thus making it the best period to
charge. When there is no current in the inductor,
VSW=VOUT so the difference between VIN and VOUT
can charge the bootstrap capacitor.
Startup and Shutdown
If both VIN and VEN exceed their respective
thresholds, the chip starts. The reference block
initiates to generate a stable reference voltage
and currents, and then the internal regulator is
enabled. The regulator provides a stable supply
for the remaining circuitries.
At higher duty cycles, the time period available
for bootstrap charging is shorter so the bootstrap
capacitor may not sufficiently charge. If the
internal circuit does not have sufficient voltage
and the bootstrap capacitor is not charged, extra
external circuitry can ensure the bootstrap
voltage is within the normal operational region.
The DC quiescent current of the floating driver is
about 20µA. Make sure the bleeding current at
the SW node exceeds this value, such that:
VO
IO 
 20A
(R1  R2)
Current Comparator and Current Limit
A current-sense MOSFET accurately senses the
power MOSFET’s current. The result goes to the
high-speed current comparator for current-mode
control.: When the power MOSFET turns ON, the
comparator is first blanked till the end of the turnon transition to avoid noise issues. The
comparator then compares the power switch
current to VCOMP. When the sensed current
exceeds VCOMP, the comparator output is LOW,
turning OFF the power MOSFET. The
cycle-by-cycle maximum current of the internal
power MOSFET is internally limited.
While the internal supply rail is up, an internal
timer holds the power MOSFET OFF for about
50µs to blank the startup noise. When the
internal soft-start block is enabled, it first holds its
SS output low to ensure the remaining circuitries
are ready and then slowly ramps up.
Three events can shut down the chip: VEN LOW,
VIN LOW and thermal shutdown. During
shutdown, the power MOSFET turns OFF first to
avoid any fault triggering. Then VCOMP and the
internal supply rail drop.
Programmable Oscillator
An external resistor (RFREQ) from the FREQ pin to
ground sets the MPQ4560 oscillating frequency.
The value of RFREQ can be calculated from:
RFREQ (kΩ) =
100000
-5
fS (kHz)
For example, for fSW=500kHz, RFREQ=195kΩ.
Short Circuit Protection
When the output is shorted to the ground, the
switching frequency folds back and the current
limit falls to lower the short-circuit current. When
VFB is zero, the current limit drops to about 50%
of its full current limit. When VFB exceeds 0.4V,
current limit reaches 100%.
MPQ4560 Rev. 1.1
3/29/2013
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MPQ4560 – 2A, 2MHz, 55V, STEP-DOWN CONVERTER
APPLICATION INFORMATION
COMPONENT SELECTION
Setting the Output Voltage
A resistive voltage divider from the output voltage
to FB pin sets the output voltage. The voltage
divider divides the output voltage down to the
feedback voltage by the ratio:
VFB =VOUT 
R2
R1+R2
Thus the output voltage is:
VOUT =VFB 
R1+R2
R2
For example, the value for R2 can be 10kΩ. With
this value, R1 is:
R1=12.5  (VOUT -0.8)(KΩ)
So for a 3.3V output voltage, R2 is 10kΩ, and R1
is 31.6kΩ.
Inductor
The inductor provides constant current to the
output load while being driven by the switched
input voltage. A larger-value inductor will result in
lower ripple current that will lower the output
ripple voltage. However, a larger inductor value
will be physically larger, have higher series
resistance, or lower saturation current.
To determine the inductance, allow the inductor’s
peak-to-peak ripple current to approximately
equal 30% of the maximum switch current limit.
Make sure that the peak inductor current is less
than the maximum switch current limit. The
inductance value can be calculated by:
L1=
VOUT
fs  ΔIL
 (1-
VOUT
VIN
)
Where VOUT is the output voltage, VIN is the input
voltage, fS is the switching frequency, and ∆IL is
the peak-to-peak inductor ripple current.
Choose an inductor that will not saturate under
the maximum inductor peak current. The peak
inductor current can be calculated by:
ILP  ILOAD 

VOUT
V
 1  OUT
2  fS  L1 
VIN



Where ILOAD is the load current.
Table 1 lists several suitable inductors from
various manufacturers. The different inductor
choices include price vs. size requirements and
any EMI requirements.
MPQ4560 Rev. 1.1
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MPQ4560 – 2A, 2MHz, 55V, STEP-DOWN CONVERTER
Table 1: Inductor Selection Guide
Inductance
Max DCR
Current Rating
Dimensions
(µH)
(Ω)
(A)
L × W × H (mm3)
7447789004
4.7
0.033
2.9
7.3×7.3×3.2
744066100
10
0.035
3.6
10×10×3.8
744771115
15
0.025
3.75
12×12×6
744771122
22
0.031
3.37
12×12×6
RLF7030T-4R7
4.7
0.031
3.4
7.3×6.8×3.2
SLF10145T-100
10
0.0364
3
10.1×10.1×4.5
SLF12565T-150M4R2
15
0.0237
4.2
12.5×12.5×6.5
SLF12565T-220M3R5
22
0.0316
3.5
12.5×12.5×6.5
FDV0630-4R7M
4.7
0.049
3.3
7.7×7×3
919AS-100M
10
0.0265
4.3
10.3×10.3×4.5
919AS-160M
16
0.0492
3.3
10.3×10.3×4.5
919AS-220M
22
0.0776
3
10.3×10.3×4.5
Part Number
Wurth Electronics
TDK
Toko
Output Rectifier Diode
The output rectifier diode supplies the current to
the inductor when the high-side switch is OFF.
Use a Schottky diode to reduce losses from the
diode forward voltage and recovery times.
Choose a diode whose maximum reverse voltage
rating exceeds the maximum input voltage, and
whose current rating exceeds the maximum load
current. Table 2 lists example Schottky diodes
and manufacturers.
Table 2: Diode Selection Guide
Diodes
Voltage/
Current
Rating
Manufacturer
B290-13-F
90V, 2A
Diodes Inc.
B380-13-F
80V, 3A
Diodes Inc.
CMSH2-100M
100V, 2A
Central Semi
CMSH3-100MA
100V, 3A
Central Semi
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MPQ4560 – 2A, 2MHz, 55V, STEP-DOWN CONVERTER
Input Capacitor
The input current to the step-down converter is
discontinuous and requires a capacitor to supply
the AC current to the step-down converter while
maintaining the DC input voltage. Use capacitors
with low equivalent series resistances (ESR) for
the best performance. Ceramic capacitors are
best, but tantalum or low-ESR electrolytic
capacitors may also suffice.
For simplification, choose the input capacitor with
an RMS current rating greater than half of the
maximum load current. The input capacitor (C1)
can be electrolytic, tantalum, or ceramic.
When using electrolytic or tantalum capacitors,
place a small, high-quality, ceramic capacitor
(0.1μF) as close to the IC as possible. When
using ceramic capacitors, make sure that they
have enough capacitance to provide sufficient
charge to prevent excessive voltage ripple at the
input. The input voltage ripple caused by
capacitance is approximately:
VIN 

ILOAD
V
V
 OUT  1  OUT
fS  C1 VIN 
VIN



Output Capacitor
The output capacitor (C2) maintains the DC
output voltage. Use ceramic, tantalum, or lowESR electrolytic capacitors. Low-ESR capacitors
are preferred to keep the output voltage ripple
low. The output voltage ripple can be estimated
as:
VOUT 
VOUT 
V
 1  OUT
fS  L 
VIN

 
1

   R ESR 

8  f S  C2 
 
Where L is the inductor value and RESR is the
ESR value of the output capacitor.
For ceramic capacitors, the capacitance
dominates the impedance at the switching
frequency and contributes the most to the output
voltage ripple. For simplification, the output
voltage ripple can be estimated by:
ΔVOUT 

V
 1  OUT
VIN
 L  C2 
VOUT
8  fS
2



For tantalum or electrolytic capacitors, the ESR
dominates the impedance at the switching
frequency. For simplification, the output ripple is
approximately:
ΔVOUT 
VOUT 
V
 1  OUT
fS  L 
VIN

  R ESR

The characteristics of the output capacitor also
affect the stability of the regulation system. The
MPQ4560 can be optimized for a wide range of
capacitances and ESR values.
Compensation Components
MPQ4560 employs current-mode control for easy
compensation and fast transient response. The
COMP pin controls the system stability and
transient response. The COMP pin is the output
of the internal error amplifier. A series capacitorresistor
combination
sets
a
pole-zero
combination to control the control system’s
characteristics. The DC gain of the voltage
feedback loop is:
A VDC  R LOAD  GCS  A VEA 
VFB
VOUT
Where

AVEA is the error-amplifier voltage gain,
400V/V;

GCS is the current-sense transconductance,
5.6A/V; and

RLOAD is the load resistor value.
The system has two important poles: One from
the compensation capacitor (C3) and the output
resistor of error amplifier, and the other due to
the output capacitor and the load resistor. These
poles are located at:
fP1 
GEA
2 π C3  A VEA
fP 2 
1
2 π C2  RLOAD
Where,
GEA
is
the
transconductance, 120μA/V.
error-amplifier
The system has one important zero due to the
compensation capacitor and the compensation
resistor (R3). This zero is located at:
fZ1 
1
2 π C3  R3
MPQ4560 Rev. 1.1
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MPQ4560 – 2A, 2MHz, 55V STEP-DOWN CONVERTER
The system may have another significant zero if
the output capacitor has a large capacitance or a
high ESR value. This zero is located at:
fESR
C3 
1

2π  C2  RESR
In this case, a third pole set by the compensation
capacitor (C5) and the compensation resistor can
compensate for the effect of the ESR zero. This
pole is located at:
fP 3 
1
2 π C5  R3
The goal of compensation design is to shape the
converter transfer function for a desired loop
gain. The system crossover frequency where the
feedback loop has unity gain is important: Lower
crossover frequencies result in slower line and
load transient responses, while higher crossover
frequencies lead to system instability. Generally,
set the crossover frequency to ~0.1×fSW.
Table 3: Compensation Values for Typical
Output Voltage/Capacitor Combinations
VOUT
(V)
L (µH)
C2
(µF)
R3
(kΩ)
C3
(pF)
C6
(pF)
1.8
4.7
33
32.4
680
None
2.5
4.7 - 6.8
22
26.1
680
None
3.3
6.8 -10
22
68.1
220
None
5
15 - 22
33
47.5
330
None
12
10
22
16
470
2
To optimize the compensation components for
conditions not listed in Table 3, follow these
steps:
1. Choose R3 to set the desired crossover
frequency:
2 π C2  fC VOUT
R3 

GEA  GCS
VFB
Where fC is the desired crossover frequency.
2. Choose C3 to achieve the desired phase
margin. For applications with typical inductor
MPQ4560 Rev. 1.1
3/29/2013
values, set the compensation zero (fZ1) <0.25 ×fC
to provide sufficient phase margin. C3 is then:
4
2 π R3  fC
3. C5 is required if the ESR zero of the output
capacitor is located at <0.5 ×fSW , or the following
relationship is valid:
f
1
 S
2π  C2  RESR 2
If this is the case, use C5 to set the pole (fP3) at
the location of the ESR zero. Determine the C5:
C5 
C2  RESR
R3
High-Frequency Operation
The switching frequency of MPQ4560 can be
programmed up to 2MHz by an external resistor.
The minimum on time of MPQ4560 is about
100ns (typ). Pulse-skipping occurs more readily
at higher switching frequencies due to the
minimum ON time.
Since the internal bootstrap circuitry has higher
impedance, which may not sufficiently charge the
bootstrap capacitor during each (1−D)×τS
charging period, add an external bootstrap
charging diode if the switching frequency is about
2MHz (see External Bootstrap Diode section for
detailed implementation information).
With higher switching frequencies, the capacitors’
inductive reactances (XL) dominate so that the
ESL of input/output capacitors determine the
input/output ripple voltages at higher switching
frequencies. As a result, use high-frequency
ceramic capacitors as input decoupling
capacitors and output filtering capacitors for highfrequency operation.
External Bootstrap Diode
An external bootstrap diode from the 5V rail to
the BST pin may enhance the efficiency under
the following conditions:

There is a 5V rail available in the system;

VIN ≤5V;

3.3V<VOUT<5V; and
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MPQ4560 – 2A, 2MHz, 55V STEP-DOWN CONVERTER

for high duty-cycle operation (when VOUT/VIN >
65%).
The bootstrap diode can be a low cost one such
as IN4148 or BAT54.
Figure 2: External Bootstrap Diode
At no-load or light-load, the converter may
operate in pulse-skipping mode in order to
maintain output-voltage regulation. Thus there is
less time to refresh the BS voltage. For sufficient
gate voltage during pulse-skipping, VIN–VOUT>3V.
For example, if the VOUT=3.3V, VIN must be
exceed 3.3V+3V=6.3V to maintain sufficient BST
voltage at no-load or light-load. To meet this
requirement, the EN pin can program the input
UVLO voltage to VOUT+3V.
MPQ4560 Rev. 1.1
3/29/2013
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MPQ4560 – 2A, 2MHz, 55V, STEP-DOWN CONVERTER
TYPICAL APPLICATION CIRCUITS
Figure 3: Typical Application, 1.8V Output
Figure 4: Typical Application, 5V Output
MPQ4560 Rev. 1.1
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MPQ4560 – 2A, 2MHz, 55V, STEP-DOWN CONVERTER
PCB LAYOUT GUIDE
3)
PCB layout is very important for stable
operation. Try to duplicate the EVB layout for
optimum performance.
Route SW away from sensitive analog
areas such as FB.
4)
Connect IN, SW, and especially GND to
large copper surfaces to cool the chip to
improve thermal performance and longterm reliability.
5)
Place the compensation components close
to the MPQ4560. Avoid placing the
compensation components close to or
under high dv/dt SW node, or inside the
high di/dt power loop. If necessary, add a
ground plane to isolate the loops.
6)
Switching loss increases at higher
frequencies.
To
improve
thermal
conduction, add a grid of thermal vias
under the exposed pad. Use small vias
(15mil barrel diameter) so that the hole fills
during the plating process: larger vias can
cause solder-wicking during the reflow
process. The pitch (distance between the
centers) between these thermal vias is
typically 40mil.
For changes, please follow these guidelines
and use Figure 5 as reference.
1) Place the input decoupling capacitor and
the catch diode as close to the MPQ4560
(VIN pin, SW pin and PGND) as possible,
with traces that are very short and fairly
wide. This can help to greatly reduce the
voltage spike on SW node, and the EMI
noise.
2)
Ensure all feedback connections are short
and direct. Place the feedback resistors
and compensation components as close to
the chip as possible. Try to run the
feedback trace as far from the inductor and
noisy power traces as possible. Run the
feedback trace on the side of the PCB
opposite of the inductor with a ground
plane separating the two.
MPQ4560 Typical Application Circuit
MPQ4560 Rev. 1.1
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MPQ4560 – 2A, 2MHz, 55V, STEP-DOWN CONVERTER
GND
R1
R5
R4
C3
R2
R3
L1
2
FB
COMP FREQ
EN
SW
VIN
BST
1
3
GND
4
SW
C4
D1
8
7
6
5
R6
C2
C1
Vin
GND
GND
Vo
TOP Layer
Bottom Layer
MPQ4560DN Layout Guide
GND
R5
R4
C3
3
SW
1
4
SW
EN
5
SW 2
FB
COMP
R1
R2
R3
L1
C4
8
9
Vin
Vin
10 BST
7 FREQ
6 GND
D1
R6
C2
C1
Vin
GND
GND
Vo
TOP Layer
Bottom Layer
MPQ4560DQ Layout Guide
Figure 5: MPQ4560 Typical Application Circuit and PCB Layout Guide
MPQ4560 Rev. 1.1
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MPQ4560 – 2A, 2MHz, 55V, STEP-DOWN CONVERTER
PACKAGE INFORMATION
3mm × 3mm QFN10 (EXPOSED PAD)
MPQ4560 Rev. 1.1
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MPQ4560 – 2A, 2MHz, 55V, STEP-DOWN CONVERTER
SOIC8E
NOTICE: The information in this document is subject to change without notice. Users should warrant and guarantee that third
party Intellectual Property rights are not infringed upon when integrating MPS products into any application. MPS will not
assume any legal responsibility for any said applications.
MPQ4560 Rev. 1.1
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21
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