Linear LT1991 Precision, 100ua gain selectable amplifier Datasheet

LT1991
Precision, 100µA
Gain Selectable Amplifier
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FEATURES
DESCRIPTIO
■
The LT®1991 combines a precision operational amplifier
with eight precision resistors to form a one-chip solution
for accurately amplifying voltages. Gains from –13 to 14
with a gain accuracy of 0.04% can be achieved using no
external components. The device is particularly well suited
for use as a difference amplifier, where the excellent
resistor matching results in a common mode rejection
ratio of greater than 75dB.
■
■
■
■
■
■
■
■
■
■
■
Pin Configurable as a Difference Amplifier,
Inverting and Noninverting Amplifier
Difference Amplifier
Gain Range 1 to 13
CMRR >75dB
Noninverting Amplifier
Gain Range 0.07 to 14
Inverting Amplifier
Gain Range –0.08 to –13
Gain Error <0.04%
Gain Drift < 3ppm/°C
Wide Supply Range: Single 2.7V to Split ±18V
Micropower: 100µA Supply Current
Precision: 50µV Maximum Input Offset Voltage
560kHz Gain Bandwidth Product
Rail-to-Rail Output
Space Saving 10-Lead MSOP and DFN Packages
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APPLICATIO S
■
■
■
■
The amplifier features a 50µV maximum input offset
voltage and a gain bandwidth product of 560kHz. The
device operates from any supply voltage from 2.7V to 36V
and draws only 100µA supply current on a 5V supply. The
output swings to within 40mV of either supply rail.
The resistors have excellent matching, 0.04% over temperature for the 450k resistors. The matching temperature
coefficent is guaranteed less than 3ppm/°C. The resistors
are extremely linear with voltage, resulting in a gain
nonlinearity of less than 10ppm.
The LT1991 is fully specified at 5V and ±15V supplies and
from –40°C to 85°C. The device is available in space
saving 10-lead MSOP and low profile (0.8mm) 3mm ×
3mm DFN packages.
Handheld Instrumentation
Medical Instrumentation
Strain Gauge Amplifiers
Differential to Single-Ended Conversion
, LTC and LT are registered trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners. Patent Pending.
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TYPICAL APPLICATIO
Rail-to-Rail Gain = 1 Difference Amplifier
50k
450k
150k
VM(IN)
∆VIN
VP(IN)
–
450k
+
450k
INPUT RANGE
–0.5V TO 5.1V
RIN = 900kΩ
–
+
150k
4pF
LT1991
450k
40
35
PERCENTAGE OF UNITS (%)
5V
Distribution of Resistor Matching
VOUT = VREF + ∆VIN
SWING 40mV TO
EITHER RAIL
ROUT <0.1Ω
450k RESISTORS
LT1991A
30
25
20
15
10
5
50k
0
– 0.04
4pF
VREF = 2.5V
0
– 0.02
0.02
RESISTOR MATCHING (%)
0.04
1991TA01b
1991 TA01
1991fb
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LT1991
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ABSOLUTE
AXI U RATI GS
(Note 1)
Total Supply Voltage (V + to V –) ............................... 40V
Input Voltage (Pins P1/M1, Note 2) ....................... ±60V
Input Voltage
(Other inputs Note 2).............. V + + 0.2V to V – – 0.2V
Output Short-Circuit Duration (Note 3) ............ Indefinite
Operating Temperature Range (Note 4) ...–40°C to 85°C
Specified Temperature Range (Note 5) ....–40°C to 85°C
Maximum Junction Temperature
DD Package ...................................................... 125°C
MS Package ..................................................... 150°C
Storage Temperature Range
DD Package .......................................–65°C to 125°C
MS Package ......................................–65°C to 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
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PACKAGE/ORDER I FOR ATIO
ORDER PART
NUMBER
TOP VIEW
P1
1
10 M1
P3
2
9 M3
P9
3
8 M9
VEE
4
7 VCC
REF
5
6 OUT
DD PACKAGE
10-LEAD (3mm × 3mm) PLASTIC DFN
ORDER PART
NUMBER
TOP VIEW
LT1991CDD
LT1991IDD
LT1991ACDD
LT1991AIDD
DD PART MARKING*
EXPOSED PAD CONNECTED TO VEE PCB
CONNECTION OPTIONAL
TJMAX = 125°C, θJA = 160°C/W
P1
P3
P9
VEE
REF
1
2
3
4
5
10
9
8
7
6
LT1991CMS
LT1991IMS
LT1991ACMS
LT1991AIMS
M1
M3
M9
VCC
OUT
MS PACKAGE
10-LEAD PLASTIC MSOP
MS PART MARKING*
TJMAX = 150°C, θJA = 230°C/W
LBMM
LTQD
*Temperature and electrical grades are identified by a label on the shipping container. Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. Difference amplifier configuration, VS = 5V, 0V or ±15V;
VCM = VREF = half supply, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
∆G
Gain Error
VS = ±15V, VOUT = ±10V; RL = 10k
G = 1; LT1991A
G = 1; LT1991
G = 3 or 9; LT1991A
G = 3 or 9; LT1991
●
●
●
●
TYP
MAX
UNITS
±0.04
±0.08
±0.06
±0.12
%
%
%
%
GNL
Gain Nonlinearity
VS = ±15V; VOUT = ±10V; RL = 10k
●
1
10
ppm
∆G/∆T
Gain Drift vs Temperature (Note 6)
VS = ±15V; VOUT = ±10V; RL = 10k
●
0.3
3
ppm/°C
CMRR
Common Mode Rejection Ratio,
Referred to Inputs (RTI)
VS = ±15V; VCM = ±15.2V
G = 9; LT1991A
G = 3; LT1991A
G = 1; LT1991A
Any Gain; LT1991
●
●
●
●
80
75
75
60
Input Voltage Range (Note 7)
P1/M1 Inputs
VS = ±15V; VREF = 0V
VS = 5V, 0V; VREF = 2.5V
VS = 3V, 0V; VREF = 1.25V
●
●
●
–28
–0.5
0.75
VCM
100
93
90
70
dB
dB
dB
dB
27.6
5.1
2.35
V
V
V
1991fb
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LT1991
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. Difference amplifier configuration, VS = 5V, 0V or ±15V;
VCM = VREF = half supply, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
VCM
Input Voltage Range (Note 7)
P1/M1 Inputs, P9/M9 Connected to REF
VS = ±15V; VREF = 0V
VS = 5V, 0V; VREF = 2.5V
VS = 3V, 0V; VREF = 1.25V
●
●
●
P3/M3 Inputs
VS = ±15V; VREF = 0V
VS = 5V, 0V; VREF = 2.5V
VS = 3V, 0V; VREF = 1.25V
P9/M9 Inputs
VS = ±15V; VREF = 0V
VS = 5V, 0V; VREF = 2.5V
VS = 3V, 0V; VREF = 1.25V
VOS
Op Amp Offset Voltage (Note 8)
MIN
MAX
UNITS
–60
–14
–1.5
60
16.8
7.3
V
V
V
●
●
●
–15.2
0.5
0.95
15.2
4.2
1.95
V
V
V
●
●
●
–15.2
0.85
1.0
15.2
3.9
1.9
V
V
V
15
50
135
µV
µV
15
80
160
µV
µV
25
100
200
µV
µV
25
150
250
µV
µV
0.3
1
2.5
5
7.5
nA
nA
50
500
750
pA
pA
50
1000
1500
pA
pA
LT1991AMS, VS = 5V, 0V
TYP
●
LT1991AMS, VS = ±15V
●
LT1991MS
●
LT1991DD
●
∆VOS/∆T
Op Amp Offset Voltage Drift (Note 6)
IB
Op Amp Input Bias Current
●
●
IOS
Op Amp Input Offset Current
LT1991A
●
LT1991
●
µV/°C
Op Amp Input Noise Voltage
0.01Hz to 1Hz
0.01Hz to 1Hz
0.1Hz to 10Hz
0.1Hz to 10Hz
0.35
0.07
0.25
0.05
µVP-P
µVRMS
µVP-P
µVRMS
en
Input Noise Voltage Density
G = 1; f = 1kHz
G = 9; f = 1kHz
180
46
nV/√Hz
nV/√Hz
RIN
Input Impedance (Note 10)
P1 (M1 = Ground)
P3 (M3 = Ground)
P9 (M9 = Ground)
●
●
●
630
420
350
900
600
500
1170
780
650
kΩ
kΩ
kΩ
M1 (P1 = Ground)
M3 (P3 = Ground)
M9 (P9 = Ground)
●
●
●
315
105
35
450
150
50
585
195
65
kΩ
kΩ
kΩ
%
%
%
%
∆R
Resistor Matching
(Note 9)
450k Resistors, LT1991A
Other Resistors, LT1991A
450k Resistors, LT1991
Other Resistors, LT1991
●
●
●
●
0.01
0.02
0.02
0.04
0.04
0.06
0.08
0.12
∆R/∆T
Resistor Temperature Coefficient (Note 6)
Resistor Matching
Absolute Value
●
●
0.3
–30
3
PSRR
Power Supply Rejection Ratio
VS = ±1.35V to ±18V (Note 8)
●
Minimum Supply Voltage
●
105
135
2.4
ppm/°C
ppm/°C
dB
2.7
V
1991fb
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LT1991
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. Difference amplifier configuration, VS = 5V, 0V or ±15V;
VCM = VREF = half supply, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
VOUT
Output Voltage Swing (to Either Rail)
No Load
VS = 5V, 0V
VS = 5V, 0V
VS = ±15V
●
●
1mA Load
VS = 5V, 0V
VS = 5V, 0V
VS = ±15V
●
●
Drive Output Positive;
Short Output to Ground
8
4
12
●
mA
mA
Drive Output Negative;
Short Output to VS or Midsupply
8
4
21
●
mA
mA
G=1
G=3
G=9
110
78
40
kHz
kHz
kHz
ISC
Output Short-Circuit Current (Sourcing)
Output Short-Circuit Current (Sinking)
BW
–3dB Bandwidth
MIN
TYP
MAX
UNITS
40
55
65
110
mV
mV
mV
150
225
275
300
mV
mV
mV
GBWP
Op Amp Gain Bandwidth Product
f = 10kHz
560
kHz
tr, tf
Rise Time, Fall Time
G = 1; 0.1V Step; 10% to 90%
G = 9; 0.1V Step; 10% to 90%
3
8
µs
µs
ts
Settling Time to 0.01%
G = 1; VS = 5V, 0V; 2V Step
G = 1; VS = 5V, 0V; –2V Step
G = 1; VS = ±15V, 10V Step
G = 1; VS = ±15V, –10V Step
42
48
114
74
µs
µs
µs
µs
SR
Slew Rate
VS = 5V, 0V; VOUT = 1V to 4V
VS = ±15V; VOUT = ±10V
0.12
0.12
V/µs
V/µs
IS
Supply Current
VS = 5V, 0V
●
●
0.06
0.08
100
110
150
µA
µA
130
160
210
µA
µA
●
VS = ±15V
●
Note 1: Absolute Maximum Ratings are those beyond which the life of the
device may be impaired.
Note 2: The P3/M3 and P9/M9 inputs should not be taken more than 0.2V
beyond the supply rails. The P1/M1 inputs can withstand ±60V if P9/M9
are grounded and VS = ±15V (see Applications Information section about
“High Voltage CM Difference Amplifiers”).
Note 3: A heat sink may be required to keep the junction temperature
below absolute maximum ratings.
Note 4: Both the LT1991C and LT1991I are guaranteed functional over the
–40°C to 85°C temperature range.
Note 5: The LT1991C is guaranteed to meet the specified performance
from 0°C to 70°C and is designed, characterized and expected to meet
specified performance from –40°C to 85°C but is not tested or QA
sampled at these temperatures. The LT1991I is guaranteed to meet
specified performance from –40°C to 85°C.
Note 6: This parameter is not 100% tested.
Note 7: Input voltage range is guaranteed by the CMRR test at VS = ±15V.
For the other voltages, this parameter is guaranteed by design and through
correlation with the ±15V test. See the Applications Information section to
determine the valid input voltage range under various operating
conditions.
Note 8: Offset voltage, offset voltage drift and PSRR are defined as
referred to the internal op amp. You can calculate output offset as follows.
In the case of balanced source resistance, VOS,OUT = VOS • NOISEGAIN +
IOS • 450k + IB • 450k • (1– RP/RN) where RP and RN are the total
resistance at the op amp positive and negative terminal respectively.
Note 9: Applies to resistors that are connected to the inverting inputs.
Resistor matching is not tested directly, but is guaranteed by the gain
error test.
Note 10: Input impedence is tested by a combination of direct
measurements and correlation to the CMRR and gain error tests.
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LT1991
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TYPICAL PERFOR A CE CHARACTERISTICS
Output Voltage Swing vs
Temperature
200
VCC
VS = 5V, 0V
NO LOAD
OUTPUT VOLTAGE SWING (mV)
175
TA = 85°C
150
SUPPLY CURRENT (µA)
Output Voltage Swing vs Load
Current (Output Low)
TA = 25°C
125
TA = –40°C
100
75
50
60
0
2
4
6 8 10 12 14 16 18 20
SUPPLY VOLTAGE (±V)
0
25
50
75
100
TA = 25°C
–500
–600
–700
–800
–900
1
2
3 4 5 6 7
LOAD CURRENT (mA)
8
9
10
5
250
0
–250
–0.03
–100
1
6 7 8 9 10 11 12 13
GAIN (V/V)
1991 G07
2
3
4
5
6 7 8 9 10 11 12 13
GAIN (V/V)
1991 G06
Slew Rate vs Temperature
0.25
GAIN = 1
VS = ±15V
VOUT = ±10V
0.20
SR– (FALLING EDGE)
0.15
SR+ (RISING EDGE)
0.10
0.05
REPRESENTATIVE UNITS
–0.04
5
–50
0.30
–0.01
–750
4
0
1991 G05
0
–0.02
3
VS = 5V, 0V
REPRESENTATIVE PARTS
50
125
0.01
–500
2
100
SLEW RATE (V/µs)
0.02
10
–150
50
25
0
75
TEMPERATURE (°C)
GAIN = 1
VS = ±15V
VOUT = ±10V
TA = 25°C
0.03
9
100
Gain Error vs Load Current
GAIN ERROR (%)
OUTPUT OFFSET VOLTAGE (µV)
SOURCING
0.04
500
1
150
15
10
8
Input Offset Voltage vs
Difference Gain
20
1991 G04
VS = 5V, 0V
REPRESENTATIVE PARTS
750
3 4 5 6 7
LOAD CURRENT (mA)
2
1
SINKING
Output Offset Voltage vs
Difference Gain
1000
0
1991 G03
VS = 5V, 0V
0
–50 –25
–1000
0
TA = –40°C
400
VEE
125
INPUT OFFSET VOLTAGE (µV)
OUTPUT SHORT-CIRCUIT CURRENT (mA)
OUTPUT VOLTAGE SWING (mV)
TA = 85°C
–400
TA = 25°C
600
1991 G02
25
TA = –40°C
–300
800
Output Short-Circuit Current vs
Temperature
VS = 5V, 0V
–200
1000
TEMPERATURE (°C)
Output Voltage Swing vs Load
Current (Output High)
VCC
TA = 85°C
200
1991 G01
–100
VS = 5V, 0V
1200
OUTPUT LOW
(LEFT AXIS)
VEE
–50 –25
0
–40
–60
20
25
–20
OUTPUT HIGH
(RIGHT AXIS)
40
1400
OUTPUT VOLTAGE (mV)
Supply Current vs Supply Voltage
–1000
(Difference Amplifier Configuration)
0
1
2
3
LOAD CURRENT (mA)
4
5
1991 G08
0
–50 –25
50
25
75
0
TEMPERATURE (°C)
100
125
1991 G09
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LT1991
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TYPICAL PERFOR A CE CHARACTERISTICS
Bandwidth vs Gain
CMRR vs Frequency
120
VS = 5V, 0V
TA = 25°C
110
100
GAIN = 1
90
GAIN = 3
80
60
40
70
60
50
3 4
40
30
20
20
10
10
10
100
1k
10k
FREQUENCY (Hz)
VS = 5V, 0V
TA = 25°C
GAIN = 1
VS = ±15V
GAIN ERROR (%)
CMRR (dB)
GAIN = 3
GAIN = 1
60
40
0.1
0
–50 –25
0.01
100
1k
FREQUENCY (Hz)
10k
100k
100
VS = 5V, 0V
TA = 25°C
0
GAIN (dB)
GAIN = 1
GAIN
–45
–2
–90
–3
–4
–135
–6
–10
–180
–7
1
10
100
FREQUENCY (kHz)
600
1991 G16
100
–8
0.5
1
10
FREQUENCY (kHz)
100
125
1991 G15
VS = 5V, 0V
TA = 25°C 0
GAIN = 1
1 PHASE
GAIN = 9
GAIN = 3
REPRESENTATIVE UNITS
50
25
75
0
TEMPERATURE (°C)
0.01Hz to 1Hz Voltage Noise
–5
–20
0
–50 –25
125
400
PHASE (deg)
0
0.010
Gain and Phase vs Frequency
2
–1
10
0.015
1991 G14
Gain vs Frequency
20
0.020
REPRESENTATIVE UNITS
50
25
75
0
TEMPERATURE (°C)
1991 G13
30
GAIN = 1
VS = ±15V
0.005
20
10
100k
0.025
80
GAIN = 9
1
1k
10k
FREQUENCY (Hz)
Gain Error vs Temperature
0.030
100
100
1
100
1991 G12
CMRR vs Temperature
120
10
10
1991 G11
Output Impedance vs Frequency
OUTPUT IMPEDANCE (Ω)
0
1M
100k
1991 G10
1000
50
30
5 6 7 8 9 10 11 12 13
GAIN SETTING (V/V)
GAIN = 3
60
VS = ±15V
TA = 25°C
MEASURED IN G =13
REFERRED TO OP AMP INPUTS
OP AMP VOLTAGE NOISE (100nV/DIV)
1 2
GAIN = 1
70
40
0
GAIN = 9
80
PSRR (dB)
CMRR (dB)
80
VS = 5V, 0V
TA = 25°C
110
100
20
GAIN (dB)
120
VS = 5V, 0V
TA = 25°C
GAIN = 9
90
–3dB BANDWIDTH (kHz)
PSRR vs Frequency
120
100
0
(Difference Amplifier Configuration)
0
10 20 30 40 50 60 70 80 90 100
TIME (s)
1991 G17
1991 G21
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LT1991
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TYPICAL PERFOR A CE CHARACTERISTICS
Small Signal Transient Response
Small Signal Transient Response
GAIN = 1
Small Signal Transient Response
GAIN = 9
GAIN = 3
50mV/DIV
50mV/DIV
50mV/DIV
1991 G18
5µs/DIV
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PI FU CTIO S
1991 G19
5µs/DIV
5µs/DIV
1991 G20
(Difference Amplifier Configuration)
P1 (Pin 1): Noninverting Gain-of-1 input. Connects a 450k
internal resistor to the op amp’s noninverting input.
OUT (Pin 6): Output. VOUT = VREF + 1 • (VP1 – VM1) + 3 •
(VP3 – VM3) + 9 • (VP9 – VM9).
P3 (Pin 2): Noninverting Gain-of-3 input. Connects a 150k
internal resistor to the op amp’s noninverting input.
VCC (Pin 7): Positive Power Supply. Can be anything from
2.7V to 36V above the VEE voltage.
P9 (Pin 3): Noninverting Gain-of-9 input. Connects a 50k
internal resistor to the op amp’s noninverting input.
M9 (Pin 8): Inverting Gain-of-9 input. Connects a 50k
internal resistor to the op amp’s inverting input.
VEE (Pin 4): Negative Power Supply. Can be either ground
(in single supply applications), or a negative voltage (in
split supply applications).
M3 (Pin 9): Inverting Gain-of-3 input. Connects a 150k
internal resistor to the op amp’s inverting input.
REF (Pin 5): Reference Input. Sets the output level when
difference between inputs is zero. Connects a 450k internal resistor to the op amp’s noninverting input.
M1 (Pin 10): Inverting Gain-of-1 input. Connects a 450k
internal resistor to the op amp’s inverting input.
Exposed Pad: Must be soldered to PCB.
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BLOCK DIAGRA
M1
M3
M9
VCC
OUT
10
9
8
7
6
50k
450k
150k
4pF
450k
INM
OUT
450k
INP
LT1991
150k
450k
50k
4pF
1
2
3
4
5
P1
P3
P9
VEE
REF
1991 BD
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LT1991
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APPLICATIO S I FOR ATIO
Introduction
The LT1991 may be the last op amp you ever have to stock.
Because it provides you with several precision matched
resistors, you can easily configure it into several different
classical gain circuits without adding external components. The several pages of simple circuits in this data
sheet demonstrate just how easy the LT1991 is to use. It
can be configured into difference amplifiers, as well as into
inverting and noninverting single ended amplifiers. The
fact that the resistors and op amp are provided together in
such a small package will often save you board space and
reduce complexity for easy probing.
admittances. Because it has 9 times the admittance, the
voltage applied to the P9 input has 9 times the effect of the
voltage applied to the P1 input.
Bandwidth
The bandwidth of the LT1991 will depend on the gain you
select (or more accurately the noise gain resulting from
the gain you select). In the lowest configurable gain of 1,
the –3dB bandwidth is limited to 450kHz, with peaking of
about 2dB at 280kHz. In the highest configurable gains,
bandwidth is limited to 32kHz.
Input Noise
The Op Amp
The op amp internal to the LT1991 is a precision device
with 15µV typical offset voltage and 3nA input bias current. The input offset current is extremely low, so matching the source resistance seen by the op amp inputs will
provide for the best output accuracy. The op amp inputs
are not rail-to-rail, but extend to within 1.2V of VCC and 1V
of VEE. For many configurations though, the chip inputs
will function rail-to-rail because of effective attenuation to
the +input. The output is truly rail-to-rail, getting to within
40mV of the supply rails. The gain bandwidth product of
the op amp is about 560kHz. In noise gains of 2 or more,
it is stable into capacitive loads up to 500pF. In noise gains
below 2, it is stable into capacitive loads up to 100pF.
The Resistors
The resistors internal to the LT1991 are very well matched
SiChrome based elements protected with barrier metal.
Although their absolute tolerance is fairly poor (±30%),
their matching is to within 0.04%. This allows the chip to
achieve a CMRR of 75dB, and gain errors within 0.04%.
The resistor values are 50k, 150k, and 2 of 450k, connected to each of the inputs. The resistors have power
limitations of 1watt for the 450k resistors, 0.3watt for the
150k resistors and 0.5watt for the 50k resistors; however,
in practice, power dissipation will be limited well below
these values by the maximum voltage allowed on the input
and REF pins. The 450k resistors connected to the M1 and
P1 inputs are isolated from the substrate, and can therefore be taken beyond the supply voltages. The naming of
the pins “P1,” “P3,” “P9,” etc., is based on their relative
The LT1991 input noise is dominated by the Johnson
noise of the internal resistors (√4kTR). Paralleling all four
resistors to the +input gives a 32.1kΩ resistance, for
23nV/√Hz of voltage noise. The equivalent network on the
–input gives another 23nV/√Hz, and taking their RMS sum
gives a total 33nV/√Hz input referred noise floor. Output
noise depends on configuration and noise gain.
Input Resistance
The LT1991 input resistances vary with configuration, but
once configured are apparent on inspection. Note that
resistors connected to the op amp’s –input are looking
into a virtual ground, so they simply parallel. Any feedback
resistance around the op amp does not contribute to input
resistance. Resistors connected to the op amp’s +input
are looking into a high impedance, so they add as parallel
or series depending on how they are connected, and
whether or not some of them are grounded. The op amp
+input itself presents a very high GΩ impedance. In the
classical noninverting op amp configuration, the LT1991
presents the high input impedance of the op amp, as is
usual for the noninverting case.
Common Mode Input Voltage Range
The LT1991 valid common mode input range is limited by
three factors:
1. Maximum allowed voltage on the pins
2. The input voltage range of the internal op amp
3. Valid output voltage
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The maximum voltage allowed on the P3, M3, P9, and M9
inputs includes the positive and negative supply plus a
diode drop. These pins should not be driven more than
0.2V outside of the supply rails. This is because they are
connected through diodes to internal manufacturing postpackage trim circuitry, and through a substrate diode to
VEE. If more than 10mA is allowed to flow through these
pins, there is a risk that the LT1991 will be detrimmed or
damaged. The P1 and M1 inputs do not have clamp diodes
or substrate diodes or trim circuitry and can be taken well
outside the supply rails. The maximum allowed voltage on
the P1 and M1 pins is ±60V.
The input voltage range of the internal op amp extends to
within 1.2V of VCC and 1V of VEE. The voltage at which the
op amp inputs common mode is determined by the
voltage at the op amp’s +input, and this is determined by
the voltages on pins P1, P3, P9 and REF. (See “Calculating
Input Voltage Range” section.) This is true provided that
the op amp is functioning and feedback is maintaining the
inputs at the same voltage, which brings us to the third
requirement.
For valid circuit function, the op amp output must not be
clipped. The output will clip if the input signals are attempting to force it to within 40mV of its supply voltages. This
usually happens due to too large a signal level, but it can
also occur with zero input differential and must therefore
be included as an example of a common mode problem.
Consider Figure 1. This shows the LT1991 configured as
a gain of 13 difference amplifier on a single supply with the
output REF connected to ground. This is a great circuit, but
it does not support VDM = 0V at any common mode
because the output clips into ground while trying to
produce 0VOUT. It can be fixed simply by declaring the
valid input differential range not to extend below +4mV, or
by elevating the REF pin above 40mV, or by providing a
negative supply.
Calculating Input Voltage Range
Figure 2 shows the LT1991 in the generalized case of a
difference amplifier, with the inputs shorted for the common mode calculation. The values of RF and RG are
dictated by how the P inputs and REF pin are connected.
By superposition we can write:
VINT = VEXT • (RF/(RF + RG)) + VREF • (RG/(RF + RG))
Or, solving for VEXT:
VEXT = VINT • (1 + RG/RF) – VREF • RG/RF
But valid VINT voltages are limited to VCC – 1.2V and VEE +
1V, so:
MAX VEXT = (VCC – 1.2) • (1 + RG/RF) – VREF • RG/RF
and:
MIN VEXT = (VEE + 1) • (1 + RG/RF) – VREF • RG/RF
RF
VCC
RG
5V
–
7
8
50k
VEXT
450k
4pF
9
VINT
RG
150k
+
VEE
VREF
RF
10
450k
–
VDM
0V+
VCM
2.5V
–
6
1
450k
2
150k
3
50k
Figure 2. Calculating CM Input Voltage Range
VOUT = 13 • VDM
+
4pF
450k
REF 5
LT1991
4
1991 F02
1991 F01
Figure 1. Difference Amplifier Cannot Produce
0V on a Single Supply. Provide a Negative
Supply, or Raise Pin 5, or Provide 4mV of VDM
These two voltages represent the high and low extremes
of the common mode input range, if the other limits have
not already been exceeded (1 and 3, above). In most cases,
the inverting inputs M1 through M9 can be taken further
than these two extremes because doing this does not
move the op amp input common mode. To calculate the
limit on this additional range, see Figure 3. Note that, with
VMORE = 0, the op amp output is at VREF. From the max
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VEXT (the high cm limit), as VMORE goes positive, the op
amp output will go more negative from VREF by the amount
VMORE • RF/RG, so:
VOUT = VREF – VMORE • RF/RG
Or:
VMORE = (VREF – VOUT) • RG/RF
The most negative that VOUT can go is VEE + 0.04V, so:
Max VMORE = (VREF – VEE – 0.04V) • RG/RF
(should be positive)
The situation where this function is negative, and therefore
problematic, when VREF = 0 and VEE = 0, has already been
dealt with in Figure 1. The strength of the equation is
demonstrated in that it provides the three solutions suggested in Figure 1: raise VREF, lower VEE, or provide some
negative VMORE.
representation of the circuit on the top. The LT1991 is
shown on the bottom configured in a precision gain of 5.5.
One of the benefits of the noninverting op amp configuration is that the input impedance is extremely high. The
LT1991 maintains this benefit. Given the finite number of
available feedback resistors in the LT1991, the number of
gain configurations is also finite. The complete list of such
Hi-Z input noninverting gain configurations is shown in
Table 1. Many of these are also represented in Figure 5 in
schematic form. Note that the P-side resistor inputs have
been connected so as to match the source impedance
seen by the internal op amp inputs. Note also that gain and
noise gain are identical, for optimal precision.
RF
RG
–
VOUT
Likewise, from the lower common mode extreme, making
the negative input more negative will raise the output
voltage, limited by VCC – 0.04V.
VIN
+
VOUT = GAIN • VIN
GAIN = 1 + RF/RG
CLASSICAL NONINVERTING OP AMP CONFIGURATION.
YOU PROVIDE THE RESISTORS.
MIN VMORE = (VREF – VCC + 0.04V) • RG/RF
(should be negative)
RF
9
150k
10
450k
1
450k
2
150k
3
50k
450k
4pF
–
VMORE
VINT
RG
50k
VCC
RG
VEXT
MAX OR MIN
8
+
–
6
VEE
VREF
RF
1991 F03
Figure 3. Calculating Additional
Voltage Range of Inverting Inputs
Again, the additional input range calculated here is only
available provided the other remaining constraint is not
violated, the maximum voltage allowed on the pin.
The Classical Noninverting Amplifier: High Input Z
Perhaps the most common op amp configuration is the
noninverting amplifier. Figure 4 shows the textbook
VOUT
+
4pF
450k
LT1991
5
VIN
CLASSICAL NONINVERTING OP AMP CONFIGURATION
IMPLEMENTED WITH LT1991. RF = 225k, RG = 50k, GAIN = 5.5.
GAIN IS ACHIEVED BY GROUNDING, FLOATING OR FEEDING BACK
THE AVAILABLE RESISTORS TO ARRIVE AT DESIRED RF AND RG.
WE PROVIDE YOU WITH <0.1% RESISTORS.
1991 F04
Figure 4. The LT1991 as a Classical Noninverting Op Amp
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Table 1. Configuring the M Pins for Simple Noninverting Gains.
The P Inputs are driven as shown in the examples on the next
page
Gain
M9
M9, M3, M1 Connection
M3
M1
1
Output
Output
Output
1.077
Output
Output
Ground
1.1
Output
Float
Ground
1.25
Float
Output
Ground
1.273
Output
Ground
Output
1.3
Output
Ground
Float
1.4
Output
Ground
Ground
2
Float
Float
Ground
2.5
Float
Ground
Output
2.8
Ground
Output
Output
3.25
Ground
Output
Float
3.5
Ground
Output
Ground
4
Float
Ground
Float
5
Float
Ground
Ground
5.5
Ground
Float
Output
7
Ground
Ground
Output
10
Ground
Float
Float
11
Ground
Float
Ground
13
Ground
Ground
Float
14
Ground
Ground
Ground
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VS+
8
M9
9
M3
10
M1
VIN
1
P1
2
P3
3
P9
VS+
8
M9
9
M3
10
M1
7
VCC
LT1991
VEE
OUT
REF
5
6
VOUT
1
P1
2
P3
3
P9
VIN
4
VS –
7
VCC
LT1991
VEE
6
VOUT
VIN
4
LT1991
VEE
OUT
REF
5
6
VOUT
1
P1
2
P3
3
P9
4
VS–
VS+
7
VCC
LT1991
VEE
OUT
REF
5
6
VOUT
1
P1
2
P3
3
P9
4
VS+
8
M9
9
M3
10
M1
7
VCC
VEE
7
VCC
LT1991
VEE
VOUT
1
P1
2
P3
3
P9
VIN
4
OUT
REF
5
6
VOUT
4
V S–
GAIN = 5.5
VS+
6
VOUT
4
8
M9
9
M3
10
M1
GAIN = 5
OUT
REF
5
6
VIN
GAIN = 4
VIN
VEE
OUT
REF
5
GAIN = 3.25
VIN
LT1991
LT1991
VS –
V S–
VIN
1
P1
2
P3
3
P9
7
VCC
VS+
8
M9
9
M3
10
M1
7
VCC
8
M9
9
M3
10
M1
1
P1
2
P3
3
P9
GAIN = 2
VS+
1
P1
2
P3
3
P9
OUT
REF
5
V S–
GAIN = 1
8
M9
9
M3
10
M1
VS+
8
M9
9
M3
10
M1
VS–
VS+
8
M9
9
M3
10
M1
7
VCC
LT1991
VEE
OUT
REF
5
6
VOUT
1
P1
2
P3
3
P9
4
V S–
7
VCC
LT1991
VEE
OUT
REF
5
6
VOUT
4
V S–
VIN
GAIN = 7
GAIN = 10
GAIN = 11
VS+
8
M9
9
M3
10
M1
VIN
1
P1
2
P3
3
P9
VS +
8
M9
9
M3
10
M1
7
VCC
LT1991
VEE
OUT
REF
5
6
VOUT
1
P1
2
P3
3
P9
4
VS–
7
VCC
LT1991
VEE
OUT
REF
5
6
VOUT
4
V S–
VIN
GAIN = 13
GAIN = 14
1991 F05
Figure 5. Some Implementations of Classical Noninverting
Gains Using the LT1991. High Input Z Is Maintained
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Attenuation Using the P Input Resistors
Attenuation happens as a matter of fact in difference
amplifier configurations, but it is also used for reducing
peak signal level or improving input common mode range
even in single ended systems. When signal conditioning
indicates a need for attenuation, the LT1991 resistors are
ready at hand. The four precision resistors can provide
several attenuation levels, and these are tabulated in
Table 2 as a design reference.
VIN
VIN
OKAY UP
TO ±60V
RA
VINT
RG
1
VINT = A • VIN
A = RG/(RA + RG)
VINT
450k
2
150k
3
50k
+
4pF
450k
LT1991
5
CLASSICAL ATTENUATOR
LT1991 ATTENUATING TO THE +INPUT BY
DRIVING AND GROUNDING AND FLOATING
INPUTS RA = 450k, RG = 50k, SO A = 0.1.
1991 F06
Figure 6. LT1991 Provides for Easy Attenuation to the Op Amp’s
+Input. The P1 Input Can Be Taken Well Outside of the Supplies
Because the attenuations and the noninverting gains are
set independently, they can be combined. This provides
high gain resolution, about 340 unique gains between
0.077 and 14, as plotted in Figure 7. This is too large a
number to tabulate, but the designer can calculate achievable gain by taking the vector product of the gains and
attenuations in Tables 1 and 2, and seeking the best match.
Average gain resolution is 1.5%, with a worst case of 7%.
100
GAIN
10
Table 2. Configuring the P Pins for Various Attenuations. Those
Shown in Bold Are Functional Even When the Input Drive
Exceeds the Supplies
P9, P3, P1, REF Connection
P3
P1
A
P9
REF
0.0714
Ground
Ground
Drive
0.0769
Ground
Ground
Drive
Float
0.0909
Ground
Float
Drive
Ground
0.1
Ground
Float
Drive
Float
0.143
Ground
Ground
Drive
Drive
0.182
Ground
Float
Drive
Drive
0.2
Float
Ground
Drive
Ground
0.214
Ground
Drive
Ground
Ground
0.231
Ground
Drive
Float
Ground
0.25
Float
Ground
Drive
Float
0.286
Ground
Drive
Drive
Ground
0.308
Ground
Drive
Drive
Float
0.357
Ground
Drive
Drive
Drive
Ground
0.4
Float
Ground
Drive
Drive
0.5
Float
Float
Drive
Ground
0.6
Float
Drive
Ground
Ground
0.643
Drive
Ground
Ground
Ground
0.692
Drive
Ground
Float
Ground
0.714
Drive
Ground
Drive
Ground
0.75
Float
Drive
Float
Ground
0.769
Drive
Ground
Drive
Float
0.786
Drive
Ground
Drive
Drive
0.8
Float
Drive
Drive
Ground
0.818
Drive
Float
Ground
Ground
0.857
Drive
Drive
Ground
Ground
0.9
Drive
Float
Float
Ground
0.909
Drive
Float
Drive
Ground
0.923
Drive
Drive
Float
Ground
0.929
Drive
Drive
Drive
Ground
1
Drive
Drive
Drive
Drive
1
0.1
0.01
0
50
100
150 200
COUNT
250
300
350
1991 F07
Figure 7. Over 346 Unique Gain Settings Achievable with the
LT1991 by Combining Attenuation with Noninverting Gain
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Table 3. Configuring the M Pins for Simple Inverting Gains
Inverting Configuration
The inverting amplifier, shown in Figure 8, is another
classical op amp configuration. The circuit is actually
identical to the noninverting amplifier of Figure 4, except
that VIN and GND have been swapped. The list of available
gains is shown in Table 3, and some of the circuits are
shown in Figure 9. Noise gain is 1+|Gain|, as is the usual
case for inverting amplifiers. Again, for the best DC performance, match the source impedance seen by the op amp
inputs.
RF
RG
VIN
–
VOUT
+
VOUT = GAIN • VIN
GAIN = – RF/RG
CLASSICAL INVERTING OP AMP CONFIGURATION.
YOU PROVIDE THE RESISTORS.
VIN
(DRIVE)
8
50k
9
150k
10
450k
1
450k
2
150k
3
50k
450k
Gain
M9
M9, M3, M1 Connection
M3
M1
–0.077
Output
Output
Drive
–0.1
Output
Float
Drive
–0.25
Float
Output
Drive
–0.273
Output
Drive
Output
–0.3
Output
Drive
Float
–0.4
Output
Drive
Drive
–1
Float
Float
Drive
–1.5
Float
Drive
Output
–1.8
Drive
Output
Output
–2.25
Drive
Output
Float
–2.5
Drive
Output
Drive
–3
Float
Drive
Float
–4
Float
Drive
Drive
–4.5
Drive
Float
Output
–6
Drive
Drive
Output
–9
Drive
Float
Float
–10
Drive
Float
Drive
–12
Drive
Drive
Float
–13
Drive
Drive
Drive
4pF
–
6
VOUT
+
4pF
450k
LT1991
5
CLASSICAL INVERTING OP AMP CONFIGURATION IMPLEMENTED
WITH LT1991. RF = 225k, RG = 50k, GAIN = –4.5.
GAIN IS ACHIEVED BY GROUNDING, FLOATING OR FEEDING BACK
THE AVAILABLE RESISTORS TO ARRIVE AT DESIRED RF AND RG.
WE PROVIDE YOU WITH <0.1% RESISTORS.
1991 F08
Figure 8. The LT1991 as a Classical Inverting Op Amp.
Note the Circuit Is Identical to the Noninverting Amplifier,
Except that VIN and Ground Have Been Swapped
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VS +
8
M9
9
M3
10
M1
VIN
1
P1
2
P3
3
P9
VS +
7
VCC
LT1991
VEE
OUT
REF
5
6
8
M9
9
M3
10
M1
VIN
VOUT
1
P1
2
P3
3
P9
4
VS–
7
VCC
LT1991
VEE
1
P1
2
P3
3
P9
LT1991
VEE
OUT
REF
5
6
8
M9
9
M3
10
M1
VIN
VOUT
1
P1
2
P3
3
P9
4
LT1991
VEE
OUT
REF
5
6
VOUT
7
VCC
LT1991
VEE
6
VOUT
1
P1
2
P3
3
P9
4
VS–
VS+
VIN
LT1991
OUT
REF
5
6
VOUT
8
M9
9
M3
10
M1
1
P1
2
P3
3
P9
4
GAIN = –9
8
M9
9
M3
10
M1
1
P1
2
P3
3
P9
VEE
OUT
REF
5
4
LT1991
VEE
OUT
REF
5
6
VOUT
4
GAIN = –10
VS +
7
VCC
LT1991
7
VCC
VS–
VS +
VIN
VOUT
GAIN = –4.5
VS–
GAIN = –6
6
VS–
7
VCC
VEE
OUT
REF
5
4
VS +
OUT
REF
5
VOUT
4
8
M9
9
M3
10
M1
1
P1
2
P3
3
P9
4
8
M9
9
M3
10
M1
VIN
6
VS+
VIN
GAIN = –4
7
VCC
VEE
VEE
OUT
REF
5
GAIN = –2.25
7
VCC
VS +
LT1991
LT1991
VS–
VS–
GAIN = –3
1
P1
2
P3
3
P9
1
P1
2
P3
3
P9
7
VCC
VS +
7
VCC
8
M9
9
M3
10
M1
VOUT
GAIN = –1
VS–
VIN
6
4
VS +
VIN
OUT
REF
5
8
M9
9
M3
10
M1
VS–
GAIN = –0.25
8
M9
9
M3
10
M1
VS+
VIN
6
VIN
VOUT
8
M9
9
M3
10
M1
1
P1
2
P3
3
P9
7
VCC
LT1991
VEE
OUT
REF
5
6
VOUT
4
VS–
VS–
GAIN = –12
GAIN = –13
1991 F09
Figure 9. It Is Simple to Get Precision Inverting Gains with the LT1991.
Input Impedance Varies from 45kΩ (Gain = –13) to 450kΩ (Gain = –1)
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RF
Difference Amplifiers
The resistors in the LT1991 allow it to easily make difference amplifiers also. Figure 10 shows the basic 4-resistor
difference amplifier and the LT1991. A difference gain of
3 is shown, but notice the effect of the additional dashed
connections. By connecting the 450k resistors in parallel,
the gain is reduced by a factor of 2. Of course, with so
many resistors, there are many possible gains. Table 4
shows the difference gains and how they are achieved.
Note that, as for inverting amplifiers, the noise gain is 1
more than the signal gain.
VIN–
VIN+
RG
RG
–
VOUT
+
RF
CLASSICAL DIFFERENCE AMPLIFIER USING THE LT1991
8 M9 50k
Table 4. Connections Giving Difference Gains for the LT1991
+
Gain
VIN
0.077
P1
0.1
P1
–
Output
GND (REF)
M1
M3, M9
P3, P9
VIN
M1
M9
P9
0.25
P1
M1
M3
P3
0.273
P3
M3
M1, M9
P1, P9
0.3
P3
M3
M9
P9
0.4
P1, P3
M1, M3
M9
P9
1
P1
M1
P3
M3
M1
P1
1.8
P9
M9
M1, M3
P1, P3
2.25
P9
M9
M3
P3
2.5
P1, P9
M1, M9
M3
P3
3
P3
M3
4
P1, P3
M1, M3
4.5
P9
M9
M1
P1
6
P3, P9
M3, M9
M1
P1
9
P9
M9
P1, P9
M1, M9
12
P3, P9
M3, M9
13
P1, P3, P9
M1, M3, M9
VIN–
VIN+
4pF
9 M3 150k
10 M1 450k
PARALLEL
TO CHANGE
R F, R G
450k
–
6
1 P1
450k
2 P3
150k
3 P9
50k
VOUT
+
4pF
450k
5
LT1991
1.5
10
VOUT = GAIN • (VIN+ – VIN–)
GAIN = RF/RG
CLASSICAL DIFFERENCE AMPLIFIER IMPLEMENTED
WITH LT1991. RF = 450k, RG = 150k, GAIN = 3.
ADDING THE DASHED CONNECTIONS CONNECTS THE
TWO 450k RESISTORS IN PARALLEL, SO RF IS REDUCED
TO 225k. GAIN BECOMES 225k/150k = 1.5.
1991 F10
Figure 10. Difference Amplifier Using the LT1991. Gain Is Set
Simply by Connecting the Correct Resistors or Combinations
of Resistors. Gain of 3 Is Shown, with Dashed Lines Modifying
It to Gain of 1.5. Noise Gain Is Optimal
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VS +
VIN
VIN
8
M9
9
M3
10
M1
–
1
P1
2
P3
3
P9
+
VS +
7
VCC
LT1991
VEE
OUT
REF
5
6
VIN
–
VIN
+
8
M9
9
M3
10
M1
VOUT
1
P1
2
P3
3
P9
4
VS–
7
VCC
LT1991
VEE
VIN+
1
P1
2
P3
3
P9
LT1991
VEE
OUT
REF
5
6
VIN–
8
M9
9
M3
10
M1
VIN+
1
P1
2
P3
3
P9
VOUT
4
VIN+
LT1991
VEE
OUT
REF
5
6
VOUT
VIN+
4
OUT
REF
5
6
VOUT
1
P1
2
P3
3
P9
VIN+
4
8
M9
9
M3
10
M1
1
P1
2
P3
3
P9
7
VCC
LT1991
VEE
VS–
4
LT1991
OUT
REF
5
6
VOUT
VIN+
4
8
M9
9
M3
10
M1
1
P1
2
P3
3
P9
VIN+
1
P1
2
P3
3
P9
VEE
OUT
REF
5
4
VS–
GAIN = 12
LT1991
VEE
OUT
REF
5
6
VOUT
4
GAIN = 10
VS +
VIN–
7
VCC
LT1991
7
VCC
VS–
GAIN = 9
8
M9
9
M3
10
M1
VOUT
VS+
VIN–
VS +
VIN–
6
GAIN = 4.5
VS–
GAIN = 6
OUT
REF
5
VS–
7
VCC
VEE
VOUT
4
VS +
8
M9
9
M3
10
M1
VIN–
6
VS+
VIN–
GAIN = 4
7
VCC
VEE
VEE
OUT
REF
5
GAIN = 2.25
7
VCC
VS +
LT1991
LT1991
VS–
VS–
GAIN = 3
1
P1
2
P3
3
P9
VIN+
1
P1
2
P3
3
P9
7
VCC
VS +
7
VCC
8
M9
9
M3
10
M1
VOUT
GAIN = 1
VS–
VIN–
6
4
VS +
VIN–
OUT
REF
5
8
M9
9
M3
10
M1
VS–
GAIN = 0.25
8
M9
9
M3
10
M1
VS+
VIN–
6
VOUT
VIN+
8
M9
9
M3
10
M1
1
P1
2
P3
3
P9
7
VCC
LT1991
VEE
OUT
REF
5
6
VOUT
4
VS–
GAIN = 13
1991 F11
Figure 11. Many Difference Gains Are Achievable Just by Strapping the Pins
1991fb
17
LT1991
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APPLICATIO S I FOR ATIO
450k
8 M9 50k
RF
VIN–
VIN+
10 M1 450k
CROSSCOUPLING
RG
VOUT
+
RF
–
6
–
RG
4pF
9 M3 150k
VIN–
VIN+
VOUT = GAIN • (VIN+ – VIN–)
GAIN = RF/RG
1 P1
450k
2 P3
150k
3 P9
50k
VOUT
+
4pF
450k
5
LT1991
CLASSICAL DIFFERENCE AMPLIFIER IMPLEMENTED
WITH LT1991. RF = 450k, RG = 150k, GAIN = 3.
GAIN CAN BE ADJUSTED BY "CROSS COUPLING." MAKING THE
DASHED CONNECTIONS REDUCE THE GAIN FROM 3 T0 2.
WHEN CROSS COUPLING, SEE WHAT IS CONNECTED TO THE
VIN+ VOLTAGE. CONNECTING P3 AND M1 GIVES +3 –1 = 2.
CONNECTIONS TO VIN– ARE SYMMETRIC: M3 AND P1.
CLASSICAL DIFFERENCE AMPLIFIER
1991 F10
Figure 12. Another Method of Selecting Difference Gain Is “Cross-Coupling.”
The Additional Method Means the LT1991 Provides All Integer Gains from 1 to 13
Difference Amplifier: Additional Integer Gains Using
Cross-Coupling
Figure 12 shows the basic difference amplifier as well as
the LT1991 in a difference gain of 3. But notice the effect
of the additional dashed connections. This is referred to as
“cross-coupling” and has the effect of reducing the differential gain from 3 to 2. Using this method, additional
integer gains are achievable, as shown in Table 5 below, so
that all integer gains from 1 to 13 are achieved with the
LT1991. Note that the equations can be written by inspection from the VIN+ connections, and that the VIN– connections are simply the opposite (swap P for M and M for P).
Noise gain, bandwidth, and input impedance specifications for the various cases are also tabulated, as these are
not obvious. Schematics are provided in Figure 13.
VS+
VIN–
+
VIN
8
M9
9
M3
10
M1
1
P1
2
P3
3
P9
7
VCC
LT1991
VEE
Gain
VIN+
VIN–
2
P3, M1
M3, P1
5
70
281
141
P9, M3, M1 M9, P3, P1 9 – 3 – 1 14
32
97
49
13
35
122
49
P9, P1, M3 M9, M1, P3 9 + 1 – 3 14
32
121
44
11
38
248
50
11 P9, P3, M1 M9, M3, P1 9 + 3 – 1 14
32
242
37
5
6*
7
8
P9, M3
P9, M1
M9, P3
M9, P1
3–1
9–3
9–1
6
VOUT
1
P1
2
P3
3
P9
VIN+
VS–
7
VCC
LT1991
VEE
GAIN = 2
VIN+
8
M9
9
M3
10
M1
1
P1
2
P3
3
P9
6
VOUT
4
GAIN = 5
VS+
VIN–
OUT
REF
5
VS–
VS+
LT1991
VEE
OUT
REF
5
8
M9
9
M3
10
M1
VIN–
7
VCC
6
VOUT
1
P1
2
P3
3
P9
VIN+
4
VS–
7
VCC
LT1991
VEE
OUT
REF
5
6
VOUT
4
VS–
GAIN = 7
GAIN = 8
VS+
VIN–
–
Noise –3dB BW RIN
RIN
Equation Gain
kHz Typ kΩ Typ kΩ
OUT
REF
5
4
Table 5. Connections Using Cross-Coupling. Note That Equations
Can Be Written by Inspection of the VIN+ Column
+
VS+
8
M9
9
M3
10
M1
VIN–
VIN+
8
M9
9
M3
10
M1
1
P1
2
P3
3
P9
7
VCC
LT1991
VEE
OUT
REF
5
6
VOUT
4
VS–
GAIN = 11
1991 F13
Figure 13. Integer Gain Difference
Amplifiers Using Cross-Coupling
*Gain of 6 is better implemented as shown previously, but is included here for completeness.
1991fb
18
LT1991
U
W
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APPLICATIO S I FOR ATIO
High Voltage CM Difference Amplifiers
This class of difference amplifier remains to be discussed.
Figure 14 shows the basic circuit on the top. The effective
input voltage range of the circuit is extended by the fact
that resistors RT attenuate the common mode voltage
seen by the op amp inputs. For the LT1991, the most
useful resistors for RG are the M1 and P1 450kΩ resistors,
because they do not have diode clamps to the supplies and
therefore can be taken outside the supplies. As before, the
input CM of the op amp is the limiting factor and is set by
the voltage at the op amp +input, VINT. By superposition
we can write:
VINT = VEXT • (RF||RT)/(RG + RF||RT) + VREF • (RG||RT)/
(RF + RG||RT) + VTERM • (RF||RG)/(RT + RF||RG)
Solving for VEXT:
Table 6. HighV CM Connections Giving Difference Gains
for the LT1991
2 • VLIM - VREF
VIN–
1
P1
M1
1
P1
M1
P3, M3
5
5 • VLIM – VREF – 3 • VTERM
1
P1
M1
P9, M9
11
11 • VLIM – VREF – 9 • VTERM
1
P1
M1
P3||P9
M3||M9
14
14 • VLIM – VREF – 12 • VTERM
RT
RF
VCC
RG
VIN–
–
VOUT
RG
VIN+
(= VEXT)
+
RT
RT
VOUT = GAIN • (VIN+ – VIN–)
VEE GAIN = RF/RG
RF
VREF
VTERM
HIGH CM VOLTAGE DIFFERENCE AMPLIFIER
INPUT CM TO OP AMP IS ATTENUATED BY
RESISTORS RT CONNECTED TO VTERM.
7
12V
8 M9 50k
450k
4pF
9 M3 150k
10 M1 450k
–
6
= 11 • (10.8V) – 2.5 – 9 • 12 = 8.3V
MIN VEXT = 11 • (VEE + 1V) – VREF – 9 • VTERM
2
VIN+
MAX VEXT = 11 • (VCC – 1.2V) – VREF – 9 • VTERM
and:
Max, Min VEXT
(Substitute VCC – 1.2,
VEE + 1 for VLIM)
Gain
VEXT = (1 + RG/(RF||RT)) • (VINT – VREF • (RG||RT)/
(RF + RG||RT) – VTERM • (RF||RG)/(RT + RF||RG))
Given the values of the resistors in the LT1991, this
equation has been simplified and evaluated, and the resulting equations provided in Table 6. As before, substituting VCC – 1.2 and VEE + 1 for VLIM will give the valid
upper and lower common mode extremes respectively.
Following are sample calculations for the case shown in
Figure 14, right-hand side. Note that P9 and M9 are
terminated so row 3 of Table 6 provides the equation:
Noise
Gain
VIN+
VIN–
INPUT CM RANGE
= –60V TO 8.3V
1 P1
450k
2 P3
150k
3 P9
50k
+
4pF
450k
= 11 • (1V) – 2.5 – 9 • 12 = –99.5V
but this exceeds the 60V absolute maximum rating of the
P1, M1 pins, so –60V becomes the de facto negative
common mode limit. Several more examples of high CM
circuits are shown in Figures 15, 16, 17 for various
supplies.
VOUT
REF 5
2.5V
LT1991
4
HIGH NEGATIVE CM VOLTAGE DIFFERENCE AMPLIFIER
IMPLEMENTED WITH LT1991.
RF = 450k, RG = 450k, RT 50k, GAIN = 1
VTERM = VCC = 12V, VREF = 2.5V, VEE = GROUND.
1991 F14
Figure 14. Extending CM Input Range
1991fb
19
LT1991
U
W
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APPLICATIO S I FOR ATIO
3V
8
M9
9
M3
10
M1
VIN –
1
P1
2
P3
3
P9
VIN +
3V
7
VCC
6
VOUT
OUT
REF
5
1.25V
LT1991
VEE
VIN –
VIN +
4
8
M9
9
M3
10
M1
1
P1
2
P3
3
P9
VCM = 0.8V TO 2.35V
7
VCC
LT1991
VEE
VIN –
1
P1
2
P3
3
P9
VIN +
OUT
REF
5
6
VIN –
VOUT
6
VOUT
OUT
REF
5
1.25V
LT1991
VEE
VIN –
VIN +
4
1
P1
2
P3
3
P9
7
VCC
LT1991
VEE
VEE
4
3V
3V
7
VCC
6
VOUT
OUT
REF
5
1.25V
OUT
REF
5
6
VOUT
3V
VCM = –1V TO 0.6V
VDM <–40mV
VCM = 2V TO 3.6V
VDM > 40mV
8
M9
9
M3
10
M1
7
VCC
LT1991
1
P1
2
P3
3
P9
VIN +
4
3V
8
M9
9
M3
10
M1
3V
8
M9
9
M3
10
M1
3V
8
M9
9
M3
10
M1
VIN –
1
P1
2
P3
3
P9
VIN +
4
7
VCC
6
VOUT
OUT
REF
5
1.25V
LT1991
VEE
4
1.25V
VCM = 3.8V TO 7.75V
VCM = 0V TO 4V
3V
8
M9
9
M3
10
M1
VIN –
1
P1
2
P3
3
P9
VIN +
VEE
3V
3V
7
VCC
LT1991
VCM = –5V TO –1.25V
6
VOUT
OUT
REF
5
1.25V
VIN –
VIN +
4
8
M9
9
M3
10
M1
1
P1
2
P3
3
P9
7
VCC
LT1991
VEE
6
VOUT
OUT
REF
5
1.25V
VIN –
3V
8
M9
9
M3
10
M1
1
P1
2
P3
3
P9
VIN +
4
7
VCC
LT1991
VEE
6
VOUT
OUT
REF
5
1.25V
4
1.25V
VCM = –1.5V TO 7.2V
VCM = 9.8V TO 18.55V
3V
VIN –
VIN +
8
M9
9
M3
10
M1
1
P1
2
P3
3
P9
VEE
3V
3V
7
VCC
LT1991
VCM = –17.2V TO –8.45V
6
VOUT
OUT
REF
5
1.25V
4
VIN –
VIN +
8
M9
9
M3
10
M1
1
P1
2
P3
3
P9
7
VCC
LT1991
VEE
6
VOUT
OUT
REF
5
1.25V
4
VIN –
VIN +
3V
8
M9
9
M3
10
M1
1
P1
2
P3
3
P9
7
VCC
LT1991
VEE
6
VOUT
OUT
REF
5
1.25V
4
1.25V
VCM = –2.25V TO 8.95V
VCM = 12.75V TO 23.95V
VCM = –23.2V TO –12V
1991 F15
Figure 15. Common Mode Ranges for Various LT1991 Configurations on VS = 3V, 0V; with Gain = 1
1991fb
20
LT1991
U
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APPLICATIO S I FOR ATIO
5V
8
M9
9
M3
10
M1
VIN –
1
P1
2
P3
3
P9
VIN +
5V
7
VCC
LT1991
OUT
REF
5
2.5V
VEE
6
VIN –
VOUT
VIN +
4
8
M9
9
M3
10
M1
1
P1
2
P3
3
P9
VCM = –0.5V TO 5.1V
7
VCC
LT1991
VEE
VIN –
1
P1
2
P3
3
P9
VIN +
OUT
REF
5
6
VIN –
VOUT
LT1991
OUT
REF
5
2.5V
VEE
6
VIN –
VOUT
VIN +
4
1
P1
2
P3
3
P9
7
VCC
LT1991
VEE
VEE
4
5V
5V
7
VCC
6
OUT
REF
5
VOUT
3V
VCM = –3V TO 2.6V
VDM <–40mV
VCM = 2V TO 7.6V
VDM > 40mV
8
M9
9
M3
10
M1
7
VCC
LT1991
1
P1
2
P3
3
P9
VIN +
4
5V
8
M9
9
M3
10
M1
5V
8
M9
9
M3
10
M1
OUT
REF
5
2.5V
6
VIN –
VOUT
5V
8
M9
9
M3
10
M1
1
P1
2
P3
3
P9
VIN +
4
7
VCC
LT1991
OUT
REF
5
2.5V
VEE
6
VOUT
4
2.5V
VCM = 2.5V TO 16.5V
VCM = –5V TO 9V
5V
8
M9
9
M3
10
M1
VIN –
1
P1
2
P3
3
P9
VIN +
VEE
5V
5V
7
VCC
LT1991
VCM = –12.5V TO 1.5V
OUT
REF
5
2.5V
6
VIN –
VOUT
VIN +
4
8
M9
9
M3
10
M1
1
P1
2
P3
3
P9
7
VCC
LT1991
VEE
OUT
REF
5
2.5V
6
VIN –
VOUT
5V
8
M9
9
M3
10
M1
1
P1
2
P3
3
P9
VIN +
4
7
VCC
LT1991
VEE
OUT
REF
5
2.5V
6
VOUT
4
2.5V
VCM = –14V TO 16.8V
VCM = 8.5V TO 39.3V
5V
VIN –
8
M9
9
M3
10
M1
1
P1
2
P3
3
P9
VIN +
VEE
5V
5V
7
VCC
LT1991
VCM = –36.5V TO –5.7V
OUT
REF
5
2.5V
4
6
VIN –
VOUT
VIN +
8
M9
9
M3
10
M1
1
P1
2
P3
3
P9
7
VCC
LT1991
VEE
OUT
REF
5
2.5V
4
6
VIN –
VOUT
VIN +
5V
8
M9
9
M3
10
M1
1
P1
2
P3
3
P9
7
VCC
LT1991
VEE
OUT
REF
5
2.5V
6
VOUT
4
2.5V
VCM = –18.5V TO 20.7V
VCM = 11.5V TO 50.7V
VCM = –48.5V TO –9.3V
1991 F16
Figure 16. Common Mode Ranges for Various LT1991 Configurations on VS = 5V, 0V; with Gain = 1
1991fb
21
LT1991
U
W
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APPLICATIO S I FOR ATIO
5V
VIN –
VIN +
8
M9
9
M3
10
M1
1
P1
2
P3
3
P9
5V
7
VCC
LT1991
OUT
REF
5
VEE
6
8
M9
9
M3
10
M1
VIN –
VOUT
4
–5V
7
VCC
LT1991
1
P1
2
P3
3
P9
VIN +
VEE
4
5V
VIN –
1
P1
2
P3
3
P9
VIN +
2.5V
LT1991
OUT
REF
5
VEE
6
8
M9
9
M3
10
M1
VIN –
VOUT
4
VEE
LT1991
VEE
OUT
REF
5
6
VIN –
VOUT
1
P1
2
P3
3
P9
VIN +
4
–5V
–5V
1
P1
2
P3
3
P9
LT1991
6
VIN –
VOUT
VEE
1
P1
2
P3
3
P9
4
VEE
5V
–5V
VCM = –56V TO 53.2V
–5V
VOUT
OUT
REF
5
6
VIN –
VOUT
5V
8
M9
9
M3
10
M1
1
P1
2
P3
3
P9
VIN +
4
7
VCC
LT1991
VEE
OUT
REF
5
6
VOUT
4
–5V
VCM = –60V TO –3.2V
5V
7
VCC
LT1991
VEE
6
4
–5V
VIN +
OUT
REF
5
VCM = –35V TO 4V
5V
OUT
REF
5
7
VCC
–5V
LT1991
8
M9
9
M3
10
M1
VOUT
–5V
LT1991
1
P1
2
P3
3
P9
VIN +
VCM = 1V TO 60V
7
VCC
VEE
VIN –
VOUT
7
VCC
5V
VIN +
6
5V
8
M9
9
M3
10
M1
6
5V
–5V
VCM = –44V TO 41.8V
VIN –
4
8
M9
9
M3
10
M1
VCM = –5V TO 34V
7
VCC
8
M9
9
M3
10
M1
OUT
REF
5
OUT
REF
5
VEE
5V
4
–5V
1
P1
2
P3
3
P9
LT1991
1
P1
2
P3
3
P9
VIN +
7
VCC
–5V
VCM = –13V TO 2.6V
VDM <–40mV
7
VCC
5V
VIN +
VIN –
VOUT
–5V
LT1991
1
P1
2
P3
3
P9
VIN +
VCM = –20V TO 19V
VIN –
6
5V
7
VCC
8
M9
9
M3
10
M1
OUT
REF
5
–5V
VCM = –3V TO 12.6V
VDM > 40mV
VCM = –8V TO 7.6V
8
M9
9
M3
10
M1
5V
8
M9
9
M3
10
M1
OUT
REF
5
4
–5V
VCM = 4V TO 60V
6
VIN –
VOUT
VIN +
5V
8
M9
9
M3
10
M1
1
P1
2
P3
3
P9
7
VCC
LT1991
VEE
OUT
REF
5
6
VOUT
4
–5V
VCM = –60V TO –6.8V
1991 F17
Figure 17. Common Mode Ranges for Various LT1991 Configurations on VS = ±5V, with Gain = 1
1991fb
22
LT1991
U
PACKAGE DESCRIPTIO
MS Package
10-Lead Plastic MSOP
(Reference LTC DWG # 05-08-1661)
0.889 ± 0.127
(.035 ± .005)
5.23
(.206)
MIN
3.20 – 3.45
(.126 – .136)
3.00 ± 0.102
(.118 ± .004)
(NOTE 3)
0.50
0.305 ± 0.038
(.0197)
(.0120 ± .0015)
BSC
TYP
RECOMMENDED SOLDER PAD LAYOUT
0.254
(.010)
0.497 ± 0.076
(.0196 ± .003)
REF
10 9 8 7 6
3.00 ± 0.102
(.118 ± .004)
(NOTE 4)
4.90 ± 0.152
(.193 ± .006)
DETAIL “A”
0° – 6° TYP
GAUGE PLANE
1 2 3 4 5
0.53 ± 0.152
(.021 ± .006)
DETAIL “A”
0.86
(.034)
REF
1.10
(.043)
MAX
0.18
(.007)
SEATING
PLANE
0.17 – 0.27
(.007 – .011)
TYP
0.50
(.0197)
BSC
0.127 ± 0.076
(.005 ± .003)
MSOP (MS) 0603
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
1991fb
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
23
LT1991
U
TYPICAL APPLICATIO
Micropower AV = 10 Instrumentation Amplifier
10
9
8
7
VOUT
6
+
VM
1/2 LT6011
–
4pF
–
+
+
VP
LT1991
1/2 LT6011
–
4pF
1
2
3
4
5
1991 TA02
Bidirectional Current Source
Single Supply AC Coupled Amplifier
VS+
8
M9
9
M3
10
M1
VIN –
VIN +
R2*
10k
1
P1
2
P3
3
P9
VS = 2.7V TO 36V
8
M9
9
M3
10
M1
7
1µF
6
LT1991
R1
10k
5
VCC
VIN
4
VS
ILOAD =
–
0.1µF
1
P1
2
P3
3
P9
7
6
LT1991
VOUT
5
4
VIN + – VIN –
10kΩ
GAIN = 12
BW = 7Hz TO 32kHz
*SHORT R2 FOR LOWEST OUTPUT
OFFSET CURRENT. INCLUDE R2 FOR
HIGHEST OUTPUT IMPEDANCE.
1991 TA03
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT1990
High Voltage, Gain Selectable Difference Amplifier
±250V Common Mode, Micropower, Pin Selectable Gain = 1, 10
LT1991
Precision Gain Selectable Difference Amplifier
Micropower, Pin Selectable Gain = –13 to 14
LT1995
High Speed, Gain Selectable Difference Amplifier
30MHz, 1000V/µs, Pin Selectable Gain = –7 to 8
LT6010/LT6011/LT6012
Single/Dual/Quad 135µA 14nV/√Hz Rail-to-Rail Out
Precision Op Amp
Similar Op Amp Performance as
Used in LT1991 Difference Amplifier
LT6013/LT6014
Single/Dual 145µA 8nV/√Hz Rail-to-Rail Out
Precision Op Amp
Lower Noise AV ≥ 5 Version of LT1991 Type Op Amp
LTC6910-X
Programmable Gain Amplifiers
3 Gain Configurations, Rail-to-Rail Input and Output
1991fb
24
Linear Technology Corporation
LT/TP 0105 1K REV B • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2004