ADS1282 AD S1 282 www.ti.com ................................................................................................................................................. SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 High-Resolution Analog-to-Digital Converter FEATURES DESCRIPTION 1 • High Resolution: 130dB SNR (250SPS, High-Resolution Mode) 127dB SNR (250SPS, Low-Power Mode) • High Accuracy: THD: –122dB INL: 0.5ppm • Low-Noise PGA • Two-Channel Input MUX • Inherently-Stable Modulator with Fast Responding Over-Range Detection • Flexible Digital Filter: Sinc + FIR + IIR (Selectable) Linear or Minimum Phase Response Programmable High-Pass Filter Selectable FIR Data Rates: 250SPS to 4kSPS • Filter Bypass Option • Low Power Consumption: High-Resolution Mode: 25mW Low-Power Mode: 17mW Shutdown: 10µW • Offset and Gain Calibration Engine • SYNC Input • Analog Supply: Unipolar (+5V) or Bipolar (±2.5V) • Digital Supply: 1.8V to 3.3V The ADS1282 is an extremely high-performance, single-chip analog-to-digital converter (ADC) with an integrated, low-noise programmable gain amplifier (PGA) and two-channel input multiplexer (MUX). The ADS1282 is suitable for the demanding needs of energy exploration and seismic monitoring environments. 2 The converter uses a fourth-order, inherently stable, delta-sigma (ΔΣ) modulator that provides outstanding noise and linearity performance. The modulator is used either in conjunction with the on-chip digital filter, or can be bypassed for use with post processing filters. The flexible input MUX provides an additional external input for measurement, as well as internal self-test connections. The PGA features outstanding low noise (5nV/√Hz) and high input impedance, allowing easy interfacing to geophones and hydrophones over a wide range of gains. The digital filter provides selectable data rates from 250 to 4000 samples per second (SPS). The high-pass filter (HPF) features an adjustable corner frequency. On-chip gain and offset scaling registers support system calibration. The synchronization input (SYNC) can be used to synchronize the conversions of multiple ADS1282s. The SYNC input also accepts a clock input for continuous alignment of conversions from an external source. APPLICATIONS • • • Two operating modes allow optimization of noise and power. Together, the amplifier, modulator, and filter dissipate 25mW and only 17mW in low-power mode. The ADS1282 is available in a compact TSSOP-28 package and is fully specified from –40°C to +85°C, with a maximum operating range to +125°C. Energy Exploration Seismic Monitoring High-Accuracy Instrumentation AVDD VREFN VREFP DVDD CLK ADS1282 MUX Input 1 Input 2 PGA 4th-Order DS Modulator Programmable Digital Filter Calibration SPI Interface SCLK DOUT DIN DRDY Control SYNC RESET PWDN VCOM Over-Range Modulator Output 3 AVSS DGND 1 2 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. All trademarks are the property of their respective owners. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 2007–2008, Texas Instruments Incorporated ADS1282 SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 ................................................................................................................................................. www.ti.com This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. ORDERING INFORMATION For the most current package and ordering information see the Package Option Addendum at the end of this document, or see the TI web site at www.ti.com. ABSOLUTE MAXIMUM RATINGS (1) Over operating free-air temperature range (unless otherwise noted). ADS1282 UNIT AVDD to AVSS –0.3 to +5.5 V AVSS to DGND –2.8 to +0.3 V DVDD to DGND –0.3 to +3.9 V 100, momentary mA 10, continuous mA AVSS – 0.3 to AVDD + 0.3 V Input current Input current Analog input voltage Digital input voltage to DGND –0.3 to DVDD + 0.3 V +150 °C Operating temperature range –40 to +125 °C Storage temperature range –60 to +150 °C Maximum junction temperature (1) 2 Stresses above these ratings may cause permanent damage. Exposure to absolute maximum conditions for extended periods may degrade device reliability. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those specified is not implied. Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 ADS1282 www.ti.com ................................................................................................................................................. SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 ELECTRICAL CHARACTERISTICS Limit specifications at –40°C to +85°C. Typical specifications at +25°C, AVDD = +2.5V, AVSS = –2.5V, fCLK (1) = 4.096MHz, VREFP = +2.5V, VREFN = –2.5V, DVDD = +3.3V, CAPN – CAPP = 10nF, PGA = 1, High-Resolution Mode, and fDATA = 1000SPS, unless otherwise noted. ADS1282 PARAMETER CONDITIONS MIN TYP MAX UNIT ANALOG INPUTS Full-scale input voltage VIN = (AINP – AINN) Absolute input range AINP or AINN ±VREF/2 × PGA AVSS + 0.7 PGA input voltage noise density V AVDD – 1.25 V 5 Differential input impedance (2) Common-mode input impedance Input bias current Crosstalk f = 31.25Hz MUX on-resistance nV/√Hz 1 GΩ 100 MΩ 1 nA –135 dB 30 Ω PGA OUTPUT (CAPP, CAPN) Absolute output range AVSS + 0.4 AVDD – 0.4 PGA differential output impedance 600 Output impedance tolerance ±10 External bypass capacitance 10 Modulator differential input impedance V Ω % 100 nF High-resolution mode 55 kΩ Low-power mode 110 kΩ AC PERFORMANCE Signal-to-noise ratio (3) SNR High-resolution mode 120 124 Low-power mode 117 121 dB High-resolution mode Total harmonic distortion (4) Spurious-free dynamic range THD PGA = 1...16 –122 –114 PGA = 32 –117 –110 PGA = 64 –115 dB Low-power mode PGA = 1...16 –122 –114 PGA = 32 –113 –108 PGA = 64 –109 SFDR dB dB 123 DC PERFORMANCE Resolution Data rate Integral nonlinearity (INL) fDATA (5) No missing codes 31 FIR filter mode 250 4000 SPS Sinc filter mode 8000 128,000 SPS 0.00005 0.0004 % FSR (6) 50 200 µV Differential input Offset error Offset error after calibration (7) Shorted input Offset drift Gain error (8) 1 µV 0.02 µV/°C High-resolution mode –1.5 –1.0 –0.5 % Low-power mode –1 –0.5 0 % Gain error after calibration (7) (1) (2) (3) (4) (5) (6) (7) (8) Bits 0.0002 % fCLK = system clock. Input impedance is improved by disabling input chopping (CHOP bit = 0). VIN = 20mVDC/PGA, see Table 1. VIN = 31.25Hz, –0.5dBFS. Best-fit method. FSR: Full-scale range = ±VREF/2 × PGA. Calibration accuracy is on the level of noise reduced by 4 (calibration averages 16 readings). The PGA output impedance and the modulator input impedance results in –1% systematic gain error (high-resolution mode) and –0.5% error (low-power mode). Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 3 ADS1282 SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 ................................................................................................................................................. www.ti.com ELECTRICAL CHARACTERISTICS (continued) Limit specifications at –40°C to +85°C. Typical specifications at +25°C, AVDD = +2.5V, AVSS = –2.5V, fCLK = 4.096MHz, VREFP = +2.5V, VREFN = –2.5V, DVDD = +3.3V, CAPN – CAPP = 10nF, PGA = 1, High-Resolution Mode, and fDATA = 1000SPS, unless otherwise noted. ADS1282 PARAMETER CONDITIONS MIN TYP MAX UNIT DC PERFORMANCE (continued) Gain drift PGA = 1 2 PGA = 16 9 Gain matching (9) 0.3 fCM = 60Hz (10) Common-mode rejection Power-supply rejection AVDD, AVSS DVDD fPS = 60Hz (10) 95 ppm/°C ppm/°C 0.8 110 80 90 90 115 0.5 5 % dB dB VOLTAGE REFERENCE INPUTS Reference input voltage (VREF = VREFP – VREFN) (AVDD – AVSS) + 0.2 V V Negative reference input VREFN AVSS – 0.1 VREFP – 0.5 Positive reference input VREFP VREFN + 0.5 AVDD + 0.1 Reference input impedance V High-resolution mode 85 kΩ Low-power mode 170 kΩ DIGITAL FILTER RESPONSE Passband ripple ±0.003 Passband (–0.01dB) 0.375 × fDATA Bandwidth (–3dB) 0.413 × fDATA High-pass filter corner 0.1 Stop band attenuation (11) 135 Hz Hz 10 0.500 × fDATA Settling time (latency) Hz dB Stop band Group delay dB Minimum phase filter 5/fDATA Linear phase filter 31/fDATA Minimum phase filter 10/fDATA Linear phase filter 62/fDATA Hz s s DIGITAL INPUT/OUTPUT VIH 0.8 × DVDD DVDD V VIL DGND 0.2 × DVDD V VOH IOH = 1mA VOL IOL = 1mA 0.2 × DVDD V 0 < VDIGITAL IN < DVDD ±10 µA 4.096 MHz fCLK/2 MHz Input leakage Clock input Serial clock rate fCLK 0.8 × DVDD 1 fSCLK V (9) Gain match relative to PGA = 1. (10) fCM is the input common-mode frequency. fPS is the power-supply frequency. (11) Input frequencies in the range of NfCLK/512 ± fDATA/2 (N = 1, 2, 3...) can mix with the modulator chopping clock. In these frequency ranges intermodulation = 120dB, typ. 4 Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 ADS1282 www.ti.com ................................................................................................................................................. SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 ELECTRICAL CHARACTERISTICS (continued) Limit specifications at –40°C to +85°C. Typical specifications at +25°C, AVDD = +2.5V, AVSS = –2.5V, fCLK = 4.096MHz, VREFP = +2.5V, VREFN = –2.5V, DVDD = +3.3V, CAPN – CAPP = 10nF, PGA = 1, High-Resolution Mode, and fDATA = 1000SPS, unless otherwise noted. ADS1282 PARAMETER CONDITIONS MIN TYP MAX UNIT POWER SUPPLY AVSS –2.6 0 V AVDD AVSS + 4.75 AVSS + 5.25 V DVDD 1.65 3.6 V AVDD, AVSS current DVDD current Power dissipation High-resolution mode 4.5 6.5 |mA| Low-power mode 3 4.2 |mA| Standby mode 1 15 |µA| Power-down mode 1 15 |µA| All modes 0.6 0.8 mA Modulator mode 0.1 Standby mode 25 50 µA Power-down mode (12) 1 15 µA High-reolution mode 25 35 mW Low-power mode 17 25 mW Standby mode 90 250 µW Power-down mode 10 125 µW mA (12) CLK input stopped. TIMING DIAGRAM tSCDL tSCLK tSPWH SCLK tDIST tSPWL tSCDL DIN tDIHD tDOHD DOUT tDOPD TIMING REQUIREMENTS At TA = –40°C to +85°C and DVDD = 1.65V to 3.6V, unless otherwise noted. PARAMETER DESCRIPTION tSCLK tSPWH, (1) (2) MIN MAX UNITS 2 16 1/fCLK SCLK pulse width, high and low (1) 0.8 10 1/fCLK SCLK period L tDIST DIN valid to SCLK rising edge: setup time 50 tDIHD Valid DIN to SCLK rising edge: hold time 50 ns tDOPD SCLK falling edge to valid new DOUT: propagation delay (2) tDOHD SCLK falling edge to DOUT invalid: hold time 0 ns tSCDL Final SCLK rising edge of command to first SCLK rising edge for register read/write data. (Also between consecutive commands.) 24 1/fCLK ns 100 ns Holding SCLK low for 64 DRDY falling edges resets the serial interface. Load on DOUT = 20pF || 100kΩ. Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 5 ADS1282 SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 ................................................................................................................................................. www.ti.com DEVICE INFORMATION TSSOP-28 Top View CLK 1 28 BYPAS SCLK 2 27 DGND DRDY 3 26 DVDD DOUT 4 25 DGND DIN 5 24 RESET DGND 6 23 PWDN MCLK 7 22 VREFP M1 8 21 VREFN ADS1282 M0 9 20 AVSS SYNC 10 19 AVDD MFLAG 11 18 AINN1 DGND 12 17 AINP1 CAPN 13 16 AINN2 CAPP 14 15 AINP2 TERMINAL FUNCTIONS 6 NAME NO. FUNCTION DESCRIPTION Master clock input CLK 1 Digital input SCLK 2 Digital input Serial clock input DRDY 3 Digital output Data ready output: read data on falling edge DOUT 4 Digital output Serial data output DIN 5 Digital input Serial data input MCLK 7 Digital I/O Modulator clock output; if in modulator mode: MCLK: Modulator clock output Otherwise, the pin is an unused input (must be tied). M1 8 Digital I/O Modulator data output 1; if in modulator mode: M1: Modulator data output 1 Otherwise, the pin is an unused input (must be tied). M0 9 Digital I/O Modulator data output 0; if in modulator mode: M0: Modulator data output 0 Otherwise, the pin is an unused input (must be tied). SYNC 10 Digital input Synchronize input MFLAG 11 Digital output Modulator Over-Range flag: 0 = normal, 1 = modulator over-range DGND 6, 12, 25, 27 Digital ground Digital ground, pin 12 is the key ground point CAPN 13 Analog PGA outputs: Connect 10nF capacitor from CAPP to CAPN PGA outputs: Connect 10nF capacitor from CAPP to CAPN CAPP 14 Analog AINP2 15 Analog input Positive analog input 2 AINN2 16 Analog input Negative analog input 2 AINP1 17 Analog input Positive analog input 1 AINN1 18 Analog input Negative analog input 1 AVDD 19 Analog supply Positive analog power supply AVSS 20 Analog supply Negative analog power supply VREFN 21 Analog input Negative reference input VREFP 22 Analog input Positive reference input PWDN 23 Digital input Power-down input, active low RESET 24 Digital input Reset input, active low DVDD 26 Digital supply Digital power supply: +1.8V to +3.3V BYPAS 28 Analog Sub-regulator output: Connect 1µF capacitor to DGND Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 ADS1282 www.ti.com ................................................................................................................................................. SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 TYPICAL CHARACTERISTICS At +25°C, AVDD = +2.5V, AVSS = –2.5V, fCLK = 4.096MHz, VREFP = +2.5V, VREFN = –2.5V, DVDD = +3.3V, CAPN – CAPP = 10nF, PGA = 1, High-Resolution Mode, and fDATA = 1000SPS, unless otherwise noted. OUTPUT SPECTRUM 8192-Point FFT VIN = -0.5dBFS, 31.25Hz High-Resolution Mode PGA = 1 THD = -124.0dB -20 -40 Amplitude (dB) OUTPUT SPECTRUM -60 -80 -100 -120 0 8192-Point FFT VIN = -0.5dBFS, 31.25Hz Low-Power Mode PGA = 1 THD = -123.0dB -20 -40 Amplitude (dB) 0 -60 -80 -100 -120 -140 -140 -160 -160 -180 -180 0 50 100 150 200 250 300 350 400 450 Frequency (Hz) 500 0 50 Figure 1. -60 -80 -100 -120 0 8192-Point FFT VIN = -20dBFS, 31.25Hz Low-Power Mode PGA = 1 THD = -117.0dB -20 -40 Amplitude (dB) -40 Amplitude (dB) OUTPUT SPECTRUM 8192-Point FFT VIN = -20dBFS, 31.25Hz High-Resolution Mode PGA = 1 THD = -120.1dB -20 -60 -80 -100 -120 -140 -140 -160 -160 -180 -180 0 50 100 150 200 250 300 350 400 450 Frequency (Hz) 500 0 50 Figure 3. -60 -80 -100 -120 0 -40 -60 -80 -100 -120 -140 -140 -160 -160 -180 8192-Point FFT VIN = -0.5dBFS, 31.25Hz Low-Power Mode PGA = 16 THD = -121.4dB -20 Amplitude (dB) -40 500 OUTPUT SPECTRUM 8192-Point FFT VIN = -0.5dBFS, 31.25Hz High-Resolution Mode PGA = 16 THD = -122.4dB -20 100 150 200 250 300 350 400 450 Frequency (Hz) Figure 4. OUTPUT SPECTRUM 0 Amplitude (dB) 500 Figure 2. OUTPUT SPECTRUM 0 100 150 200 250 300 350 400 450 Frequency (Hz) -180 0 50 100 150 200 250 300 350 400 450 Frequency (Hz) 500 0 Figure 5. 50 100 150 200 250 300 350 400 450 Frequency (Hz) 500 Figure 6. Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 7 ADS1282 SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 ................................................................................................................................................. www.ti.com TYPICAL CHARACTERISTICS (continued) At +25°C, AVDD = +2.5V, AVSS = –2.5V, fCLK = 4.096MHz, VREFP = +2.5V, VREFN = –2.5V, DVDD = +3.3V, CAPN – CAPP = 10nF, PGA = 1, High-Resolution Mode, and fDATA = 1000SPS, unless otherwise noted. OUTPUT SPECTRUM 0 -20 -60 -80 -100 -120 8192-Point FFT Shorted Input Low-Power Mode SNR = 121.0dB -20 -40 Amplitude (dB) -40 Amplitude (dB) OUTPUT SPECTRUM 0 8192-Point FFT Shorted Input High-Resolution Mode SNR = 124.0dB -60 -80 -100 -120 -140 -140 -160 -160 -180 -180 0 50 100 150 200 250 300 350 400 450 Frequency (Hz) 500 0 50 100 150 200 250 300 350 400 450 Frequency (Hz) Figure 7. Figure 8. OUTPUT SPECTRUM 0 -60 -80 -100 -120 8192-Point FFT 20mVDC Low-Power Mode SNR = 121.3dB -20 -40 Amplitude (dB) -40 Amplitude (dB) OUTPUT SPECTRUM 0 8192-Point FFT 20mVDC High-Resolution Mode SNR = 124.2dB -20 -60 -80 -100 -120 -140 -140 -160 -160 -180 -180 0 50 100 150 200 250 300 350 400 450 Frequency (Hz) 500 0 50 100 150 200 250 300 350 400 450 Frequency (Hz) Figure 9. SNR vs TEMPERATURE 126 -105 High-Resolution Mode, PGA = 1 Low-Power Mode, PGA = 1 High-Resolution Mode, PGA = 8 Low-Power Mode, PGA = 8 -115 -120 -125 Signal-to-Noise Ratio (dB) THD Limited by Signal Generator -110 500 Figure 10. THD vs INPUT FREQUENCY -100 Total Harmonic Distortion (dB) 500 VIN = 20mVDC 125 High-Resolution Mode 124 123 122 Low-Power Mode 121 VIN = -0.5dBFS -130 10 20 30 40 50 60 70 Input Frequency (Hz) 80 90 100 120 -55 Figure 11. 8 -35 -15 5 25 45 65 Temperature (°C) 85 105 125 Figure 12. Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 ADS1282 www.ti.com ................................................................................................................................................. SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 TYPICAL CHARACTERISTICS (continued) At +25°C, AVDD = +2.5V, AVSS = –2.5V, fCLK = 4.096MHz, VREFP = +2.5V, VREFN = –2.5V, DVDD = +3.3V, CAPN – CAPP = 10nF, PGA = 1, High-Resolution Mode, and fDATA = 1000SPS, unless otherwise noted. THD vs TEMPERATURE SNR vs REFERENCE VOLTAGE 130 PGA = 8 VIN = 31.25Hz, -0.5dBFS High-Resolution Mode, PGA = 1 Signal-to-Noise Ratio (dB) Total Harmonic Distortion (dB) -110 -115 Low-Power Mode -120 -125 High-Resolution Mode 125 High-Resolution Mode, PGA = 8 120 115 Low-Power Mode, PGA = 1 110 105 Low-Power Mode, PGA = 8 -130 -55 100 -35 -15 5 25 45 65 Temperature (°C) 85 105 0 125 1 2 Figure 13. 3 VREF (V) 4 5 5.5 Figure 14. THD vs REFERENCE VOLTAGE SNR vs CLOCK FREQUENCY 125 -110 Signal-to-Noise Ratio (dB) Total Harmonic Distortion (dB) High-Resolution Mode, PGA = 8 -115 Low-Power Mode, PGA = 8 -120 -125 Low-Power Mode, PGA = 1 124 High-Resolution Mode 123 122 121 Low-Power Mode 120 VIN = 20mVDC Data Rate = fCLK/4096 High-Resolution Mode, PGA = 1 119 -130 0 1 2 3 4 5 0.5 6 1.0 1.5 VREF (V) Figure 15. 4.0 4.5 -120 High-Resolution Mode 130 Common-Mode Rejection (dB) Total Harmonic Distortion (dB) Low-Power Mode -125 3.5 CMR vs INPUT FREQUENCY PGA = 8 VIN = 31.25Hz, -0.5dBFS Data Rate = fCLK/4096 -115 2.5 3.0 fCLK (MHz) Figure 16. THD vs CLOCK FREQUENCY -110 2.0 120 110 100 90 Low-Power Mode 80 High-Resolution Mode -130 70 0.5 1.0 1.5 2.0 2.5 3.0 fCLK (MHz) 3.5 4.0 4.5 10 Figure 17. 100 1k 10k Input Frequency (Hz) 100k 1M Figure 18. Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 9 ADS1282 SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 ................................................................................................................................................. www.ti.com TYPICAL CHARACTERISTICS (continued) At +25°C, AVDD = +2.5V, AVSS = –2.5V, fCLK = 4.096MHz, VREFP = +2.5V, VREFN = –2.5V, DVDD = +3.3V, CAPN – CAPP = 10nF, PGA = 1, High-Resolution Mode, and fDATA = 1000SPS, unless otherwise noted. POWER-SUPPLY REJECTION vs FREQUENCY INL vs INPUT AMPLITUDE 4 140 AVDD 3 Integral Nonlinearity (ppm) Power-Supply Rejection (dB) DVDD 120 100 80 AVSS 60 40 20 2 PGA = 2 1 0 -1 PGA = 8 -3 0 10 100 1k 10k 100k Power-Supply Frequency (Hz) -4 -100 1M -50 -25 0 25 50 Input Amplitude (% Full-Scale) -75 Figure 19. 75 100 Figure 20. INL vs TEMPERATURE CROSSTALK OUTPUT SPECTRUM 4 0 Shorted Input 8192-Point FFT Adjacent Channel VIN = -0.5dBFS, 31.25Hz -20 -40 3 Amplitude (dB) Integral Nonlinearity (ppm) PGA = 32 -2 2 1 -60 -80 -100 -120 -140 -160 0 -55 -180 -35 -15 5 25 45 65 Temperature (°C) 85 105 125 0 100 150 200 250 300 350 400 450 Frequency (Hz) 50 Figure 21. 500 Figure 22. POWER vs TEMPERATURE POWER vs CLOCK FREQUENCY 30 30 High-Resolution Mode High-Resolution Mode 25 Power (mW) Power (mW) 25 20 Low-Power Mode 20 Low-Power Mode 15 10 15 5 10 -55 0 -35 -15 5 25 45 65 Temperature (°C) 85 105 125 1.0 Figure 23. 10 1.5 2.0 2.5 3.0 fCLK (MHz) 3.5 4.0 4.5 Figure 24. Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 ADS1282 www.ti.com ................................................................................................................................................. SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 TYPICAL CHARACTERISTICS (continued) At +25°C, AVDD = +2.5V, AVSS = –2.5V, fCLK = 4.096MHz, VREFP = +2.5V, VREFN = –2.5V, DVDD = +3.3V, CAPN – CAPP = 10nF, PGA = 1, High-Resolution Mode, and fDATA = 1000SPS, unless otherwise noted. OFFSET HISTOGRAM GAIN ERROR HISTOGRAM 30 10 25 Units 25 Units 25 8 High-Resolution Mode 20 Occurrences Occurrences PGA = 8 15 10 6 4 PGA = 1 Low-Power Mode 2 5 0 -100 -80 -60 -40 -20 0 20 Offset (mV) 40 60 80 0 -1.2 100 -1.1 -1.0 -0.9 Figure 25. 80 Occurrences 30 60 50 PGA = 8, 64 30 20 PGA =1 -15 -14 -13 -12 -11 -10 -9 -8 -7 -6 -5 -4 -3 -2 -1 0 1 2 3 4 5 0.10 0.08 0.06 0.04 0.02 0 0 0 -0.02 10 -0.04 PGA = 16 40 10 -0.08 PGA = 1, 2, 4 PGA = 32 40 -0.10 -0.3 70 50 20 25 Units Based on +20°C Intervals Over the Range of -40°C to +85°C 80 PGA = 8 60 -0.06 Occurrences 70 -0.4 GAIN DRIFT HISTOGRAM 90 25 Units Based on +20°C Intervals Over the Range of -40°C to +85°C -0.5 Figure 26. OFFSET DRIFT HISTOGRAM 90 -0.8 -0.7 -0.6 Gain Error (%) Gain Drift (ppm/°C) Offset Drift (mV/°C) Figure 27. Figure 28. GAIN MATCH HISTOGRAM 8 Worst-Case Gain Match Relative PGA = 1 (25 Units) Occurrences 6 4 2 0.02 0.06 0.10 0.14 0.18 0.22 0.26 0.30 0.34 0.38 0.42 0.46 0.50 0 Gain Error (%) Figure 29. Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 11 ADS1282 SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 ................................................................................................................................................. www.ti.com OVERVIEW The ADS1282 is a high-performance analog-to-digital converter (ADC) intended for energy exploration, seismic monitoring, chomatography, and other exacting applications. The converter provides 24- or 32-bit output data in data rates from 250SPS to 4000SPS. Figure 30 shows the block diagram of the ADS1282. The two-channel input MUX allows five configurations: Input 1; Input 2; Input 1 and Input 2 shorted together; shorted with 400Ω test; and common-mode test. The input MUX is followed by a continuous time PGA, featuring very low noise of 5nV/√Hz. The PGA is controlled by register settings, allowing gains of 1 to 64. The inherently-stable, fourth-order, delta-sigma modulator measures the differential input signal VIN = (AINP – AINN) PGA against the differential reference VREF = (VREFP – VREFN). A digital output (MFLAG) indicates that the modulator is in overload as a result of an overdrive condition. The modulator output is available directly on the MCLK, M0, and M1 output pins. The modulator connects to an on-chip digital filter that provides the output code readings. The SYNC input resets the operation of both the digital filter and the modulator, allowing synchronization conversions of multiple ADS1282 devices to an external event. The SYNC input supports a continuously-toggled input mode that accepts an external data frame clock locked to the conversion rate. The RESET input resets the register settings and also restarts the conversion process. The PWDN input sets the device into a micro-power state. Note that register settings are not retained in PWDN mode. Use the STANDBY command in its place if it is desired to retain register settings (the quiescent current in the Standby mode is slightly higher). Noise-immune Schmitt-trigger and clock-qualified inputs (RESET and SYNC) provide increased reliability in high-noise environments. The serial interface is used to read conversion data, in addition to reading from and writing to the configuration registers. VREFP VREFN AVDD CAPN CAPP The digital filter consists of a variable decimation rate, fifth-order sinc filter followed by a variable phase, decimate-by-32, finite-impulse response (FIR) low-pass filter with programmable phase, and then by an adjustable high-pass filter for dc removal of the output reading. The output of the digital filter can be taken from the sinc, the FIR low-pass, or the infinite impulse response (IIR) high-pass sections. Gain and offset registers scale the digital filter output to produce the final code value. The scaling feature can be used for calibration and sensor gain matching. The output data word is provided as either a 24-bit word or a full 32-bit word, allowing complete utilization of the inherently high resolution. ADS1282 AINP2 AINN2 AINP1 AINN1 BYPAS CLK +1.8V (Digital core) DVDD LDO MUX 300W 400W PGA 4th-Order DS Modulator 300W 400W Programmable Digital Filter Calibration Serial Interface Over-Range Detection SYNC Control RESET PWDN AVDD + AVSS 2 AVSS DRDY SCLK DIN DOUT MFLAG MCLK M0 DGND M1 Figure 30. ADS1282 Block Diagram 12 Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 ADS1282 www.ti.com ................................................................................................................................................. SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 The device features unipolar and bipolar analog power supplies (AVDD and AVSS, respectively) for input range flexibility and a digital supply accepting 1.8V to 3.3V. The analog supplies may be set to +5V to accept unipolar signals (with input offset) or set lower in the range of ±2.5V to accept true bipolar input signals (ground referenced). INPUT-REFERRED NOISE An internal sub-regulator is used to supply the digital core from DVDD. The BYPAS pin (pin 28) is the sub-regulator output and requires a 1µF capacitor for noise reduction. BYPAS should not be used to drive external circuitry. where: NOISE PERFORMANCE IDLE TONES The ADS1282 offers outstanding noise performance (SNR). SNR depends on the data rate, the PGA setting, and the mode (high-resolution or low-power). As the bandwidth is reduced by decreasing the data rate, the SNR improves correspondingly. Similarly, as the PGA gain is increased, the SNR decreases. In low-power mode, the SNR decreases 3dB for each setting. Table 1 summarizes the noise performance versus data rate, PGA setting, and mode. The ADS1282 modulator incorporates an internal dither signal that randomizes the idle tone energy. Low-level idle tones may still be present, typically –137dB below full-scale. The low-level idle tones can be shifted out of the passband with an external offset = 20mV/PGA. See the Application Information section for the recommended offset circuit. The input-referred noise is related to SNR by Equation 1: FSRRMS SNR = 20log NRMS (1) FSRRMS = Full-scale range RMS = VREF/2 × √2 × PGA NRMS = Noise RMS (input-referred) OPERATING MODE For applications where minimal power consumption is important, the low-power mode can be selected (register bit MODE = 0). In low-power mode, the power is reduced to 17mW (from 25mW) and SNR decreases an average of 3dB. The default mode is high-resolution. Table 1. Signal-to-Noise Ratio (dB) (1) (1) PGA (High-Resolution Mode) PGA (Low-Power Mode) DATA RATE (SPS) 1 2 4 8 16 32 64 1 2 4 8 16 32 64 250 130 130 129 128 125 119 114 127 127 126 125 122 116 111 500 127 127 126 125 122 116 111 124 124 123 122 119 113 108 1000 124 124 123 122 119 113 108 121 121 120 119 116 110 105 2000 121 121 120 119 116 111 106 118 118 117 116 113 108 103 4000 118 118 117 116 113 108 103 115 115 114 113 110 105 100 VIN = 20mVDC/PGA. Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 13 ADS1282 SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 ................................................................................................................................................. www.ti.com Table 2. Multiplexer Modes ANALOG INPUTS AND MULTIPLEXER A diagram of the input multiplexer is shown in Figure 31. MUX[2:0] ESD diodes protect the multiplexer inputs. If either input is taken below AVSS – 0.3V or above AVDD + 0.3V, the ESD protection diodes may turn on. If these conditions are possible, external Schottky clamp diodes and/or series resistors may be required to limit the input current to safe values (see the Absolute Maximum Ratings table). Also, overdriving one unused input may affect the conversions of the other input. If overdriven inputs are possible, it is recommended to clamp the signal with external Schottky diodes. AVDD S1 ESD Diodes S2 AINP2 400W 2 To PGA 400W S4 S5 AINN1 ESD Diodes AINN2 S1, S5 001 S2, S6 AINP2 and AINN2 connected to preamplifier 010 S3, S4 Preamplifier inputs shorted together through 400Ω internal resistors 011 S1, S5, S2, S6 AINP1, AINN1 and AINP2, AINN2 connected together and to the preamplifier 100 S6, S7 External short, preamplifier inputs shorted to AINN2 (common-mode test) 0 S3 AVDD + AVSS AVDD 000 (+) S7 AVSS DESCRIPTION AINP1 and AINN1 connected to preamplifer The typical on-resistance (RON) of the multiplexer switch is 30Ω. When the multiplexer is used to drive an external load on one input by a signal generator on the other input, on-resistance and on-resistance amplitude dependency can lead to measurement errors. Figure 32 shows THD versus load resistance and amplitude. Note that THD improves with high-impedance loads and with lower amplitude drive signals. The data are measured with the circuit from Figure 33 with MUX[2:0] = 011. (-) S6 Total Harmonic Distortion (dB) AINP1 SWITCHES PGA = 1 PGA = 2 PGA = 4 PGA = 8 PGA = 16 PGA = 32 PGA = 64 -20 -40 -60 -80 -100 -120 -140 0.1k 1k 10k 100k 1M 10M RLOAD (W) AVSS Figure 31. Analog Inputs and Multiplexer Figure 32. THD versus External Load and Signal Magnitude (PGA) (see Figure 33) The specified input operating range of the PGA is shown in Equation 2: AVSS + 0.7V < (AINN or AINP) < AVDD - 1.25V (2) 500W Absolute input levels (input signal level and common-mode level) should be maintained within these limits for best operation. 500W The multiplexer connects one of the two external differential inputs to the preamplifier inputs, in addition to internal connections for various self-test modes. Table 2 summarizes the multiplexer configurations for Figure 31. RLOAD 14 ADS1282 Input 1 Input 2 Figure 33. Driving an External Load Through the MUX Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 ADS1282 www.ti.com ................................................................................................................................................. SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 PGA (Programmable Gain Amplifier) AVDD Referring to Figure 34, amplifiers A1 and A2 are chopped to remove the offset, offset drift, and the 1/f noise. Chopping moves the effects to fCLK/128 (8kHz), which is safely out of the passband. Chopping can be disabled by setting the CHOP register bit = 0. With chopping disabled, the impedance of the PGA increases substantially (>> 1GΩ). As shown in Figure 35, chopping maintains flat noise density; if chopping is disabled, however, it results in a rising 1/f noise profile. The PGA has programmable gains from 1 to 64. Table 3 shows the register bit setting for the PGA and resulting full-scale differential range. The specified output operating range of the PGA is shown in Equation 4: AVSS + 0.4V < (CAPN or CAPP) < AVDD - 0.4V (4) PGA output levels (signal plus common-mode) should be maintained within these limits for best operation. MUX (+) Gain Control PGA[2:0] Bits GAIN 000 1 ±2.5 001 2 ±1.25 010 4 ±0.625 011 8 ±0.312 100 16 ±0.156 101 32 ±0.078 110 64 ±0.039 10nF (1) 300W CAPN A2 (55kW, typ ) Modulator Effective Impedance MUX (-) Chopping Control CHOP Bit AVSS (1) Low-power mode = 110kΩ. Figure 34. PGA Block Diagram 100 PGA CHOP Off 10 PGA CHOP On 1 DIFFERENTIAL INPUT RANGE (V)(1) PGA[2:0] CAPP CHOP 10 Table 3. PGA Gain Settings 300W A1 PGA Noise (nV/ÖHz) The PGA of the ADS1282 is a low-noise, continuous-time, differential-in/differential-out CMOS amplifier. The gain is programmable from 1 to 64, set by register bits, PGA[2:0]. The PGA differentially drives the modulator through 300Ω internal resistors. A COG capacitor (10nF typical) must be connected to CAPP and CAPN to filter modulator sampling glitches. The external capacitor also serves as an anti-alias filter. The corner frequency is given in Equation 3: 1 fP = 6.3 ´ 600 ´ C (3) 100 Frequency (Hz) 1k Figure 35. PGA Noise (1) VREF = 5V. Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 15 ADS1282 SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 ................................................................................................................................................. www.ti.com The ADC block of the ADS1282 is composed of two sections: a high-accuracy modulator and a programmable digital filter. MODULATOR The high-performance modulator is an inherently-stable, fourth-order, ΔΣ, 2 + 2 pipelined structure, as Figure 36 shows. It shifts the quantization noise to a higher frequency (out of the passband) where digital filtering can easily remove it. The modulator can be filtered either by the on-chip digital filter or by use of post-processing filters. fCLK/4 MCLK passband. The noise moves out of the passband and appears at the chopping frequency (fCLK/512 = 8kHz). The component at 5.8kHz is the tone frequency, shifted out of band by an external 20mV/PGA offset. The frequency of the tone is proportional to the applied dc input and is given by PGA × VIN/0.003 (in kHz). 0 1Hz Resolution VIN = 20mVDC -20 -40 Magnitude (dB) ADC -60 -80 -100 -120 -140 Analog Input (VIN) 2nd-Order DS 1st-Stage M0 -160 -180 1 10 100 1k 10k 100k Frequency (Hz) 2nd-Order DS 2nd-Stage M1 Figure 37. Modulator Output Spectrum 4th-Order Modulator MODULATOR OVER-RANGE Figure 36. Fourth-Order Modulator The modulator first stage converts the analog input voltage into a pulse-code modulated (PCM) stream. When the level of differential analog input (AINP – AINN) is near one-half the level of the reference voltage 1/2 × (VREFP – VREFN), the ‘1’ density of the PCM data stream is at its highest. When the level of the differential analog input is near zero, the PCM ‘0’ and ‘1’ densities are nearly equal. At the two extremes of the analog input levels (+FS and –FS), the ‘1’ density of the PCM streams is approximately +90% and +10%, respectively. The modulator second stage produces a '1' density data stream designed to cancel the quantization noise of the first stage. The data streams of the two stages are then combined before the digital filter stage, as shown in Equation 5. Y[n] = 3M0[n - 2] - 6M0[n - 3] + 4M0[n - 4] + 9(M1[n] - 2M1[n - 1] + M1[n - 2]) (5) M0[n] represents the most recent first-stage output while M0[n – 1] is the previous first-stage output. When the modulator output is enabled, the digital filter shuts down to save power. The modulator is optimized for input signals within a 4kHz passband. As Figure 37 shows, the noise shaping of the modulator results in a sharp increase in noise above 6kHz. The modulator has a chopped input structure that further reduces noise within the 16 The ADS1282 modulator is inherently stable, and therefore, has predictable recovery behavior resulting from an input overdrive condition. The modulator does not exhibit self-resetting behavior, which often results in an unstable output data stream. The ADS1282 modulator outputs a 1s density data stream at 90% duty cycle with the positive full-scale input signal applied (10% duty cycle with the negative full-scale signal). If the input is overdriven past 90% modulation, but below 100% modulation (10% and 0% for negative overdrive, respectively), the modulator remains stable and continues to output the 1s density data stream. The digital filter may or may not clip the output codes to +FS or –FS, depending on the duration of the overdrive. When the input returns to the normal range from a long duration overdrive (worst case), the modulator returns immediately to the normal range, but the group delay of the digital filter delays the return of the conversion result to within the linear range (31 readings for linear phase FIR). 31 additional readings (62 total) are required for completely settled data. If the inputs are sufficiently overdriven to drive the modulator to full duty cycle (that is, all 1s, all 0s, or ±110%FSR), the modulator enters a stable saturated state. The digital output code may clip to +FS or –FS, again depending on the duration. A small duration overdrive may not always clip the output code. When Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 ADS1282 www.ti.com ................................................................................................................................................. SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 the input returns to the normal range, the modulator requires up to 12 modulator clock cycles (fMOD) to exit saturation and return to the linear region. The digital filter requires an additional 62 conversions for fully settled data (linear phase FIR). In the extreme case of over-range, either input is overdriven, exceeding the voltage of either analog supply voltage plus an internal ESD diode drop. The internal diodes begin to conduct and the signal on the input is clipped. When the input overdrive is removed, the diodes recover quickly. Keep in mind that the input current must be limited to 100mA peak or 10mA continuous if an overvoltage condition is possible. MODULATOR OVER-RANGE DETECTION (MFLAG) The ADS1282 has a fast-responding over-range detection that indicates when the differential input exceeds approximately 100% over-range. The threshold tolerance is ±2.5%.The MFLAG output asserts high when in an over-range condition. As Figure 38 and Figure 39 illustrate, the absolute differential input is compared to 100% of range. The output of the comparator is sampled at the rate of fMOD/2, yielding the MFLAG output. The minimum MFLAG pulse width is fMOD/2. AINP MODULATOR INPUT IMPEDANCE The modulator samples the buffered input voltage with an internal capacitor to perform conversions. The charging of the input sampling capacitor draws a transient current from the PGA output. The average value of the current can be used to calculate an effective input impedance of REFF = 1/(fMOD × CS). Where: fMOD = Modulator sample frequency (High-resolution mode = CLK/4) (Low-power mode = CLK/8) CS = Input sampling capacitor (17pF, typ) The resulting modulator input impedance for CLK = 4.096MHz is 55kΩ (110kΩ low-power mode). The modulator input impedance and the PGA output resistors result in a systematic gain error of –1% (–0.5% in low-power mode). CS can vary ±20% or more over production lots, affecting the gain error. å IABSI P 100% FS AINN Q MFLAG Pin fMOD/2 Figure 38. Modulator Over-Range Block Diagram +100% (AINP - AINN) 0% Time -100% MFLAG Pin Figure 39. Modulator Over-Range Flag Operation Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 17 ADS1282 SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 ................................................................................................................................................. www.ti.com MODULATOR OUTPUT MODE The modulator digital stream output is accessible directly, bypassing and disabling the internal digital filter. The modulator output mode is activated by setting the CONFIG0 register bits FILTR[1:0] = 00. Pins M0 and M1 then become the modulator data outputs and the MCLK becomes the modulator clock output. When not in the modulator mode, these pins are inputs and must be tied. The modulator output is composed of three signals: one output for the modulator clock (MCLK) and two outputs for the modulator data (M0 and M1). The modulator clock output rate is fMOD (fCLK/4 for high-resolution mode, fCLK/8 for low-power mode). The SYNC input resets the MCLK phase, as shown in Figure 40. The SYNC input is latched on the rising edge of CLK. The MCLK resets and the next rising edge of MCLK occurs five CLK periods later. The modulator output data are two bits wide, which must be merged together before being filtered. Use the time domain equation of Equation 5 to merge the data outputs. VOLTAGE REFERENCE INPUTS (VREFP, VREFN) The voltage reference for the ADS1282 is the differential voltage between VREFP and VREFN: VREF = VREFP – VREFN. The reference inputs use a structure similar to that of the analog inputs with the circuitry of the reference inputs shown in Figure 41. The average load presented by the switched capacitor reference input can be modeled with an effective differential impedance of REFF = tSAMPLE/CIN (tSAMPLE = 1/fMOD). Note that the effective impedance of the reference inputs loads the external reference. AVDD ESD Diodes VREFP VREFN ESD Diodes tCSHD CLK tCMD tSYMD MCLK REFF = 1 fMOD ´ CX AVSS tSCSU SYNC REFF = 85kW (170kW Low-Power Mode) (fMOD = 1.024MHz) 11.5pF Figure 41. Simplified Reference Input Circuit (1) tMCD0, 1 M0 M1 (1) High-resolution mode: MCLK = fCLK/4; low-power mode: MCLK = fCLK/8. Figure 40. Modulator Mode Timing Modulator Output Timing for Figure 40 PARAMETER tMCD0, (1) 18 1 DESCRIPTION MIN TYP MCLK rising edge to M0, M1 valid propagation delay (1) MAX UNIT 100 ns tCMD CLK rising edge (after SYNC rising edge) to MCLK rising edge reset time tCSHD CLK to SYNC hold time to not latch on CLK edge 10 ns tSCSU SYNC to CLK setup time to latch on CLK edge 10 ns tSYMD SYNC to stable bit stream 5 1/fCLK 16 1/fMOD Load on M0 and M1 = 20pF || 100kΩ. Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 ADS1282 www.ti.com ................................................................................................................................................. SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 Table 4. Digital Filter Selection The ADS1282 reference inputs are protected by ESD diodes. In order to prevent these diodes from turning on, the voltage on either input must stay within the range shown in Equation 6: FILTR[1:0] BITS DIGITAL FILTERS SELECTED 00 Bypass; modulator output mode 01 Sinc 10 Sinc + FIR 11 Sinc + FIR + HPF (low-pass and high-pass) AVSS - 300mV < (VREFP or VREFN) < AVDD + 300mV (6) Note that the minimum valid input for VREFN is AVSS – 0.1V and maximum valid input for VREFP is AVDD + 0.1V. A high-quality +5V reference voltage is necessary for achieving the best performance from the ADS1282. Noise and drift on the reference degrade overall system performance, and it is critical that special care be given to the circuitry generating the reference voltages in order to achieve full performance. See the Application Information section for reference recommendations. DIGITAL FILTER Sinc Filter Stage (sinx/x) The sinc filter is a variable decimation rate, fifth-order, low-pass filter. Data are supplied to this section of the filter from the modulator at the rate of fMOD (fCLK/4). The sinc filter attenuates the high-frequency noise of the modulator, then decimates the data stream into parallel data. The decimation rate affects the overall data rate of the converter; it is set by the DR[2:0] register bits, as shown in Table 5. Equation 7 shows the scaled Z-domain transfer function of the sinc filter. The digital filter receives the modulator output and decimates the data stream. By adjusting the amount of filtering, tradeoffs can be made between resolution and data rate: filter more for higher resolution, filter less for higher data rate. The digital filter is comprised of three cascaded filter stages: a variable-decimation, fifth-order sinc filter; a fixed-decimation FIR, low-pass filter (LPF) with selectable phase; and a programmable, first-order, high-pass filter (HPF), as shown in Figure 42. The output can be taken from one of the three filter blocks, as Figure 42 shows. To implement the digital filter completely off-chip, select the filter bypass setting (modulator output). For partial filtering by the ADS1282, select the sinc filter output. For complete on-chip filtering, activate both the sinc and FIR stages. The HPF can then be included to remove dc and low frequencies from the data. Table 4 shows the filter options. N H(Z) = 1-Z -1 1-Z 5 (7) Table 5. Sinc Filter Data Rates (CLK = 4.096MHz) DR[2:0] REGISTER DECIMATION RATIO (N) SINC DATA RATE (SPS) 000 128 8,000 001 64 16,000 010 32 32,000 011 16 64,000 100 8 128,000 3 Direct Modulator Bit Stream Filter Mode (Register Select) 30 Filter MUX From Modulator Sinc Filter (Decimate by 8 to 128) Coefficient Filter (FIR) (Decimate by 32) High-Pass Filter (IIR) CAL Block Code Clip To Output Register 31 Figure 42. Digital Filter and Output Code Processing Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 19 ADS1282 SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 ................................................................................................................................................. www.ti.com Equation 8 shows the frequency domain transfer function of the sinc filter. 0 -0.5 5 -1.0 ½H(f)½ = N sin Gain (dB) 2pN ´ f sin fMOD 2p ´ f fMOD -2.0 (8) where: N = decimation ratio (see Table 5) 0 0 -20 Gain (dB) -40 -60 -80 -100 -120 -140 1 -2.5 -3.0 The sinc filter has notches (or zeroes) that occur at the output data rate and multiples thereof. At these frequencies, the filter has zero gain. Figure 43 shows the frequency response of the sinc filter and Figure 44 shows the roll-off of the sinc filter. 0 -1.5 2 3 4 Normalized Frequency (fIN/fDATA) 5 0.05 0.10 0.15 0.20 Normalized Frequency (fIN/fDATA) Figure 44. Sinc Filter Roll-Off FIR Stage The second stage of the ADS1282 digital filter is an FIR low-pass filter. Data are supplied to this stage from the sinc filter. The FIR stage is segmented into four sub-stages, as shown in Figure 45. The first two sub-stages are half-band filters with decimation ratios of 2. The third sub-stage decimates by 4 and the fourth sub-stage decimates by 2. The overall decimation of the FIR stage is 32. Note that two coefficient sets are used for the third and fourth sections, depending on the phase selection. Table 22 (in the Appendix section at the end of this document) lists the FIR stage coefficients. Table 6 lists the data rates and overall decimation ratio of the FIR stage. Table 6. FIR Filter Data Rates Figure 43. Sinc Filter Frequency Response Sinc Filter FIR Stage 1 Decimate by 2 FIR Stage 2 Decimate by 2 DR[2:0] REGISTER DECIMATION RATIO (N) FIR DATA RATE (SPS) 000 4096 250 001 2048 500 010 1024 1000 011 512 2000 100 256 4000 FIR Stage 4 Decimate by 2 FIR Stage 3 Decimate by 4 Output Coefficients Linear Minimum PHASE Select Figure 45. FIR Filter Sub-Stages 20 Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 ADS1282 www.ti.com ................................................................................................................................................. SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 As shown in Figure 46, the FIR frequency response provides a flat passband to 0.375 of the data rate (±0.003dB passband ripple). Figure 47 shows the transition from passband to stop band. 0.003 Magnitude (dB) 0.002 GROUP DELAY AND STEP RESPONSE The FIR block is implemented as a multi-stage FIR structure with selectable linear or minimum phase response. The passband, transition band, and stop band responses of the filters are nearly identical but differ in the respective phase responses. Linear Phase Response 0.001 0 -0.001 -0.002 -0.003 0 0.08 0.16 0.24 0.32 Normalized Input Frequency (fIN/fDATA) 0.40 Linear phase filters exhibit constant delay time versus input frequency (that is, constant group delay). Linear phase filters have the property that the time delay from any instant of the input signal to the same instant of the output data is constant and is independent of the signal nature. This filter behavior results in essentially zero phase error when analyzing multi-tone signals. However, the group delay and settling time of the linear phase filter are somewhat larger than the minimum phase filter, as shown in Figure 48. Figure 46. FIR Passband Magnitude Response (fDATA = 500Hz) 1.4 Minimum Phase Filter 1.2 0 Amplitude (dB) 1.0 Magnitude (dB) -20 -40 -60 0.8 0.6 0.4 0.2 Linear Phase Filter -80 0 -100 -0.2 0 -120 -140 0.35 0.40 0.45 0.50 0.55 0.60 Normalized Input Frequency (fIN/fDATA) 0.65 5 10 15 20 25 30 35 40 45 50 55 60 65 Time Index (1/fDATA) Figure 48. FIR Step Response Figure 47. FIR Transition Band Magnitude Response Although not shown in Figure 47, the passband response repeats at multiples of the modulator frequency (NfMOD – f0 and NfMOD + f0, where N = 1, 2, etc. and f0 = passband). These image frequencies, if present in the signal and not externally filtered, fold back (or alias) into the passband and cause errors. A low-pass signal filter reduces the effect of aliasing. Often, the RC low-pass filter provided by the PGA output resistors and the external capacitor connected to CAPP and CAPN provides sufficient signal attenuation. Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 21 ADS1282 SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 ................................................................................................................................................. www.ti.com Minimum Phase Response HPF Stage The minimum phase filter provides a short delay from the arrival of an input signal to the output, but the relationship (phase) is not constant versus frequency, as shown in Figure 49. The filter phase is selected by the PHS bit, as Table 7 shows. The last stage of the ADS1282 filter block is a first-order HPF implemented as an IIR structure. This filter stage blocks dc signals and rolls off low-frequency components below the cut-off frequency. The transfer function for the filter is shown in Equation 9: 1 - Z-1 2-a ´ HPF(Z) = -1 2 1 - bZ (9) 35 Linear Phase Filter Group Delay (1/fDATA) 30 where b is calculated as shown in Equation 10: 25 2 2 b= 20 (1 + (1 - a) ) 2 (10) The high-pass corner frequency is programmed by registers HPF[1:0], in hexadecimal. Equation 11 is used to set the high-pass corner frequency. Table 8 lists example values for the high-pass filter. 15 10 Minimum Phase Filter 5 0 20 40 60 80 100 120 Frequency (Hz) 140 160 180 200 HPF[1:0] = 65,536 1 - 1-2 cos wN + sin wN - 1 cos wN Figure 49. FIR Group Delay (fDATA = 500Hz) Table 7. FIR Phase Selection PHS BIT FILTER PHASE 0 Linear 1 Minimum (11) Where: HPF = High-pass filter register value (converted to hexadecimal) ωN = 2πfHP/fDATA (normalized frequency, radians) fHP = High-pass corner frequency (Hz) fDATA = Data rate (Hz) Table 8. High-Pass Filter Value Examples 22 fHP (Hz) DATA RATE (SPS) HPF[1:0] 0.5 250 0337h 1.0 500 0337h 1.0 1000 019Ah Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 ADS1282 www.ti.com ................................................................................................................................................. SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 The HPF causes a small gain error, in which case the magnitude of the error depends on the ratio of fHP/fDATA. For many common values of (fHP/fDATA), the gain error is negligible. Figure 50 shows the gain error of the HPF. The gain error factor is illustrated in Equation 15 (see the Appendix at the end of this document). Gain Error (dB) -0.10 -0.20 SYNCHRONIZATION (SYNC PIN AND SYNC COMMAND) -0.30 The ADS1282 can be synchronized to an external event, as well as synchronized to other ADS1282 devices if the sync event is applied simultaneously. -0.40 -0.50 0.0001 0.001 0.01 0.1 Frequency Ratio (fHP/fDATA) Figure 50. HPF Gain Error 0 90 -7.5 75 The ADS1282 has two sources for synchronization: the SYNC input pin and the SYNC command. The ADS1282 also has two synchronizing modes: Pulse-sync and Continuous-sync. In Pulse-sync mode, the ADS1282 synchronizes to a single sync event. In Continuous-sync mode, either a single SYNC event is used to synchronize conversions or a continuous clock is applied to the pin with a period equal to integer multiples of the data rate. When the periods of the sync input and the DRDY output do not match, the ADS1282 re-synchronizes and conversions are restarted. 60 Amplitude 45 -22.5 Phase -30.0 30 -37.5 15 Phase (°) Amplitude (dB) Figure 51 shows the first-order amplitude and phase response of the HPF. Note that in the case of applying step inputs or synchronizing, the settling time of the filter should be taken into account. -45.0 0.01 The ADS1282 requires a clock input for operation. The clock is applied to the CLK pin. The data conversion rate scales directly with the CLK frequency. Power consumption versus CLK frequency is relatively constant (see the Typical Characteristics). As with any high-speed data converter, a high-quality, low-jitter clock is essential for optimum performance. Crystal clock oscillators are the recommended clock source. Make sure to avoid excess ringing on the clock input; keep the clock trace as short as possible and use a 50Ω series resistor close to the source. 0 -15.0 MASTER CLOCK INPUT (CLK) 0 0.1 1 10 Normalized Frequency (f/fC) 100 Figure 51. HPF Amplitude and Phase Response Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 23 ADS1282 SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 ................................................................................................................................................. www.ti.com PULSE-SYNC MODE tCSHD In Pulse-sync mode, the ADS1282 stops and restarts the conversion process when a sync event occurs (by pin or command). When the sync event occurs, the device resets the internal memory; DRDY goes high (pulse SYNC mode) otherwise in Continuous SYNC mode, DRDY continues to toggle, and after the digital filter has settled, new conversion data are available, as shown in Figure 52 and Table 9. Note that resynchronization occurs on the next rising CLK edge after the rising edge of the SYNC pin or after the eighth rising SCLK edge for opcode SYNC commands. To be effective, the SYNC opcode should be broadcast to all devices simultaneously. CONTINUOUS-SYNC MODE In Continuous-sync mode, either a single sync pulse or a continuous clock may be applied. When a single sync pulse is applied (rising edge), the device behaves similar to the Pulse-sync mode. However, in this mode, DRDY continues to toggle unaffected but the DOUT output is held low until data are ready, 63 DRDY periods later. When the conversion data are non-zero, new conversion data are ready (as shown in Figure 52). When a continuous clock is applied to the SYNC pin, the period must be an integral multiple of the output data rate or the device re-synchronizes. When the sync input is first applied, the device re-synchronizes (under the condition tSYNC ≠ N/fDATA). DRDY continues to output but DOUT is held low until the new data are ready. Then, if SYNC is applied again and the period matches an integral multiple of the output data rate, the device freely runs without re-synchronization. Note that the phase of the applied clock and output data rate (DRDY) are not matched because of the initial delay of DRDY after SYNC is first applied. Figure 53 shows the timing for Continuous-Sync mode. System Clock (fCLK) tSCSU SYNC Command tSPWH SYNC Pin New Data Ready tSPWL tDR DRDY (Pulse-Sync) New Data Ready tDR DRDY (Continuous-Sync) DOUT Figure 52. Pulse-Sync Timing, Continuous-Sync Timing with Single Sync tSCSU tCSHD System Clock (fCLK) tSPWL tSPWH SYNC tSYNC DRDY 1/fDATA Figure 53. Continuous-Sync Timing with Sync Clock Note that a SYNC clock input should be applied after the Continuous-Sync mode is set. The first rising edge of SYNC then causes a synchronization. Table 9. Pulse-Sync Timing for Figure 52 and Figure 53 PARAMETER DESCRIPTION MIN MAX UNITS tSYNC SYNC period (1) 1 Infinite n/fDATA tCSHD CLK to SYNC hold time to not latch on CLK edge 10 tSCSU SYNC to CLK setup time to latch on CLK edge 10 ns SYNC pulse width, high or low 2 1/fCLK tSPWH, tDR (1) 24 L Time for data ready (SINC filter) ns See Appendix, Table 23 Time for data ready (FIR filter) 62.98046875/fDATA + 466/fCLK Continuous-Sync mode; a free-running SYNC clock input without causing re-synchronization. Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 ADS1282 www.ti.com ................................................................................................................................................. SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 RESET (RESET Pin and Reset Command) The ADS1282 may be reset in two ways: toggle the RESET pin low or send a Reset command. When using the RESET pin, take it low and hold for at least 2/fCLK to force a reset. The ADS1282 is held in reset until the pin is released. By command, RESET takes effect on the next rising edge of fCLK after the eighth rising edge of SCLK of the command. Note that in order to ensure the Reset command can function, the SPI interface may require resetting itself; see the Serial Interface section. In reset, registers are set to default and the conversions are synchronized on the next rising edge of CLK. New conversion data are available, as shown in Figure 54 and Table 10. Settled Data DRDY tDR tCRHD Wakeup Command DRDY tDR Figure 55. PWDN Pin and Wake-Up Command Timing (Table 11 shows tDR) The ADS1282 has three power supplies: AVDD, AVSS, and DVDD. Figure 56 shows the power-on sequence of the ADS1282. The power supplies can be sequenced in any order. The supplies [the difference of (AVDD – AVSS) and DVDD] generate an internal reset whose outputs are summed to generate a global internal reset. After the supplies have crossed the minimum thresholds, 216 fCLK cycles are counted before releasing the internal reset. After the internal reset is released, new conversion data are available, as shown in Figure 56 and Table 11. tRCSU RESET Pin or RESET Command Figure 54. Reset Timing Table 10. Reset Timing for Figure 54 PARAMETER DESCRIPTION PWDN Pin POWER-ON SEQUENCE System Clock (fCLK) tRST In power-down, note that the device outputs remain active and the device inputs must not float. When the Standby command is sent, the SPI port and the configuration registers are kept active. Figure 55 and Table 11 show the timing. MIN UNITS tCRHD CLK to RESET hold time 10 ns tRCSU RESET to CLK setup time 10 ns tRST RESET low 2 1/fCLK tDR Time for data ready 62.98046875/ fDATA + 468/fCLK AVDD - AVSS DVDD 3.5V nom 1V nom CLK 16 POWER-DOWN (PWDN Pin and Standby Command) Internal Reset There are two ways to power-down the ADS1282: take the PWDN pin low or send a Standby command. When the PWDN pin is pulled low, the internal circuitry is disabled to minimize power and the contents of the register settings are reset. 2 fCLK DRDY tDR Figure 56. Power-On Sequence Table 11. Power-On, PWDN Pin, and Wake-Up Command Timing for New Data PARAMETER DESCRIPTION FILTER MODE 16 tDR (1) (2) Time for data ready 2 CLK cycles after power-on; and new data ready after PWDN pin or Wake-Up command See Appendix, Table 23 SINC (1) 62.98046875/fDATA + 468/fCLK (2) FIR Supply power-on and PWDN pin default is 1000SPS FIR. Subtract two CLK cycles for the Wake-Up command. The Wake-Up command is timed from the next rising edge of CLK to after the eighth rising edge of SCLK during command to DRDY falling. Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 25 ADS1282 SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 ................................................................................................................................................. www.ti.com DVDD POWER SUPPLY Serial Clock (SCLK) The DVDD supply operates over the range of +1.65V to +3.6V. If DVDD is operated at less than 2.25V, connect the DVDD pin to the BYPAS pin. If DVDD is greater than or equal to 2.25V, do not connect DVDD to the BYPAS pin. Figure 57 shows this connection. The serial clock (SCLK) is an input that is used to clock data into (DIN) and out of (DOUT) the ADS1282. This input is a Schmitt-trigger input that has a high degree of noise immunity. However, it is recommended to keep SCLK as clean as possible to prevent possible glitches from inadvertently shifting the data. 1.65V to 3.6V DVDD Data are shifted into DIN on the rising edge of SCLK and data are shifted out of DOUT on the falling edge of SCLK. If SCLK is held low for 64 DRDY cycles, data transfer or commands in progress terminate and the SPI interface resets. The next SCLK pulse starts a new communication cycle. This timeout feature can be used to recover the interface when a transmission is interrupted or SCLK inadvertently glitches. SCLK should remain low when not active. ADS1282 Tie DVDD to BYPAS if DVDD power is < 2.25V. Otherwise float BYPAS. BYPAS 1m F Figure 57. DVDD Power Data Input (DIN) SERIAL INTERFACE A serial interface is used to read the conversion data and access the configuration registers. The interface consists of three basic signals: SCLK, DIN, and DOUT. An additional output, DRDY, transitions low in Read Data Continuous mode when data are ready for retrieval. Figure 58 shows the connection when multiple converters are used. FPGA or Processor SCLK DOUT1 ADS1282 DIN2 DRDY1 SCLK DOUT2 ADS1282 DIN2 DRDY2 SCLK The data input pin (DIN) is used to input register data and commands to the ADS1282. Keep DIN low when reading conversion data in the Read Data Continuous mode (except when issuing a STOP Read Data Continuous command). Data on DIN are shifted into the converter on the rising edge of SCLK. In Pin mode, DIN is not used. Data Output (DOUT) The data output pin (DOUT) is used to output data from the ADS1282. Data are shifted out on DOUT on the falling edge of SCLK. DOUT1 DIN1 IRQ SCLK (optional) DOUT2 DIN2 IRQ (optional) Figure 58. Interface for Multiple Devices 26 Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 ADS1282 www.ti.com ................................................................................................................................................. SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 Data Ready (DRDY) DATA FORMAT DRDY is an output; when it transitions low, this transition indicates new conversion data are ready, as shown in Figure 59. When reading data by the continuous mode, the data must be read within four CLK periods before DRDY goes low again or the data are overwritten with new conversion data. When reading data by the command mode, the read operation can overlap the occurrence of the next DRDY without data corruption. The ADS1282 provides 32 bits of conversion data in binary twos complement format, as shown in Table 12. The LSB of the data is a redundant sign bit: '0' for positive numbers and '1' for negative numbers. However, when the output is clipped to +FS, the LSB = 1; when the output is clipped to –FS, the LSB = 0. If desired, the data readback may be stopped at 24 bits. Note that in sinc filter mode, the output data are scaled by 1/2. Table 12. Ideal Output Code versus Input Signal DRDY INPUT SIGNAL VIN (AINP – AINN) DOUT Bit 31 Bit 30 32-BIT IDEAL OUTPUT CODE(1) Bit 29 FIR FILTER SCLK VREF > 2 x PGA VREF Figure 59. DRDY with Data Retrieval 2 x PGA DRDY resets high on the first falling edge of SCLK. Figure 59 and Figure 60 show the function of DRDY with and without data readback, respectively. If data are not retrieved (no SCLK provided), DRDY pulses high for four fCLK periods during the update time, as shown in Figure 60. 4/fCLK VREF 2PGA ´ (230 - 1) 0 -VREF 2PGA ´ (230 - 1) Data Updating -VREF 2PGA DRDY Figure 60. DRDY With No Data Retrieval < -VREF 2PGA ´ 230 230 - 1 ´ 230 230 - 1 SINC FILTER 7FFFFFFFh (2) 7FFFFFFEh 3FFFFFFFh 00000002h 00000001h 00000000h 00000000h FFFFFFFFh FFFFFFFFh 80000001h C0000000h 80000000h (2) (1) Excludes effects of noise, linearity, offset, and gain errors. (2) In sinc filter mode, the output does not clip at half-scale code when the full-scale range is exceeded. Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 27 ADS1282 SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 ................................................................................................................................................. www.ti.com READING DATA The ADS1282 has two ways to read conversion data: Read Data Continuous and Read Data By Command. When a Stop Read Data Continuous command is issued, the DRDY output is blocked but the ADS1282 continues conversions. In stop continuous mode, the data can only be read by command. Read Data Continuous Read Data By Command In the Read Data Continuous mode, the conversion data are shifted out directly from the device without the need for sending a read command. This mode is the default mode at power-on. This mode is also enabled by the RDATAC command. When DRDY goes low, indicating that new data are available, the MSB of data appears on DOUT, as shown in Figure 61. The data are normally read on the rising edge of SCLK, at the occurrence of the first falling edge of SCLK, DRDY returns high. After 32 bits of data have been shifted out, further SCLK transitions cause DOUT to go low. If desired, the read operation may be stopped at 24 bits. The data shift operation must be completed within four CLK periods before DRDY falls again or the data may be corrupted. The Read Data Continuous mode is stopped by the SDATAC command. In this mode, conversion data are read by command. In the Read Data By Command mode, a read data command must be sent to the device for each data conversion (as shown in Figure 62). When the read data command is received (on the eighth SCLK rising edge), data are available to read only when DRDY goes low (tDR). When DRDY goes low, conversion data appear on DOUT. The data may be read on the rising edge of SCLK. DRDY 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 25 26 27 28 29 30 31 32 SCLK DOUT Data Byte 1 (MSB) Data Byte 2 (MSB - 1) Data Byte 4 (LSB) tDDPD DIN Figure 61. Read Data Continuous Table 13. Timing Data for Figure 61 PARAMETER DESCRIPTION (1) MIN TYP DRDY to valid MSB on DOUT propagation delay (1) tDDPD MAX UNITS 100 ns Load on DOUT = 20pF || 100kΩ. DRDY tDR 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 33 34 35 36 37 38 39 40 SCLK DOUT Don't Care DIN Data Byte 1 (MSB) Date Byte 4 (LSB) tDDPD Command Byte (0001 0010) Figure 62. Read Data By Command, RDATA (tDDPD timing is given in Table 13) Table 14. Read Data Timing for Figure 62 PARAMETER tDR 28 DESCRIPTION MIN Time for new data after data read command Submit Documentation Feedback 0 TYP MAX UNITS 1 fDATA Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 ADS1282 www.ti.com ................................................................................................................................................. SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 ONE-SHOT OPERATION The ADS1282 can perform very power-efficient, one-shot conversions using the STANDBY command while under software control. Figure 63 shows this sequence. First, issue the STANDBY command to set the Standby mode. When ready to make a measurement, issue the WAKEUP command. Monitor DRDY; when it goes low, the fully settled conversion data are ready and may be read directly in Read Data Continuous mode. Afterwards, issue another STANDBY command. When ready for the next measurement, repeat the cycle starting with another WAKEUP command. OFFSET AND FULL-SCALE CALIBRATION REGISTERS The conversion data can be scaled for offset and gain before yielding the final output code. As shown in Figure 64, the output of the digital filter is first subtracted by the offset register (OFC) and then multiplied by the full-scale register (FSC). Equation 12 shows the scaling: FSC[2:0] Final Output Data = (Input - OFC[2:0]) ´ 400000h (12) The values of the offset and full-scale registers are set by writing to them directly, or they are set automatically by calibration commands. Note that the offset and full-scale calibrations apply to specific PGA settings. When the PGA changes, the contents of these registers may have to be recalculated. Calibration is bypassed in the sinc filter mode. Standby ADS1282 Status Performing One-Shot Conversion Standby DRDY DIN (1) STANDBY STANDBY WAKEUP Settled Data DOUT (1) See Figure 55 and Table 11 for time to new data. Figure 63. One-Shot Conversions Using the STANDBY Command AINP Modulator AINN Digital Filter + S ´ OFC Register FSC Register 400000h - Output Data Clipped to 32 Bits Final Output Figure 64. Calibration Block Diagram Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 29 ADS1282 SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 ................................................................................................................................................. www.ti.com OFC[2:0] Registers FSC[2:0] Registers The offset calibration is a 24-bit word, composed of three 8-bit registers, as shown in Table 17. The offset register is left-justified to align with the 32-bits of conversion data. The offset is in twos complement format with a maximum positive value of 7FFFFFh and a maximum negative value of 800000h. This value is subtracted from the conversion data. A register value of 00000h has no offset correction (default value). Note that while the offset calibration register value can correct offsets ranging from –FS to +FS (as shown in Table 15), to avoid input overload, the analog inputs cannot exceed the full-scale range. The full-scale calibration is a 24-bit word, composed of three 8-bit registers, as shown in Table 18. The full-scale calibration value is 24-bit, straight offset binary, normalized to 1.0 at code 400000h. Table 16 summarizes the scaling of the full-scale register. A register value of 400000h (default value) has no gain correction (gain = 1). Note that while the gain calibration register value corrects gain errors above 1 (gain correction < 1), the full-scale range of the analog inputs cannot be exceeded to avoid input overload. Table 16. Full-Scale Calibration Register Values Table 15. Offset Calibration Values FSC REGISTER GAIN CORRECTION OFC REGISTER FINAL OUTPUT CODE(1) 800000h 2.0 7FFFFFh 80000000h 400000h 1.0 000001h FFFFFF00h 200000h 0.5 000000h 00000000h 000000h 0 FFFFFFh 00000100h 800000h 7FFFFF00h (1) Full 32-bit final output code with zero code input. Table 17. Offset Calibration Word REGISTER BYTE OFC0 LSB B7 B6 B5 B4 BIT ORDER B3 B2 B1 OFC1 MID B15 B14 B13 B12 B11 B10 B9 B8 OFC2 MSB B23 (MSB) B22 B21 B20 B19 B18 B17 B16 B0 (LSB) B0 (LSB) Table 18. Full-Scale Calibration Word 30 REGISTER BYTE FSC0 LSB B7 B6 B5 B4 BIT ORDER B3 B2 B1 FSC1 MID B15 B14 B13 B12 B11 B10 B9 B8 FSC2 MSB B23 (MSB) B22 B21 B20 B19 B18 B17 B16 Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 ADS1282 www.ti.com ................................................................................................................................................. SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 CALIBRATION COMMANDS OFSCAL Command Calibration commands may be sent to the ADS1282 to calibrate the conversion data. The values of the offset and gain calibration registers are internally written to perform calibration. The appropriate input signals must be applied to the ADS1282 inputs before sending the commands. Use slower data rates to achieve more consistent calibration results; this effect is a byproduct of the lower noise that these data rates provide. Also, if calibrating at power-on, be sure the reference voltage is fully settled. The OFSCAL command performs an offset calibration. Before sending the offset calibration command, a zero input signal must be applied to the ADS1282 and the inputs allowed to stabilize. When the command is sent, the ADS1282 averages 16 readings and then writes this value to the OFC register. The contents of the OFC register may be subsequently read or written. During offset calibration, the full-scale correction is bypassed. Figure 65 shows the calibration command sequence. After the analog input voltage (and reference) have stabilized, send the Stop Data Continuous command followed by the SYNC and Read Data Continuous commands. 64 data periods later, DRDY goes low. After DRDY goes low, send the Stop Data Continuous, then the Calibrate command followed by the Read Data Continuous command. After 16 data periods, calibration is complete and conversion data may be read at this time. The SYNC input must remain high during the calibration sequence. Note that the calibration commands apply to specific PGA settings. If the PGA is changed, recalibration is necessary. Calibration is bypassed in the sinc filter mode. GANCAL Command The GANCAL command performs a gain calibration. Before sending the GANCAL command, a dc input signal must be applied that is in the range of, but not exceeding, positive or negative full-scale. After the signal has stabilized, the command can be sent. The ADS1282 averages 16 readings, then computes the value that compensates for the gain error. The gain correction value is then written to the FSC register. The contents of the GANCAL register may be subsequently read or written. Note that while the gain calibration command corrects for gain errors above 1 (gain correction < 1), to avoid input overload, the analog inputs cannot exceed full-scale range. The gain calibration should be performed after the offset calibration. VIN Fully stable input and reference voltage. Commands SDATAC DRDY SYNC RDATAC SDATAC OFSCAL or GANCAL RDATAC 16 Data Periods 64 Data Periods Calibration Complete SYNC Figure 65. Offset/Gain Calibration Timing Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 31 ADS1282 SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 ................................................................................................................................................. www.ti.com USER CALIBRATION System calibration of the ADS1282 can be performed without using the calibration commands. This procedure requires the calibration values to be externally calculated and then written to the calibration registers. The steps for this procedure are: 1. Set the OFSCAL[2:0] register = 0h and GANCAL[2:0] = 400000h. These values set the offset and gain registers to 0 and 1, respectively. 2. Apply a zero differential input to the input of the system. Wait for the system to settle and then average n output readings. Higher numbers of averaged readings result in more consistent calibration. Write the averaged value to the OFC register. 3. Apply a differential positive or negative dc signal, or an ac signal, less than the full-scale input to the system. Wait for the system to settle and then average the n output readings. 32 The value written to the FSC registers is calculated by Equation 13. DC signal calibration is shown in Equation 13. The expected output code is based on 31-bit output data. FSC[2:0] = 400000h ´ Expected Output Code Actual Output Code (13) For ac signal calibration, use an RMS value of collected data (as shown in Equation 14). Expected RMS Value FSC[2:0] = 400000h ´ Actual RMS Value (14) Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 ADS1282 www.ti.com ................................................................................................................................................. SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 COMMANDS The commands listed in Table 19 control the operation of the ADS1282. Most commands are stand-alone (that is, 1 byte in length); the register reads and writes require a second command byte in addition to the actual data bytes. In Read Data Continuous mode, the ADS1282 places conversion data on the DOUT pin as SCLK is applied. As a consequence of the potential conflict of conversion data on DOUT and data placed on DOUT resulting from a register or Read Data By Command operation, it is necessary to send a STOP Read Data Continuous command before Register or Data Read By Command. The STOP Read Data Continuous command disables the direct output of conversion data on the DOUT pin. A delay of 24 fCLK cycles between commands and between bytes within a command is required, starting from the last SCLK rising edge of one command to the first SCLK rising edge of the following command. This delay is shown in Figure 66. Command Byte DIN Command Byte SCLK (1) tSCLKDLY (1) tSCLKDLY = 24/fCLK (min). Figure 66. Consecutive Commands Table 19. Command Descriptions COMMAND TYPE DESCRIPTION 1st COMMAND BYTE (1) (2) WAKEUP Control Wake-up from Standby mode 0000 000X (00h or 01h) STANDBY Control Enter Standby mode 0000 001X (02h or 03h) SYNC Control Synchronize the A/D conversion 0000 010X (04h or 5h) RESET Control Reset registers to default values 0000 011X (06h or 07h) RDATAC Control Read data continuous 0001 0000 (10h) SDATAC Control Stop read data continuous 0001 0001 (11h) RDATA Data RREG Register Read nnnnn register(s) at address rrrrr (4) 001r rrrr (20h + 000r rrrr) 000n nnnn (00h + n nnnn) WREG Register Write nnnnn register(s) at address rrrrr 010r rrrr (40h + 000r rrrr) 000n nnnn (00h + n nnnn) OFSCAL Calibration Offset calibration 0110 0000 (60h) GANCAL Calibration Gain calibration 0110 0001 (61h) (1) (2) (3) (4) Read data by command (4) 2nd COMMAND BYTE (3) 0001 0010 (12h) X = don't care. rrrrr = starting address for register read and write commands. nnnnn = number of registers to be read/written – 1. For example, to read/write three registers, set nnnnn = 2 (00010). Required to cancel Read Data Continuous mode before sending a command. Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 33 ADS1282 SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 ................................................................................................................................................. www.ti.com WAKEUP: Wake-Up From Standby Mode SDATAC: Stop Read Data Continuous Description: This command is used to exit the standby mode. Upon sending the command, the time for the first data to be ready is illustrated in Figure 55 and Table 12. Sending this command during normal operation has no effect; for example, reading data by the Read Data Continuous method with DIN held low. Description: This command stops the Read Data Continuous mode. Exiting the Read Data Continuous mode is required before sending Register and Data read commands. This command suppresses the DRDY output, but the ADS1282 continues conversions. STANDBY: Standby Mode RDATA: Read Data By Command Description: This command places the ADS1282 into Standby mode. In Standby, the device enters a reduced power state where a low quiescent current remains to keep the register settings and SPI interface active. For complete device shutdown, take the PWDN pin low (register settings are not saved). To exit Standby mode, issue the WAKEUP command. The operation of Standby mode is shown in Figure 67. Description: This command reads the conversion data. See the Read Data By Command section for more details. DIN 0000 001X (STANDBY) 0000 000X (WAKEUP) Standby Mode Operating Figure 67. STANDBY Command Sequence SYNC: Synchronize the A/D Conversion Description: This command synchronizes the A/D conversion. Upon receipt of the command, the reading in progress is cancelled and the conversion process is re-started. In order to synchronize multiple ADS1282s, the command must be sent simultaneously to all devices. Note that the SYNC pin must be high for this command. RESET: Reset the Device Description: The RESET command resets the registers to default values, enables the Read Data Continuous mode, and restarts the conversion process; the RESET command is functionally the same as the RESET pin. See Figure 54 for the RESET command timing. RDATAC: Read Data Continuous Description: This command enables the Read Data Continuous mode (default mode). In this mode, conversion data can be read from the device directly without the need to supply a data read command. Each time DRDY falls low, new data are available to read. See the Read Data Continuous section for more details. 34 Description: This command is used to read single or multiple register data. The command consists of a two-byte op-code argument followed by the output of register data. The first byte of the op-code includes the starting address, and the second byte specifies the number of registers to read – 1. First command byte: 001r rrrr, where rrrrr is the starting address of the first register. SCLK Operating RREG: Read Register Data Second command byte: 000n nnnn, where nnnnn is the number of registers – 1 to read. Starting with the 16th falling edge of SCLK, the register data appear on DOUT. The RREG command is illustrated in Figure 68. Note that a delay of 24 fCLK cycles is required between each byte transaction. WREG: Write to Register Description: This command writes single or multiple register data. The command consists of a two-byte op-code argument followed by the input of register data. The first byte of the op-code contains the starting address and the second byte specifies the number of registers to write – 1. First command byte: 001r rrrr, where rrrrr is the starting address of the first register. Second command byte: 000n nnnn, where nnnnn is the number of registers – 1 to write. Data byte(s): one or more register data bytes, depending on the number of registers specified. Figure 69 illustrates the WREG command. Note that a delay of 24 fCLK cycles is required between each byte transaction. Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 ADS1282 www.ti.com ................................................................................................................................................. SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 OFSCAL: Offset Calibration GANCAL: Gain Calibration Description: This command performs an offset calibration. The inputs to the converter (or the inputs to the external pre-amplifier) should be zeroed and allowed to stabilize before sending this command. The offset calibration register updates after this operation. See the Calibration Commands section for more details. Description: This command performs a gain calibration. The inputs to the converter should have a stable dc input, preferably close to (but not exceeding) positive full-scale. The gain calibration register updates after this operation. See the Calibration Commands section for more details. tDLY 1 2 3 4 5 6 7 8 9 tDLY 10 11 12 13 14 15 16 tDLY 17 18 19 20 21 22 23 24 25 26 SCLK DIN Command Byte 1 DOUT Command Byte 2 Don't Care Register Data 5 Register Data 6 Example: Read six registers, starting at register 05h (OFC0) Command Byte 1 = 0010 0101 Command Byte 2 = 0000 0101 Figure 68. Read Register Data (Table 20 shows tDLY) tDLY 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 tDLY tDLY 17 18 19 20 21 22 23 24 25 26 SCLK DIN Command Byte 1 Command Byte 2 Register Data 5 Register Data 6 Example: Write six registers, starting at register 05h (OFC0) Command Byte 1 = 0100 0101 Command Byte 2 = 0000 0101 Figure 69. Write Register Data (Table 20 shows tDLY) Table 20. tDRY Value PARAMETER MIN tDLY 24/fCLK Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 35 ADS1282 SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 ................................................................................................................................................. www.ti.com REGISTER MAP Collectively, the registers contain all the information needed to configure the part, such as data rate, filter selection, calibration, etc. The registers are accessed by the RREG and WREG commands. The registers can be accessed individually or as a block of registers by sending or receiving consecutive bytes. Table 21. Register Map ADDRESS REGISTER RESET VALUE BIT 7 BIT 6 BIT 5 BIT 4 BIT 3 BIT 2 BIT 1 00h ID X0h ID3 ID2 ID1 ID0 0 0 0 0 01h CONFIG0 52h SYNC MODE DR2 DR1 DR0 PHS FILTR1 FILTR0 02h CONFIG1 08h 0 MUX2 MUX1 MUX0 CHOP PGA2 PGA1 PGA0 03h HPF0 32h HPF07 HPF06 HPF05 HPF04 HPF03 HPF02 HPF01 HPF00 04h HPF1 03h HPF15 HPF14 HPF13 HPF12 HPF11 HPF10 HPF09 HPF08 05h OFC0 00h OFC07 OFC06 OFC05 OFC04 OFC03 OFC02 OFC01 OFC00 06h OFC1 00h OFC15 OFC14 OFC13 OFC12 OFC11 OFC10 OFC09 OFC08 07h OFC2 00h OFC23 OFC22 OFC21 OFC20 OFC19 OFC18 OFC17 OFC16 08h FSC0 00h FSC07 FSC06 FSC05 FSC04 FSC03 FSC02 FSC01 FSC00 09h FSC1 00h FSC15 FSC14 FSC13 FSC12 FSC11 FSC10 FSC09 FSC08 0Ah FSC2 40h FSC23 FSC22 FSC21 FSC20 FSC19 FSC18 FSC17 FSC16 BIT 0 ID : ID REGISTER (ADDRESS 00h) 7 6 5 4 3 2 1 0 ID3 ID2 ID1 ID0 0 0 0 0 Reset value = X0h. Bit[7:4] ID[3:0] Factory-programmed identification bits (read-only) Bit[3:0] Reserved Always write '0' 36 Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 ADS1282 www.ti.com ................................................................................................................................................. SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 CONFIG0 : CONFIGURATION REGISTER 0 (ADDRESS 01h) 7 6 5 4 3 2 1 0 SYNC MODE DR2 DR1 DR0 PHASE FILTR1 FILTR0 Reset value = 52h. Bit[7] SYNC Synchronization mode 0: Pulse SYNC mode (default) 1: Continuous SYNC mode Bit[6] MODE 0: Low-power mode 1: High-resolution mode (default) Bit[5:3] Data Rate Select DR[2:0] 000: 250SPS 001: 500SPS 010: 1000SPS (default) 011: 2000SPS 100: 4000SPS Bit[2] FIR Phase Response PHASE 0: Linear phase (default) 1: Minimum phase Bit[1:0] Digital Filter Select FILTR[1:0] Digital filter configuration 00: On-chip filter bypassed, modulator output mode 01: Sinc filter block only 10: Sinc + LPF filter blocks (default) 11: Sinc + LPF + HPF filter blocks Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 37 ADS1282 SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 ................................................................................................................................................. www.ti.com CONFIG1 : CONFIGURATION REGISTER 1 (ADDRESS 02h) 7 6 5 4 3 2 1 0 0 MUX2 MUX1 MUX0 CHOP PGA2 PGA1 PGA0 Reset value = 08h. Bit[7] Reserved Always write '0' Bit[6:4] MUX Select MUX[2:0] 000: AINP1 and AINN1 (default) 001: AINP2 and AINN2 010: Internal short via 400Ω 011:AINP1 and AINN1 connected to AINP2 and AINN2 100: External short to AINN2 Bit[3] PGA Chopping Enable CHOP 0: PGA chopping disabled 1: PGA chopping enabled (default) Bit[2:0] PGA Gain Select PGA[2:0] 000: G 001: G 010: G 011: G 100: G 101: G 110: G 38 = = = = = = = 1 (default) 2 4 8 16 32 64 Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 ADS1282 www.ti.com ................................................................................................................................................. SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 HPF1 and HPF0 These two bytes (high-byte and low-byte, respectively) set the corner frequency of the high-pass filter. HPF0: High-Pass Filter Corner Frequency, Low Byte (Address 03h) 7 6 5 4 3 2 1 0 HP07 HP06 HP05 HP04 HP03 HP02 HP01 HP00 Reset value = 32h. HPF1: High-Pass Filter Corner Frequency, High Byte (Address 04h) 7 6 5 4 3 2 1 0 HP15 HP14 HP13 HP12 HP11 HP10 HP09 HP08 Reset value = 03h. OFC2, OFC1, OFC0 These three bytes set the offset calibration value. OFC0: Offset Calibration, Low Byte (Address 05h) 7 6 5 4 3 2 1 0 OC07 OC06 OC05 OC04 OC03 OC02 OC01 OC00 Reset value = 00h. OFC1: Offset Calibration, Mid Byte (Address 06h) 7 6 5 4 3 2 1 0 OC15 OC14 OC13 OC12 OC11 OC10 OC09 OC08 Reset value = 00h. OFC2: Offset Calibration, High Byte (Address 07h) 7 6 5 4 3 2 1 0 OC23 OC22 OC21 OC20 OC19 OC18 OC17 OC16 Reset value = 00h. FSC2, FSC1, FSC0 These three bytes set the full-scale calibration value. FSC0: Full-Scale Calibration, Low Byte (Address 08h) 7 6 5 4 3 2 1 0 FSC07 FSC06 FSC05 FSC04 FSC03 FSC02 FSC01 FSC00 Reset value = 00h. FSC1: Full-Scale Calibration, Mid Byte (Address 09h) 7 6 5 4 3 2 1 0 FSC15 FSC14 FSC13 FSC12 FSC11 FSC10 FSC09 FSC08 Reset value = 00h. Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 39 ADS1282 SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 ................................................................................................................................................. www.ti.com FSC2: Full-Scale Calibration, High Byte (Address 0Ah) 7 6 5 4 3 2 1 0 FSC23 FSC22 FSC21 FSC20 FSC19 FSC18 FSC17 FSC16 Reset value = 40h. CONFIGURATION GUIDE After RESET or power-on, the registers can be configured using the following procedure: 1. Reset the serial interface. Before using the serial interface, it may be necessary to recover the serial interface (undefined I/O power-up sequencing may cause false SCLK detection). To reset the SPI interface, toggle the RESET pin or, when in Read Data Continuous mode, hold SCLK low for 64 DRDY periods. 2. Configure the registers. The registers are configured by either writing to them individually or as a group. Software may be configured in either mode. The SDATAC command must be sent before register read/write operations to cancel the Read Data Continuous mode. 3. Verify register data. The register may be read back for verification of device communications. 40 4. Set the data mode. After register configuration, the device may be configured for Read Data Continuous mode, either by the Read Data Continuous command or configured in Read Data By Register mode using SDATAC command. 5. Synchronize readings. Whenever SYNC is high, the ADS1282 freely runs the data conversions. To stop and re-sync the conversions, take SYNC low and then high. 6. Read data. If the Read Data Continuous mode is active, the data are read directly after DRDY falls by applying SCLK pulses. If the Read Data Continuous mode is inactive, the data can only be read by Read Data By Command. The Read Data opcode command must be sent in this mode to read each conversion result (note that DRDY only asserts after each read data command is sent). Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 ADS1282 www.ti.com ................................................................................................................................................. SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 APPLICATION INFORMATION The ADS1282 is a very high-resolution ADC. Optimal performance requires giving special attention to the support circuitry and printed circuit board (PCB) design. Locate noisy digital components, such as microcontrollers, oscillators, etc, in an area of the PCB away from the converter or front-end components. Locating the digital components close to the power-entry point keeps the digital current path short and separate from sensitive analog components. A typical geophone front-end application is shown in Figure 70. The application shows the ADS1282 operation with dual ±2.5V analog supplies. The ADS1282 can also operate with a single +5V analog supply. The geophone input signal is filtered both differentially, by components C4 and R1 to R4 and filtered independently by components C2, C3 and R1, R2. The differential filter removes high-frequency normal mode components from the input signal. The independent filters remove high-frequency components that are common to both input signals leads (common-mode filter). The recommended input filters may not be required for all applications depending on the system requirements. Resistors R5 and R6 bias the signals inputs to midsupply (ground), and also provide the bias current return path for the ADS1282 inputs. For single-supply operation, set the bias to a low impedance +2.5V (AVDD/2). Resistors R5 and R6 can also influence common-mode attenuation. To maintain good CMR performance, resistors R5 and R6 may require matching. The REF02 +5V reference provides the reference to the ADS1282. The reference output is filtered by R7 and C5. The filter requires several seconds to settle after power-on. Capacitor C7 provides high-frequency bypassing of the reference inputs and should be placed close to the ADS1282 pins. Note that R7 (1kΩ) results in a systematic gain error (–1.2% in high-resolution mode, and –0.6% in low-power mode). Alternatively, the REF5050 (5V) or REF5045 (4.5V) reference can be used. The REF5045 reference has the advantage of operating from the +5V power supply. The REF5050 requires +5.2V minimum power supply. The noise filter is also required on these references. Optional components R8, and R9 provides a 20mV offset to the ADS1282. The internal 300Ω resistors form a voltage divider with the external resistors to provide the offset. The offset moves the low level idle tones out of the passsband. Note that the offset is independent of the PGA setting. To maintain good CMR performance, R10 and R11 should be matched to 0.1%, and the traces routed back directly to the reference. Capacitor C6 (10nF) filters the PGA output glitches caused by sampling of the modulator. The capacitor also forms a low-pass filter on the input signal with a cut-off frequency ≈25kHz. Diode clamps protect the ADS1282 inputs from voltage transients and overloads. Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 41 ADS1282 SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 ................................................................................................................................................. www.ti.com +2.5V 15 +2.5V Test Source -2.5V 19 20 AVDD AVSS AINP2 16 AINN2 R1 100W R3 100W C2 1nF, COG Geophone R2 100W C3 1nF, COG R4 100W 17 AINP1 R5 20kW C4 10nF COG R6 20kW 18 AINN1 ADS1282 -2.5V R8 (1) 75kW +6.5V (+2.8V REF5050) 1m F 14 C6 10nF COG 13 CAPN R9 R7 1kW REF02 (REF5050) CAPP (1) 75kW 22 VREFP 1m F + C5 100mF C7 0.1mF 21 -2.5V VREFN DGND 6, 12, 25, 27 (1) Optional 20mV offset. Match to 0.1% to maintain CMR. Figure 70. Geophone Interface Application 42 Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 ADS1282 www.ti.com ................................................................................................................................................. SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 Figure 71 shows the digital connection to a field programmable gate array (FPGA) device. In this example, two ADS1282s are shown connected. The DRDY output from each ADS1282 can be used; however, when the devices are synchronized, the DRDY output from only one device is sufficient. A shared SCLK line between the devices is optional. The modulator over-range flag (MFLAG) from each device ties to the FPGA. For synchronization, one SYNC control line connects all ADS1282 devices. The RESET line also connects to all ADS1282 devices. For best performance, the FPGA and the ADS1282s should operate from the same clock. Avoid ringing on the digital inputs. 47Ω resistors in series with the digital traces can help to reduce ringing by controlling impedances. Place the resistors at the source (driver) end of the trace. Unused digital inputs should not float; tie them to DVDD or GND. This includes the modulator data pins, M0, M1, and MCLK. 4.096MHz Clock 47W +3.3V 26 (1) DVDD 1mF 28 CLK 1 ADS1282 24 RESET BYPAS DOUT DIN 1mF SCLK SYNC DGND MFLAG 4 47W RESET 47W DOUT1 47W 5 DIN1 47W 2 SCLK1 47W 10 SYNC 47W 11 CLK Input MFLAG1 4, 12, 23 +3.3V 26 (1) 1mF CLK DVDD ADS1282 RESET 28 BYPAS DOUT DIN 1mF SCLK SYNC MFLAG DGND DRDY 1 FPGA 24 4 47W DOUT2 5 DIN2 2 SCLK2 47W 10 47W 11 3 47W MFLAG2 DRDY 6, 12, 25 NOTE: Dashed line is optional. (1) For DVDD < 2.25V, see the DVDD Power Supply section. Figure 71. Microcontroller Interface with Dual ADS1282s Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 43 ADS1282 SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 ................................................................................................................................................. www.ti.com APPENDIX Table 22. FIR Stage Coefficients SECTION 1 COEFFICIENT Scaling = 1/8388608 SECTION 3 SECTION 4 Scaling = 134217728 Scaling = 134217728 LINEAR PHASE MINIMUM PHASE LINEAR PHASE MINIMUM PHASE b0 –10944 –774 –73 819 –132 11767 b1 0 0 –874 8211 –432 133882 b2 103807 8994 –4648 44880 –75 769961 b3 0 0 –16147 174712 2481 2940447 b4 –507903 –51663 –41280 536821 6692 8262605 b5 0 0 –80934 1372637 7419 17902757 b6 2512192 199523 –120064 3012996 –266 30428735 b7 4194304 0 –118690 5788605 –10663 40215494 b8 2512192 –629120 –18203 9852286 –8280 39260213 b9 0 0 224751 14957445 10620 23325925 b10 –507903 2570188 580196 20301435 22008 –1757787 b11 0 4194304 893263 24569234 348 –21028126 b12 103807 2570188 891396 26260385 –34123 –21293602 b13 0 0 293598 24247577 –25549 –3886901 b14 –10944 –629120 –987253 18356231 33460 14396783 b15 0 –2635779 9668991 61387 16314388 b16 199523 –3860322 327749 –7546 1518875 b17 0 –3572512 –7171917 –94192 –12979500 b18 –51663 –822573 –10926627 –50629 –11506007 b19 0 4669054 –10379094 101135 2769794 b20 8994 12153698 –6505618 134826 12195551 b21 0 19911100 –1333678 –56626 6103823 b22 –774 25779390 2972773 –220104 –6709466 b23 27966862 5006366 –56082 –9882714 b24 Only half shown; symmetric starting with b22. 4566808 263758 –353347 2505652 231231 8629331 126331 –215231 5597927 b27 –1496514 –430178 –4389168 b28 –1933830 34715 –7594158 b29 –1410695 580424 –428064 b30 –502731 283878 6566217 b31 245330 –588382 4024593 b32 565174 –693209 –3679749 b33 492084 366118 –5572954 b34 231656 1084786 332589 5136333 b25 b26 44 SECTION 2 b35 –9196 132893 b36 –125456 –1300087 2351253 b37 –122207 –878642 –3357202 b38 –61813 1162189 –3767666 b39 –4445 1741565 1087392 b40 22484 –522533 3847821 b41 22245 –2490395 919792 b42 10775 –688945 –2918303 Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 ADS1282 www.ti.com ................................................................................................................................................. SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 Table 22. FIR Stage Coefficients (continued) SECTION 1 SECTION 2 SECTION 3 SECTION 4 Scaling = 134217728 Scaling = 134217728 MINIMUM PHASE LINEAR PHASE MINIMUM PHASE b43 940 2811738 –2193542 b44 –2953 2425494 1493873 b45 –2599 –2338095 2595051 b46 –1052 –4511116 –79991 b47 –43 641555 –2260106 b48 214 6661730 –963855 b49 132 2950811 1482337 b50 33 –8538057 1480417 COEFFICIENT Scaling = 1/8388608 LINEAR PHASE b51 –10537298 –586408 b52 9818477 –1497356 b53 41426374 –168417 b54 56835776 1166800 b55 Only half shown; symmetric starting with b53. 644405 b56 b57 –675082 –806095 b58 211391 b59 740896 b60 141976 b61 –527673 b62 –327618 b63 278227 b64 363809 b65 –70646 b66 –304819 b67 –63159 b68 205798 b69 124363 b70 –107173 b71 –131357 b72 31104 b73 107182 b74 15644 b75 –71728 b76 –36319 b77 38331 b78 38783 b79 –13557 b80 –31453 b81 –1230 b82 20983 b83 7729 b84 –11463 b85 –8791 b86 4659 Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 45 ADS1282 SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 ................................................................................................................................................. www.ti.com Table 22. FIR Stage Coefficients (continued) SECTION 1 SECTION 2 SECTION 3 SECTION 4 Scaling = 134217728 COEFFICIENT LINEAR PHASE Scaling = 1/8388608 MINIMUM PHASE Scaling = 134217728 LINEAR PHASE b87 MINIMUM PHASE 7126 b88 –732 b89 –4687 b90 –976 b91 2551 b92 1339 b93 –1103 b94 –1085 b95 314 b96 681 b97 16 b98 –349 b99 –96 b100 144 b101 78 b102 –46 b103 –42 b104 9 b105 16 b106 0 b107 –4 1- 1-2 HPF Gain Error Factor = 2- cos wN + sin wN - 1 cos wN cos wN + sin wN - 1 cos wN (15) See the HPF Stage section for an example of how to use this equation. Table 23. tDR Time for Data Ready (Sinc Filter) (1) fDATA fCLK (1) 128k 440 64k 616 32k 968 16k 1672 8k 2824 For SYNC and Wake-Up commands, fCLK = number of CLK cycles from next rising CLK edge directly after eighth rising SCLK edge to DRDY falling edge. For Wake-Up command only, subtract two fCLK cycles. Table 23 is referenced by Table 9 and Table 11. 46 Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 ADS1282 www.ti.com ................................................................................................................................................. SBAS418D – SEPTEMBER 2007 – REVISED JULY 2008 Revision History Changes from Revision C (May 2008) to Revision D ...................................................................................................... Page • • • • • • • Changed fCLK/128 to fCLK/512 in last paragraph of Modulator section ................................................................................. Updated second paragraph of Modulator Input Impedance section.................................................................................... Updated second paragraph in Modulator Output Mode section .......................................................................................... Updated Figure 40 and added footnote 1............................................................................................................................ Updated Figure 42 ............................................................................................................................................................... Updated Table 12 and added footnote 2 ............................................................................................................................. Updated Figure 70 ............................................................................................................................................................... Submit Documentation Feedback Copyright © 2007–2008, Texas Instruments Incorporated Product Folder Link(s): ADS1282 16 17 18 18 19 27 42 47 PACKAGE OPTION ADDENDUM www.ti.com 11-Jul-2008 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Drawing Pins Package Eco Plan (2) Qty ADS1282IPW ACTIVE TSSOP PW 28 50 Green (RoHS & no Sb/Br) CU NIPDAU Level-1-260C-UNLIM ADS1282IPWG4 ACTIVE TSSOP PW 28 50 Green (RoHS & no Sb/Br) CU NIPDAU Level-1-260C-UNLIM ADS1282IPWR ACTIVE TSSOP PW 28 2000 Green (RoHS & no Sb/Br) CU NIPDAU Level-1-260C-UNLIM ADS1282IPWRG4 ACTIVE TSSOP PW 28 2000 Green (RoHS & no Sb/Br) CU NIPDAU Level-1-260C-UNLIM Lead/Ball Finish MSL Peak Temp (3) (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material) (3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. 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Addendum-Page 1 PACKAGE MATERIALS INFORMATION www.ti.com 10-Jun-2008 TAPE AND REEL INFORMATION *All dimensions are nominal Device ADS1282IPWR Package Package Pins Type Drawing TSSOP PW 28 SPQ Reel Reel Diameter Width (mm) W1 (mm) 2000 330.0 16.4 Pack Materials-Page 1 A0 (mm) B0 (mm) K0 (mm) P1 (mm) 6.95 8.3 1.6 8.0 W Pin1 (mm) Quadrant 16.0 Q1 PACKAGE MATERIALS INFORMATION www.ti.com 10-Jun-2008 *All dimensions are nominal Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm) ADS1282IPWR TSSOP PW 28 2000 346.0 346.0 33.0 Pack Materials-Page 2 MECHANICAL DATA MTSS001C – JANUARY 1995 – REVISED FEBRUARY 1999 PW (R-PDSO-G**) PLASTIC SMALL-OUTLINE PACKAGE 14 PINS SHOWN 0,30 0,19 0,65 14 0,10 M 8 0,15 NOM 4,50 4,30 6,60 6,20 Gage Plane 0,25 1 7 0°– 8° A 0,75 0,50 Seating Plane 0,15 0,05 1,20 MAX PINS ** 0,10 8 14 16 20 24 28 A MAX 3,10 5,10 5,10 6,60 7,90 9,80 A MIN 2,90 4,90 4,90 6,40 7,70 9,60 DIM 4040064/F 01/97 NOTES: A. B. C. D. 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