IRS23365DM Product Summary Features • • • • • • • • • • • • • • • • • • • Drives up to six IGBT/MOSFET power devices Gate drive supplies up to 20 V per channel Integrated bootstrap functionality Over-current protection Over-temperature shutdown input Advanced input filter Integrated deadtime protection Shoot-through (cross-conduction) protection Undervoltage lockout for VCC & VBS Enable/disable input and fault reporting Adjustable fault clear timing Separate logic and power grounds 3.3 V input logic compatible Tolerant to negative transient voltage Designed for use with bootstrap power supplies Matched propagation delays for all channels -40°C to 125°C operating range RoHS compliant Lead-Free Topology 3 Phase VOFFSET ≤ 600 V VOUT 10 V – 20 V Io+ & I o- (typical) 180 mA & 380 mA tON & tOFF (typical) 530 ns & 530 ns Deadtime (typical) 275 ns Package Options 48-Lead MLPQ7X7 (without 14 leads) Typical Applications • • • • Appliance motor drives Servo drives Micro inverter drives General purpose three phase inverters Typical Connection Diagram 1 www.irf.com © 2012 International Rectifier 3 December, 2012 IRS23365DM Description The IRS23365DM is a high voltage, high speed, power MOSFET and IGBT gate drivers with three high-side and three lowside referenced output channels for 3-phase applications. This IC is designed to be used with low-cost bootstrap power supplies; the bootstrap diode functionality has been integrated into this device to reduce the component count and the PCB size. Proprietary HVIC and latch immune CMOS technologies have been implemented in a rugged monolithic structure. The floating logic input is compatible with standard CMOS or LSTTL outputs (down to 3.3 V logic). A current trip function which terminates all six outputs can be derived from an external current sense resistor. Enable functionality is available to terminate all six outputs simultaneously. An open-drain FAULT signal is provided to indicate that a fault (e.g., over-current, overtemperature, or undervoltage shutdown event) has occurred. Fault conditions are cleared automatically after a delay programmed externally via an RC network connected to the RCIN input. The output drivers feature a high-pulse current buffer stage designed for minimum driver cross-conduction. Shoot-through protection circuitry and a minimum deadtime circuitry have been integrated into this IC. Propagation delays are matched to simplify the HVIC’s use in high frequency applications. The floating channels can be used to drive N-channel power MOSFETs or IGBTs in the high-side configuration, which operate up to 600 V. Simplified Block Diagram 2 www.irf.com © 2012 International Rectifier 3 December, 2012 IRS23365DM Typical Application Diagram DC Bus ≤ 600 V + To Load Input Voltage - IRS2336xD VCC Control Inputs, EN, & FAULT Qualification Information† Industrial Qualification Level †† Comments: This family of ICs has passed JEDEC’s Industrial qualification. IR’s Consumer qualification level is granted by extension of the higher Industrial level. ††† Moisture Sensitivity Level MLPQ7X7 Human Body Model ESD Machine Model Charged Device Model †††† IC Latch-Up Test RoHS Compliant † †† MSL3 , 260°C (per IPC/JEDEC J-STD-020) Class 1B (per JEDEC standard JESD22-A114) Class B (per EIA/JEDEC standard EIA/JESD22-A115) Class IV (per JEDEC standard JESD22-C101) Class I, Level A (per JESD78) Yes Qualification standards can be found at International Rectifier’s web site http://www.irf.com/ Higher qualification ratings may be available should the user have such requirements. Please contact your International Rectifier sales representative for further information. Higher MSL ratings may be available for the specific package types listed here. Please contact your International Rectifier sales representative for further information. Charged Device Model classification is based on SOIC28W package. ††† †††† 3 www.irf.com © 2012 International Rectifier 3 December, 2012 IRS23365DM Absolute Maximum Ratings Absolute maximum ratings indicate sustained limits beyond which damage to the device may occur. All voltage parameters are absolute voltages referenced to VSS unless otherwise stated in the table. The thermal resistance and power dissipation ratings are measured under board mounted and still air conditions. Voltage clamps are included between VCC & COM (25 V), VCC & VSS (25 V), and VB & VS (25 V). Definition Symbol VCC Low side supply voltage VIN Logic input voltage (HIN, LIN, ITRIP, EN) VRCIN VB VS VHO VLO VFLT COM dVS/dt PW HIN PD Package power dissipation @ TA ≤+25ºC RthJA TJ TS TL † RCIN input voltage High-side floating well supply voltage High-side floating well supply return voltage Floating gate drive output voltage Low-side output voltage Fault output voltage Power ground Allowable VS offset supply transient relative to VSS High-side input pulse width Thermal resistance, junction to ambient Junction temperature Storage temperature Lead temperature (soldering, 10 seconds) Min Max Units † -0.3 25 VSS-0.3 VSS+5.2 VSS-0.3 -0.3 † VB-25 VS-0.3 COM-0.3 VSS-0.3 VCC-25 — 500 — VCC+0.3 † 625 VB+0.3 VB+0.3 VCC+0.3 VCC+0.3 VCC+0.3 50 — V/ns ns 2.0 W 63 ºC/W 150 150 300 ºC — — -55 — V All supplies are tested at 25 V. An internal 25 V clamp exists for each supply. 4 www.irf.com © 2012 International Rectifier 3 December, 2012 IRS23365DM Recommended Operating Conditions For proper operation, the device should be used within the recommended conditions. All voltage parameters are absolute voltages referenced to VSS unless otherwise stated in the table. The offset rating is tested with supplies of (VCC-COM) = (VBVS) = 15 V. Definition Min Max 10 VSS 20 VSS+5 VS+10 COM-8 -50 Vs COM -5 VSS VSS VSS -40 VS+20 600 600 VB VCC 5 VCC VCC VSS+5 125 Units Symbol VCC VIN Low-side supply voltage HIN, LIN, & EN input voltage VB VS VS(t) VHO VLO COM VFLT VRCIN VITRIP TA High-side floating well supply voltage † High-side floating well supply offset voltage †† Transient high-side floating supply voltage Floating gate drive output voltage Low-side output voltage Power ground FAULT output voltage RCIN input voltage ITRIP input voltage Ambient temperature V ºC † Logic operation for VS of –8 V to 600 V. Logic state held for VS of –8 V to –VBS. Please refer to Design Tip DT97-3 for more details. †† Operational for transient negative VS of VSS - 50 V with a 50 ns pulse width. Guaranteed by design. Refer to the Application Information section of this datasheet for more details. 5 www.irf.com © 2012 International Rectifier 3 December, 2012 IRS23365DM Static Electrical Characteristics o (VCC-COM) = (VB-VS) = 15 V. TA = 25 C unless otherwise specified. The VIN and IIN parameters are referenced to VSS and are applicable to all six channels. The VO and IO parameters are referenced to respective VS and COM and are applicable to the respective output leads HO or LO. The VCCUV parameters are referenced to VSS. The VBSUV parameters are referenced to VS. Symbol VCCUV+ Definition VCC supply undervoltage positive going threshold VCC supply undervoltage negative going threshold VCC supply undervoltage hysteresis VBS supply undervoltage positive going threshold VBS supply undervoltage negative going threshold VBS supply undervoltage hysteresis High-side floating well offset supply leakage Quiescent VBS supply current Quiescent VCC supply current High level output voltage drop, VBIAS-VO Low level output voltage drop, VO Min 8 Typ 8.9 7.4 8.2 9 0.3 0.7 — 8 8.9 9.8 7.4 8.2 9 0.3 — — — — — — 0.7 — 70 2 0.90 0.40 50 120 4 1.4 0.6 Io+ Output high short circuit pulsed current 120 180 — Io- Output low short circuit pulsed current 250 380 — VIH Logic “1” input voltage 2.5 — — VIL Logic “0” input voltage Input voltage clamp (HIN, LIN, ITRIP and EN) Input bias current (HO = High) — — 0.8 4.8 5.2 5.65 IIN = 100 µA — 165 220 VIN = 0 V Input bias current (HO = Low) Input bias current (LO = High) Input bias current (LO = Low) RCIN positive going threshold RCIN hysteresis RCIN input bias current RCIN low on resistance ITRIP positive going threshold ITRIP negative going threshold ITRIP hysteresis “High” ITRIP input bias current “Low” ITRIP input bias current Enable positive going threshold Enable negative going threshold “High” enable input bias current “Low” enable input bias current FAULT low on resistance Internal BS diode Ron — — — — — — — 0.37 — — — — — 0.8 — — — — 120 165 120 8 3 — 50 0.46 0.4 0.07 5 — — — 5 — 50 200 165 220 165 — — 1 100 0.55 — — 20 1 2.5 — 20 1 100 — VCCUVVCCUVHY VBSUV+ VBSUVVBSUVHY ILK IQBS IQCC VOH VOL Max 9.8 Units Test Conditions V NA µA mA V V mA VIN,CLAMP IHIN+ IHINILIN+ ILINVRCIN,TH VRCIN,HY IRCIN RON,RCIN VIT,TH+ VIT,THVIT,HYS IITRIP+ IITRIPVEN,TH+ VEN,THIEN+ IENRON,FLT RBS 6 www.irf.com © 2012 International Rectifier VB = VS = 600 V All inputs are in the off state IO= 20 mA VO=0 V,VIN=0 V, PW ≤ 10 µs VO=15 V,VIN=5 V, PW ≤ 10 µs NA V µA VIN = 4 V VIN = 0 V VIN = 4 V V NA µA Ω V VRCIN = 0 V or 15 V I = 1.5 mA NA µA VIN = 4 V VIN = 0 V NA V µA Ω VIN = 4 V VIN = 0 V I = 1.5 mA NA 3 December, 2012 IRS23365DM Dynamic Electrical Characteristics o VCC= VB = 15 V, VS = VSS = COM, TA = 25 C, and CL = 1000 pF unless otherwise specified. Symbol tON tOFF tR tF tFIL,IN tEN tFILTER,EN tFLTCLR tITRIP Definition Turn-on propagation delay Turn-off propagation delay Turn-on rise time Turn-off fall time † Input filter time (HIN, LIN, ITRIP) Min 400 400 — — Typ 530 530 125 50 Max 750 750 190 75 200 350 510 Enable low to output shutdown propagation delay 350 460 650 Enable input filter time 100 200 — VIN = 0 V & 5 V ns VIN, VEN = 0 V or 5 V NA FAULT clear time RCIN: R = 2 MΩ, C = 1 nF 1.3 ITRIP to output shutdown propagation delay 500 750 1200 VITRIP = 5 V — 400 — VIN = 0 V or 5 V VITRIP = 5 V 1.65 2 tFLT ITRIP to FAULT propagation delay 400 600 950 Deadtime 190 — 275 — 420 DT matching ms VITRIP = 0 V ITRIP blanking time MDT Test Conditions VIN = 0 V or 5 V tBL DT Units †† VIN = 0 V & 5 V without 60 ns external deadtime MT Delay matching time (tON, tOFF) PM Pulse width distortion † ††† †† — — — — 50 75 VIN = 0 V & 5 V with external deadtime larger than DT PW input=10 µs The minimum width of the input pulse is recommended to exceed 500 ns to ensure the filtering time of the input filter is exceeded. This parameter applies to all of the channels. Please see the application section for more details. PM is defined as PW IN - PW OUT. †† ††† 7 www.irf.com © 2012 International Rectifier 3 December, 2012 IRS23365DM 8 www.irf.com © 2012 International Rectifier 3 December, 2012 IRS23365DM Functional Block Diagram: IRS23365D 9 www.irf.com © 2012 International Rectifier 3 December, 2012 IRS23365DM Input/Output Pin Equivalent Circuit Diagrams: IRS23365D VB ESD Diode 20 V Clamp HO ESD Diode V CC ESD Diode VS RPU 20 V Clamp HIN or LIN ESD Diode 600V VC C V IN Clamp ESD Diode VSS 25 V Clamp LO ESD Diode COM 10 www.irf.com © 2012 International Rectifier 3 December, 2012 IRS23365DM Lead Definitions: IRS23365DM Symbol Description VCC Low-side supply voltage VSS Logic ground VB1 High-side gate drive floating supply (phase 1) VB2 High-side gate drive floating supply (phase 2) VB3 High-side gate drive floating supply (phase 3) VS1 High voltage floating supply return (phase 1) VS2 High voltage floating supply return (phase 2) VS3 High voltage floating supply return (phase 3) HIN1/N Logic inputs for high-side gate driver outputs (phase 1); input is out-of-phase with output HIN2/N Logic inputs for high-side gate driver outputs (phase 2); input is out-of-phase with output HIN3/N Logic inputs for high-side gate driver outputs (phase 3); input is out-of-pha LIN1/N Logic inputs for low-side gate driver outputs (phase 1); input is out-of-phase with output LIN2/N Logic inputs for low-side gate driver outputs (phase 2); input is out-of-phase with output LIN3/N Logic inputs for low-side gate driver outputs (phase 3); input is out-of-phase with output se with output HO1 High-side driver outputs (phase 1) HO2 High-side driver outputs (phase 2) HO3 High-side driver outputs (phase 3) LO1 Low-side driver outputs (phase 1) LO2 Low-side driver outputs (phase 2) LO3 Low-side driver outputs (phase 3) COM Low-side gate drive return Indicates over-current, over-temperature (ITRIP), or low-side undervoltage lockout has occurred. This pin has negative logic and an open-drain output. The use of over-current and over-temperature protection requires the use of external components. Logic input to shutdown functionality. Logic functions when EN is high (i.e., positive logic). No effect on FAULT and not latched. Analog input for over-current shutdown. When active, ITRIP shuts down outputs and activates FAULT and RCIN low. When ITRIP becomes inactive, FAULT stays active low for an externally set time tFLTCLR, then automatically becomes inactive (open-drain high impedance). An external RC network input used to define the FAULT CLEAR delay (tFLTCLR) approximately equal to R*C. When RCIN > 8 V, the FAULT pin goes back into an open-drain high-impedance state. FAULT/N EN ITRIP RCIN 11 www.irf.com © 2012 International Rectifier 3 December, 2012 IRS23365DM Lead Assignments The central exposed pad (35) has to be connected to COM for better electrical performance. 12 www.irf.com © 2012 International Rectifier 3 December, 2012 IRS23365DM Application Information and Additional Details Information regarding the following topics are included as subsections within this section of the datasheet. • • • • • • • • • • • • • • • • • • • • • IGBT/MOSFET Gate Drive Switching and Timing Relationships Deadtime Matched Propagation Delays Input Logic Compatibility Undervoltage Lockout Protection Shoot-Through Protection Enable Input Fault Reporting and Programmable Fault Clear Timer Over-Current Protection Over-Temperature Shutdown Protection Truth Table: Undervoltage lockout, ITRIP, and ENABLE Advanced Input Filter Short-Pulse / Noise Rejection Integrated Bootstrap Functionality Bootstrap Power Supply Design Separate Logic and Power Grounds Tolerant to Negative VS Transients PCB Layout Tips Integrated Bootstrap FET limitation Additional Documentation IGBT/MOSFET Gate Drive The IRS23365D HVICs are designed to drive up to six MOSFET or IGBT power devices. Figures 1 and 2 illustrate several parameters associated with the gate drive functionality of the HVIC. The output current of the HVIC, used to drive the gate of the power switch, is defined as IO. The voltage that drives the gate of the external power switch is defined as VHO for the high-side power switch and VLO for the low-side power switch; this parameter is sometimes generically called VOUT and in this case does not differentiate between the high-side or low-side output voltage. VB (or VCC) VB (or VCC) IO+ HO (or LO) HO (or LO) + IO- VHO (or VLO) VS (or COM) - VS (or COM) Figure 1: HVIC sourcing current 13 www.irf.com © 2012 International Rectifier Figure 2: HVIC sinking current 3 December, 2012 IRS23365DM Switching and Timing Relationships The relationship between the input and output signals of the IRS23365D are illustrated below in Figures 3 . From these figures, we can see the definitions of several timing parameters (i.e., PW IN, PW OUT, tON, tOFF, tR, and tF) associated with this device. LINx (or HINx) 50% 50% PWIN tON LOx (or HOx) tOFF tR tF PWOUT 90% 10% 90% 10% Figure 3: Switching time waveforms The following two figures illustrate the timing relationships of some of the functionality of the IRS23365D; this functionality is described in further detail later in this document. During interval A of Figure 4, the HVIC has received the command to turn-on both the high- and low-side switches at the same time; as a result, the shoot-through protection of the HVIC has prevented this condition and both the high- and low-side output are held in the off state. Interval B of Figures 4 and 5 shows that the signal on the ITRIP input pin has gone from a low to a high state; as a result, all of the gate drive outputs have been disabled (i.e., see that HOx has returned to the low state; LOx is also held low), the voltage on the RCIN pin has been pulled to 0 V, and a fault is reported by the FAULT output transitioning to the low state. Once the ITRIP input has returned to the low state, the output will remain disabled and the fault condition reported until the voltage on the RCIN pin charges up to VRCIN,TH (see interval C in Figure 6); the charging characteristics are dictated by the RC network attached to the RCIN pin. During intervals D and E of Figure 4, we can see that the enable (EN) pin has been pulled low (as is the case when the driver IC has received a command from the control IC to shutdown); this results in the outputs (HOx and LOx) being held in the low state until the enable pin is pulled high. 14 www.irf.com © 2012 International Rectifier 3 December, 2012 IRS23365DM Figure 4: Input/output timing diagram Interval B Interval C VIT,TH- VIT,TH+ ITRIP FAULT tFLT 50% 50% RCIN VRCIN,TH 90% HOx tFLTCLR tITRIP Figure 5: Detailed view of B & C intervals Deadtime This family of HVICs features integrated deadtime protection circuitry. The deadtime for these ICs is fixed; other ICs within IR’s HVIC portfolio feature programmable deadtime for greater design flexibility. The deadtime feature inserts a time period (a minimum deadtime) in which both the high- and low-side power switches are held off; this is done to ensure that the power switch being turned off has fully turned off before the second power switch is turned on. This minimum deadtime is automatically inserted whenever the external deadtime is shorter than DT; external deadtimes larger than DT are not modified by the gate driver. Figure 6 illustrates the deadtime period and the relationship between the output gate signals. The deadtime circuitry of the IRS23365D is matched with respect to the high- and low-side outputs of a given channel; additionally, the deadtimes of each of the three channels are matched. Figure 6 defines the two deadtime parameters (i.e., DT1 and DT2) of a specific channel; the deadtime matching parameter (MDT) associated with the IRS23365D specifies the maximum difference between DT1 and DT2. The MDT parameter also applies when comparing the DT of one channel of the IRS23365D to that of another. 15 www.irf.com © 2012 International Rectifier 3 December, 2012 IRS23365DM Figure 6: Illustration of deadtime Matched Propagation Delays The IRS23365D is designed with propagation delay matching circuitry. With this feature, the IC’s response at the output to a signal at the input requires approximately the same time duration (i.e., tON, tOFF) for both the low-side channels and the highside channels; the maximum difference is specified by the delay matching parameter (MT). Additionally, the propagation delay for each low-side channel is matched when compared to the other low-side channels and the propagation delays of the highside channels are matched with each other; the MT specification applies as well. The propagation turn-on delay (tON) of the IRS23365D is matched to the propagation turn-on delay (tOFF). Input Logic Compatibility The inputs of this IC are compatible with standard CMOS and TTL outputs. The IRS23365D has been designed to be compatible with 3.3 V and 5 V logic-level signals. Figure 7 illustrates an input signal to the IRS23365D, its input threshold values, and the logic state of the IC as a result of the input signal. Figure 7: HIN & LIN input thresholds 16 www.irf.com © 2012 International Rectifier 3 December, 2012 IRS23365DM Undervoltage Lockout Protection This family of ICs provides undervoltage lockout protection on both the VCC (logic and low-side circuitry) power supply and the VBS (high-side circuitry) power supply. Figure 8 is used to illustrate this concept; VCC (or VBS) is plotted over time and as the waveform crosses the UVLO threshold (VCCUV+/- or VBSUV+/-) the undervoltage protection is enabled or disabled. Upon power-up, should the VCC voltage fail to reach the VCCUV+ threshold, the IC will not turn-on. Additionally, if the VCC voltage decreases below the VCCUV- threshold during operation, the undervoltage lockout circuitry will recognize a fault condition and shutdown the high- and low-side gate drive outputs, and the FAULT pin will transition to the low state to inform the controller of the fault condition. Upon power-up, should the VBS voltage fail to reach the VBSUV threshold, the IC will not turn-on. Additionally, if the VBS voltage decreases below the VBSUV threshold during operation, the undervoltage lockout circuitry will recognize a fault condition, and shutdown the high-side gate drive outputs of the IC. The UVLO protection ensures that the IC drives the external power devices only when the gate supply voltage is sufficient to fully enhance the power devices. Without this feature, the gates of the external power switch could be driven with a low voltage, resulting in the power switch conducting current while the channel impedance is high; this could result in very high conduction losses within the power device and could lead to power device failure. Figure 8: UVLO protection 17 www.irf.com © 2012 International Rectifier 3 December, 2012 IRS23365DM Shoot-Through Protection The IRS23365D is equipped with shoot-through protection circuitry (also known as cross-conduction prevention circuitry). Figure 9 shows how this protection circuitry prevents both the high- and low-side switches from conducting at the same time. Table 1 illustrates the input/output relationship of the devices in the form of a truth table. Figure 9: Illustration of shoot-through protection circuitry IRS23365D HIN LIN HO LO 0 0 0 0 0 1 1 0 1 0 0 1 1 1 0 0 Table 1: Input/output truth table for IRS23365D Enable Input The IRS23365D is equipped with an enable input pin that is used to shutdown or enable the HVIC. When the EN pin is in the high state the HVIC is able to operate normally (assuming no other fault conditions). When a condition occurs that should shutdown the HVIC, the EN pin should see a low logic state. The enable circuitry of the IRS23365D features an input filter; the minimum input duration is specified by tFILTER,EN. Please refer to the EN pin parameters VEN,TH+, VEN,TH-, and IEN for the details of its use. Table 2 gives a summary of this pin’s functionality and Figure 10 illustrates the outputs’ response to a shutdown command. EN VEN,TH- Enable Input tEN * Enable input high Outputs enabled Enable input low Outputs disabled Table 2: Enable functionality truth table HOx (or LOx) 90% Figure 10: Output enable timing waveform (*assumes no other fault condition) 18 www.irf.com © 2012 International Rectifier 3 December, 2012 IRS23365DM Fault Reporting and Programmable Fault Clear Timer The IRS23365D provides an integrated fault reporting output and an adjustable fault clear timer. There are two situations that would cause the HVIC to report a fault via the FAULT pin. The first is an undervoltage condition of VCC and the second is if the ITRIP pin recognizes a fault. Once the fault condition occurs, the FAULT pin is internally pulled to VSS and the fault clear timer is activated. The fault output stays in the low state until the fault condition has been removed and the fault clear timer expires; once the fault clear timer expires, the voltage on the FAULT pin will return to VCC. The length of the fault clear time period (tFLTCLR) is determined by exponential charging characteristics of the capacitor where the time constant is set by RRCIN and CRCIN. In Figure 11 where we see that a fault condition has occurred (UVLO or ITRIP), RCIN and FAULT are pulled to VSS, and once the fault has been removed, the fault clear timer begins. Figure 12 shows that RRCIN is connected between the VCC and the RCIN pin, while CRCIN is placed between the RCIN and VSS pins. Figure 11: RCIN and FAULT pin waveforms Figure 12: Programming the fault clear timer The design guidelines for this network are shown in Table 3. ≤1 nF CRCIN Ceramic 0.5 MΩ to 2 MΩ RRCIN >> RON,RCIN Table 3: Design guidelines The length of the fault clear time period can be determined by using the formula below. vC(t) = Vf(1-e -t/RC ) tFLTCLR = -(RRCINCRCIN)ln(1-VRCIN,TH/VCC) 19 www.irf.com © 2012 International Rectifier 3 December, 2012 IRS23365DM Over-Current Protection The IRS23365D HVICs are equipped with an ITRIP input pin. This functionality can be used to detect over-current events in the DC- bus. Once the HVIC detects an over-current event through the ITRIP pin, the outputs are shutdown, a fault is reported through the FAULT pin, and RCIN is pulled to VSS. The level of current at which the over-current protection is initiated is determined by the resistor network (i.e., R0, R1, and R2) connected to ITRIP as shown in Figure 13, and the ITRIP threshold (VIT,TH+). The circuit designer will need to determine the maximum allowable level of current in the DC- bus and select R0, R1, and R2 such that the voltage at node VX reaches the over-current threshold (VIT,TH+) at that current level. VIT,TH+ = R0IDC-(R1/(R1+R2)) Figure 13: Programming the over-current protection For example, a typical value for resistor R0 could be 50 mΩ. The voltage of the ITRIP pin should not be allowed to exceed 5 V; if necessary, an external voltage clamp may be used. Over-Temperature Shutdown Protection The ITRIP input of the IRS23365D can also be used to detect over-temperature events in the system and initiate a shutdown of the HVIC (and power switches) at that time. In order to use this functionality, the circuit designer will need to design the resistor network as shown in Figure 14 and select the maximum allowable temperature. This network consists of a thermistor and two standard resistors R3 and R4. As the temperature changes, the resistance of the thermistor will change; this will result in a change of voltage at node VX. The resistor values should be selected such the voltage VX should reach the threshold voltage (VIT,TH+) of the ITRIP functionality by the time that the maximum allowable temperature is reached. The voltage of the ITRIP pin should not be allowed to exceed 5 V. When using both the over-current protection and over-temperature protection with the ITRIP input, OR-ing diodes (e.g., DL4148) can be used. This network is shown in Figure 15; the OR-ing diodes have been labeled D1 and D2. 20 www.irf.com © 2012 International Rectifier 3 December, 2012 IRS23365DM Figure 14: Programming over-temperature protection Figure 15: Using over-current protection and overtemperature protection Truth Table: Undervoltage lockout, ITRIP, and ENABLE Table 4 provides the truth table for the IRS23365D. The first line shows that the UVLO for VCC has been tripped; the FAULT output has gone low and the gate drive outputs have been disabled. VCCUV is not latched in this case and when VCC is greater than VCCUV, the FAULT output returns to the high impedance state. The second case shows that the UVLO for VBS has been tripped and that the high-side gate drive outputs have been disabled. After VBS exceeds the VBSUV threshold, HO will stay low until the HVIC input receives a new rising transition of HIN. The third case shows the normal operation of the HVIC. The fourth case illustrates that the ITRIP trip threshold has been reached and that the gate drive outputs have been disabled and a fault has been reported through the fault pin. In the last case, the HVIC has received a command through the EN input to shutdown; as a result, the gate drive outputs have been disabled. UVLO VCC UVLO VBS Normal operation ITRIP fault EN command VCC <VCCUV 15 V 15 V 15 V 15 V VBS — <VBSUV 15 V 15 V 15 V ITRIP — 0V 0V >VITRIP 0V EN — 5V 5V 5V 0V RCIN High High High Low High FAULT 0 High impedance High impedance 0 High impedance LO 0 LIN LIN 0 0 HO 0 0 HIN 0 0 Table 4: IRS23365D UVLO, ITRIP, EN, RCIN, & FAULT truth table Advanced Input Filter The advanced input filter allows an improvement in the input/output pulse symmetry of the HVIC and helps to reject noise spikes and short pulses. This input filter has been applied to the HIN, LIN, and EN inputs. The working principle of the new filter is shown in Figures 16 and 17. Figure 16 shows a typical input filter and the asymmetry of the input and output. The upper pair of waveforms (Example 1) show an input signal with a duration much longer then tFIL,IN; the resulting output is approximately the difference between the input signal and tFIL,IN. The lower pair of waveforms (Example 2) show an input signal with a duration slightly longer then tFIL,IN; the resulting output is approximately the difference between the input signal and tFIL,IN. Figure 17 shows the advanced input filter and the symmetry between the input and output. The upper pair of waveforms (Example 1) show an input signal with a duration much longer then tFIL,IN; the resulting output is approximately the same duration as the input signal. The lower pair of waveforms (Example 2) show an input signal with a duration slightly longer then tFIL,IN; the resulting output is approximately the same duration as the input signal. 21 www.irf.com © 2012 International Rectifier 3 December, 2012 IRS23365DM Figure 16: Typical input filter Figure 17: Advanced input filter Short-Pulse / Noise Rejection Example 2 Example 1 This device’s input filter provides protection against short-pulses (e.g., noise) on the input lines. If the duration of the input signal is less than tFIL,IN, the output will not change states. Example 1 of Figure 18 shows the input and output in the low state with positive noise spikes of durations less than tFIL,IN; the output does not change states. Example 2 of Figure 18 shows the input and output in the high state with negative noise spikes of durations less than tFIL,IN; the output does not change states. Figure 18: Noise rejecting input filters Figures 19 and 20 present lab data that illustrates the characteristics of the input filters while receiving ON and OFF pulses. The input filter characteristic is shown in Figure 19; the left side illustrates the narrow pulse ON (short positive pulse) characteristic while the left shows the narrow pulse OFF (short negative pulse) characteristic. The x-axis of Figure 19 shows the duration of PW IN, while the y-axis shows the resulting PW OUT duration. It can be seen that for a PW IN duration less than tFIL,IN, that the resulting PW OUT duration is zero (e.g., the filter rejects the input signal/noise). We also see that once the PW IN duration exceed tFIL,IN, that the PW OUT durations mimic the PW IN durations very well over this interval with the symmetry improving as the duration increases. To ensure proper operation of the HVIC, it is suggested that the input pulse width for the high-side inputs be ≥ 500 ns. The difference between the PW OUT and PW IN signals of both the narrow ON and narrow OFF cases is shown in Figure 20; the careful reader will note the scale of the y-axis. The x-axis of Figure 20 shows the duration of PW IN, while the y-axis shows the resulting PW OUT–PW IN duration. This data illustrates the performance and near symmetry of this input filter. 22 www.irf.com © 2012 International Rectifier 3 December, 2012 IRS23365DM Figure 19: IRS23365D input filter characteristic Figure 20: Difference between the input pulse and the output pulse Integrated Bootstrap Functionality The new IRS23365D family features integrated high-voltage bootstrap MOSFETs that eliminate the need of the external bootstrap diodes and resistors in many applications. There is one bootstrap MOSFET for each high-side output channel and it is connected between the VCC supply and its respective floating supply (i.e., VB1, VB2, VB3); see Figure 21 for an illustration of this internal connection. The integrated bootstrap MOSFET is turned on only during the time when LO is ‘high’, and it has a limited source current due to RBS. The VBS voltage will be charged each cycle depending on the on-time of LO and the value of the CBS capacitor, the drain-source (collector-emitter) drop of the external IGBT (or MOSFET), and the low-side free-wheeling diode drop. The bootstrap MOSFET of each channel follows the state of the respective low-side output stage (i.e., the bootstrap MOSFET is ON when LO is high, it is OFF when LO is low), unless the VB voltage is higher than approximately 110% of VCC. In that case, the bootstrap MOSFET is designed to remain off until VB returns below that threshold; this concept is illustrated in Figure 22. 23 www.irf.com © 2012 International Rectifier 3 December, 2012 IRS23365DM VB1 VCC VB2 VB3 Figure 21: Internal bootstrap MOSFET connection Figure 22: Bootstrap MOSFET state diagram A bootstrap MOSFET is suitable for most of the PWM modulation schemes and can be used either in parallel with the external bootstrap network (i.e., diode and resistor) or as a replacement of it. The use of the integrated bootstrap as a replacement of the external bootstrap network may have some limitations. An example of this limitation may arise when this functionality is used in non-complementary PWM schemes (typically 6-step modulations) and at very high PWM duty cycle. In these cases, superior performances can be achieved by using an external bootstrap diode in parallel with the internal bootstrap network. Bootstrap Power Supply Design For information related to the design of the bootstrap power supply while using the integrated bootstrap functionality of the IRS23365D, please refer to Application Note 1123 (AN-1123) entitled “Bootstrap Network Analysis: Focusing on the Integrated Bootstrap Functionality.” This application note is available at www.irf.com. For information related to the design of a standard bootstrap power supply (i.e., using an external discrete diode) please refer to Design Tip 04-4 (DT04-4) entitled “Using Monolithic High Voltage Gate Drivers.” This design tip is available at www.irf.com. Separate Logic and Power Grounds The IRS23365D has separate logic and power ground pin (VSS and COM respectively) to eliminate some of the noise problems that can occur in power conversion applications. Current sensing shunts are commonly used in many applications for power inverter protection (i.e., over-current protection), and in the case of motor drive applications, for motor current measurements. In these situations, it is often beneficial to separate the logic and power grounds. Figure 23 shows a HVIC with separate VSS and COM pins and how these two grounds are used in the system. The VSS is used as the reference point for the logic and over-current circuitry; VX in the figure is the voltage between the ITRIP pin and the VSS pin. Alternatively, the COM pin is the reference point for the low-side gate drive circuitry. The output voltage used to drive the low-side gate is VLO-COM; the gate-emitter voltage (VGE) of the low-side switch is the output voltage of the driver minus the drop across RG,LO. 24 www.irf.com © 2012 International Rectifier 3 December, 2012 IRS23365DM DC+ BUS DBS VB (x3) VCC CBS HO (x3) HVIC ITRIP RG,HO VS (x3) VS1 LO (x3) VS3 RG,LO + + VSS VS2 VGE1 COM + VGE2 - - VGE3 - R2 R0 + VX R1 - DC- BUS Figure 23: Separate VSS and COM pins Tolerant to Negative VS Transients A common problem in today’s high-power switching converters is the transient response of the switch node’s voltage as the power switches transition on and off quickly while carrying a large current. A typical 3-phase inverter circuit is shown in Figure 24; here we define the power switches and diodes of the inverter. If the high-side switch (e.g., the IGBT Q1 in Figures 25 and 26) switches off, while the U phase current is flowing to an inductive load, a current commutation occurs from high-side switch (Q1) to the diode (D2) in parallel with the low-side switch of the same inverter leg. At the same instance, the voltage node VS1, swings from the positive DC bus voltage to the negative DC bus voltage. Figure 24: Three phase inverter 25 www.irf.com © 2012 International Rectifier 3 December, 2012 IRS23365DM DC+ BUS Q1 ON IU VS1 Q2 OFF D2 DC- BUS Figure 25: Q1 conducting Figure 26: D2 conducting Also when the V phase current flows from the inductive load back to the inverter (see Figures 27 and 28), and Q4 IGBT switches on, the current commutation occurs from D3 to Q4. At the same instance, the voltage node, VS2, swings from the positive DC bus voltage to the negative DC bus voltage. Figure 27: D3 conducting Figure 28: Q4 conducting However, in a real inverter circuit, the VS voltage swing does not stop at the level of the negative DC bus, rather it swings below the level of the negative DC bus. This undershoot voltage is called “negative VS transient”. The circuit shown in Figure 29 depicts one leg of the three phase inverter; Figures 30 and 31 show a simplified illustration of the commutation of the current between Q1 and D2. The parasitic inductances in the power circuit from the die bonding to the PCB tracks are lumped together in LC and LE for each IGBT. When the high-side switch is on, VS1 is below the DC+ voltage by the voltage drops associated with the power switch and the parasitic elements of the circuit. When the high-side power switch turns off, the load current momentarily flows in the low-side freewheeling diode due to the inductive load connected to VS1 (the load is not shown in these figures). This current flows from the DC- bus (which is connected to the COM pin of the HVIC) to the load and a negative voltage between VS1 and the DC- Bus is induced (i.e., the COM pin of the HVIC is at a higher potential than the VS pin). 26 www.irf.com © 2012 International Rectifier 3 December, 2012 IRS23365DM Figure 29: Parasitic Elements Figure 30: VS positive Figure 31: VS negative In a typical motor drive system, dV/dt is typically designed to be in the range of 3-5 V/ns. The negative VS transient voltage can exceed this range during some events such as short circuit and over-current shutdown, when di/dt is greater than in normal operation. International Rectifier’s HVICs have been designed for the robustness required in many of today’s demanding applications. The IRS23365D has been seen to withstand large negative VS transient conditions on the order of -50 V for a period of 50 ns. An illustration of the IRS23365D’s performance can be seen in Figure 32. This experiment was conducted using various loads to create this condition; the curve shown in this figure illustrates the successful operation of the IRS2336D under these stressful conditions. In case of -VS transients greater then -20 V for a period of time greater than 100 ns; the HVIC is designed to hold the high-side outputs in the off state for 4.5 µs in order to ensure that the high- and low-side power switches are not on at the same time. Figure 32: Negative VS transient results for an International Rectifier HVIC Even though the IRS23365D has been shown able to handle these large negative VS transient conditions, it is highly recommended that the circuit designer always limit the negative VS transients as much as possible by careful PCB layout and component use. 27 www.irf.com © 2012 International Rectifier 3 December, 2012 IRS23365DM PCB Layout Tips Distance between high and low voltage components: It’s strongly recommended to place the components tied to the floating voltage pins (VB and VS) near the respective high voltage portions of the device. Ground Plane: In order to minimize noise coupling, the ground plane should not be placed under or near the high voltage floating side. The central exposed pad has to be connected to COM for better electrical performance. Gate Drive Loops: Current loops behave like antennas and are able to receive and transmit EM noise (see Figure 33). In order to reduce the EM coupling and improve the power switch turn on/off performance, the gate drive loops must be reduced as much as possible. Moreover, current can be injected inside the gate drive loop via the IGBT collector-to-gate parasitic capacitance. The parasitic auto-inductance of the gate loop contributes to developing a voltage across the gate-emitter, thus increasing the possibility of a self turn-on effect. Figure 33: Antenna Loops Supply Capacitor: It is recommended to place a bypass capacitor (CIN) between the VCC and VSS pins. This connection is shown in Figure 34. A ceramic 1 µF ceramic capacitor is suitable for most applications. This component should be placed as close as possible to the pins in order to reduce parasitic elements. 28 www.irf.com © 2012 International Rectifier 3 December, 2012 IRS23365DM Figure 34: Supply capacitor Routing and Placement: Power stage PCB parasitic elements can contribute to large negative voltage transients at the switch node; it is recommended to limit the phase voltage negative transients. In order to avoid such conditions, it is recommended to 1) minimize the high-side emitter to low-side collector distance, and 2) minimize the low-side emitter to negative bus rail stray inductance. However, where negative VS spikes remain excessive, further steps may be taken to reduce the spike. This includes placing a resistor (5 Ω or less) between the VS pin and the switch node (see Figure 35), and in some cases using a clamping diode between VSS and VS (see Figure 36). See DT04-4 at www.irf.com for more detailed information. Figure 36: VS resistor Figure 37: VS clamping diode Additional Documentation Several technical documents related to the use of HVICs are available at www.irf.com; use the Site Search function and the document number to quickly locate them. Below is a short list of some of these documents. DT97-3: Managing Transients in Control IC Driven Power Stages AN-1123: Bootstrap Network Analysis: Focusing on the Integrated Bootstrap Functionality DT04-4: Using Monolithic High Voltage Gate Drivers AN-978: HV Floating MOS-Gate Driver ICs 29 www.irf.com © 2012 International Rectifier 3 December, 2012 IRS23365DM Parameter Temperature Trends 1000 1000 800 800 Exp. Exp. 600 tOFF (ns) tON (ns) Figures 38-59 provide information on the experimental performance of the IRS23365D HVIC. The line plotted in each figure is generated from actual lab data. A small number of individual samples were tested at three temperatures (-40 ºC, 25 ºC, and 125 ºC) in order to generate the experimental (Exp.) curve. The line labeled Exp. consist of three data points (one data point at each of the tested temperatures) that have been connected together to illustrate the understood temperature trend. The individual data points on the curve were determined by calculating the averaged experimental value of the parameter (for a given temperature). 400 600 400 200 200 0 -50 -25 0 25 50 75 100 0 -50 125 -25 0 Temperature (o C) 25 50 75 100 125 100 125 Temperature (o C) Figure 38: tON vs. temperature Figure 39: tOFF vs. temperature 600 1500 1200 450 tITRIP (ns) DT (ns) Exp. 300 Exp. 900 600 150 300 0 -50 -25 0 25 50 75 100 125 Temperature (o C) Figure 40: DT vs. temperature 30 www.irf.com © 2012 International Rectifier 0 -50 -25 0 25 50 75 Temperature (o C) Figure 41: tITRIP vs. temperature 3 December, 2012 IRS23365DM 1200 1000 1000 800 800 tEN (ns) tFLT (ns) Exp. 600 600 Exp. 400 400 200 200 0 -50 -25 0 25 50 75 100 0 -50 125 -25 0 Temperature (o C) 25 50 75 100 125 100 125 Temperature (o C) Figure 42: tFLT vs. temperature Figure 43: tEN vs. temperature 60 60 MDT (ns) 40 MT (ns) 40 Exp. 20 Exp. 20 0 0 -50 -25 0 25 50 75 100 125 -50 -25 0 Temperature (o C) 25 50 75 Temperature (o C) Figure 44: MT vs. temperature Figure 45: MDT vs. temperature 60 16 12 I TRIP+ (µA) PM (ns) 40 Exp. 8 20 Exp. 4 0 -50 -25 0 25 50 75 100 125 Temperature (o C) Figure 46: PM vs. temperature 31 www.irf.com © 2012 International Rectifier 0 -50 -25 0 25 50 75 100 125 Temperature (o C) Figure 47: IITRIP+ vs. temperature 3 December, 2012 IRS23365DM 5 120 100 4 IQBS (µA) I QCC (mA) 80 3 Exp. 2 60 Exp. 40 1 20 0 -50 -25 0 25 50 75 100 0 -50 125 -25 0 o 25 50 75 100 125 100 125 100 125 Temperature (o C) Temperature ( C) Figure 48: IQCC vs. temperature Figure 49: IQBS vs. temperature 0.60 0.60 0.40 0.40 IO- (A) I O+ (A) Exp. Exp. p. 0.20 0.00 -50 0.20 -25 0 25 50 75 100 0.00 -50 125 -25 0 Temperature (o C) 25 50 75 Temperature (oC) Figure 50: IO+ vs. temperature Figure 51: IO- vs. temperature 12 12 10 10 Exp. Exp. 8 VCCUV- (V) VCCUV+ (V) 8 6 4 4 2 2 0 -50 -25 0 25 50 75 100 125 Temperature (oC) Figure 52: VCCUV+ vs. temperature 32 6 www.irf.com © 2012 International Rectifier 0 -50 -25 0 25 50 75 Temperature (oC) Figure 53: VCCUV- vs. temperature 3 December, 2012 IRS23365DM 10 10 9 9 Exp. VBSUV- (V) VBSUV+ (V) Exp. 8 7 6 8 7 6 5 -50 -25 0 25 50 75 100 5 -50 125 -25 0 Temperature (o C) 25 50 75 100 125 Temperature (o C) Figure 54: VBSUV+ vs. temperature Figure 55: VBSUV- vs. temperature 800 800 600 VIT,TH- (mV) VIT,TH+ (mV) 600 EXP. p. Exp. 400 400 200 0 200 -50 -25 0 25 50 75 100 -50 125 -25 0 Figure 56: VIT,TH+ vs. temperature 80 80 RON,FLT (Ω) 100 RON,RCIN (Ω) 50 75 100 125 Figure 57: VIT,TH- vs. temperature 100 60 40 60 Exp. 40 Exp. 20 20 0 -50 0 -25 0 25 50 75 100 125 Temperature (o C) Figure 58: RON,RCIN vs. temperature 33 25 Temperature (o C) Temperature (oC) www.irf.com © 2012 International Rectifier -50 -25 0 25 50 75 100 125 Temperature (oC) Figure 59: RON,FLT vs. temperature 3 December, 2012 IRS23365DM Case outline drawing for: MLPQ7X7 34 www.irf.com © 2012 International Rectifier 3 December, 2012 IRS23365DM Tape and Reel Details: MLPQ7X7 LOADED TAPE FEED DIRECTION A B H D F C NOTE : CONTROLLING DIMENSION IN MM E G CARRIER TAPE DIMENSION FOR 48MLPQ7X7 Metric Imperial Code Min Max Min Max A 11.90 12.10 0.474 0.476 B 3.90 4.10 0.153 0.161 C 15.70 16.30 0.618 0.641 D 7.40 7.60 0.291 0.299 E 7.15 7.35 0.281 0.289 F 7.15 7.35 0.281 0.289 G 1.50 n/a 0.059 n/a H 1.50 1.60 0.059 0.062 F D C B A E G H REEL DIMENSIONS FOR 48MLPQ7X7 Metric Imperial Code Min Max Min Max A 329.60 330.25 12.976 13.001 B 20.95 21.45 0.824 0.844 C 12.80 13.20 0.503 0.519 D 1.95 2.45 0.767 0.096 E 98.00 102.00 3.858 4.015 F n/a 22.4 n/a 0.881 G 18.5 21.1 0.728 0.83 H 16.4 18.4 0.645 0.724 35 www.irf.com © 2012 International Rectifier 3 December, 2012 IRS23365DM Part Marking Information 36 www.irf.com © 2012 International Rectifier 3 December, 2012 IRS23365DM Ordering Information Standard Pack Base Part Number IRS23365D Package Type Complete Part Number Form Quantity Tube/Bulk 52 Tape and Reel 3000 IRS23365DMPbF MLPQ7x7 48L IRS23365DMTRPbF The information provided in this document is believed to be accurate and reliable. However, International Rectifier assumes no responsibility for the consequences of the use of this information. International Rectifier assumes no responsibility for any infringement of patents or of other rights of third parties which may result from the use of this information. No license is granted by implication or otherwise under any patent or patent rights of International Rectifier. The specifications mentioned in this document are subject to change without notice. This document supersedes and replaces all information previously supplied. For technical support, please contact IR’s Technical Assistance Center http://www.irf.com/technical-info/ WORLD HEADQUARTERS: 233 Kansas St., El Segundo, California 90245 Tel: (310) 252-7105 Revision History Date 09/28/11 10/11/11 03/27/2012 04/27/2012 37 Comment Original document. Iin+ and Iin- updated New datasheet format Add explanation about exposed pad www.irf.com © 2012 International Rectifier 3 December, 2012