LT1534/LT1534-1 Ultralow Noise 2A Switching Regulators U FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ DESCRIPTIO The LT ®1534/LT1534-1 are a new class of switching regulator designed to reduce conducted and radiated electromagnetic interference (EMI). Ultralow noise and EMI are achieved by providing user control of the output switch slew rates. Voltage and current slew rates can be independently programmed to optimize switcher harmonic content versus efficiency. The LT1534/LT1534-1 can reduce high frequency harmonic power by as much as 40dB with only minor losses in efficiency. The LT1534/LT1534-1 utilize a current mode architecture optimized for low noise boost topologies. The ICs include a 2A power switch along with all necessary oscillator, control and protection circuitry. Unique error amp circuitry can regulate both positive and negative voltages. The internal oscillator may be synchronized to an external clock for more accurate placement of switching harmonics. Protection features include cycle-by-cycle current limit protection, undervoltage lockout and thermal shutdown. Low minimum supply voltage and low supply current during shutdown make the LT1534/LT1534-1 well suited for portable applications. The LT1534/LT1534-1 are available in the 16-pin narrow SO package. Greatly Reduced Conducted and Radiated EMI Low Switching Harmonic Content Independent Control of Switch Voltage and Current Slew Rates 2A Current Limited Power Switch Regulates Positive and Negative Voltages 20kHz to 250kHz Oscillator Frequency Easily Synchronized to External Clock Wide Input Voltage Range: 2.7V to 23V Low Shutdown Current: 12µA Typical Easier Layout than with Conventional Switchers U APPLICATIO S ■ ■ ■ ■ ■ Precision Instrumentation Systems Isolated Supplies for Industrial Automation Medical Instruments Wireless Communications Single Board Data Acquisition Systems , LTC and LT are registered trademarks of Linear Technology Corporation. U TYPICAL APPLICATION L1 50µH 3.3V + CIN 33µF 6.3V 15 12 4 5 16.9k 6 VIN SHDN COL CT PGND LT1534-1 RVSL RT 6.8k 15nF VC GND 1,8, 9,16 C1 47µF 6.3V ×2 10Ω L2 RCSL 11 2 NFB FB 10 L3 50µH A + SYNC 3300pF 220pF 1N5817 1nF B + 5V 650mA C2 47µF 5V Output Noise (BW = 100MHz) OPTIONAL 3 14 24k 13 24k A 50mV/DIV 7.5k 7 1534 TA01 2.49k CIN: MATSUSHITA ECGCOJB330 C1, C2: MATSUSHITA ECGCOJB47O L1, L3: COILTRONICS CTX50-4 L2: COILCRAFT B08T (28nH) OR PC TRACE B 2mV/DIV 10µs/DIV 1534 TA02 Figure 1. Low Noise 3.3V to 5V Boost Converter 1 LT1534/LT1534-1 U W W W ABSOLUTE MAXIMUM RATINGS (Note 1) Input Voltage (VIN) .................................................. 30V Switch Voltage (COL) .............................................. 35V SHDN Pin Voltage .................................................... 30V Feedback Pin Current (FB) .................................... 10mA Negative Feedback Pin Current (NFB) .................. ±10mA Operating Junction Temperature Range LT1534C ................................................ 0°C to 125°C LT1534I ............................................ – 40°C to 125°C Maximum Junction Temperature .......................... 125°C Storage Temperature Range ................. – 65°C to 150°C Lead Temperature (Soldering, 10 sec.)................. 300°C W U U PACKAGE/ORDER INFORMATION ORDER PART NUMBER TOP VIEW NC 1 16 PGND COL 2 15 COL *NC 3 14 VIN LT1534CS LT1534IS ORDER PART NUMBER TOP VIEW GND 1 16 GND COL 2 15 VIN PGND 3 14 RVSL SYNC 4 13 RCSL SYNC 4 13 RVSL CT 5 12 SHDN CT 5 12 RCSL RT 6 11 VC RT 6 11 SHDN FB 7 10 NFB FB 7 10 VC NFB 8 9 GND 8 9 LT1534CS-1 LT1534IS-1 GND GND S PACKAGE 16-LEAD PLASTIC SO **FOUR CORNER PINS ARE FUSED TO INTERNAL DIE ATTACH PADDLE FOR HEAT SINKING. CONNECT THESE FOUR PINS TO EXPANDED PC LANDS FOR PROPER HEAT SINKING. S PACKAGE 16-LEAD PLASTIC SO *DO NOT CONNECT TJMAX = 125°C, θJA = 100°C/ W TJMAX = 125°C, θJA = 50°C/ W Consult factory for Military grade parts. ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C, VIN = 5V, VC = 0.9V, VFB = VREF. COL, SHDN, NFB, all other pins open unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS Supply and Protection VIN Recommended Operating Range VIN(MIN) Minimum Input Voltage ● 2.55 2.7 V IVIN Operating Supply Current 2.7V ≤ VIN ≤ 23V, RVSL, RCSL, RT = 17k ● 12 30 mA IVIN(OFF) Shutdown Supply Current 2.7V ≤ VIN ≤ 23V, VSHDN = 0V 2.7V ≤ VIN ≤ 23V, VSHDN = 0V ● 12 12 50 30 µA µA VSHDN Shutdown Threshold 2.7V ≤ VIN ≤ 23V ● 0.8 1.2 V ISHDN Shutdown Input Current ● 2.7 0.4 23 V µA –2 Error Amplifiers VREF Reference Voltage Measured at Feedback Pin ● 1.235 1.215 1.250 1.250 1.265 1.275 V V IFB Feedback Input Current VFB = VREF ● 250 900 nA FBREG Reference Voltage Line Regulation 2.7V ≤ VIN ≤ 23V ● 0.003 0.03 %/V 2 LT1534/LT1534-1 ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C, VIN = 5V, VC = 0.9V, VFB = VREF. COL, SHDN, NFB, all other pins open unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX – 2.500 – 2.420 UNITS Error Amplifiers VNFR Negative Feedback Reference Voltage Measured at Negative Feedback Pin with Feedback Pin Open ● – 2.550 INFR Negative Feedback Input Current VNFB = VNFR ● – 37 NFBREG Negative Feedback Reference Voltage Line Regulation 2.7V ≤ VIN ≤ 23V ● gm Error Amplifier Transconductance ∆IC = ±25µA V µA – 25 0.002 0.05 %/V 1100 700 1500 ● 1900 2300 µmho µmho 200 350 µA 200 350 µA IESK Error Amplifier Sink Current VFB = VREF + 150mV, VC = 0.9V, VSHDN = 1V ● 120 IESRC Error Amplifier Source Current VFB = VREF – 150mV, VC = 0.9V, VSHDN = 1V ● 120 VCLH Error Amplifier Clamp Voltage High Clamp, VFB = 1V 1.33 V VCLL Error Amplifier Clamp Voltage Low Clamp, VFB = 1.5V 0.1 V AV Error Amplifier Voltage Gain 250 V/V 250 kHz 180 Oscillator and Sync f MAX Maximum Switch Frequency f SYNC Synchronization Frequency Range RSYNC SYNC Pin Input Resistance VFBfs FB Pin Threshold for Frequency Shift fOSC = 250kHz 375 ● 5% Reduction from Nominal kHz 40 kΩ 0.4 V 91 % 200 ns Output Switches DCMAX Maximum Switch Duty Cycle RVSL = RCSL = 4.9k, fOSC = 25kHz tIBL Switch Current Limit Blanking Time BVCOL Output Switch Breakdown Voltage 2.7V ≤ VIN ≤ 23V ● RON Output Switch-On Resistance ICOL = 1.5A, Both COL Pins Tied Together ● ILIM Switch Current Limit Duty Cycle = 30% Duty Cycle = 80% ∆IIN/∆ISW Supply Current Increase During Switch-On Time ● 88 35 V 0.25 2 1.6 0.43 Ω A A 16 mA/A Slew Control VSLEWR Output Voltage Slew Rising Edge RVSL, RCSL = 17k 11 V/µs VSLEWF Output Voltage Slew Falling Edge RVSL, RCSL = 17k 14.5 V/µs ISLEWR Output Current Slew Rising Edge RVSL, RCSL = 17k 1.3 A/µs ISLEWF Output Current Slew Falling Edge RVSL, RCSL = 17k 1.3 A/µs Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. 3 LT1534/LT1534-1 U W TYPICAL PERFORMANCE CHARACTERISTICS Minimum Input Voltage (VIN) vs Temperature 0 2.70 0.8 0.7 –100 2.65 150°C – 200 2.60 2.55 125°C 0.6 SWITCH VOLTAGE (V) 25°C ∆ILIM (mA) INPUT VOLTAGE (V) Output Switch Saturation Voltage vs Switch Current Change in Maximum Switch Current (ILIM) vs Duty Cycle 85°C – 300 – 400 85°C 0.5 25°C 0.4 – 40°C 0.3 0.2 2.50 – 500 2.45 –50 –25 – 600 25 50 0 75 100 125 150 JUNCTION TEMPERATURE (°C) 0.1 0 20 0 40 60 DUTY CYCLE (%) 80 100 1.8 1.28 1.6 1.27 1.4 1.2 VFB 1.0 1.24 0.8 1.23 0.6 IFB 0.4 –2.30 0 –2.40 30 VNFB –2.45 –2.50 25 –2.55 –2.60 20 INFB –2.65 0.2 1.21 35 –2.35 NEGATIVE FB VOLTAGE (V) 2.0 1.29 1.20 –50 –25 1534 G03 –2.70 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 0 15 25 50 75 100 125 150 TEMPERATURE (°C) 1533 G04 1533 G05 Error Amplifier Transconductance vs Temperature Switching Frequency vs Feedback Pin Voltage 100 gm = ∆IVC /∆VFB SWITCHING FREQUENCY (% TYPICAL) TRANSCONDUCTANCE (mho) 1900 1800 1700 1600 1500 1400 1300 1200 1100 1000 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 1534 G08 4 Error Amplifier Output Current (VC) 500 TA = 25°C 90 400 ERROR AMPLIFIER OUTPUT (µA) 2000 NFB IINPUT CURRENT (µA) 1.30 1.22 2.0 Negative Feedback Voltage and Input Current vs Temperature FEEDBACK INPUT CURRENT (µA) FEEDBACK VOLTAGE (V) Feedback Voltage and Input Current vs Temperature 1.25 1.0 0.5 1.5 SWITCH CURRENT (A) 1533 G02 1534 G01 1.26 0 80 70 60 50 40 30 20 300 200 100 –40°C 125°C 25°C 0 –100 –200 –300 –400 10 0 0 0.1 0.3 0.4 0.5 0.2 FEEDBACK PIN VOLTAGE (V) 0.6 1533 G06 –500 –400 –300 –200 –100 0 100 200 300 400 FEEDBACK PIN VOLTAGE FROM NOMINAL (mV) 1534 G07 LT1534/LT1534-1 U W TYPICAL PERFORMANCE CHARACTERISTICS VC Pin Threshold and Clamp Voltage vs Temperature Load Transient Response 1.6 ILOAD 0.5A/DIV VC PIN VOLTAGE (V) 1.4 VC PIN CLAMP VOLTAGE 1.2 1.0 VOUT 100mV/DIV 0.8 0.6 0.4 VC PIN THRESHOLD VIN = 3.3V 500µs/DIV VOUT = 5V (NODE A) ILOAD = 0.1A TO 0.5A FIGURE 1 CIRCUIT 0.2 0 –50 –25 25 0 50 75 100 125 1534 G10 TEMPERATURE (°C) 1534 G09 Percent Change in Oscillator Frequency vs Temperature (NPO Cap and Metal Film R) Typical Output Switch Voltage and Current Slew Rates vs Slew Setting Resistance (RVSL = RCSL) 4 40 IRISE 1.5 30 VRISE 25 VFALL 3 2 20 15 1 10 CHANGE IN FREQUENCY (%) IFALL CURRENT SLEW (A/µs) 5 0 1.0 0.5 0 –0.5 –1.0 –1.5 0 1/20k 1/10k 0 1/5k 1/6.7k –2.0 –50 1/R (mmho) 1534 G11 –25 50 25 0 75 TEMPERATURE (°C) 100 125 1534 G12 Load Regulation 2.0 FIGURE 1 CIRCUIT 1.5 1.0 ∆VOUT (%) VOLTAGE SLEW (V/µs) 35 2.0 TA = 25°C 0.5 0 –0.5 –1.0 –1.5 0 100 200 300 400 500 LOAD CURRENT (mA) 600 700 1534-1 G13 5 LT1534/LT1534-1 U U U PIN FUNCTIONS (LT1534/LT1534-1) COL (Pins 2, 15/Pin 2): These two pins should be connected together externally to create the collector of the power switch. The emitter returns to PGND through a sense resistor. Large currents flow into these pins so it is desirable to keep external trace lengths short to minimize radiation. SYNC (Pin 4): The SYNC pin can be used to synchronize the oscillator to an external clock (see Oscillator Sync in Applications Information section for more details). The SYNC pin may either be floated or tied to ground if not used. CT (Pin 5): The oscillator capacitor pin is used in conjunction with RT to set the oscillator frequency. For RT = 16.9k, CT(NF) = 129/fOSC(kHz) RT (Pin 6): The oscillator resistor pin is used to set the charge and discharge currents of the oscillator capacitor. The nominal value is 16.9k. It is possible to adjust this resistance ±25% to get a more accurate oscillator frequency. FB (Pin 7): The feedback pin is used for positive voltage sensing and oscillator frequency shifting during start-up and short-circuit conditions. It is the inverting input to the error amplifier. The noninverting input of this amplifier connects internally to a 1.25V reference. This pin should be left open if not used. NFB (Pin 8/Pin 10): The negative voltage feedback pin is used for sensing a negative output voltage. The pin is connected to the inverting input of the negative feedback amplifier through a 100k source resistor. The negative feedback amplifier provides a gain of – 0.5 to the feedback amplifier; therefore, the nominal regulation point is – 2.5V on NFB. This pin should be left open if not used. GND (Pin 9/Pins 1, 8, 9, 16): Signal Ground. The internal error amplifier, negative feedback amplifier, oscillator, slew control circuitry and the bandgap reference are 6 referred to this ground. Keep the connection to the feedback divider and VC compensation network free of large ground currents. VC (Pin 10/Pin 11): The compensation pin is used for frequency compensation and current limiting. It is the output of the error amplifier and the input of the current comparator. Loop frequency compensation can be performed with an RC network connected from the VC pin to ground. SHDN (Pin 11/Pin 12): The shutdown pin is used for disabling the switcher. Grounding this pin will disable all internal circuitry. Normally this output can be tied high (to VIN) or may be left floating. RCSL (Pin 12/Pin 13): A resistor to ground sets the current slew rate for the power switch. The minimum resistor value is 3.9k and the maximum value is 68k. Current slew will be approximately: ISLEW(A/µs) = 33/RCSL(kΩ) RVSL (Pin 13/Pin 14): A resistor to ground sets the voltage slew rate for the power switch collector. The minimum resistor value is 3.9k and the maximum value is 68k. Voltage slew will be approximately: VSLEW(V/µs) = 220/RVSL(kΩ) VIN (Pin 14/Pin 15): Input Supply Pin. Bypass this pin with a ≥ 4.7µF low ESR capacitor. When VIN is below 2.55V the part will go into undervoltage lockout where it will stop output switching and pull the VC pin low. PGND (Pin 16/Pin 3): Power Switch Ground. This ground comes from the emitters of the power switches. In normal operation this pin should have approximately 25nH inductance to ground. This can be done by trace inductance (approximately 1") or with wire or a specific inductive component (e.g., small ferrite bead). This inductance ensures stability in the current slew control loop during turn-off. Too much inductance (>50nH) may produce oscillation on the output voltage slew edges. LT1534/LT1534-1 W BLOCK DIAGRA SHDN VIN PGND COL COL VC LDO REGULATOR + NEGATIVE FEEDBACK AMP INTERNAL VCC – 100k + OUTPUT DRIVER 50k NFB – – RVSL SLEW CONTROL – FB gm ERROR AMP + + RCSL COMP + S 1.25V Q FF R RT OSCILLATOR CT SYNC GND 1534 BD U OPERATIO In noise sensitive applications, switching regulators tend to be ruled out as a power supply option due to their propensity for generating unwanted noise. When switching supplies are required due to efficiency or input/output voltage constraints, great pains must be taken to work around the noise generated by a typical supply. These steps may include precise synchronization of the power supply oscillator to an external clock, synchronizing the rest of the circuit to the power supply oscillator, or halting power supply switching during noise sensitive operations. The LT1534 greatly simplifies the task of eliminating supply noise by enabling the design of an inherently low noise switching regulator power supply. The LT1534 is a fixed frequency, current mode switching regulator with unique circuitry to control the voltage and current slew rates of the output switch. Slew control capability provides much greater control over power supply components that can create conducted and radiated electromagnetic interference. The current mode control provides excellent AC and DC line regulation and simplifies loop compensation. Current Mode Control A switching cycle begins with an oscillator discharge pulse which resets the RS flip-flop, turning on the output driver (refer to Block Diagram). The switch current is sensed across an internal resistor and the resulting voltage is amplified and compared to the output of the error amplifier (VC pin). The driver is turned off once the output of the current sense amplifier exceeds the voltage on the VC pin. Internal slope compensation is provided to ensure stability under high duty cycle conditions. Output regulation is obtained using the error amp to set the switch current trip point. The error amp is a transconductance amplifier that integrates the difference between the feedback output voltage and an internal 1.25V reference. The output of the error amp adjusts the switch current trip point to provide the required load current at the desired regulated output voltage. This method of controlling current rather than voltage provides faster input transient response, cycle by cycle current limiting for better output switch protection and greater ease in compensating the feedback loop. The VC pin serves three different purposes. It is used for loop compensation, current limit adjustment and soft starting. During normal operation the VC voltage will be between 0.2V and 1.33V. An external clamp may be used for lowering the current limit. A capacitor coupled to an external clamp can be used for soft starting. 7 LT1534/LT1534-1 U OPERATIO The negative feedback amplifier allows for direct regulation of negative output voltages. The voltage on the NFB pin gets amplified by a gain of – 0.5 and driven onto the FB input, i.e., the NFB pin regulates to – 2.5V while the amplifier output internally drives the FB pin to 1.25V as in normal operation. The negative feedback amplifier input impedance is 100k (typ) referred to ground. Slew Control Control of output voltage and current slew rates is done via two feedback loops. One loop controls the output switch collector voltage dV/dt and the other loop controls the emitter current dI/dt. Output slew control is achieved by comparing the currents generated by these two slewing events to currents created by external resistors RVSL and RCSL. The two control loops are combined internally to provide a smooth transition from current slew control to voltage slew control. Internal Regulator Most of the control circuitry operates from an internal 2.4V low dropout regulator that is powered from VIN. The internal low dropout design allows VIN to vary from 2.7V to 23V with virtually no change in device performance. When the part is put into shutdown, the internal regulator is turned off, leaving only a small (12µA typ) current drain from VIN. Protection Features There are three modes of protection in the LT1534. The first is overcurrent limit. This is achieved via the clamping action of the VC pin. The second is thermal shutdown that disables both output drivers and pulls the VC pin low in the event of excessive chip temperature. The third is undervoltage lockout that also disables both outputs and pulls the VC pin low whenever VIN drops below 2.5V. U W U U APPLICATIONS INFORMATION Reducing EMI from switching power supplies has traditionally invoked fear in designers. Many switchers are designed solely on efficiency and as such produce waveforms filled with high frequency harmonics that then propagate through the rest of the power supply. The LT1534 provides control over two of the more important variables for controlling EMI with switching inductive loads: switch voltage slew rate and switch current slew rate. The use of this part will reduce noise and EMI over conventional switch mode controllers. Because these variables are under control, a supply built with this part will exhibit far less tendency to create EMI and less chance of wandering into problems during production. It is beyond the scope of this data sheet to get into EMI fundamentals. AN70 contains much information concerning noise in switching regulators and should be consulted. Oscillator Frequency The oscillator determines the switching frequency and therefore the fundamental positioning of all harmonics. The use of good quality external components is important to ensure oscillator frequency stability. The oscillator is a 8 sawtooth design. A current defined by external resistor RT is used to charge and discharge the capacitor CT. The discharge rate is approximately ten times the charge rate. By allowing the user to have control over both components, trimming of oscillator frequency can be more easily achieved. The external capacitance CT is chosen by: CT(nF) = 2180/[fOSC(kHz) • RT(kΩ)] where fOSC is the desired oscillator frequency in kHz. For RT equal to 16.9k, this simplifies to: CT(nF) = 129/fOSC(kHz) (e.g., CT = 1.29nF for fOSC = 100kHz) A good quality temperature stable capacitor should be chosen. Nominally RT should be 16.9k. Since it sets up current, its temperature coefficient should be selected to compliment the capacitor. Ideally, both should have low temperature coefficients. LT1534/LT1534-1 U W U U APPLICATIONS INFORMATION If the FB pin is below 0.4V the oscillator discharge time will increase, causing the oscillation frequency to decrease by approximately 6:1. This feature helps minimize power dissipation during start-up and short-circuit conditions. Oscillator frequency is important for noise reduction in two ways: 1) the lower the oscillator frequency the lower the harmonics of waveforms are, making it easier to filter them, 2) the oscillator will control the placement of output frequency harmonics which can aid in specific problems where you might be trying to avoid a certain frequency bandwidth that is used for detection elsewhere. Oscillator Sync If a more precise frequency is desired (e.g., to accurately place harmonics) the oscillator can be synchronized to an external clock. Set the RC timing components for an oscillator frequency 10% lower than the desired sync frequency. Drive the SYNC pin with a square wave (with greater than 1.4V amplitude). The rising edge of the sync square wave will initiate clock discharge. The sync pulse should have a minimum of 0.5µs pulse width. Be careful in synchronizing to frequencies much different from the part since the internal oscillator charge slope determines slope compensation. It would be possible to get into subharmonic oscillation if the sync doesn’t allow for the charge cycle of the capacitor to initiate slope compensation. In general, this will not be a problem until the sync frequency is greater than 1.5 times the oscillator free-run frequency. Usually it will be desirable to keep the voltage and current slew resistors approximately the same. There are circumstances where a better optimization can be found by adjusting each separately, but as these values are separated further, a loss of independence of control will occur. Starting from the lowest resistor setting adjust the pots until the noise level meets your guidelines. Note that slower slewing waveforms will dissipate more power so that efficiency will drop. You can also monitor this as you make your slew adjustment. It is possible to use a single slew setting resistor. In this case the RVSL and RCSL pins are tied together. A resistor with a value of 2k to 34k (one half the individual resistors) can then be tied from these pins to ground. Emitter Inductance A small inductance in the power ground minimizes a potential dip in the output current falling edge that can occur under fast slewing, 25nH is usually sufficient. Greater than 50nH may produce unwanted oscillations in the voltage output. The inductance can be created by wire or board trace with the equivalent of one inch of straight length. A spiral board trace will require less length. Positive Output Voltage Setting Sensing of a positive output voltage is usually done using a resistor divider from the output to the FB pin. The positive input to the error amp is connected internally to a 1.25V bandgap reference. The FB pin will regulate to this voltage. R1 VOUT FB PIN Slew Rate Setting R2 Setting the voltage and current slew rates is easy. External resistors to ground on the RVSL and RCSL pins determine the slew rates. Determining what slew rate to use is more difficult. There are several ways to approach the problem. First start by putting a 50k resistor pot with a 3.9k series resistance on each pin. In general, the next step will be to monitor the noise that you are concerned with. Be careful in measurement technique (consult AN70). Keep probe ground leads very short. 1534 F01 Figure 2 Referring to Figure 2, R1 is determined by: V R1 = R2 OUT − 1 1.25 The FB bias current represents a small error and can usually be ignored for values of R1|| R2 up to 10k. 9 LT1534/LT1534-1 U U W U APPLICATIONS INFORMATION One word of caution. Sometimes a feedback zero is added to the control loop by placing a capacitor across R1 above. If the feedback zero capacitively pulls the FB pin above the internal regulator voltage (2.4V typ), output regulation may be disrupted. A series resistance with the feedback pin can eliminate this potential problem. Thermal Considerations Negative Output Voltage Setting Power dissipation is a function of topology, input voltage, switch current and slew rates. It is impractical to come up with an all-encompassing formula. It is therefore recommended that package temperature be measured in each application. The part has an internal thermal shutdown to prevent device destruction, but this should not replace careful thermal design. Negative output voltage can be sensed using the NFB pin. In this case regulation will occur when the NFB pin is at – 2.5V. The input bias current for the NFB pin is –25µA (INFB) and must be accounted for when selecting divider resistor values. Computing power dissipation for this IC requires careful attention to detail. Reduced output slewing causes the part to dissipate more power than would occur with fast edges. However, much improvement in noise can be produced with modest decrease in supply efficiency. R1 –VOUT NFB PIN INFB R2 1534 F02 Figure 3 Referring to Figure 3, R1 is chosen such that: R1 = R2 • VOUT − 2.5 2.5 + R2 • 25µA A suggested value for R2 is 2.5k. The NFB pin is normally left open if the FB pin is being used. 1. Dissipation due to input current: I PVIN = VIN11mA + 60 where I is the average switch current. 2. Dissipation due to the driver saturation: PVSAT = (VSAT)(I)(DCMAX) where VSAT is the output saturation voltage which is approximately 0.1 + (0.2)(I), DCMAX is the maximum duty cycle. 3. Dissipation due to output slew using approximations for slew rates: Dual Polarity Output Voltage Sensing Certain applications may benefit from sensing both positive and negative output voltages. When doing this each output voltage resistor divider is individually set as previously described. When both FB and NFB pins are used, the LT1534 will act to prevent either output from going beyond its set output voltage. The highest output (lightest load) will dominate control of the regulator. This technique would prevent either output from going unregulated high at no load. However, this technique will also compromise output load regulation. Shutdown If the shutdown pin is pulled low, the regulator will turn off. The supply current will be reduced to less than 20µA. 10 2 2 2 V I2 + ∆I VIN − VSAT I IN 4 4 PSLEW = RCSL + RVSL fOSC 9 220 109 33 10 ( ) ( ) ( ) () ( ) ( )( ) Note if VSAT and ∆I are small with respect to VIN and I, then: ()( ) ( )( ( ) ( ) VIN R VSL I RCSL + PSLEW = fOSC VIN I 9 9 33 10 220 10 ) ( )( )() LT1534/LT1534-1 U U W U APPLICATIONS INFORMATION where ∆I is the ripple current in the switch, RCSL and RVSL are the slew resistors and fOSC is the oscillator frequency. Power dissipation PD is the sum of these three terms. Die junction temperature is then computed as: poor quality (high ESR) output capacitors are used. The addition of a 0.0047µF capacitor on the VC pin reduces switching frequency ripple to only a few millivolts. A low value for RVC will also reduce VC pin ripple, but loop phase margin may be inadequate. TJ = TAMB + (PD)(θJA) RVC 2k where TAMB is ambient temperature and θJA is the package thermal resistance. For the 16-pin SO with fused leads the θJA is 50°C/W. For example, with fOSC = 40kHz, 0.4A average current and 0.1A of ripple, the maximum duty cycle is 88%. Assume slew resistors are both 17k and VSAT is 0.26V, then: PD = 0.176W + 0.094W + 0.158W = 0.429W In an S16 fused lead package the die junction temperature would be 21°C above ambient. Frequency Compensation Loop frequency compensation is accomplished by way of a series RC network on the output of the error amplifier (VC pin). Referring to Figure 4, the main pole is formed by capacitor CVC and the output impedance of the error amplifier (approximately 400kΩ). The series resistor RVC creates a “zero” which improves loop stability and transient response. A second capacitor CVC2, typically onetenth the size of the main compensation capacitor, is sometimes used to reduce the switching frequency ripple on the VC pin. VC pin ripple is caused by output voltage ripple attenuated by the output divider and multiplied by the error amplifier. Without the second capacitor, VC pin ripple is: VCPIN RIPPLE = (1. 25)(VRIPPLE)(gm)(RVC) VOUT where VRIPPLE = Output ripple (VP-P) gm = Error amplifier transconductance RVC = Series resistor on VC pin VOUT = DC output voltage To prevent irregular switching, VC pin ripple should be kept below 50mVP-P. Worst-case VC pin ripple occurs at maximum output load current and will also be increased if CVC 0.01µF VC PIN CVC2 4.7nF 1534 F03 Figure 4 Capacitors While the IC reduces the source of switcher noise, it is essential for the lowest noise, that the filter capacitors should have low parasitic impedance. Sanyo OS-CON, Panasonic Specialty Polymer and tantalum capacitors are the preferred types. Aluminum electrolytics are not suitable for this application. In general, ESR is more critical than capacitance. At higher frequencies, ESL can also be important. Paralleling capacitors can reduce both ESR and ESL. Design Note 95 offers more information about capacitor selection. The following is a brief summary: Solid tantalum capacitors have small size and low impedance. Typically they are available for voltages below 50V. They may have a problem with surge currents (AVX TPS line addresses this issue). OS-CON capacitors have very low impedance but are only available for 25V or less. Form factor may be a problem. Sometimes their very low ESR can cause loop stability problems. Ceramic capacitors are generally used for high frequency and high voltage bypass. They too can have such a low ESR as to cause loop stability problems. Often they can resonate with their ESL before ESR becomes effective. Specialty Polymer Aluminum: Panasonic has come out with their series CD capacitors. While they are only available for voltages below 16V, they have very low ESR and good surge capability. 11 LT1534/LT1534-1 U W U U APPLICATIONS INFORMATION Input Capacitor Fast Voltage Slew Edges The ESR of this capacitor acts with high frequency current components to produce much of the conducted noise of the switcher. Values of 1µF to 47µF are typical with ESR less than 0.3Ω. Place the capacitor close to the IC and inductor. A very fast voltage slew under certain operating conditions may produce ringing on the COL voltage waveform. While there is small harmonic energy in this, it can be eliminated by placing an RC network of 10Ω in series with 1000pF from the COL pin to ground. The input capacitor can see a high surge current when a battery of high capacitance source is connected “live.” Some solid tantalum capacitors can fail under this condition. Several manufacturers have developed a line of solid tantalum capacitors specially tested for surge capability (e.g., AVX TPS series). However, even these units may fail if the input voltage approaches the maximum voltage rating of the capacitor. AVX recommends derating capacitor voltage by 2:1 for high surge applications. Switching Diodes Output Filter Capacitor Output capacitors are usually chosen on the basis of ESR since this will determine output ripple. However, low ESR is also needed for low output noise and this will typically be the tougher requirement. Typically required ESR will be less than 0.2Ω . Typical capacitance values are in the 47µF to 500µF range. Again keep connection length as short as possible. Table 1 shows some typical surface mount capacitors. Table 1 ESR (MAX Ω) SIZE CAPACITOR E CASE AVX TPS, Sprague 593D 0.1 to 0.3 AVX TAJ 0.7 to 0.9 AVX TPS, Sprague 593D 0.1 to 0.3 D CASE AVX TAJ Panasonic CD C CASE B CASE 12 0.9 to 2.0 0.05 to 0.18 AVX TPS 0.2 (Typ) AVX TAJ 1.8 to 3.0 AVX TAJ 2.5 to 10 In general, switching diodes should be Schottky diodes such as 1N5817-19 or MBR320-330. Choosing the Inductor For a boost converter, inductor selection involves tradeoffs of size, maximum output power, transient response and filtering characteristics. Higher inductor values provide more output power and lower input ripple. However, they are physically larger and can impede transient response. Low inductor values have high magnetizing current, which can reduce maximum power and increase input current ripple. The following procedure can be used to handle these trade-offs: 1. Assume that the average inductor current for a boost converter is equal to load current times VOUT/VIN and decide whether the inductor must withstand continuous overload conditions. If average inductor current at maximum load current is 0.5A, for instance, a 0.5A inductor may not survive a continuous 1.5A overload condition. Also be aware that boost converters are not short-circuit protected, and under output short conditions, only the available current of the input supply limits inductor current. LT1534/LT1534-1 U U W U APPLICATIONS INFORMATION 2. Calculate peak inductor current at full load current to ensure that the inductor will not saturate. Peak current can be significantly higher than output current, especially with smaller inductors and lighter loads, so don’t omit this step. Powdered iron cores are forgiving because they saturate softly, whereas ferrite cores saturate abruptly. Other core material falls in between. The following formula assumes continuous mode operation but it errors only slightly on the high side for discontinuous mode, so it can be used for all conditions. ( 3. Choose a core geometry. For low EMI problems a closed structure should be used such as a pot core, ER core, E core or toroid (see AN70 appendix I). 4. Select an inductor that can handle peak current, average current (heating effects) and fault current. 5. Finally, double check output voltage ripple. The experts in the Linear Technology Applications department have experience with a wide range of inductor types and can assist you in making a good choice. ) VIN VOUT – VIN V IPEAK = IOUT OUT + VIN 2 • • • L f V OUT Further Help AN70 has more information on noise in switching regulators and its measurement. AN19 has general information on switcher design. The Linear Technology applications group is always ready to lend a helping hand. L = inductance value VIN = supply voltage VOUT = output voltage I = output current f = oscillator frequency U TYPICAL APPLICATIO S Low Noise ±12V Dual Output Flyback Converter with Dual Polarity Output Voltage Sensing L1 50µH MBR330 T1 2.7V TO 10V 1 10µF 7 + P6KE-20A 10 4 2 8 9 3 1N4148 12 4 15 VIN SHDN 16.9k 6 CT 0.047µF + C2 47µF 47µF 3 L2 50µH NOTE 1 – 12V 50mA OPTIONAL 14 1nF RT 10k 47µF 10Ω LT1534-1 RVSL VC + C1 47µF MBR330 PGND RCSL 11 2 SYNC 2700pF 5 + COL 12V 50mA GND 1,8, 9,16 NFB FB 13 21.5k 1% 7 2.49k 1% 10 2.49k 1% 9.31k 1% T1: PHILIPS EFD-15-3F3 CORE GAP FOR PRIMARY L = 100µH PIN 1 TO TO 10, 16 TURNS 24AWG PIN 3 TO 8, 45 TURNS 38AWG NOTE 1: 25nH TRACE INDUCTANCE OR COILCRAFT B10T L1, L2: COILTRONICS CTX50-4 PIN 4 TO 7, 45 TURNS 36AWG PIN 2 TO 9, 16 TURNS 24AWG 1534 TA04 13 LT1534/LT1534-1 U TYPICAL APPLICATIO S Ultralow Noise Regulator for a Thermo-Electric Cooler, Maintaining Sensitive Electronics at Low Temperatures L3 22µH R1 50mΩ 12V 0.1µF C2 + 47µF 16V C1 + 47µF 16V D1 1000pF – U2A + L1 100µH 10Ω 1 TO 5 R2 10k + 10k L4 22µH + 6 TO 10 100Ω COL VIN 3.3k RCSL RT 2200pF GND R4 5.1k 18k R5 1.6k 10k 499k R6 25k 100k 24k Q5 2N3904 RVSL C4 22µF 16V 499k + U2B – 2N3904 FB CT 0.1µF 1k VC LT1534-1 U1 110k 10k 0A TO 1.5A THERMOELECTRIC COOLER 0.01µF L2 22nF PGND C3 22µF 16V 10k R3 690k 5.6V 0.1µF D1: MOTOROLA MBR5320T3 L1: MIDCOM 38440 L2: COILCRAFT B07T L3, L4: SUMIDA CD43-220 U2A, U2B: LT1490 3.3M RT 10k NTC 1534 TA05 Ultralow Noise 5V to –3V Cuk Converter INPUT VOLTAGE 4V TO 12V –3V 0.7A 4.7µF 12 4 SHDN COL 16.9k 6 PGND CT LT1534-1 RVSL RT RCSL 11 L1B 20µH VC GND 1,8, 9,16 2k 33nF 2 NFB FB 3 NOTE 1 14 C1 47µF MBR320 13 7 10 4.99k 1% 1k 1% NOTE 1: 25nH TRACE INDUCTANCE OR COILCRAFT B10T L1: COILTRONICS CTX20-4 14 4.7µF SYNC 1300pF 5 L1A 20µH 15 VIN 1534 TA06 A Cuk converter is a natural topology for a low noise converter. The Cuk converter is a dual of a buck boost converter. C1 is the primary means of storing and transferring energy. Like a buck boost, the DC transfer function is approximately VOUT/VIN = DC/(1 – DC). The output voltage, though negative, can be higher or lower in magnitude from the input. The two inductors can be separate however, by placing them on the same winding input and output current ripple can be greatly reduced. The additional slew control provided by the LT1534 will reduce the high frequency content even further. LT1534/LT1534-1 U PACKAGE DESCRIPTION Dimensions in inches (millimeters) unless otherwise noted. S Package 16-Lead Plastic Small Outline (Narrow 0.150) (LTC DWG # 05-08-1610) 0.386 – 0.394* (9.804 – 10.008) 16 15 14 13 12 11 10 9 0.150 – 0.157** (3.810 – 3.988) 0.228 – 0.244 (5.791 – 6.197) 1 0.010 – 0.020 × 45° (0.254 – 0.508) 2 3 4 5 6 0.053 – 0.069 (1.346 – 1.752) 0.008 – 0.010 (0.203 – 0.254) 0.014 – 0.019 (0.355 – 0.483) TYP 8 0.004 – 0.010 (0.101 – 0.254) 0° – 8° TYP 0.016 – 0.050 (0.406 – 1.270) 7 0.050 (1.270) BSC S16 1098 *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 15 LT1534/LT1534-1 U TYPICAL APPLICATION Low Noise Wide Input Range ±5V Supply C2 10µF 16V + C1 10µF 16V 1 15 12 10 11 6.8k 220pF VIN COL SYNC L2 28nH LT1534-1 VC RT 0.01µF 16.9k 6 PGND FB CT 5 GND RVSL 3 7 RCSL 1,8 14 9,16 4k TO 1500pF 25k VOUT1 5V 150mA* 6 4 2 3 12 5 9 + C4 47µF 6.3V 2 SHDN NFB T1 + 4 1N5817 + VIN 3V TO 12V 4k TO 25k C3 10µF 1nF 16V 11 1N5817 T1 8 10 7.5k VOUT1 2.49k 13 10Ω + C5 47µF 6.3V T1 7 *TOTAL OUTPUT CURRENT ≤ 300mA C1, C2, C3: MATSUSHITA ECGCICB6R8 C4, C5: MATSUSHITA ECGC0JB470 L2: COILCRAFT B08T OR PC TRACE T1: COILTRONICS VP2-0216 VOUT2 – 5V 150mA* 1534 TA03 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1129 700mA Micropower Low Dropout Regulator 0.4V Dropout Voltage, Reverse Battery Protection LT1175 500mA Negative Low Dropout Micropower Regulator Positive or Negative Shutdown Logic LT1370 500kHz High Efficiency 6A Switching Regulator 90% Efficiency, Constant Frequency, High Power LT1371 500kHz High Efficiency 3A Switching Regulator 90% Efficiency, Constant Frequency, Synchronizable LT1377 1MHz High Efficiency 1.5A Switching Regulator High Frequency, Small Inductor LT1425 Isolated Flyback Switching Regulator Excellent Regulation Without Transformer “Third Winding” LT1533 Ultralow Noise 1A Switching Regulator Push-Pull Design for Low Noise Isolated Supplies LT1763 500mA Low Noise Micropower LDO 20µVRMS(10Hz to 100kHz), 30µA Quiescent Current LT1777 700mA Low Noise Step-Down Switching Regulator Programmable dI/dt Limit, 48VMax VIN 16 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408)432-1900 ● FAX: (408) 434-0507 ● www.linear-tech.com 1534fa LT/TP 0300 2K REV A • PRINTED IN USA LINEAR TECHNOLOGY CORPORATION 1998