TI1 LM34926SDX/NOPB Lm34926 integrated secondary-side bias regulator for isolated dc-dc converter Datasheet

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LM34926
SNVS847E – JUNE 2012 – REVISED DECEMBER 2014
LM34926 Integrated Secondary-Side Bias Regulator for Isolated DC-DC Converters
1 Features
3 Description
•
•
The LM34926 regulator features all of the functions
needed to implement a low-cost, efficient, isolated
bias regulator. This high-voltage regulator contains
two 100-V N-channel MOSFET switches — a highside buck switch and a low-side synchronous switch.
The constant on-time (COT) control scheme
employed in the LM34926 requires no loop
compensation and provides excellent transient
response. The regulator operates with an on-time
control that is inversely proportional to the input
voltage. This feature allows the operating frequency
to remain relatively constant. An intelligent peak
current limit is implemented with integrated sense
circuit. Other features include a programmable input
undervoltage comparator to inhibit operation during
low-voltage conditions. Protection features include
thermal shutdown and VCC Undervoltage Lockout
(UVLO). The LM34926 device is offered in WSON-8
and SO PowerPAD-8 plastic packages.
1
•
•
•
•
•
•
•
•
•
•
•
•
•
Wide 7.5-V to 100-V Input Range
Integrated 300-mA High-Side
and Low-Side Switches
No Schottky Required
Constant On-Time Control
No Loop Compensation Required
Ultra-Fast Transient Response
Nearly Constant Operating Frequency
Intelligent Peak Current Limit
Adjustable Output Voltage From 1.225 V
Precision 2% Feedback Reference
Frequency Adjustable to 1 MHz
Adjustable Undervoltage Lockout (UVLO)
Remote Shutdown
Thermal Shutdown
Packages:
– WSON-8
– SO PowerPAD™-8 Packages
Device Information(1)
PART NUMBER
LM34926
2 Applications
•
•
PACKAGE
BODY SIZE (NOM)
HSOP (8)
4.89 mm x 3.90 mm
WSON (8)
4.00 mm x 4.00 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
Isolated Telecom Bias Supply
Isolated Automotive and Industrial Electronics
Typical Application
VOUT2
D1
+
LM34926
VIN
7.5V-100V
CIN
2
+
4
RUV2
RON
3
BST
VIN
SW
RON
VCC
UVLO
FB
RUV1
RTN
1
NS
COUT2
7
+
8
X1
CBST
VOUT1
NP
6
D2
5
+
CVCC
Rr
RFB2
+
RFB1
COUT1
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
LM34926
SNVS847E – JUNE 2012 – REVISED DECEMBER 2014
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Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
4
6.1
6.2
6.3
6.4
6.5
6.6
6.7
4
4
4
4
5
6
6
Absolute Maximum Ratings .....................................
Handling Ratings.......................................................
Recommended Operating Ratings............................
Thermal Information ..................................................
Electrical Characteristics...........................................
Switching Characteristics ..........................................
Typical Characteristics ..............................................
Detailed Description .............................................. 8
7.1 Overview ................................................................... 8
7.2 Functional Block Diagram ......................................... 8
7.3 Feature Description................................................... 9
7.4 Device Functional Modes........................................ 13
8
Application and Implementation ........................ 14
8.1 Application Information............................................ 14
8.2 Typical Application .................................................. 14
9 Power Supply Recommendations...................... 20
10 Layout................................................................... 20
10.1 Layout Guidelines ................................................. 20
10.2 Layout Example .................................................... 20
11 Device and Documentation Support ................. 21
11.1 Trademarks ........................................................... 21
11.2 Electrostatic Discharge Caution ............................ 21
11.3 Glossary ................................................................ 21
12 Mechanical, Packaging, and Orderable
Information ........................................................... 21
4 Revision History
Changes from Revision D (December 2013) to Revision E
Page
•
Added Pin Configuration and Functions section, Handling Rating table, Switching Characteristics table, Feature
Description section, Device Functional Modes, Application and Implementation section, Power Supply
Recommendations section, Layout section, Device and Documentation Support section, and Mechanical,
Packaging, and Orderable Information section ..................................................................................................................... 1
•
CMS Request No.: C1411181 ................................................................................................................................................ 1
•
Changed Thermal Information table ....................................................................................................................................... 4
•
Changed Control Overview section ........................................................................................................................................ 9
•
Changed Soft-Start Circuit, Isolated Fly-Buck Converter graphics. ..................................................................................... 13
•
Deleted Lowest Part Count Isolated Application Schematic ............................................................................................... 19
Changes from Revision C (December 2013) to Revision D
•
Page
Added Thermal Parameters ................................................................................................................................................... 4
Changes from Revision B (March 2013) to Revision C
Page
•
Changed formatting to the TI standard................................................................................................................................... 1
•
Changed minimum operating input voltage from 9 V to 7.5 V in Features, Typical Application, Pin Descriptions,
Recommended Operating Ratings ........................................................................................................................................ 1
•
Added Absolute Maximum Junction Temperature.................................................................................................................. 4
Changes from Revision A (March 2013) to Revision B
•
Added SW to RTN (100 ns transient) in Absolute Maximum Ratings ................................................................................... 4
Changes from Original (March 2013) to Revision A
•
2
Page
Page
Changed layout of National Data Sheet to the TI standards................................................................................................ 20
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5 Pin Configuration and Functions
8-Pin HSOP
DDA Package
Top View
RTN
1
VIN
2
8
SW
7
BST
6
VCC
5
FB
HSOP
UVLO
3
RON
4
Exp Pad
8-Pin WSON
NGU Package
Top View
RTN
1
VIN
2
UVLO
3
RON
4
8 SW
WSON-8
Exp Pad
7 BST
6 VCC
5 FB
Pin Functions
PIN
I/O
DESCRIPTION
APPLICATION INFORMATION
NO.
NAME
1
RTN
–
Ground
Ground connection of the integrated circuit.
2
VIN
I
Input Voltage
Operating input range is 7.5 V to 100 V.
Resistor divider from VIN to UVLO to GND programs the
undervoltage detection threshold. An internal current source is
enabled when UVLO is above 1.225 V to provide hysteresis.
When UVLO pin is pulled below 0.66 V externally, the parts goes
in shutdown mode.
3
UVLO
I
Input Pin of Undervoltage
Comparator
4
RON
I
On-Time Control
A resistor between this pin and VIN sets the switch on-time as a
function of VIN. Minimum recommended on-time is 100 ns at max
input voltage.
5
FB
I
Feedback
This pin is connected to the inverting input of the internal
regulation comparator. The regulation level is 1.225 V.
6
VCC
O
Output from the Internal High
Voltage Series Pass Regulator.
Regulated at 7.6 V.
The internal VCC regulator provides bias supply for the gate
drivers and other internal circuitry. A 1.0-μF decoupling capacitor
is recommended.
7
BST
I
Bootstrap Capacitor
An external capacitor is required between the BST and SW pins
(0.01-μF ceramic). The BST pin capacitor is charged by the VCC
regulator through an internal diode when the SW pin is low.
8
SW
O
Switching Node
Power switching node. Connect to the output inductor and
bootstrap capacitor.
–
EP
–
Exposed Pad
Exposed pad must be connected to RTN pin. Connect to system
ground plane on application board for reduced thermal resistance.
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6 Specifications
6.1 Absolute Maximum Ratings (1) (2)
MIN
MAX
UNIT
VIN, UVLO to RTN
–0.3
100
V
SW to RTN
–1.5
VIN + 0.3
V
–5
VIN + 0.3
V
BST to VCC
100
V
BST to SW
13
V
SW to RTN (100 ns transient)
RON to RTN
–0.3
100
V
VCC to RTN
–0.3
13
V
FB to RTN
–0.3
5
V
200
°C
150
°C
Lead Temperature
(3)
Maximum Junction Temperature (4)
(1)
(2)
(3)
(4)
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
The RTN pin is the GND reference electrically connected to the substrate.
For detailed information on soldering plastic SO PowerPAD package, refer to the Packaging Data Book. Maximum solder time not to
exceed 4 seconds.
High junction temperatures degrade operating lifetimes. Operating lifetime is derated for junction temperatures greater than 125°C.
6.2 Handling Ratings
Tstg
Storage temperature range
MIN
MAX
–55
150
°C
2
kV
750
V
Human body model (HBM), per ANSI/ESDA/JEDEC JS-001, all pins (1)
V(ESD)
(1)
(2)
Electrostatic discharge
Charged device model (CDM), per JEDEC specification JESD22C101, all pins (2)
UNIT
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
6.3 Recommended Operating Ratings
MIN
MAX
VIN Voltage
7.5
100
V
Operating Junction Temperature (1)
–40
125
°C
(1)
UNIT
High junction temperatures degrade operating lifetimes. Operating lifetime is derated for junction temperatures greater than 125°C.
6.4 Thermal Information
LM34926
THERMAL METRICS (1)
NGU
DDA
UNIT
8 PINS
RθJA
Junction-to-ambient thermal resistance
RθJC(bot)
Junction-to-case (bottom) thermal resistance
3.2
2.4
ΨJB
Junction-to-board thermal characteristic parameter
19.2
24.4
RθJB
Junction-to-board thermal resistance
19.1
30.6
RθJC(top)
Junction-to-case (top) thermal resistance
34.7
37.3
ΨJT
Junction-to-top thermal characteristic parameter
0.3
6.7
(1)
4
41.3
41.1
°C/W
For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report (SPRA953).
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6.5 Electrical Characteristics
Typical values correspond to TJ = 25°C. Minimum and maximum limits apply over –40°C to 125°C junction temperature range
unless otherwise stated. VIN = 48V unless stated otherwise. See (1).
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
6.25
7.6
8.55
UNIT
VCC SUPPLY
VCC Reg
VCC Regulator Output
VIN = 48 V, ICC = 20 mA
VCC Current Limit
VIN = 48 V (2)
VCC UVLO Threshold (VCC increasing)
26
4.15
VCC UVLO Hysteresis
V
mA
4.5
4.9
300
V
mV
VCC Drop Out Voltage
VIN = 8 V, ICC = 20 mA
2.3
V
IIN Operating Current
Nonswitching, FB = 3 V
1.75
mA
IIN Shutdown Current
UVLO = 0 V
50
225
µA
UNDERVOLTAGE SENSING FUNCTION
UV Threshold
UV Rising
1.19
1.225
1.26
V
UV Hysteresis Input Current
UV = 2.5 V
–10
–20
-29
µA
Remote Shutdown Threshold
Voltage at UVLO Falling
0.32
0.66
V
110
mV
Remote Shutdown Hysteresis
REGULATION AND OVERVOLTAGE COMPARATORS
FB Regulation Level
Internal Reference Trip Point for Switch
ON
FB Overvoltage Threshold
Trip Point for Switch OFF
1.2
FB Bias Current
1.225
1.25
V
1.62
V
60
nA
SWITCH CHARACTERISTICS
Buck Switch RDS(ON)
ITEST = 200 mA, BST-SW = 7 V
Synchronous RDS(ON)
ITEST = 200 mA
Gate Drive UVLO
VBST − VSW Rising
2.4
Gate Drive UVLO Hysteresis
0.8
1.8
Ω
0.45
1
Ω
3
3.6
260
V
mV
CURRENT LIMIT
Current Limit Threshold
390
575
750
mA
Current Limit Response Time
Time to Switch Off
150
ns
OFF-Time Generator (Test 1)
FB = 0.1 V, VIN = 48 V
12
µs
OFF-Time Generator (Test 2)
FB = 1.0 V, VIN = 48 V
2.5
µs
165
°C
20
°C
THERMAL SHUTDOWN
Tsd
Thermal Shutdown Temperature
Thermal Shutdown Hysteresis
(1)
(2)
All limits are specified by design. All electrical characteristics having room temperature limits are tested during production at TA = 25°C.
All hot and cold limits are ensured by correlating the electrical characteristics to process and temperature variations and applying
statistical process control.
VCC provides self bias for the internal gate drive and control circuits. Device thermal limitations limit external loading.
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6.6 Switching Characteristics
Typical values correspond to TJ = 25°C. Minimum and maximum limits apply over –40°C to 125°C junction temperature range
unless otherwise stated. VIN = 48 V unless otherwise stated.
MIN
TYP
MAX
UNIT
ON-TIME GENERATOR
TON Test 1
VIN = 32 V, RON = 100 kΩ
270
350
460
ns
TON Test 2
VIN = 48 V, RON = 100 kΩ
188
250
336
ns
TON Test 3
VIN = 75 V, RON = 250 kΩ
250
370
500
ns
TON Test 4
VIN = 10 V, RON = 250 kΩ
1880
3200
4425
ns
MINIMUM OFF-TIME
Minimum Off-Timer
FB = 0 V
144
ns
6.7 Typical Characteristics
100
EFFICIENCY (%)
90
VIN=36V
VIN=24V
80
VIN=48V
70
60
VOUT2=10V, IOUT1=0
50
50
6
100
150
200
250
LOAD CURRENT (mA)
300
Figure 1. Efficiency at 750 kHz, VOUT1 = 10 V
Figure 2. VCC vs VIN
Figure 3. VCC vs ICC
Figure 4. ICC vs External VCC
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Typical Characteristics (continued)
Figure 5. TON vs VIN and RON
Figure 6. TOFF (ILIM) vs VFB and VIN
Figure 7. IIN vs VIN (Operating, Nonswitching)
Figure 8. IIN vs VIN (Shutdown)
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7 Detailed Description
7.1 Overview
The LM34926 step-down switching regulator features all the functions needed to implement a low-cost, efficient,
isolated bias supply. This high-voltage regulator contains 100-V, N-channel buck and synchronous switches, is
easy to implement, and is provided in thermally enhanced SO PowerPAD-8 and WSON-8 packages. The
regulator operation is based on a constant on-time control scheme using an on-time inversely proportional to VIN.
This control scheme does not require loop compensation. Current limit is implemented with forced off-time
inversely proportional to VOUT. This scheme ensures short circuit protection while providing minimum foldback.
The simplified block diagram of the LM34926 device is shown in Functional Block Diagram.
The LM34926 device can be applied in numerous applications to efficiently regulate down higher voltages. This
regulator is well suited for 48-V telecom and automotive power bus ranges. Protection features include: thermal
shutdown, undervoltage lockout, minimum forced off-time, and an intelligent current limit.
7.2 Functional Block Diagram
LM34926
START-UP
REGULATOR
VIN
VCC
V UVLO
20 µA
4.5V
UVLO
THERMAL
SHUTDOWN
UVLO
1.225V
SD
VDD REG
BST
0.66V
SHUTDOWN
BG REF
VIN
DISABLE
ON/OFF
TIMERS
RON
1.225V
SW
COT CONTROL
LOGIC
FEEDBACK
FB
OVER-VOLTAGE
1.62V
RTN
8
CURRENT
LIMIT
ONE-SHOT
ILIM
COMPARATOR
+
VILIM
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7.3 Feature Description
7.3.1 Control Overview
The LM34926 regulator employs a control principle based on a comparator and a one-shot on-timer, with the
output voltage feedback (FB) compared to an internal reference (1.225 V). If the FB voltage is below the
reference the internal buck switch is switched on for the one-shot timer period, which is a function of the input
voltage and the programming resistor (RT). Following the on-time the switch remains off until the FB voltage falls
below the reference, and the forced minimum off-time has expired. When the FB pin voltage falls below the
reference and the off-time one-shot period expires, the buck switch is then turned on for another on-time oneshot period. This continues until regulation is achieved and the FB voltage is approximately equal to 1.225 V
(typical).
In a synchronous buck converter, the low-side (sync) FET is on when the high-side (buck) FET is off. The
inductor current ramps up when the high-side switch is on and ramps down when the high-side switch is off.
There is no diode emulation feature in this IC, and therefore, the inductor current may ramp in the negative
direction at light load. This causes the converter to operate in continuous conduction mode (CCM) regardless of
the output loading. The operating frequency remains relatively constant with load and line variations. The
operating frequency can be determined from Equation 1.
VOUT1
f SW =
.x RON
where
•
K = 9 × 10–11
(1)
The output voltage (VOUT) is set by two external resistors (RFB1, RFB2). The regulated output voltage is
determined from Equation 2.
VOUT = 1.225V x
RFB2 + RFB1
RFB1
(2)
This regulator regulates the output voltage based on ripple voltage at the feedback input, requiring a minimum
amount of ESR for the output capacitor (COUT). A minimum of 25 mV of ripple voltage at the feedback pin (FB) is
required for the LM34926 device. In cases where the capacitor ESR is too small, additional series resistance
may be required (RC in Figure 9).
For applications where lower output voltage ripple is required the output can be taken directly from a low ESR
output capacitor, as shown in Figure 9. However, RC slightly degrades the load regulation.
L1
VOUT
SW
LM34926
RFB2
FB
RC
+
RFB1
COUT
VOUT
(low ripple)
Figure 9. Low Ripple Output Configuration
7.3.2 VCC Regulator
The LM34926 device contains an internal high-voltage linear regulator with a nominal output of 7.6 V. The input
pin (VIN) can be connected directly to the line voltages up to 100 V. The VCC regulator is internally current limited
to 30 mA. The regulator sources current into the external capacitor at VCC. This regulator supplies current to
internal circuit blocks including the synchronous MOSFET driver and the logic circuits. When the voltage on the
VCC pin reaches the UVLO threshold of 4.5 V, the IC is enabled.
The VCC regulator contains an internal diode connection to the BST pin to replenish the charge in the gate drive
boot capacitor when the SW pin is low.
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Feature Description (continued)
At high input voltages, the power dissipated in the high-voltage regulator is significant and can limit the overall
achievable output power. As an example, with the input at 48 V and switching at high frequency, the VCC
regulator may supply up to 7 mA of current resulting in 48 V × 7 mA = 336 mW of power dissipation. If the VCC
voltage is driven externally by an alternate voltage source, from 8.55 V to 14 V, the internal regulator is disabled.
This reduces the power dissipation in the IC.
7.3.3 Regulation Comparator
The feedback voltage at FB is compared to an internal 1.225 V reference. In normal operation, when the output
voltage is in regulation, an on-time period is initiated when the voltage at FB falls below 1.225 V. The high-side
switch stays on for the on-time, causing the FB voltage to rise above 1.225 V. After the on-time period, the highside switch stays off until the FB voltage again falls below 1.225 V. During start-up, the FB voltage is below
1.225 V at the end of each on-time, causing the high-side switch to turn on immediately after the minimum forced
off-time of 144 ns. The high-side switch can be turned off before the on-time is complete if peak current in the
inductor reaches the current limit threshold.
7.3.4 Overvoltage Comparator
The feedback voltage at FB is compared to an internal 1.62 V reference. If the voltage at FB rises above 1.62-V
the on-time pulse is immediately terminated. This condition can occur if the input voltage and/or the output load
changes suddenly. The high-side switch will not turn on again until the voltage at FB falls below 1.225 V.
7.3.5 On-Time Generator
The on-time for the LM34926 device is determined by the RON resistor, and is inversely proportional to the input
voltage (VIN), resulting in a nearly constant frequency as VIN is varied over its range. The on-time equation for the
LM34926 is determined be Equation 3.
TON =
10-10 x RON
VIN
(3)
See Figure 5. RON should be selected for a minimum on-time (at maximum VIN) greater than 100 ns, for proper
operation. This requirement limits the maximum frequency for each application.
7.3.6 Current Limit
The LM34926 device contains an intelligent current limit off-timer. If the current in the buck switch exceeds
575 mA, the present cycle is immediately terminated, and a non-resetable off-timer is initiated. The length of offtime is controlled by the FB voltage and the input voltage VIN. As an example, when FB = 0 V and VIN = 48 V, a
maximum off-time is set to 16 μs. This condition occurs when the output is shorted, and during the initial part of
start-up. This amount of time ensures safe short circuit operation up to the maximum input voltage of
100 V.
In cases of overload where the FB voltage is above zero volts (not a short circuit) the current limit off-time is
reduced. Reducing the off-time during less severe overloads reduces the amount of foldback, recovery time, and
start-up time. The off-time is calculated from Equation 4.
0.07 x VIN
Ps
TOFF(ILIM) =
VFB + 0.2V
(4)
The current limit protection feature is peak limited, the maximum average output will be less than the peak.
7.3.7 N-Channel Buck Switch and Driver
The LM34926 device integrates an N-channel buck switch and associated floating high voltage gate driver. The
gate driver circuit works in conjunction with an external bootstrap capacitor and an internal high-voltage diode. A
0.01-uF ceramic capacitor connected between the BST and SW pins provides the voltage to the driver during the
on-time. During each off-time, the SW pin is at approximately 0 V, and the bootstrap capacitor charges from VCC
through the internal diode. The minimum off-timer, set to 144 ns, ensures a minimum time each cycle to recharge
the bootstrap capacitor.
10
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Feature Description (continued)
7.3.8 Synchronous Rectifier
The LM34926 device provides an internal synchronous N-Channel MOSFET rectifier. This MOSFET provides a
path for the inductor current to flow when the high-side MOSFET is turned off.
The synchronous rectifier has no diode emulation mode, and is designed to keep the regulator in continuous
conduction mode even during light loads which would otherwise result in discontinuous operation. This feature
specifically allows the user to design a secondary regulator using a transformer winding off the main inductor to
generate the alternate regulated output voltage.
7.3.9 Undervoltage Detector
The LM34926 device contains a dual-level UVLO circuit. A summary of threshold voltages and operational states
is provided in Device Functional Modes. When the UVLO pin voltage is below 0.66 V, the controller is in a low
current shutdown mode. When the UVLO pin voltage is greater than 0.66 V but less than 1.225 V, the controller
is in standby mode. In standby mode the VCC bias regulator is active while the regulator output is disabled. When
the VCC pin exceeds the VCC undervoltage thresholds and the UVLO pin voltage is greater than 1.225 V, normal
operation begins. An external set-point voltage divider from VIN to GND can be used to set the minimum
operating voltage of the regulator.
UVLO hysteresis is accomplished with an internal 20-μA current source that is switched on or off into the
impedance of the set-point divider. When the UVLO threshold is exceeded, the current source is activated to
quickly raise the voltage at the UVLO pin. The hysteresis is equal to the value of this current times the resistance
RUV2.
If the UVLO pin is wired directly to the VIN pin, the regulator will begin operation once the VCC undervoltage is
satisfied.
VIN
2
VIN
CIN
+
RUV2
LM34926
3
UVLO
RUV1
Figure 10. UVLO Resistor Setting
7.3.10 Thermal Protection
The LM34926 device should be operated so the junction temperature does not exceed 150°C during normal
operation. An internal Thermal Shutdown circuit is provided to protect the LM34926 device in the event of a
higher than normal junction temperature. When activated, typically at 165°C, the controller is forced into a lowpower reset state, disabling the buck switch and the VCC regulator. This feature prevents catastrophic failures
from accidental device overheating. When the junction temperature falls below 145°C (typical hysteresis = 20°C),
the VCC regulator is enabled, and normal operation is resumed.
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Feature Description (continued)
7.3.11 Ripple Configuration
LM34926 uses constant on-time (COT) control scheme, in which the on-time is terminated by an on-timer, and
the off-time is terminated by the feedback voltage (VFB) falling below the reference voltage (VREF). Therefore, for
stable operation, the feedback voltage must decrease monotonically, in phase with the inductor current during
the off-time. Furthermore this change in feedback voltage (ΔVFB) during off-time must be large enough to
suppress any noise component present at the feedback node.
Table 1 shows three different methods for generating appropriate voltage ripple at the feedback node. Type 1
and Type 2 ripple circuits couple the ripple at the output of the converter to the feedback node (FB). The output
voltage ripple has two components:
1. Capacitive ripple caused by the inductor current ripple charging and discharging the output capacitor.
2. Resistive ripple caused by the inductor current ripple flowing through the ESR of the output capacitor.
The capacitive ripple is not in phase with the inductor current. As a result, the capacitive ripple does not
decrease monotonically during the off-time. The resistive ripple is in phase with the inductor current and
decreases monotonically during off-time. The resistive ripple must exceed the capacitive ripple at the output node
(VOUT) for stable operation. If this condition is not satisfied, unstable switching behavior is observed in COT
converters, with multiple on-time bursts in close succession followed by a long off-time.
Type 3 ripple method uses Rr and Cr and the switch node (SW) voltage to generate a triangular ramp. This
triangular ramp is AC-coupled using Cac to the feedback node (FB). Because this circuit does not use the output
voltage ripple, it is ideally suited for applications where low output voltage ripple is required. See AN-1481
Controlling Output Ripple and Achieving ESR Independence in Constant On-Time (COT) Regulator Designs
(SNVA166) for more details for each ripple generation method.
Table 1. Ripple Configuration
TYPE 1
LOWEST COST CONFIGURATION
TYPE 2
REDUCED RIPPLE CONFIGURATION
TYPE 3
MINIMUM RIPPLE CONFIGURATION
VOUT
VOUT
L1
VOUT
L1
L1
Cac
R FB2
R FB2
RC
To FB
R FB1
R FB1
GND
25 mV VOUT
x
ûIL(MIN) VREF
Cr
R FB2
C OUT
Cac
To FB
C OUT
RC >
Rr
RC
C OUT
GND
To FB
R FB1
GND
C>
5
gsw(RFB2||RFB1)
25 mV
RC >
ûIL(MIN)
Cr = 3300 pF
Cac = 100 nF
(VIN(MIN) - VOUT) x TON
R rC r <
25 mV
7.3.12 Soft Start
A soft-start feature can be implemented with the LM34926 device using an external circuit. As shown in
Figure 11, the soft-start circuit consists of one capacitor C1, two resistors R1 and R2, and a diode D. During the
initial start-up, the VCC voltage is established before the VOUT voltage. Capacitor C1 is discharged and diode D is
thereby forward biased to pull up the FB voltage. The FB voltage exceeds the reference voltage (1.225 V) and
switching is therefore disabled. As capacitor C1 charges, the voltage at node B gradually decreases and
switching commences. VOUT will gradually rise to maintain the FB voltage at the reference voltage. Once the
voltage at node B is less than a diode drop above the FB voltage, the soft-start sequence is finished and D is
reverse-biased.
12
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During the initial part of the start-up, the FB voltage can be approximated as shown in Equation 5. The effect of
R1 has been ignored to simplify the calculation.
VFB = (VCC - VD) x
RFB1 x RFB2
R2 x (RFB1 + RFB2) + RFB1 x RFB2
(5)
C1 is charged after the first start up. Diode D1 is optional and can be added to discharge C1 when the input
voltage experiences a momentary drop to initialize the soft-start sequence.
To achieve the desired soft start, the following design guidance is recommended:
(1) R2 is selected so that VFB is higher than 1.225 V for a VCC of 4.5 V, but is lower than 5 V when VCC is 8.55 V.
If an external VCC is used, VFB should not exceed 5 V at maximum VCC.
(2) C1 is selected to achieve the desired start-up time which can be determined from .
RFB1 x RFB2
)
tS = C1 x (R2 +
RFB1 + RFB2
(3) R1 is used to maintain the node B voltage at zero after the soft start is finished. A value larger than the
feedback resistor divider is preferred. The effect of resistor R1 is ignored.
Using the component values shown in Figure 12, selecting C1 = 1 uF, R2 = 1 kΩ, R1 = 30 kΩ results in a softstart time of about 2 ms.
VOUT
VCC
C1
RFB2
R2
To FB
D
D1
B
RFB1
R1
Figure 11. Soft-Start Circuit
7.4 Device Functional Modes
The UVLO pin controls the operating mode of the LM34926 device (see Table 2 for the detailed functional
states).
Table 2. UVLO Mode
UVLO
VCC
MODE
DESCRIPTION
< 0.66 V
Disabled
Shutdown
VCC regulator disabled.
Switching disabled.
0.66 V — 1.225 V
Enabled
Standby
VCC regulator enabled.
Switching disabled.
VCC < 4.5 V
Standby
VCC regulator enabled.
Switching disabled.
VCC > 4.5 V
Operating
> 1.225 V
VCC enabled.
Switching enabled.
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The LM34926 device is step-down DC-DC converter. The device is typically used to convert a higher DC voltage
to a lower DC voltage with a maximum available output current of 300 mA. Use the following design procedure to
select component values for the LM34926 device. Alternately, use the WEBENCH® software to generate a
complete design. The WEBENCH software uses an iterative design procedure and accesses a comprehensive
database of components when generating a design. This section presents a simplified discussion of the design
process.
8.2 Typical Application
Application Circuit: 20-V to 95-V Input and 10-V, 250-mA Output Isolated Fly-Buck™ Converter
VOUT2
D1
+
N2
COUT2
1 µF
X1
LM34926
BST
VIN
20V-95V
0.01 µF
+
CBST
+
47 µH
VOUT1
SW
46.4 kΩ 1 nF
Rr
Cr
VIN
CIN
1 µF
N1
CBYP
0.1 µF
+
RON
RUV2
127 kΩ
RUV1
8.25 kΩ
RON
130 kΩ
0.1 µF
RTN
COUT1
1 µF
RFB2
VCC
UVLO
+
Cac
FB
+
D2
7.32 kΩ
CVCC
1 µF
RFB1
1 kΩ
Figure 12. Isolated Fly-Buck Converter Using LM34926
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Typical Application (continued)
8.2.1 Design Requirements
Selection of external components is illustrated through a design example. Table 3 lists the design example
specifications.
Table 3. Buck Converter Design Specifications
DESIGN PARAMETERS
VALUE
Input Voltage Range
20 V to 95 V
Primary Output Voltage
10 V
Secondary (Isolated) Output Voltage
9.5 V
Maximum Output Current (Primary + Secondary)
250 mA
Maximum Power Output
2.5 W
Nominal Switching Frequency
750 kHz
8.2.2 Detailed Design Procedure
8.2.2.1 Transformer Turns Ratio
The transformer turns ratio is selected based on the ratio of the primary output voltage to the secondary
(isolated) output voltage. In this design example, the two outputs are nearly equal and a 1:1 turns ratio
transformer is selected. Therefore, N2 / N1 = 1.
If the secondary (isolated) output voltage is significantly higher or lower than the primary output voltage, a turns
ratio less than or greater than 1 is recommended. The primary output voltage is normally selected based on the
input voltage range such that the duty cycle of the converter does not exceed 50% at the minimum input voltage.
This condition is satisfied if VOUT1 < VIN_MIN / 2
8.2.2.2 Total IOUT
The total primary referred load current is calculated by multiplying the isolated output loads by the turns ratio of
the transformer as shown in Equation 6.
N2
I OUT(MAX) = I OUT1 + I OUT2 ´
= 0.250 A
(6)
N1
8.2.2.3 RFB1, RFB2
The feedback resistors are selected to set the primary output voltage. The selected value for RFB1 is 1 kΩ. RFB2
can be calculated using the following equations to set VOUT1 to the specified value of 10 V. A standard resistor
value of 7.32 kΩ is selected for RFB2.
VOUT1 = 1.225V x (1 +
RFB2
)
RFB1
(7)
VOUT1
- 1) x RFB1 = 7.16 k:
:RFB2 = (
1.225
(8)
8.2.2.4 Frequency Selection
Equation 1 is used to calculate the value of RON required to achieve the desired switching frequency.
VOUT1
f SW =
.x RON
where
•
K = 9 × 10–11
(9)
For VOUT1 of 10 V and fSW of 750 kHz, the calculated value of RON is 148 kΩ. A lower value of 130 kΩ is selected
for this design to allow for second-order effects at high switching frequency that are not included in Equation 9.
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8.2.2.5 Transformer Selection
A coupled inductor or a flyback-type transformer is required for this topology. Energy is transferred from primary
to secondary when the low-side synchronous switch of the buck converter is conducting.
The maximum inductor primary ripple current that can be tolerated without exceeding the buck switch peak
current limit threshold (0.39-A minimum) is given by Equation 10.
N2 ö
æ
DI L1 = ç 0.39 A - I OUT1 - I OUT2 ´
´ 2 = 0.28 A
N1 ÷ø
è
(10)
Using the maximum peak-to-peak inductor ripple current ΔIL1 from Equation 10, the minimum inductor value is
given by Equation 11.
VIN(MAX) - VOUT
VOUT
´
= 42.6 mH
L1 =
DI L1 ´ fSW
VIN(MAX)
(11)
A higher value of 47 µH is selected to insure the high-side switch current does not exceed the minimum peak
current limit threshold.
8.2.2.6 Primary Output Capacitor
In a conventional buck converter the output ripple voltage is calculated as shown in Equation 12.
f
'VOUT =
'IL1
x f x COUT1
(12)
To limit the primary output ripple voltage ΔVOUT1 to approximately 50 mV, an output capcitor COUT1 of 0.93 µF is
required.
Figure 13 shows the primary winding current waveform (IL1) of a fly-buck converter. The reflected secondary
winding current adds to the primary winding current during the buck switch off-time. Because of this increased
current, the output voltage ripple is not the same as in conventional buck converter. The output capacitor value
calculated in Equation 12 should be used as the starting point. Optimization of output capacitance over the entire
line and load range must be done experimentally. If the majority of the load current is drawn from the secondary
isolated output, a better approximation of the primary output voltage ripple is given by Equation 13.
N2 ö
æ
ç I OUT2 ´ N1 ÷ ´ TON(MAX)
è
ø
DVOUT1 =
» 0.16 V
COUT1
(13)
TON(MAX) x IOUT2 x N2/N1
IL1
IOUT2
IL2
TON(MAX) x IOUT2
Figure 13. Current Waveforms for COUT1 Ripple Calculation
A standard 1-µF, 25-V capacitor is selected for this design. If lower output voltage ripple is required, a higher
value should be selected for COUT1 and/or COUT2.
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8.2.2.7 Secondary Output Capacitor
A simplified waveform for secondary output current (IOUT2) is shown in Figure 14.
IOUT2
IL2
TON(MAX) x IOUT2
Figure 14. Secondary Current Waveforms for COUT2 Ripple Calculation
The secondary output current (IOUT2) is sourced by COUT2 during on-time of the buck switch, TON. Ignoring the
current transition times in the secondary winding, the secondary output capacitor ripple voltage can be calculated
using Equation 14.
IOUT2 x TON (MAX)
'VOUT2 =
COUT2
(14)
For a 1:1 transformer turns ratio, the primary and secondary voltage ripple equations are identical. Therefore,
COUT2 is chosen to be equal to COUT1 (1 µF) to achieve comparable ripple voltages on primary and secondary
outputs.
If lower output voltage ripple is required, a higher value should be selected for COUT1 and/or COUT2.
8.2.2.8 Type III Feedback Ripple Circuit
Type III ripple circuit as described in Ripple Configuration is required for the Fly-Buck topology. Type I and Type
II ripple circuits use series resistance and the triangular inductor ripple current to generate ripple at VOUT and the
FB pin. The primary ripple current of a Fly-Buck is the combination or primary and reflected secondary currents
as shown in Figure 13. In the fly-buck topology, Type I and Type II ripple circuits suffer from large jitter as the
reflected load current affects the feedback ripple.
VOUT
L1
Rr
Cac
C OUT
Cr
R FB2
GND
To FB
R FB1
Figure 15. Type III Ripple Circuit
Selecting the Type III ripple components using the equations from Ripple Configuration ensures that the FB pin
ripple is be greater than the capacitive ripple from the primary output capacitor COUT1. The feedback ripple
component values are chosen as shown in Equation 15.
Cr = 1000 pF
Cac = 0.1 PF
RrCr d
(VIN (MIN) - VOUT) x TON
50 mV
(15)
The calculated value for Rr is 66 kΩ. This value provides the minimum ripple for stable operation. A smaller
resistance should be selected to allow for variations in TON, COUT1 and other components. For this design, Rr
value of 46.4 kΩ is selected.
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8.2.2.9 Secondary Diode
The reverse voltage across secondary-rectifier diode D1 when the high-side buck switch is off can be calculated
using Equation 16.
VD1 =
N2
VIN
N1
(16)
For a VIN_MAX of 95 V and the 1:1 turns ratio of this design, a 100-V Schottky is selected.
8.2.2.10 VCC and Bootstrap Capacitor
A 1-µF capacitor of 16-V or higher rating is recommended for the VCC regulator bypass capacitor.
A good value for the BST pin bootstrap capacitor is 0.01-µF with a 16-V or higher rating.
8.2.2.11 Input Capacitor
The input capacitor is typically a combination of a smaller bypass capacitor located near the regulator IC and a
larger bulk capacitor. The total input capacitance should be large enough to limit the input voltage ripple to a
desired amplitude. For input ripple voltage ΔVIN, CIN can be calculated using Equation 17.
I OUT(MAX)
CIN ³
4 ´ f ´ DVIN
(17)
Choosing a ΔVIN of 0.5 V gives a minimum CIN of 0.167 μF. A standard value of 0.1 μF is selected for for CBYP in
this design. A bulk capacitor of higher value reduces voltage spikes due to parasitic inductance between the
power source to the converter. A standard value of 1 μF is selected for for CIN in this design. The voltage ratings
of the two input capacitors should be greater than the maximum input voltage under all conditions.
8.2.2.12 UVLO Resistors
UVLO resistors RUV1 and RUV2 set the undervoltage lockout threshold and hysteresis according to Equation 18
and Equation 19.
VIN (HYS) = IHYS x RUV2
(18)
VIN (UVLO, rising) = 1.225V x
R
( RUV2
+ 1)
UV1
where
•
IHYS = 20 μA, typical
(19)
For a UVLO hysteresis of 2.5 V and UVLO rising threshold of 20 V, Equation 18 and Equation 19 require RUV1 of
8.25 kΩ and RUV2 of 127 kΩ and these values are selected for this design example.
8.2.2.13 VCC Diode
Diode D2 is an optional diode connected between VOUT1 and the VCC regulator output pin. When VOUT1 is more
than one diode drop greater than the VCC voltage, the VCC bias current is supplied from VOUT1. This results in
reduced power losses in the internal VCC regulator which improves converter efficiency. VOUT1 must be set to a
voltage at least one diode drop higher than 8.55 V (the maximum VCC voltage) if D2 is used to supply bias
current.
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8.2.3 Application Curves
VIN = 48 V
IOUT1 = 0 mA
VIN = 48 V
IOUT1 = 0
Step Load on IOUT2 = 80 to 180 mA
IOUT2 = 100 mA
Figure 16. Steady State Waveform
Figure 17. Step Load Response
100
EFFICIENCY (%)
90
VIN=36V
VIN=24V
80
VIN=48V
70
60
VOUT2=10V, IOUT1=0
50
50
100
150
200
250
LOAD CURRENT (mA)
300
VOUT1 = 10 V
Figure 18. Efficiency at 750 kHz
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9 Power Supply Recommendations
LM34926 is a power-management device. The power supply for the device is any DC voltage source within the
specified input range.
10 Layout
10.1 Layout Guidelines
A proper layout is essential for optimum performance of the circuit. To ensure proper layout, observe the
following guidelines:
1. CIN: The loop consisting of input capacitor (CIN), VIN pin, and RTN pin carries switching currents. Therefore,
place the input capacitor close to the IC, directly across VIN and RTN pins, and the connections to these two
pins should be direct to minimize the loop area. In general it is not possible to accommodate all of input
capacitance near the IC. A good practice is to use a 0.1-μF or 0.47-μF capacitor directly across the VIN and
RTN pins close to the IC, and the remaining bulk capacitor as close as possible (see Figure 19).
2. CVCC and CBST: The VCC and bootstrap (BST) bypass capacitors supply switching currents to the high and
low-side gate drivers. Place these two capacitors as close to the IC as possible, and the connecting trace
lengths and loop area should be minimized (see Figure 19).
3. The Feedback trace carries the output voltage information and a small ripple component that is necessary for
proper operation of LM34926. Therefore take care while routing the feedback trace to avoid coupling any
noise to this pin. In particular, feedback trace should not run close to magnetic components, or parallel to any
other switching trace.
4. SW trace: SW node switches rapidly between VIN and GND every cycle and is therefore a possible source of
noise. SW node area should be minimized. In particular SW node should not be inadvertently connected to a
copper plane or pour.
10.2 Layout Example
8
SW
7
BST
3
6
VCC
4
5
FB
RTN
1
VIN
2
UVLO
RON
CIN
HSOP
CVCC
Figure 19. Placement of Bypass Capacitors
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11 Device and Documentation Support
11.1 Trademarks
PowerPAD, Fly-Buck are trademarks of Texas Instruments.
WEBENCH is a registered trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
11.2 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.3 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
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20-Oct-2014
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
Lead/Ball Finish
MSL Peak Temp
(2)
(6)
(3)
Op Temp (°C)
Device Marking
(4/5)
LM34926MR/NOPB
ACTIVE SO PowerPAD
DDA
8
95
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
S000XB
LM34926MRX/NOPB
ACTIVE SO PowerPAD
DDA
8
2500
Green (RoHS
& no Sb/Br)
CU SN
Level-3-260C-168 HR
S000XB
LM34926SD/NOPB
ACTIVE
WSON
NGU
8
1000
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
L34926
LM34926SDX/NOPB
ACTIVE
WSON
NGU
8
4500
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
L34926
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3)
MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4)
There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5)
Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6)
Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish
value exceeds the maximum column width.
Addendum-Page 1
Samples
PACKAGE OPTION ADDENDUM
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20-Oct-2014
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com
20-Oct-2014
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
Diameter Width (mm)
(mm) W1 (mm)
B0
(mm)
K0
(mm)
P1
(mm)
W
Pin1
(mm) Quadrant
LM34926MRX/NOPB
SO
Power
PAD
DDA
8
2500
330.0
12.4
6.5
5.4
2.0
8.0
12.0
Q1
LM34926SD/NOPB
WSON
NGU
8
1000
178.0
12.4
4.3
4.3
1.3
8.0
12.0
Q1
LM34926SDX/NOPB
WSON
NGU
8
4500
330.0
12.4
4.3
4.3
1.3
8.0
12.0
Q1
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
20-Oct-2014
*All dimensions are nominal
Device
Package Type
Package Drawing
Pins
SPQ
Length (mm)
Width (mm)
Height (mm)
LM34926MRX/NOPB
LM34926SD/NOPB
SO PowerPAD
DDA
8
2500
367.0
367.0
35.0
WSON
NGU
8
1000
210.0
185.0
35.0
LM34926SDX/NOPB
WSON
NGU
8
4500
367.0
367.0
35.0
Pack Materials-Page 2
MECHANICAL DATA
DDA0008B
MRA08B (Rev B)
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MECHANICAL DATA
NGU0008B
SDC08B (Rev A)
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