Cherry CS51313GDR16 Synchronous cpu buck controller capable of implementing multiple linear regulator Datasheet

CS51313
CS51313
Synchronous CPU Buck Controller
Capable of Implementing Multiple Linear Regulators
Features
Description
The CS51313 is a synchronous dual
NFET Buck Regulator Controller. It is
designed to power the core logic of the
latest high performance CPUs. It uses the
V2TM control method to achieve the
fastest possible transient response and
best overall regulation. It incorporates
many additional features required to
ensure the proper operation and protection of the CPU and Power system. The
CS51313 provides the industry’s most
highly integrated solution, minimizing
external component count, total solution
size, and cost.
The CS51313 is specifically designed to
power Intel’s Pentium® II processor and
includes the following features: 5-bit
DAC with 1.2% tolerance, Power-Good
output, overcurrent hiccup mode protection, over voltage protection, VCC monitor, Soft Start, adaptive voltage positioning and adaptive FET non-overlap time.
A precision reference trimmed to 1% is
also externally available for use by other
regulators. The CS51313 will operate
over an 8.4V to 14V range and is available in 16 lead narrow body surface
mount package.
■ Synchronous Switching
Regulator Controller for CPU
VCORE
■ Dual N-Channel MOSFET
Synchronous Buck Design
■ V2
TM
Control Topology
■ 200ns Transient Loop Response
■ 5-bit DAC with 1.2% Tolerance
■ Hiccup Mode Overcurrent
Protection
■ 40ns Gate Rise and Fall Times
(3.3nF load)
■ 65ns Adaptive FET Non-overlap
Time
■ Adaptive Voltage Positioning
■ Power-Good Output Monitors
Regulator Output
Application Diagram
+12V
■ VCC Monitor Provides Under
Voltage Lockout
+5V
+3.3V +3.3V
1200µF/10V
1200µF/10V
1µF
FS70VSJ-03
VCC
VID0
GATEH
GATEL
VID1
3.3mΩ
VCORE
2.0V@19A
PWRGD
VID4
■ +1.23V Reference Voltage
Available Externally
1200µF/10V
x5
VOUT
VID3
OVP
1.2µH
■ Enable Through use of the
COMP pin
FS70VSJ-03
VFB
VID2
■ OVP Output Monitors Regulator
Output
1200µF/10V
x3
510Ω
COMP
VREF
0.1
µF
COFF
GND
+12
0.01
µF
1µF
3
2
+
-
Package Options
510Ω
10K
680pF
18K
1%
0.1µF
PWRGD
VID0
IRL3103S
1
LM358A
22.1K
51K
1%
100K
VGTL+
1.5V@3A
1%
1%
1200µF/ 10V
x2
LM358A
5
6
+
-
7
TIP 31
102K
100K
1%
1%
16 Lead SO Narrow
100Ω
VCLOCK
2.5V@1A
47µF
COMP
1
VID1
COFF
VID2
PWRGD
VID3
OVP
VREF
GATE(L)
VID4
Gnd
VFB
GATE(H)
VOUT
VCC
V2 is a trademark of Switch Power, Inc.
Pentium is a registered trademark of Intel Corporation.
Cherry Semiconductor Corporation
2000 South County Trail, East Greenwich, RI 02818
Tel: (401)885-3600 Fax: (401)885-5786
Email: [email protected]
Web Site: www.cherry-semi.com
Rev. 3/11/99
1
A
®
Company
CS51313
Absolute Maximum Ratings
Operating Junction Temperature, TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 150°C
Lead Temperature Soldering
Reflow (SMD styles only) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .60 sec. max above 183°C, 230°C peak
Storage Temperature Range, TS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −65° to 150°C
ESD Susceptibility (Note 1) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2kV
Pin Symbol
Pin Name
VMAX
VMIN
ISOURCE
ISINK
1mA
1.5A Peak
200mA DC
5mA
1mA
VREF
VCC
Bandgap Reference Voltage
IC Power Input
6V
16V
-0.3V
-0.3V
1mA
N/A
COMP
VFB, VOUT, VID0-4
6V
6V
-0.3V
-0.3V
1mA
1mA
COFF
GATE(H), GATE(L)
Compensation Pin
Voltage Feedback Input, Output
Voltage Sense Pin, Voltage
ID DAC Inputs
Off-Time Pin
High-Side, Low Side FET Drivers
6V
16V
-0.3V
-0.3V
PWRGD
OVP
Gnd
Power-Good Output
Overvoltage Protection
Ground
6V
15V
0V
-0.3V
-0.3V
0V
1mA
50mA
1.5APeak
1.5A Peak
200mA DC 200mA DC
1mA
30mA
30mA
1mA
1.5A Peak
N/A
200mA DC
Package Pin Description
PACKAGE PIN #
PIN SYMBOL
FUNCTION
1,2,3,4,6
VIDO – VID4
9
VCC
10
11
12
14
GATE(H)
Gnd
GATE(L)
PWRGD
16
COMP
15
COFF
8
7
VOUT
VFB
5
VREF
13
OVP
Voltage ID DAC inputs. These pins are internally pulled up to
5.65V if left open. VID4 selects the DAC range. When VID4 is
high (logic one), the Error Amp reference range is 2.125V to
3.525V with 100mV increments. When VID4 is low (logic zero),
the Error amp reference voltage is 1.325V to 2.075V with 50mV
increments.
Input power supply pin for the internal circuitry.
Decouple with filter capacitor to Gnd.
High side switch FET driver pin
Ground pin.
Low side synchronous FET driver pin.
Power-Good Output. Open collector output drives low when
VFB is out of regulation.
Error amp output. PWM comparator inverting input.
A capacitor to Gnd provides error amp compensation.
Off-Time Capacitor Pin. A capacitor from this pin to Gnd sets
the off time for the regulator
Current limit comparator inverting input.
Error amp inverting input, PWM comparator non-inverting
input, current limit comparator non-inverting input, PWRGD
and OVP comparator input.
Bandgap Reference Voltage. It can be used to generate other
regulated output voltages.
Overvoltage protection pin. Goes high when overvoltage
condition is detected on VFB.
2
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
-7.0
15
30
0.1
30
60
80
50
70
3.2
0.5
7.0
60
120
µA
µA
µA
dB
kHz
dB
mmho
MΩ
Error Amplifier
■
VFB Bias Current
COMP Source Current
COMP Sink Current
Open Loop Gain
Unity Gain Bandwidth
PSRR @ 1kHz
Transconductance
Output Impedance
0.2V ≤VFB ≤ 3.5V
VCOMP = 1.2V to 3.6V; VFB = 1.9 V
VCOMP = 1.2V; VFB = 2.1V
CCOMP = 0.1µF
CCOMP = 0.1µF
CCOMP = 0.1µF
■ Voltage Identification DAC
Measure VFB = VCOMP, VCC = 12V (Note 2)
75°C ≤ TJ ≤ 125°C
VID4 VID3 VID2 VID1 VID0
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
1
25°C ≤ TJ ≤ 75°C
MIN
TYP
MAX
± TOL
MIN
TYP
MAX
± TOL
UNIT
3.483
3.384
3.285
3.186
3.087
2.989
2.890
2.791
2.692
2.594
2.495
2.396
2.297
2.198
2.099
2.050
2.001
1.953
1.904
1.854
1.805
1.755
1.706
1.656
1.607
1.558
1.508
1.459
1.409
1.360
1.310
1.225
3.525
3.425
3.325
3.225
3.125
3.025
2.925
2.825
2.725
2.625
2.525
2.425
2.325
2.225
2.125
2.075
2.025
1.975
1.925
1.875
1.825
1.775
1.725
1.675
1.625
1.575
1.525
1.475
1.425
1.375
1.325
1.250
3.567
3.466
3.365
3.264
3.163
3.061
2.960
2.859
2.758
2.657
2.555
2.454
2.353
2.252
2.151
2.100
2.049
1.997
1.946
1.896
1.845
1.795
1.744
1.694
1.643
1.593
1.542
1.491
1.441
1.390
1.340
1.275
1.2%
1.2%
1.2%
1.2%
1.2%
1.2%
1.2%
1.2%
1.2%
1.2%
1.2%
1.2%
1.2%
1.2%
1.2%
1.2%
1.2%
1.1%
1.1%
1.1%
1.1%
1.1%
1.1%
1.1%
1.1%
1.1%
1.1%
1.1%
1.1%
1.1%
1.1%
2.0%
3.455
3.357
3.259
3.161
3.063
2.965
2.875
2.777
2.679
2.580
2.482
2.389
2.290
2.192
2.093
2.044
1.995
1.945
1.896
1.847
1.798
1.748
1.699
1.650
1.601
1.551
1.502
1.453
1.404
1.354
1.305
1.225
3.525
3.425
3.325
3.225
3.125
3.025
2.925
2.825
2.725
2.625
2.525
2.425
2.325
2.225
2.125
2.075
2.025
1.975
1.925
1.875
1.825
1.775
1.725
1.675
1.625
1.575
1.525
1.475
1.425
1.375
1.325
1.250
3.596
3.494
3.392
3.290
3.188
3.086
2.975
2.873
2.771
2.670
2.568
2.461
2.360
2.258
2.157
2.106
2.055
2.005
1.954
1.903
1.852
1.802
1.751
1.700
1.649
1.599
1.548
1.497
1.446
1.396
1.345
1.275
2.0%
2.0%
2.0%
2.0%
2.0%
2.0%
1.7%
1.7%
1.7%
1.7%
1.7%
1.5%
1.5%
1.5%
1.5%
1.5%
1.5%
1.5%
1.5%
1.5%
1.5%
1.5%
1.5%
1.5%
1.5%
1.5%
1.5%
1.5%
1.5%
1.5%
1.5%
2.0%
V
V
V
V
V
V
V
V
V
V
V
V
V
V
V
V
V
V
V
V
V
V
V
V
V
V
V
V
V
V
V
V
3
CS51313
Electrical Characteristics: 0˚C < TA < 70˚C; 0˚C < TJ < 125˚C; 9V < VCC < 14V;
2.0V DAC Code (VID4 = VID3 =VID2 = VID1 = 0, VID0 = 1), CGATE(H) = CGATE(L) = 3.3nF, COFF = 390pF; Unless otherwise stated.
CS51313
Electrical Characteristics: 0˚C < TA < 70˚C; 0˚C < TJ < 125˚C; 9V < VCC < 14V;
2.0V DAC Code (VID4 = VID3 =VID2 = VID1 = 0, VID0 = 1), CGATE(H) = CGATE(L) = 3.3nF, COFF = 390pF; Unless otherwise stated.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
0.01
UNIT
Line Regulation
9V ≤ VCC ≤ 14V
Input Threshold
VID4, VID3, VID2, VID1, VID0
1.00
1.25
2.40
V
Input Pull-up Resistance
VID4, VID3, VID2, VID1, VID0
25
50
100
kΩ
5.48
5.65
5.82
V
1.211
1.230
1.248
V
Pull-up Voltage
%/V
■ Bandgap Reference Voltage
VREF
IVREF = 10µA sourcing,VCC = 12V
■ GATE(H) and GATE(L)
High Voltage at 100mA
Measure VCC –GATE(L)/(H)
1.2
2.1
V
Low Voltage at 100mA
Measure GATE(L)/H
1.0
1.5
V
Rise Time
1.6V < GATE(H)/(L) < (VCC – 2.5V)
40
80
ns
Fall Time
(VCC – 2.5V) > GATE(L)/(H) > 1.6V
40
80
ns
GATE(H) to GATE(L) Delay
GATE(H)<2V, GATE(L)>2V
VCC = 12V
30
65
110
ns
GATE(L) to GATE(H) Delay
GATE(L)<2V, GATE(H)>2V
VCC = 12V
30
65
110
ns
GATE pull-down
Resistance to Gnd (Note 3)
20
50
115
kΩ
0V ≤ VOUT ≤ 3.5V
77
86
101
mV
0.2
0.25
0.3
V
VOUT Bias Current
0.2V ≤ VOUT ≤ 3.5V
-7.0
0.1
7.0
µA
OVC Latch Discharge Current
VCOMP = 1V
100
800
2500
µA
PWM Comparator Offset Voltage
0V ≤ VFB ≤ 3.5V
0.99
1.10
1.23
V
Transient Response
VFB = 0 to 3.5V
200
300
ns
1.6
2.3
■ Overcurrent Protection
OVC Comparator Offset Voltage
Discharge Threshold Voltage
■ PWM Comparator
■ COFF
Off-Time
1.0
µs
Charge Current
VCOFF = 1.5V
550
µA
Discharge Current
VCOFF = 1.5V
25
mA
■ Power-Good Output
PWRGD Sink Current
VFB = 1.7V, VPWRGD = 1V
0.5
4
15
mA
PWRGD Upper Threshold
% of nominal DAC code
5
8.5
12
%
PWRGD Lower Threshold
% of nominal DAC code
-12
-8.5
-5
%
PWRGD Output Low Voltage
VFB = 1.7V, IPWRGD = 500uA
0.2
0.3
V
10
25
mA
8.5
12
%
1.1
1.5
V
■ Overvoltage Protection (OVP) Output
OVP Source Current
OVP = 1V
1
OVP Threshold
% of nominal DAC code
5
OVP Pull-up Voltage
IOVP = 1mA, VCC - VOVP
4
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
■ General Electrical Specifications
VCC Monitor Start Threshold
7.9
8.4
8.9
V
VCC Monitor Stop Threshold
7.6
8.1
8.6
V
0.15
0.30
0.60
V
12
20
mA
Hysteresis
Start - Stop
VCC Supply Current
No Load on GATE(H), GATE(L)
Note 1: All pins are rated 2kV except for the VREF pin (Pin 5) which is typically rated at 800V.
Note 2: The IC power dissipation in a typical application with VCC = 12V, switching frequency fSW = 250kHz, 50nc
MOSFETs and RθJA = 115°C/W yields an operating junction temperature rise of approximately 52°C, and a junction temperature of 77°C with an ambient temperature of 25°C.
Note 3: Guaranteed by design, not 100% tested in production.
Block Diagram
VFB
COFF
COMP
1.10V
-
PWM COMP
+
+
-
EA
+
-
DISCHARGE
COMP
+
-
VREF
CURRENT LIMIT
+
86mV
+
VOUT
OFF
TIME
+
- 0.25V
BANDGAP
REFERENCE
R
Q
FAULT
LATCH
S
VID0
VID1
UVLO
DAC
VID2
VCC
VID3
GATE(H)
VID4
NONOVERLAP
LOGIC
+
-
GATE(L)
+
VCC
OVP
PWRGD
5
Gnd
CS51313
Electrical Characteristics: 0˚C < TA < 70˚C; 0˚C < TJ < 125˚C; 9V < VCC < 14V;
2.0V DAC Code (VID4 = VID3 =VID2 = VID1 = 0, VID0 = 1), CGATE(H) = CGATE(L) = 3.3nF, COFF = 390pF; Unless otherwise stated.
Figure 1: Gate(H) and Gate(L) Falltime vs. Load Capacitance.
Figure 4: Percent Output Error vs. DAC Output
Voltage Setting, VID4 = 0.
VCC = 12V
TA = 25°C
125
100
0.10
Output Error (%)
Falltime (ns)
150
75
50
25
0
0
2000
4000
6000
8000
10000
12000
14000
16000
0.05
0
−0.05
VCC = 12V
TA = 25°C
VID4 = 0
−0.10
−0.15
−0.20
1.325 1.375 1.425 1.475 1.525 1.575 1.625 1.675 1.725 1.775 1.825 1.875 1.925 1.975 2.025 2.075
Load Capacitance (pF)
DAC Output Voltage Setting (V)
Figure 2: Gate(H) and Gate(L) Risetime vs. Load Capacitance.
Figure 5: Percent Output Error vs. DAC Output
Voltage Setting, VID4 = 1.
VCC = 12V
TA = 25°C
125
100
0.35
0.30
75
0.25
50
0.20
25
0.15
0
0
2000
4000
6000
8000
10000
12000
14000
Output Error (%)
Risetime (ns)
150
16000
Load Capacitance (pF)
0.10
0.05
0
−0.05
−0.10
Figure 3: DAC Output Voltage vs. Temperature,
DAC Code = 00001.
VCC = 12V
TA = 25°C
VID4 = 1
−0.15
−0.20
DAC Output Voltage
Deviation (%)
−0.25
2.125 2.225 2.325 2.425 2.525 2.625 2.725 2.825 2.925 3.025 3.125 3.225 3.335 3.425 3.525
0.10
DAC Output Voltage Setting (V)
VCC = 12V
0.05
0
−0.05
−0.10
−0.15
0
20
40
60
80
100
120
Junction Temperature (°C)
Application Information
Theory Of Operation
PWM
Comparator
+
CS51313
Typical Performance Characteristics
V2TM Control Method
The V2TM method of control uses a ramp signal that is generated by the ESR of the output capacitors. This ramp is
proportional to the AC current through the main inductor
and is offset by the value of the DC output voltage. This
control scheme inherently compensates for variation in
either line or load conditions, since the ramp signal is generated from the output voltage itself. This control scheme
differs from traditional techniques such as voltage mode,
which generates an artificial ramp, and current mode,
which generates a ramp from inductor current.
GATE(H)
C
–
GATE(L)
Output
Voltage
Feedback
Ramp Signal
VFB
Error
Amplifier
COMP
Figure 6: V2
TM
6
Error
Signal
Control Diagram
–
E
+
Reference
Voltage
CS51313
Application Information: continued
Programmable Output
The V2TM control method is illustrated in Figure 6. The output voltage is used to generate both the error signal and
the ramp signal. Since the ramp signal is simply the output
voltage, it is affected by any change in the output regardless of the origin of that change. The ramp signal also contains the DC portion of the output voltage, which allows
the control circuit to drive the main switch to 0% or 100%
duty cycle as required.
A change in line voltage changes the current ramp in the
inductor, affecting the ramp signal, which causes the V2TM
control scheme to compensate the duty cycle. Since the
change in inductor current modifies the ramp signal, as in
current mode control, the V2TM control scheme has the same
advantages in line transient response.
A change in load current will have an affect on the output
voltage, altering the ramp signal. A load step immediately
changes the state of the comparator output, which controls
the main switch. Load transient response is determined
only by the comparator response time and the transition
speed of the main switch. The reaction time to an output
load step has no relation to the crossover frequency of the
error signal loop, as in traditional control methods.
The error signal loop can have a low crossover frequency,
since transient response is handled by the ramp signal
loop. The main purpose of this ‘slow’ feedback loop is to
provide DC accuracy. Noise immunity is significantly
improved, since the error amplifier bandwidth can be
rolled off at a low frequency. Enhanced noise immunity
improves remote sensing of the output voltage, since the
noise associated with long feedback traces can be effectively filtered.
Line and load regulation are drastically improved because
there are two independent voltage loops. A voltage mode
controller relies on a change in the error signal to compensate for a deviation in either line or load voltage. This
change in the error signal causes the output voltage to
change corresponding to the gain of the error amplifier,
which is normally specified as line and load regulation.
A current mode controller maintains fixed error signal
under deviation in the line voltage, since the slope of the
ramp signal changes, but still relies on a change in the error
signal for a deviation in load. The V2TM method of control
maintains a fixed error signal for both line and load variation, since the ramp signal is affected by both line and load.
The CS51313 is designed to provide two methods for programming the output voltage of the power supply. A five
bit on board digital to analog converter (DAC) is used to
program the output voltage within two different ranges.
The first range is 2.125V to 3.525V in 100mV steps, the second is 1.325V to 2.075V in 50mV steps, depending on the
digital input code. If all five bits are left open, the CS51313
enters adjust mode. In adjust mode, the designer can
choose any output voltage by using resistor divider feedback to the VFB pin, as in traditional controllers. The
CS51313 is specifically designed to meet or exceed Intel’s
Pentium® II specifications.
Error Amplifier
An inherent benefit of the V2TM control topology is that
there is no large bandwidth requirement on the error
amplifier design. The reaction time to an output load step
has no relation to the crossover frequency, since transient
response is handled by the ramp signal loop. The main
purpose of this”slow”feedback loop is to provide DC accuracy. Noise immunity is significantly improved, since the
error amplifier bandwidth can be rolled off at a low frequency. Enhanced noise immunity improves remote sensing of the output voltage, since the noise associated with
long feedback traces can be effectively filtered. The COMP
pin is the output of the error amplifier and a capacitor to
Gnd compensates the error amplifier loop. Additionally,
through the built-in offset on the PWM Comparator noninverting input, the COMP pin provides the hiccup timing
for the Overcurrent Protection, the Soft Start function that
minimizes inrush currents during regulator power-up and
switcher output enable.
Reference Voltage
The CS51313 has a precision reference trimmed to 1.5%
over temperature, which is externally available for use by
other power supplies on the motherboard. For instance, the
VREF pin can be used to configure an LDO controller that
drives either a MOSFET or a bipolar transistor. The compensation criteria on this LDO controller is set by the
dynamic performance requirement on the overall power
supply. The following circuit demonstrates the typical connections required to implement an LDO controller using
the CS51313 VREF pin.
Constant Off-Time
To minimize transient response, the CS51313 uses a
Constant Off-Time method to control the rate of output
pulses. During normal operation, the Off-Time of the high
side switch is terminated after a fixed period, set by the
COFF capacitor. Every time the VFB pin exceeds the COMP
pin voltage an Off-Time is initiated. To maintain regulation, the V2TM Control Loop varies switch On-Time. The
PWM comparator monitors the output voltage ramp, and
terminates the switch On-Time.
Constant Off-Time provides a number of advantages.
Switch duty Cycle can be adjusted from 0 to 100% on a
pulse-by pulse basis when responding to transient conditions. Both 0% and 100% Duty Cycle operation can be
maintained for extended periods of time in response to
Load or Line transients.
+1.5V
+3.3V
External N-FET
+12V
CIN
R1
21.9K
0.5%
R2
100K
0.5%
CO
+
VREF
Figure 7: VREF used in an N-FET LDO regulator.
The applications diagram shows a pair of linear regulators
for VGTL and VCLOCK. The 1.23V VREF of the CS51313 is
used as the reference for both regulators. The feedback
7
CS51313
Application Information: continued
limit threshold voltage. In this case, the PWM control loop
has achieved regulation and the initial pulse is then followed by a constant off time as programmed by the COFF
capacitor. The COMP capacitor will continue to slowly
charge and the regulator output voltage will follow it, less
the 1.1V PWM offset, until it achieves the voltage programmed by the DAC’s VID input. The Error Amp will
then source or sink current to the COMP cap as required to
maintain the correct regulator DC output voltage. Since the
rate of increase of the COMP pin voltage is typically set
much slower than the regulator’s slew capability, inrush
current, output voltage, and duty cycle all gradually
increase from zero. (See Figures 8, 9, and 10).
resistors determine the output voltage for each regulator.
In this case, it will be 1.5V @ 3A for VGTL and 2.5V @ 1A for
VCLOCK. In Figure 7 the ratio of resistor R1 to resistor R2 is
(VOUT/VREF) - 1, where VOUT = 1.5V and VREF = 1.23V. The
same formula can be used to determine the ratio of the
feedback resistors needed to implement a 2.5V linear regulator (VOUT = 2.5V).To negate the bias current of the operational amplifier, a resistor with a value equal to the parallel
combination of the feedback resistors (R1//R2) is connected in series with the non-inverting input of this operational
amplifier. R2 sets the minimum output current, (IMIN =
VREF/R2).
The pass transistor must be able to dissipate the power
adequately while keeping the junction temperature below
the maximum specified by the manufacturer. For example,
with VGTL output of 1.5V, input voltage of 3.3V, and output DC current of 3A, the pass transistor dissipates (3.3V 1.5V) × 3A = 5.4W.
Sufficient output capacitance must be added to ensure that
the output voltage remains within specification during
transient loading. For example, the GTL bus load can ramp
from 0 to 2.7A at a rate of 8A/µs. The designer needs to
verify that the circuit will meet these requirements using
the transistor and operational amplifier chosen.
Start-up @
VCC > 8.4V
Startup
The CS51313 provides a controlled startup of regulator output voltage and features Programmable Soft Start implemented through the Error Amp and external Compensation
Capacitor. This feature, combined with overcurrent protection, prevents stress to the regulator power components
and overshoot of the output voltage during startup.
As Power is applied to the regulator, the CS51313
Undervoltage Lockout circuit (UVL) monitors the IC’s supply voltage (VCC) which is typically connected to the +12V
output of the AC-DC power supply. The UVL circuit prevents the NFET gates from being activated until VCC
exceeds the 8.4V (typ) threshold. Hysteresis of 300mV (typ)
is provided for noise immunity. The Error Amp Capacitor
connected to the COMP pin is charged by a 30µA current
source. This capacitor must be charged to 1.1V (typ) so that
it exceeds the PWM comparator’s offset before the V2TM
PWM control loop permits switching to occur.
When VCC has exceeded 8.4V and COMP has charged to
1.1V, the upper Gate driver (GATE(H)) is activated, turning on the upper FET. This causes current to flow through
the output inductor and into the output capacitors and
load according to the following equation:
I = (VIN – VOUT) ×
Figure 8: Normal Startup (2ms/div).
Channel 1 - Regulator Output Voltage (1V/div)
Channel 2 - COMP Pin (1V/div)
Channel 3 - VCC (10V/div)
Channel 4 - Regulator Input Voltage (5V/div)
Start-up @
VCC > 8.4V
Initial Pulse until VOUT
> COMP + PWM Offset
T
L
GATE(H) and the upper NFET remain on and inductor
current ramps up until the initial pulse is terminated by
either the PWM control loop or the overcurrent protection.
This initial surge of in-rush current minimizes startup time,
but avoids overstressing of the regulator’s power components.
The PWM comparator will terminate the initial pulse if the
regulator output exceeds the voltage on the COMP pin
plus the 1.1V PWM comparator offset before the voltage
drop across the current sense resistor exceeds the current
Figure 9: Normal Startup showing initial pulse followed by Soft Start
(20µs/div).
Channel 1 - Regulator Output Voltage (0.2V/div)
Channel 2 – Inductor Switching Node (5V/div)
Channel 3 - VCC (10V/div)
Channel 4 - Regulator Input Voltage (5V/div)
8
When driving large capacitive loads, the COMP must
charge slowly enough to avoid tripping the CS51313 overcurrent protection. The following equation can be used to
ensure unconditional startup:
ICHG
CCOMP
<
ILIM − ILOAD
COUT
where
ICHG = COMP Source Current (30µA typical);
CCOMP = COMP Capacitor value (0.1µF typical);
ILIM = Current Limit Threshold;
ILOAD = Load Current during startup;
COUT = Total Output Capacitance.
Duty Cycle = VOUT / VIN
0.27V / 3.54V = 7% ≈ 5.2%
Normal Operation
During Normal operation, Switch Off-Time is constant and
set by the COFF capacitor. Switch On-Time is adjusted by
the V2TM Control loop to maintain regulation. This results in
changes in regulator switching frequency, duty cycle, and
output ripple in response to changes in load and line.
Output voltage ripple will be determined by inductor ripple current and the ESR of the output capacitors
Figure 10: Pulse-by-Pulse Regulation during Soft Start (2µs/div).
Channel 1 - Regulator Output Voltage (0.2V/div)
Channel 2 – Inductor Switching Node (5V/div)
Channel 3 - VCC (10V/div)
Channel 4 - Regulator Input Voltage (5V/div)
If the voltage across the Current Sense resistor generates a
voltage difference between the VFB and VOUT pins that
exceeds the OVC Comparator Offset Voltage (86mV typical), the Fault latch is set. This causes the COMP pin to be
quickly discharged, turning off GATE(H) and the upper
NFET since the voltage on the COMP pin is now less than
the 1.1V PWM comparator offset. The Fault latch is reset
when the voltage on the COMP decreases below the
Discharge threshold voltage (0.25V typical). The COMP
capacitor will again begin to charge, and when it exceeds
the 1.1V PWM comparator offset, the regulator output will
Soft Start normally (see Figure 11).
Because the start-up circuitry depends on the current sense
function, a current sense resistor should always be used.
Transient Response
The CS51313 V2TM Control Loop’s 200ns reaction time provides unprecedented transient response to changes in
input voltage or output current. Pulse-by-pulse adjustment
of duty cycle is provided to quickly ramp the inductor current to the required level. Since the inductor current cannot
be changed instantaneously, regulation is maintained by
the output capacitor(s) during the time required to slew the
inductor current.
Overall load transient response is further improved
through a feature called “Adaptive Voltage Positioning”.
This technique pre-positions the output voltage to reduce
total output voltage excursions during changes in load.
Holding tolerance to 1% allows the error amplifiers reference voltage to be targeted +25mV high without compromising DC accuracy. A “Droop Resistor”, implemented
through a PC board trace, connects the Error Amps feedback pin (VFB) to the output capacitors and load and carries
the output current. With no load, there is no DC drop
across this resistor, producing an output voltage tracking
the Error amps, including the +25mV offset. When the full
load current is delivered, a 50mV drop is developed across
this resistor. This results in output voltage being offset 25mV low.
OCP @
VCC > 8.4V
The result of Adaptive Voltage Positioning is that additional margin is provided for a load transient before reaching
the output voltage specification limits. When load current
suddenly increases from its minimum level, the output is
pre-positioned +25mV. Conversely, when load current
suddenly decreases from its maximum level, the output is
pre-positioned -25mV. For best Transient Response, a combination of a number of high frequency and bulk output
capacitors are usually used.
Soft Start @
COMP > 1.1V
Figure 11: Startup with COMP pre-charged to 2V (2ms/div).
Channel 1 - Regulator Output Voltage (1V/div)
Channel 2 - COMP Pin (1V/div)
Channel 3 - VCC (10V/div)
Channel 4 - Regulator Input Voltage (5V/div)
9
CS51313
Application Information: continued
CS51313
Application Information: continued
Slope Compensation
The V2TM control method uses a ramp signal, generated by
the ESR of the output capacitors, that is proportional to the
ripple current through the inductor. To maintain regulation, the V2TM control loop monitors this ramp signal,
through the PWM comparator, and terminates the switch
on-time.
The stringent load transient requirements of modern
microprocessors require the output capacitors to have very
low ESR. The resulting shallow slope presented to the
PWM comparator, due to the very low ESR, can lead to
pulse width jitter and variation caused by both random or
synchronous noise.
Adding slope compensation to the control loop, avoids
erratic operation of the PWM circuit, particularly at lower
duty cycles and higher frequencies, where there is not
enough ramp signal, and provides a more stable switchpoint.
The scheme that prevents that switching noise prematurely
triggers the PWM circuit consists of adding a positive voltage slope to the output of the Error Amplifier (COMP pin)
during an off-time cycle.
The circuit that implements this function is shown in
Figure 12.
16
COMP
CCOMP
CS51313
R2
C1
R1
12
GATE(L)
To Synchronous FET
Figure 12: Small RC filter provides the proper voltage ramp at the
beginning of each on-time cycle.
The ramp waveform is generated through a small RC filter
that provides the proper voltage ramp at the beginning of
each on-time cycle. The resistors R1 and R2 in the circuit of
Figure 12 form a voltage divider from the GATE(L) output,
superimposing a small artificial ramp on the output of the
error amplifier. It is important that the series combination
R1/R2 is high enough in resistance not to load down and
negatively affect the slew rate on the GATE(L) pin.
Protection and Monitoring Features
threshold, the current sense comparator allows the fault
latch to be set. This causes the regulator to stop switching.
During this over current condition, the CS51313 stays off
for the time it takes the COMP pin capacitor to discharge
to its lower 0.25V threshold. As soon as the COMP pin
reaches 0.25V, the Fault latch is reset (no overcurrent condition present) and the COMP pin is charged with a 30µA
current source to a voltage 1.1V greater than the VFB voltage. Only at this point the regulator attempts to restart normally. The CS51313 will operate initially with a duty cycle
whose value depends on how low the VFB voltage was
during the overcurrent condition (whether hiccup mode
was due to excessive current or hard short). This protection scheme minimizes thermal stress to the regulator components, input power supply, and PC board traces, as the
over current condition persists. Upon removal of the overload, the fault latch is cleared, allowing normal operation
to resume.
Overvoltage Protection
Overvoltage protection (OVP) is provided as result of the
normal operation of the V2TM control topology and requires
no additional external components. The control loop
responds to an overvoltage condition within 200ns, causing the top MOSFET to shut off, disconnecting the regulator from its input voltage. This results in a “crowbar”
action to clamp the output voltage and prevents damage to
the load. The regulator will remain in this state until the
overvoltage condition ceases or the input voltage is pulled
low. Additionally, a dedicated Overvoltage protection
(OVP) output pin (pin 13) is provided in the CS51313. The
OVP signal will go high (overvoltage condition), if the output voltage (VCC(CORE)) exceeds the regulation voltage by
8.5% of the voltage set by the particular DAC code. The
OVP pin can source up to 25mA of current that can be
used to drive an SCR to crowbar the power supply.
Power-Good Circuit
The Power-Good pin (pin 14) is an open-collector signal
consistent with TTL DC specifications. It is externally
pulled up, and is pulled low (below 0.3V) when the regulator output voltage typically exceeds ± 8.5% of the nominal
output voltage. Maximum output voltage deviation before
Power-Good is pulled low is ± 12%.
Output Enable
On/off control of the regulator outputs can be implemented by pulling the COMP pins low. It is required to pull the
COMP pins below the 1.1V PWM comparator offset voltage in order to disable switching on the GATE drivers.
Overcurrent Protection
A loss-less hiccup mode current limit protection feature is
provided, requiring only the COMP capacitor to implement. The CS51313 provides overcurrent protection by
sensing the current through a “Droop” resistor, using an
internal current sense comparator. The comparator compares the voltage drop across the “Droop” resistor to an
internal reference voltage of 86mV (typical).
If the voltage drop across the “Droop” resistor exceeds this
CS51313-based VCC(CORE)
Buck Regulator Design Example
Step 1: Definition of the design specifications
In computer motherboard applications the input voltage
comes from the “silver box” power supply. 5V ± 5% is
used for conversion to output voltage, and 12V ± 5% is
used for the external NFET gate voltage and circuit bias.
10
The CPU VCC(CORE) tolerance can be affected by any or all
of the following reasons:
1) buck regulator output voltage setpoint accuracy;
2) output voltage change due to discharging or charging of
the bulk decoupling capacitors during a load current transient;
3) output voltage change due to the ESR and ESL of the
bulk and high frequency decoupling capacitors, circuit
traces, and vias;
4) output voltage ripple and noise.
Budgeting the tolerance is left up to the designer who must
take into account all of the above effects and provide a
VCC(CORE) that will meet the specified tolerance at the
CPU’s inputs.
The designer must also ensure that the regulator component junction temperatures are kept within the manufacturer’s specified ratings at full load and maximum ambient
temperature. As computer motherboards become increasingly complex, regulator size also becomes important, as
there is less space available for the CPU power supply.
ESRMAX =
Number of capacitors =
ESRCAP
,
ESRMAX
where
ESRCAP = maximum ESR per capacitor (specified in
manufacturer’s data sheet);
ESRMAX = maximum allowable ESR.
The actual output voltage deviation due to ESR can then be
verified and compared to the value assigned by the designer:
∆VESR = ∆IOUT × ESRMAX
Similarly, the maximum allowable ESL is calculated from
the following formula:
These components must be selected and placed carefully to
yield optimal results. Capacitors should be chosen to provide acceptable ripple on the regulator output voltage. Key
specifications for output capacitors are their ESR
(Equivalent Series Resistance), and ESL (Equivalent Series
Inductance). For best transient response, a combination of
low value/high frequency and bulk capacitors placed close
to the load will be required.
In order to determine the number of output capacitors the
maximum voltage transient allowed during load transitions has to be specified. The output capacitors must hold
the output voltage within these limits since the inductor
current can not change with the required slew rate. The
output capacitors must therefore have a very low ESL and
ESR.
The voltage change during the load current transient is:
(
∆VESR
,
∆IOUT
where ∆VESR = change in output voltage due to ESR
(assigned by the designer).
Once the maximum allowable ESR is determined, the
number of output capacitors can be found by using the formula
Step 2: Selection of the Output Capacitors
∆VOUT = ∆IOUT ×
CS51313
Application Information: continued
ESLMAX =
∆VESL × ∆t
,
∆I
where
∆I/∆T = load current slew rate (as high as 20A/µs);
∆VESL = change in output voltage due to ESL.
The actual maximum allowable ESL can be determined by
using the equation:
ESLMAX =
ESLCAP
Number of output capacitors
,
where ESLCAP = maximum ESL per capacitor (it is estimated that a 10 × 12mm Aluminum Electrolytic capacitor has
approximately 4nH of package inductance).
The actual output voltage deviation due to the actual maximum ESL can then be verified:
)
tTR
ESL
+ ESR +
,
COUT
∆t
where
∆IOUT / ∆t = load current slew rate;
∆IOUT = load transient;
∆t = load transient duration time;
ESL = Maximum allowable ESL including capacitors,
circuit traces, and vias;
ESR = Maximum allowable ESR including capacitors
and circuit traces;
tTR = output voltage transient response time.
The designer has to independently assign values for the
change in output voltage due to ESR, ESL, and output
capacitor discharging or charging. Empirical data indicates
that most of the output voltage change (droop or spike
depending on the load current transition) results from the
total output capacitor ESR.
The maximum allowable ESR can then be determined
according to the formula
∆VESL =
ESLMAX × ∆I
.
∆t
The designer now must determine the change in output
voltage due to output capacitor discharge during the transient:
∆VCAP =
∆I × ∆tTR
COUT
,
where
∆tTR = the output voltage transient response time
(assigned by the designer);
∆VCAP = output voltage deviation due to output capacitor discharge;
∆I = Load step.
11
CS51313
Application Information: continued
The total change in output voltage as a result of a load current transient can be verified by the following formula:
Step 4: Selection of the Output Inductor
The inductor should be selected based on its inductance,
current capability, and DC resistance. Increasing the inductor value will decrease output voltage ripple, but degrade
transient response. There are many factors to consider in
selecting the inductor including cost, efficiency, EMI and
ease of manufacture. The inductor must be able to handle
the peak current at the switching frequency without saturating, and the copper resistance in the winding should be
kept as low as possible to minimize resistive power loss.
There are a variety of materials and types of magnetic
cores that could be used for this application. Among them
are ferrites, molypermalloy cores (MPP), amorphous and
powdered iron cores. Powdered iron cores are very commonly used. Powdered iron cores are very suitable due to
their high saturation flux density and have low loss at high
frequencies, a distributed gap and exhibit very low EMI.
The inductor value can be determined by:
∆VOUT = ∆VESR + ∆VESL + ∆VCAP
Step 3: Selection of the Duty Cycle,
Switching Frequency, Switch On-Time (TON)
and Switch Off-Time (TOFF)
The duty cycle of a buck converter (including parasitic
losses) is given by the formula:
Duty Cycle = D =
VOUT + (VHFET + VL + VDROOP)
,
VIN + VLFET − VHFET − VL
where
VOUT = buck regulator output voltage;
VHFET = high side FET voltage drop due to RDS(ON);
VL = output inductor voltage drop due to inductor wire
DC resistance;
VDROOP = droop (current sense) resistor voltage drop;
VIN = buck regulator input voltage;
VLFET = low side FET voltage drop due to RDS(ON).
The switch On-Time (time during which the switching
MOSFET in a synchronous buck topology is conducting) is
determined by:
Duty Cycle
,
FSW
∆IL =
∆VOUT
ESRMAX
ESRMAX =
− TON,
,
,
∆VOUT
∆IL
,
where
ESRMAX = maximum allowable ESR;
∆VOUT = 1% × VOUT = maximum allowable output voltage ripple ( budgeted by the designer );
∆IL = inductor ripple current;
VOUT = output voltage.
The number of output capacitors is determined by:
The COFF capacitor value has to be selected in order to set
the Off-Time, TOFF, above:
COFF =
L
Rearranging we have:
The switch Off-Time (time during which the switching
MOSFET is not conducting) can be determined by:
FSW
VOUT × TOFF
∆IL =
Step 3b: Calculation of Switch Off-Time
TOFF =
,
where
∆IL = inductor ripple current;
VOUT = output voltage;
TOFF = switch Off-Time;
L = inductor value.
The designer can now verify if the number of output
capacitors from step 2 will provide an acceptable output
voltage ripple (1% of output voltage is common). The formula below is used:
where FSW = regulator switching frequency selected by the
designer.
Higher operating frequencies allow the use of smaller
inductor and capacitor values. Nevertheless, it is common
to select lower frequency operation because a higher frequency results in lower efficiency due to MOSFET gate
charge losses. Additionally, the use of smaller inductors at
higher frequencies results in higher ripple current, higher
output voltage ripple, and lower efficiency at light load
currents.
1
∆Ι
where
VIN = input voltage;
VOUT = output voltage;
tTR = output voltage transient response time (assigned
by the designer);
∆I = load transient.
The inductor ripple current can then be determined:
Step3a: Calculation of Switch On-Time
TON =
(VIN − VOUT) × tTR
L=
Period × (1 − D)
,
3980
where
3980 is a characteristic factor of the CS51313;
D = Duty Cycle.
Number of capacitors =
ESRCAP
,
ESRMAX
where ESRCAP = maximum ESR per capacitor (specified in
manufacturer’s data sheet).
12
The designer must also verify that the inductor value
yields reasonable inductor peak and valley currents (the
inductor current is a triangular waveform):
IL(PEAK) = IOUT +
∆IL
2
NCIN =
where
NCIN = number of input capacitors;
ICIN(RMS) = total input RMS current;
IRIPPLE = input capacitor ripple current rating (specified
in manufacturer’s data sheets).
The total input capacitor ESR needs to be determined in
order to calculate the power dissipation of the input capacitors:
,
where
IL(PEAK) = inductor peak current;
IOUT = load current;
∆IL = inductor ripple current.
IL(VALLEY) = IOUT −
∆IL
2
ICIN(RMS)
,
IRIPPLE
ESRCIN =
,
ESRCAP
,
NCIN
where
ESRCIN = total input capacitor ESR;
ESRCAP = maximum ESR per capacitor (specified in
manufacturer’s data sheets);
NCIN = number of input capacitors.
Once the total ESR of the input capacitors is known, the
input capacitor ripple voltage can be determined using the
formula:
where IL(VALLEY) = inductor valley current.
Given the requirements of an application such as a buck
converter, it is found that a toroid powdered iron core is
quite suitable due to its low cost, low core losses at the
switching frequency, and low EMI.
Step 5: Selection of the Input Capacitors
VCIN(RMS) = ICIN(RMS) × ESRCIN,
These components must be selected and placed carefully to
yield optimal results. Capacitors should be chosen to provide acceptable ripple on the input supply lines. A key
specification for input capacitors is their ripple current rating. The input capacitor should also be able to handle the
input RMS current IIN(RMS).
The combination of the input capacitors CIN discharges
during the on-time.
The input capacitor discharge current is given by:
where
VCIN(RMS) = input capacitor RMS voltage;
ICIN(RMS) = total input RMS current;
ESRCIN = total input capacitor ESR.
The designer must determine the input capacitor power
loss in order to ensure there isn’t excessive power dissipation through these components. The following formula is
used:
ICINDIS(RMS) =
PCIN(RMS) = ICIN(RMS)2 × ESRCIN
(IL(PEAK)2 + (IL(PEAK) × IL(VALLEY)) + IL(VALLEY)2 × D
3
where
PCIN(RMS) = input capacitor RMS power dissipation;
ICIN(RMS) = total input RMS current;
ESRCIN = total input capacitor ESR.
,
where
ICINDIS(RMS) = input capacitor discharge current;
IL(PEAK) = inductor peak current;
IL(VALLEY) = inductor valley current.
CIN charges during the off-time, the average current
through the capacitor over one switching cycle is zero:
ICIN(CH) = ICIN(DIS) ×
D
1−D
Step 6: Selection of the Input Inductor
A CPU switching regulator, such as the one in a buck
topology, must not disturb the primary +5V supply. One
method of achieving this is by using an input inductor and
a bypass capacitor. The input inductor isolates the +5V
supply from the noise generated in the switching portion
of the microprocessor buck regulator and also limits the
inrush current into the input capacitors upon power up.
The inductor’s limiting effect on the input current slew rate
becomes increasingly beneficial during load transients. The
worst case is when the CPU load changes from no load to
full load (load step), a condition under which the highest
voltage change across the input capacitors is also seen by
the input inductor. The inductor successfully blocks the
ripple current while placing the transient current requirements on the input bypass capacitor bank, which has to
initially support the sudden load change.
The minimum inductance value for the input inductor is
therefore:
,
where
ICIN(CH) = input capacitor charge current;
ICIN(DIS) = input capacitor discharge current;
D = Duty Cycle.
The total Input RMS current is:
ICIN(RMS) =
(ICIN(DIS)2 × D) + (ICIN(CH)2 × (1 − D))
The number of input capacitors required is then determined by:
13
CS51313
Application Information: continued
CS51313
Application Information: continued
LIN =
Two considerations complicate the task of estimating
switching times. First, since the magnitude of the input
capacitance, CISS, varies with VDS, the RC time constant
determined by the gate-drive impedance and CISS changes
during the switching cycle. Consequently, computation of
the rise time of the gate voltage by using a specific gatedrive impedance and input capacitance yields only a
rough estimate. The second consideration is the effect of
the “Miller” capacitance, CRSS, which is referred to as CDG
in the following discussion. For example, when a device is
on, VDS(ON) is fairly small and VGS is about 12V. CDG is
charged to VDS(ON) − VGS, which is a negative potential if
the drain is considered the positive electrode. When the
drain is “off”, CDG is charged to quite a different potential.
In this case the voltage across CDG is a positive value since
the potential from gate-to-source is near zero volts and VDS
is essentially the drain supply voltage. During turn-on and
turn-off, these large swings in gate-to-drain voltage tax the
current sourcing and sinking capabilities of the gate drive.
In addition to charging and discharging CGS, the gate drive
must also supply the displacement current required by
CDG(IGATE = Cdg dVdg/dt). Unless the gate-drive
impedance is very low, the VGS waveform commonly
plateaus during rapid changes in the drain-to-source voltage.
The most important aspect of FET performance is the Static
Drain-To-Source On-Resistance (RDS(ON)), which effects
regulator efficiency and FET thermal management requirements. The On-Resistance determines the amount of current a FET can handle without excessive power dissipation
that may cause overheating and potentially catastrophic
failure. As the drain current rises, especially above the continuous rating, the On-Resistance also increases. Its positive temperature coefficient is between +0.6%/C and
+0.85%/C. The higher the On-Resistance the larger the
conduction loss is. Additionally, the FET gate charge
should be low in order to minimize switching losses and
reduce power dissipation.
Both logic level and standard FETs can be used. The reference designs derive gate drive from the 12V supply, which
is generally available in most computer systems and utilizes logic level FETs.
Voltage applied to the FET gates depends on the application circuit used. Both upper and lower gate driver outputs
are specified to drive to within 1.5V of ground when in the
low state and to within 2V of their respective bias supplies
when in the high state. In practice, the FET gates will be
driven rail-to-rail due to overshoot caused by the capacitive load they present to the controller IC.
∆V
,
(dI/dt)MAX
where
LIN = input inductor value;
∆V = voltage seen by the input inductor during a full
load swing;
(dI/dt)MAX = maximum allowable input current slew
rate (0.1A/µs for a Pentium® II power supply).
The designer must select the LC filter pole frequency so
that at least 40dB attenuation is obtained at the regulator
switching frequency. The LC filter is a double-pole network with a slope of −2, a roll-off rate of –40dB/dec, and a
corner frequency:
fC =
1
2π LC
,
where
L = input inductor;
C = input capacitor(s).
Step 7: Selection of the Switching FET
FET Basics
The use of the MOSFET as a power switch is propelled by
two reasons: 1) Its very high input impedance; and 2) Its very
fast switching times. The electrical characteristics of a MOSFET are considered to be those of a perfect switch. Control
and drive circuitry power is therefore reduced. Because the
input impedance is so high, it is voltage driven. The input
of the MOSFET acts as if it were a small capacitor, which
the driving circuit must charge at turn on. The lower the
drive impedance, the higher the rate of rise of VGS, and the
faster the turn- on time. Power dissipation in the switching
MOSFET consists of 1) conduction losses, 2) leakage losses,
3) turn-on switching losses, 4) turn-off switching losses,
and 5) gate-transitions losses. The latter three losses are
proportional to frequency. For the conducting power dissipation rms values of current and resistance are used for
true power calculations. The fast switching speed of the
MOSFET makes it indispensable for high-frequency power
supply applications. Not only are switching power losses
minimized, but also the maximum usable switching frequency is considerably higher. Switching time is independent of temperature. Also, at higher frequencies, the use of
smaller and lighter components (transformer, filter choke,
filter capacitor) reduces overall component cost while
using less space for more efficient packaging at lower
weight.
The MOSFET has purely capacitive input impedance. No
DC current is required. It is important to keep in mind the
drain current of the FET has a negative temperature coefficient. Increase in temperature causes higher on-resistance
and greater leakage current. For switching circuits, VDS(ON)
should be low to minimize power dissipation at a given ID,
and VGS should be high to accomplish this. MOSFET
switching times are determined by device capacitance,
stray capacitance, and the impedance of the gate drive circuit. Thus the gate driving circuit must have high momentary peak current sourcing and sinking capability for
switching the MOSFET. The input capacitance, output
capacitance and reverse-transfer capacitance also increase
with increased device current rating.
Step 7a - Selection of the switching (upper) FET
The designer must ensure that the total power dissipation
in the FET switch does not cause the power component’s
junction temperature to exceed 150°C.
The maximum RMS current through the switch can be
determined by the following formula:
IRMS(H) =
(IL(PEAK)2 + (IL(PEAK) × IL(VALLEY)) + IL(VALLEY)2 × D
3
14
,
where
IRMS(H) = maximum switching MOSFET RMS current;
IL(PEAK) = inductor peak current;
IL(VALLEY) = inductor valley current;
D = Duty Cycle.
Once the RMS current through the switch is known, the
switching MOSFET conduction losses can be calculated:
where
PRMSL = lower MOSFET conduction losses;
IOUT = load current;
D = Duty Cycle;
RDS(ON) = lower FET drain-to-source on-resistance.
The synchronous MOSFET has no switching losses, except
for losses in the internal body diode, because it turns on
into near zero voltage conditions. The MOSFET body
diode will conduct during the non-overlap time and the
resulting power dissipation (neglecting reverse recovery
losses) can be calculated as follows:
PRMS(H) = IRMS(H)2 × RDS(ON)
where
PRMS(H) = switching MOSFET conduction losses;
IRMS(H) = maximum switching MOSFET RMS current;
RDS(ON) = FET drain-to-source on-resistance
The upper MOSFET switching losses are caused during
MOSFET switch-on and switch-off and can be determined
by using the following formula:
PSWL = VSD × ILOAD × non-overlap time × FSW,
where
PSWL = lower FET switching losses;
VSD = lower FET source-to-drain voltage;
ILOAD = load current
Non-overlap time = GATE(L)-to-GATE(H) or GATE(H)to-GATE(L) delay (from CS51313 data sheet Electrical
Characteristics section);
FSW = switching frequency.
The total power dissipation in the synchronous (lower)
MOSFET can then be calculated as:
PSWH = PSWH(ON) + PSWH(OFF)
=
VIN × IOUT × (tRISE + tFALL)
6T
,
where
PSWH(ON) = upper MOSFET switch-on losses;
PSWH(OFF) = upper MOSFET switch-off losses;
VIN = input voltage;
IOUT = load current;
tRISE = MOSFET rise time (from FET manufacturer’s
switching characteristics performance curve);
tFALL = MOSFET fall time (from FET manufacturer’s
switching characteristics performance curve);
T = 1/FSW = period.
The total power dissipation in the switching MOSFET can
then be calculated as:
PLFET(TOTAL) = PRMSL + PSWL,
where
PLFET(TOTAL) = Synchronous (lower) FET total losses;
PRMSL = Switch Conduction Losses;
PSWL = Switching losses.
Once the total power dissipation in the synchronous FET is
known the maximum FET switch junction temperature can
be calculated:
TJ = TA + [PLFET(TOTAL) × RθJA],
PHFET(TOTAL) = PRMSH + PSWH(ON) + PSWH(OFF),
where
TJ = MOSFET junction temperature;
TA = ambient temperature;
PLFET(TOTAL) = total synchronous (lower) FET losses;
RθJA = lower FET junction-to-ambient thermal resistance.
where
PHFET(TOTAL) = total switching (upper) MOSFET losses;
PRMSH = upper MOSFET switch conduction Losses;
PSWH(ON) = upper MOSFET switch-on losses;
PSWH(OFF) = upper MOSFET switch-off losses.
Once the total power dissipation in the switching FET is
known, the maximum FET switch junction temperature
can be calculated:
Step 8: Control IC Power Dissipation
The power dissipation of the IC varies with the MOSFETs
used, VCC, and the CS51313 operating frequency. The average MOSFET gate charge current typically dominates the
control IC power dissipation.
The IC power dissipation is determined by the formula:
TJ = TA + [PHFET(TOTAL) × RθJA],
where
TJ = FET junction temperature;
TA = ambient temperature;
PHFET(TOTAL) = total switching (upper) FET losses;
RθJA = upper FET junction-to-ambient thermal resistance
PCONTROLIC = ICCVCC + PGATE(H) + PGATE(L),
The switch conduction losses for the lower FET can be calculated as follows:
where
PCONTROLIC = control IC power dissipation;
ICC = IC quiescent supply current;
VCC = IC supply voltage;
PGATE(H) = upper MOSFET gate driver (IC) losses;
PGATE(L) = lower MOSFET gate driver (IC) losses.
The upper (switching) MOSFET gate driver (IC) losses are:
PRMSL = IRMS2 × RDS(ON) = [IOUT × (1 − D)]2 × RDS(ON),
PGATE(H) = QGATE(H) × FSW × VGATE(H),
Step 7b: Selection of the synchronous (lower) FET
15
CS51313
Application Information: continued
CS51313
Application Information: continued
where
PGATE(H) = upper MOSFET gate driver (IC) losses;
QGATE(H) = total upper MOSFET gate charge;
FSW = switching frequency;
VGATE(H) = upper MOSFET gate voltage.
The lower (synchronous) MOSFET gate driver (IC) losses
are:
caused by sudden and fast load changes. These load transients can have slew rates as high as 20A/µs.
“Droop” Resistor for Adaptive Voltage Positioning
and Current Limit
Adaptive voltage positioning is used to help keep the output voltage within specification during load transients. To
implement adaptive voltage positioning a “Droop
Resistor” must be connected between the output inductor
and output capacitors and load. This resistor carries the
full load current and should be chosen so that both DC and
AC tolerance limits are met. An embedded PC trace resistor has the distinct advantage of near zero cost implementation. However, this droop resistor can vary due to three
reasons: 1) the sheet resistivity variation caused by variation in the thickness of the PCB layer; 2) the mismatch of
L/W; and 3) temperature variation.
PGATE(L) = QGATE(L) × FSW × VGATE(L),
where
PGATE(L) = lower MOSFET gate driver (IC) losses;
QGATE(L) = total lower MOSFET gate charge;
FSW = switching frequency;
VGATE(L) = lower MOSFET gate voltage.
The junction temperature of the control IC is primarily a
function of the PCB layout, since most of the heat is
removed through the traces connected to the pins of the
IC.
1) Sheet Resistivity
Step 9: Slope Compensation
For one ounce copper, the thickness variation is typically
1.26 mil to 1.48 mil. Therefore the error due to sheet resistivity is:
1.48 - 1.26
= ±8%.
1.37
Voltage regulators for today’s advanced processors are
expected to meet very stringent load transient requirements. One of the key factors in achieving tight dynamic
voltage regulation is low ESR at the CPU input supply
pins. Low ESR at the regulator output results in low output voltage ripple. The consequence is, however, that
there’s very little voltage ramp at the control IC feedback
pin (VFB) and regulator sensitivity to noise and loop instability are two undesirable effects that can surface. The performance of the CS51313-based CPU VCC(CORE) regulator is
improved when a fixed amount of slope compensation is
added to the output of the PWM Error Amplifier (COMP
pin) during the regulator Off-Time. Referring to Figure 12,
the amount of voltage ramp at the COMP pin is dependent
on the gate voltage of the lower (synchronous) FET and the
value of resistor divider formed by R1and R2.
VSLOPECOMP = VGATE(L) ×
(
2) Mismatch due to L/W
The variation in L/W is governed by variations due to the
PCB manufacturing process. The error due to L/W mismatch is typically 1%.
3) Thermal Considerations
Due to I2 × R power losses the surface temperature of the
droop resistor will increase causing the resistance to
increase. Also, the ambient temperature variation will contribute to the increase of the resistance, according to the
formula:
R = R20 [1+ α20(Τ−20)],
)
-t
R2
× (1 − e τ ),
R1 + R2
where
R20 = resistance at 20˚C;
where
VSLOPECOMP = amount of slope added;
VGATE(L) = lower MOSFET gate voltage;
R1, R2 = voltage divider resistors;
t = tOFF (switch off-time);
τ = RC constant determined by C1 and the parallel combination of R1, R2 (Figure 12), neglecting the low driver
output impedance
The artificial voltage ramp created by the slope compensation scheme results in improved control loop stability provided that the RC filter time constant is smaller than the
off-time cycle duration (time during which the lower MOSFET is conducting).
α=
0.00393
;
˚C
T= operating temperature;
R = desired droop resistor value.
For temperature T = 50˚C, the % R change = 12%.
Droop Resistor Tolerance
Tolerance due to sheet resistivity variation
±8%
Tolerance due to L/W error
1%
Tolerance due to temperature variation
12%
Total tolerance for droop resistor
21%
In order to determine the droop resistor value the nominal
voltage drop across it at full load has to be calculated. This
voltage drop has to be such that the output voltage at full
load is above the minimum DC tolerance spec:
Step 10: Selection of Current Limit Filter Components
The current limit filter is implemented by a 0.1µF ceramic
capacitor across and two 510Ω resistors in series with the
VFB and VOUT current limit comparator input pins. They
provide a time constant τ = RC = 100µs, which enables the
circuit to filter out noise and be immune to false triggering,
16
VDROOP(TYP) =
VDAC(MIN)-VDC(MIN)
1+RDROOP(TOLERANCE)
point:
.
RSENSE(MIN) = RSENSE(TYP) × 0.79,
Example: for a 450MHz Pentium®II, the DC accuracy spec
is 1.93 < VCC(CORE) < 2.07V, and the AC accuracy spec is
1.9V < VCC(CORE) < 2.1V. The CS51313 DAC output voltage
is +2.001V < VDAC < +2.049V. In order not to exceed the
DC accuracy spec, the voltage drop developed across the
resistor must be calculated as follows:
VDROOP(TYP) =
=
RSENSE(MAX) = RSENSE(TYP) × 1.21,
RSENSE(MAX) =
VTH(MIN)
77mV
=
= 4.8mΩ.
ICL(MIN)
16A
We select,
[VDAC(MIN)-VDC (MIN)]
1+RDROOP(TOLERANCE)
+2.001V-1.93V
1.21
CS51313
Application Information :continued
RSENSE(TYP) = 3.3mΩ.
We calculate the range of load currents that will cause the
internal current sense comparator to detect an overload
condition.
= 71mV.
With the CS51313 DAC accuracy being 1%, the internal
error amplifier’s reference voltage is trimmed so that the
output voltage will be 25mV high at no load. With no load,
there is no DC drop across the resistor, producing an output voltage tracking the error amplifier output voltage,
including the offset. When the full load current is delivered, a drop of -50mV is developed across the resistor.
Therefore, the regulator output is pre-positioned at 25mV
above the nominal output voltage before a load turn-on.
The total voltage drop due to a load step is ∆V-25mV and
the deviation from the nominal output voltage is 25mV
smaller than it would be if there was no droop resistor.
Similarly at full load the regulator output is pre-positioned
at 25mV below the nominal voltage before a load turn-off.
the total voltage increase due to a load turn-off is ∆V-25mV
and the deviation from the nominal output voltage is
25mV smaller than it would be if there was no droop resistor. This is because the output capacitors are pre-charged
to a value that is either 25mV above the nominal output
voltage before a load turn-on or, 25mV below the nominal
output voltage before a load turn-off .
Obviously, the larger the voltage drop across the droop
resistor (the larger the resistance), the worse the DC and
load regulation, but the better the AC transient response.
Nominal Current Limit Setpoint
From the overcurrent detection data in the electrical characteristics table:
VTH(TYP) = 86mV,
ICL(NOM) =
VTH(TYP)
= 86mV = 26A.
RSENSE(NOM)
3.3mΩ
Maximum Current Limit Setpoint
From the overcurrent detection data in the electrical characteristics table:
VTH(MAX) = 101mV,
ICL(MAX)=
=
Current Limit
The current limit setpoint has to be higher than the normal
full load current. Attention has to be paid to the current
rating of the external power components as these are the
first to fail during an overload condition. The MOSFET
continuous and pulsed drain current rating at a given case
temperature has to be accounted for when setting the current limit trip point.
Temperature curves on MOSFET manufacturers’ data
sheets allow the designer to determine the MOSFET drain
current at a particular VGS and TJ (junction temperature).
This, in turn, will assist the designer to set a proper current
limit, without causing device breakdown during an overload condition.
Let’s assume the full CPU load is 16A. The internal
current sense comparator current limit voltage limits are:
77mV < VTH < 101mV. Also, there is a 21% total variation
in RSENSE as discussed in the previous section.
We compute the value of the current sensing element
(embedded PCB trace) for the minimum current limit set-
VTH(MAX)
VTH(MAX)
=
RSENSE(MIN) RSENSE(NOM) × 0.79
101mV
= 38.7A.
3.3mΩ × 0.79
Therefore, the range of load currents that will cause the
internal current sense comparator to detect an overload
condition through a 3.3mΩ embedded PCB trace is:
19.3A < ICL < 38.7A, with 26A being the nominal overload
condition.
Design Rules for Using a Droop Resistor
The basic equation for laying an embedded resistor is:
RAR = ρ ×
L
A
or R = ρ ×
where
A= W × t = cross-sectional area;
ρ= the copper resistivity (µΩ-mil);
L= length (mils);
W = width (mils);
t = thickness (mils).
17
L
(W × t)
,
CS51313
Application Information: continued
For most PCBs the copper thickness, t, is 35µm (1.37 mils)
for one ounce copper; ρ = 717.86µΩ-mil.
For a CPU load of 16A the resistance needed to create a
50mV drop at full load is:
RDROOP =
Thermal Management
Thermal Considerations for Power MOSFETs
In order to maintain good reliability, the junction temperature of the semiconductor components should be kept to a
maximum of 150°C or lower. The thermal impedance
(junction to ambient) required to meet this requirement
can be calculated as follows:
TJ(MAX) - TA
Thermal Impedance =
Power
50mV
50mV
=
= 3.1mΩ.
IOUT
16A
The resistivity of the copper will drift with the temperature
according to the following guidelines:
∆R = 12% @ TA = +50˚C;
∆R = 34% @TA = +100˚C.
A heatsink may be added to TO-220 components to reduce
their thermal impedance. A number of PC board layout
techniques such as thermal vias and additional copper foil
area can be used to improve the power handling capability
of surface mount components.
Droop Resistor Length, Width, and Thickness
The minimum width and thickness of the droop resistor
should primarily be determined on the basis of the currentcarrying capacity required, and the maximum permissible
droop resistor temperature rise. PCB manufacturer design
charts can be used in determining current- carrying capacity and sizes of etched copper conductors for various temperature rises above ambient.
For single conductor applications, such as the use of the
droop resistor, PCB design charts show that for a droop
resistor with a required current-carrying capacity of 16A,
and a 45˚C temperature rise above ambient, the recommended cross section is 275 mil2.
EMI Management
As a consequence of large currents being turned on and off
at high frequency, switching regulators generate noise as a
consequence of their normal operation. When designing
for compliance with EMI/EMC regulations, additional
components may be added to reduce noise emissions.
These components are not required for regulator operation
and experimental results may allow them to be eliminated.
The input filter inductor may not be required because bulk
filter and bypass capacitors, as well as other loads located
on the board will tend to reduce regulator di/dt effects on
the circuit board and input power supply. Placement of the
power component to minimize routing distance will also
help to reduce emissions.
W × t = 275 mil2,
where
W = droop resistor width;
t = droop resistor thickness.
For 1oz. copper, t= 1.37 mils, therefore W = 201 mils =
0.201 in.
R=ρ×
L
,
W×t
Layout Guidelines
where
R = droop resistor value;
ρ = 0.71786mΩ-mil (1 oz. copper);
L = droop resistor length;
W = droop resistor width.
When laying out the CPU buck regulator on a printed circuit board, the following checklist should be used to
ensure proper operation of the CS51313.
1) Rapid changes in voltage across parasitic capacitors and
abrupt changes in current in parasitic inductors are major
concerns for a good layout.
RDROOP = 3.3mΩ.
3.3mΩ = 0.71786mΩ-mil ×
2) Keep high currents out of sensitive ground connections.
L
.
201 mils × 1.37 mils
3) Avoid ground loops as they pick up noise. Use star or
single point grounding.
4) For high power buck regulators on double-sided PCBs a
single ground plane (usually the bottom) is recommended.
Hence, L = 1265 mils = 1.265 in.
In layouts where it is impractical to lay out a droop resistor
in a straight line 1265 mils long, the embedded PCB trace
can be “snaked” to fit within the available space.
5) Even though double sided PCBs are usually sufficient
for a good layout, four-layer PCBs are the optimum
approach to reducing susceptibility to noise. Use the two
internal layers as the power and Gnd planes, the top layer
for power connections and component vias, and the bottom layer for the noise sensitive traces.
18
6) Keep the inductor switching node small by placing the
output inductor, switching and synchronous FETs close
together.
12) Place the COFF and COMP capacitors as close as possible to the COFF and COMP pins.
13) Place the current limit filter capacitor between the VFB
and VOUT pins, as close as possible to the pins.
7) The MOSFET gate traces to the IC must be as short,
straight, and wide as possible.
8) Use fewer, but larger output capacitors, keep the capacitors clustered, and use multiple layer traces with heavy
copper to keep the parasitic resistance low.
14) Connect the filter components of the following pins:
VFB, VOUT, COFF, and COMP to the Gnd pin with a single
trace, and connect this local Gnd trace to the output capacitor Gnd.
9) Place the switching MOSFET as close to the +5V input
capacitors as possible.
15) The “Droop” Resistor (embedded PCB trace) has to be
wide enough to carry the full load current.
10) Place the output capacitors as close to the load
as possible.
16) Place the VCC bypass capacitor as close as possible to
the IC.
11) Place the VFB,VOUT filter resistors (510Ω) in series with
the VFB and VOUT pins as close as possible to the pins.
19
CS51313
Application Information: continued
CS51313
Package Specification
PACKAGE DIMENSIONS IN mm (INCHES)
PACKAGE THERMAL DATA
D
Lead Count
Metric
Max
Min
10.00
9.80
16L SO Narrow
Thermal Data
English
Max Min
.394 .386
RΘJC
RΘJA
typ
typ
16L
SO Narrow
28
115
˚C/W
˚C/W
Surface Mount Narrow Body (D); 150 mil wide
4.00 (.157)
3.80 (.150)
6.20 (.244)
5.80 (.228)
0.51 (.020)
0.33 (.013)
1.27 (.050) BSC
1.75 (.069) MAX
1.57 (.062)
1.37 (.054)
1.27 (.050)
0.40 (.016)
0.25 (.010)
0.19 (.008)
D
0.25 (0.10)
0.10 (.004)
REF: JEDEC MS-012
Ordering Information
Part Number
CS51313GD16
CS51313GDR16
Rev. 3/11/99
Cherry Semiconductor Corporation reserves the right to
make changes to the specifications without notice. Please
contact Cherry Semiconductor Corporation for the latest
available information.
Description
16L SO Narrow
16L SO Narrow (tape & reel)
20
© 1999 Cherry Semiconductor Corporation