LINER LTC3703EGN-5 60v synchronous switching regulator controller Datasheet

LTC3703-5
60V Synchronous
Switching Regulator Controller
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FEATURES
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DESCRIPTIO
The LTC®3703-5 is a synchronous step-down switching
regulator controller that can directly step-down voltages
from up to 60V, making it ideal for telecom and automotive
applications. The LTC3703-5 drives external logic level
N-channel MOSFETs using a constant frequency (up to
600kHz), voltage mode architecture.
High Voltage Operation: Up to 60V
Large 1Ω Gate Drivers (with 5V Supply)
No Current Sense Resistor Required
Step-Up or Step-Down DC/DC Converter
Dual N-Channel MOSFET Synchronous Drive
Excellent Transient Response and DC Line Regulation
Programmable Constant Frequency: 100kHz to
600kHz
±1% Reference Accuracy
Synchronizable up to 600kHz
Selectable Pulse Skip Mode Operation
Low Shutdown Current: 25µA Typ
Programmable Current Limit
Undervoltage Lockout
Programmable Soft-Start
16-Pin Narrow SSOP and 28-Pin SSOP Packages
A precise internal reference provides 1% DC accuracy. A
high bandwidth error amplifier and patented* line feed
forward compensation provide very fast line and load
transient response. Strong 1Ω gate drivers allow the
LTC3703-5 to drive multiple MOSFETs for higher current
applications. The operating frequency is user programmable from 100kHz to 600kHz and can also be synchronized to an external clock for noise-sensitive applications.
Current limit is programmable with an external resistor
and utilizes the voltage drop across the synchronous
MOSFET to eliminate the need for a current sense resistor.
For applications requiring up to 100V operation, refer to
the LTC3703 data sheet.
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APPLICATIO S
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48V Telecom and Base Station Power Supplies
Networking Equipment, Servers
Automotive and Industrial Control
PARAMETER
Maximum VIN
MOSFET Gate Drive
VCC UV+
VCC UV–
, LTC and LT are registered trademarks of Linear Technology Corporation.
*U.S. Patent Numbers: 5408150, 5055767, 6677210, 5847554, 5481178, 6304066, 6580258;
Others Pending.
LTC3703-5
60V
4.5V to 15V
3.7V
3.1V
LTC3703
100V
9.3V to 15V
8.7V
6.2V
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TYPICAL APPLICATIO
High Efficiency High Voltage Step-Down Converter
VCC
5V
Efficiency vs Load Current
22µF
MMDL770T1
VIN
6V TO 60V
MODE/SYNC VIN
+
30k
FSET
COMP
TG
LTC3703-5
FB
SW
INV
0.1µF
100Ω
0.1µF
8µH
12k
IMAX
VCC
10Ω
270µF
16V
DRVCC
+
VOUT
5V
5A
VIN = 42V
90
85
Si7850DP
RUN/SS
BG
D1
MBR1100
10µF
113k
1%
VIN = 24V
95
Si7850DP
470pF
21.5k
1%
VIN = 12V
22µF
×2
BOOST
10k
1000pF
100
EFFICIENCY (%)
+
GND
80
0
BGRTN
1µF
2200pF
1
3
2
LOAD CURRENT (A)
4
5
37053 TA04b
37035 TA04
37035f
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LTC3703-5
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ABSOLUTE
AXI U RATI GS (Note 1)
Supply Voltages
VCC, DRVCC .......................................... –0.3V to 15V
(DRVCC – BGRTN), (BOOST – SW) ...... –0.3V to 15V
BOOST (Continuous) ............................ –0.3V to 85V
BOOST (400ms) ................................... –0.3V to 95V
BGRTN ...................................................... –5V to 0V
VIN Voltage (Continuous) .......................... –0.3V to 70V
VIN Voltage (400ms) ................................. –0.3V to 80V
SW Voltage (Continuous) ............................ –1V to 70V
SW Voltage (400ms) ................................... –1V to 80V
Run/SS Voltage .......................................... –0.3V to 5V
MODE/SYNC, INV Voltages ....................... –0.3V to 15V
fSET, FB, IMAX, COMP Voltages ................... –0.3V to 3V
Driver Outputs
TG ................................ SW – 0.3V to BOOST + 0.3V
BG ........................... BGRTN – 0.3V to DRVCC + 0.3V
Peak Output Current <10µs BG,TG ............................ 5A
Operating Temperature Range (Note 2) .. –40°C to 85°C
Junction Temperature (Notes 3, 7) ....................... 125°C
Storage Temperature Range ................. –65°C to 150°C
Lead Temperature (Soldering, 10 sec.)................. 300°C
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PACKAGE/ORDER I FOR ATIO
ORDER PART
NUMBER
TOP VIEW
MODE/SYNC 1
fSET 2
16 VIN
LTC3703EGN-5
15 B00ST
TOP VIEW
VIN
1
28 BOOST
NC
2
27 TG
NC
3
26 SW
NC
4
25 NC
NC
5
24 NC
COMP 3
14 TG
MODE/SYNC
6
23 NC
FB 4
13 SW
fSET
7
22 NC
12 VCC
COMP
8
21 VCC
FB
9
20 DRVCC
IMAX 5
INV 6
RUN/SS 7
GND 8
11 DRVCC
10 BG
9
GN PART
MARKING
BGRTN
GN PACKAGE
16-LEAD NARROW PLASTIC SSOP
37035
IMAX 10
19 BG
INV 11
18 NC
NC 12
17 NC
RUN/SS 13
16 NC
GND 14
ORDER PART
NUMBER
LTC3703EG-5
15 BGRTN
TJMAX = 125°C, θJA = 110°C/W
G PACKAGE
28-LEAD PLASTIC SSOP
TJMAX = 125°C, θJA = 100°C/W
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VCC = DRVCC = VBOOST = VIN = 5V, VMODE/SYNC = VINV = VSW =
BGRTN = 0V, RUN/SS = IMAX = open, RSET = 25k, unless otherwise specified.
SYMBOL
VCC, DRVCC
VIN
ICC
PARAMETER
VCC, DRVCC Supply Voltage
VIN Pin Voltage
VCC Supply Current
IDRVCC
DRVCC Supply Current
IBOOST
BOOST Supply Current
CONDITIONS
●
MIN
4.1
TYP
●
VFB = 0V
RUN/SS = 0V
(Note 5)
RUN/SS = 0V
(Note 5)
RUN/SS = 0V
●
●
1.7
25
0
0
360
0
MAX
15
60
2.5
40
5
5
500
5
UNITS
V
V
mA
µA
µA
µA
µA
µA
37035f
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LTC3703-5
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VCC = DRVCC = VBOOST = VIN = 5V, VMODE/SYNC = VINV = VSW =
BGRTN = 0V, RUN/SS = IMAX = open, RSET = 25k, unless otherwise specified.
SYMBOL
PARAMETER
Main Control Loop
VFB
Feedback Voltage
CONDITIONS
(Note 4)
●
∆VFB, LINE
∆VFB, LOAD
VMODE/SYNC
∆VMODE/SYNC
IMODE/SYNC
VINV
IINV
IVIN
Feedback Voltage Line Regulation
Feedback Voltage Load Regulation
MODE/SYNC Threshold
MODE/SYNC Hysteresis
MODE/SYNC Current
Invert Threshold
Invert Current
VIN Sense Input Current
IMAX
VOS, IMAX
VRUN/SS
IRUN/SS
IMAX Source Current
VIMAX Offset Voltage
Shutdown Threshold
RUN/SS Source Current
Maximum RUN/SS Sink Current
Undervoltage Lockout
VUV
Oscillator
fOSC
fSYNC
tON, MIN
DCMAX
Driver
IBG, PEAK
RBG, SINK
ITG, PEAK
RTG, SINK
Feedback Amplifier
AVOL
fU
IFB
ICOMP
Oscillator Frequency
External Sync Frequency Range
Minimum On-Time
Maximum Duty Cycle
BG Driver Peak Source Current
BG Driver Pull-Down RDS, ON
TG Driver Peak Source Current
TG Driver Pull-Down RDS, ON
5V < VCC < 15V (Note 4)
1V < VCOMP < 2V (Note 4)
MODE/SYNC Rising
MIN
TYP
MAX
UNITS
0.792
0.788
0.800
0.808
0.812
0.05
0.1
0.87
V
V
%/V
%
V
mV
µA
V
µA
µA
µA
µA
mV
V
µA
µA
V
V
V
●
●
0.75
0 ≤ VMODE/SYNC ≤ 15V
1
0 ≤ VINV ≤ 15V
VIN = 60V
RUN/SS = 0V, VIN = 10V
VIMAX = 0V
|VSW| – VIMAX at IRUN/SS = 0µA
RUN/SS = 0V
|VSW| – VIMAX > 100mV
VCC Rising
VCC Falling
Hysteresis
●
●
●
10.5
– 25
0.7
2.3
9
3.4
2.8
0.45
RSET = 25kΩ
270
100
f < 200kHz
89
0.75
(Note 8)
0.75
(Note 8)
Op Amp DC Open Loop Gain
(Note 4)
Op Amp Unity Gain Crossover Frequency (Note 6)
FB Input Current
0 ≤ VFB ≤ 3V
COMP Sink/Source Current
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: The LTC3703-5 is guaranteed to meet performance specifications
from 0°C to 70°C. Specifications over the –40°C to 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls.
Note 3: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formula:
LTC3703-5: TJ = TA + (PD • 100 °C/W) G Package
Note 4: The LTC3703-5 is tested in a feedback loop that servos VFB to the
reference voltage with the COMP pin forced to a voltage between 1V and 2V.
74
±5
0.007
0.01
0.8
20
0
1.5
0
80
0
12
10
0.9
3.8
17
3.7
3.1
0.65
300
200
93
1
1.2
1
1.2
85
25
0
±10
1
2
1
130
1
13.5
55
1.2
5.3
25
4.1
3.4
0.85
330
600
96
1.8
1.8
1
kHz
kHz
ns
%
A
Ω
A
Ω
dB
MHz
µA
mA
Note 5: The dynamic input supply current is higher due to the power
MOSFET gate charging being delivered at the switching frequency
(QG • fOSC).
Note 6: Guaranteed by design. Not subject to test.
Note 7: This IC includes overtemperature protection that is intended to
protect the device during momentary overload conditions. Junction
temperature will exceed 125°C when overtemperature protection is active.
Continuous operation above the specified maximum operating junction
temperature may impair device reliability.
Note 8: RDS(ON) guaranteed by correlation to wafer level measurement.
37035f
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LTC3703-5
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TYPICAL PERFOR A CE CHARACTERISTICS
Efficiency vs Input Voltage
TA = 25°C (unless otherwise noted).
Load Transient Response
Efficiency vs Load Current
100
100
VIN = 24V
95
IOUT = 1A
EFFICIENCY (%)
IOUT = 5A
90
85
VOUT
50mV/DIV
VIN = 42V
90
IOUT
2A/DIV
85
VOUT = 5V
f = 250kHz
FORCED CONTINUOUS
80
0
10
20
30
40
INPUT VOLTAGE (V)
VOUT = 12V
f = 250kHz
PULSE SKIP ENABLED
80
50
60
0
1
50µs/DIV
VIN = 50V
VOUT = 12V
1A TO 5A LOAD STEP
2
3
LOAD CURRENT (A)
37035 G01
37035 G02
VCC Current vs VCC Voltage
3.5
4
120
COMP = 1.5V
100
3
2.5
2.0
VCC CURRENT (mA)
VCC CURRENT (mA)
VCC Shutdown Current vs VCC
Voltage
VCC Current vs Temperature
VCC RISING
3.0
VFB = 0V
1.5
1.0
COMP = 1.5V
2
VFB = 0V
60
40
20
0
0
2.5
5
10
7.5
VCC VOLTAGE (V)
12.5
0
20 40 60
–60 –40 –20 0
TEMPERATURE (°C)
15
80
0
100
0
VCC Shutdown Current
vs Temperature
4
10 12
6
8
VCC VOLTAGE (V)
14
16
37035 G06
Normalized Frequency
vs Temperature
Reference Voltage
vs Temperature
0.803
VCC = 5V
1.20
1.15
25
20
15
10
0.802
NORMALIZED FREQUENCY
REFERENCE VOLTAGE (V)
30
0.801
0.800
0.799
5
0
20 40 60
–60 –40 –20 0
TEMPERATURE (°C)
2
37035 G05
37035 G04
VCC CURRENT (µA)
80
1
0.5
35
37035 G03
5
4
VCC CURRENT (µA)
EFFICIENCY (%)
95
1.10
1.05
1.00
0.95
0.90
0.85
80
100
37035 G07
0.798
20 40 60
–60 –40 –20 0
TEMPERATURE (°C)
80
100
37035 G08
0.80
–60 –40 –20
0
20 40 60
TEMPERATURE (°C)
80
100
37035 G09
37035f
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LTC3703-5
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TYPICAL PERFOR A CE CHARACTERISTICS
Driver Pull-Down RDS(ON)
vs Temperature
Driver Peak Source Current
vs Temperature
1.2
Driver Peak Source Current
vs Supply Voltage
1.6
VCC = 5V
3.0
VCC = 5V
PEAK SOURCE CURRENT (A)
1.2
1.1
RDS(ON) (Ω)
PEAK SOURCE CURRENT (A)
1.4
1.0
1.0
0.8
0.6
0.4
0.9
0
20 40 60
TEMPERATURE (°C)
80
0
–60 –40 –20 0
20 40 60
TEMPERATURE (°C)
100
2.0
1.5
1.0
0.5
0.2
0.8
–60 –40 –20
2.5
37035 G10
0
80
100
0
5
2.5
7.5
10
12.5
DRVCC/BOOST VOLTAGE (V)
37035 G12
37035 G11
Rise/Fall Time
vs Gate Capacitance
Driver Pull-Down RDS(ON)
vs Supply Voltage
200
1.3
15
RUN/SS Pull-Up Current
vs Temperature
5
VCC = 5V
VCC = 5V
1.2
1.0
0.9
0.8
4
150
RUN/SS CURRENT (µA)
RISE/FALL TIME (ns)
RDS(ON) (Ω)
1.1
RISE TIME
100
50
0.6
2.5
5
7.5
10
12.5
DRVCC/BOOST VOLTAGE (V)
0
15
10
15
5
GATE CAPACITANCE (nF)
0
0
–60 –40 –20
20
RUN/SS Pull-Up Current
vs VCC Voltage
100
IMAX = 0.3V
90
1
80
MAX DUTY CYCLE (%)
RUN/SS SINK CURRENT (µA)
20
2
100
Max % DC vs RUN/SS Voltage
25
5
3
80
15735 G15
RUN/SS Sink Current
vs SW Voltage
4
0 20 40 60
TEMPERATURE (°C)
37035 G14
37035 G13
RUN/SS PULL-UP CURRENT (µA)
2
1
FALL TIME
0.7
3
15
10
5
0
70
60
50
40
30
20
10
–5
0
0
–10
0
2.5
5
7.5
10
VCC VOLTAGE (V)
12.5
15
37035 G16
0
0.1
0.5
0.2 0.3 0.4
|SW| VOLTAGE (V)
0.6
0.7
37035 G17
–10
0.5
1.0
2.0
1.5
RUN VOLTAGE (V)
2.5
3.0
37035 G18
37035f
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LTC3703-5
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TYPICAL PERFOR A CE CHARACTERISTICS
IMAX Current vs Temperature
Max % DC vs Frequency and
Temperature
% Duty Cycle vs COMP Voltage
100
13
100
95
12
MAX DUTY CYCLE (%)
DUTY CYCLE (%)
IMAX SOURCE CURRENT (µA)
VIN = 10V
80
VIN = 50V
60
VIN = 25V
40
80
100
0.5
0.75
1.00
1.25 1.50
COMP (V)
1.75
Shutdown Threshold vs
Temperature
70
0
100
200 300 400 500
FREQUENCY (kHz)
600
700
37035 G21
tON(MIN) vs Temperature
1.4
160
1.2
140
120
1.0
tON(MIN) (ns)
SHUTDOWN THRESHOLD (V)
2.00
37035 G20
37035 G19
0.8
0.6
0.4
100
80
60
40
0.2
0
–60 –40 –20
25°C
80
75
0
0
20 40 60
TEMPERATURE (°C)
–45°C
85
90°C
20
11
–60 –40 –20
90
20
0
20 40 60
TEMPERATURE (°C)
80
100
37035 G22
0
–60 –40 –20
0
20 40 60
TEMPERATURE (°C)
80
100
37035 G23
37035f
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LTC3703-5
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PI FU CTIO S
(GN16/G28)
MODE/SYNC (Pin 1/Pin 6): Pulse Skip Mode Enable/Sync
Pin. This multifunction pin provides Pulse Skip Mode enable/disable control and an external clock input for synchronization of the internal oscillator. Pulling this pin below 0.8V
or to an external logic-level synchronization signal disables
Pulse Skip Mode operation and forces continuous operation. Pulling the pin above 0.8V enables Pulse Skip Mode
operation. This pin can also be connected to a feedback
resistor divider from a secondary winding on the inductor
to regulate a second output voltage.
fSET (Pin 2/Pin 7): Frequency Set. A resistor connected to
this pin sets the free running frequency of the internal oscillator. See applications section for resistor value selection details.
COMP (Pin 3/Pin 8): Loop Compensation. This pin is connected directly to the output of the internal error amplifier.
An RC network is used at the COMP pin to compensate the
feedback loop for optimal transient response.
FB (Pin 4/Pin 9): Feedback Input. Connect FB through a
resistor divider network to VOUT to set the output voltage.
Also connect the loop compensation network from COMP
to FB.
IMAX (Pin 5/Pin 10): Current Limit Set. The IMAX pin sets
the current limit comparator threshold. If the voltage drop
across the bottom MOSFET exceeds the magnitude of the
voltage at IMAX, the controller goes into current limit. The
IMAX pin has an internal 12µA current source, allowing the
current threshold to be set with a single external resistor
to ground. See the Current Limit Programming section for
more information on choosing RIMAX.
INV (Pin 6/Pin 11): Top/Bottom Gate Invert. Pulling this pin
above 2V sets the controller to operate in step-up (boost)
mode with the TG output driving the synchronous MOSFET
and the BG output driving the main switch. Below 1V, the
controller will operate in step-down (buck) mode.
RUN/SS (Pin 7/Pin 13): Run/Soft-Start. Pulling RUN/SS below 0.9V will shut down the LTC3703-5, turn off both of the
external MOSFET switches and reduce the quiescent supply current to 25µA. A capacitor from RUN/SS to ground
will control the turn-on time and rate of rise of the output
voltage at power-up. An internal 4µA current source pullup at the RUN/SS pin sets the turn-on time at approximately
750ms/µF.
GND (Pin 8/Pin 14): Ground Pin.
BGRTN (Pin 9/Pin 15): Bottom Gate Return. This pin connects to the source of the pull-down MOSFET in the BG
driver and is normally connected to ground. Connecting a
negative supply to this pin allows the synchronous
MOSFET’s gate to be pulled below ground to help prevent
false turn-on during high dV/dt transitions on the SW node.
See the Applications Information section for more details.
BG (Pin 10/Pin 19): Bottom Gate Drive. The BG pin drives
the gate of the bottom N-channel synchronous switch
MOSFET. This pin swings from BGRTN to DRVCC.
DRVCC (Pin 11/Pin 20): Driver Power Supply Pin. DRVCC
provides power to the BG output driver. This pin should be
connected to a voltage high enough to fully turn on the
external MOSFETs, normally 4.5V to 15V for logic level
threshold MOSFETs. DRVCC should be bypassed to BGRTN
with a 10µF, low ESR (X5R or better) ceramic capacitor.
VCC (Pin 12/Pin 21) : Main Supply Pin. All internal circuits
except the output drivers are powered from this pin. VCC
should be connected to a low noise power supply voltage
between 4.5V and 15V and should be bypassed to GND
(Pin␣ 8) with at least a 0.1µF capacitor in close proximity to
the LTC3703-5.
SW (Pin 13/Pin 26): Switch Node Connection to Inductor
and Bootstrap Capacitor. Voltage swing at this pin is from
a Schottky diode (external) voltage drop below ground to
VIN.
TG (Pin 14/Pin 27): Top Gate Drive. The TG pin drives the
gate of the top N-channel synchronous switch MOSFET. The
TG driver draws power from the BOOST pin and returns to
the SW pin, providing true floating drive to the top MOSFET.
BOOST (Pin 15/Pin 28): Top Gate Driver Supply. The BOOST
pin supplies power to the floating TG driver. The BOOST pin
should be bypassed to SW with a low ESR (X5R or better)
0.1µF ceramic capacitor. An additional fast recovery Schottky diode from DRVCC to BOOST will create a complete floating charge-pumped supply at BOOST.
VIN (Pin 16/Pin 1): Input Voltage Sense Pin. This pin is connected to the high voltage input of the regulator and is used
by the internal feedforward compensation circuitry to improve line regulation. This is not a supply pin.
37035f
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LTC3703-5
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FU CTIO AL DIAGRA
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RSET
FSET
2
OVERCURRENT
12µA
4µA
IMAX
–
5
RMAX
+
50mV
–
±
+
RUN/SS
±
–
5
CSS
1V
CHIP
SD
+
3.2V
INV
UVSD OTSD
SYNC
DETECT
MODE/SYNC 1
EXT SYNC
VCC
+
OSC
DB
–
FORCED CONTINUOUS
INV
REVERSE
CURRENT
15
COMP
14
3
0.8V
FB
R2
R1
4
+
+ FB
–
÷
% DC
LIMIT
–
PWM
+
13
DRIVE
LOGIC
11
VIN 16
+MIN–
10
+MAX–
9
VCC
(<15V)
0.76V
VIN
BOOST
TG
CB
M1
SW
DRVCC
BG
M2
BGRTN
6 INV
0.84V
12
L1
OVER
TEMP
BANDGAP
VCC
VOUT
8 GND
UVLO
COUT
OT SD
0.8V
REFERENCE
INTERNAL
3.2V VCC
UV SD
GN16
VCC
CVCC
37035 FD
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OPERATIO
(Refer to Functional Diagram)
The LTC3703-5 is a constant frequency, voltage mode
controller for DC/DC step-down converters. It is designed
to be used in a synchronous switching architecture with
two external N-channel MOSFETs. Its high operating voltage capability allows it to directly step down input voltages
up to 60V without the need for a step-down transformer.
For circuit operation, please refer to the Functional
Diagram of the IC and the circuit on the first page of this
data sheet. The LTC3703-5 uses voltage mode control in
which the duty ratio is controlled directly by the error
amplifier output and thus requires no current sense resistor. The VFB pin receives the output voltage feedback and
is compared to the internal 0.8V reference by the error
amplifier, which outputs an error signal at the COMP pin.
37035f
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LTC3703-5
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OPERATIO
(Refer to Functional Diagram)
When the load current increases, it causes a drop in the
feedback voltage relative to the reference. The COMP voltage then rises, increasing the duty ratio until the output
feedback voltage again matches the reference voltage. In
normal operation, the top MOSFET is turned on when the
RS latch is set by the on-chip oscillator and is turned off
when the PWM comparator trips and resets the latch. The
PWM comparator trips at the proper duty ratio by comparing the error amplifier output (after being “compensated”
by the line feedforward multiplier) to a sawtooth waveform
generated by the oscillator. When the top MOSFET is turned
off, the bottom MOSFET is turned on until the next cycle
begins or, if Pulse Skip Mode operation is enabled, until
the inductor current reverses as determined by the reverse
current comparator. MAX and MIN comparators ensure
that the output never exceed ±5% of nominal value by
monitoring VFB and forcing the output back into regulation
quickly by either keeping the top MOSFET off or forcing
maximum duty cycle. The operation of its other features—
fast transient response, outstanding line regulation, strong
gate drivers, short-circuit protection, and shutdown/
soft-start—are described below.
Fast Transient Response
The LTC3703-5 uses a fast 25MHz op amp as an error amplifier. This allows the compensation network to be optimized for better load transient response. The high
bandwidth of the amplifier, along with high switching frequencies and low value inductors, allow very high loop
crossover frequencies. The 800mV internal reference allows
regulated output voltages as low as 800mV without external level shifting amplifiers.
Line Feedforward Compensation
The LTC3703-5 achieves outstanding line transient response using a patented feedforward correction scheme.
With this circuit the duty cycle is adjusted instantaneously
to changes in input voltage, thereby avoiding unacceptable overshoot or undershoot. It has the added advantage
of making the DC loop gain independent of input voltage.
Figure 1 shows how large transient steps at the input have
little effect on the output voltage.
VOUT
50mV/DIV
VIN
20V/DIV
IL
2A/DIV
20µs/DIV
VOUT = 12V
ILOAD = 1A
25V TO 60V VIN STEP
37035 F01
Figure 1. Line Transient Performance
Strong Gate Drivers
The LTC3703-5 contains very low impedance drivers
capable of supplying amps of current to slew large MOSFET
gates quickly. This minimizes transition losses and allows
paralleling MOSFETs for higher current applications. A
60V floating high side driver drives the top side MOSFET
and a low side driver drives the bottom side MOSFET (see
Figure 2). They can be powered from either a separate DC
supply or a voltage derived from the input or output
voltage (see MOSFET Driver Supplies section). The bottom side driver is supplied directly from the DRVCC pin.
The top MOSFET drivers are biased from floating bootstrap capacitor CB, which normally is recharged during
each off cycle through an external diode from DRVCC when
the top MOSFET turns off. In Pulse Skip Mode operation,
where it is possible that the bottom MOSFET will be off for
an extended period of time, an internal counter guarantees
that the bottom MOSFET is turned on at least once every
10 cycles for 10% of the period to refresh the bootstrap
capacitor. An undervoltage lockout keeps the LTC3703-5
shut down unless this voltage is above 4.1V.
The bottom driver has an additional feature that helps
minimize the possibility of external MOSFET shoot-thru.
When the top MOSFET turns on, the switch node dV/dt
pulls up the bottom MOSFET’s internal gate through the
Miller capacitance, even when the bottom driver is holding
the gate terminal at ground. If the gate is pulled up high
enough, shoot-thru between the top side and bottom side
37035f
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MOSFETs can occur. To prevent this from occuring, the
bottom driver return is brought out as a separate pin
(BGRTN) so that a negative supply can be used to reduce
the effect of the Miller pull-up. For example, if a –2V supply
is used on BGRTN, the switch node dV/dt could pull the
gate up 2V before the VGS of the bottom MOSFET has more
than 0V across it.
VIN
DRVCC
LTC3703-5
DRVCC
+
DB
CIN
BOOST
TG
cycle control set to 0%. As CSS continues to charge, the
duty cycle is gradually increased, allowing the output
voltage to rise. This soft-start scheme smoothly ramps the
output voltage to its regulated value, with no overshoot.
The RUN/SS voltage will continue ramping until it reaches
an internal 4V clamp. Then the MIN feedback comparator
is enabled and the LTC3703-5 is in full operation. When the
RUN/SS is low, the supply current is reduced to 25µA.
VOUT
CB
M1
SW
0V
SHUTDOWN START-UP
L
VOUT
BG
M2
OUTPUT VOLTAGE
IN REGULATION
COUT
BGRTN
3V
RUN/SS SOFT-STARTS
OUTPUT VOLTAGE AND
INDUCTOR CURRENT
VRUN/SS
0V TO –5V
37035 F02
1.4V
1V
Figure 2. Floating TG Driver Supply and Negative BG Return
MINIMUM
DUTY CYCLE
0V
Constant Frequency
The internal oscillator can be programmed with an external resistor connected from fSET to ground to run between
100kHz and 600kHz, thereby optimizing component size,
efficiency, and noise for the specific application. The
internal oscillator can also be synchronized to an external
clock applied to the MODE/SYNC pin and can lock to a
frequency in the 100kHz to 600kHz range. When locked to
an external clock, Pulse Skip Mode operation is automatically disabled. Constant frequency operation brings with it
a number of benefits: Inductor and capacitor values can be
chosen for a precise operating frequency and the feedback
loop can be similarly tightly specified. Noise generated by
the circuit will always be at known frequencies.
Subharmonic oscillation and slope compensation, common headaches with constant frequency current mode
switchers, are absent in voltage mode designs like the
LTC3703-5.
Shutdown/Soft-Start
The main control loop is shut down by pulling RUN/SS pin
low. Releasing RUN/SS allows an internal 4µA current
source to charge the soft-start capacitor CSS. When CSS
reaches 1V, the main control loop is enabled with the duty
CURRENT
LIMIT
MIN COMPARATOR ENABLED
4V
+
NORMAL OPERATION
LTC3703-5 POWER
ENABLE DOWN MODE
37035 F03
Figure 3. Soft-Start Operation in Start Up and Current Limit
Current Limit
The LTC3703-5 includes an onboard current limit circuit
that limits the maximum output current to a user-programmed level. It works by sensing the voltage drop across
the bottom MOSFET and comparing that voltage to a userprogrammed voltage at the IMAX pin. Since the bottom
MOSFET looks like a low value resistor during its on-time,
the voltage drop across it is proportional to the current
flowing in it. In a buck converter, the average current in the
inductor is equal to the output current. This current also
flows through the bottom MOSFET during its on-time.
Thus by watching the drain-to-source voltage when the
bottom MOSFET is on, the LTC3703-5 can monitor the
output current. The LTC3703-5 senses this voltage and
inverts it to allow it to compare the sensed voltage (which
becomes more negative as peak current increases) with a
positive voltage at the IMAX pin. The IMAX pin includes a
12µA pull-up, enabling the user to set the voltage at IMAX
with a single resistor (RIMAX) to ground. See the Current
Limit Programming section for RIMAX selection.
37035f
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(Refer to Functional Diagram)
For maximum protection, the LTC3703-5 current limit
consists of a steady-state limit circuit and an instantaneous limit circuit. The steady-state limit circuit is a gm
amplifier that pulls a current from the RUN/SS pin proportional to the difference between the SW and IMAX voltages.
This current begins to discharge the capacitor at RUN/SS,
reducing the duty cycle and controlling the output voltage
until the current regulates at the limit. Depending on the
size of the capacitor, it may take many cycles to discharge
the RUN/SS voltage enough to properly regulate the
output current. This is where the instantaneous limit
circuit comes into play. The instantaneous limit circuit is
a cycle-by-cycle comparator which monitors the bottom
MOSFET’s drain voltage and keeps the top MOSFET from
turning on whenever the drain voltage is 50mV above the
programmed max drain voltage. Thus the cycle-by-cycle
comparator will keep the inductor current under control
until the gm amplifier gains control.
skip cycles to maintain regulation. The frequency drops
but this further improves efficiency by minimizing gate
charge losses. In forced continuous mode, the bottom
MOSFET is always on when the top MOSFET is off,
allowing the inductor current to reverse at low currents.
This mode is less efficient due to resistive losses, but has
the advantage of better transient response at low currents,
constant frequency operation, and the ability to maintain
regulation when sinking current. See Figure 4 for a comparison of the effect on efficiency at light loads for each
mode. The MODE/SYNC threshold is 0.8V ±7.5%, allowing the MODE/SYNC to act as a feedback pin for regulating
a second winding. If the feedback voltage drops below
0.8V, the LTC3703-5 reverts to continuous operation to
maintain regulation in the secondary supply.
100
90
The LTC3703-5 can operate in one of two modes selectable with the MODE/SYNC pin—Pulse Skip Mode or
forced continuous mode. Pulse Skip Mode is selected
when increased efficiency at light loads is desired. In this
mode, the bottom MOSFET is turned off when inductor
current reverses to minimize the efficiency loss due to
reverse current flow. As the load is decreased (see Figure␣ 5), the duty cycle is reduced to maintain regulation
until its minimum on-time (~200ns) is reached. When the
load decreases below this point, the LTC3703-5 begins to
PULSE SKIP MODE
EFFICIENCY (%)
80
Pulse Skip Mode
VIN = 12V
VIN = 42V
70
60
50
VIN = 12V
40
VIN = 42V
30
20
VOUT = 5V
FORCED CONTINUOUS
PULSE SKIP MODE
10
0
10
100
1000
LOAD (mA)
10000
37035 F04
Figure 4. Efficiency in Pulse Skip/Forced Continuous Modes
FORCED CONTINUOUS
DECREASING
LOAD
CURRENT
37035 F05
Figure 5. Comparison of Inductor Current Waveforms for Pulse Skip Mode and Forced Continuous Operation
37035f
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Buck or Boost Mode Operation
The LTC3703-5 has the capability of operating both as a
step-down (buck) and step-up (boost) controller. In boost
mode, output voltages as high as 60V can be tightly
regulated. With the INV pin grounded, the LTC3703-5
operates in buck mode with TG driving the main (top side)
switch and BG driving the synchronous (bottom side)
switch. If the INV pin is pulled above 2V, the LTC3703-5
operates in boost mode with BG driving the main (bottom
side) switch and TG driving the synchronous (top side)
switch. Internal circuit operation is very similar regardless
of the operating mode with the following exceptions: In
boost mode, Pulse Skip Mode operation is always disabled regardless of the level of the MODE/SYNC pin and
the line feedforward compensation is also disabled. The
overcurrent circuitry continues to monitor the load current
by looking at the drain voltage of the main (bottom side)
MOSFET. In boost mode, however, the peak MOSFET
current does not equal the load current but instead
ID = ILOAD/(1 – D). This factor needs to be taken into
account when programming the IMAX voltage.
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The basic LTC3703-5 application circuit is shown on the first
page of this data sheet. External component selection is determined by the input voltage and load requirements as
explained in the following sections. After the operating
frequency is selected, RSET and L can be chosen. The
operating frequency and the inductor are chosen for a
desired amount of ripple current and also to optimize efficiency and component size. Next, the power MOSFETs and
D1 are selected based on voltage, load and efficiency requirements. CIN is selected for its ability to handle the large
RMS currents in the converter and COUT is chosen with low
enough ESR to meet the output voltage ripple and transient
specifications. Finally, the loop compensation components
are chosen to meet the desired transient specifications.
noise-sensitive communications systems, it is often desirable to keep the switching noise out of a sensitive
frequency band.
The LTC3703-5 uses a constant frequency architecture
that can be programmed over a 100kHz to 600kHz range
with a single resistor from the fSET pin to ground, as shown
in the circuit on the first page of this data sheet. The
nominal voltage on the fSET pin is 1.2V, and the current that
flows from this pin is used to charge and discharge an
internal oscillator capacitor. The value of RSET for a given
operating frequency can be chosen from Figure 6 or from
the following equation:
RSET (kΩ) =
7100
f(kHz ) – 25
Operating Frequency
1000
100
RSET (kΩ)
The choice of operating frequency and inductor value is a
trade off between efficiency and component size. Low
frequency operation improves efficiency by reducing
MOSFET switching losses and gate charge losses. However, lower frequency operation requires more inductance for a given amount of ripple current, resulting in a
larger inductor size and higher cost. If the ripple current
is allowed to increase, larger output capacitors may be
required to maintain the same output ripple. For converters with high step-down VIN to VOUT ratios, another
consideration is the minimum on-time of the LTC3703-5
(see the Minimum On-time Considerations section). A
final consideration for operating frequency is that in
10
1
0
200
400
600
FREQUENCY (kHz)
800
1000
37035 F06
Figure 6. Timing Resistor (RSET) Value
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The oscillator can also be synchronized to an external
clock applied to the MODE/SYNC pin with a frequency in
the range of 100kHz to 600kHz (refer to the MODE/SYNC
Pin section for more details). In this synchronized mode,
Pulse Skip Mode operation is disabled. The clock high
level must exceed 2V for at least 25ns. As shown in
Figure␣ 7, the top MOSFET turn-on will follow the rising
edge of the external clock by a constant delay equal to onetenth of the cycle period.
2V TO 10V
MODE/
SYNC
tMIN = 25ns
0.8T
TG
T
T = 1/fO
D = 40%
ripple current occurs at the highest VIN. To guarantee that
ripple current does not exceed a specified maximum, the
inductor in buck mode should be chosen according to:
L≥

VOUT 
V
1 – OUT 

f ∆IL(MAX)  VIN(MAX) 
The inductor also has an affect on low current operation
when Pulse Skip Mode operation is enabled. The frequency begins to decrease when the output current drops
below the average inductor current at which the LTC3703-5
is operating at its tON(MIN) in discontinuous mode (see
Figure 5). Lower inductance increases the peak inductor
current that occurs in each minimum on-time pulse and
thus increases the output current at which the frequency
starts decreasing.
0.1T
Power MOSFET Selection
IL
37035 F07
Figure 7. MODE/SYNC Clock Input and Switching
Waveforms for Synchronous Operation
Inductor
The inductor in a typical LTC3703-5 circuit is chosen for
a specific ripple current and saturation current. Given an
input voltage range and an output voltage, the inductor
value and operating frequency directly determine the
ripple current. The inductor ripple current in the buck
mode is:
∆IL =
VOUT  VOUT 
1–

VIN 
(f)(L) 
Lower ripple current reduces core losses in the inductor,
ESR losses in the output capacitors and output voltage
ripple. Thus highest efficiency operation is obtained at low
frequency with small ripple current. To achieve this however, requires a large inductor.
A reasonable starting point is to choose a ripple current
between 20% and 40% of IO(MAX). Note that the largest
The LTC3703-5 requires at least two external N-channel
power MOSFETs, one for the top (main) switch and one or
more for the bottom (synchronous) switch. The number,
type and “on” resistance of all MOSFETs selected take into
account the voltage step-down ratio as well as the actual
position (main or synchronous) in which the MOSFET will
be used. A much smaller and much lower input capacitance MOSFET should be used for the top MOSFET in
applications that have an output voltage that is less than
1/3 of the input voltage. In applications where VIN >> VOUT,
the top MOSFETs’ “on” resistance is normally less important for overall efficiency than its input capacitance at
operating frequencies above 300kHz. MOSFET manufacturers have designed special purpose devices that provide
reasonably low “on” resistance with significantly reduced
input capacitance for the main switch application in switching regulators.
Selection criteria for the power MOSFETs include the “on”
resistance RDS(ON), input capacitance, breakdown voltage
and maximum output current.
The most important parameter in high voltage applications is breakdown voltage BVDSS. Both the top and
bottom MOSFETs will see full input voltage plus any
additional ringing on the switch node across its drain-tosource during its off-time and must be chosen with the
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appropriate breakdown specification. Since most MOSFETs
in the 30V to 60V range have logic level thresholds
(VGS(MIN) ≥ 4.5V), the LTC3703-5 is designed to be used
with a 4.5V to 15V gate drive supply (DRVCC pin).
For maximum efficiency, on-resistance RDS(ON) and input
capacitance should be minimized. Low RDS(ON) minimizes
conduction losses and low input capacitance minimizes
transition losses. MOSFET input capacitance is a combination of several components but can be taken from the
typical “gate charge” curve included on most data sheets
(Figure 8).
VIN
MILLER EFFECT
V
VGS
a
b
QIN
CMILLER = (QB – QA)/VDS
+
VGS
+V
DS
–
–
MainSwitchDutyCycle =
VOUT
VIN
SynchronousSwitchDutyCycle =
VIN – VOUT
VIN
The power dissipation for the main and synchronous
MOSFETs at maximum output current are given by:
VOUT
2
IMAX ) (1 + δ)RDR(ON) +
(
VIN
I
VIN2 MAX (RDR )(CMILLER ) •
2

1
1 
+

( f)
 VCC – VTH(IL) VTH(IL) 
V –V
PSYNC = IN OUT (IMAX )2 (1 + δ)RDS(0N)
VIN
PMAIN =
37035 F08
Figure 8. Gate Charge Characteristic
The curve is generated by forcing a constant input current
into the gate of a common source, current source loaded
stage and then plotting the gate voltage versus time. The
initial slope is the effect of the gate-to-source and the gateto-drain capacitance. The flat portion of the curve is the
result of the Miller multiplication effect of the drain-to-gate
capacitance as the drain drops the voltage across the
current source load. The upper sloping line is due to the
drain-to-gate accumulation capacitance and the gate-tosource capacitance. The Miller charge (the increase in
coulombs on the horizontal axis from a to b while the curve
is flat) is specified for a given VDS drain voltage, but can be
adjusted for different VDS voltages by multiplying by the
ratio of the application VDS to the curve specified VDS
values. A way to estimate the CMILLER term is to take the
change in gate charge from points a and b on a manufacturers data sheet and divide by the stated VDS voltage
specified. CMILLER is the most important selection criteria
for determining the transition loss term in the top MOSFET
but is not directly specified on MOSFET data sheets. CRSS
and COS are specified sometimes but definitions of these
parameters are not included.
When the controller is operating in continuous mode the
duty cycles for the top and bottom MOSFETs are given by:
where δ is the temperature dependency of RDS(ON), RDR is
the effective top driver resistance (approximately 2Ω at
VGS = VMILLER), VIN is the drain potential and the change
in drain potential in the particular application. VTH(IL) is the
data sheet specified typical gate threshold voltage specified in the power MOSFET data sheet at the specified drain
current. CMILLER is the calculated capacitance using the
gate charge curve from the MOSFET data sheet and the
technique described above.
Both MOSFETs have I2R losses while the topside N-channel
equation includes an additional term for transition losses,
which peak at the highest input voltage. For VIN < 25V, the
high current efficiency generally improves with larger
MOSFETs, while for VIN > 25V, the transition losses
rapidly increase to the point that the use of a higher
RDS(ON) device with lower CMILLER actually provides higher
efficiency. The synchronous MOSFET losses are greatest
at high input voltage when the top switch duty factor is low
or during a short circuit when the synchronous switch is
on close to 100% of the period.
The term (1 + δ) is generally given for a MOSFET in the
form of a normalized RDS(ON) vs temperature curve, and
typically varies from 0.005/°C to 0.01/°C depending on
the particular MOSFET used.
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Multiple MOSFETs can be used in parallel to lower RDS(ON)
and meet the current and thermal requirements if desired.
The LTC3703-5 contains large low impedance drivers
capable of driving large gate capacitances without significantly slowing transition times. In fact, when driving
MOSFETs with very low gate charge, it is sometimes
helpful to slow down the drivers by adding small gate
resistors (10Ω or less) to reduce noise and EMI caused by
the fast transitions.
Schottky Diode Selection
The Schottky diode D1 shown in the circuit on the first
page of this data sheet. conducts during the dead time
between the conduction of the power MOSFETs. This
prevents the body diode of the bottom MOSFET from
turning on and storing charge during the dead time and
requiring a reverse recovery period that could cost as
much as 1% to 2% in efficiency. A 1A Schottky diode is
generally a good size for 3A to 5A regulators. Larger
diodes result in additional losses due to their larger
junction capacitance. The diode can be omitted if the
efficiency loss can be tolerated.
Input Capacitor Selection
In continuous mode, the drain current of the top MOSFET
is approximately a square wave of duty cycle VOUT/VIN
which must be supplied by the input capacitor. To prevent
large input transients, a low ESR input capacitor sized for
the maximum RMS current is given by:
ICIN(RMS) ≅ IO(MAX)

VOUT  VIN
– 1

VIN  VOUT 
1/ 2
This formula has a maximum at VIN = 2VOUT, where IRMS
= IO(MAX)/2. This simple worst-case condition is commonly
used for design because even significant deviations do not
offer much relief. Note that the ripple current ratings from
capacitor manufacturers are often based on only 2000 hours
of life. This makes it advisable to further derate the capacitor or to choose a capacitor rated at a higher temperature
than required. Several capacitors may also be placed in
parallel to meet size or height requirements in the design.
Because tantalum and OS-CON capacitors are not available in voltages above 30V, for regulators with input sup-
plies above 30V, choice of input capacitor type is limited
to ceramics or aluminum electrolytics. Ceramic capacitors
have the advantage of very low ESR and can handle high
RMS current, however ceramics with high voltage ratings
(>50V) are not available with more than a few microfarads
of capacitance. Furthermore, ceramics have high voltage
coefficients which means that the capacitance values decrease even more when used at the rated voltage. X5R and
X7R type ceramics are recommended for their lower voltage and temperature coefficients. Another consideration
when using ceramics is their high Q which if not properly
damped, may result in excessive voltage stress on the
power MOSFETs. Aluminum electrolytics have much higher
bulk capacitance, however, they have higher ESR and lower
RMS current ratings.
A good approach is to use a combination of aluminum
electrolytics for bulk capacitance and ceramics for low
ESR and RMS current. If the RMS current cannot be
handled by the aluminum capacitors alone, when used
together, the percentage of RMS current that will be
supplied by the aluminum capacitor is reduced to
approximately:
% IRMS,ALUM ≈
1
1 + (8fCRESR )2
• 100%
where RESR is the ESR of the aluminum capacitor and C is
the overall capacitance of the ceramic capacitors. Using an
aluminum electrolytic with a ceramic also helps damp the
high Q of the ceramic, minimizing ringing.
Output Capacitor Selection
The selection of COUT is primarily determined by the ESR
required to minimize voltage ripple. The output ripple
(∆VOUT) is approximately equal to:

1 
∆VOUT ≤ ∆IL  ESR +


8fC OUT 
Since ∆IL increases with input voltage, the output ripple is
highest at maximum input voltage. ESR also has a significant effect on the load transient response. Fast load
transitions at the output will appear as voltage across the
ESR of COUT until the feedback loop in the LTC3703-5 can
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change the inductor current to match the new load current
value. Typically, once the ESR requirement is satisfied the
capacitance is adequate for filtering and has the required
RMS current rating.
Manufacturers such as Nichicon, United Chemicon and
Sanyo should be considered for high performance
throughhole capacitors. The OS-CON (organic semiconductor dielectric) capacitor available from Sanyo has the
lowest product of ESR and size of any aluminum electrolytic at a somewhat higher price. An additional ceramic
capacitor in parallel with OS-CON capacitors is recommended to reduce the effect of their lead inductance.
In surface mount applications, multiple capacitors placed
in parallel may be required to meet the ESR, RMS current
handling and load step requirements. Dry tantalum, special polymer and aluminum electrolytic capacitors are
available in surface mount packages. Special polymer
capacitors offer very low ESR but have lower capacitance
density than other types. Tantalum capacitors have the
highest capacitance density but it is important to only use
types that have been surge tested for use in switching
power supplies. Several excellent surge-tested choices
are the AVX TPS and TPSV or the KEMET T510 series.
Aluminum electrolytic capacitors have significantly higher
ESR, but can be used in cost-driven applications providing
that consideration is given to ripple current ratings and
long term reliability. Other capacitor types include Panasonic SP and Sanyo POSCAPs.
Output Voltage
The LTC3703-5 output voltage is set by a resistor divider
according to the following formula:
 R1
VOUT = 0.8V 1 + 
 R2
The external resistor divider is connected to the output as
shown in the Functional Diagram, allowing remote voltage
sensing. The resultant feedback signal is compared with
the internal precision 800mV voltage reference by the
error amplifier. The internal reference has a guaranteed
tolerance of ±1%. Tolerance of the feedback resistors will
add additional error to the output voltage. 0.1% to 1%
resistors are recommended.
MOSFET Driver Supplies (DRVCC and BOOST)
The LTC3703-5 drivers are supplied from the DRVCC and
BOOST pins (see Figure 2), which have an absolute
maximum voltage of 15V. If the main supply voltage, VIN,
is higher than 15V a separate supply with a voltage
between 5V and 15V must be used to power the drivers. If
a separate supply is not available, one can easily be
generated from the main supply using one of the circuits
shown in Figure␣ 9. If the output voltage is between 5V and
15V, the output can be used to directly power the drivers
as shown in Figure 9a. If the output is below 5V, Figure 9b
shows an easy way to boost the supply voltage to a
sufficient level. This boost circuit uses the LT1613 in a
ThinSOTTM package and a chip inductor for minimal extra
area (<0.2 in2). Two other possible schemes are an extra
winding on the inductor (Figure 9c) or a capacitive charge
pump (Figure 9d). All the circuits shown in Figure␣ 9
require a start-up circuit (Q1, D1 and R1) to provide driver
power at initial start-up or following a short-circuit. The
resistor R1 must be sized so that it supplies sufficient base
current and zener bias current at the lowest expected value
of VIN. When using an existing supply, the supply must be
capable of supplying the required gate driver current
which can be estimated from:
IDRVCC = (f)(QG(TOP) + QG(BOTTOM))
This equation for IDRVCC is also useful for properly sizing
the circuit components shown in Figure 9.
An external bootstrap capacitor, CB, connected to the
BOOST pin supplies the gate drive voltage for the topside
MOSFETs. Capacitor CB is charged through external
diode, DB, from the DRVCC supply when SW is low. When
the top side MOSFET is turned on, the driver places the C B
voltage across the gate-source of the top MOSFET. The
switch node voltage, SW, rises to VIN and the BOOST pin
follows. With the topside MOSFET on, the boost voltage
is above the input supply: VBOOST = VIN + VDRVCC. The
value of the boost capacitor CB needs to be 100 times that
of the total input capacitance of the top side MOSFET(s).
The reverse breakdown of the external diode, DB, must be
greater than VIN(MAX). Another important consideration
for the external diode is the reverse recovery and reverse
leakage, either of which may cause excessive reverse
ThinSOT is a tradmark of Linear Technology Corporation.
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D2
ZHCS400
VIN
VIN
+
C10
1µF
10V
1µF
+
CIN
5.1V
+
CIN
VIN
VIN
LTC3703-5
TG
SW
DRVCC
BG
TG
L1
VOUT
5V TO
15V
VCC
SW
COUT
DRVCC
BG
VOUT
<5V
L1
+
+
BGRTN
COUT
BGRTN
37035 F09a
3703 F09b
Figure 9a. VCC Generated from 5V < VOUT < 15V
Figure 9b. VCC Generated from VOUT < 5V
VIN (<40V)
VIN
+
+
1µF
CIN
+
OPTIONAL VCC
CONNECTION
5V < VSEC < 15V
CIN
5.1V
BAT85
5.1V
VIN
TG1
TG
T1
DRVCC
SW
FCB
BG1
GND
BGRTN
R1
0.22µF
VN2222LL
LTC3703-5
1µF
N
BAT85
VIN
VSEC
+
LTC3703-5
VCC
C9
4.7µF
6.3V
R17
37.4k
VIN
SW
1%
LT1613
SHDN
FB
R17
GND
12.1k
1%
LTC3703-5
5.1V
VCC
L2
4.7µH
BAT85
VOUT
1
VOUT
VCC
SW
DRVCC
BG
L1
+
COUT
+
COUT
R2
BGRTN
3703 F09c
Figure 9c. Secondary Output Loop and VCC Connection
current to flow at full reverse voltage. If the reverse
current times reverse voltage exceeds the maximum
allowable power dissipation, the diode may be damaged.
For best results, use an ultrafast recovery diode such as
the MMDL770T1.
An internal undervoltage lockout (UVLO) monitors the
voltage on DRVCC to ensure that the LTC3703-5 has
sufficient gate drive voltage. If the DRVCC voltage falls
3703 F09d
Figure 9d. Capacitive Charge Pump for VCC (VIN < 40V)
below the UVLO threshold, the LTC3703-5 shuts down
and the gate drive outputs remain low.
Bottom MOSFET Source Supply (BGRTN)
The bottom gate driver, BG, switches from DRVCC to BGRTN
where BGRTN can be a voltage between ground and –5V.
Why not just keep it simple and always connect BGRTN to
ground? In high voltage switching converters, the switch
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node dV/dt can be many volts/ns, which will pull up on the
gate of the bottom MOSFET through its Miller capacitance.
If this Miller current, times the internal gate resistance of
the MOSFET plus the driver resistance, exceeds the threshold of the FET, shoot-through will occur. By using a negative supply on BGRTN, the BG can be pulled below ground
when turning the bottom MOSFET off. This provides a few
extra volts of margin before the gate reaches the turn-on
threshold of the MOSFET. Be aware that the maximum
voltage difference between DRVCC and BGRTN is 15V. If,
for example, VBGRTN = –2V, the maximum voltage on
DRVCC pin is now 13V instead of 15V.
Current Limit Programming
Programming current limit on the LTC3703-5 is straight
forward. The IMAX pin sets the current limit by setting the
maximum allowable voltage drop across the bottom
MOSFET. The voltage across the MOSFET is set by its onresistance and the current flowing in the inductor, which
is the same as the output current. The LTC3703-5 current
limit circuit inverts the negative voltage across the MOSFET
before comparing it to the voltage at IMAX, allowing the
current limit to be set with a positive voltage.
To set the current limit, calculate the expected voltage
drop across the bottom MOSFET at the maximum desired
current and maximum junction temperature:
VPROG = (ILIMIT)(RDS(ON))(1 + δ)
where δ is explained in the MOSFET Selection section.
VPROG is then programmed at the IMAX pin using the
internal 12µA pull-up and an external resistor:
RIMAX = VPROG/12µA
The current limit value should be checked to ensure that
ILIMIT(MIN) > IOUT(MAX). The minimum value of current limit
generally occurs with the largest VIN at the highest ambient temperature, conditions that cause the largest power
loss in the converter. Note that it is important to check for
self-consistency between the assumed MOSFET junction
temperature and the resulting value of ILIMIT which heats
the MOSFET switches.
Caution should be used when setting the current limit
based upon the RDS(ON) of the MOSFETs. The maximum
current limit is determined by the minimum MOSFET on-
resistance. Data sheets typically specify nominal and
maximum values for RDS(ON), but not a minimum. A
reasonable assumption is that the minimum RDS(ON) lies
the same amount below the typical value as the maximum
lies above it. Consult the MOSFET manufacturer for further
guidelines.
For best results, use a VPROG voltage between 100mV and
500mV. Values outside of this range may give less accurate current limit. The current limit can also be disabled by
floating the IMAX pin.
FEEDBACK LOOP/COMPENSATION
Feedback Loop Types
In a typical LTC3703-5 circuit, the feedback loop consists
of the modulator, the external inductor, the output capacitor and the feedback amplifier with its compensation
network. All of these components affect loop behavior and
must be accounted for in the loop compensation. The
modulator consists of the internal PWM generator, the
output MOSFET drivers and the external MOSFETs themselves. From a feedback loop point of view, it looks like a
linear voltage transfer function from COMP to SW and has
a gain roughly equal to the input voltage. It has fairly
benign AC behavior at typical loop compensation frequencies with significant phase shift appearing at half the
switching frequency.
The external inductor/output capacitor combination makes
a more significant contribution to loop behavior. These
components cause a second order LC roll off at the output,
with the attendant 180° phase shift. This rolloff is what filters
the PWM waveform, resulting in the desired DC output
voltage, but the phase shift complicates the loop compensation if the gain is still higher than unity at the pole frequency. Eventually (usually well above the LC pole
frequency), the reactance of the output capacitor will approach its ESR and the rolloff due to the capacitor will stop,
leaving 6dB/octave and 90° of phase shift (Figure 10).
So far, the AC response of the loop is pretty well out of the
user’s control. The modulator is a fundamental piece of the
LTC3703-5 design and the external L and C are usually
chosen based on the regulation and load current requirements without considering the AC loop response. The
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FREQ
–6dB/OCT
–6dB/OCT
–
OUT
RB
–90
PHASE
–6dB/OCT
GAIN
–12dB/OCT
0
C1
R2
R1
FB
GAIN (dB)
GAIN (dB)
GAIN
IN
PHASE (DEG)
PHASE (DEG)
AV
C2
0
FREQ
+
VREF
–90
–180
–180
PHASE
–270
–270
–360
–360
37035 F10
37035 F12
Figure 12. Type 2 Schematic and Transfer Function
feedback amplifier, on the other hand, gives us a handle
with which to adjust the AC response. The goal is to have
180° phase shift at DC (so the loop regulates) and something less than 360° phase shift at the point that the loop
gain falls to 0dB. The simplest strategy is to set up the
feedback amplifier as an inverting integrator, with the 0dB
frequency lower than the LC pole (Figure 11). This “Type
1” configuration is stable but transient response is less
than exceptional if the LC pole is at a low frequency.
for an extended frequency range. LTC3703-5 circuits
using conventional switching grade electrolytic output
capacitors can often get acceptable phase margin with
Type 2 compensation.
R1
FB
–
–6dB/OCT
OUT
RB
VREF
GAIN
0
FREQ
+
–90
–180
PHASE
–270
“Type 3” loops (Figure 13) use two poles and two zeros to
obtain a 180° phase boost in the middle of the frequency
band. A properly designed Type 3 circuit can maintain
acceptable loop stability even when low output capacitor
ESR causes the LC section to approach 180° phase shift
well above the initial LC roll-off. As with a Type 2 circuit,
the loop should cross through 0dB in the middle of the
phase bump to maximize phase margin. Many LTC3703-5
circuits using low ESR tantalum or OS-CON output capacitors need Type 3 compensation to obtain acceptable phase
margin with a high bandwidth feedback loop.
IN
–360
C3
R2
C1
37035 F11
Figure 11. Type 1 Schematic and Transfer Function
Figure 12 shows an improved “Type 2” circuit that uses an
additional pole-zero pair to temporarily remove 90° of
phase shift. This allows the loop to remain stable with 90°
more phase shift in the LC section, provided the loop
reaches 0dB gain near the center of the phase “bump.”
Type 2 loops work well in systems where the ESR zero in
the LC roll-off happens close to the LC pole, limiting the
total phase shift due to the LC. The additional phase
compensation in the feedback amplifier allows the 0dB
point to be at or above the LC pole frequency, improving
loop bandwidth substantially over a simple Type 1 loop. It
has limited ability to compensate for LC combinations
where low capacitor ESR keeps the phase shift near 180°
R1
R3
FB
–
VREF
–6dB/OCT
GAIN
OUT
RB
PHASE (DEG)
C2
GAIN (dB)
C1
IN
PHASE (DEG)
GAIN (dB)
Figure 10. Transfer Function of Buck Modulator
+6dB/OCT
–6dB/OCT
0
FREQ
+
–90
–180
PHASE
–270
–360
37035 F13
Figure 13. Type 3 Schematic and Transfer Function
Feedback Component Selection
Selecting the R and C values for a typical Type 2 or Type␣ 3
loop is a nontrivial task. The applications shown in this
data sheet show typical values, optimized for the power
components shown. They should give acceptable performance with similar power components, but can be way off
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if even one major power component is changed significantly. Applications that require optimized transient response will require recalculation of the compensation
values specifically for the circuit in question. The underlying mathematics are complex, but the component values
can be calculated in a straightforward manner if we know
the gain and phase of the modulator at the crossover
frequency.
Modulator gain and phase can be measured directly from
a breadboard or can be simulated if the appropriate
parasitic values are known. Measurement will give more
accurate results, but simulation can often get close enough
to give a working system. To measure the modulator gain
and phase directly, wire up a breadboard with an LTC3703-5
and the actual MOSFETs, inductor and input and output
capacitors that the final design will use. This breadboard
should use appropriate construction techniques for high
speed analog circuitry: bypass capacitors located close to
the LTC3703-5, no long wires connecting components,
appropriately sized ground returns, etc. Wire the feedback
amplifier as a simple Type 1 loop, with a 10k resistor from
VOUT to FB and a 0.1µF feedback capacitor from COMP to
FB. Choose the bias resistor (RB) as required to set the
desired output voltage. Disconnect RB from ground and
connect it to a signal generator or to the source output of
a network analyzer (Figure 14) to inject a test signal into
the loop. Measure the gain and phase from the COMP pin
to the output node at the positive terminal of the output
capacitor. Make sure the analyzer’s input is AC coupled so
that the DC voltages present at both the COMP and VOUT
5V
VIN
+
+
10µF
CIN
VCC
VCOMP
TO
ANALYZER 0.1µF
RB
VIN
fSET
TG
COMP
SW
LTC3703-5
FB
BG
NC
AC
SOURCE
FROM
ANALYZER
BOOST
DRVCC
10k
RUN/SS
M1
LEXT
+
M2
VOUT
TO
ANALYZER
COUT
INV
MODE/SYNC
GND BGRTN
nodes don’t corrupt the measurements or damage the
analyzer.
If breadboard measurement is not practical, a SPICE
simulation can be used to generate approximate gain/
phase curves. Plug the expected capacitor, inductor and
MOSFET values into the following SPICE deck and generate an AC plot of V(VOUT )/V(COMP) in dB and phase of
VOUT in degrees. Refer to your SPICE manual for details of
how to generate this plot.
*3703-5 modulator gain/phase
*2003 Linear Technology
*this file written to run with PSpice 8.0
*may require modifications for other
SPICE simulators
*MOSFETs
rfet mod sw 0.02
;MOSFET rdson
*inductor
lext sw out1 10u
rl out1 out 0.015
;inductor value
;inductor series R
*output cap
cout out out2 540u
resr out2 0 0.01
;capacitor value
;capacitor ESR
*3703-5 internals
emod mod 0 value = {43*v(comp)}
;3703-5multiplier
vstim comp 0 0 ac 1 ;ac stimulus
.ac dec 100 1k 1meg
.probe
.end
With the gain/phase plot in hand, a loop crossover frequency can be chosen. Usually the curves look something
like Figure 10. Choose the crossover frequency in the
rising or flat parts of the phase curve, beyond the external
LC poles. Frequencies between 10kHz and 50kHz usually
work well. Note the gain (GAIN, in dB) and phase (PHASE,
in degrees) at this point. The desired feedback amplifier
gain will be –GAIN to make the loop gain at 0dB at this
frequency. Now calculate the needed phase boost, assuming 60° as a target phase margin:
BOOST = – (PHASE + 30°)
37035 F14
Figure 14. Modulator Gain/Phase Measurement Set-Up
If the required BOOST is less than 60°, a Type 2 loop can
be used successfully, saving two external components.
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BOOST values greater than 60° usually require Type 3
loops for satisfactory performance.
Finally, choose a convenient resistor value for R1 (10k is
usually a good value). Now calculate the remaining values:
(K is a constant used in the calculations)
f = chosen crossover frequency
G = 10(GAIN/20) (this converts GAIN in dB to G in
absolute gain)
TYPE 2 Loop:
 BOOST

K = tan
+ 45°
 2

1
2π • f • G • K • R1
C1 = C 2 K2 − 1
C2 =
(
)
K
2π • f • C1
VREF (R1)
RB =
VOUT − VREF
R2 =
TYPE 3 Loop:
 BOOST

K = tan2 
+ 45°
 4

1
C2 =
2π • f • G • R1
C1 = C 2 K − 1
( )
K
2π • f • C1
R1
R3 =
K −1
1
C3 =
2πf K • R3
VREF (R1)
RB =
VOUT − VREF
R2 =
Boost Converter Design
The following sections discuss the use of the LTC3703-5
as a step-up (boost) converter. In boost mode, the
LTC3703-5 can step-up output voltages as high as 60V.
These sections discuss only the design steps specific to a
boost converter. For the design steps common to both a
buck and a boost, see the applicable section in the buck
mode section. An example of a boost converter circuit is
shown in the Typical Applications section. To operate the
LTC3703-5 in boost mode, the INV pin should be tied to
the VCC voltage (or a voltage above 2V). Note that in boost
mode, pulse-skip operation and the line feedforward compensation are disabled.
For a boost converter, the duty cycle of the main switch is:
VOUT – VIN
VOUT
For high VOUT to VIN ratios, the maximum VOUT is limited
by the LTC3703-5’s maximum duty cycle which is typically
93%. The maximum output voltage is therefore:
D=
VOUT (MAX) =
VIN(MIN)
≅ 14VIN(MIN)
1 – DMAX
Boost Converter: Inductor Selection
In a boost converter, the average inductor current equals
the average input current. Thus, the maximum average
inductor current can be calculated from:
IL(MAX) =
IO(MAX)
VO
= IO(MAX) •
1 − DMAX
VIN(MIN)
Similar to a buck converter, choose the ripple current to be
20% to 40% of IL(MAX). The ripple current amplitude then
determines the inductor value as follows:
L=
VIN(MIN)
• DMAX
∆IL • f
The minimum required saturation current for the inductor
is:
IL(SAT) > IL(MAX) + ∆IL/2
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Boost Converter: Power MOSFET Selection
For information about choosing power MOSFETs for a
boost converter, see the Power MOSFET Selection section for the buck converter, since MOSFET selection is
similar. However, note that the power dissipation equations for the MOSFETs at maximum output current in a
boost converter are:
2
 I

PMAIN = DMAX  MAX  (1 + δ )RDS(ON) +
 1 – DMAX 
 I

1
VOUT2  MAX  (RDR )(CMILLER ) •
 1 – DMAX 
2

1
1 
+
( f )

 VCC – VTH(IL) VTH(IL) 

1 
2
PSYNC = – 
 (IMAX ) (1 + δ )RDS(ON)
 1 – DMAX 
Boost Converter: Output Capacitor Selection
In boost mode, the output capacitor requirements are
more demanding due to the fact that the current waveform
is pulsed instead of continuous as in a buck converter. The
choice of component(s) is driven by the acceptable ripple
voltage which is affected by the ESR, ESL and bulk
capacitance as shown in Figure 15. The total output ripple
voltage is:

1
ESR 
∆VOUT = IO(MAX) 
+

 f • C OUT 1 – DMAX 
where the first term is due to the bulk capacitance and
second term due to the ESR.
The choice of output capacitor is driven also by the RMS
ripple current requirement. The RMS ripple current is:
IRMS(COUT ) ≈ IO(MAX) •
VO – VIN(MIN)
VIN(MIN)
At lower output voltages (less than 30V), it may be
possible to satisfy both the output ripple voltage and RMS
ripple current requirements with one or more capacitors of
∆VCOUT
VOUT
(AC)
∆VESR
RINGING DUE TO
TOTAL INDUCTANCE
(BOARD + CAP)
Figure 15. Output Voltage Ripple
Waveform for a Boost Converter
a single capacitor type. However, at output voltages above
30V where capacitors with both low ESR and high bulk
capacitance are hard to find, the best approach is to use a
combination of aluminum and ceramic capacitors (see
discussion in Input Capacitor section for the buck converter). With this combination, the ripple voltage can be
improved significantly. The low ESR ceremic capacitor
will minimize the ESR step, while the electrolytic will
supply the required bulk capacitance.
Boost Converter: Input Capacitor Selection
The input capacitor of a boost converter is less critical than
the output capacitor, due to the fact that the inductor is in
series with the input and the input current waveform is
continuous. The input voltage source impedance determines the size of the input capacitor, which is typically in
the range of 10µF to 100µF. A low ESR capacitor is
recommended though not as critical as for the output
capacitor.
The RMS input capacitor ripple current for a boost converter is:
IRMS(CIN) = 0.3 •
VIN(MIN)
• DMAX
L• f
Please note that the input capacitor can see a very high
surge current when a battery is suddenly connected to the
input of the converter and solid tantalum capacitors can
fail catastrophically under these conditions. Be sure to
specify surge-tested capacitors!
Boost Converter: Current Limit Programming
The LTC3703-5 provides current limiting in boost mode by
monitoring the VDS of the main switch during its on-time
and comparing it to the voltage at IMAX. To set the current
limit, calculate the expected voltage drop across the
MOSFET at the maximum desired inductor current and
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maximum junction temperature. The maximum inductor
current is a function of both duty cycle and maximum load
current, so the limit must be set for the maximum expected
duty cycle (minimum VIN) in order to ensure that the
current limit does not kick in at loads < IO(MAX):
VPROG =
GAIN
(dB)
PHASE
(DEG)
GAIN
AV
–12dB/OCT
0
0
IO(MAX)
RDS(ON) (1 + δ)
1 – DMAX
–90
PHASE
 V

=  OUT  IO(MAX) • RDS(ON) (1 + δ)
 VIN(MIN) 
–180
37035 F16
Once VPROG is determined, RIMAX is chosen as follows:
Figure 16. Transfer Function of Boost Modulator
RIMAX = VPROG/12µA
Note that in a boost mode architecture, it is only possible
to provide protection for “soft” shorts where VOUT > VIN.
For hard shorts, the inductor current is limited only by the
input supply capability. Refer to Current Limit Programming for buck mode for further considerations for current
limit programming.
quency so that the overall loop gain is 0dB here. The
compensation component to achieve this, using a Type 1
amplifier (see Figure 11), is:
Boost Converter: Feedback Loop/Compensation
Run/Soft-Start Function
Compensating a voltage mode boost converter is unfortunately more difficult than for a buck converter. This is due
to an additional right-half plane (RHP) zero that is present
in the boost converter but not in a buck. The additional phase
lag resulting from the RHP zero is difficult if not impossible
to compensate even with a Type 3 loop, so the best approach
is usually to roll off the loop gain at a lower frequency than
what could be achievable in buck converter.
The RUN/SS pin is a multipurpose pin that provide a softstart function and a means to shut down the LTC3703-5.
Soft-start reduces the input supply’s surge current by
gradually increasing the duty cycle and can also be used
for power supply sequencing.
A typical gain/phase plot of a voltage-mode boost converter is shown in Figure 16. The modulator gain and
phase can be measured as described for a buck converter
or can be estimated as follows:
GAIN (COMP-to-VOUT DC gain) = 20Log(VOUT2/VIN)
Dominant Pole: fP =
1
VIN
•
VOUT 2π LC
Since significant phase shift begins at frequencies above
the dominant LC pole, choose a crossover frequency no
greater than about half this pole frequency. The gain of the
compensation network should equal –GAIN at this fre-
G = 10–GAIN/20
C1 = 1/(2π • f • G • R1)
Pulling RUN/SS below 1V puts the LTC3703-5 into a low
quiescent current shutdown (IQ ≅ 25µA). This pin can be
driven directly from logic as shown in Figure 17. Releasing
RUN/SS
2V/DIV
VOUT
5V/DIV
IL
2A/DIV
VIN = 50V
ILOAD = 2A
CSS = 0.01µF
2ms/DIV
37035 F17
Figure 17. LTC3703-5 Startup Operation
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the RUN/SS pin allows an internal 4µA current source to
charge up the soft-start capacitor CSS. When the voltage
on RUN/SS reaches 1V, the LTC3703-5 begins operating
at its minimum on-time. As the RUN/SS voltage increases
from 1V to 3V, the duty cycle is allowed to increase from
0% to 100%. The duty cycle control minimizes input
supply inrush current and elimates output voltage overshoot at start-up and ensures current limit protection even
with a hard short. The RUN/SS voltage is internally clamped
at 4V.
If RUN/SS starts at 0V, the delay before starting is
approximately:
1V
C SS = (0.25s / µF )C SS
4µA
plus an additional delay, before the output will reach its
regulated value, of:
tDELAY,START =
3V – 1V
C SS = (0.5s / µF )C SS
4µA
The start delay can be reduced by using diode D1 in
Figure 18.
tDELAY,REG ≥
3.3V
OR 5V
RUN/SS
RUN/SS
D1
CSS
CSS
37035 F18
Figure 18. RUN/SS Pin Interfacing
MODE/SYNC Pin (Operating Mode and Secondary
Winding Control)
The MODE/SYNC pin is a dual function pin that can be used
for enabling or disabling Pulse Skip Mode operation and
also as an external clock input for synchronizing the internal oscillator (see next section). Pulse Skip Mode is enabled
when the MODE/SYNC pin is above 0.8V and is disabled,
i.e. forced continuous, when the pin is below 0.8V.
In addition to providing a logic input to force continuous
operation and external synchronization, the MODE/SYNC
pin provides a means to regulate a flyback winding output
as shown in Figure 9c. The auxiliary output is taken from
a second winding on the core of the inductor, converting
it to a transformer. The auxiliary output voltage is set by
the main output voltage and the turns ratio of the extra
winding to the primary winding as follows:
VSEC ≈ (N + 1)VOUT
Since the secondary winding only draws current when the
synchronous switch is on, load regulation at the auxiliary
output will be relatively good as long as the main output is
running in continuous mode. As the load on the primary
output drops and the LTC3703-5 switches to Pulse Skip
Mode operation, the auxiliary output may not be able to
maintain regulation, especially if the load on the auxiliary
output remains heavy. To avoid this, the auxiliary output
voltage can be divided down with a conventional feedback
resistor string with the divided auxiliary output voltage fed
back to the MODE/SYNC pin. The MODE/SYNC threshold
is trimmed to 800mV with 20mV of hysteresis, allowing
precise control of the auxiliary voltage and is set as
follows:
 R1
VSEC(MIN) ≈ 0.8V 1 + 
 R2
where R1 and R2 are shown in Figure 9c.
If the LTC3703-5 is operating in Pulse Skip Mode and the
auxiliary output voltage drops below VSEC(MIN), the MODE/
SYNC pin will trip and the LTC3703-5 will resume continuous operation regardless of the load on the main output.
Thus, the MODE/SYNC pin removes the requirement that
power must be drawn from the inductor primary in order
to extract power from the auxiliary winding. With the loop
in continuous mode (MODE/SYNC < 0.8V), the auxiliary
outputs may nominally be loaded without regard to the
primary output load.
The following table summarizes the possible states available on the MODE/SYNC pin:
Table 1.
MODE/SYNC Pin
DC Voltage: 0V to 0.75V
DC Voltage: ≥ 0.87V
Feedback Resistors
Ext. Clock: 0V to ≥ 2V
Condition
Forced Continuous
Current Reversal Enabled
Pulse Skip Mode Operation
No Current Reversal
Regulating a Secondary Winding
Forced Continuous
No Current Reversal
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MODE/SYNC Pin (External Synchronization)
The internal LTC3703-5 oscillator can be synchronized to
an external oscillator by applying and clocking the MODE/
SYNC pin with a signal above 2VP-P. The internal oscillator
locks to the external clock after the second clock transition is received. When external synchronization is detected, LTC3703-5 will operate in forced continuous
mode. If an external clock transition is not detected for
three successive periods, the internal oscillator will revert
to the frequency programmed by the RSET resistor. The
internal oscillator can synchronize to frequencies between 100kHz and 600kHz, independent of the frequency
programmed by the RSET resistor. However, it is recommended that an RSET resistor be chosen such that the
frequency programmed by the RSET resistor is close to the
expected frequency of the external clock. In this way, the
best converter operation (ripple, component stress, etc)
is achieved if the external clock signal is lost.
Minimum On-Time Considerations (Buck Mode)
Minimum on-time tON(MIN) is the smallest amount of time
that the LTC3703-5 is capable of turning the top MOSFET
on and off again. It is determined by internal timing delays
and the amount of gate charge required to turn on the top
MOSFET. Low duty cycle applications may approach this
minimum on-time limit and care should be taken to ensure
that:
V
tON = OUT > tON(MIN)
VIN • f
where tON(MIN) is typically 200ns.
If the duty cycle falls below what can be accommodated by
the minimum on-time, the LTC3703-5 will begin to skip
cycles. The output will be regulated, but the ripple current
and ripple voltage will increase. If lower frequency operation is acceptable, the on-time can be increased above
tON(MIN) for the same step-down ratio.
PC board trace clearance between high and low voltage
pins in higher voltage applications. Where clearance is an
issue, the G28 package should be used. The G28 package
has 4 unconnected pins between the all adjacent high
voltage and low voltage pins, providing 5(0.0106”) =
0.053” clearance which will be sufficient for most applications up to 60V. For more information, refer to the printed
circuit board design standards described in IPC-2221
(www.ipc.org).
Efficiency Considerations
The efficiency of a switching regulator is equal to the
output power divided by the input power (x100%). Percent efficiency can be expressed as:
%Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage
of input power. It is often useful to analyze the individual
losses to determine what is limiting the efficiency and
what change would produce the most improvement. Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC3703-5 circuits: 1) LTC3703-5 VCC current,
2) MOSFET gate current, 3) I2R losses, 4) Topside MOSFET
transition losses.
1. VCC Supply current. The VCC current is the DC supply
current given in the Electrical Characteristics table which
powers the internal control circuitry of the LTC3703-5.
Total supply current is typically about 2.5mA and usually
results in a small (<1%) loss which is proportional to VCC.
Pin Clearance/Creepage Considerations
2. DRVCC current is MOSFET driver current. This current
results from switching the gate capacitance of the power
MOSFETs. Each time a MOSFET gate is switched on and
then off, a packet of gate charge QG moves from DRVCC to
ground. The resulting dQ/dt is a current out of the DRVCC
supply. In continuous mode, IDRVCC = f(QG(TOP) + QG(BOT)),
where QG(TOP) and QG(BOT) are the gate charges of the top
and bottom MOSFETs.
The LTC3703-5 is available in two packages (GN16 and
G28) both with identical functionality. The GN16 package
gives the smallest size solution, however the 0.013”
(minimum) space between pins may not provide sufficient
3. I2R losses are predicted from the DC resistances of
MOSFETs, the inductor and input and output capacitor
ESR. In continuous mode, the average output current
flows through L but is “chopped” between the topside
37035f
25
LTC3703-5
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APPLICATIO S I FOR ATIO
MOSFET and the synchronous MOSFET. If the two
MOSFETs have approximately the same RDS(ON), then the
resistance of one MOSFET can simply be summed with the
DCR resistance of L to obtain I2R losses. For example, if
each RDS(ON) = 25mΩ and RL = 25mΩ, then total resistance is 50mΩ. This results in losses ranging from 1% to
5% as the output current increases from 1A to 5A for a 5V
output.
4. Transition losses apply only to the topside MOSFET in
buck mode and they become significant when operating at
higher input voltages (typically 20V or greater). Transition
losses can be estimated from the second term of the PMAIN
equation found in the Power MOSFET Selection section.
The transition losses can become very significant at the
high end of the LTC3703-5 operating voltage range. To
improve efficiency, one may consider lowering the frequency and/or using MOSFETs with lower CRSS at the
expense of higher RDS(ON).
Other losses including CIN and COUT ESR dissipative
losses, Schottky conduction losses during dead-time, and
inductor core losses generally account for less than 2%
total additional loss.
Transient Response
Due to the high gain error amplifier and line feedforward
compensation of the LTC3703-5, the output accuracy due
to DC variations in input voltage and output load current
will be almost negligible. For the few cycles following a
load transient, however, the output deviation may be
larger while the feedback loop is responding. Consider a
typical 48V input to 5V output application circuit,
subjected to a 1A to 5A load transient. Initially, the loop is
in regulation and the DC current in the output capacitor is
zero. Suddenly, an extra 4A (= 5A-1A) flows out of the
output capacitor while the inductor is still supplying only
1A. This sudden change will generate a (4A) • (RESR)
voltage step at the output; with a typical 0.015Ω output
capacitor ESR, this is a 60mV step at the output.
The feedback loop will respond and will move at the bandwidth allowed by the external compensation network
towards a new duty cycle. If the unity gain crossover frequency is set to 50kHz, the COMP pin will get to 60% of the
way to 90% duty cycle in 3µs. Now the inductor is seeing
43V across itself for a large portion of the cycle and its
current will increase from 1A at a rate set by di/dt = V/L. If
the inductor value is 10µH, the peak di/dt will be 43V/10µH
or 4.3A/µs. Sometime in the next few micro-seconds after
the switch cycle begins, the inductor current will have
risen to the 5A level of the load current and the output
voltage will stop dropping. At this point, the inductor current will rise somewhat above the level of the output current to replenish the charge lost from the output capacitor
during the load transient. With a properly compensated
loop, the entire recovery time will be inside of 10µs.
Most loads care only about the maximum deviation from
ideal, which occurs somewhere in the first two cycles after
the load step hits. During this time, the output capacitor
does all the work until the inductor and control loop regain
control. The initial drop (or rise if the load steps down) is
entirely controlled by the ESR of the capacitor and amounts
to most of the total voltage drop. To minimize this drop,
choose a low ESR capacitor and/or parallel multiple capacitors at the output. The capacitance value accounts for
the rest of the voltage drop until the inductor current rises.
With most output capacitors, several devices paralleled to
get the ESR down will have so much capacitance that this
drop term is negligible. Ceramic capacitors are an exception; a small ceramic capacitor can have suitably low ESR
with relatively small values of capacitance, making this
second drop term more significant.
Optimizing Loop Compensation
Loop compensation has a fundamental impact on transient recovery time, the time it takes the LTC3703-5 to
recover after the output voltage has dropped due to a load
step. Optimizing loop compensation entails maintaining
the highest possible loop bandwidth while ensuring loop
stability. The feedback component selection section describes in detail the techniques used to design an optimized Type 3 feedback loop, appropriate for most
LTC3703-5 systems.
37035f
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APPLICATIO S I FOR ATIO
Measurement Techniques
Measuring transient response presents a challenge in two
respects: obtaining an accurate measurement and generating a suitable transient to test the circuit. Output measurements should be taken with a scope probe directly
across the output capacitor. Proper high frequency probing techniques should be used. In particular, don’t use the
6" ground lead that comes with the probe! Use an adapter
that fits on the tip of the probe and has a short ground clip
to ensure that inductance in the ground path doesn’t cause
a bigger spike than the transient signal being measured.
Conveniently, the typical probe tip ground clip is spaced
just right to span the leads of a typical output capacitor.
Now that we know how to measure the signal, we need to
have something to measure. The ideal situation is to use
the actual load for the test and switch it on and off while
watching the output. If this isn’t convenient, a current step
generator is needed. This generator needs to be able to
turn on and off in nanoseconds to simulate a typical
switching logic load, so stray inductance and long clip
leads between the LTC3703-5 and the transient generator
must be minimized.
Figure 19 shows an example of a simple transient generator. Be sure to use a noninductive resistor as the load
element—many power resistors use an inductive spiral
pattern and are not suitable for use here. A simple solution
is to take ten 1/4W film resistors and wire them in parallel
LTC3703-5
to get the desired value. This gives a noninductive resistive
load which can dissipate 2.5W continuously or 50W if
pulsed with a 5% duty cycle, enough for most LTC3703-5
circuits. Solder the MOSFET and the resistor(s) as close to
the output of the LTC3703-5 circuit as possible and set up
the signal generator to pulse at a 100Hz rate with a 5% duty
cycle. This pulses the LTC3703-5 with 500µs
transients10ms apart, adequate for viewing the entire
transient recovery time for both positive and negative
transitions while keeping the load resistor cool.
Design Example
As a design example, take a supply with the following
specifications: VIN = 20V to 60V (48V nominal), VOUT =
12V ±5%, IOUT(MAX) = 10A, f=250kHz. First, calculate RSET
to give the 250kHz operating frequency:
RSET = 7100/(250-25) = 31.6k
Next, choose the inductor value for about 40% ripple
current at maximum VIN:
L=
12V
 12 
 1 –  = 10µH
(250kHz)(0.4)(10 A)  60 
With 10µH inductor, ripple current will vary from 1.9A to
3.8A (19% to 38%) over the input supply range.
Next, verify that the minimum on-time is not violated. The
minimum on-time occurs at maximum VIN:
tON(MIN) =
VOUT
VOUT
VIN(MIN)( f)
=
12
= 800ns
60(250kHz)
RLOAD
IRFZ44 OR
EQUIVALENT
PULSE
GENERATOR
50Ω
0V TO 10V
100Hz, 5%
DUTY CYCLE
37035 F19
LOCATE CLOSE TO THE OUTPUT
Figure 19. Transient Load Generator
which is above the LTC3703-5’s 200ns minimum on-time.
Next, choose the top and bottom MOSFET switch. Since
the drain of each MOSFET will see the full supply voltage
60V(max) plus any ringing, choose a 60V MOSFET.
Si7850DP has a 60V BVDSS, RDS(ON) = 22mΩ(max), δ =
0.007/°C, CMILLER = (9nC – 3nC)/30V = 200pF, VGS(MILLER)
= 3.8V, θJA = 20°C/W. The power dissipation can be
37035f
27
LTC3703-5
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APPLICATIO S I FOR ATIO
estimated at maximum input voltage, assuming a junction
temperature of 100°C (30°C above an ambient of 70°C):
12
(10)2 [1 + 0.007(100 – 25)](0.022)
60
1
1
 10 

+ (60)2   (2)(200pF) • 
+
 (250k)
 2
 10 – 3.8 3.8 
= 0.67W + 0.76W = 1.43W
PMAIN =
And double check the assumed TJ in the MOSFET:
TJ = 70°C + (1.43W)(20°C/W) = 99°C
Since the synchronous MOSFET will be conducting over
twice as long each period (almost 100% of the period in
short circuit) as the top MOSFET, use two Si7850DP
MOSFETs on the bottom:
 60 − 12 
PSYNC = 
 (10)2[1 + 0.007(100 – 25)] •
 60 
 0.022 

 = 1.34W
 2 
TJ = 70°C + (1.34W)(20°C/W) = 97°C
However, a 0A to 10A load step will cause an output
voltage change of up to:
∆VOUT(STEP) = ∆ILOAD(ESR) = (10A)(0.009Ω)
= 90mV
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC3703-5. These items are also illustrated graphically in
the layout diagram of Figure 20. For layout of a Boost Mode
Converter, layout is similar with VIN and VOUT swapped.
Check the following in your layout:
1. Keep the signal and power grounds separate. The signal
ground consists of the LTC3703-5 GND pin, the ground
return of CVCC, and the (–) terminal of VOUT. The power
ground consists of the Schottky diode anode, the source
of the bottom side MOSFET, and the (–) terminal of the
input capacitor and DRVCC capacitor. Connect the signal
and power grounds together at the (–) terminal of the
output capacitor. Also, try to connect the (–) terminal of
the output capacitor as close as possible to the (–)
terminals of the input and DRVCC capacitor and away from
the Schottky loop described in (2).
Next, set the current limit resistor. Since IMAX = 10A, the
limit should be set such that the minimum current limit is
>10A. Minimum current limit occurs at maximum RDS(ON).
Using the above calculation for bottom MOSFET TJ, the
max RDS(ON) = (22mΩ/2) [1 + 0.007 (97-25)] = 16.5mΩ
2. The high di/dt loop formed by the top N-channel
MOSFET, the bottom MOSFET and the CIN capacitor
should have short leads and PC trace lengths to minimize
high frequency noise and voltage stress from inductive
ringing.
Therefore, IMAX pin voltage should be set to (10A)(0.0165)
= 0.165V. The RSET resistor can now be chosen to be
0.165V/12µA = 14kΩ.
3. Connect the drain of the top side MOSFET directly to the
(+) plate of CIN, and connect the source of the bottom side
MOSFET directly to the (–) terminal of CIN. This capacitor
provides the AC current to the MOSFETs.
CIN is chosen for an RMS current rating of about 5A
(IMAX/2) at 85°C. For the output capacitor, two low ESR
OSCON capacitors (18mΩ each) are used to minimize
output voltage changes due to inductor current ripple and
load steps. The ripple voltage will be:
∆VOUT(RIPPLE) = ∆IL(MAX) (ESR) = (4A)(0.018Ω/2)
= 36mV
4. Place the ceramic CDRVCC decoupling capacitor immediately next to the IC, between DRVCC and BGRTN. This
capacitor carries the MOSFET drivers’ current peaks.
Likewise the CB capacitor should also be next to the IC
between BOOST and SW.
37035f
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LTC3703-5
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APPLICATIO S I FOR ATIO
5. Place the small-signal components away from high
frequency switching nodes (BOOST, SW, TG, and BG). In
the layout shown in Figure 20, all the small signal components have been placed on one side of the IC and all of the
power components have been placed on the other. This
also helps keep the signal ground and power ground
isolated.
6. A separate decoupling capacitor for the supply, VCC, is
useful with an RC filter between the DRVCC supply and VCC
pin to filter any noise injected by the drivers. Connect this
capacitor close to the IC, between the VCC and GND pins
and keep the ground side of the VCC capacitor (signal
ground) isolated from the ground side of the DRVCC
capacitor (power ground).
7. For optimum load regulation and true remote sensing,
the top of the output resistor divider should connect
independently to the top of the output capacitor (Kelvin
connection), staying away from any high dV/dt traces.
Place the divider resistors near the LTC3703-5 in order to
keep the high impedance FB node short.
8. For applications with multiple switching power converters connected to the same input supply, make sure that the
input filter capacitor for the LTC3703-5 is not shared with
other converters. AC input current from another converter
could cause substantial input voltage ripple, and this could
interfere with the operation of the LTC3703-5. A few
inches of PC trace or wire (L ≅ 100nH) between CIN of the
LTC3703-5 and the actual source VIN should be sufficient
to prevent input noise interference problems.
VCC
VIN
DB
1
RSET
RC1
3
CC1
CC2
2
4
RMAX
R2
CC3
BOOST
COMP
TG
FB
SW
8
RUN/SS
GND
BG
BGRTN
CIN
CB
13
11
DRVCC
+
14
6
INV
M1
15
LTC3703-5
12
IMAX
VCC
7
R1
FSET
16
5
CSS
RC2
MODE/SYNC VIN
L1
+
RF
+
VOUT
COUT
10
CDRVCC
X5R
9
D1
M2
–
CVCC
X5R
37035 F18
Figure 20. LTC3703-5 Buck Converter Suggested Layout
37035f
29
LTC3703-5
U
TYPICAL APPLICATIO S
15V-60V Input Voltage to 12V/10A Step-Down Converter with Pulse Skip Mode Enabled
VCC
5V TO 15V
+
1
16
MODE/SYNC VIN
RSET
25k 2
15
FSET
BOOST
RC1
10k
3
CC1
470pF
CC2
1000pF
RC2
100Ω
CC3
2200pF
COMP
4
RMAX 15k
R2
8.06k
1%
VIN
DB
15V TO 60V
MMDL770T1
22µF
25V
CSS
0.1µF
R1
113k
1%
FB
TG
SW
14
LTC3703-5
12
IMAX
VCC
6
11
INV
RUN/SS
8
GND
DRVCC
BG
BGRTN
M1
Si7850DP
CB
0.1µF
13
5
7
+
L1
8µH
RF
10Ω
COUT
220µF
25V
×2
M2
Si7460DP
10
VOUT
12V
10A
+
D1
MBR1100
CDRVCC
10µF
9
CIN
22µF
100V
×2
CVCC
1µF
37035 TA01
Single Input Supply 5V/5A Output Step-Down Converter
100Ω
10k
FZT600
*
VIN
6V TO 60V
5.1V
+
1
RSET 25k
RC1
10k
CC2
1000pF
3
CC1
470pF
RMAX 15k
R2
21.5k
1%
RC2
100Ω
CC3
2200pF
R1
113k
1%
2
4
FSET
COMP
BOOST
TG
7
8
DB1
MMDL770T1
16
+
15
4.7Ω
14
CB
0.1µF
13
SW
LTC3703-5
5
12
VCC
IMAX
6
CSS
0.1µF
MODE/SYNC VIN
22µF
25V
FB
INV
RUN/SS
GND
DRVCC
BG
BGRTN
11
CMDSH-3
DB2
MMDL770T1
L1 4.7µH
RF
10Ω
M2
Si7850DP
10
9
M1
Si7850DP
CIN
22µF
100V
CDRVCC
10µF
COUT
220µF
25V
+
VOUT
5V
5A
D1
MBR1100
CVCC
1µF
*OPTIONAL ZENER PROVIDES UNDERVOLTAGE LOCKOUT ON INPUT SUPPLY, VUVLO ≅ 5 + V Z
3703 TA02
37035f
30
LTC3703-5
U
PACKAGE DESCRIPTIO
GN Package
16-Lead Plastic SSOP (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1641)
.189 – .196*
(4.801 – 4.978)
.045 ±.005
.009
(0.229)
REF
16 15 14 13 12 11 10 9
.254 MIN
.150 – .165
.229 – .244
(5.817 – 6.198)
.0165 ± .0015
.150 – .157**
(3.810 – 3.988)
.0250 BSC
RECOMMENDED SOLDER PAD LAYOUT
1
.015 ± .004
× 45°
(0.38 ± 0.10)
.007 – .0098
(0.178 – 0.249)
2 3
4
5 6
7
.0532 – .0688
(1.35 – 1.75)
8
.004 – .0098
(0.102 – 0.249)
0° – 8° TYP
.016 – .050
(0.406 – 1.270)
NOTE:
1. CONTROLLING DIMENSION: INCHES
INCHES
2. DIMENSIONS ARE IN
(MILLIMETERS)
3. DRAWING NOT TO SCALE
.008 – .012
(0.203 – 0.305)
TYP
.0250
(0.635)
BSC
GN16 (SSOP) 0204
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
G Package
28-Lead Plastic SSOP (5.3mm)
(Reference LTC DWG # 05-08-1640)
9.90 – 10.50*
(.390 – .413)
28 27 26 25 24 23 22 21 20 19 18 17 16 15
1.25 ±0.12
7.8 – 8.2
5.3 – 5.7
0.42 ±0.03
7.40 – 8.20
(.291 – .323)
0.65 BSC
1 2 3 4 5 6 7 8 9 10 11 12 13 14
RECOMMENDED SOLDER PAD LAYOUT
2.0
(.079)
MAX
5.00 – 5.60**
(.197 – .221)
0° – 8°
0.09 – 0.25
(.0035 – .010)
0.55 – 0.95
(.022 – .037)
NOTE:
1. CONTROLLING DIMENSION: MILLIMETERS
MILLIMETERS
2. DIMENSIONS ARE IN
(INCHES)
3. DRAWING NOT TO SCALE
0.65
(.0256)
BSC
0.22 – 0.38
(.009 – .015)
TYP
*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED .152mm (.006") PER SIDE
**DIMENSIONS DO NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED .254mm (.010") PER SIDE
0.05
(.002)
MIN
G28 SSOP 0204
37035f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
31
LTC3703-5
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TYPICAL APPLICATIO
5V to 24V/5A Synchronous Boost Converter
+
1
RSET 25k
10k
3
CC1
100pF
0.1µF
RMAX 15k
R2
3.92k
1%
R1
113k
1%
2
CSS
0.1µF
4
MODE/SYNC VIN
FSET
BOOST
COMP
TG
FB
SW
15
14
6
11
8
DRVCC
RUN/SS
GND
BG
BGRTN
CB
0.1µF
13
LTC3703-5
12
IMAX
VCC
INV
DB
CMDSH-3
16
5
7
22µF
25V
RF
10Ω
COUT +
220µF
30V
×3
MBRS140T3
L1
3.3µH
M2
Si7892DP
10
9
M1
Si7390DP
VOUT
24V
5A
CIN
100µF
20V
VIN
4.5V TO 15V
+
CDRVCC
10µF
CVCC
1µF
37035 TA03
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37035f
32
Linear Technology Corporation
LT/TP 0404 1K • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900
●
FAX: (408) 434-0507 ● www.linear.com
 LINEAR TECHNOLOGY CORPORATION 2004