EL7516 NS ESI G D W R NE NT I S D FO A C E M E E D EPL M EN COM TIBLE R 16 Data Sheet E R NOT COMPA ISL975 PI N ® October 9, 2007 600kHz/1.2MHz PWM Step-Up Regulator Features The EL7516 is a high frequency, high efficiency step-up voltage regulator operated at constant frequency PWM mode. With an internal 1.5A, 200mΩ MOSFET, it can deliver up to 600mA output current at over 90% efficiency. The selectable 600kHz and 1.2MHz allows smaller inductors and faster transient response. An external compensation pin gives the user greater flexibility in setting frequency compensation allowing the use of low ESR Ceramic output capacitors. • >90% efficiency When shut down, it draws <10µA of current and can operate down to 2.5V input supply. These features along with 1.2MHz switching frequency makes it an ideal device for portable equipment and TFT-LCD displays. The EL7516 is available in an 8 Ld MSOP package with a maximum height of 1.1mm. The device is specified for operation over the full -40°C to +85°C temperature range. FN7333.6 • 1.6A, 200mΩ power MOSFET • VIN > 2.5V • 600kHz/1.2MHz switching frequency selection • Adjustable soft-start • Internal thermal protection • 1.1mm max height 8 Ld MSOP package • Pb-free plus anneal available (RoHS compliant) Applications • TFT-LCD displays • DSL modems • PCMCIA cards • Digital cameras Pinout • GSM/CDMA phones EL7516 (8 LD MSOP) TOP VIEW • Portable equipment • Handheld devices COMP 1 8 SS FB 2 7 FSEL SHDN 3 6 VDD GND 4 5 LX Ordering Information PART NUMBER PART MARKING TAPE & REEL PACKAGE PKG. DWG. # EL7516IY f - 8 Ld MSOP MDP0043 EL7516IY-T7 f 7” 8 Ld MSOP MDP0043 EL7516IY-T13 f 13” 8 Ld MSOP MDP0043 EL7516IYZ (Note) BARAA - 8 Ld MSOP (Pb-Free) MDP0043 EL7516IYZ-T7 (Note) BARAA 7” 8 Ld MSOP (Pb-Free) MDP0043 EL7516IYZ-T13 BARAA (Note) 13” 8 Ld MSOP (Pb-Free) MDP0043 NOTE: Intersil Pb-free plus anneal products employ special Pb-free material sets; molding compounds/die attach materials and 100% matte tin plate termination finish, which are RoHS compliant and compatible with both SnPb and Pb-free soldering operations. Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020. 1 CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc. Copyright Intersil Americas Inc. 2002, 2004-2007. All Rights Reserved All other trademarks mentioned are the property of their respective owners. EL7516 Absolute Maximum Ratings (TA = +25°C) Thermal Information LX to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .18V VDD to GND. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6.5V COMP, FB, SHDN, SS, FSEL to GND . . . . . . . -0.3V to (VDD +0.3V) Storage Temperature . . . . . . . . . . . . . . . . . . . . . . . .-65°C to +150°C Operating Ambient Temperature . . . . . . . . . . . . . . . .-40°C to +85°C Operating Junction Temperature . . . . . . . . . . . . . . . . . . . . . . +135°C Power Dissipation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . See Curves CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied. IMPORTANT NOTE: All parameters having Min/Max specifications are guaranteed. Typical values are for information purposes only. Unless otherwise noted, all tests are at the specified temperature and are pulsed tests, therefore: TJ = TC = TA Electrical Specifications PARAMETER VIN = 3.3V, VOUT = 12V, IOUT = 0mA, FSEL = GND, TA = +25°C unless otherwise specified. DESCRIPTION CONDITIONS MIN TYP MAX UNIT 10 µA IQ1 Quiescent Current - Shut-down SHDN = 0V 0.6 IQ2 Quiescent Current - Not Switching SHDN = VDD, FB = 1.3V 0.7 IQ3 Quiescent Current - Switching SHDN = VDD, FB = 1.0V 1.3 2 mA VFB Feedback Voltage 1.294 1.309 V IB-FB Feedback Input Bias Current 0.01 0.5 µA VDD Start-Up Input Voltage Range 5.5 V 1.272 2.6 mA DMAX - 600kHz Maximum Duty Cycle FSEL = 0V 84 90 % DMAX - 1.2MHz Maximum Duty Cycle FSEL = VDD 84 90 % 1.3 1.5 A ILIM Current Limit - Max Peak Input Current ISHDN Shut-down Input Bias Current SHDN = 0V 0.01 rDS-ON Switch ON-Resistance VDD = 2.7V, ILX = 1A 0.2 Switch Leakage Current VSW = 18V 0.01 ΔVOUT/ΔVIN Line Regulation 3V < VIN < 5.5V, VOUT = 12V 0.1 % ΔVOUT/ΔIOUT Load Regulation VIN = 3.3V, VOUT = 12V, IO = 30mA to 200mA 6.7 mV/A fOSC1 Switching Frequency Accuracy FSEL = 0V 500 620 740 kHz fOSC2 Switching Frequency Accuracy FSEL = VDD 1000 1250 1500 kHz VIL SHDN, FSEL Input Low Level 0.5 V VIH SHDN, FSEL Input High Level VIL SHDN, Input Low Level 5V Input Supply VIH SHDN, Input High Level 5V Input Supply 4.5 GM Error Amp Tranconductance ΔI = 5µA 90 AV Voltage Gain ILX-LEAK 0.1 µA Ω 3 µA 2.7 V 1.25 V V 130 170 1µ/Ω 350 V/V VDD-ON VDD UVLO On Threshold 2.40 2.51 2.60 V VDD-OFF VDD UVLO Off Threshold 2.20 2.30 2.40 V ISS Soft-start Charge Current 4 6 8 µA RCS Current Sense Transresistance 0.08 V/A OTP Over-temperature Protection 130 °C 2 FN7333.6 October 9, 2007 EL7516 Block Diagram SHDN FSEL REFERENCE GENERATOR VDD OSCILLATOR SS SHUTDOWN AND START-UP CONTROL LX PWM LOGIC CONTROLLER FET DRIVER COMPARATOR CURRENT SENSE GND FB GM AMPLIFIER COMP Pin Descriptions PIN NUMBER PIN NAME DESCRIPTION 1 COMP Compensation pin. Output of the internal error amplifier. Capacitor and resistor from COMP pin to ground. 2 FB Voltage feedback pin. Internal reference is 1.294V nominal. Connect a resistor divider from VOUT. VOUT = 1.294V (1 + R1/R2). See Typical Application Circuit. 3 SHDN 4 GND 5 LX 6 VDD Analog power supply input pin. 7 FSEL Frequency select pin. When FSEL is set low, switching frequency is set to 620kHz. When connected to high or VDD, switching frequency is set to 1.25MHz. 8 SS Shutdown control pin. Pull SHDN low to turn off the device. Analog and power ground. Power switch pin. Connected to the drain of the internal power MOSFET. Soft-start control pin. Connect a capacitor to control the converter start-up. Typical Application Circuit 1 COMP R3 3.9kΩ R1 2 FB C5 R2 4.7nF 10kΩ 3 SHDN 4 GND S1 3 SS 8 85.2kΩ FSEL 7 VDD 6 LX 5 C3 27nF C4 2.7V TO 5.5V + C1 0.1µF 22µF 10µH D1 + C2 12V 22µF FN7333.6 October 9, 2007 EL7516 Typical Performance Curves 95 0.6 LOAD REGULATION (%) EFFICIENCY (%) 0.4 90 85 80 0.2 0 -0.2 -0.4 -0.6 -0.8 75 -1.0 0 100 200 300 400 0 50 100 IOUT (mA) 250 300 350 FIGURE 2. LOAD REGULATION - 3.3V VIN TO 12V VOUT @ 1.3MHz 1.0 LOAD REGULATION (%) 90 EFFICIENCY (%) 200 IOUT (mA) FIGURE 1. EFFICIENCY - 3.3V VIN TO 12V VOUT @ 1.3MHz 85 80 0.5 0 -0.5 -1.0 75 0 100 200 300 400 0 50 100 150 200 250 300 350 IOUT (mA) IOUT (mA) FIGURE 3. EFFICIENCY - 3.3V VIN TO 12V VOUT @ 620kHz FIGURE 4. LOAD REGULATION - 3.3V VIN TO 12V VOUT @ 620kHz 1.0 LOAD REGULATION (%) 95 90 EFFICIENCY (%) 150 85 80 75 0.5 0 -0.5 -1.0 70 0 100 300 200 400 500 IOUT (mA) FIGURE 5. EFFICIENCY - 3.3V VIN TO 9V VOUT @ 1.2MHz 4 0 100 200 300 400 500 IOUT (mA) FIGURE 6. LOAD REGULATION - 3.3V VIN TO 9V VOUT @ 1.2MHz FN7333.6 October 9, 2007 EL7516 Typical Performance Curves (Continued) 1.0 LOAD REGULATION (%) EFFICIENCY (%) 90 85 80 0.6 0.2 -0.2 -0.6 -1.0 75 0 100 200 300 400 500 0 100 200 IOUT (mA) 300 400 500 IOUT (mA) FIGURE 7. EFFICIENCY - 3.3V VIN TO 9V VOUT @ 600kHz FIGURE 8. LOAD REGULATION - 3.3V VIN TO 9V VOUT @ 600kHz 0.8 95 LOAD REGULATION (%) EFFICIENCY (%) 0.6 90 85 80 0.4 0.2 1.0 -0.2 -0.4 -0.6 -0.8 75 -1 0 100 200 300 400 500 600 0 100 200 IOUT (mA) 300 400 500 600 IOUT (mA) FIGURE 9. EFFICIENCY - 5V VIN TO 12V VOUT @ 1.2MHz FIGURE 10. LOAD REGULATION - 5V VIN TO 12V VOUT @ 1.2MHz 0.8 92 LOAD REGULATION (%) EFFICIENCY (%) 0.6 90 88 86 0.4 0.2 1.0 -0.2 -0.4 -0.6 -0.8 84 -1 0 100 200 300 400 500 600 IOUT (mA) FIGURE 11. EFFICIENCY - 5V VIN TO 12V VOUT @ 600kHz 5 0 100 200 300 400 500 600 IOUT (mA) FIGURE 12. LOAD REGULATION - 5V VIN TO 12V VOUT @ 600kHz FN7333.6 October 9, 2007 EL7516 Typical Performance Curves (Continued) 0.6 95 LOAD REGULATION (%) EFFICIENCY (%) 0.4 90 85 80 0.2 0 -0.2 -0.4 -0.6 -0.8 75 -1 0 200 600 400 800 1k 0 200 400 IOUT (mA) FIGURE 13. EFFICIENCY - 5V VIN TO 9V VOUT @ 1.2MHz 0.10 VOUT = 8V IOUT = 80mA LINE REGULATION (%) VOUT=12V IOUT=80mA 0.1 1.2MHz 0 600kHz -0.1 2 4 3 6 5 0.05 1.2MHz 0 600kHz -0.05 -0.10 2.5 -0.2 4.5 3.5 FIGURE 15. LINE REGULATION FIGURE 16. LINE REGULATION 95 0.5 1.2MHz LOAD REGULATION (%) 600kHz 90 85 1.2MHz 80 75 70 10 6.5 5.5 VIN (V) VIN (V) EFFICIENCY (%) 1k FIGURE 14. LOAD REGULATION - 5V VIN TO 9V VOUT @ 1.2MHz 0.2 LINE REGULATION (%) 800 600 IOUT (mA) 0.3 0.1 -0.1 600kHz -0.3 -0.5 110 210 310 410 510 610 IOUT (mA) FIGURE 17. EFFICIENCY vs IOUT - 3.3V TO 8V 6 0 100 200 300 400 500 600 IOUT (mA) FIGURE 18. LOAD REGULATION - 3.3V TO 8V FN7333.6 October 9, 2007 EL7516 (Continued) 94 1.29 92 1.28 90 1.27 FREQUENCY (MHz) EFFICIENCY (%) Typical Performance Curves 88 1.2MHz 86 84 82 80 600kHz 78 200 600 400 800 1.25 1.24 1.23 1.22 1.21 76 0 1.26 1k 1.2 2.5 1.2k 3 IOUT (mA) 4.5 4 5 5.5 VIN (V) FIGURE 19. EFFICIENCY vs IOUT FIGURE 20. FREQUENCY (1.2MHz) vs VIN 670 93 660 91 EFFICIENCY (kHz) FREQUENCY (kHz) 3.5 650 640 630 620 89 87 85 83 610 600 2.5 81 3 3.5 4.5 4 5 5.5 0 VIN (V) 200 400 600 800 1k IOUT (mA) FIGURE 21. FREQUENCY (600kHz) vs VIN FIGURE 22. EFFICIENCY - 5V VIN TO 9V VOUT @ 600kHz LOAD REGULATION (%) 0.4 VIN = 3.3V VOUT = 12V IOUT = 50mA TO 300mA 0.2 0 200mV/DIV -0.2 -0.4 0 200 400 600 800 1k 0.1ms/DIV IOUT (mA) FIGURE 23. LOAD REGULATION - 5V VIN TO 9V VOUT @ 600kHz 7 FIGURE 24. TRANSIENT REPONSE - 600kHz FN7333.6 October 9, 2007 EL7516 Typical Performance Curves (Continued) 5 VIN = 3.3V VOUT = 12V IOUT = 50mA TO 300mA SHDN LEVEL (V) 4 200mV/DIV SHDN TURN ON SHDN TURN OFF 3 2 1 0 3 3.5 4 4.5 VIN (V) 0.1ms/DIV FIGURE 25. TRANSIENT RESPONSE - 1.2MHz 5 5.5 6 FIGURE 26. TYPICAL SHDN INPUT LEVEL vs VIN JEDEC JESD51-7 HIGH EFFECTIVE THERMAL CONDUCTIVITY TEST BOARD JEDEC JESD51-3 LOW EFFECTIVE THERMAL CONDUCTIVITY TEST BOARD 1.0 0.6 POWER DISSIPATION (W) POWER DISSIPATION (W) 0.9 870mW 0.8 0.7 M SO JA P =1 15 8 °C /W θ 0.6 0.5 0.4 0.3 0.2 0.5 486mW 0.4 θ JA = 0.3 M SO P8 20 6° C/ W 0.2 0.1 0.1 0 0 0 25 75 85 50 100 0 125 Applications Information The EL7516 is a high frequency, high efficiency boost regulator operated at constant frequency PWM mode. The boost converter stores energy from an input voltage source and delivers it to a higher output voltage. The input voltage range is 2.5V to 5.5V and the output voltage range is 5V to 18V. The switching frequency is selectable between 600KHz and 1.2MHz, allowing smaller inductors and faster transient response. An external compensation pin gives the user greater flexibility in setting output transient response and tighter load regulation. The converter soft-start characteristic can also be controlled by external CSS capacitor. The SHDN pin allows the user to completely shut-down the device. Boost Converter Operations Figure 28 shows a boost converter with all the key components. In steady state operating and continuous conduction mode where the inductor current is continuous, 8 50 75 85 100 125 AMBIENT TEMPERATURE (°C) AMBIENT TEMPERATURE (°C) FIGURE 27. PACKAGE POWER DISSIPATION vs AMBIENT TEMPERATURE 25 FIGURE 28. PACKAGE POWER DISSIPATION vs AMBIENT TEMPERATURE the boost converter operates in two cycles. During the first cycle, as shown in Figure 29, the internal power FET turns on and the Schottky diode is reverse biased and cuts off the current flow to the output. The output current is supplied from the output capacitor. The voltage across the inductor is VIN and the inductor current ramps up in a rate of VIN / L, L is the inductance. The inductance is magnetized and energy is stored in the inductor. The change in inductor current is: V IN ΔI L1 = Δt1 × --------L D Δt1 = ---------f SW D = Duty Cycle I OUT ΔV O = ---------------- × Δt 1 C OUT (EQ. 1) FN7333.6 October 9, 2007 EL7516 During the second cycle, the power FET turns off and the Schottky diode is forward biased, Figure 30. The energy stored in the inductor is pumped to the output supplying output current and charging the output capacitor. The Schottky diode side of the inductor is clamp to a Schottky diode above the output voltage, so the voltage drop across the inductor is VIN - VOUT. The change in inductor current during the second cycle is: L D VOUT VIN COUT CIN EL7516 IL ΔIL2 Δt2 V IN – V OUT ΔI L = Δt2 × -------------------------------L ΔVO 1–D Δt2 = ------------f SW (EQ. 2) For stable operation, the same amount of energy stored in the inductor must be taken out. The change in inductor current during the two cycles must be the same. ΔI1 + ΔI2 = 0 V IN 1 – D V IN – V OUT D ---------- × --------+ ------------- × -------------------------------- = 0 L f SW L f SW V OUT 1 ---------------- = ------------1–D V IN FIGURE 31. BOOST CONVERTER - CYCLE 2, POWER SWITCH OPEN Output Voltage An external feedback resistor divider is required to divide the output voltage down to the nominal 1.294V reference voltage. The current drawn by the resistor network should be limited to maintain the overall converter efficiency. The maximum value of the resistor network is limited by the feedback input bias current and the potential for noise being coupled into the feedback pin. A resistor network less than 100k is recommended. The boost converter output voltage is determined by the relationship: (EQ. 3) R 1⎞ ⎛ V OUT = V FB × ⎜ 1 + -------⎟ R 2⎠ ⎝ L (EQ. 4) D VOUT VIN COUT CIN Inductor Selection EL7516 FIGURE 29. BOOST CONVERTER L VOUT VIN COUT CIN The nominal VFB voltage is 1.294V. The inductor selection determines the output ripple voltage, transient response, output current capability, and efficiency. Its selection depends on the input voltage, output voltage, switching frequency, and maximum output current. For most applications, the inductance should be in the range of 2µH to 33µH. The inductor maximum DC current specification must be greater than the peak inductor current required by the regulator. The peak inductor current can be calculated: V IN × ( V OUT – V IN ) I OUT × V OUT I L ( PEAK ) = ------------------------------------ + 1 ⁄ 2 × ----------------------------------------------------V IN L × V OUT × FREQ EL7516 (EQ. 5) Output Capacitor IL ΔIL1 Δt1 ΔVO FIGURE 30. BOOST CONVERTER - CYCLE 1, POWER SWITCH CLOSED Low ESR capacitors should be used to minimize the output voltage ripple. Multilayer ceramic capacitors (X5R and X7R) are preferred for the output capacitors because of their lower ESR and small packages. Tantalum capacitors with higher ESR can also be used. The output ripple can be calculated as: I OUT × D ΔV O = ------------------------- + I OUT × ESR f SW × C O (EQ. 6) For noise sensitive application, a 0.1µF placed in parallel with the larger output capacitor is recommended to reduce the switching noise coupled from the LX switching node. 9 FN7333.6 October 9, 2007 EL7516 In selecting the Schottky diode, the reverse break down voltage, forward current and forward voltage drop must be considered for optimum converter performance. The diode must be rated to handle 1.5A, the current limit of the EL7516. The breakdown voltage must exceed the maximum output voltage. Low forward voltage drop, low leakage current, and fast reverse recovery will help the converter to achieve the maximum efficiency. Input Capacitor The value of the input capacitor depends on the input and output voltages, the maximum output current, the inductor value and the noise allowed to put back on the input line. For most applications, a minimum 10µF is required. For applications that run close to the maximum output current limit, input capacitor in the range of 22µF to 47µF is recommended. The EL7516 is powered from the VIN. High frequency 0.1µF by-pass cap is recommended to be close to the VIN pin to reduce supply line noise and ensure stable operation. Loop Compensation EL7516 does not use a level translator or ground-referenced threshold for the SHDN input. For different supply voltages, please refer to Figure 32 to choose the right input threshold voltages for SHDN, where VTP is about 1V. It is recommended that VIH = (VIN - VTP/2) and VIL = (VIN/4). If the consistent SHDN threshold is desired in the application, an external active level shifter must be used. The simplest circuit requires 1 NMOS and 1 resistor, as shown in Figure 33 where the gate of the NMOS is connected to supply of PWRON logic circuit, and the source of the NMOS goes to PWRON pin of the converter. SHDN INPUT THRESHOLDS Schottky Diode VIN = 3.3V VIN = 5.5V VIH, UPPER LOGIC THRESHOLD VIL, LOWER LOGIC THRESHOLD KEEP OUT 0V The EL7516 incorporates an transconductance amplifier in its feedback path to allow the user some adjustment on the transient response and better regulation. The EL7516 uses current mode control architecture, which has a fast current sense loop and a slow voltage feedback loop. The fast current feedback loop does not require any compensation. The slow voltage loop must be compensated for stable operation. The compensation network is a series RC network from COMP pin to ground. The resistor sets the high frequency integrator gain for fast transient response and the capacitor sets the integrator zero to ensure loop stability. For most applications, the compensation resistor in the range of 2k to 7.5k and the compensation capacitor in the range of 3nF to 10nF. Soft-Start The soft-start is provided by an internal 6µA current source, which charges the external CSS. The peak MOSFET current is limited by the voltage on the capacitor. This in turn controls the rising rate of the output voltage. The regulator goes through the start-up sequence as well after the SHDN pin is pulled to HI. Frequency Selection The EL7516 switching frequency can be user selected to operate at either at constant 620kHz or 1.25MHz. Connecting FSEL pin to ground sets the PWM switching frequency to 620kHz. When connect FSEL high or VDD, switching frequency is set to 1.25MHz. VIN (VIN - VTP) (VIN/2) VIN FIGURE 32. SHDN INPUT THRESHOLD vs INPUT SUPPLY VOLTAGE SUPPLY INPUT VOLTAGE 20k TO EL7516 3VD 20k PIN3 SHDN PWRON PIN OF THE CONVERTER FIGURE 33. LEVEL SHIFTER CIRCUIT Maximum Output Current The MOSFET current limit is nominally 1.5A and guaranteed 1.3A. This restricts the maximum output current IOMAX based on the following formula: I L = I L-AVG + ( 1 ⁄ 2 × ΔI L ) (EQ. 7) Shut-Down Control When the Shut-down pin is pulled down, the EL7516 is shutdown, reducing the supply current to <3µA. 10 FN7333.6 October 9, 2007 EL7516 Thermal Performance where: The EL7516 uses a fused-lead package, which has a reduced θJA of 100°C/W on a four-layer board and 115°C/W on a two-layer board. Maximizing copper around the ground pins will improve the thermal performance. IL = MOSFET current limit IL-AVG = average inductor current ΔIL = inductor ripple current V IN × [ ( V O + V DIODE ) – V IN ] ΔI L = -----------------------------------------------------------------------------L × ( V O + V DIODE ) × f S This device also has internal thermal shut-down set at around +130°C to protect the component. (EQ. 8) Layout Considerations VDIODE = Schottky diode forward voltage, typically, 0.6V fS = switching frequency, 600kHz or 1.2MHz I OUT I L-AVG = ------------1–D (EQ. 9) To achieve highest efficiency, best regulation and the most stable operation, a good printed circuit board layout is essential. It is strongly recommended that the demoboard layout be followed as closely as possible. Use the following general guidelines when laying out the print circuit board: 1. Place C4 as close to the VDD pin as possible. C4 is the supply bypass capacitor of the device. D = MOSFET turn-on ratio: V IN D = 1 – -------------------------------------------V OUT + V DIODE (EQ. 10) Table 1 gives typical maximum IOUT values for 1.2MHz switching frequency and 22µH inductor: TABLE 1. 2. Keep the C1 ground, GND pin and C2 ground as close as possible. 3. Keep the two high current paths a) from C1 through L1, to the LX pin and GND and b) from C1 through L1, D1, and C2 as short as possible. 4. High current traces should be as short and as wide as possible. VIN (V) VOUT (V) IOMAX (mA) 2.5 5 570 5. Place the feedback resistor close to the FB pin to avoid noise pickup. 2.5 9 325 6. Place the compensation network close to the COMP pin. 2.5 12 250 3.3 5 750 The demo board is a good example of layout based on these principles; it is available upon request. 3.3 9 435 Differences Between EL7516 and ISL97516 3.3 12 330 5 9 650 ISL97516 is the replacement for EL7516, and it is pin-to-pin compatible to EL7516, but there are differences between the two parts, as shown in the Table 2: 5 12 490 TABLE 2. DIFFERENCES BETWEEN EL7516 AND ISL97516 ISL97516 EL7516 Current Limit 2.0A (typical value) 1.5A (typical value) Over-Temperature Protection +150°C +130°C Logic High or Low Level Refer to Ground, Fixed. Refer to input voltage, Varying From Table 2, it shows that ISL97516 can provide more output current at the same conditions, and work in higher ambient temperature. The fixed logic level also helps reduce the system design complexity. 11 FN7333.6 October 9, 2007 EL7516 Mini SO Package Family (MSOP) 0.25 M C A B D MINI SO PACKAGE FAMILY (N/2)+1 N E MDP0043 A E1 MILLIMETERS PIN #1 I.D. 1 B (N/2) e H C SEATING PLANE 0.10 C N LEADS SYMBOL MSOP8 MSOP10 TOLERANCE NOTES A 1.10 1.10 Max. - A1 0.10 0.10 ±0.05 - A2 0.86 0.86 ±0.09 - b 0.33 0.23 +0.07/-0.08 - c 0.18 0.18 ±0.05 - D 3.00 3.00 ±0.10 1, 3 E 4.90 4.90 ±0.15 - E1 3.00 3.00 ±0.10 2, 3 e 0.65 0.50 Basic - L 0.55 0.55 ±0.15 - L1 0.95 0.95 Basic - N 8 10 Reference - 0.08 M C A B b Rev. D 2/07 NOTES: 1. Plastic or metal protrusions of 0.15mm maximum per side are not included. L1 2. Plastic interlead protrusions of 0.25mm maximum per side are not included. A 3. Dimensions “D” and “E1” are measured at Datum Plane “H”. 4. Dimensioning and tolerancing per ASME Y14.5M-1994. c SEE DETAIL "X" A2 GAUGE PLANE L A1 0.25 3° ±3° DETAIL X All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems. Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries. For information regarding Intersil Corporation and its products, see www.intersil.com 12 FN7333.6 October 9, 2007