LINER LT1374IR-5 4.5a, 500khz step-down switching regulator Datasheet

LT1374
4.5A, 500kHz Step-Down
Switching Regulator
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FEATURES
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DESCRIPTIO
The LT ®1374 is a 500kHz monolithic buck mode switching
regulator. A 4.5A switch is included on the die along with
all the necessary oscillator, control and logic circuitry. High
switching frequency allows a considerable reduction in the
size of external components. The topology is current mode
for fast transient response and good loop stability. Both
fixed output voltage and adjustable parts are available.
Constant 500kHz Switching Frequency
High Power 16-Pin TSSOP Package Available
Uses All Surface Mount Components
Inductor Size Reduced to 1.8µH
Saturating Switch Design: 0.07Ω
Effective Supply Current: 2.5mA
Shutdown Current: 20µA
Cycle-by-Cycle Current Limiting
Easily Synchronizable
A special high speed bipolar process and new design techniques achieve high efficiency at high switching frequency.
Efficiency is maintained over a wide output current range
by using the output to bias the circuitry and by utilizing a
supply boost capacitor to saturate the power switch.
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APPLICATIO S
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Portable Computers
Battery-Powered Systems
Battery Chargers
Distributed Power
The LT1374 is available in standard 7-pin DD, TO-220, fused
lead SO-8 and 16-pin exposed pad TSSOP packages. Full
cycle-by-cycle short-circuit protection and thermal shutdown are provided. Standard surface mount external parts
may be used, including the inductor and capacitors. There
is the optional function of shutdown or synchronization. A
shutdown signal reduces supply current to 20µA. Synchronization allows an external logic level signal to increase the
internal oscillator from 580kHz to 1MHz.
, LTC and LT are registered trademarks of Linear Technology Corporation.
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TYPICAL APPLICATIO
5V Buck Converter
Efficiency vs Load Current
D2
CMDSH3 OR FMMD914
INPUT
6V TO 25V
C3*
10µF TO
50µF
VIN
+
BOOST
L1**
5µH
OUTPUT**
5V, 4.25A
VSW
LT1374-5 BIAS
DEFAULT
= ON
SHDN
GND
SENSE
VC
CC
1.5nF
VOUT = 5V
VIN = 10V
L = 10µH
95
D1
MBRS330T3
+
* RIPPLE CURRENT RATING ≥ IOUT/2
** INCREASE L1 TO 10µH FOR LOAD CURRENTS ABOVE 3.5A AND TO 20µH ABOVE 4A
SEE APPLICATIONS INFORMATION
C1
100µF, 10V
SOLID
TANTALUM
1374 TA01
EFFICIENCY (%)
C2
0.27µF
100
90
85
80
75
70
0
0.5
1.0
1.5 2.0 2.5 3.0
LOAD CURRENT (A)
3.5
4.0
1374 TA02
1374fb
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LT1374
W W
W
AXI U
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ABSOLUTE
RATI GS
(Note 1)
Input Voltage
LT1374 ............................................................... 25V
LT1374HV .......................................................... 32V
BOOST Pin Voltage ................................................. 38V
BOOST Pin Above Input Voltage ............................. 15V
SHDN Pin Voltage ..................................................... 7V
BIAS Pin Voltage ...................................................... 7V
FB Pin Voltage (Adjustable Part) ............................ 3.5V
FB Pin Current (Adjustable Part) ............................ 1mA
SENSE Voltage (Fixed 5V Part) ................................. 7V
SYNC Pin Voltage ..................................................... 7V
Operating Junction Temperature Range
LT1374C ............................................... 0°C to 125° C
LT1374I ........................................... – 40°C to 125°C
Storage Temperature Range ................ – 65°C to 150°C
Lead Temperature (Soldering, 10 sec)................. 300°C
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PACKAGE/ORDER I FOR ATIO
FRONT VIEW
7
6
5
4
3
2
1
TAB
IS
GND
FB OR SENSE*
BOOST
VIN
GND
VSW
SYNC OR SHDN*
VC
R PACKAGE
7-LEAD PLASTIC DD
TJMAX = 125°C, θJA = 30°C/ W
WITH PACKAGE SOLDERED TO 0.5 SQUARE INCH
COPPER AREA OVER BACKSIDE GROUND PLANE
OR INTERNAL POWER PLANE. θJA CAN VARY
FROM 20°C/W TO > 40°C/W DEPENDING ON
MOUNTING TECHNIQUES
GND
1
16 GND
NC
2
15 VSW
VIN
3
14 VSW
VIN
4
13 SYNC
BOOST
5
12 SHDN
FB/SENSE
6
11 VC
NC
7
10 BIAS
8
ORDER
PART NUMBER
LT1374CR
LT1374CR-5
LT1374CR-SYNC
LT1374CR-5 SYNC
LT1374HVCR
LT1374IR
LT1374IR-5
LT1374IR-SYNC
LT1374IR-5 SYNC
LT1374HVIR
LT1374CS8
LT1374CS8-5
LT1374CS8-SYNC
LT1374CS8-5 SYNC
LT1374HVCS8
LT1374IS8
LT1374IS8-5
LT1374IS8-SYNC
LT1374IS8-5 SYNC
LT1374HVIS8
TOP VIEW
VIN 1
9
GND
FE16 PACKAGE
16-LEAD PLASTIC TSSOP
θJA = 40°C/ W
EXPOSED PAD SOLDERED TO
GROUND PLANE
8 VSW
SYNC
7
OR SHDN*
6 VC
BOOST 2
FB OR
3
SENSE*
FGND 4
5 BIAS
S8 PACKAGE
8-LEAD PLASTIC SO
θJA = 80°C/ W WITH FUSED (FGND)
GROUND PIN CONNECTED TO GROUND
PLANE OR LARGE LANDS
ORDER
PART NUMBER
TOP VIEW
GND
ORDER
PART NUMBER
LT1374CFE
LT1374IFE
TAB IS
GND
FRONT VIEW
7
6
5
4
3
2
1
FB OR SENSE*
BOOST
VIN
GND
VSW
SHDN
VC
T7 PACKAGE
7-LEAD PLASTIC TO-220
S8 PART MARKING
1374
1374I
13745 1374I5
1374SN 374ISN
3745SN 74I5SN
1374HV 1374HVI
ORDER
PART NUMBER
LT1374CT7
LT1374CT7-5
LT1374IT7
LT1374IT7-5
TJMAX = 125°C, θJA = 50°C/ W, θJC = 4°C/ W
*Default is the adjustable output voltage device with FB pin and shutdown function. Option -5 replaces FB with SENSE pin for fixed 5V output applications.
-SYNC replaces SHDN with SYNC pin for applications requiring synchronization. Consult LTC Marketing for parts specified with wider operating
temperature ranges.
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LT1374
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are at TJ = 25°C. VIN = 15V, VC = 1.5V, Boost = VIN + 5V, switch open, unless otherwise noted.
PARAMETER
Feedback Voltage (Adjustable)
CONDITIONS
All Conditions
●
All Conditions
●
Sense Voltage (Fixed 5V)
SENSE Pin Resistance
Reference Voltage Line Regulation
5V ≤ VIN ≤ 25V (5V ≤ VIN ≤ 32V for LT1374HV)
Feedback Input Bias Current
Error Amplifier Voltage Gain
Error Amplifier Transconductance
(Notes 2, 8)
∆I (VC) = ±10µA (Note 8)
●
●
VC Pin to Switch Current Transconductance
Error Amplifier Source Current
Error Amplifier Sink Current
VC Pin Switching Threshold
VC Pin High Clamp
Switch Current Limit
Slope Compensation (Note 9)
Switch On Resistance (Note 7)
MIN
2.39
2.36
4.94
4.90
7
200
1500
1000
VFB = 2.1V or VSENSE = 4.4V
VFB = 2.7V or VSENSE = 5.6V
Duty Cycle = 0
●
●
140
140
VC Open, VFB = 2.1V or VSENSE = 4.4V, DC ≤ 50%
DC = 80%
ISW = 4.5A
●
4.5
TYP
2.42
5.0
10
0.01
0.5
400
2000
5.3
225
225
0.9
2.1
6
0.8
0.07
●
Maximum Switch Duty Cycle
VFB = 2.1V or VSENSE = 4.4V
●
Switch Frequency
VC Set to Give 50% Duty Cycle
●
Switch Frequency Line Regulation
Frequency Shifting Threshold on FB Pin
Minimum Input Voltage (Note 3)
Minimum Boost Voltage (Note 4)
Boost Current (Note 5)
VIN Supply Current (Note 6)
BIAS Supply Current (Note 6)
Shutdown Supply Current
5V ≤ VIN ≤ 25V, (5V ≤ VIN ≤ 32V for LT1374HV)
∆f = 10kHz
90
86
460
440
●
●
0.8
●
ISW ≤ 4.5A
ISW = 1A
ISW = 4.5A
VBIAS = 5V
VBIAS = 5V
VSHDN = 0V, VIN ≤ 25V, VSW = 0V, VC Open
●
●
●
●
●
93
93
500
0
1.0
5.0
2.3
20
90
0.9
3.2
20
●
VSHDN = 0V, VIN ≤ 32V, VSW = 0V, VC Open
30
●
Lockout Threshold
Shutdown Thresholds
VC Open
VC Open Device Shutting Down
Device Starting Up
Synchronization Threshold
Synchronizing Range
SYNC Pin Input Resistance
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: Gain is measured with a VC swing equal to 200mV above the
switching threshold level to 200mV below the upper clamp level.
Note 3: Minimum input voltage is not measured directly, but is guaranteed
by other tests. It is defined as the voltage where internal bias lines are still
●
●
●
2.3
0.13
0.25
●
2.38
0.37
0.45
1.5
580
40
MAX
2.45
2.48
5.06
5.10
14
0.03
2
2700
3100
320
320
8.5
0.1
0.13
540
560
0.15
1.3
5.5
3.0
35
140
1.4
4.0
50
75
75
100
2.46
0.60
0.7
2.2
1000
UNITS
V
V
V
V
kΩ
%/V
µA
µMho
µMho
A/ V
µA
µA
V
V
A
A
Ω
Ω
%
%
kHz
kHz
%/ V
V
V
V
mA
mA
mA
mA
µA
µA
µA
µA
V
V
V
V
kHz
kΩ
regulated so that the reference voltage and oscillator frequency remain
constant. Actual minimum input voltage to maintain a regulated output will
depend on output voltage and load current. See Applications Information.
Note 4: This is the minimum voltage across the boost capacitor needed to
guarantee full saturation of the internal power switch.
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LT1374
ELECTRICAL CHARACTERISTICS
Note 5: Boost current is the current flowing into the boost pin with the pin
held 5V above input voltage. It flows only during switch on time.
Note 6: VIN supply current is the current drawn when the BIAS pin is held
at 5V and switching is disabled. If the BIAS pin is unavailable or open
circuit, the sum of VIN and BIAS supply currents will be drawn by the VIN
pin.
Note 7: Switch on resistance is calculated by dividing VIN to VSW voltage
by the forced current (4.5A). See Typical Performance Characteristics for
the graph of switch voltage at other currents.
Note 8: Transconductance and voltage gain refer to the internal amplifier
exclusive of the voltage divider. To calculate gain and transconductance,
refer to the SENSE pin on the fixed voltage parts. Divide values shown by
the ratio VOUT/2.42.
Note 9: Slope compensation is the current subtracted from the switch
current limit at 80% duty cycle. See Maximum Output Load Current in the
Applications Information section for further details.
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TYPICAL PERFOR A CE CHARACTERISTICS
Switch Peak Current Limit
Switch Voltage Drop
6.0
125°C
SWITCH PEAK CURRENT (A)
350
25°C
300
250
– 40°C
200
150
100
TYPICAL
5.5
5.0
MINIMUM
4.5
4.0
3.0
0
1
2
4
3
SWITCH CURRENT (A)
5
0
20
60
40
DUTY CYCLE (%)
Shutdown Pin Bias Current
80
100
2.410
– 50
200
AT 2.38V STANDBY THRESHOLD
(CURRENT FLOWS OUT OF PIN)
4
0
–50 –25
25
50
75
100
125
Shutdown Supply Current
25
VSHDN = 0V
STANDBY
SHUTDOWN PIN VOLTAGE (V)
300
0
1374 G03
2.40
CURRENT REQUIRED TO FORCE SHUTDOWN
(FLOWS OUT OF PIN). AFTER SHUTDOWN,
CURRENT DROPS TO A FEW µA
–25
TEMPERATURE (°C)
Standby and Shutdown Thresholds
500
CURRENT (µA)
2.415
1374 G02
1374 G18
8
2.420
3.5
50
400
2.425
INPUT SUPPLY CURRENT (µA)
SWITCH VOLTAGE (mV)
400
FEEDBACK VOLTAGE (V)
450
0
Feedback Pin Voltage
2.430
6.5
500
2.36
2.32
0.8
START-UP
0.4
20
15
10
5
SHUTDOWN
50
25
75
0
TEMPERATURE (°C)
100
125
1374 G04
0
–50 –25
0
50
100
25
75
0
JUNCTION TEMPERATURE (°C)
125
1374 G05
0
5
10
15
INPUT VOLTAGE (V)
20
25
1374 G06
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LT1374
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TYPICAL PERFOR A CE CHARACTERISTICS
Shutdown Supply Current
Error Amplifier Transconductance
VIN = 25V
50
VIN = 10V
40
30
20
3000
2000
2500
200
PHASE
GAIN (µMho)
TRANSCONDUCTANCE (µMho)
60
Error Amplifier Transconductance
2500
1500
1000
500
150
GAIN
2000
100
VC
(
)
COUT
12pF
ROUT
200k
1500
VFB 2 × 10–3
1000
ERROR AMPLIFIER EQUIVALENT CIRCUIT
50
PHASE (DEG)
INPUT SUPPLY CURRENT (µA)
70
0
10
RLOAD = 50Ω
0
50
0
75 100
25
–50 –25
JUNCTION TEMPERATURE (°C)
0
0.1
0.2
0.3
SHUTDOWN VOLTAGE (V)
0
0.4
125
1k
10k
100k
FREQUENCY (Hz)
Frequency Foldback
1374 G09
Minimum Input Voltage
with 5V Output
Switching Frequency
6.4
550
500
540
6.2
FREQUENCY (kHz)
300
200
INPUT VOLTAGE (V)
530
400
SWITCHING
FREQUENCY
520
510
500
490
480
470
100
FEEDBACK PIN
CURRENT
0.5
1.5
2.0
1.0
FEEDBACK PIN VOLTAGE (V)
5.8
5.6
MINIMUM
RUNNING
VOLTAGE
5.4
5.2
450
– 50
2.5
5.0
–25
0
25
50
75
100
1
125
10
100
LOAD CURRENT (mA)
TEMPERATURE (°C)
1000
1374 G12
1374 G11
1374 G10
Maximum Load Current
at VOUT = 10V
Maximum Load Current
at VOUT = 3.3V
Maximum Load Current
at VOUT = 5V
4.5
4.5
4.5
VOUT = 10V
MINIMUM
STARTING
VOLTAGE
6.0
460
0
0
–50
10M
1M
1374 G08
1374 G07
SWITCHING FREQUENCY (kHz) OR CURRENT (µA)
500
100
L = 20µH
L = 20µH
L = 20µH
L = 10µH
L = 10µH
L = 5µH
4.0
3.5
3.5
3.0
3.0
L = 5µH
CURRENT (A)
4.0
CURRENT (A)
CURRENT (A)
L = 10µH
4.0
L = 5µH
3.5
VOUT = 5V
VOUT = 3.3V
0
5
10
15
INPUT VOLTAGE (V)
20
25
1374 G13
3.0
0
5
10
15
INPUT VOLTAGE (V)
20
25
1374 G14
0
5
10
15
INPUT VOLTAGE (V)
20
25
1374 G15
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LT1374
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TYPICAL PERFOR A CE CHARACTERISTICS
VC Pin Shutdown Threshold
BOOST Pin Current
DUTY CYCLE = 100%
THRESHOLD VOLTAGE (V)
70
60
50
40
30
20
1.2
1.0
0.8
0
1
3
2
4
SWITCH CURRENT (A)
5
Kool Mµ
0.4
PERMALLOY
µ = 125
0.01
0.2
CORE LOSS IS
INDEPENDENT OF LOAD
CURRENT UNTIL LOAD CURRENT FALLS
LOW ENOUGH FOR CIRCUIT TO GO INTO
DISCONTINUOUS MODE
0.6
0.4
–50
2
1.2
0.8
®
10
0
4
TYPE 52
POWDERED IRON
0.1
0.12
0.08
0.04
0.001
–25
0
25
50
75
100
JUNCTION TEMPERATURE (°C)
1374 G16
125
1374 G11
CORE LOSS (% OF 5W LOAD)
80
20
12
8
VOUT = 5V, VIN = 10V, IOUT = 1A
SHUTDOWN
90
BOOST PIN CURRENT (mA)
Inductor Core Loss
1.0
1.4
CORE LOSS (W)
100
0.02
0
5
10
15
INDUCTANCE (µH)
20
25
1374 G01
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PI FU CTIO S
FB/SENSE: The feedback pin is the input to the error
amplifier which is referenced to an internal 2.42V source.
An external resistive divider is used to set the output
voltage. The fixed voltage (-5) parts have the divider
included on-chip and the FB pin is used as a SENSE pin,
connected directly to the 5V output. Three additional
functions are performed by the FB pin. When the pin
voltage drops below 1.7V, switch current limit is reduced.
Below 1.5V the external sync function is disabled. Below
1V, switching frequency is also reduced. See Feedback Pin
Function section in Applications Information for details.
BOOST: The BOOST pin is used to provide a drive voltage,
higher than the input voltage, to the internal bipolar NPN
power switch. Without this added voltage, the typical
switch voltage loss would be about 1.5V. The additional
boost voltage allows the switch to saturate and voltage
loss approximates that of a 0.07Ω FET structure. Efficiency improves from 75% for conventional bipolar designs to > 89% for these new parts.
VIN: This is the collector of the on-chip power NPN switch.
This pin powers the internal circuitry and internal regulator
when the BIAS pin is not present. At NPN switch on and off,
high dI/dt edges occur on this pin. Keep the external
bypass and catch diode close to this pin. All trace inductance on this path will create a voltage spike at switch off,
adding to the VCE voltage across the internal NPN. Both VIN
6
pins of the 16-lead TSSOP package must be shorted
together on the PC board.
GND: The GND pin connection needs consideration for
two reasons. First, it acts as the reference for the regulated
output, so load regulation will suffer if the “ground” end of
the load is not at the same voltage as the GND pin of the
IC. This condition will occur when load current or other
currents flow through metal paths between the GND pin
and the load ground point. Keep the ground path short
between the GND pin and the load and use a ground plane
when possible. The second consideration is EMI caused
by GND pin current spikes. Internal capacitance between
the VSW pin and the GND pin creates very narrow (<10ns)
current spikes in the GND pin. If the GND pin is connected
to system ground with a long metal trace, this trace may
radiate excess EMI. Keep the path between the input
bypass and the GND pin short.
VSW: The switch pin is the emitter of the on-chip power
NPN switch. This pin is driven up to the input pin voltage
during switch on time. Inductor current drives the switch
pin negative during switch off time. Negative voltage is
clamped with the external catch diode. Maximum negative
switch voltage allowed is – 0.8V. Both VSW pins of the
16-lead TSSOP package must be shorted together on the
PC board.
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LT1374
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PI FU CTIO S
SYNC: (Excludes T7 package) The sync pin is used to
synchronize the internal oscillator to an external signal. It
is directly logic compatible and can be driven with any
signal between 10% and 90% duty cycle. The synchronizing range is equal to initial operating frequency, up to
1MHz. This pin replaces SHDN on -SYNC option parts. See
Synchronizing section in Applications Information for
details.
SHDN: The shutdown pin is used to turn off the regulator
and to reduce input drain current to a few microamperes.
Actually, this pin has two separate thresholds, one at
2.38V to disable switching, and a second at 0.4V to force
complete micropower shutdown. The 2.38V threshold
functions as an accurate undervoltage lockout (UVLO).
This can be used to prevent the regulator from operating
until the input voltage has reached a predetermined level.
used for frequency compensation, but can do double duty
as a current clamp or control loop override. This pin sits
at about 1V for very light loads and 2V at maximum load.
It can be driven to ground to shut off the regulator, but if
driven high, current must be limited to 4mA.
BIAS: (SO-8 and FE16 Packages) The BIAS pin is used to
improve efficiency when operating at higher input voltages and light load current. Connecting this pin to the
regulated output voltage forces most of the internal circuitry to draw its operating current from the output voltage
rather than the input supply. This is a much more efficient
way of doing business if the input voltage is much higher
than the output. Minimum output voltage setting for this
mode of operation is 3.3V. Efficiency improvement at
VIN = 20V, VOUT = 5V, and IOUT = 25mA is over 10%.
NC: No Connect. Leave floating or solder to any node.
VC: The VC pin is the output of the error amplifier and the
input of the peak switch current comparator. It is normally
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BLOCK DIAGRA
The LT1374 is a constant frequency, current mode buck
converter. This means that there is an internal clock and
two feedback loops that control the duty cycle of the power
switch. In addition to the normal error amplifier, there is a
current sense amplifier that monitors switch current on a
cycle-by-cycle basis. A switch cycle starts with an oscillator pulse which sets the RS flip-flop to turn the switch on.
When switch current reaches a level set by the inverting
input of the comparator, the flip-flop is reset and the
switch turns off. Output voltage control is obtained by
using the output of the error amplifier to set the switch
current trip point. This technique means that the error
amplifier commands current to be delivered to the output
rather than voltage. A voltage fed system will have low
phase shift up to the resonant frequency of the inductor
and output capacitor, then an abrupt 180° shift will occur.
The current fed system will have 90° phase shift at a much
lower frequency, but will not have the additional 90° shift
until well beyond the LC resonant frequency. This makes
it much easier to frequency compensate the feedback loop
and also gives much quicker transient response.
Most of the circuitry of the LT1374 operates from an
internal 2.9V bias line. The bias regulator normally draws
power from the regulator input pin, but if the BIAS pin is
connected to an external voltage higher than 3V, bias
power will be drawn from the external source (typically the
regulated output voltage). This will improve efficiency if
the BIAS pin voltage is lower than regulator input voltage.
High switch efficiency is attained by using the BOOST pin
to provide a voltage to the switch driver which is higher
than the input voltage, allowing the switch to saturate. This
boosted voltage is generated with an external capacitor
and diode. Two comparators are connected to the shutdown pin. One has a 2.38V threshold for undervoltage
lockout and the second has a 0.4V threshold for complete
shutdown.
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LT1374
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BLOCK DIAGRA
0.01Ω
INPUT
+
BIAS*
2.9V BIAS
REGULATOR
–
CURRENT
SENSE
AMPLIFIER
VOLTAGE GAIN = 20
INTERNAL
VCC
Σ
SLOPE COMP
BOOST
0.9V
500kHz
OSCILLATOR
SYNC
S
+
SHUTDOWN
COMPARATOR
DRIVER
CIRCUITRY
RS
FLIP-FLOP
CURRENT
COMPARATOR
R
–
Q1
POWER
SWITCH
VSW
–
+
0.4V
FREQUENCY
SHIFT CIRCUIT
SHDN
3.5µA
FOLDBACK
CURRENT
LIMIT
CLAMP
+
Q2
–
LOCKOUT
COMPARATOR
ERROR
AMPLIFIER
gm = 2000µMho
VC
2.38V
FB
+
–
2.42V
GND
*BIAS PIN IS AVAILABLE ONLY ON THE S0-8 AND FE16 PACKAGES
1374 BD
Figure 1. Block Diagram
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APPLICATIO S I FOR ATIO
FEEDBACK PIN FUNCTIONS
The feedback (FB) pin on the LT1374 is used to set output
voltage and provide several overload protection features.
The first part of this section deals with selecting resistors
to set output voltage and the remaining part talks about
foldback frequency and current limiting created by the FB
pin. Please read both parts before committing to a final
design. The fixed 5V LT1374-5 has internal divider resistors and the FB pin is renamed SENSE, connected directly
to the output.
The suggested value for the output divider resistor (see
Figure 2) from FB to ground (R2) is 5k or less, and a
formula for R1 is shown below. The output voltage error
caused by ignoring the input bias current on the FB pin is
less than 0.25% with R2 = 5k. A table of standard 1%
values is shown in Table 1 for common output voltages.
Please read the following if divider resistors are increased
above the suggested values.
R1 =
(
)
R2 VOUT − 2.42
2.42
Table 1
OUTPUT
VOLTAGE
(V)
R2
(kΩ)
R1
(NEAREST 1%)
(kΩ)
% ERROR AT OUTPUT
DUE TO DISCREET 1%
RESISTOR STEPS
3
4.99
1.21
+ 0.23
3.3
4.99
1.82
+ 0.08
5
4.99
5.36
+ 0.39
6
4.99
7.32
– 0.5
8
4.99
11.5
– 0.04
10
4.99
15.8
+ 0.83
12
4.99
19.6
– 0.62
15
4.99
26.1
+ 0.52
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More Than Just Voltage Feedback
foldback operation. Again, it is nearly transparent to the
user under normal load conditions. The only loads that may
be affected are current source loads which maintain full
load current with output voltage less than 50% of final value.
In these rare situations the feedback pin can be clamped
above 1.5V with an external diode to defeat foldback current limit. Caution: clamping the feedback pin means that
frequency shifting will also be defeated, so a combination
of high input voltage and dead shorted output may cause
the LT1374 to lose control of current limit.
The feedback pin is used for more than just output voltage
sensing. It also reduces switching frequency and current
limit when output voltage is very low (see the Frequency
Foldback graph in Typical Performance Characteristics).
This is done to control power dissipation in both the IC and
in the external diode and inductor during short-circuit
conditions. A shorted output requires the switching regulator to operate at very low duty cycles, and the average
current through the diode and inductor is equal to the
short-circuit current limit of the switch (typically 6A for the
LT1374, folding back to less than 3A). Minimum switch on
time limitations would prevent the switcher from attaining
a sufficiently low duty cycle if switching frequency were
maintained at 500kHz, so frequency is reduced by about
5:1 when the feedback pin voltage drops below 1V (see
Frequency Foldback graph). This does not affect operation
with normal load conditions; one simply sees a gear shift
in switching frequency during start-up as the output
voltage rises.
The internal circuitry which forces reduced switching
frequency also causes current to flow out of the feedback
pin when output voltage is low. The equivalent circuitry is
shown in Figure 2. Q1 is completely off during normal
operation. If the FB pin falls below 1V, Q1 begins to
conduct current and reduces frequency at the rate of
approximately 5kHz/µA. To ensure adequate frequency
foldback (under worst-case short-circuit conditions), the
external divider Thevinin resistance must be low enough
to pull 150µA out of the FB pin with 0.6V on the pin (RDIV
≤ 4k). The net result is that reductions in frequency and
current limit are affected by output voltage divider impedance. Although divider impedance is not critical, caution
should be used if resistors are increased beyond the
suggested values and short-circuit conditions will occur
with high input voltage. High frequency pickup will
increase and the protection accorded by frequency and
current foldback will decrease.
In addition to lower switching frequency, the LT1374 also
operates at lower switch current limit when the feedback
pin voltage drops below 1.7V. Q2 in Figure 2 performs this
function by clamping the VC pin to a voltage less than its
normal 2.1V upper clamp level. This foldback current limit
greatly reduces power dissipation in the IC, diode and
inductor during short-circuit conditions. External synchronization is also disabled to prevent interference with
LT1374
VSW
TO FREQUENCY
SHIFTING
ERROR
AMPLIFIER
1.6V
+
R5
5k
OUTPUT
5V
–
Q1
2.4V
R1
R3
1k
R4
1k
FB
Q2
TO SYNC CIRCUIT
VC
GND
+
R2
5k
1374 F02
Figure 2. Frequency and Current Limit Foldback
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MAXIMUM OUTPUT LOAD CURRENT
Maximum load current for a buck converter is limited by
the maximum switch current rating (IP) of the LT1374.
This current rating is 4.5A up to 50% duty cycle (DC),
decreasing to 3.7A at 80% duty cycle. This is shown
graphically in Typical Performance Characteristics and as
shown in the formula below:
IP = 4.5A for DC ≤ 50%
IP = 3.21 + 5.95(DC) – 6.75(DC)2 for 50% < DC < 90%
Note that there is less load current available at the higher
input voltage because inductor ripple current increases.
This is not always the case. Certain combinations of
inductor value and input voltage range may yield lower
available load current at the lowest input voltage due to
reduced peak switch current at high duty cycles. If load
current is close to the maximum available, please check
maximum available current at both input voltage
extremes. To calculate actual peak switch current with a
given set of conditions, use:
DC = Duty cycle = VOUT/VIN
Example: with VOUT = 5V, VIN = 8V; DC = 5/8 = 0.625, and;
ISW(MAX) = 3.21 + 5.95(0.625) – 6.75(0.625)2 = 4.3A
ISW(PEAK ) = IOUT +
(
)
2(L)(f)(V )
VOUT VIN − VOUT
IN
Current rating decreases with duty cycle because the
LT1374 has internal slope compensation to prevent current
mode subharmonic switching. For more details, read Application Note 19. The LT1374 is a little unusual in this regard
because it has nonlinear slope compensation which gives
better compensation with less reduction in current limit.
For lighter loads where discontinuous operation can be
used, maximum load current is equal to:
Maximum load current would be equal to maximum
switch current for an infinitely large inductor, but with
finite inductor size, maximum load current is reduced by
one-half peak-to-peak inductor current. The following
formula assumes continuous mode operation, implying
that the term on the right is less than one-half of IP.
Example: with L = 1.2µH, VOUT = 5V, and VIN(MAX) = 15V,
IOUT(MAX) =
IP −
Continuous Mode
(V )(V − V )
2(L)(f)(V )
OUT
IN
OUT
IN
For the conditions above and L = 3.3µH,
IOUT (MAX ) = 4.3 −
(5)(8 − 5)
()
2 3.3 • 10 − 6  500 • 10 3  8
= 4.3 − 0.57 = 3.73 A
At VIN = 15V, duty cycle is 33%, so IP is just equal to a fixed
4.5A, and IOUT(MAX) is equal to:
4.5 −
(5)(15 − 5)
= 4.5 − 1 = 3.5A
10
( )
2 3.3 • 10 − 6  500 • 10 3  15
(IP) (f)(L)(VIN)
2(VOUT )(VIN − VOUT )
2
IOUT(MAX) =
Discontinuous mode
4.5)  500 • 10   1.2 • 10  (15)
(
= 1.82A
)=
2(5)(15 − 5)
2
IOUT (MAX
3
−6
The main reason for using such a tiny inductor is that it is
physically very small, but keep in mind that peak-to-peak
inductor current will be very high. This will increase output
ripple voltage. If the output capacitor has to be made larger
to reduce ripple voltage, the overall circuit could actually
wind up larger.
CHOOSING THE INDUCTOR AND OUTPUT CAPACITOR
For most applications the output inductor will fall in the
range of 3µH to 20µH. Lower values are chosen to reduce
physical size of the inductor. Higher values allow more
output current because they reduce peak current seen by
the LT1374 switch, which has a 4.5A limit. Higher values
also reduce output ripple voltage, and reduce core loss.
Graphs in the Typical Performance Characteristics section
show maximum output load current versus inductor size
and input voltage. A second graph shows core loss versus
inductor size for various core materials.
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When choosing an inductor you might have to consider
maximum load current, core and copper losses, allowable
component height, output voltage ripple, EMI, fault current in the inductor, saturation, and of course, cost. The
following procedure is suggested as a way of handling
these somewhat complicated and conflicting requirements.
1. Choose a value in microhenries from the graphs of
maximum load current and core loss. Choosing a small
inductor may result in discontinuous mode operation
at lighter loads, but the LT1374 is designed to work
well in either mode. Keep in mind that lower core loss
means higher cost, at least for closed core geometries
like toroids. The core loss graphs show both absolute
loss and percent loss for a 5W output, so actual percent
losses must be calculated for each situation.
Assume that the average inductor current is equal to
load current and decide whether or not the inductor
must withstand continuous fault conditions. If maximum load current is 0.5A, for instance, a 0.5A inductor
may not survive a continuous 4.5A overload condition.
Dead shorts will actually be more gentle on the inductor because the LT1374 has foldback current limiting.
2. Calculate peak inductor current at full load current to
ensure that the inductor will not saturate. Peak current
can be significantly higher than output current, especially with smaller inductors and lighter loads, so don’t
omit this step. Powdered iron cores are forgiving
because they saturate softly, whereas ferrite cores
saturate abruptly. Other core materials fall somewhere
in between. The following formula assumes continuous mode of operation, but it errs only slightly on the
high side for discontinuous mode, so it can be used for
all conditions.
IPEAK = IOUT +
(
)
2(f)(L)(V )
VOUT VIN − VOUT
IN
VIN = Maximum input voltage
f = Switching frequency, 500kHz
3. Decide if the design can tolerate an “open” core geometry like a rod or barrel, with high magnetic field
radiation, or whether it needs a closed core like a toroid
to prevent EMI problems. One would not want an open
core next to a magnetic storage media, for instance!
This is a tough decision because the rods or barrels are
temptingly cheap and small and there are no helpful
guidelines to calculate when the magnetic field radiation will be a problem.
4. Start shopping for an inductor (see representative
surface mount units in Table 2) which meets the
requirements of core shape, peak current (to avoid
Table 2
VENDOR/
PART NO.
VALUE
DC
(µH) (Amps)
CORE
TYPE
SERIES
CORE
RESIS- MATER- HEIGHT
TANCE(Ω)
IAL
(mm)
Coiltronics
CTX2-1
2
4.1
Tor
0.011
KMµ
4.2
CTX5-4
5
4.4
Tor
0.019
KMµ
6.4
CTX8-4
8
3.5
Tor
0.020
KMµ
6.4
CTX2-1P
2
3.4
Tor
0.014
52
4.2
CTX2-3P
2
4.6
Tor
0.012
52
4.8
CTX5-4P
5
3.3
Tor
0.027
52
6.4
CDRH125
10
4.0
SC
0.025
Fer
6
CDRH125
12
3.5
SC
0.027
Fer
6
CDRH125
15
3.3
SC
0.030
Fer
6
CDRH125
18
3.0
SC
0.034
Fer
6
DT3316-222
2.2
5
SC
0.035
Fer
5.1
DT3316-332
3.3
5
SC
0.040
Fer
5.1
DT3316-472
4.7
3
SC
0.045
Fer
5.1
PE-53650
4
4.8
Tor
0.017
Fer
9.1
PE-53651
5
5.4
Tor
0.018
Fer
9.1
Sumida
Coilcraft
Pulse
PE-53652
9
5.5
Tor
0.022
Fer
10
PE-53653
16
5.1
Tor
0.032
Fer
10
IHSM-4825
2.7
5.1
Open
0.034
Fer
5.6
IHSM-4825
4.7
4.0
Open
0.047
Fer
5.6
IHSM-5832
10
4.3
Open
0.053
Fer
7.1
IHSM-5832
15
3.5
Open
0.078
Fer
7.1
IHSM-7832
22
3.8
Open
0.054
Fer
7.1
Dale
Tor = Toroid
SC = Semi-closed geometry
Fer = Ferrite core material
52 = Type 52 powdered iron core material
KMµ = Kool Mµ
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saturation), average current (to limit heating), and fault
current (if the inductor gets too hot, wire insulation will
melt and cause turn-to-turn shorts). Keep in mind that
all good things like high efficiency, low profile, and high
temperature operation will increase cost, sometimes
dramatically. Get a quote on the cheapest unit first to
calibrate yourself on price, then ask for what you really
want.
5. After making an initial choice, consider the secondary
things like output voltage ripple, second sourcing, etc.
Use the experts in the Linear Technology’s applications department if you feel uncertain about the final
choice. They have experience with a wide range of
inductor types and can tell you about the latest developments in low profile, surface mounting, etc.
Output Capacitor
The output capacitor is normally chosen by its Effective
Series Resistance (ESR), because this is what determines
output ripple voltage. At 500kHz, any polarized capacitor
is essentially resistive. To get low ESR takes volume, so
physically smaller capacitors have high ESR. The ESR
range for typical LT1374 applications is 0.05Ω to 0.2Ω. A
typical output capacitor is an AVX type TPS, 100µF at 10V,
with a guaranteed ESR less than 0.1Ω. This is a “D” size
surface mount solid tantalum capacitor. TPS capacitors
are specially constructed and tested for low ESR, so they
give the lowest ESR for a given volume. The value in
microfarads is not particularly critical, and values from
22µF to greater than 500µF work well, but you cannot
cheat mother nature on ESR. If you find a tiny 22µF solid
tantalum capacitor, it will have high ESR, and output ripple
voltage will be terrible. Table 3 shows some typical solid
tantalum surface mount capacitors.
Table 3. Surface Mount Solid Tantalum Capacitor ESR
and Ripple Current
E Case Size
ESR (Max., Ω )
Ripple Current (A)
AVX TPS, Sprague 593D
0.1 to 0.3
0.7 to 1.1
AVX TAJ
0.7 to 0.9
0.4
0.1 to 0.3
0.7 to 1.1
0.2 (typ)
0.5 (typ)
D Case Size
AVX TPS, Sprague 593D
C Case Size
AVX TPS
Many engineers have heard that solid tantalum capacitors
are prone to failure if they undergo high surge currents.
This is historically true, and type TPS capacitors are
specially tested for surge capability, but surge ruggedness
is not a critical issue with the output capacitor. Solid
tantalum capacitors fail during very high turn-on surges,
which do not occur at the output of regulators. High
discharge surges, such as when the regulator output is
dead shorted, do not harm the capacitors.
Unlike the input capacitor, RMS ripple current in the
output capacitor is normally low enough that ripple current rating is not an issue. The current waveform is
triangular with a typical value of 200mARMS. The formula
to calculate this is:
Output Capacitor Ripple Current (RMS):
IRIPPLE(RMS) =
( )(
(L)(f)(V )
0.29 VOUT VIN − VOUT
)
IN
Ceramic Capacitors
Higher value, lower cost ceramic capacitors are now
becoming available in smaller case sizes. These are tempting for switching regulator use because of their very low
ESR. Unfortunately, the ESR is so low that it can cause
loop stability problems. Solid tantalum capacitor’s ESR
generates a loop “zero” at 5kHz to 50kHz that is instrumental in giving acceptable loop phase margin. Ceramic
capacitors remain capacitive to beyond 300kHz and usually resonate with their ESL before ESR becomes effective.
They are appropriate for input bypassing because of their
high ripple current ratings and tolerance of turn-on surges.
Linear Technology plans to issue a design note on the use
of ceramic capacitors in the near future.
OUTPUT RIPPLE VOLTAGE
Figure 3 shows a typical output ripple voltage waveform
for the LT1374. Ripple voltage is determined by the high
frequency impedance of the output capacitor, and ripple
current through the inductor. Peak-to-peak ripple current
through the inductor into the output capacitor is:
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IP-P
(V )(V − V )
=
(V )(L)(f)
OUT
IN
CATCH DIODE
OUT
IN
For high frequency switchers, the sum of ripple current
slew rates may also be relevant and can be calculated
from:
Σ
dI VIN
=
dt L
Peak-to-peak output ripple voltage is the sum of a triwave
created by peak-to-peak ripple current times ESR, and a
square wave created by parasitic inductance (ESL) and
ripple current slew rate. Capacitive reactance is assumed
to be small compared to ESR or ESL.
( )( ) ( )
VRIPPLE = IP-P ESR + ESL Σ
dI
dt
Example: with VIN =10V, VOUT = 5V, L = 10µH, ESR = 0.1Ω,
ESL = 10nH:
(5)(10 − 5)
= 0.5A
10  10 • 10 − 6  500 • 103 
dI
10
= 10 6
Σ =
dt 10 • 10 − 6
VRIPPLE = 0.5A 0.1 +  10 • 10 − 9   10 6 
= 0.05 + 0.01 = 60mVP-P
IP-P =
( )
The suggested catch diode (D1) is a 1N5821 Schottky, or
its Motorola equivalent, MBR330. It is rated at 3A average
forward current and 30V reverse voltage. Typical forward
voltage is 0.5V at 3A. The diode conducts current only
during switch off time. Peak reverse voltage is equal to
regulator input voltage. Average forward current in normal
operation can be calculated from:
ID(AVG) =
)
VIN
This formula will not yield values higher than 3A with
maximum load current of 4.25A unless the ratio of input to
output voltage exceeds 3.4:1. The only reason to consider
a larger diode is the worst-case condition of a high input
voltage and overloaded (not shorted) output. Under shortcircuit conditions, foldback current limit will reduce diode
current to less than 2.6A, but if the output is overloaded
and does not fall to less than 1/3 of nominal output voltage,
foldback will not take effect. With the overloaded condition, output current will increase to a typical value of 5.7A,
determined by peak switch current limit of 6A. With
VIN = 15V, VOUT = 4V (5V overloaded) and IOUT = 5.7A:
ID(AVG) =
( )( )
(
IOUT VIN − VOUT
(
) = 4.18A
5.7 15 − 4
15
This is safe for short periods of time, but it would be
prudent to check with the diode manufacturer if continuous operation under these conditions must be tolerated.
BOOST␣ PIN␣ CONSIDERATIONS
VOUT AT
IOUT = 1A
20mV/DIV
VOUT AT
IOUT = 50mA
INDUCTOR
CURRENT
AT IOUT = 1A
0.5A/DIV
0.5µs/DIV
1374 F03
INDUCTOR
CURRENT
AT IOUT = 50mA
Figure 3. LT1374 Ripple Voltage Waveform
For most applications, the boost components are a 0.27µF
capacitor and a FMMD914 diode. The anode is connected
to the regulated output voltage and this generates a
voltage across the boost capacitor nearly identical to the
regulated output. In certain applications, the anode may
instead be connected to the unregulated input voltage.
This could be necessary if the regulated output voltage is
very low (< 3V) or if the input voltage is less than 5V.
Efficiency is not affected by the capacitor value, but the
capacitor should have an ESR of less than 1Ω to ensure
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that it can be recharged fully under the worst-case condition of minimum input voltage. Almost any type of film or
ceramic capacitor will work fine.
CMIN =
(I
OUT / 50
( f )( V
)(V
OUT
WARNING! Peak voltage on the BOOST pin is the sum of
unregulated input voltage plus the voltage across the
boost capacitor. This normally means that peak BOOST
pin voltage is equal to input voltage plus output voltage,
but when the boost diode is connected to the regulator
input, peak BOOST pin voltage is equal to twice the input
voltage. Be sure that BOOST pin voltage does not exceed
its maximum rating.
OUT / VIN
− 3V
)
)
f = Switching frequency
VOUT = Regulated output voltage
VIN = Minimum input voltage
This formula can yield capacitor values substantially less
than 0.27µF, but it should be used with caution since it
does not take into account secondary factors such as
capacitor series resistance, capacitance shift with temperature and output overload.
For nearly all applications, a 0.27µF boost capacitor works
just fine, but for the curious, more details are provided
here. The size of the boost capacitor is determined by
switch drive current requirements. During switch on time,
drain current on the capacitor is approximately IOUT/ 50. At
peak load current of 4.25A, this gives a total drain of 85mA.
Capacitor ripple voltage is equal to the product of on time
and drain current divided by capacitor value;
∆V = (tON)(85mA/C). To keep capacitor ripple voltage to
less than 0.6V (a slightly arbitrary number) at the worstcase condition of tON = 1.8µs, the capacitor needs to be
0.27µF. Boost capacitor ripple voltage is not a critical
parameter, but if the minimum voltage across the capacitor drops to less than 3V, the power switch may not
saturate fully and efficiency will drop. An approximate
formula for absolute minimum capacitor value is:
SHUTDOWN FUNCTION AND
UNDERVOLTAGE LOCKOUT
Figure 4 shows how to add undervoltage lockout (UVLO)
to the LT1374. Typically, UVLO is used in situations where
the input supply is current limited, or has a relatively high
source resistance. A switching regulator draws constant
power from the source, so source current increases as
source voltage drops. This looks like a negative resistance
load to the source and can cause the source to current limit
or latch low under low source voltage conditions. UVLO
prevents the regulator from operating at source voltages
where these problems might occur.
RFB
LT1374
IN
INPUT
OUTPUT
VSW
2.38V
+
STANDBY
RHI
–
3.5µA
+
SHDN
+
TOTAL
SHUTDOWN
C1
RLO
0.4V
–
GND
1374 F04
Figure 4. Undervoltage Lockout
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Threshold voltage for lockout is about 2.38V, slightly less
than the internal 2.42V reference voltage. A 3.5µA bias
current flows out of the pin at threshold. This internally
generated current is used to force a default high state on
the shutdown pin if the pin is left open. When low shutdown current is not an issue, the error due to this current
can be minimized by making RLO 10k or less. If shutdown
current is an issue, RLO can be raised to 100k, but the error
due to initial bias current and changes with temperature
should be considered.
(
)
input rises back to 13.5V. ∆V is therefore 1.5V and
VIN = 12V. Let RLO = 25k.
) ]
2.38 − 25k (3.5µA)
25k (10.41)
=
= 114k
RHI =
[
(
25k 12 − 2.38 1.5 / 5 + 1 + 1.5
2.29
RFB = 114k 5 / 1.5 = 380k
(
)
RLO = 10k to 100k 25k suggested
RHI =
(
RLO VIN − 2.38V
(
)
SWITCH NODE CONSIDERATIONS
)
2.38V − RLO 3.5 µA
VIN = Minimum input voltage
Keep the connections from the resistors to the shutdown
pin short and make sure that interplane or surface capacitance to the switching nodes are minimized. If high
resistor values are used, the shutdown pin should be
bypassed with a 1000pF capacitor to prevent coupling
problems from the switch node. If hysteresis is desired in
the undervoltage lockout point, a resistor RFB can be
added to the output node. Resistor values can be calculated from:
RHI =
[
(
)
RLO VIN − 2.38 ∆V / VOUT + 1 + ∆V
( )(
(
2.38 − R2 3.5µA
RFB = RHI VOUT / ∆V
)
)
]
25k suggested for RLO
VIN = Input voltage at which switching stops as input
voltage descends to trip level
∆V = Hysteresis in input voltage level
Example: output voltage is 5V, switching is to stop if input
voltage drops below 12V and should not restart unless
For maximum efficiency, switch rise and fall times are
made as short as possible. To prevent radiation and high
frequency resonance problems, proper layout of the components connected to the switch node is essential. B field
(magnetic) radiation is minimized by keeping catch diode,
switch pin, and input bypass capacitor leads as short as
possible. E field radiation is kept low by minimizing the
length and area of all traces connected to the switch pin
and BOOST pin. A ground plane should always be used
under the switcher circuitry to prevent interplane coupling. A suggested layout for the critical components is
shown in Figure 5. Note that the feedback resistors and
compensation components are kept as far as possible
from the switch node. Also note that the high current
ground path of the catch diode and input capacitor are kept
very short and separate from the analog ground line.
The high speed switching current path is shown schematically in Figure 6. Minimum lead length in this path is
essential to ensure clean switching and low EMI. The path
including the switch, catch diode, and input capacitor is
the only one containing nanosecond rise and fall times. If
you follow this path on the PC layout, you will see that it is
irreducibly short. If you move the diode or input capacitor
away from the LT1374, get your resumé in order. The
other paths contain only some combination of DC and
500kHz triwave, so are much less critical.
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CONNECT TO
GROUND PLANE
MINIMIZE LT1374, C3, D1 LOOP
VIN
C3
D1
C5
GND
C6
VOUT
1
GND
C1
CONNECT TO
GROUND PLANE
R3
TAKE OUTPUT
DIRECTLY FROM
END OF OUTPUT
CAPACITOR
L1
U1
D2
PLACE FEEDTHROUGHS
AROUND GND PIN FOR GOOD
THERMAL CONDUCTIVITY
KEEP FB AND VC COMPONENTS
AWAY FROM HIGH FREQUENCY,
HIGH CURRENT COMPONENTS
R2
C4
KELVIN SENSE
VOUT
13745 F05a
Figure 5a. Suggested Layout (Topside Only Shown) SO-8
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SOLDER EXPOSED PAD AND
USE FEEDTHROUGHS FOR
BETTER THERMAL CONDUCTIVITY
KEEP FB AND VC
COMPONENTS AWAY
FROM HIGH FREQUENCY,
HIGH CURRENT
COMPONENTS
GND
VIN
C4
C2
R3
KELVIN
SENSE
OUTPUT
C3
D2
R2
R1
C1
VOUT
D1
C1
1374 F05b
Figure 5b. Suggested Layout (Topside Only Shown) TSSOP
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MINIMIZE LT1374 C3, D1 LOOP
CONNECT TO GND PLANE
C3
D1
VOUT
VIN
TAKE OUTPUT
DIRECTLY FROM
END OF OUTPUT
CAPACITOR
L1
PLACE FEEDTHROUGHS
AROUND GND TAB,
C3, D1 FOR GOOD
THERMAL
CONDUCTIVITY
KELVIN SENSE
VOUT
VOUT
D2
C4
C5
C6
CC2
R2
R3
CC
7 6 5 4 3 2 1
U1
RC
TAB IS GND
GND
1374 F05c
KEEP FB AND VC COMPONENTS
AWAY FROM ANY HIGH FREQUENCY,
HIGH CURRENT CMPONENTS OR PATHS
Figure 5c. Suggested Layout (Topside Only Shown) DD
SWITCH NODE
L1
5V
VIN
HIGH
FREQUENCY
CIRCULATING
PATH
LOAD
1374 F06
Figure 6. High Speed Switching Path
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PARASITIC RESONANCE
Resonance or “ringing” may sometimes be seen on the
switch node (see Figure 7). Very high frequency ringing
following switch rise time is caused by switch/diode/input
capacitor lead inductance and diode capacitance. Schottky diodes have very high “Q” junction capacitance that
can ring for many cycles when excited at high frequency.
If total lead length for the input capacitor, diode and switch
path is 1 inch, the inductance will be approximately 25nH.
At switch off, this will produce a spike across the NPN
output device in addition to the input voltage. At higher
currents this spike can be in the order of 10V to 20V or
higher with a poor layout, potentially exceeding the absolute max switch voltage. The path around switch, catch
diode and input capacitor must be kept as short as
possible to ensure reliable operation. When looking at this,
a >100MHz oscilloscope must be used, and waveforms
should be observed on the leads of the package. This
switch off spike will also cause the SW node to go below
ground. The LT1374 has special circuitry inside which
mitigates this problem, but negative voltages over 1V
lasting longer than 10ns should be avoided. Note that
100MHz oscilloscopes are barely fast enough to see the
details of the falling edge overshoot in Figure 7.
A second, much lower frequency ringing is seen during
switch off time if load current is low enough to allow the
inductor current to fall to zero during part of the switch off
time (see Figure 8). Switch and diode capacitance resonate with the inductor to form damped ringing at 1MHz to
10 MHz. This ringing is not harmful to the regulator and it
has not been shown to contribute significantly to EMI. Any
attempt to damp it with a resistive snubber will degrade
efficiency.
INPUT BYPASSING AND VOLTAGE RANGE
Input Bypass Capacitor
Step-down converters draw current from the input supply
in pulses. The average height of these pulses is equal to
load current, and the duty cycle is equal to VOUT/ VIN. Rise
and fall time of the current is very fast. A local bypass
capacitor across the input supply is necessary to ensure
proper operation of the regulator and minimize the ripple
RISE AND FALL
WAVEFORMS ARE
SUPERIMPOSED
(PULSE WIDTH IS
NOT 120ns)
5V/DIV
20ns/DIV
1374 F07
Figure 7. Switch Node Resonance
5V/DIV
SWITCH NODE
VOLTAGE
INDUCTOR
CURRENT
100mA/DIV
20ns/DIV
1375/76 F11
0.5µs/DIV
1374 F08
Figure 8. Discontinuous Mode Ringing
current fed back into the input supply. The capacitor also
forces switching current to flow in a tight local loop,
minimizing EMI.
Do not cheat on the ripple current rating of the Input
bypass capacitor, but also don’t get hung up on the value
in microfarads. The input capacitor is intended to absorb
all the switching current ripple, which can have an RMS
value as high as one half of load current. Ripple current
ratings on the capacitor must be observed to ensure
reliable operation. In many cases it is necessary to parallel
two capacitors to obtain the required ripple rating. Both
capacitors must be of the same value and manufacturer to
guarantee power sharing. The actual value of the capacitor
in microfarads is not particularly important because at
500kHz, any value above 5µF is essentially resistive. RMS
ripple current rating is the critical parameter. Actual RMS
current can be calculated from:
(
)
IRIPPLE(RMS) = IOUT VOUT VIN − VOUT / VIN
2
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The term inside the radical has a maximum value of 0.5
when input voltage is twice output, and stays near 0.5 for
a relatively wide range of input voltages. It is common
practice therefore to simply use the worst-case value and
assume that RMS ripple current is one half of load current.
At maximum output current of 4.5A for the LT1374, the
input bypass capacitor should be rated at 2.25A ripple
current. Note however, that there are many secondary
considerations in choosing the final ripple current rating.
These include ambient temperature, average versus peak
load current, equipment operating schedule, and required
product lifetime. For more details, see Application Notes
19 and 46, and Design Note 95.
Larger capacitors may be necessary when the input voltage is very close to the minimum specified on the data
sheet. Small voltage dips during switch on time are not
normally a problem, but at very low input voltage they may
cause erratic operation because the input voltage drops
below the minimum specification. Problems can also
occur if the input-to-output voltage differential is near
minimum. The amplitude of these dips is normally a
function of capacitor ESR and ESL because the capacitive
reactance is small compared to these terms. ESR tends to
be the dominate term and is inversely related to physical
capacitor size within a given capacitor type.
SYNCHRONIZING (Available as -SYNC Option)
Input Capacitor Type
Some caution must be used when selecting the type of
capacitor used at the input to regulators. Aluminum
electrolytics are lowest cost, but are physically large to
achieve adequate ripple current rating, and size constraints (especially height), may preclude their use.
Ceramic capacitors are now available in larger values, and
their high ripple current and voltage rating make them
ideal for input bypassing. Cost is fairly high and footprint
may also be somewhat large. Solid tantalum capacitors
would be a good choice, except that they have a history of
occasional spectacular failures when they are subjected to
large current surges during power-up. The capacitors can
short and then burn with a brilliant white light and lots of
nasty smoke. This phenomenon occurs in only a small
percentage of units, but it has led some OEM companies
to forbid their use in high surge applications. The input
bypass capacitor of regulators can see these high surges
when a battery or high capacitance source is connected.
Several manufacturers have developed a line of solid
tantalum capacitors specially tested for surge capability
(AVX TPS series for instance, see Table 3), but even these
units may fail if the input voltage surge approaches the
maximum voltage rating of the capacitor. AVX recommends derating capacitor voltage by 2:1 for high surge
applications. The highest voltage rating is 50V, so 25V
may be a practical upper limit when using solid tantalum
capacitors for input bypassing.
The LT1374-SYNC has the SHDN pin replaced with a
SYNC pin, which is used to synchronize the internal
oscillator to an external signal. The SYNC input must pass
from a logic level low, through the maximum synchronization threshold with a duty cycle between 10% and 90%.
The input can be driven directly from a logic level output.
The synchronizing range is equal to initial operating frequency up to 1MHz. This means that minimum practical
sync frequency is equal to the worst-case high selfoscillating frequency (550kHz), not the typical operating
frequency of 500kHz. Caution should be used when synchronizing above 700kHz because at higher sync frequencies the amplitude of the internal slope compensation
used to prevent subharmonic switching is reduced. This
type of subharmonic switching only occurs at input voltages less than twice output voltage. Higher inductor
values will tend to eliminate this problem. See Frequency
Compensation section for a discussion of an entirely
different cause of subharmonic switching before assuming that the cause is insufficient slope compensation.
Application Note 19 has more details on the theory of slope
compensation.
At power-up, when VC is being clamped by the FB pin (see
Figure 2, Q2), the sync function is disabled. This allows the
frequency foldback to operate in the shorted output condition. During normal operation, switching frequency is
controlled by the internal oscillator until the FB pin reaches
1.5V, after which the SYNC pin becomes operational. If no
synchronization is required, this pin should be connected
to ground.
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THERMAL CALCULATIONS
Power dissipation in the LT1374 chip comes from four
sources: switch DC loss, switch AC loss, boost circuit
current, and input quiescent current. The following formulas show how to calculate each of these losses. These
formulas assume continuous mode operation, so they
should not be used for calculating efficiency at light load
currents.
Switch loss:
PSW =
( ) (VOUT ) + 24ns(I )(V )(f)
OUT IN
VIN
RSW IOUT
PBOOST =
(
VOUT IOUT / 50
TJ = TA + θJA (PTOT)
With the TSSOP16 package (θJA = 40°C/W), at an ambient
temperature of 50°C,
TJ = 50 + 40 (0.87) = 85°C
2
For the DD package with a good copper plane under the
device, thermal resistance will be about 30°C/W. For the
conditions above:
Boost current loss:
2
Thermal resistance for LT1374 package is influenced by
the presence of internal or backside planes. With a full
plane under the 16-lead TSSOP package, thermal resistance will be about 40°C/W. To calculate die temperature,
use the proper thermal resistance number for the desired
package and add in worst-case ambient temperature:
TJ = 50 + 30 (0.87) = 76°C
)
Die temperature is highest at low input voltage, so use
lowest continuous input operating voltage for thermal
calculations.
VIN
Quiescent current loss:
(
)
PQ = VIN 0.001 + VOUT
(
)
FREQUENCY COMPENSATION
2

 VOUT  0.002


0.005 +
VIN
(
)
RSW = Switch resistance (≈ 0.07)
24ns = Equivalent switch current/voltage overlap time
f = Switch frequency
Example: with VIN = 10V, VOUT = 5V and IOUT = 3A:
(0.07)(3) (5) +  24 • 10  (3)(10) 500 • 10 
=




10
2
PSW
−9
3
= 0.32 + 0.36 = 0.68W
(5) (3 / 50) = 0.15W
=
2
PBOOST
10
(5) (0.002) = 0.04W
= 10(0.001) + 5(0.005) +
2
PQ
10
Total power dissipation is 0.68 + 0.15 + 0.04 = 0.87W.
Loop frequency compensation of switching regulators
can be a rather complicated problem because the reactive
components used to achieve high efficiency also introduce multiple poles into the feedback loop. The inductor
and output capacitor on a conventional step-down converter actually form a resonant tank circuit that can exhibit
peaking and a rapid 180° phase shift at the resonant
frequency. By contrast, the LT1374 uses a “current mode”
architecture to help alleviate phase shift created by the
inductor. The basic connections are shown in Figure 9.
Figure 10 shows a Bode plot of the phase and gain of the
power section of the LT1374, measured from the VC pin to
the output. Gain is set by the 5.3A/V transconductance of
the LT1374 power section and the effective complex
impedance from output to ground. Gain rolls off smoothly
above the 600Hz pole frequency set by the 100µF output
capacitor. Phase drop is limited to about 70°. Phase
recovers and gain levels off at the zero frequency (≈16kHz)
set by capacitor ESR (0.1Ω).
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Error amplifier transconductance phase and gain are
shown in Figure 11. The error amplifier can be modeled
as a transconductance of 2000µMho, with an output
impedance of 200kΩ in parallel with 12pF. In all practical
applications, the compensation network from VC pin to
ground has a much lower impedance than the output
impedance of the amplifier at frequencies above 500Hz.
This means that the error amplifier characteristics themselves do not contribute excess phase shift to the loop, and
the phase/gain characteristics of the error amplifier section are completely controlled by the external compensation network.
In Figure 12, full loop phase/gain characteristics are
shown with a compensation capacitor of 1.5nF, giving the
error amplifier a pole at 530Hz, with phase rolling off to 90°
and staying there. The overall loop has a gain of 74dB at
low frequency, rolling off to unity-gain at 100kHz. Phase
shows a two-pole characteristic until the ESR of the output
capacitor brings it back above 10kHz. Phase margin is
about 75° at unity-gain.
Analog experts will note that around 4.4kHz, phase dips
very close to the zero phase margin line. This is typical of
switching regulators, especially those that operate over a
wide range of loads. This region of low phase is not a
problem as long as it does not occur near unity-gain. In
practice, the variability of output capacitor ESR tends to
dominate all other effects with respect to loop response.
Variations in ESR will cause unity-gain to move around,
but at the same time phase moves with it so that adequate
phase margin is maintained over a very wide range of ESR
(≥ ±3:1).
LT1374
3000
VSW
150
ESR
2.42V
+
GAIN
2000
1500
–
+
C1
100
(
)
ROUT
200k
VFB 2 × 10–3
1000
R2
VC
COUT
12pF
ERROR AMPLIFIER EQUIVALENT CIRCUIT
1k
10k
100k
FREQUENCY (Hz)
Figure 9. Model for Loop Response
1374 F11
20
40
0
–40
PHASE
–20
–80
100
1k
10k
FREQUENCY (Hz)
100k
–120
1M
1374 F10
Figure 10. Response from VC Pin to Output
200
GAIN
60
150
40
100
PHASE
20
50
VIN = 10V
VOUT = 5V, IOUT = 2A
COUT = 100µF, 10V, AVX TPS
CC = 1.5nF, RC = 0, L = 10µH
0
–20
10
100
1k
10k
FREQUENCY (Hz)
LOOP PHASE (DEG)
0
80
LOOP GAIN (dB)
GAIN
VIN = 10V
VOUT = 5V
IOUT = 2A
Figure 11. Error Amplifier Gain and Phase
PHASE: VC PIN TO OUTPUT (DEG)
GAIN: VC PIN TO OUTPUT (dB)
40
–50
10M
1M
1374 F09
10
0
RLOAD = 50Ω
500
100
CC
–40
50
PHASE (DEG)
FB
RC
CF
PHASE
2500
R1
VC
GND
200
OUTPUT
ERROR
AMPLIFIER
GAIN (µMho)
CURRENT MODE
POWER STAGE
gm = 5.3A/V
0
100k
–50
1M
1374 F12
Figure 12. Overall Loop Characteristics
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What About a Resistor in the Compensation Network?
It is common practice in switching regulator design to
add a “zero” to the error amplifier compensation to
increase loop phase margin. This zero is created in the
external network in the form of a resistor (RC) in series
with the compensation capacitor. Increasing the size of
this resistor generally creates better and better loop
stability, but there are two limitations on its value. First,
the combination of output capacitor ESR and a large
value for RC may cause loop gain to stop rolling off
altogether, creating a gain margin problem. An approximate formula for RC where gain margin falls to zero is:
(
) (G )(G )(ESR)(2.42)
VOUT
R C Loop Gain = 1 =
MP
MA
GMP = Transconductance of power stage = 5.3A/V
GMA = Error amplifier transconductance = 2(10–3)
ESR = Output capacitor ESR
2.42 = Reference voltage
With VOUT = 5V and ESR = 0.03Ω, a value of 6.5k for RC
would yield zero gain margin, so this represents an upper
limit. There is a second limitation however which has
nothing to do with theoretical small signal dynamics.
This resistor sets high frequency gain of the error amplifier, including the gain at the switching frequency. If
switching frequency gain is high enough, output ripple
voltage will appear at the VC pin with enough amplitude
to muck up proper operation of the regulator. In the
marginal case, subharmonic switching occurs, as evidenced by alternating pulse widths seen at the switch
node. In more severe cases, the regulator squeals or
hisses audibly even though the output voltage is still
roughly correct. None of this will show on a theoretical
Bode plot because Bode is an amplitude insensitive
analysis. Tests have shown that if ripple voltage on the VC
is held to less than 100mVP-P, the LT1374 will be well
behaved. The formula below will give an estimate of VC
ripple voltage when RC is added to the loop, assuming
that RC is large compared to the reactance of CC at
500kHz.
VC(RIPPLE ) =
(R )(G )(V − V )(ESR)(2.4)
(V )(L)(f)
C
MA
IN
OUT
IN
GMA = Error amplifier transconductance (2000µMho)
If a computer simulation of the LT1374 showed that a
series compensation resistor of 3k gave best overall loop
response, with adequate gain margin, the resulting VC pin
ripple voltage with VIN = 10V, VOUT = 5V, ESR = 0.1Ω,
L = 10µH, would be:
3k ) 2 • 10  (10 − 5)(0.1)(2.4)
(
= 0.144V
)=
 10 • 10   500 • 10 
10
( )


−3
VC (RIPPLE
−6
3
This ripple voltage is high enough to possibly create
subharmonic switching. In most situations a compromise
value (< 2k in this case) for the resistor gives acceptable
phase margin and no subharmonic problems. In other
cases, the resistor may have to be larger to get acceptable
phase response, and some means must be used to control
ripple voltage at the VC pin. The suggested way to do this
is to add a capacitor (CF) in parallel with the RC /CC network
on the VC pin. Pole frequency for this capacitor is typically
set at one-fifth of switching frequency so that it provides
significant attenuation of switching ripple, but does not
add unacceptable phase shift at loop unity-gain frequency.
With RC = 3k,
CF =
5
(2π)(f)(R )
C
=
5
( )
2π  500 • 103  3k
= 531pF
How Do I Test Loop Stability?
The “standard” compensation for LT1374 is a 1.5nF
capacitor for CC, with RC = 0. While this compensation will
work for most applications, the “optimum” value for loop
compensation components depends, to various extent, on
parameters which are not well controlled. These include
inductor value (±30% due to production tolerance, load
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current and ripple current variations), output capacitance
(±20% to ±50% due to production tolerance, temperature, aging and changes at the load), output capacitor ESR
(±200% due to production tolerance, temperature and
aging), and finally, DC input voltage and output load
current . This makes it important for the designer to check
out the final design to ensure that it is “robust” and tolerant
of all these variations.
After verifying that the setup is working correctly, I start
varying load current and input voltage to see if I can find
any combination that makes the transient response look
suspiciously “ringy.” This procedure may lead to an adjustment for best loop stability or faster loop transient
response. Nearly always you will find that loop response
looks better if you add in several kΩ for RC. Do this only
if necessary, because as explained before, RC above 1k
may require the addition of CF to control VC pin ripple.
If everything looks OK, I use a heat gun and cold spray on
the circuit (especially the output capacitor) to bring out
any temperature-dependent characteristics.
I check switching regulator loop stability by pulse loading
the regulator output while observing transient response at
the output, using the circuit shown in Figure 13. The
regulator loop is “hit” with a small transient AC load
current at a relatively low frequency, 50Hz to 1kHz. This
causes the output to jump a few millivolts, then settle back
to the original value, as shown in Figure 14. A well behaved
loop will settle back cleanly, whereas a loop with poor
phase or gain margin will “ring” as it settles. The number
of rings indicates the degree of stability, and the frequency
of the ringing shows the approximate unity-gain frequency of the loop. Amplitude of the signal is not particularly important, as long as the amplitude is not so high that
the loop behaves nonlinearly.
Keep in mind that this procedure does not take initial
component tolerance into account. You should see fairly
clean response under all load and line conditions to ensure
that component variations will not cause problems. One
note here: according to Murphy, the component most
VOUT AT
IOUT = 500mA
BEFORE FILTER
VOUT AT
IOUT = 500mA
AFTER FILTER
VOUT AT
IOUT = 50mA
AFTER FILTER
LOAD PULSE
THROUGH 50Ω
f ≈ 780Hz
10mV/DIV
The output of the regulator contains both the desired low
frequency transient information and a reasonable amount
of high frequency (500kHz) ripple. The ripple makes it
difficult to observe the small transient, so a two-pole,
100kHz filter has been added. This filter is not particularly
critical; even if it attenuated the transient signal slightly,
this wouldn’t matter because amplitude is not critical.
5A/DIV
0.2ms/DIV
1374 F14
Figure 14. Loop Stability Check
RIPPLE FILTER
SWITCHING
REGULATOR
ADJUSTABLE
INPUT SUPPLY
ADJUSTABLE
DC LOAD
470Ω
+
100µF TO
1000µF
3300pF
TO X1
OSCILLOSCOPE
PROBE
4.7k
330pF
50Ω
TO
OSCILLOSCOPE
SYNC
100Hz TO 1kHz
100mV TO 1VP-P
1374 F13
Figure 13. Loop Stability Test Circuit
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likely to be changed in production is the output capacitor,
because that is the component most likely to have manufacturer variations (in ESR) large enough to cause problems. It would be a wise move to lock down the sources of
the output capacitor in production.
A possible exception to the “clean response” rule is at very
light loads, as evidenced in Figure 14 with ILOAD = 50mA.
Switching regulators tend to have dramatic shifts in loop
response at very light loads, mostly because the inductor
current becomes discontinuous. One common result is very
slow but stable characteristics. A second possibility is low
phase margin, as evidenced by ringing at the output with
transients. The good news is that the low phase margin at
light loads is not particularly sensitive to component variation, so if it looks reasonable under a transient test, it will
probably not be a problem in production. Note that frequency of the light load ringing may vary with component
tolerance but phase margin generally hangs in there.
POSITIVE-TO-NEGATIVE CONVERTER
The circuit in Figure 15 is a classic positive-to-negative
topology using a grounded inductor. It differs from the
standard approach in the way the IC chip derives its
feedback signal, however, because the LT1374 accepts
only positive feedback signals, the ground pin must be tied
to the regulated negative output. A resistor divider to
ground or, in this case, the sense pin, then provides the
proper feedback voltage for the chip.
D1
CMDSH-3
INPUT
5.5V TO
20V
C3
10µF TO
50µF
BOOST
VIN
C2
0.27µF
L1*
5µH
VSW
LT1374-5
+
GND
SENSE
VC
CC
RC
+
D2
MBRS330T3
C1
100µF
10V TANT
×2
* INCREASE L1 TO 10µH OR 20µH FOR HIGHER CURRENT APPLICATIONS.
SEE APPLICATIONS INFORMATION
** MAXIMUM LOAD CURRENT DEPENDS ON MINIMUM INPUT VOLTAGE
AND INDUCTOR SIZE. SEE APPLICATIONS INFORMATION
OUTPUT**
–5V, 1.8A
1374 F15
Inverting regulators differ from buck regulators in the
basic switching network. Current is delivered to the output
as square waves with a peak-to-peak amplitude much
greater than load current. This means that maximum load
current will be significantly less than the LT1374’s 4.5A
maximum switch current, even with large inductor values.
The buck converter in comparison, delivers current to the
output as a triangular wave superimposed on a DC level
equal to load current, and load current can approach 4.5A
with large inductors. Output ripple voltage for the positiveto-negative converter will be much higher than a buck
converter. Ripple current in the output capacitor will also
be much higher. The following equations can be used to
calculate operating conditions for the positive-to-negative
converter.
Maximum load current:
( )(
) (V )(V − 0.35 )
)( )( ) OUT IN
)(VOUT + VF)


VIN VOUT

IP −

2 VOUT + VIN f L 


IMAX =
VOUT + VIN − 0.35
(
(
IP = Maximum rated switch current
VIN = Minimum input voltage
VOUT = Output voltage
VF = Catch diode forward voltage
0.35 = Switch voltage drop at 4.5A
Example: with VIN(MIN) = 5.5V, VOUT = 5V, L = 10µH,
VF = 0.5V, IP = 4.5A: IMAX = 2A. Note that this equation does
not take into account that maximum rated switch current
(IP) on the LT1374 is reduced slightly for duty cycles
above 50%. If duty cycle is expected to exceed 50% (input
voltage less than output voltage), use the actual IP value
from the Electrical Characteristics table.
Operating duty cycle:
DC =
VOUT + VF
VIN − 0.3 + VOUT + VF
(This formula uses an average value for switch loss, so it
may be several percent in error.)
Figure 15. Positive-to-Negative Converter
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With the conditions above:
DC =
5 + 0.5
= 51%
5.5 − 0.3 + 5 + 0.5
This duty cycle is close enough to 50% that IP can be
assumed to be 4.5A.
OUTPUT DIVIDER
If the adjustable part is used, the resistor connected to
VOUT (R2) should be set to approximately 5k. R1 is
calculated from:
R1 =
(
)
R2 VOUT – 2.42
2.42
INDUCTOR VALUE
lowest value of inductance that can be used, but in some
cases (lower output load currents) it may give a value that
creates unnecessarily high output ripple voltage. A compromise value is often chosen that reduces output ripple.
As you can see from the graph, large inductors will not
give arbitrarily low ripple, but small inductors can give
high ripple.
The difficulty in calculating the minimum inductor size
needed is that you must first know whether the switcher
will be in continuous or discontinuous mode at the critical
point where switch current is 4.5A. The first step is to use
the following formula to calculate the load current where
the switcher must use continuous mode. If your load
current is less than this, use the discontinuous mode
formula to calculate minimum inductor needed. If load
current is higher, use the continuous mode formula.
Output current where continuous mode is needed:
Unlike buck converters, positive-to-negative converters
cannot use large inductor values to reduce output ripple
voltage. At 500kHz, values larger than 25µH make almost
no change in output ripple. The graph in Figure 16 shows
peak-to-peak output ripple voltage for a 5V to – 5V converter versus inductor value. The criteria for choosing the
inductor is therefore typically based on ensuring that peak
switch current rating is not exceeded. This gives the
(V ) (I )
4(V + V )(V + V
2
IN
ICONT =
IN
OUT + VF
IN
)
Minimum inductor discontinuous mode:
L MIN =
( )( )
(f)(I )
2 VOUT IOUT
2
P
250
OUTPUT RIPPLE VOLTAGE (mVP-P)
OUT
2
P
5V TO – 5V CONVERTER
OUTPUT CAPACITOR’S
ESR = 0.05Ω
200
Minimum inductor continuous mode:
(V )(V )
IN
L MIN =
150
ILOAD = 1A
( )(
2 f VIN + VOUT
100
)
ILOAD = 0.25A
50
OUT
(


V +V
I − I  1 + OUT F
 P OUT 
VIN


)


For the example above, with maximum load current of 1A:
0
0
5
15
10
INDUCTOR SIZE (µH)
20
(5.5) (4.5)
= 1.15 A
4 (5.5 + 5)(5.5 + 5 + 0.5 )
2
1374 F16
Figure 16. Ripple Voltage on Positive-to-Negative Converter
ICONT =
2
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26
LT1374
U
W
U
U
APPLICATIONS INFORMATION
This says that discontinuous mode can be used and the
minimum inductor needed is found from:
L MIN =
( )( )
 500 • 10  4.5

( )
25 1
3
2
= 1µH
In practice, the inductor should be increased by about 30%
over the calculated minimum to handle losses and variations in value. This suggests a minimum inductor of 1.3µH
for this application, but looking at the ripple voltage chart
shows that output ripple voltage could be reduced by a factor of two by using a 15µH inductor. There is no rule of thumb
here to make a final decision. If modest ripple is needed and
the larger inductor does the trick, go for it. If ripple is noncritical use the smaller inductor. If ripple is extremely critical, a second filter may have to be added in any case, and
the lower value of inductance can be used. Keep in mind
that the output capacitor is the other critical factor in determining output ripple voltage. Ripple shown on the graph
(Figure 16) is with two parallel capacitor’s ESR of 0.1Ω. This
is reasonable for AVX type TPS “D” or “E” size surface mount
solid tantalum capacitors, but the final capacitor chosen
must be looked at carefully for ESR characteristics.
Ripple Current in the Input and Output Capacitors
Positive-to-negative converters have high ripple current in
both the input and output capacitors. For long capacitor
lifetime, the RMS value of this current must be less than
the high frequency ripple current rating of the capacitor.
The following formula will give an approximate value for
RMS ripple current. This formula assumes continuous
mode and large inductor value. Small inductors will give
somewhat higher ripple current, especially in discontinuous mode. The exact formulas are very complex and
appear in Application Note 44, pages 30 and 31. For our
purposes here I have simply added a fudge factor (ff). The
value for ff is about 1.2 for higher load currents and
L ≥10µH. It increases to about 2.0 for smaller inductors at
lower load currents.
( )( )
Capacitor IRMS = ff IOUT
VOUT
VIN
ff = Fudge factor (1.2 to 2.0)
Diode Current
Average diode current is equal to load current. Peak diode
current will be considerably higher.
Peak diode current:
Continuous Mode =
IOUT
(VIN + VOUT ) + (VIN)(VOUT )
VIN
2(L)( f)( VIN + VOUT )
Discontinuous Mode =
( )( )
(L)(f)
2 IOUT VOUT
Keep in mind that during start-up and output overloads,
average diode current may be much higher than with
normal loads. Care should be used if diodes rated less than
3A are used, especially if continuous overload conditions
must be tolerated.
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27
LT1374
U
W
U
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APPLICATIONS INFORMATION
FE Package
16-Lead Plastic TSSOP (4.4mm)
(Reference LTC DWG # 05-08-1663,
Exposed Pad Variation BB)
4.95 – 5.05*
(.196 – .204)
3.8
(.149)
16 1514 13 12 1110
6.60 ±0.10
9
EXPOSED
PAD HEAT SINK
ON BOTTOM OF
PACKAGE
4.50 ±0.10
3.0
6.25 – 6.50
(.118) (.246 – .256)
0.45 ±0.05
1.05 ±0.10
0.65 BSC
1 2 3 4 5 6 7 8
RECOMMENDED SOLDER PAD
1.15
(.0453)
MAX
4.30 – 4.48*
(.169 – .176)
0° – 8°
0.105 – 0.180
(.0041 – .0071)
0.50 – 0.70
(.020 – .028)
0.65
(.0256)
BSC
0.195 – 0.30
(.0077 – .0118)
0.05 – 0.15
(.002 – .006)
FE16 TSSOP 1101
NOTE:
1. CONTROLLING DIMENSION: MILLIMETERS
MILLIMETERS
2. DIMENSIONS ARE IN
(INCHES)
3. DRAWING NOT TO SCALE
*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.150mm (.006") PER SIDE
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28
LT1374
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PACKAGE DESCRIPTION
R Package
7-Lead Plastic DD Pak
(LTC DWG # 05-08-1462)
0.256
(6.502)
0.060
(1.524)
0.060
(1.524)
TYP
0.390 – 0.415
(9.906 – 10.541)
0.165 – 0.180
(4.191 – 4.572)
15° TYP
0.060
(1.524)
0.183
(4.648)
0.059
(1.499)
TYP
0.330 – 0.370
(8.382 – 9.398)
BOTTOM VIEW OF DD PAK
HATCHED AREA IS SOLDER PLATED
COPPER HEAT SINK
(
+0.008
0.004 –0.004
+0.203
0.102 –0.102
)
0.095 – 0.115
(2.413 – 2.921)
0.075
(1.905)
0.300
(7.620)
0.045 – 0.055
(1.143 – 1.397)
(
+0.012
0.143 –0.020
+0.305
3.632 –0.508
)
0.040 – 0.060
(1.016 – 1.524)
0.026 – 0.036
(0.660 – 0.914)
0.013 – 0.023
(0.330 – 0.584)
0.050 ± 0.012
(1.270 ± 0.305)
R (DD7) 0396
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29
LT1374
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PACKAGE DESCRIPTION
S8 Package
8-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
0.189 – 0.197*
(4.801 – 5.004)
8
7
6
5
0.150 – 0.157**
(3.810 – 3.988)
0.228 – 0.244
(5.791 – 6.197)
1
0.010 – 0.020
× 45°
(0.254 – 0.508)
0.008 – 0.010
(0.203 – 0.254)
0.053 – 0.069
(1.346 – 1.752)
0°– 8° TYP
0.016 – 0.050
0.406 – 1.270
0.014 – 0.019
(0.355 – 0.483)
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
2
3
4
0.004 – 0.010
(0.101 – 0.254)
0.050
(1.270)
TYP
SO8 0996
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30
LT1374
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PACKAGE DESCRIPTION
T7 Package
7-Lead Plastic TO-220 (Standard)
(LTC DWG # 05-08-1422)
0.390 – 0.415
(9.906 – 10.541)
0.165 – 0.180
(4.191 – 4.572)
0.147 – 0.155
(3.734 – 3.937)
DIA
0.045 – 0.055
(1.143 – 1.397)
0.230 – 0.270
(5.842 – 6.858)
0.460 – 0.500
(11.684 – 12.700)
0.570 – 0.620
(14.478 – 15.748)
0.330 – 0.370
(8.382 – 9.398)
0.620
(15.75)
TYP
0.700 – 0.728
(17.780 – 18.491)
0.152 – 0.202
0.260 – 0.320 (3.860 – 5.130)
(6.604 – 8.128)
0.040 – 0.060
(1.016 – 1.524)
0.095 – 0.115
(2.413 – 2.921)
0.013 – 0.023
(0.330 – 0.584)
0.026 – 0.036
(0.660 – 0.914)
0.135 – 0.165
(3.429 – 4.191)
0.155 – 0.195
(3.937 – 4.953)
T7 (TO-220) (FORMED) 1197
1374fb
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
31
LT1374
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TYPICAL APPLICATION
Dual Output SEPIC␣ Converter
losses. C4 provides a low impedance path to maintain an
equal voltage swing in L1B, improving regulation. In a
flyback converter, during switch on time, all the converter’s
energy is stored in L1A only, since no current flows in L1B.
At switch off, energy is transferred by magnetic coupling
into L1B, powering the – 5V rail. C4 pulls L1B positive
during switch on time, causing current to flow, and energy
to build in L1B and C4. At switch off, the energy stored in
both L1B and C4 supply the – 5V rail. This reduces the
current in L1A and changes L1B current waveform from
square to triangular. For details on this circuit see Design
Note 100.
The circuit in Figure 17 generates both positive and
negative 5V outputs with a single piece of magnetics. The
two inductors shown are actually just two windings on a
standard BH Electronics inductor. The topology for the 5V
output is a standard buck converter. The – 5V topology
would be a simple flyback winding coupled to the buck
converter if C4 were not present. C4 creates the SEPIC
(Single-Ended Primary Inductance Converter) topology
which improves regulation and reduces ripple current in
L1. Without C4, the voltage swing on L1B compared to
L1A would vary due to relative loading and coupling
INPUT
6V TO 25V
VIN
BOOST
C2
0.27µF
+
L1A*
6.8µH
SENSE
VC
C3
22µF
35V TANT
OUTPUT
5V
VSW
LT1374-5 BIAS
SHDN
GND
D2
CMDSH-3
RC
470Ω
CC
0.01µF
+
C1**
100µF
10V TANT
+
C5**
100µF
10V TANT
D1
MBRD340
GND
* L1 IS A SINGLE CORE WITH TWO WINDINGS
BH ELECTRONICS #501-0726
** TOKIN IE475ZY5U-C304
† IF LOAD CAN GO TO ZERO, AN OPTIONAL
PRELOAD OF 1k TO 5k MAY BE USED TO
IMPROVE LOAD REGULATION
C4**
4.7nF
+
L1B* D3
MBRD340
OUTPUT
–5V†
1374 F17
Figure 17. Dual Output SEPIC Converter
RELATED PARTS
PART NUMBER
DESCRIPTION
LT1074/LT1076
Step-Down Switching Regulators
LTC®1148
High Efficiency Synchronous Step-Down Switching Regulator
LTC1149
High Efficiency Synchronous Step-Down Switching Regulator
LTC1174
High Efficiency Step-Down and Inverting DC/DC Converter
LT1176
Step-Down Switching Regulator
LT1370
High Efficiency DC/DC Converter
LT1371
High Efficiency DC/DC Converter
LT1372/LT1377
500kHz and 1MHz High Efficiency 1.5A Switching Regulators
LTC1735
High Efficiency Step-Down Converter
LT1765
3A Step-Down Switching Regulator
LT1766
1.5A Step-Down Switching Regulator
LT1767
1.5A Step-Down Switching Regulator
Burst Mode is a registered trademark of Linear Technology Corporation.
COMMENTS
40V Input, 100kHz, 5A and 2A
External FET Switches
External FET Switches
0.5A, 150kHz Burst Mode® Operation
PDIP LT1076
42V, 6A, 500kHz Switch
35V, 3A, 500kHz Switch
Boost Topology
External Switches, Very High Efficiency
1.25MHz, 3A, 25V Input, SO-8 and TSSOP16 Packages
200kHz, 1.5A, 60V Input, SO-8 and GN16 Packages
1.25MHz, 1.5A, 25V Input, MS8 Package
1374fb
32
Linear Technology Corporation
LT/TP 0102 1.5K REV B • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
 LINEAR TECHNOLOGY CORPORATION 1998