LINER LT3971 38v, 1.2a, 2mhz step-down regulator with 2.8 quiescent current Datasheet

LT3971
38V, 1.2A, 2MHz
Step-Down Regulator with
2.8µA Quiescent Current
FEATURES
DESCRIPTION
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The LT®3971 is an adjustable frequency monolithic buck
switching regulator that accepts a wide input voltage range
up to 38V. Low quiescent current design consumes only
2.8μA of supply current while regulating with no load. Low
ripple Burst Mode operation maintains high efficiency at
low output currents while keeping the output ripple below
15mV in a typical application. An internally compensated
current mode topology is used for fast transient response
and good loop stability. A high efficiency 0.33Ω switch
is included on the die along with a boost Schottky diode
and the necessary oscillator, control and logic circuitry.
An accurate 1V threshold enable pin can be used to shut
down the LT3971, reducing the input supply current to
700nA. A capacitor on the SS pin provides a controlled
inrush current (soft-start). A power good flag signals
when VOUT reaches 91% of the programmed output voltage. The LT3971 is available in small 10-pin MSOP and
3mm × 3mm DFN packages with exposed pads for low
thermal resistance.
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Ultralow Quiescent Current:
2.8μA IQ Regulating 12VIN to 3.3VOUT
Low Ripple Burst Mode® Operation:
Output Ripple < 15mVP-P
Wide Input Voltage Range: 4.3V to 38V
1.2A Maximum Output Current
Adjustable Switching Frequency: 200kHz to 2MHz
Synchronizable Between 250kHz to 2MHz
Fast Transient Response
Accurate 1V Enable Pin Threshold
Low Shutdown Current: IQ = 700nA
Power Good Flag
Soft-Start Capability
Internal Compensation
Saturating Switch Design: 0.33Ω On-Resistance
Output Voltage: 1.19V to 30V
Small Thermally Enhanced 10-Pin MSOP Package
and (3mm × 3mm) DFN Packages
APPLICATIONS
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L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks
of Linear Technology Corporation. All other trademarks are the property of their respective
owners.
Automotive Battery Regulation
Power for Portable Products
Industrial Supplies
TYPICAL APPLICATION
3.3V Step Down Converter
No Load Supply Current
VIN
4.5V TO 38V
4.0
VIN
EN
0.47μF
PG
SS
4.7μF
3.5
BOOST
4.7μH
SW
LT3971
VOUT
3.3V
1.2A
RT
BD
10pF
1.78M
49.9k
SYNC
GND
INPUT CURRENT (μA)
OFF ON
3.0
2.5
2.0
1.5
FB
22μF
1M
3480 TA01
1.0
0
10
20
30
INPUT VOLTAGE (V)
40
3971 TA01b
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1
LT3971
ABSOLUTE MAXIMUM RATINGS
(Note 1)
VIN, EN Voltage .........................................................38V
BOOST Pin Voltage ...................................................55V
BOOST Pin Above SW Pin.........................................30V
FB, RT, SYNC, SS Voltage ...........................................6V
PG, BD Voltage .........................................................30V
Boost Diode Current....................................................1A
Operating Junction Temperature Range (Note 2)
LT3971E ............................................. –40°C to 125°C
LT3971I .............................................. –40°C to 125°C
Storage Temperature Range .............. –65°C to 150°C
Lead Temperature (Soldering, 10 sec)
(MSE Only) ....................................................... 300°C
PIN CONFIGURATION
TOP VIEW
TOP VIEW
10 SYNC
BD
1
BOOST
2
SW
3
VIN
4
7 SS
EN
5
6 FB
11
GND
BD
BOOST
SW
VIN
EN
9 PG
8 RT
DD PACKAGE
10-LEAD (3mm s 3mm) PLASTIC DFN
θJA = 45°C, θJC = 10°C/W
EXPOSED PAD (PIN 11) IS GND, MUST BE SOLDERED TO PCB
1
2
3
4
5
11
GND
10
9
8
7
6
SYNC
PG
RT
SS
FB
MSE PACKAGE
10-LEAD PLASTIC MSOP
θJA = 45°C, θJC = 10°C/W
EXPOSED PAD (PIN 11) IS GND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT3971EDD#PBF
LT3971EDD#TRPBF
LFJF
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
LT3971IDD#PBF
LT3971IDD#TRPBF
LFJF
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
LT3971EMSE#PBF
LT3971EMSE#TRPBF
LTFJG
10-Lead Plastic MSOP
–40°C to 125°C
LT3971IMSE#PBF
LT3971IMSE#TRPBF
LTFJG
10-Lead Plastic MSOP
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
3971f
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LT3971
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, VEN = 12V, VBD = 3.3V unless otherwise noted. (Note 2)
PARAMETER
CONDITIONS
Quiescent Current from VIN
FB Pin Current
MIN
l
Minimum Input Voltage
VEN Low
VEN High, VSYNC Low
VEN High, VSYNC Low
l
VFB = 1.19V
l
Feedback Voltage
l
FB Voltage Line Regulation
4.3V < VIN < 40V
Switching Frequency
RT = 11k
RT = 35.7k
RT = 255k
1.175
1.165
1.6
0.8
160
TYP
MAX
4
4.3
V
0.7
1.7
1.2
2.7
4.5
μA
μA
μA
0.1
12
nA
1.19
1.19
1.205
1.215
V
V
0.0002
0.01
%/V
2
1
200
2.4
1.2
240
MHz
MHz
kHz
Minimum Switch On Time
80
Minimum Switch Off Time
110
150
2.4
3
Switch Current Limit
Switch VCESAT
1.8
ISW = 1A
ns
330
Switch Leakage Current
UNITS
0.02
ns
A
mV
1
μA
Boost Schottky Forward Voltage
ISH = 100mA
770
Boost Schottky Reverse Leakage
VREVERSE = 12V
0.02
1
1.4
1.8
V
20
28
mA
1.01
1.07
Minimum Boost Voltage (Note 3)
VIN = 5V
BOOST Pin Current
ISW = 1A, VBOOST = 15V
EN Voltage Threshold
EN Rising
l
l
0.95
mV
μA
V
EN Voltage Hysteresis
30
EN Pin Current
0.2
20
nA
100
140
mV
PG Threshold Offset from VFB
VFB Rising
60
PG Hysteresis
20
PG Leakage
VPG = 3V
PG Sink Current
VPG = 0.4V
SYNC Threshold
0.02
l
300
570
0.6
0.8
SYNC Pin Current
SS Source Current
mV
mV
1
μA
1.0
0.1
VSS = 1V
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LT3971E is guaranteed to meet performance specifications
from 0°C to 125°C junction temperature. Specifications over the –40°C
to 125°C operating junction temperature range are assured by design,
0.6
1
μA
V
nA
1.6
μA
characterization, and correlation with statistical process controls. The
LT3971I is guaranteed over the full –40°C to 125°C operating junction
temperature range. High junction temperatures degrade operating
lifetimes. Operating lifetime is derated at junction temperatures greater
than 125°C.
Note 3: This is the minimum voltage across the boost capacitor needed to
guarantee full saturation of the switch.
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3
LT3971
TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency, VOUT = 5V
Efficiency, VOUT = 3.3V
100
VIN = 12V
90
80
VIN = 12V
70
VIN = 36V
VIN = 24V
60
50
70
VIN = 36V
60
EFFICIENCY (%)
80
EFFICIENCY (%)
EFFICIENCY (%)
Efficiency, VOUT = 5V
100
100
90
VIN = 24V
50
FRONT PAGE APPLICATION
90 VOUT = 5V
R1 = 1M
80
R2 = 309k
VIN = 12V
70
60
VIN = 24V
VIN = 36V
50
40
30
40
FRONT PAGE APPLICATION
= 5V
V
30 OUT
R1 = 1M
R2 = 309k
20
0
0.2
0.4
0.6
0.8
LOAD CURRENT (A)
40
20
30
1
20
1.2
10
FRONT PAGE APPLICATION
0
0.2
0.4
0.6
0.8
LOAD CURRENT (A)
3971 G01
0
0.01
1.2
0.1
1
10
100
LOAD CURRENT (mA)
1000
3971 G03
No Load Supply Current
100
FRONT PAGE APPLICATION
80
1
3971 G02
Efficiency, VOUT = 3.3V
90
TA = 25°C, unless otherwise noted.
No Load Supply Current
4.0
DIODES, INC.
DFLS2100
VIN = 12V
FRONT PAGE APPLICATION
VOUT = 3.3V
3.5
VIN = 24V
50
VIN = 36V
40
30
INPUT CURRENT (μA)
60
INPUT CURRENT (μA)
EFFICIENCY (%)
70
10
20
3.0
2.5
2.0
1.5
10
0.1
1
10
100
LOAD CURRENT (mA)
1
–55
1000
1.0
–25
5
35
65
95
TEMPERATURE (°C)
3.0
1.200
2.5
LOAD CURRENT (A)
1.205
1.195
1.190
1.185
1.180
5
35
65
95
TEMPERATURE (°C)
125
155
3971 G07
10
20
30
INPUT VOLTAGE (V)
Maximum Load Current
FRONT PAGE APPLICATION
VOUT = 3.3V
FRONT PAGE APPLICATION
VOUT = 5V
2.0
TYPICAL
2.0
MINIMUM
1.5
1.0
0
40
3971 G06
2.5
TYPICAL
1.5
MINIMUM
1.0
0.5
0.5
–25
0
Maximum Load Current
Feedback Voltage
FEEDBACK VOLTAGE (V)
155
3971 G05
3971 G04
1.175
–55
125
LOAD CURRENT (A)
0
0.01
5
10
15
20
25
30
INPUT VOLTAGE (V)
35
40
3971 G08
0
5
10
25
30
15
20
INPUT VOLTAGE (V)
35
40
3971 G09
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LT3971
TYPICAL PERFORMANCE CHARACTERISTICS
Load Regulation
Switching Frequency
0.30
Switch Current Limit
3.0
1000
0.25
0.15
SWITCH CURRENT LIMIT (A)
950
0.20
900
FREQUENCY (kHz)
LOAD REGULATION (%)
TA = 25°C, unless otherwise noted.
0.10
0.05
0
–0.05
–0.10
–0.15
850
800
750
700
–0.20
600
–55
1200
2.0
1.5
1.0
0.5
650
FRONT PAGE APPLICATION
–0.25 REFERENCED FROM V
OUT AT 0.5A LOAD
–0.30
0
200
400
600
800 1000
LOAD CURRENT (mA)
2.5
0
–25
5
35
65
95
TEMPERATURE (°C)
125
3971 G10
155
0
20
40
60
DUTY CYCLE (%)
Switch Current Limit
100
3971 G12
3971 G11
Boost Pin Current
Switch VCESAT
2.5
80
600
30
500
25
2.2
BOOST PIN CURRENT (mA)
2.3
400
VCESAT (mV)
SWITCH CURRENT LIMIT (A)
2.4
2.1
2.0
1.9
300
200
1.8
1.7
100
20
15
10
5
1.6
125
0
155
0
250
500
750 1000 1250
SWITCH CURRENT (mA)
3971 G13
400
800
350
SWITCH ON/OFF TIME (ns)
SWITCHING FREQUENCY (kHz)
900
600
500
400
300
200
250
500
750 1000 1250
SWITCH CURRENT (mA)
Soft-Start
2.5
300
MIN TOFF 1A LOAD
250
200
1500
3971 G15
Minimum Switch On-Time/
Switch Off-Time
700
0
3971 G14
Frequency Foldback
MIN TOFF 0.5A LOAD
150
100
MIN TON
2.0
1.5
1.0
0.5
50
100
0
0
1500
SWITCH CURRENT LIMIT (A)
DUTY CYCLE = 30%
1.5
–55 –25
5
35
65
95
TEMPERATURE (°C)
0
0.2
0.4
0.8
0.6
FB PIN VOLTAGE (V)
1
1.2
3971 G16
0
–55
0
–25
35
95
5
65
TEMPERATURE (°C)
125
155
3971 G17
0
0.25 0.5 0.75 1 1.25 1.5 1.75
SS PIN VOLTAGE (V)
2
3971 G18
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LT3971
TYPICAL PERFORMANCE CHARACTERISTICS
Minimum Input Voltage
TA = 25°C, unless otherwise noted.
EN Threshold
Minimum Input Voltage
5.0
6.4
FRONT PAGE APPLICATION
4.8 VOUT = 3.3V
1.05
FRONT PAGE APPLICATION
VOUT = 5V
6.2
1.04
4.2
TO START
4.0
3.8
TO RUN
3.6
6.0
TO START
5.8
5.6
5.4
3.4
3.0
0
200
400
800 1000
600
LOAD CURRENT (mA)
5.0
1200
1.4
94
1.0
0.8
0.6
0.4
0.2
0
200
400
800 1000
600
LOAD CURRENT (mA)
1200
0.99
250
500
750 1000 1250
BOOST DIODE CURRENT (mA)
1500
93
0.95
–55
125
155
3971 G21
90
89
88
87
IL
500mA/DIV
85
–55
–25
5
35
65
95
TEMPERATURE (°C)
125
10μs/DIV
FRONT PAGE APPLICATION
VIN = 12V, VOUT = 3.3V
COUT = 47μF
155
3971 G23
Switching Waveforms;
Burst Mode Operation
3971 G24
Switching Waveforms; Full
Frequency Continuous Operation
VSW
5V/DIV
VSW
5V/DIV
3971 G25
5
35
65
95
TEMPERATURE (°C)
91
Transient Load Response,
Load Current Stepped from
0.5A to 1A
IL
500mA/
DIV
–25
VOUT
100mV/DIV
92
3971 G22
VOUT
100mV/
DIV
FALLING THRESHOLD
0.98
Transient Load Response,
Load Current Stepped from 25mA
(Burst Mode Operation) to 525mA
86
10μs/DIV
FRONT PAGE APPLICATION
VIN = 12V, VOUT = 3.3V
COUT = 47μF
1.00
Power Good Threshold
95
THRESHOLD VOLTAGE (%)
BOOST DIODE VF (V)
Boost Diode Forward Voltage
1.2
RISING THRESHOLD
1.01
3971 G20
1.6
0
1.02
0.96
3971 G19
0
1.03
0.97
TO RUN
5.2
3.2
THRESHOLD VOLTAGE (V)
4.4
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
4.6
IL
500mA/DIV
IL
500mA/DIV
VOUT
20mV/DIV
VOUT
20mV/DIV
5μs/DIV
FRONT PAGE APPLICATION
VIN = 12V, VOUT = 3.3V
ILOAD = 10mA
COUT = 22μF
3971 G26
1μs/DIV
FRONT PAGE APPLICATION
VIN = 12V, VOUT = 3.3V
ILOAD = 1A
COUT = 22μF
3971 G27
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LT3971
PIN FUNCTIONS
BD (Pin 1): This pin connects to the anode of the boost
diode. The BD pin is normally connected to the output.
BOOST (Pin 2): This pin is used to provide a drive voltage, higher than the input voltage, to the internal bipolar
NPN power switch.
SW (Pin 3): The SW pin is the output of an internal power
switch. Connect this pin to the inductor, catch diode, and
boost capacitor.
VIN (Pin 4): The VIN pin supplies current to the LT3971’s
internal circuitry and to the internal power switch. This
pin must be locally bypassed.
EN (Pin 5): The part is in shutdown when this pin is low
and active when this pin is high. The hysteretic threshold
voltage is 1.005V going up and 0.975V going down. The EN
threshold is only accurate when VIN is above 4.3V. If VIN is
lower than 4.2V, ground EN to place the part in shutdown.
Tie to VIN if shutdown feature is not used.
FB (Pin 6): The LT3971 regulates the FB pin to 1.19V.
Connect the feedback resistor divider tap to this pin. Also,
connect a phase lead capacitor between FB and VOUT.
Typically this capacitor is 10pF.
SS (Pin 7): A capacitor is tied between SS and ground to
slowly ramp up the peak current limit of the LT3971 on
start-up. The soft-start capacitor is only actively discharged
when EN is low. The SS pin is released when the EN pin
goes high. Float this pin to disable soft-start. For applications with input voltages above 25V, add a 100k resistor
in series with the soft-start capacitor.
RT (Pin 8): A resistor is tied between RT and ground to
set the switching frequency.
PG (Pin 9): The PG pin is the open-drain output of an
internal comparator. PGOOD remains low until the FB pin
is within 9% of the final regulation voltage. PGOOD is valid
when the LT3971 is enabled and VIN is above 4.3V.
SYNC (Pin 10): This is the external clock synchronization
input. Ground this pin for low ripple Burst Mode operation
at low output loads. Tie to a clock source for synchronization, which will include pulse-skipping at low output
loads. When in pulse-skipping mode, quiescent current
increases to 1.5mA.
GND (Exposed Pad Pin 11): Ground. The exposed pad
must be soldered to PCB.
3971f
7
LT3971
BLOCK DIAGRAM
VIN
C1
INTERNAL 1.19V REF
1V
EN
RT
–
+
VIN
+
–
3
SHDN
BD
SLOPE COMP
SWITCH
LATCH
BOOST
R
OSCILLATOR
200kHz TO 2MHz
RT
C3
Q
S
L1
VOUT
SW
Burst Mode
DETECT
SYNC
PG
ERROR AMP
+
–
+
–
1.09V
VC
D1
C2
VC CLAMP
1μA
SS
SHDN
GND
FB
R2
R3
C4
R1
3991 BD
C5
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LT3971
OPERATION
The LT3971 is a constant frequency, current mode stepdown regulator. An oscillator, with frequency set by RT,
sets an RS flip-flop, turning on the internal power switch.
An amplifier and comparator monitor the current flowing
between the VIN and SW pins, turning the switch off when
this current reaches a level determined by the voltage at
VC (see Block Diagram). An error amplifier measures the
output voltage through an external resistor divider tied to
the FB pin and servos the VC node. If the error amplifier’s
output increases, more current is delivered to the output;
if it decreases, less current is delivered. An active clamp
on the VC node provides current limit. The VC node is
also clamped by the voltage on the SS pin; soft-start is
implemented by generating a voltage ramp at the SS pin
using an external capacitor.
If the EN pin is low, the LT3971 is shut down and draws
700nA from the input. When the EN pin exceeds 1V, the
switching regulator will become active.
The switch driver operates from either VIN or from the
BOOST pin. An external capacitor is used to generate a
voltage at the BOOST pin that is higher than the input
supply. This allows the driver to fully saturate the internal
bipolar NPN power switch for efficient operation.
To further optimize efficiency, the LT3971 automatically
switches to Burst Mode operation in light load situations.
Between bursts, all circuitry associated with controlling
the output switch is shut down, reducing the input supply
current to 1.7μA. In a typical application, 2.8μA will be consumed from the supply when regulating with no load.
The oscillator reduces the LT3971’s operating frequency
when the voltage at the FB pin is low. This frequency
foldback helps to control the output current during startup and overload.
The LT3971 contains a power good comparator which
trips when the FB pin is at 91% of its regulated value. The
PG output is an open-drain transistor that is off when the
output is in regulation, allowing an external resistor to pull
the PG pin high. Power good is valid when the LT3971 is
enabled and VIN is above 4.2V.
APPLICATIONS INFORMATION
Achieving Ultralow Quiescent Current
1000
FRONT PAGE APPLICATION
VIN = 12V
VOUT = 3.3V
Figure 1. Switching Frequency in Burst Mode Operation
As the output load decreases, the frequency of single current pulses decreases (see Figure 1) and the percentage
of time the LT3971 is in sleep mode increases, resulting
in much higher light load efficiency. By maximizing the
time between pulses, the converter quiescent current
gets closer to the 1.7μA ideal. Therefore, to optimize the
quiescent current performance at light loads, the current
in the feedback resistor divider and the reverse current
in the catch diode must be minimized, as these appear
to the output as load currents. Use the largest possible
SWITCHING FREQUENCY (kHz)
To enhance efficiency at light loads, the LT3971 operates
in low ripple Burst Mode, which keeps the output capacitor
charged to the desired output voltage while minimizing
the input quiescent current. In Burst Mode operation the
LT3971 delivers single pulses of current to the output capacitor followed by sleep periods where the output power
is supplied by the output capacitor. When in sleep mode
the LT3971 consumes 1.7μA, but when it turns on all the
circuitry to deliver a current pulse, the LT3971 consumes
1.5mA of input current in addition to the switch current.
Therefore, the total quiescent current will be greater than
1.7μA when regulating.
800
600
400
200
0
0
20
40
60
80
LOAD CURRENT (mA)
100
120
3971 F01
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9
LT3971
APPLICATIONS INFORMATION
feedback resistors and a low leakage Schottky catch diode
in applications utilizing the ultralow quiescent current
performance of the LT3971. The feedback resistors should
preferably be on the order of MΩ and the Schottky catch
diode should have less than 1μA of typical reverse leakage at room temperature. These two considerations are
reiterated in the FB Resistor Network and Catch Diode
Selection sections.
To ensure proper Burst Mode operation, the SYNC pin
must be grounded. When synchronized with an external
clock, the LT3971 will pulse skip at light loads. The quiescent current will significantly increase to 1.5mA in light
load situations when synchronized with an external clock.
Holding the SYNC pin high yields no advantages in terms
of output ripple or minimum load to full frequency, so is
not recommended.
It is important to note that another way to decrease the
pulse frequency is to increase the magnitude of each
single current pulse. However, this increases the output
voltage ripple because each cycle delivers more power to
the output capacitor. The magnitude of the current pulses
was selected to ensure less than 15mV of output ripple in
a typical application. See Figure 2.
FB Resistor Network
The output voltage is programmed with a resistor divider
between the output and the FB pin. Choose the resistor
values according to:
⎞
⎛ V
R1= R2 ⎜ OUT − 1⎟
⎝ 1.19 V ⎠
Reference designators refer to the Block Diagram. 1%
resistors are recommended to maintain output voltage
accuracy.
VSW
5V/DIV
IL
500mA/DIV
VOUT
20mV/DIV
5μs/DIV
FRONT PAGE APPLICATION
VIN = 12V
VOUT = 3.3V
ILOAD = 10mA
3971 F02
Figure 2. Burst Mode Operation
While in Burst Mode operation, the burst frequency and
the charge delivered with each pulse will not change with
output capacitance. Therefore, the output voltage ripple will
be inversely proportional to the output capacitance. In a
typical application with a 22μF output capacitor, the output
ripple is about 10mV, and with a 47μF output capacitor
the output ripple is about 5mV. The output voltage ripple
can continue to be decreased by increasing the output
capacitance.
At higher output loads (above 92mA for the front page
application) the LT3971 will be running at the frequency
programmed by the RT resistor, and will be operating in
standard PWM mode. The transition between PWM and low
ripple Burst Mode operation will exhibit slight frequency
jitter, but will not disturb the output voltage.
The total resistance of the FB resistor divider should be
selected to be as large as possible to enhance low current
performance. The resistor divider generates a small load
on the output, which should be minimized to optimize the
low supply current at light loads.
When using large FB resistors, a 10pF phase lead capacitor
should be connected from VOUT to FB.
Setting the Switching Frequency
The LT3971 uses a constant frequency PWM architecture
that can be programmed to switch from 200kHz to 2MHz
by using a resistor tied from the RT pin to ground. A table
showing the necessary RT value for a desired switching
frequency is in Table 1.
Table 1. Switching Frequency vs RT Value
SWITCHING FREQUENCY (MHz)
RT VALUE (kΩ)
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
1.8
2.0
255
118
71.5
49.9
35.7
28.0
22.1
17.4
14.0
11.0
3971f
10
LT3971
APPLICATIONS INFORMATION
Operating Frequency Tradeoffs
Input Voltage Range
Selection of the operating frequency is a tradeoff between
efficiency, component size, minimum dropout voltage, and
maximum input voltage. The advantage of high frequency
operation is that smaller inductor and capacitor values may
be used. The disadvantages are lower efficiency, lower
maximum input voltage, and higher dropout voltage. The
highest acceptable switching frequency (fSW(MAX)) for a
given application can be calculated as follows:
The minimum input voltage is determined by either the
LT3971’s minimum operating voltage of 4.3V or by its
maximum duty cycle (see equation in Operating Frequency
Tradeoffs section). The minimum input voltage due to
duty cycle is:
fSW(MAX ) =
VOUT + VD
tON(MIN)( VIN − VSW + VD )
where VIN is the typical input voltage, VOUT is the output
voltage, VD is the catch diode drop (~0.5V), and VSW is
the internal switch drop (~0.5V at max load). This equation
shows that slower switching frequency is necessary to
safely accommodate high VIN/VOUT ratio. Also, as shown
in the Input Voltage Range section, lower frequency allows
a lower dropout voltage. The input voltage range depends
on the switching frequency because the LT3971 switch has
finite minimum on and off times. The minimum switch on
and off times are strong functions of temperature. Use
the typical minimum on and off curves to design for an
application’s maximum temperature, while adding about
30% for part-to-part variation. The minimum and maximum
duty cycles that can be achieved taking minimum on and
off times into account are:
DCMIN = fSW tON(MIN)
DCMAX = 1− fSW tOFF(MIN)
where fSW is the switching frequency, the tON(MIN) is the
minimum switch on-time, and the tOFF(MIN) is the minimum
switch off-time. These equations show that duty cycle range
increases when switching frequency is decreased.
A good choice of switching frequency should allow adequate input voltage range (see Input Voltage Range section) and keep the inductor and capacitor values small.
VIN(MIN) =
VOUT + VD
−V +V
1− fSW tOFF(MIN) D SW
where VIN(MIN) is the minimum input voltage, VOUT is
the output voltage, VD is the catch diode drop (~0.5V),
VSW is the internal switch drop (~0.5V at max load), fSW
is the switching frequency (set by RT), and tOFF(MIN) is
the minimum switch off-time. Note that higher switching frequency will increase the minimum input voltage.
If a lower dropout voltage is desired, a lower switching
frequency should be used.
The maximum input voltage for LT3971 applications
depends on switching frequency, the Absolute Maximum
Ratings of the VIN and BOOST pins, and the operating
mode. For a given application where the switching frequency and the output voltage are already selected, the
maximum input voltage (VIN(OP-MAX)) that guarantees
optimum output voltage ripple for that application can be
found by applying the following equation:
VIN(OP-MAX ) =
VOUT + VD
–V +V
fSW • tON(MIN) D SW
where tON(MIN) is the minimum switch on-time. Note that
a higher switching frequency will decrease the maximum
operating input voltage. Conversely, a lower switching
frequency will be necessary to achieve normal operation
at higher input voltages.
The circuit will tolerate inputs above the maximum operating input voltage and up to the Absolute Maximum
Ratings of the VIN and BOOST pins, regardless of chosen
switching frequency. However, during such transients
3971f
11
LT3971
APPLICATIONS INFORMATION
where VIN is higher than VIN(OP-MAX), the LT3971 will enter
pulse-skipping operation where some switching pulses are
skipped to maintain output regulation. The output voltage
ripple and inductor current ripple will be higher than in
typical operation. Do not overload when VIN is greater
than VIN(OP-MAX).
Table 2. Inductor Vendors
VENDOR
URL
PART SERIES
TYPE
Murata
www.murata.com
LQH55D
Open
TDK
www.componenttdk.com SLF7045
SLF10145
Shielded
Shielded
Toko
www.toko.com
D62CB
D63CB
D73C
D75F
Shielded
Shielded
Shielded
Open
Coilcraft
www.coilcraft.com
MSS7341
MSS1038
Shielded
Shielded
Sumida
www.sumida.com
CR54
CDRH74
CDRH6D38
CR75
Open
Shielded
Shielded
Open
Inductor Selection and Maximum Output Current
A good first choice for the inductor value is:
L=
VOUT + VD
fSW
where fSW is the switching frequency in MHz, VOUT is the
output voltage, VD is the catch diode drop (~0.5V) and L
is the inductor value in μH.
The inductor’s RMS current rating must be greater than the
maximum load current and its saturation current should be
about 30% higher. For robust operation in fault conditions
(start-up or short-circuit) and high input voltage (>30V),
the saturation current should be above 2.8A. To keep the
efficiency high, the series resistance (DCR) should be less
than 0.1Ω, and the core material should be intended for
high frequency applications. Table 2 lists several vendors
and suitable types.
The inductor value must be sufficient to supply the desired
maximum output current (IOUT(MAX)), which is a function
of the switch current limit (ILIM) and the ripple current.
IOUT(MAX ) = ILIM –
ΔIL
2
The LT3971 limits its peak switch current in order to protect
itself and the system from overload faults. The LT3971’s
switch current limit (ILIM) is at least 2.4A at low duty cycles
and decreases linearly to 1.75A at DC = 0.8.
When the switch is off, the potential across the inductor
is the output voltage plus the catch diode drop. This gives
the peak-to-peak ripple current in the inductor:
ΔIL =
(1− DC)•( VOUT + VD)
L • fSW
Where fSW is the switching frequency of the LT3971, DC is
the duty cycle and L is the value of the inductor. Therefore,
the maximum output current that the LT3971 will deliver
depends on the switch current limit, the inductor value,
and the input and output voltages. The inductor value may
have to be increased if the inductor ripple current does
not allow sufficient maximum output current (IOUT(MAX))
given the switching frequency, and maximum input voltage
used in the desired application.
The optimum inductor for a given application may differ
from the one indicated by this simple design guide. A larger
value inductor provides a higher maximum load current and
reduces the output voltage ripple. If your load is lower than
the maximum load current, than you can relax the value of
the inductor and operate with higher ripple current. This
allows you to use a physically smaller inductor, or one with
a lower DCR resulting in higher efficiency. Be aware that if
the inductance differs from the simple rule above, then the
maximum load current will depend on the input voltage. In
addition, low inductance may result in discontinuous mode
operation, which further reduces maximum load current.
3971f
12
LT3971
APPLICATIONS INFORMATION
For details of maximum output current and discontinuous
operation, see Linear Technology’s Application Note 44.
Finally, for duty cycles greater than 50% (VOUT/VIN>0.5),
a minimum inductance is required to avoid sub-harmonic
oscillations. See Application Note 19.
One approach to choosing the inductor is to start with
the simple rule given above, look at the available inductors, and choose one to meet cost or space goals. Then
use the equations above to check that the LT3971 will be
able to deliver the required output current. Note again
that these equations assume that the inductor current is
continuous. Discontinuous operation occurs when IOUT
is less than ΔIL/2.
Input Capacitor
Bypass the input of the LT3971 circuit with a ceramic
capacitor of X7R or X5R type. Y5V types have poor
performance over temperature and applied voltage, and
should not be used. A 4.7μF to 10μF ceramic capacitor
is adequate to bypass the LT3971 and will easily handle
the ripple current. Note that larger input capacitance is
required when a lower switching frequency is used (due
to longer on-times). If the input power source has high
impedance, or there is significant inductance due to
long wires or cables, additional bulk capacitance may be
necessary. This can be provided with a low performance
electrolytic capacitor.
Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input
capacitor is required to reduce the resulting voltage
ripple at the LT3971 and to force this very high frequency
switching current into a tight local loop, minimizing EMI.
A 4.7μF capacitor is capable of this task, but only if it is
placed close to the LT3971 (see the PCB Layout section).
A second precaution regarding the ceramic input capacitor
concerns the maximum input voltage rating of the LT3971.
A ceramic input capacitor combined with trace or cable
inductance forms a high quality (under damped) tank circuit. If the LT3971 circuit is plugged into a live supply, the
input voltage can ring to twice its nominal value, possibly
exceeding the LT3971’s voltage rating. This situation is
easily avoided (see the Hot Plugging Safely section).
Output Capacitor and Output Ripple
The output capacitor has two essential functions. Along
with the inductor, it filters the square wave generated by the
LT3971 to produce the DC output. In this role it determines
the output ripple, so low impedance (at the switching
frequency) is important. The second function is to store
energy in order to satisfy transient loads and stabilize the
LT3971’s control loop. Ceramic capacitors have very low
equivalent series resistance (ESR) and provide the best
ripple performance. A good starting value is:
100
COUT =
VOUT fSW
where fSW is in MHz, and COUT is the recommended output
capacitance in μF. Use X5R or X7R types. This choice will
provide low output ripple and good transient response.
Transient performance can be improved with a higher
value capacitor. Increasing the output capacitance will
also decrease the output voltage ripple. A lower value of
output capacitor can be used to save space and cost but
transient performance will suffer.
When choosing a capacitor, look carefully through the
data sheet to find out what the actual capacitance is under
operating conditions (applied voltage and temperature). A
physically larger capacitor or one with a higher voltage rating
may be required. Table 3 lists several capacitor vendors.
Table 3. Recommended Ceramic Capacitor Vendors
MANUFACTURER
WEBSITE
AVX
www.avxcorp.com
Murata
www.murata.com
Taiyo Yuden
www.t-yuden.com
Vishay Siliconix
www.vishay.com
TDK
www.tdk.com
Catch Diode Selection
The catch diode (D1 from Block Diagram) conducts current only during switch off time. Average forward current
in normal operation can be calculated from:
ID( AVG) = IOUT
VIN – VOUT
VIN
where IOUT is the output load current. The only reason to
consider a diode with a larger current rating than necessary
3971f
13
LT3971
APPLICATIONS INFORMATION
for nominal operation is for the worst-case condition of
shorted output. The diode current will then increase to the
typical peak switch current. Peak reverse voltage is equal
to the regulator input voltage. Use a diode with a reverse
voltage rating greater than the input voltage.
Table 4. Schottky Diodes. The Reverse Current Values Listed Are
Estimates Based Off of Typical Curves for Reverse Current
vs Reverse Voltage at 25°C.
VR
(V)
IAVE
(A)
MBR0520L
20
0.5
MBR0540
40
0.5
PART NUMBER
VF at 1A
(mV)
VF at 2A
(mV)
IR at VR =
20V 25°C
(μA)
On Semiconductor
30
620
0.4
MBRM120E
20
1
530
MBRM140
40
1
550
595
0.5
B0530W
30
0.5
B0540W
40
0.5
620
1
B120
20
1
500
1.1
B130
30
1
500
1.1
B140
40
1
500
1.1
B150
50
1
700
B220
20
2
500
20
B230
30
2
500
0.6
B140HB
40
1
DFLS240L
40
2
DFLS140
40
1.1
510
1
DFLS160
60
1
500
2.5
DFLS2100
100
2
770
B240
40
2
20
Diodes Inc.
15
0.4
1
500
4
860
0.01
500
0.45
Central Semiconductor
CMSH1 - 40M
40
1
500
CMSH1 - 60M
60
1
700
CMSH1 - 40ML
40
1
400
CMSH2 - 40M
40
2
550
CMSH2 - 60M
60
2
700
CMSH2 - 40L
40
2
400
CMSH2 - 40
40
2
500
CMSH2 - 60M
60
2
700
An additional consideration is reverse leakage current.
When the catch diode is reversed biased, any leakage
current will appear as load current. When operating under
light load conditions, the low supply current consumed
by the LT3971 will be optimized by using a catch diode
with minimum reverse leakage current. Low leakage
Schottky diodes often have larger forward voltage drops
at a given current, so a trade-off can exist between low
load and high load efficiency. Often Schottky diodes with
larger reverse bias ratings will have less leakage at a given
output voltage than a diode with a smaller reverse bias
rating. Therefore, superior leakage performance can be
achieved at the expense of diode size. Table 4 lists several
Schottky diodes and their manufacturers.
Ceramic Capacitors
Ceramic capacitors are small, robust and have very low
ESR. However, ceramic capacitors can cause problems
when used with the LT3971 due to their piezoelectric nature.
When in Burst Mode operation, the LT3971’s switching
frequency depends on the load current, and at very light
loads the LT3971 can excite the ceramic capacitor at audio
frequencies, generating audible noise. Since the LT3971
operates at a lower current limit during Burst Mode operation, the noise is typically very quiet to a casual ear. If
this is unacceptable, use a high performance tantalum or
electrolytic capacitor at the output.
A final precaution regarding ceramic capacitors concerns
the maximum input voltage rating of the LT3971. As previously mentioned, a ceramic input capacitor combined
with trace or cable inductance forms a high quality (under
damped) tank circuit. If the LT3971 circuit is plugged into a
live supply, the input voltage can ring to twice its nominal
value, possibly exceeding the LT3971’s rating. This situation
is easily avoided (see the Hot Plugging Safely section).
BOOST and BD Pin Considerations
Capacitor C3 and the internal boost Schottky diode (see
the Block Diagram) are used to generate a boost voltage that is higher than the input voltage. In most cases
a 0.47μF capacitor will work well. Figure 3 shows three
ways to arrange the boost circuit. The BOOST pin must
be more than 2.3V above the SW pin for best efficiency.
3971f
14
LT3971
APPLICATIONS INFORMATION
For outputs of 3V and above, the standard circuit (Figure 3a)
is best. For outputs between 2.8V and 3V, use a 1μF boost
capacitor. A 2.5V output presents a special case because it
is marginally adequate to support the boosted drive stage
while using the internal boost diode. For reliable BOOST pin
operation with 2.5V outputs use a good external Schottky
diode (such as the ON Semi MBR0540), and a 1μF boost
capacitor (Figure 3b). For output voltages below 2.5V,
the boost diode can be tied to the input (Figure 3c), or to
another external supply greater than 2.8V. However, the
circuit in Figure 3a is more efficient because the BOOST pin
current comes from a lower voltage source. You must also
be sure that the maximum voltage ratings of the BOOST
and BD pins are not exceeded.
BD
VIN
VIN
BOOST
LT3971
4.7μF
GND
The minimum operating voltage of an LT3971 application
is limited by the minimum input voltage (4.3V) and by
the maximum duty cycle as outlined in the Input Voltage
Range section. For proper start-up, the minimum input
voltage is also limited by the boost circuit. If the input
voltage is ramped slowly, the boost capacitor may not
be fully charged. Because the boost capacitor is charged
with the energy stored in the inductor, the circuit will rely
on some minimum load current to get the boost circuit
running properly. This minimum load will depend on input
and output voltages, and on the arrangement of the boost
circuit. The minimum load generally goes to zero once the
circuit has started. Figure 4 shows a plot of minimum load
to start and to run as a function of input voltage. In many
cases the discharged output capacitor will present a load
to the switcher, which will allow it to start. The plots show
the worst-case situation where VIN is ramping very slowly.
C3
SW
5.0
VOUT
4.8
INPUT VOLTAGE (V)
4.6
(3a) For VOUT > 2.8V
D2
BD
VIN
VIN
4.7μF
GND
C3
SW
TO START
4.2
4.0
TO RUN
3.8
3.6
3.4 VOUT = 3.3V
TA = 25°C
3.2 L = 4.7μH
f = 800kHz
3.0
10
BOOST
LT3971
4.4
VOUT
100
LOAD CURRENT (mA)
1000
100
LOAD CURRENT (mA)
1000
6.4
6.2
INPUT VOLTAGE (V)
(3b) For 2.5V < VOUT < 2.8V
BD
VIN
VIN
BOOST
LT3971
4.7μF
GND
C3
VOUT
SW
TO START
6.0
5.8
5.6
TO RUN
5.4
5.2
VOUT = 5V
TA = 25°C
L = 4.7μH
f = 800kHz
5.0
3971 FO3
(3c) For VOUT < 2.5V; VIN(MAX) = 27V
Figure 3. Three Circuits for Generating the Boost Voltage
10
3971 F04
Figure 4. The Minimum Input Voltage Depends on
Output Voltage, Load Current and Boost Circuit
3971f
15
LT3971
APPLICATIONS INFORMATION
For lower start-up voltage, the boost diode can be tied to
VIN; however, this restricts the input range to one-half of
the absolute maximum rating of the BOOST pin.
At light loads, the inductor current becomes discontinuous and this reduces the minimum input voltage to approximately 400mV above VOUT. At higher load currents,
the inductor current is continuous and the duty cycle is
limited by the maximum duty cycle of the LT3971, requiring
a higher input voltage to maintain regulation.
Enable Pin
Be aware that when the input voltage is below 4.3V, the
input current may rise to several hundred μA. And the part
may be able to switch at cold or for VIN(EN) thresholds less
than 7V. Figure 6 shows the magnitude of the increased
input current in a typical application with different programmed VIN(EN).
When operating in Burst Mode for light load currents, the
current through the VIN(EN) resistor network can easily be
greater than the supply current consumed by the LT3971.
Therefore, the VIN(EN) resistors should be large to minimize
their effect on efficiency at low loads.
The LT3971 is in shutdown when the EN pin is low and
active when the pin is high. The rising threshold of the EN
comparator is 1.01V, with 30mV of hysteresis. The EN pin
can be tied to VIN if the shutdown feature is not used.
400
INPUT CURRENT (μA)
Adding a resistor divider from VIN to EN programs the
LT3971 to regulate the output only when VIN is above a
desired voltage (see Figure 5). Typically, this threshold,
VIN(EN), is used in situations where the input supply is current limited, or has a relatively high source resistance. A
switching regulator draws constant power from the source,
so source current increases as source voltage drops. This
looks like a negative resistance load to the source and can
cause the source to current limit or latch low under low
source voltage conditions. The VIN(EN) threshold prevents
the regulator from operating at source voltages where the
problems might occur. This threshold can be adjusted by
setting the values R3 and R4 such that they satisfy the
following equation:
12V VIN(EN) Input Current
500
300
200
100
0
0
1
2
3
4 5 6 7 8 9 10 11 12
INPUT VOLTAGE (V)
VIN(EN) = 12V
R3 = 11M
R4 = 1M
6V VIN(EN) Input Current
500
where output regulation should not start until VIN is above
VIN(EN). Due to the comparator’s hysteresis, switching will
not stop until the input falls slightly below VIN(EN).
R3
1V
EN
+
–
300
200
100
0
LT3971
VIN
INPUT CURRENT (μA)
400
R3
VIN(EN) =
+1
R4
0
SHDN
1
VIN(EN) = 6V
R3 = 5M
R4 = 1M
2
3
4
INPUT VOLTAGE (V)
5
6
3971 F06
R4
3971 F05
Figure 5. Programmed Enable Threshold
Figure 6. Input Current vs Input Voltage
for a Programmed VIN(EN) of 6V and 12V
3971f
16
LT3971
APPLICATIONS INFORMATION
Soft-Start
The SS pin can be used to soft-start the LT3971 by throttling
the maximum input current during start-up. An internal 1μA
current source charges an external capacitor generating a
voltage ramp on the SS pin. The SS pin clamps the internal
VC node, which slowly ramps up the current limit. Maximum
current limit is reached when the SS pin is about 1.5V or
higher. By selecting a large enough capacitor, the output
can reach regulation without overshoot. For applications
with input voltages above 25V, a 100k resistor in series
with the soft-start capacitor is recommended. Figure 7
shows start-up waveforms for a typical application with
a 10nF capacitor on SS for a 3.3Ω load when the EN pin
is pulsed high for 13ms.
The external SS capacitor is only actively discharged when
EN is low. With EN low, the external SS cap is discharged
through approximately 150Ω. The EN pin needs to be low
long enough for the external cap to completely discharge
through the 150Ω pull-down prior to start-up.
VSS
1V/DIV
The LT3971 will not enter Burst Mode operation at low
output loads while synchronized to an external clock, but
instead will pulse skip to maintain regulation.
The LT3971 may be synchronized over a 250kHz to 2MHz
range. The RT resistor should be chosen to set the LT3971
switching frequency 20% below the lowest synchronization
input. For example, if the synchronization signal will be
250kHz and higher, the RT should be selected for 200kHz.
To assure reliable and safe operation the LT3971 will only
synchronize when the output voltage is near regulation as
indicated by the PG flag. It is therefore necessary to choose
a large enough inductor value to supply the required output
current at the frequency set by the RT resistor (see the
Inductor Selection section). The slope compensation is set
by the RT value, while the minimum slope compensation
required to avoid subharmonic oscillations is established
by the inductor size, input voltage, and output voltage.
Since the synchronization frequency will not change the
slopes of the inductor current waveform, if the inductor
is large enough to avoid subharmonic oscillations at the
frequency set by RT, than the slope compensation will be
sufficient for all synchronization frequencies.
Shorted and Reversed Input Protection
VOUT
2V/DIV
IL
0.5A/DIV
2ms/DIV
3971 F07
Figure 7. Soft-Start Waveforms for Front-Page Application
with 10nF Capacitor on SS. EN is Pulsed High for About
13ms with a 3.3Ω Load Resistor
Synchronization
To select low ripple Burst Mode operation, tie the SYNC pin
below 0.6V (this can be ground or a logic low output).
Synchronizing the LT3971 oscillator to an external frequency can be done by connecting a square wave (with
20% to 80% duty cycle) to the SYNC pin. The square
wave amplitude should have valleys that are below 0.6V
and peaks above 1.0V (up to 6V).
If the inductor is chosen so that it won’t saturate excessively, a LT3971 buck regulator will tolerate a shorted
output. There is another situation to consider in systems
where the output will be held high when the input to the
LT3971 is absent. This may occur in battery charging applications or in battery backup systems where a battery
or some other supply is diode ORed with the LT3971’s
output. If the VIN pin is allowed to float and the EN pin
is held high (either by a logic signal or because it is tied
to VIN), then the LT3971’s internal circuitry will pull its
quiescent current through its SW pin. This is fine if your
system can tolerate a few μA in this state. If you ground
the EN pin, the SW pin current will drop to essentially
zero. However, if the VIN pin is grounded while the output
is held high, regardless of EN, parasitic diodes inside the
LT3971 can pull current from the output through the SW
pin and the VIN pin. Figure 8 shows a circuit that will run
only when the input voltage is present and that protects
against a shorted or reversed input.
3971f
17
LT3971
APPLICATIONS INFORMATION
D4
MBRS140
VIN
VIN
BOOST
EN
SW
L1
C2
VOUT
VOUT
LT3971
GND
BD
FB
+
BACKUP
RPG
C3
C4
3971 F07
Figure 8. Diode D4 Prevents a Shorted Input from Discharging a
Backup Battery Tied to the Output. It Also Protects the Circuit from
a Reversed Input. The LT3971 Runs Only When the Input is Present
PCB Layout
For proper operation and minimum EMI, care must be taken
during printed circuit board layout. Figure 9 shows the
recommended component placement with trace, ground
plane and via locations. Note that large, switched currents
flow in the LT3971’s VIN and SW pins, the catch diode
(D1), and the input capacitor (C1). The loop formed by
these components should be as small as possible. These
components, along with the inductor and output capacitor,
should be placed on the same side of the circuit board,
and their connections should be made on that layer. Place
a local, unbroken ground plane below these components.
The SW and BOOST nodes should be as small as possible.
Finally, keep the FB and RT nodes small so that the ground
traces will shield them from the SW and BOOST nodes.
The Exposed Pad on the bottom of the package must be
soldered to ground so that the pad acts as a heat sink. To
keep thermal resistance low, extend the ground plane as
much as possible, and add thermal vias under and near
the LT3971 to additional ground planes within the circuit
board and on the bottom side.
Hot Plugging Safely
The small size, robustness and low impedance of ceramic
capacitors make them an attractive option for the input
bypass capacitor of LT3971 circuits. However, these capacitors can cause problems if the LT3971 is plugged into
a live supply. The low loss ceramic capacitor, combined
with stray inductance in series with the power source,
forms an under damped tank circuit, and the voltage at
C5
D1
GND
RT
R1
R2
C1
GND
3971 F09
VIAS TO LOCAL GROUND PLANE
VIAS TO VOUT
VIAS TO SYNC
VIAS TO RUN/SS
VIAS TO PG
VIAS TO VIN
OUTLINE OF LOCAL
GROUND PLANE
Figure 9. A Good PCB Layout Ensures Proper, Low EMI Operation
the VIN pin of the LT3971 can ring to twice the nominal
input voltage, possibly exceeding the LT3971’s rating and
damaging the part. If the input supply is poorly controlled
or the user will be plugging the LT3971 into an energized
supply, the input network should be designed to prevent
this overshoot. See Linear Technology Application Note
88 for a complete discussion.
High Temperature Considerations
For higher ambient temperatures, care should be taken in
the layout of the PCB to ensure good heat sinking of the
LT3971. The Exposed Pad on the bottom of the package
must be soldered to a ground plane. This ground should be
tied to large copper layers below with thermal vias; these
layers will spread heat dissipated by the LT3971. Placing
additional vias can reduce thermal resistance further. The
maximum load current should be derated as the ambient
temperature approaches the maximum junction rating.
Power dissipation within the LT3971 can be estimated by
calculating the total power loss from an efficiency measurement and subtracting the catch diode loss and inductor
3971f
18
LT3971
APPLICATIONS INFORMATION
loss. The die temperature is calculated by multiplying the
LT3971 power dissipation by the thermal resistance from
junction to ambient.
avoid excessive increase in light load supply current at
high temperatures.
Other Linear Technology Publications
Also keep in mind that the leakage current of the power
Schottky diode goes up exponentially with junction temperature. When the power switch is closed, the power
Schottky diode is in parallel with the power converter’s
output filter stage. As a result, an increase in a diode’s
leakage current results in an effective increase in the load,
and a corresponding increase in input power. Therefore,
the catch Schottky diode must be selected with care to
Application Notes 19, 35 and 44 contain more detailed
descriptions and design information for buck regulators
and other switching regulators. The LT1376 data sheet
has a more extensive discussion of output ripple, loop
compensation and stability testing. Design Note 318
shows how to generate a bipolar output supply using a
buck regulator.
TYPICAL APPLICATIONS
5V Step-Down Converter
VIN
7V TO 38V
VIN
EN
OFF ON
BOOST
0.47μF
PG
4.7μF
4.7μH
SW
SS
LT3971
RT
BD
10pF
1M
49.9k
SYNC
FB
GND
309k
f = 800kHz
22μF
VOUT
5V
1.2A
3971 TA02
3.3V Step-Down Converter
VIN
4.3V TO 38V
VIN
EN
OFF ON
BOOST
0.47μF
PG
4.7μF
SS
4.7μH
SW
LT3971
RT
BD
10pF
71.5k
1M
SYNC
f = 600kHz
GND
FB
562k
22μF
VOUT
3.3V
1.2A
3971 TA03
3971f
19
LT3971
TYPICAL APPLICATIONS
1.8V Step-Down Converter
2.5V Step-Down Converter
VIN
4.3V TO 38V
VIN
4.3V TO 27V
VIN
OFF ON
BOOST
EN
OFF ON
1μF
PG
4.7μF
BOOST
0.47μF
PG
4.7μH
SW
SS
BD
VIN
EN
SS
LT3971
4.7μH
SW
LT3971
4.7μF
RT
RT
BD
10pF
10pF
VOUT
2.5V
1.2A
1M
118k
SYNC
GND
FB
47μF
909k
f = 400kHz
118k
511k
SYNC
GND
FB
1M
f = 400kHz
3971 TA04
100μF
VOUT
1.8V
1.2A
3971 TA05
12V Step-Down Converter
VIN
15V TO 38V
VIN
EN
OFF ON
BOOST
0.47μF
PG
10μF
SW
SS
10μH
LT3971
RT
BD
10pF
VOUT
12V
1.2A
1M
49.9k
SYNC
GND
FB
110k
f = 800kHz
10μF
3971 TA06
3.3V Step-Down Converter with Undervoltage Lockout, Soft-Start, and Power Good
VIN
6V TO 38V
5M
VIN
BOOST
EN
0.47μF
4.7μH
SW
4.7μF
SS
100k
150k
LT3971
RT
PG
BD
1M
PGOOD
10pF
1nF
49.9k
1M
SYNC
f = 800kHz
GND
FB
562k
22μF
VOUT
3.3V
1.2A
3971 TA07
3971f
20
LT3971
TYPICAL APPLICATIONS
5V, 2MHz Step-Down Converter with Soft-Start
VIN
9V TO 25V
VIN
EN
OFF ON
BOOST
0.47μF
PG
2.2μH
SW
SS
LT3971
2.2μF
RT
BD
1nF
10pF
11k
1M
SYNC
GND
FB
309k
f = 2MHz
22μF
VOUT
5V
1.2A
3971 TA08
4V Step-Down Converter with a High Impedance Input Source
+
11M
24V
–
+
VIN
EN
CBULK
100μF
1M
BOOST
0.47μF
PG
SS
* AVERAGE OUTPUT POWER CANNOT
EXCEED THAT WHICH CAN BE PROVIDED
BY HIGH IMPEDANCE SOURCE.
NAMELY,
V2
•H
POUT(MAX) =
4R
4.7μH
SW
LT3971
4.7μF
RT
BD
1nF
10pF
49.9k
SYNC
f = 800kHz
VOUT
4V
1.2A*
1M
GND
FB
412k
100μF
WHERE V IS VOLTAGE OF SOURCE, R IS
INTERNAL SOURCE IMPEDANCE, AND N IS
LT3971 EFFICIENCY. MAXIMUM OUTPUT
CURRENT OF 1.2A CAN BE SUPPLIED FOR A
SHORT TIME BASED ON THE ENERGY
WHICH CAN BE SOURCED BY THE BULK
INPUT CAPACITANCE.
3971 TA09a
Sourcing a Maximum Load Pulse
VOUT
200mV/DIV
Start-Up from High Impedance Input Source
VIN
1V/DIV
VIN
5V/DIV
VOUT
2V/DIV
IL
1A/DIV
IL
500mA/DIV
500μs/DIV
3971 TA09b
2ms/DIV
3971 TA09c
3971f
21
LT3971
PACKAGE DESCRIPTION
DD Package
10-Lead Plastic DFN (3mm × 3mm)
(Reference LTC DWG # 05-08-1699)
R = 0.115
TYP
6
0.38 ± 0.10
10
0.675 ±0.05
3.50 ±0.05
1.65 ±0.05
2.15 ±0.05 (2 SIDES)
3.00 ±0.10
(4 SIDES)
PACKAGE
OUTLINE
1.65 ± 0.10
(2 SIDES)
PIN 1
TOP MARK
(SEE NOTE 6)
(DD) DFN 1103
5
0.200 REF
0.25 ± 0.05
0.50
BSC
2.38 ±0.05
(2 SIDES)
1
0.25 ± 0.05
0.50 BSC
0.75 ±0.05
0.00 – 0.05
2.38 ±0.10
(2 SIDES)
BOTTOM VIEW—EXPOSED PAD
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
NOTE:
1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-2).
CHECK THE LTC WEBSITE DATA SHEET FOR CURRENT STATUS OF VARIATION ASSIGNMENT
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE
TOP AND BOTTOM OF PACKAGE
3971f
22
LT3971
PACKAGE DESCRIPTION
MSE Package
10-Lead Plastic MSOP, Exposed Die Pad
(Reference LTC DWG # 05-08-1664 Rev C)
BOTTOM VIEW OF
EXPOSED PAD OPTION
2.794 p 0.102
(.110 p .004)
5.23
(.206)
MIN
0.889 p 0.127
(.035 p .005)
1
0.05 REF
10
3.00 p 0.102
(.118 p .004)
(NOTE 3)
DETAIL “B”
CORNER TAIL IS PART OF
DETAIL “B” THE LEADFRAME FEATURE.
FOR REFERENCE ONLY
NO MEASUREMENT PURPOSE
10 9 8 7 6
DETAIL “A”
0o – 6o TYP
1 2 3 4 5
GAUGE PLANE
0.53 p 0.152
(.021 p .006)
DETAIL “A”
0.18
(.007)
0.497 p 0.076
(.0196 p .003)
REF
3.00 p 0.102
(.118 p .004)
(NOTE 4)
4.90 p 0.152
(.193 p .006)
0.254
(.010)
0.29
REF
1.83 p 0.102
(.072 p .004)
2.083 p 0.102 3.20 – 3.45
(.082 p .004) (.126 – .136)
0.50
0.305 p 0.038
(.0197)
(.0120 p .0015)
BSC
TYP
RECOMMENDED SOLDER PAD LAYOUT
2.06 p 0.102
(.081 p .004)
SEATING
PLANE
0.86
(.034)
REF
1.10
(.043)
MAX
0.17 – 0.27
(.007 – .011)
TYP
0.50
(.0197)
BSC
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
0.1016 p 0.0508
(.004 p .002)
MSOP (MSE) 0908 REV C
3971f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
23
LT3971
RELATED PARTS
PART
DESCRIPTION
COMMENTS
LT3980
58V, 80V Transient Protection, 2A, 2.4MHz High Efficiency Micropower VIN(MIN) = 3.6V, VIN(MAX) = 58V, Transient to 80V, VOUT(MIN) = 0.79V,
IQ = 75μA, ISD <1μA, MSOP-16E 3mm × 4mm DFN-16 Package
Step-Down DC/DC Converter
LT3970
40V, 350mA High Efficiency Micropower Step-Down DC/DC Converter
with IQ = 2.5μA
LT3695
36V, 60V Transient Protection, 1A, 2.2MHz High Efficiency Micropower VIN(MIN) = 3.6V, VIN(MAX) = 36V, Transient to 60V, VOUT(MIN) = 0.8V,
IQ = 75μA, ISD <1μA, MSOP-16E Package
Step-Down DC/DC Converter with 1A Fault Tolerance
LT3689
36V, 60V Transient Protection, 800mA, 2.2MHz High Efficiency
Micropower Step-Down DC/DC Converter with POR Reset and
Watchdog Timer
VIN(MIN) = 3.6V, VIN(MAX) = 36V, Transient to 60V, VOUT(MIN) = 0.8V,
IQ = 75μA, ISD <1μA, 3mm × 3mm QFN-16 Package
LT3682
36V, 60VMAX, 1A, 2.2MHz High Efficiency Micropower Step-Down
DC/DC Converter
VIN(MIN) = 3.6V, VIN(MAX) = 36V, VOUT(MIN) = 0.8V,
IQ = 75μA, ISD <1μA, 3mm × 3mm DFN-12 Package
LT3480
36V with Transient Protection to 60V, 2A (IOUT), 2.4MHz, High
Efficiency Step-Down DC/DC Converter with Burst Mode Operation
VIN(MIN) = 3.6V, VIN(MAX) = 38V, VOUT(MIN) = 0.78V,
IQ = 70μA, ISD <1μA, 3mm × 3mm DFN-10, MSOP-10E Package
LT3685
36V with Transient Protection to 60V, 2A (IOUT), 2.4MHz,
High Efficiency Step-Down DC/DC Converter
VIN(MIN) = 3.6V, VIN(MAX) = 38V, VOUT(MIN) = 0.78V, IQ = 70μA,
ISD <1μA, 3mm × 3mm DFN-10, MSOP-10E Package
LT3481
34V with Transient Protection to 36V, 2A (IOUT), 2.8MHz, High
Efficiency Step-Down DC/DC Converter with Burst Mode Operation
VIN(MIN) = 3.6V, VIN(MAX) = 34V, VOUT(MIN) = 1.26V, IQ = 50μA,
ISD <1μA 3mm × 3mm DFN-10, MSOP-10E Package
LT3684
34V with Transient Protection to 36V, 2A (IOUT), 2.8MHz,
High Efficiency Step-Down DC/DC Converter
VIN(MIN) = 3.6V, VIN(MAX)= 34V, VOUT(MIN) = 1.26V, IQ = 850μA,
ISD <1μA, 3mm × 3mm DFN-10, MSOP-10E Package
LT3508
36V with Transient Protection to 40V, Dual 1.4A (IOUT), 3MHz,
High Efficiency Step-Down DC/DC Converter
VIN(MIN) = 3.7V, VIN(MAX) = 37V, VOUT(MIN) = 0.8V, IQ = 4.6mA,
ISD = 1μA, 4mm × 4mm QFN-24, TSSOP-16E Package
LT3505
36V with Transient Protection to 40V, 1.4A (IOUT), 3MHz,
High Efficiency Step-Down DC/DC Converter
VIN(MIN) = 3.6V, VIN(MAX) = 34V, VOUT(MIN) = 0.78V, IQ = 2mA,
ISD = 2μA, 3mm × 3mm DFN-8, MSOP-8E Package
LT3500
36V, 40VMAX, 2A, 2.5MHz High Efficiency Step-Down DC/DC Converter VIN(MIN) = 3.6V, VIN(MAX) = 36V, VOUT(MIN) = 0.8V, IQ = 2.5mA,
and LDO Controller
ISD <10μA, 3mm × 3mm DFN-10 Package
LT3507
36V 2.5MHz, Triple (2.4A + 1.5A + 1.5A (IOUT)) with LDO Controller
High Efficiency Step-Down DC/DC Converter
VIN(MIN) = 4.0V, VIN(MAX) = 36V, VOUT(MIN) = 0.8V, IQ = 7mA,
ISD = 1μA, 5mm × 7mm QFN-38 Package
LT3437
60V, 400mA (IOUT), Micropower Step-Down DC/DC Converter with
Burst Mode Operation
VIN(MIN) = 3.3V, VIN(MAX) = 60V, VOUT(MIN) = 1.25V, IQ = 100μA,
ISD <1μA, 3mm × 3mm DFN-10, TSSOP-16E Package
LT1976/
LT1977
60V, 1.2A (IOUT), 200kHz/500kHz, High Efficiency Step-Down DC/DC
Converter with Burst Mode Operation
VIN(MIN) = 3.3V, VIN(MAX) = 60V, VOUT(MIN) = 1.20V, IQ = 100μA,
ISD <1μA, TSSOP16E Package
LT3434/
LT3435
60V, 2.4A (IOUT), 200kHz/500kHz, High Efficiency Step-Down DC/DC
Converter with Burst Mode Operation
VIN(MIN) = 3.3V, VIN(MAX) = 60V, VOUT(MIN) = 1.20V, IQ = 100μA,
ISD <1μA, TSSOP16E Package
LT1936
36V, 1.4A(IOUT) , 500kHz High Efficiency Step-Down DC/DC Converter
VIN(MIN) = 3.6V, VIN(MAX) = 36V, VOUT(MIN) = 1.2V, IQ = 1.9mA,
ISD <1μA, MS8E Package
LT3493
36V, 1.4A(IOUT), 750kHz High Efficiency Step-Down DC/DC Converter
VIN(MIN) = 3.6V, VIN(MAX) = 36V, VOUT(MIN) = 0.8V, IQ = 1.9mA,
ISD <1μA, 2mm × 3mm DFN-6 Package
LT1766
60V, 1.2A (IOUT), 200kHz, High Efficiency Step-Down DC/DC Converter VIN(MIN) = 5.5V, VIN(MAX) = 60V, VOUT(MIN) = 1.20V, IQ = 2.5mA,
ISD = 25μA, TSSOP16E Package
VIN(MIN) = 4.2V, VIN(MAX) = 40V, VOUT(MIN) = 1.21V, IQ = 2.2μA,
ISD <1μA, MSOP-10 3mm × 2mm DFN-10 Package
3971f
24 Linear Technology Corporation
LT 1109 • PRINTED IN USA
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