LTC1626 Low Voltage, High Efficiency Step-Down DC/DC Converter U DESCRIPTION FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ The LTC ®1626 is a monolithic, low voltage, step-down current mode DC/DC converter featuring Burst ModeTM operation at low output current. The input supply voltage range of 2.5V to 6V makes the LTC1626 ideal for single cell Li-Ion and 3- or 4-cell NiCd/ NiMH applications. A built-in 0.32Ω switch (VIN = 4.5V) allows up to 0.6A of output current. Wide Input Supply Voltage Range: 2.5V to 6V High Efficiency: Up to 95% Low RDS(ON) Internal Switch: 0.32Ω (VIN = 4.5V) Current Mode Operation for Excellent Line and Load Transient Response Short-Circuit Protected Low Dropout Operation: 100% Duty Cycle Built-In Low-Battery Detector Low Quiescent Current at Light Loads: IQ = 165µA Ultralow Shutdown Current: IQ = 0.5µA Peak Inductor Current Independent of Inductor Value Available in 14-Pin SO Package The LTC1626 incorporates automatic power saving Burst Mode operation to reduce gate charge losses when the load current drops below the level required for continuous operation. With no load, the converter draws only 165µA. In shutdown, it draws a mere 0.5µA—making it ideal for current sensitive applications. The inductor current is user-programmable via an external current sense resistor. In dropout, the internal P-channel MOSFET switch is turned on continuously, maximizing battery life. U APPLICATIONS ■ ■ ■ ■ ■ ■ ■ Single Cell Li-Ion Step-Down Converters 3- or 4-Cell NiMH Step-Down Converters Cellular Telephones 5V to 3.3V Conversion 3.3V to 2.5V Conversion Inverting Converters Portable Instruments , LTC and LT are registered trademarks of Linear Technology Corporation. Burst Mode is a trademark of Linear Technology Corporation. U TYPICAL APPLICATION VIN 2.7V TO 6V Efficiency + CIN† 100 0.1µF PWR VIN VIN SHDN SW LTC1626 3900pF L* 33µH RSENSE** 0.1Ω VOUT 2.5V 0.25A D1 MBRS130LT + PGND 470Ω ITH CT CT 270pF * COILTRONICS CTX33-4 ** IRC 1206-R100F † AVX TPSD476KO16 †† AVX TPSC107M006R0150 SENSE + SENSE – SGND 1000pF COUT†† 100µF 6.3V 10k 95 EFFICIENCY (%) 47µF 16V 90 85 80 VFB 100pF 75 10k 1626 F01 70 0.01 VIN = 3.5V L1 = 33µH VOUT = 2.5V RSENSE = 0.1Ω CT = 270pF 0.1 OUTPUT CURRENT (A) 1 1626 F01a Figure 1. High Efficiency 2.5V Step-Down Converter 1 LTC1626 U U W W W ELECTRICAL CHARACTERISTICS PACKAGE/ORDER INFORMATION 14 SW PWR VIN 1 13 PWR VIN VIN 2 LBO 3 12 PGND LBI 4 11 SGND CT 5 10 SHDN ITH 6 9 VFB SENSE – 7 LTC1626CS 8 SENSE + S PACKAGE 14-LEAD PLASTIC SO TJMAX = 125°C, θJA = 110°C/ W Consult factory for Industrial and Military grade parts. TA = 25°C, VIN = 4.5V, VOUT = 2.5V, VSHDN = 0V, unless otherwise specified. PARAMETER IFB Feedback Pin Current VFB Feedback Voltage 0°C to 70°C – 40°C to 85°C ∆VOUT Output Voltage Line Regulation VIN = 3.5V to 5.5V, ILOAD = 250mA Output Voltage Load Regulation Burst Mode Output Ripple 10mA ≤ ILOAD ≤ 250mA ILOAD = 0 Input DC Supply Current (Note 2) Active Mode Sleep Mode Shutdown ORDER PART NUMBER TOP VIEW SYMBOL IQ W (Voltages Referred to GND Pin) Input Supply Voltage (Pins 1, 2, 13) ............– 0.3V to 7V Shutdown Input Voltage (Pin 10) ................– 0.3V to 7V Sense–, Sense+ (Pins 7, 8)........... – 0.3V to (VIN + 0.3V) LBO, LBI (Pins 3, 4) .................................... – 0.3V to 7V CT, ITH, VFB (Pins 5, 6, 9) ............. – 0.3V to (VIN + 0.3V) DC Switch Current (Pin 14) .................................... 1.2A Peak Switch Current (Pin 14) ................................. 1.6A Switch Voltage (Pin 14) .......(VIN – 7.5V) to (VIN + 0.3V) Operating Temperature Range ..................... 0°C to 70°C Extended Commercial Operating Temperature Range (Note 4) ............. – 40°C to 85°C Junction Temperature (Note 1) ............................. 125°C Storage Temperature Range ................. – 65°C to 150°C Lead Temperature (Soldering, 10 sec).................. 300°C U ABSOLUTE MAXIMUM RATINGS CONDITIONS MIN ● ● TYP MAX 0.1 1 µA 1.22 1.2 1.25 1.28 1.3 V V – 40 0 40 mV 25 50 50 mV mVP-P 1.9 165 0.5 3.0 300 5 mA µA µA 1.15 1.25 1.35 V ● ● VSHDN = VIN UNITS VLBTRIP Low-Battery Trip Point ILBI Low-Battery Input Bias Current ILBO Low-Battery Output Sink Current VLBO = 0.4V 0.4 1.4 VSENSE Current Sense Threshold Voltage VSENSE + – VSENSE– VSENSE – = 2.5V, VFB = VOUT/2 + 25mV (Forced) VSENSE – = 2.5V, VFB = VOUT/2 – 25mV (Forced) 130 25 155 180 mV mV 0.32 0.45 Ω 5 6 µs ±0.5 RON ON Resistance of Switch tOFF Switch Off-Time (Note 3) CT = 390pF, ILOAD = 400mA VIHSD SHDN Pin High Minimum Voltage for Device to Be Shut Down VILSD SHDN Pin Low Maximum Voltage for Device to Be Active IINSD SHDN Pin Input Current 0V ≤ VSHDN ≤ 7V The ● denotes specifications that apply over the specified operating temperature range. Note 1: TJ is calculated from the ambient temperature TA and power dissipation according to the following formula: TJ = TA + (PD • 110°C/W) Note 2: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. 2 4 VIN – 0.4 ● µA mA V 0.4 V ±1 µA Note 3: In applications where RSENSE is placed at ground potential, the off-time increases by approximately 40%. Note 4: C grade device specifications are guaranteed over the 0°C to 70°C temperature range. In addition, C grade device specifications are assured over the – 40°C to 85°C temperature range by design or correlation, but are not production tested. LTC1626 U W TYPICAL PERFORMANCE CHARACTERISTICS Efficiency vs Output Current (VOUT = 3.3V) Efficiency vs Input Voltage (VOUT = 2.5V) 100 100 L1 = 33µH RSENSE = 0.1Ω CT = 270pF 98 95 IOUT = 100mA EFFICIENCY (%) 92 90 I OUT = 250mA 88 86 84 90 85 80 75 82 80 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 6.5 7.0 7.5 INPUT VOLTAGE (V) 96 L1 = 33µH VIN = 5V VOUT = 3.3V RSENSE = 0.1Ω CT = 270pF 70 0.01 Operating Frequency 0.1 OUTPUT CURRENT (A) 80 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 6.5 7.0 7.5 INPUT VOLTAGE (V) 1 1626 G03 Switch Leakage Current 0.9 90 1.6 0.8 80 1.4 0.7 FIGURE 1 CIRCUIT 0.8 0.6 TJ = 70°C 0.5 0.4 0.6 0.3 0.4 0.2 0.2 0.1 TJ = 25°C TJ = 0°C DC Supply Current* 60 50 40 30 10 0 0 Supply Current in Shutdown TJ = 25°C * DOES NOT INCLUDE GATE CHARGE CURRENT 0.45 Low Voltage Behavior 5.0 TJ = 25°C SHUTDOWN = VIN 4.5 SUPPLY CURRENT (µA) 0.40 3.0 ACTIVE MODE 2.0 1.5 4.0 0.35 0.30 0.25 0.20 0.15 3.5 3.0 2.5 1.0 0.5 0.05 0.5 1626 G07 0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 6.5 7.0 7.5 INPUT VOLTAGE (V) 1626 G08 VOUT = 3.3V VOUT = 2.5V 1.5 0.10 SLEEP MODE L1 = 33µH RSENSE = 0.1Ω CT = 270pF TJ = 25°C ILOAD = 250mA 2.0 1.0 0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 6.5 7.0 7.5 INPUT VOLTAGE (V) 10 20 30 40 50 60 70 80 90 100 JUNCTION TEMPERATURE (°C) 1626 G06 0.50 3.5 2.5 70 1626 G05 5.0 4.0 VIN = 4.5V 20 0 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 6.5 7.0 INPUT VOLTAGE (V) 1626 G04 4.5 LEAKAGE CURRENT (µA) 100 1.0 L1 = 33µH RSENSE = 0.1Ω CT = 270pF 82 Switch Resistance RDS(ON) (Ω) NORMALIZED FREQUENCY 88 1.0 0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 6.5 7.0 7.5 INPUT VOLTAGE (V) SUPPLY CURRENT (mA) 90 1626 G02 2.0 1.2 IOUT = 250mA 92 84 1626 G01 1.8 IOUT = 100mA 94 86 OUTPUT VOLTAGE (V) EFFICIENCY (%) 96 100 EFFICIENCY (%) 98 94 Efficiency vs Input Voltage (VOUT = 3.3V) 0 0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 INPUT VOLTAGE (V) 1626 G09 3 LTC1626 U U U PIN FUNCTIONS PWR VIN (Pins 1, 13): Supply for the Power MOSFET and Its Driver. Decouple this pin properly to ground. SENSE – (Pin 7): Connects to the (–) Input of the Current Comparator. VIN (Pin 2): Main Supply for All the Control Circuitry in the LTC1626. SENSE + (Pin 8): The (+) Input to the Current Comparator. A built-in offset between Pins 7 and 8 in conjunction with RSENSE sets the current trip threshold. LBO (Pin 3): Open-Drain Output of the Low-Battery Comparator. This pin will sink current when Pin 4 (LBI) goes below 1.25V. During shutdown, this pin is high impedance. VFB (Pin 9): This pin serves as the feedback pin from an external resistive divider used to set the output voltage. SHDN (Pin 10): Shutdown Pin. Pulling this pin to VIN keeps the internal switch off and puts the LTC1626 in micropower shutdown. If not used, connect to SGND. LBI (Pin 4): The (–) Input of the Low-Battery Comparator. The (+) input is connected to a reference voltage of 1.25V. If not used, connect to VIN. SGND (Pin 11): Small-Signal Ground. Must be routed separately from other grounds to the (–) terminal of COUT. CT (Pin 5): External capacitor CT from Pin 5 to ground sets the switch off-time. The operating frequency is dependent on the input voltage and CT. PWR GND (Pin 12): Switch Driver Ground. Connects to the (–) terminal of CIN. ITH (Pin 6): Feedback Amplifier Decoupling Point. The current comparator threshold is proportional to Pin 6 voltage. SW (Pin 14): Drain of the P-Channel MOSFET Switch. Cathode of the Schottky diode must be connected closely to this pin. W BLOCK DIAGRAM PWR VIN 1 13 SENSE + SENSE – 8 7 PWR GND 14 SW 12 – 9 VFB V + SLEEP – R C Q + + S S – ITH VTH2 25mV TO 150mV + VOS – 13k G 6 VTH1 – T + 4 LBO 2 3 + + OFF-TIME CONTROL 5 CT VIN SENSE – VFB SGND 11 REFERENCE 10 SHDN A3 – 4 LBI 1626 BD LTC1626 U OPERATIO The nominal off-time of the LTC1626 is set by an external timing capacitor connected between the CT pin and ground. The operating frequency is then determined by the offtime and the difference between VIN and VOUT. The output voltage is set by an external divider returned to the VFB pin. A voltage comparator V and a gain block G compare the divided output voltage with a reference voltage of 1.25V. To optimize efficiency, the LTC1626 automatically switches between continuous and Burst Mode operation. The voltage comparator is the primary control element when the device is in Burst Mode operation, while the gain block controls the output voltage in continuous mode. When the load is heavy, the LTC1626 is in continuous operation. During the switch “ON” time, current comparator C monitors the voltage between the SENSE + and SENSE – pins connected across an external shunt in series with the inductor. When the voltage across the shunt reaches the comparator’s threshold value, its output signal changes state, resetting the flip-flop and turning the internal P-channel MOSFET off. The timing capacitor connected to the CT pin is now allowed to discharge at a rate determined by the off-time controller. When the voltage on the timing capacitor has discharged past VTH1, comparator T trips, sets the flip-flop and causes the switch to turn on. Also, the timing capacitor is recharged. The inductor current will again ramp up until the current comparator C trips. The cycle then repeats. When the load current increases, the output voltage decreases slightly. This causes the output of the gain stage (Pin 6) to increase the current comparator threshold, thus tracking the load current. When the load is relatively light, the LTC1626 automatically switches to Burst Mode operation. The current loop is interrupted when the output voltage reaches the desired regulated value. The hysteretic voltage comparator V trips when VOUT is above the desired output voltage, turning off the switch and causing the timing capacitor to discharge. This capacitor discharges past VTH1 until its voltage drops below VTH2. Comparator S then trips and a sleep signal is generated. The circuit now enters into sleep mode with the power MOSFET turned off. In sleep mode, the LTC1626 is in standby and the load current is supplied by the output capacitor. All unused circuitry is shut off, reducing quiescent current from 1.9mA to 165µA. When the output capacitor discharges by the amount of the hysteresis of the comparator V, the P-channel switch turns on again and the process repeats itself. During Burst Mode operation, the peak inductor’s current is set at 25mV/RSENSE. To avoid the operation of the current loop interfering with Burst Mode operation, a built-in offset VOS is incorporated in the gain stage. This prevents the current from increasing until the output voltage has dropped below a minimum threshold. In dropout, the P-channel MOSFET is turned on continuously (100% duty cycle) providing low dropout operation with VOUT ≅ VIN. U W U U APPLICATIONS INFORMATION The basic LTC1626 application circuit is shown in Figure 1. External component selection is driven by the load requirement and begins with the selection of RSENSE. Once RSENSE is known, CT and L can be chosen. Next, the Schottky diode D1 is selected followed by CIN and COUT. determines the peak inductor current. Depending upon the load current condition, the threshold of the comparator lies between 25mV/RSENSE and 150mV/RSENSE. The maximum output current of the LTC1626 is: RSENSE Selection for Output Current Where IRIPPLE is the peak-to-peak inductor ripple current. At a relatively light load, the LTC1626 is in Burst Mode operation. In this mode, the peak current is set at 25mV/ RSENSE. To fully benefit from Burst Mode operation, the RSENSE is chosen based on the required output current. With the current comparator monitoring the voltage developed across RSENSE, the threshold of the comparator IOUT(MAX) = 150mV/RSENSE – IRIPPLE/2 (A) 5 LTC1626 U W U U APPLICATIONS INFORMATION inductor current should be continuous during burst periods. Hence, the peak-to-peak inductor ripple current must not exceed 25mV/RSENSE. To account for light load conditions, the IOUT(MAX) is then given by: IOUT(MAX) = 150mV/RSENSE – 25mV/2RSENSE (A) = 137.5mV/RSENSE (A) Solving for RSENSE and allowing a margin of variations in the LTC1626 and external component values yields: RSENSE = 100mV/IOUT(MAX) (Ω) The LTC1626 switch is capable of supplying a maximum of 1.2A of output current. Therefore, the minimum value of RSENSE that can be used is 0.083Ω. A graph for selecting RSENSE versus maximum output current is given in Figure 2. 0.4 RSENSE (Ω) For most applications, the LTC1626 should be operated in the 100kHz to 300kHz range. This range can be extended, however, up to 600kHz, to accommodate smaller size/ valued inductors, such as low profile types, with a slight decrease in efficiency due to gate charge losses. Some experimentation may be required to determine the optimum operating frequency for a particular set of external components and operating conditions. CT and L Selection The value of CT is calculated from the desired continuous mode operating frequency: CT = (VIN − VOUT) F (VIN + VD)(3300)(VIN − VBE)(fO) ( ) where VD is the drop across the Schottky diode D1 and VBE is a base-emitter voltage drop (0.6V). 0.5 The complete expression for operating frequency is given by: 0.3 1 VIN − VOUT fO ≈ t OFF VIN + VD 0.2 where: 0.1 0 0 0.2 0.4 0.6 0.8 MAXIMUM OUTPUT CURRENT (A) 1.0 1626 F02 Figure 2. Selecting RSENSE During a short circuit of the regulator output to ground, the peak current is determined by: ISC = 150mV/RSENSE (A) In this condition, the LTC1626 automatically extends the off-time period of the P-channel MOSFET switch to allow the inductor current to decay far enough to prevent any current buildup. The resulting ripple current causes the average current to be approximately IOUT(MAX). 6 Operating Frequency Considerations ( )( )( (Hz) ) (sec) t OFF = 3300 CT VIN − VBE Figure 3 is a graph of operating frequency versus power supply voltage for the 2.5V regulator circuit shown in Figure 1 (CT = 270pF). Note that the frequency is relatively constant with supply voltage but drops as the supply voltage approaches the regulated output voltage. To maintain continuous inductor current at light load, the inductor must be chosen to provide no more than 25mV/ RSENSE of peak-to-peak ripple current. This results in the following expression for L: ( ) ( )( )( ) (H) L ≥ 5.2 105 RSENSE CT VREG Using an inductance smaller than the above value will result in inductor current being discontinuous. As a con- LTC1626 U W U U APPLICATIONS INFORMATION sequence, the LTC1626 will delay entering Burst Mode operation and efficiency will be degraded at low currents. 200 180 FIGURE 1 CIRCUIT the P-channel switch duty cycle. At high input voltages, the diode conducts most of the time. As VIN approaches VOUT, the diode conducts only a small fraction of the time. The most stressful condition for the diode is when the regulator output is shorted to ground. FREQUENCY (kHz) 160 Under short-circuit conditions, the diode must safely handle ISC(PK) at close to 100% duty cycle. Most LTC1626 circuits will be well served by either an MBRM5819 or an MBRS130LT3. An MBR0520LT1 is a good choice for IOUT(MAX) ≤ 500mA. 140 120 100 80 60 40 20 0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 6.5 7.0 7.5 INPUT VOLTAGE (V) 1626 F03 Figure 3. Operating Frequency vs Supply Voltage for Circuit Shown in Figure 1 Inductor Core Selection With the value of L selected, the type of inductor must be chosen. Basically, there are two kinds of losses in an inductor—core and copper losses. Core losses are dependent on the peak-to-peak ripple current and core material. However, they are independent of the physical size of the core. By increasing inductance, the peak-to-peak inductor ripple current will decrease, therefore reducing core loss. Utilizing low core loss material, such as molypermalloy or Kool Mµ® will allow the user to concentrate on reducing copper loss and preventing saturation. Although higher inductance reduces core loss, it increases copper loss as it requires more windings. When space is not a premium, larger wire can be used to reduce the wire resistance. This also prevents excessive heat dissipation in the inductor. Catch Diode Selection Losses in the catch diode depend on forward drop and switching times. Therefore, Schottky diodes are a good choice for low drop and fast switching times. The catch diode carries the load current during the offtime. The average diode current is therefore dependent on Input Capacitor (CIN) Selection In continuous mode, the input current of the converter is a square wave of duty cycle VOUT/VIN. To prevent large voltage transients, a low effective series resistance (ESR) input capacitor must be used. In addition, the capacitor must handle a high RMS current. The CIN RMS current is given by: IRMS ≈ [ ( IOUT VOUT VIN − VOUT )] 1/ 2 VIN ( A) This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst case is commonly used to design because even significant deviations do not offer much relief. Note that capacitor manufacturer’s ripple current ratings are often based on only 2000 hours lifetime. This make it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Do not underspecify this component. An additional 0.1µF ceramic capacitor is also required on PWR VIN for high frequency decoupling. Output Capacitor (COUT) Selection The selection of COUT is driven by the ESR for proper operation of the LTC1626. The required ESR of COUT is: ESRCOUT < 50mV/IRIPPLE where IRIPPLE is the ripple current of the inductor. For the case where the IRIPPLE is 25mV/RSENSE, the required ESR of COUT is: Kool Mµ is a registered trademark of Magnetics, Inc. 7 LTC1626 U W U U APPLICATIONS INFORMATION ESRCOUT < 2RSENSE To avoid overheating, the output capacitor must be sized to handle the ripple current generated by the inductor. The worst-case RMS ripple current in the output capacitor is given by: IRMS < 150mV/2RSENSE (ARMS) Generally, once the ESR requirements for COUT have been met, the RMS current rating far exceeds the IRIPPLE requirement. In some surface mount applications, multiple capacitors may have to be paralleled to meet the capacitance, ESR or RMS current handling requirement of the application. Aluminum electrolyte and dry tantalum capacitors are both available in surface mount configurations. In the case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalums, available in case heights ranging from 2mm to 4mm. Other capacitor types include Sanyo OS-CON, Nichicon PL series and Sprague 595D series. Consult the manufacturer for other specific recommendations. When the capacitance of COUT is made too small, the output ripple at low frequencies will be large enough to trip the voltage comparator. This causes Burst Mode operation to be activated when the LTC1626 would normally be in continuous mode operation. The effect will be most pronounced with low RSENSE values and can be improved at higher frequencies. Low-Battery Detection The low-battery detector senses the input voltage through an external resistive divider. This divided voltage connects to the (–) input of a voltage comparator (LBI) and is compared to an internal 1.25V reference voltage. Neglecting LBI input bias current, the following expression is used for setting the trip voltage threshold: The LBO is an N-channel open drain that goes low when the battery voltage drops below the threshold voltage. In shutdown, the comparator is disabled and LBO is in the high impedance state. Figure 4 is a schematic diagram detailing the low-battery comparator connection and operation. VIN LTC1626 R4 1% 1.25V LBI CFILTER 0.01µF – R3 1% 1626 F04 Figure 4. Low-Battery Comparator Setting the Output Voltage The LTC1626 develops a 1.25V reference voltage between the feedback pin VFB and the signal ground as shown in Figure 5. By selecting resistor R1, a constant current is caused to flow through R1 and R2 which sets the desired output voltage. The regulated output voltage is determined by: R2 VOUT = 1.25 1 + R1 R1 should be ≤ 10k to ensure that sufficient current flows through the divider to maintain accuracy and to provide a minimum load for the regulator output at elevated temperatures. (See Switch Leakage Current curve in Typical Performance Characteristics section.) To prevent stray pickup, a 100pF capacitor is suggested across R1, located close to the LTC1626. VOUT R2 1% LTC1626 SGND R4 VLB _ TRIP = 1.25 1 + R3 LBO + VFB 100pF R1 10k 1% 1626 F05 Figure 5. Setting the Output Voltage 8 LTC1626 U U W U APPLICATIONS INFORMATION Thermal Considerations In a majority of applications, the LTC1626 does not dissipate much heat due to its high efficiency. However, in applications where the switching regulator is running at high duty cycles or the part is in dropout with the switch turned on continuously (DC), some thermal analysis is required. The goal of the thermal analysis is to determine whether the power dissipated by the regulator exceeds the maximum junction temperature. The temperature rise is given by: TRISE = PD • θJA where PD is the power dissipated by the regulator and θJA is the thermal resistance from the junction of the die to the ambient temperature. The junction temperature is given by: TJ = TRISE + TAMBIENT As an example, consider the case when the LTC1626 is in dropout at an input voltage of 3V with a load current of 0.5A. From the Typical Performance Characteristics graph of Switch Resistance, the ON resistance of the P-channel switch is 0.45Ω. Therefore, power dissipated by the part is: PD = I2 • RDS(ON) = 113mW The SO package junction-to-ambient thermal resistance θJA is 110°C/W. Therefore, the junction temperature of the regulator when it is operating in a 25°C ambient temperature is: TJ = (0.113 • 110) + 25 = 38°C Remembering that the above junction temperature is obtained from an RDS(ON) at 25°C, we might recalculate the junction temperature based on a higher RDS(ON) since it increases with temperature. However, we can safely assume that the actual junction temperature will not exceed the absolute maximum junction temperature of 125°C. Board Layout Considerations When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC1626. These items are also illustrated graphically in the layout diagram of Figure 6. Check the following in your layout: 1. Are the signal and power grounds separated? The LTC1626 signal ground (Pin 11) must return to the (–) plate of COUT. The power ground (Pin 12) returns to the anode of the Schottky diode and the (–) plate of CIN. 2. Does the (+) plate of CIN connect to the power VIN (Pins 1, 13) as close as possible? This capacitor provides the AC current to the internal P-channel MOSFET and its driver. VIN 1 PWR VIN 2 VIN 14 SW PWR VIN 13 LTC1626 3 1k 3900pF CT 4 5 6 7 LBO LBI 0.1µF PGND SGND CT SHDN ITH VFB SENSE – BOLD LINES INDICATE HIGH CURRENT PATHS D1 + CIN L 12 11 10 SHUTDOWN R1 9 8 SENSE + + COUT R2 1000pF RSENSE VOUT 1626 F06 Figure 6. LTC1626 Layout Diagram (See Board Layout Checklist) 9 LTC1626 U U W U APPLICATIONS INFORMATION 3. Is the input decoupling capacitor (0.1µF) connected closely between power VIN (Pins 1, 13) and power ground (Pin 12)? This capacitor carries the high frequency peak currents. divider R1-R2 must be connected between the (+) plate of COUT and the signal ground. 6. Are the SENSE – and SENSE + leads routed together with minimum PC trace spacing? The 1000pF capacitor between Pin 7 and Pin 8 should be as close as possible to the LTC1626. 4. Is the Schottky diode closely connected between the power ground (Pin 12) and switch output (Pin 14)? 5. Does the LTC1626 SENSE – (Pin 7) connect to a point close to RSENSE and the (+) plate of COUT? The resistor 7. Is SHDN (Pin 10) actively pulled to ground during normal operation? The shutdown pin is high impedance and must not be allowed to float. U TYPICAL APPLICATIONS Single Cell Li-Ion to 2.5V Converter (VIN = 2.7V TO 4.5V) SINGLE Li-ION CELL + + 0.1µF VIN PWR VIN LBI CIN† 47µF 16V L1* 22µH VOUT 2.5V 0.25A SW D1 MBR0520LT1 LBO SHDN SHUTDOWN RSENSE** 0.1Ω PGND + LTC1626 SENSE + ITH 1k 1000pF CT 270pF 3900pF SENSE – COUT†† 100µF 10V 10k VFB CT SGND 10k 100pF 1626 TA01 * SUMIDA CDRH62-220 ** IRC 1206-R100F † AVX TPSD476K016 †† AVX TPSD107K010 3- to 4-Cell NiCd/NiMH to 2.5V Converter (VIN = 2.7V TO 6V) + + 3- OR 4-CELL NiCd OR NiMH R4 PWR VIN LBI 0.1µF VIN SHUTDOWN SHDN VOUT 2.5V 0.25A PGND + SENSE + ITH 3900pF RSENSE** 0.1Ω D1 MBR0520LT1 LTC1626 1k L1* 22µH SW LBO R3 CIN† 47µF 16V 1000pF CT 270pF SENSE – COUT†† 100µF 10V R1††† 10k VFB CT SGND 100pF R2††† 10k 1626 TA02 * SUMIDA CDRH62-220 ** IRC 1206-R100F † AVX TPSD476K016 †† AVX TPSD107K010 10 ††† FOR 3.3V: R1 = 15k, 1% R2 = 9.09k, 1% LTC1626 U TYPICAL APPLICATIONS Low Profile (3mm Maximum Height) 2.8V Converter VIN 3V TO 6V PWR VIN LBI 4.7µF††† CERAMIC VIN CIN† 22µF 16V TANT + SW L1* 15µH VOUT 2.8V 0.25A D1 MBR0520LT1 LBO SHDN SHUTDOWN RSENSE** 0.1Ω PGND + LTC1626 SENSE + ITH 1k 3900pF 1000pF CT 56pF R1 15k 1% SENSE – COUT†† 100µF 6.3V VFB CT SGND R2 12.1k 1% 100pF 1626 TA03 * COILCRAFT DO3308-153 ** IRC 1206-R100F † AVX TPSC226M016R0375 †† AVX TPSC107M006R0150 ††† MURATA GRM230Y5V475Z16 U PACKAGE DESCRIPTION Dimensions in inches (millimeters) unless otherwise noted. S Package 14-Lead Plastic Small Outline (Narrow 0.150) (LTC DWG # 05-08-1610) 0.337 – 0.344* (8.560 – 8.738) 14 13 12 11 10 9 8 0.228 – 0.244 (5.791 – 6.197) 0.150 – 0.157** (3.810 – 3.988) 1 0.010 – 0.020 × 45° (0.254 – 0.508) 2 3 4 5 6 0.053 – 0.069 (1.346 – 1.752) 0.008 – 0.010 (0.203 – 0.254) 0.004 – 0.010 (0.101 – 0.254) 0° – 8° TYP 0.016 – 0.050 0.406 – 1.270 0.014 – 0.019 (0.355 – 0.483) 7 0.050 (1.270) TYP *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. S14 0695 11 LTC1626 U TYPICAL APPLICATIONS Single Li-Ion to 3.3V Buck-Boost Converter L1B (VIN = 2.5V TO 4.2V) L1A SINGLE Li-ION CELL 3 2 TOP VIEW + + PWR VIN 1 4 PART NO. COILTRONICS DALE CTX33-4 LPT4545-330LA SHDN PGND 3 VFB SENSE + – COUT†† 100µF 10V 9.09k 1% 100pF 1626 TA05 SGND CT 200 350 500* 500* 500* 15k 1% L1B 33µH CT 75pF 3900pF VOUT 3.3V 2 4 ITH 1k L1A 33µH 1 D1 MBRS130LT1 LTC1626 I OUT (mA) 2.5 3.0 3.5 4.0 4.2 SW CIN† 100µF† 100µF 16V 16V + LBO SHUTDOWN MANUFACTURER VIN (V) VIN LBI L1A L1B 0.1µF SENSE + RSENSE* 0.1Ω 1000pF * IRC 1206-R100F † AVX TPSE107M016R0100 †† AVX TPSD107M010R0065 *DESIGN LIMIT 5V to 3.3V Converter VIN 5V + PWR VIN LBI 0.1µF VIN CIN† 100µF 10V RSENSE** 0.1Ω VOUT 3.3V 0.5A SW D1 MBRS130LT1 LBO SHUTDOWN L1* 47µH SHDN PGND + LTC1626 SENSE + ITH 1k 3900pF 1000pF CT 270pF SENSE – COUT†† 220µF 10V 15k 1% VFB CT SGND 9.09k 1% 100pF 1626 TA04 * COILCRAFT DO3316-473 ** IRC 1206-R100F † AVX TPSD107K010 †† AVX TPSE227K010 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LTC1174/LTC1174-3.3 LTC1174-5 High Efficiency Step-Down and Inverting DC/DC Converters Monolithic Switching Regulators, Burst Mode Operation LTC1265 1.2A, High Efficiency Step-Down DC/DC Converter Constant Off-Time Monolithic, Burst Mode Operation LT1375/LT1376 1.5A, 500kHz Step-Down Switching Regulators High Frequency, Small Inductor, High Efficiency LTC1435 High Efficiency, Low Noise, Synchronous Step-Down Converter 16-Pin Narrow SO and SSOP LTC1436/LTC1436-PLL High Efficiency, Low Noise, Synchronous Step-Down Converters 24-Pin Narrow and 28-Pin SSOP LTC1438/LTC1439 Dual, Low Noise, Synchronous Step-Down Converters Multiple Output Capability LTC1474/LTC1475 Low Quiescent Current Step-Down DC/DC Converters Monolithic, IQ = 10µA, 8-Pin MSOP 12 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417● (408)432-1900 FAX: (408) 434-0507● TELEX: 499-3977 ● www.linear-tech.com 1626f LT/TP 0398 4K • PRINTED IN USA LINEAR TECHNOLOGY CORPORATION 1997