LINER ELJPC3R3MF 1.2mhz/2.2mhz inverting dc/dc converters in thinsot Datasheet

LT1931/LT1931A
1.2MHz/2.2MHz Inverting
DC/DC Converters in ThinSOT
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FEATURES
DESCRIPTIO
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The LT®1931/LT1931A are the industry’s highest power
inverting SOT-23 current mode DC/DC converters. Both
parts include a 1A integrated switch allowing high current
outputs to be generated in a small footprint. The LT1931
switches at 1.2MHz while the LT1931A switches at 2.2MHz.
These high speeds enable the use of tiny, low cost
capacitors and inductors 2mm or less in height. The
LT1931 is capable of generating – 5V at 350mA or –12V
at 150mA from a 5V supply, while the LT1931A can
generate –5V at 300mA using significantly smaller inductors. Both parts are easy pin-for-pin upgrades for higher
power LT1611 applications.
■
■
■
■
■
■
■
■
■
Fixed Frequency 1.2MHz/2.2MHz Operation
Very Low Noise: 1mVP-P Output Ripple
– 5V at 350mA from 5V Input
–12V at 150mA from 5V Input
Uses Small Surface Mount Components
Wide Input Range: 2.6V to 16V
Low Shutdown Current: <1µA
Low VCESAT Switch: 400mV at 1A
Pin-for-Pin Compatible with the LT1611
Low Profile (1mm) ThinSOTTM Package
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APPLICATIO S
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■
■
■
■
Disk Drive MR Head Bias
Digital Camera CCD Bias
LCD Bias
GaAs FET Bias
Local Low Noise/Low Impedance Negative Supply
The LT1931/LT1931A operate in a dual inductor inverting
topology that filters both the input side and output side
current. Very low output voltage ripple approaching 1mVP-P
can be achieved when ceramic output capacitors are used.
Fixed frequency switching ensures a clean output free
from low frequency noise typically present with charge
pump solutions. The low impedance output remains within
1% of nominal during large load steps. The 36V switch
allows VIN to VOUT differential of up to 34V.
The LT1931/LT1931A are available in the 5-lead ThinSOT
package.
, LTC and LT are registered trademarks of Linear Technology Corporation. ThinSOT is a
trademark of Linear Technology Corporation. All other trademarks are the property of their
respective owners.
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TYPICAL APPLICATIO
C2
1µF
L1A
10µH
VIN
5V
Efficiency
L1B
10µH
100
95
VIN
SW
SHDN
C1
4.7µF
R1
29.4k
LT1931
C4
220pF
NFB
GND
R2
10k
VOUT
–5V
350mA
C3
22µF
90
EFFICIENCY (%)
D1
85
80
75
70
65
60
C1: TAIYO YUDEN X5R JMK212BJ475MG
C2: TAIYO YUDEN X5R LMK212BJ105MG
C3: TAIYO YUDEN X5R JMK325BJ226MM
D1: ON SEMICONDUCTOR MBR0520
L1: SUMIDA CLS62-100
Figure 1. 5V to –5V, 350mA Inverting DC/DC Converter
1931 F01
55
50
0
50
100 150 200 250
LOAD CURRENT (mA)
300
350
1931 TA01
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LT1931/LT1931A
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ABSOLUTE
RATI GS
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PACKAGE/ORDER I FOR ATIO
(Note 1)
VIN Voltage .............................................................. 16V
SW Voltage ................................................– 0.4V to 36V
NFB Voltage ............................................................. – 2V
Current Into NFB Pin ............................................ ±1mA
SHDN Voltage .......................................................... 16V
Maximum Junction Temperature .......................... 125°C
Operating Temperature Range (Note 2) .. – 40°C to 85°C
Storage Temperature Range ................. – 65°C to 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
TOP VIEW
SW 1
5 VIN
GND 2
4 SHDN
NFB 3
S5 PACKAGE
5-LEAD PLASTIC TSOT-23
TJMAX = 125°C, θJA = 150°C/ W
ORDER PART NUMBER
LT1931ES5
LT1931AES5
LT1931IS5
LT1931AIS5
S5 PART MARKING
LTRA
LTSP
LTBZF
LTBZG
Order Options Tape and Reel: Add #TR
Lead Free: Add #PBF Lead Free Tape and Reel: Add #TRPBF
Lead Free Part Marking: http://www.linear.com/leadfree/
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C.
VIN = 3V, VSHDN = VIN, unless otherwise noted. (Note 2)
PARAMETER
CONDITIONS
MIN
Minimum Operating Voltage
LT1931
TYP
MAX
2.45
2.6
Maximum Operating Voltage
MIN
LT1931A
TYP
MAX
2.45
16
Feedback Voltage
●
NFB Pin Bias Current
VNFB = –1.255V
Quiescent Current
VSHDN = 2.4V, Not Switching
– 1.275 – 1.255 – 1.235
– 1.280
– 1.230
●
UNITS
2.6
V
16
V
–1.275 –1.255 –1.235
–1.280
–1.230
V
V
4
8
8
16
µA
4.2
6
5.8
8
mA
Quiescent Current in Shutdown
VSHDN = 0V, VIN = 3V
0.01
1
0.01
1
µA
Reference Line Regulation
2.6V ≤ VIN ≤ 16V
0.01
0.05
0.01
0.05
%/V
1.4
1.6
1.8
1.6
2.2
2.6
2.9
MHz
MHz
75
82
Switching Frequency
Maximum Duty Cycle
Switch Current Limit
(Note 3)
Switch VCESAT
ISW = 1A
Switch Leakage Current
VSW = 5V
SHDN Input Voltage, High
1
0.85
1.2
●
●
84
90
1
1.2
2
400
600
0.01
1
2.4
1.2
2.5
A
400
600
mV
0.01
1
µA
2.4
SHDN Input Voltage, Low
SHDN Pin Bias Current
1
V
0.5
VSHDN = 3V
VSHDN = 0V
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: The LT1931E/LT1931AE are guaranteed to meet performance
specifications from 0°C to 70°C. Specifications over the – 40°C to 85°C
%
16
0
32
0.1
35
0
0.5
V
70
0.1
µA
µA
operating temperature range are assured by design, characterization and
correlation with statistical process controls. LT1931I/LT1931AI are
guaranteed over the –40°C to 85°C temperature range.
Note 3: Current limit guaranteed by design and/or correlation to static test.
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LT1931/LT1931A
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TYPICAL PERFOR A CE CHARACTERISTICS
Quiescent Current
Feedback Pin Voltage
7.0
90
NOT SWITCHING
80
6.5
LT1931A
5.5
5.0
4.5
LT1931
4.0
SHDN PIN CURRENT (µA)
–1.27
6.0
FEEDBACK VOLTAGE (V)
QUIESCENT CURRENT (mA)
Shutdown Pin Current
–1.28
–1.26
–1.25
–1.24
60
50
40
30
LT1931
20
0
3.0
–50
–25
0
50
25
TEMPERATURE (°C)
75
–1.22
–50
100
–25
0
25
50
TEMPERATURE (°C)
75
–10
100
0
Current Limit
1.2
VCESAT (V)
0.6
TA = 25°C
0.40
2.3
0.35
2.1
0.25
0.20
0.15
1.9
1.5
1.3
0.9
0.2
0.05
0.7
0
30
40 50 60 70
DUTY CYCLE (%)
80
0
90
0.2
0.4
0.6
0.8
SWITCH CURRENT (A)
1.0
1.2
LT1931
1.1
0.10
20
LT1931A
1.7
0.4
10
6
Oscillator Frequency
0.30
0.8
5
2.5
FREQUENCY (MHz)
TA = 25°C
1.4
1.0
3
4
2
SHDN PIN VOLTAGE (V)
1931 G03
Switch Saturation Voltage
0.45
1.6
1
1931 G02
1931 G01
CURRENT LIMIT (A)
LT1931A
70
10
–1.23
3.5
0
TA = 25°C
0.5
–50
–25
25
50
0
TEMPERATURE (°C)
100
1931 G06
1931 G05
1931 G04
75
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PI FU CTIO S
SW (Pin 1): Switch Pin. Connect inductor/diode here.
Minimize trace area at this pin to keep EMI down.
GND (Pin 2): Ground. Tie directly to local ground plane.
NFB (Pin 3): Feedback Pin. Reference voltage is –1.255V.
Connect resistive divider tap here. Minimize trace area.
The NFB bias current flows out of the pin. Set R1 and R2
according to:
For LT1931: R1 =
| VOUT | – 1.255
1.255
+ 4 • 10 – 6
R2
(
For LT1931A: R1 =
| VOUT | – 1.255
1.255
+ 8 • 10 – 6
R2
(
)
SHDN (Pin 4): Shutdown Pin. Tie to 2.4V or more to enable
device. Ground to shut down.
VIN (Pin 5): Input Supply Pin. Must be locally bypassed.
)
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LT1931/LT1931A
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BLOCK DIAGRA
VIN 5
VIN
R5
80k
R6
80k
1 SW
+
–
Q1
VOUT
CPL
(OPTIONAL)
Q2
x10
–
A1
gm
RC
Σ
RAMP
GENERATOR
+
COMPARATOR
A2
R
LATCH
S
DRIVER
Q3
Q
+
CC
R3
30k
R1
(EXTERNAL)
NFB
R2
(EXTERNAL)
0.01Ω
–
1.2MHz
OSCILLATOR
R4
150k
SHDN
3 NFB
4
SHUTDOWN
2 GND
1931 BD
Figure 2
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OPERATIO
The LT1931 uses a constant frequency, current mode
control scheme to provide excellent line and load regulation. Operation can be best understood by referring to the
Block Diagram in Figure 2. At the start of each oscillator
cycle, the SR latch is set, turning on the power switch Q3.
A voltage proportional to the switch current is added to a
stabilizing ramp and the resulting sum is fed into the
positive terminal of the PWM comparator A2. When this
voltage exceeds the level at the negative input of A2, the SR
latch is reset, turning off the power switch. The level at the
negative input of A2 is set by the error amplifier (gm) and
is simply an amplified version of the difference between
the feedback voltage and the reference voltage of –1.255V.
In this manner, the error amplifier sets the correct peak
current level to keep the output in regulation. If the error
amplifier’s output increases, more current is taken from
the output; if it decreases, less current is taken. One
function not shown in Figure 2 is the current limit. The
switch current is constantly monitored and not allowed to
exceed the nominal value of 1.2A. If the switch current
reaches 1.2A, the SR latch is reset regardless of the state
of comparator A2. This current limit protects the power
switch as well as various external components connected
to the LT1931.
The Block Diagram for the LT1931A is identical except that
the oscillator is 2.2MHz and resistors R3 to R6 are one-half
the LT1931 values.
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LT1931/LT1931A
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LT1931A AND LT1931 DIFFERENCES:
Switching Frequency
The key difference between the LT1931A and LT1931 is
the faster switching frequency of the LT1931A. At 2.2MHz,
the LT1931A switches at nearly twice the rate of the
LT1931. Care must be taken in deciding which part to use.
The high switching frequency of the LT1931A allows
smaller cheaper inductors and capacitors to be used in a
given application, but with a slight decrease in efficiency
and maximum output current when compared to the
LT1931. Generally, if efficiency and maximum output
current are critical, the LT1931 should be used. If application size and cost are more important, the LT1931A will be
the better choice. In many applications, tiny inexpensive
chip inductors can be used with the LT1931A, reducing
solution cost.
core losses at frequencies above 1MHz are much lower for
ferrite cores than for powdered-iron units. When using
coupled inductors, choose one that can handle at least 1A
of current without saturating, and ensure that the inductor
has a low DCR (copper-wire resistance) to minimize I2R
power losses. If using uncoupled inductors, each inductor
need only handle one-half of the total switch current so
that 0.5A per inductor is sufficient. A 4.7µH to 15µH
coupled inductor or a 15µH to 22µH uncoupled inductor
will usually be the best choice for most LT1931 designs.
For the LT1931A, a 2.2µH to 4.7µH coupled inductor or a
3.3µH to 10µH uncoupled inductor will usually suffice. In
certain applications such as the “Charge Pump” inverting
DC/DC converter, only a single inductor is used. In this
case, the inductor must carry the entire 1A switch current.
Table 1. Recommended Inductors—LT1931
L
(µH)
Size
(L × W × H) mm
CLS62-100
CR43-150
CR43-220
10
15
22
6.8 × 6.6 × 2.5
4.5 × 4.0 × 3.2
Sumida
(847) 956-0666
www.sumida.com
CTX10-1
CTX15-1
10
15
8.9 × 11.4 × 4.2
Coiltronics
(407) 241-7876
www. coiltronics.com
LQH3C100K24
LQH4C150K04
10
15
3.2 × 2.5 × 2.0
Murata
(404) 436-1300
www.murata.com
PART
Duty Cycle
The maximum duty cycle (DC) of the LT1931A is 75%
compared to 84% for the LT1931. The duty cycle for a
given application using the dual inductor inverting topology is given by:
| VOUT |
DC =
| VIN | + | VOUT |
For a 5V to –5V application, the DC is 50% indicating that
the LT1931A can be used. A 5V to –16V application has a
DC of 76.2% making the LT1931 the right choice. The
LT1931A can still be used in applications where the DC, as
calculated above, is above 75%. However, the part must
be operated in the discontinuous conduction mode so that
the actual duty cycle is reduced.
INDUCTOR SELECTION
Several inductors that work well with the LT1931 are listed
in Table 1 and those for the LT1931A are listed in Table 2.
Besides these, there are many other inductors that can be
used. Consult each manufacturer for detailed information
and for their entire selection of related parts. Ferrite core
inductors should be used to obtain the best efficiency, as
VENDOR
Table 2. Recommended Inductors—LT1931A
PART
L
(µH)
Size
(L × W × H) mm
ELJPC3R3MF
ELJPC4R7MF
3.3
4.7
2.5 × 2.0 × 1.6
Panasonic
(408) 945-5660
www.panasonic.com
CLQ4D10-4R71
CLQ4D10-6R82
4.7
6.8
7.6 × 4.8 × 1.8
Sumida
(847) 956-0666
www.sumida.com
LB20164R7M
LB20163R3M
4.7
3.3
2.0 × 1.6 × 1.6
Taiyo Yuden
(408) 573-4150
www.t-yuden.com
LQH3C4R7K24
LQH4C100K24
4.7
10
3.2 × 2.5 × 2.0
Murata
(404) 436-1300
www.murata.com
VENDOR
1Use drawing #5382-T039
2Use drawing #5382-T041
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LT1931/LT1931A
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APPLICATIO S I FOR ATIO
The inductors shown in Table 2 for use with the LT1931A
were chosen for their small size. For better efficiency, use
similar valued inductors with a larger volume. For instance, the Sumida CR43 series, in values ranging from
3.3µH to 10µH, will give a LT1931A application a few
percentage points increase in efficiency.
CAPACITOR SELECTION
Low ESR (equivalent series resistance) capacitors should
be used at the output to minimize the output ripple voltage.
Multilayer ceramic capacitors are an excellent choice, as
they have an extremely low ESR and are available in very
small packages. X5R dielectrics are preferred, followed by
X7R, as these materials retain their capacitance over wide
voltage and temperature ranges. A 10µF to 22µF output
capacitor is sufficient for most LT1931 applications while
a 4.7µF to 10µF capacitor will suffice for the LT1931A.
Solid tantalum or OS-CON capacitors can be used, but
they will occupy more board area than a ceramic and will
have a higher ESR. Always use a capacitor with a sufficient
voltage rating.
Ceramic capacitors also make a good choice for the input
decoupling capacitor, which should be placed as close as
possible to the LT1931/LT1931A. A 1µF to 4.7µF input
capacitor is sufficient for most applications. Table 3 shows
a list of several ceramic capacitor manufacturers. Consult
the manufacturers for detailed information on their entire
selection of ceramic parts.
Table 3. Ceramic Capacitor Manufacturers
Taiyo Yuden
(408) 573-4150
www.t-yuden.com
AVX
(803) 448-9411
www.avxcorp.com
Murata
(714) 852-2001
www.murata.com
The decision to use either low ESR (ceramic) capacitors or
the higher ESR (tantalum or OS-CON) capacitors can
effect the stability of the overall system. The ESR of any
capacitor, along with the capacitance itself, contributes a
zero to the system. For the tantalum and OS-CON capacitors, this zero is located at a lower frequency due to the
higher value of the ESR, while the zero of a ceramic
capacitor is at a much higher frequency and can generally
be ignored.
A phase lead zero can be intentionally introduced by
placing a capacitor (C4) in parallel with the resistor (R1)
between VOUT and VNFB as shown in Figure 1. The
frequency of the zero is determined by the following
equation.
ƒZ =
1
2π • R1 • C4
By choosing the appropriate values for the resistor and
capacitor, the zero frequency can be designed to improve
the phase margin of the overall converter. The typical
target value for the zero frequency is between 20kHz to
60kHz. Figure 3 shows the transient response of the
inverting converter from Figure 1 without the phase lead
capacitor C4. The phase margin is reduced as evidenced
by more ringing in both the output voltage and inductor
current. A 220pF capacitor for C4 results in better phase
margin, which is revealed in Figure 4 as a more damped
response and less overshoot. Figure 5 shows the transient
response when a 22µF tantalum capacitor with no phase
lead capacitor is used on the output. The higher output
voltage ripple is revealed in the upper waveform as a
thicker line. The transient response is adequate which
implies that the ESR zero is improving the phase margin.
VOUT
20mV/DIV
AC COUPLED
IL1A + IL1B
0.5A/DIV
AC COUPLED
LOAD 200mA
CURRENT 100mA
100µs/DIV
1931 F03
Figure 3. Transient Response of Inverting Converter
Without Phase Lead Capacitor
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LT1931/LT1931A
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APPLICATIO S I FOR ATIO
VOUT
20mV/DIV
AC COUPLED
VOUT
2V/DIV
IL1A + IL1B
0.5A/DIV
AC COUPLED
IIN
0.5A/DIV
AC COUPLED
LOAD 200mA
CURRENT 100mA
VSHDN 5V
0V
100µs/DIV
1931 F04
500µs/DIV
Figure 4. Transient Response of Inverting Converter
with 220pF Phase Lead Capacitor
Figure 6. Start-Up Waveforms for 5V to – 5V Application
(Figure 1). No Soft-Start Circuit. VOUT Reaches – 5V in
500µs; Input Current Peaks at 800mA
VOUT
0.1V/DIV
AC COUPLED
regulator tries to charge up the output capacitor as quickly
as possible, which results in a large inrush current. Figure 6 shows a typical oscillograph of the start-up waveform for the application of Figure 1 starting into a load of
33Ω. The lower waveform shows SHDN being pulsed
from 0V to 5V. The middle waveform shows the input
current, which reaches as high as 0.8A. The total time
required for the output to reach its final value is approximately 500µs. For some applications, this initial inrush
current may not be acceptable. If a longer start-up time is
acceptable, a soft-start circuit consisting of RSS and CSS,
as shown in Figure 7, can be used to limit inrush current
to a lower value. Figure 8 shows the relevant waveforms
with RSS = 15k and CSS = 33nF. Input current, measured
at VIN, is limited to a peak value of 0.5A as the time required
to reach final value increases to 1ms. In Figure 9, CSS is
IL1A + IL1B
0.5A/DIV
AC COUPLED
LOAD 200mA
CURRENT 100mA
50µs/DIV
1931 F05
Figure 5. Transient Response of Inverting Converter with 22µF
Tantalum Output Capacitor and No Phase Lead Capacitor
START-UP/SOFT-START
For most LT1931/LT1931A applications, the start-up inrush current can be high. This is an inherent feature of
switching regulators in general since the feedback loop is
saturated due to VOUT being far from its final value. The
CURRENT
PROBE
+
RSS
15k
C2
1µF
L1A
10µH
VIN
5V
VIN
SW
SHDN
R1
29.4k
NFB
GND
D2
1N4148
CSS
33nF/68nF
VOUT
L1B
10µH
D1
C1
4.7µF
LT1931
VSS
1931 F06
R2
10k
C1: TAIYO YUDEN X5R JMK212BJ475MG
C2: TAIYO YUDEN X5R LMK212BJ105MG
C3: TAIYO YUDEN XR5 JMK325BJ226MM
D1: ON SEMICONDUCTOR MBR0520
L1: SUMIDA CLS62-100
C4
220pF
VOUT
–5V
C3
22µF
1931 F07
Figure 7. RSS and CSS at SHDN Pin Provide Soft-Start to LT1931 Inverting Converter
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LT1931/LT1931A
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APPLICATIO S I FOR ATIO
DIODE SELECTION
VOUT
2V/DIV
IIN
0.5A/DIV
AC COUPLED
VSS 5V
0V
200µs/DIV
1931 F08
A Schottky diode is recommended for use with the LT1931/
LT1931A. The Motorola MBR0520 is a very good choice.
Where the input to output voltage differential exceeds 20V,
use the MBR0530 (a 30V diode). These diodes are rated to
handle an average forward current of 0.5 A. In applications
where the average forward current of the diode exceeds
0.5A, a Microsemi UPS5817 rated at 1A is recommended.
Figure 8. RSS = 15k, CSS = 33nF; VOUT Reaches – 5V in 1ms;
Input Current Peaks at 500mA
LAYOUT HINTS
The high-speed operation of the LT1931/LT1931A demands careful attention to board layout. You will not get
advertised performance with careless layout. Figure 10
shows the recommended component placement. The
ground cut at the cathode of D1 is essential for low noise
operation.
VOUT
2V/DIV
IIN
0.5A/DIV
AC COUPLED
VSS 5V
0V
500µs/DIV
1931 F09
Figure 9. RSS = 15k, CSS = 68nF; VOUT Reaches – 5V in 1.6ms;
Input Current Peaks at 350mA
L1B
L1A
C1
D1
+
C2
VIN
C3
+
increased to 68nF, resulting in a lower peak input current
of 350mA with a VOUT ramp time of 1.6ms. CSS or RSS can
be increased further for an even slower ramp, if desired.
Diode D2 serves to quickly discharge CSS when VSS is
driven low to shut down the device. D2 can be omitted,
resulting in a “soft-stop” slow discharge of the output
capacitor.
–VOUT
5
1
2
3
4
SHUTDOWN
R2
GND
R1
1931 F10
Figure 10. Suggested Component Placement.
Note Cut in Ground Copper at D1’s Cathode
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LT1931/LT1931A
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TYPICAL APPLICATIO S
5V to –12V Inverting Converter
C2
1µF
L1A
10µH
VIN
5V
Efficiency
100
L1B
10µH
95
90
D1
SHDN
R1
84.5k
LT1931
C1
4.7µF
VOUT
–12V
150mA
SW
C3
10µF
NFB
GND
R2
10k
85
EFFICIENCY (%)
VIN
80
75
70
65
60
55
C1: TAIYO YUDEN X5R JMK212BJ475MG
C2: TAIYO YUDEN X5R TMK316BJ105ML
C3: TAIYO YUDEN X5R EMK325BJ106MM
D1: ON SEMICONDUCTOR MBR0520
L1: SUMIDA CLS62-100
50
1931 TA02
0
25
75
100
50
LOAD CURRENT (mA)
125
150
1931 TA03
5V to – 5V Inverting Converter Using Uncoupled Inductors
C2
1µF
L1
10µH
VIN
5V
L2
10µH
D1
VIN
SHDN
C1
4.7µF
VOUT
–5V
300mA
SW
R1
29.4k
LT1931
220pF
C3
22µF
NFB
GND
R2
10k
C1: TAIYO YUDEN X5R JMK212BJ475MG
C2: TAIYO YUDEN X5R LMK212BJ105MG
C3: TAIYO YUDEN X5R JMK212BJ226MM
D1: ON SEMICONDUCTOR MBR0520
L1, L2: MURATA LQH3C100K04
1931 TA04
2.2MHz, 5V to – 5V Inverting Converter
C2
1µF
L1
4.7µH
VIN
5V
Efficiency
80
L2
4.7µH
VIN
SW
SHDN
C1
4.7µF
R1
28.7k
LT1931A
NFB
GND
R2
10k
C4
180pF
VOUT
–5V
300mA
C3
4.7µF
EFFICIENCY (%)
75
D1
70
65
60
55
C1: TAIYO YUDEN X5R JMK212BJ475MG
C2: TAIYO YUDEN X5R LMK212BJ105MG
C3: TAIYO YUDEN X5R JMK212BJ475MG
D1: ON SEMICONDUCTOR MBR0520
L1, L2: MURATA LQH3C4R7M24
1931 TA05a
50
0
50
100 150 200 250
LOAD CURRENT (mA)
300
350
1931 TA05b
1931fa
9
LT1931/LT1931A
U
TYPICAL APPLICATIO S
2.2MHz, 5V to –5V Converter Uses Tiny Chip Inductors
C2
1µF
L1
3.3µH
VIN
5V
Efficiency
80
L2
3.3µH
D1
VIN
SW
SHDN
C1
2.2µF
R1
28.7k
LT1931A
C4
68pF
C3
4.7µF
NFB
GND
VOUT
–5V
200mA
R2
10k
EFFICIENCY (%)
75
70
65
60
55
C1: TAIYO YUDEN X5R JMK212BJ225MG
C2: TAIYO YUDEN X5R LMK212BJ105MG
C3: TAIYO YUDEN X5R JMK212BJ475MG
D1: ON SEMICONDUCTOR MBR0520
L1, L2: PANASONIC ELJPC3R3MF
50
1931 TA06a
0
50
100
150
200
LOAD CURRENT (mA)
250
1931 TA06b
SLIC Power Supply with – 33V and – 68V Outputs, Uses Soft-Start
VIN
12V
L1
22µH
C1
4.7µF
16V
VIN
R1
1Ω
C2
1µF
35V
SW
D1
RSS
15k
VSS
LT1931
SHDN
3
2
NFB
GND
CSS
68nF
1
R2
1k
COM
C4
4.7µF
35V
VOUT1
–33V
100mA*
R3
25.5k
C6
1000pF
R4
2.7k
C3
1µF
35V
3
*TOTAL OUTPUT POWER NOT TO EXCEED 3.3W
C1 TO C5: X5R OR X7R
D1, D2: BAV99 OR EQUIVALENT
L1: SUMIDA CR43-220
D2
2
1
C5
4.7µF
35V
1931 TA08
VOUT2
–66V
48mA*
1931fa
10
LT1931/LT1931A
U
TYPICAL APPLICATIO S
SLIC Power Supply with – 21.6V and – 65V Outputs, Uses Soft-Start
L1
10µH
VIN
5V
C1
4.7µF
16V
R1
1Ω
VIN
C2
1µF
35V
SW
D1
RSS
15k
3
LT1931
SHDN
VSS
2
NFB
GND
1
VOUT1
–21.6V
48mA*
R3
16.2k
R2
1k
C8
1000pF
CSS
68nF
COM
C5
4.7µF
25V
C3
1µF
35V
R4
2.7k
D2
3
*TOTAL OUTPUT POWER NOT TO EXCEED 1.3W
C1 TO C7: X5R OR X7R
D1, D2: BAV99 OR EQUIVALENT
L1: SUMIDA CR43-100
2
1
C4
1µF
35V
C6
4.7µF
25V
D3
3
2
1
C7
4.7µF
25V
1931 TA09
VOUT2
– 65V
20mA*
U
PACKAGE DESCRIPTIO
S5 Package
5-Lead Plastic TSOT-23
(Reference LTC DWG # 05-08-1635)
0.62
MAX
0.95
REF
2.90 BSC
(NOTE 4)
1.22 REF
3.85 MAX 2.62 REF
1.4 MIN
2.80 BSC
1.50 – 1.75
(NOTE 4)
PIN ONE
RECOMMENDED SOLDER PAD LAYOUT
PER IPC CALCULATOR
0.30 – 0.45 TYP
5 PLCS (NOTE 3)
0.95 BSC
0.80 – 0.90
0.20 BSC
0.01 – 0.10
1.00 MAX
DATUM ‘A’
0.30 – 0.50 REF
0.09 – 0.20
(NOTE 3)
NOTE:
1. DIMENSIONS ARE IN MILLIMETERS
2. DRAWING NOT TO SCALE
3. DIMENSIONS ARE INCLUSIVE OF PLATING
4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
5. MOLD FLASH SHALL NOT EXCEED 0.254mm
6. JEDEC PACKAGE REFERENCE IS MO-193
1.90 BSC
S5 TSOT-23 0302 REV B
1931fa
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
11
LT1931/LT1931A
U
TYPICAL APPLICATIO
2.2MHz, 12V to – 5V Converter Uses Low Profile Coupled Inductor
C2
0.1µF
L1A
4.7µH
VIN
12V
L1B
4.7µH
D1
VIN
SHDN
R1
28.7k
LT1931A
C1
2.2µF
VOUT
–5V
450mA
SW
C3
4.7µF
NFB
GND
R2
10k
C1: TAIYO YUDEN Y5V EMK212F225ZG
C2: 0.1µF 25V X5R
C3: TAIYO YUDEN X5R JMK212BJ475MG
D1: ON SEMICONDUCTOR MBR0520
L1: SUMIDA CLQ4D10-4R7 DRAWING #5382-T039
1931 TA07a
Efficiency
80
EFFICIENCY (%)
75
70
65
60
55
50
0
100
200
300
400
LOAD CURRENT (mA)
500
1931 TA07b
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT1307
Single Cell Micropower 600kHz PWM DC/DC Converter
3.3V at 75mA from One Cell, MSOP Package
LT1316
Burst ModeTM Operation DC/DC with Programmable Current Limit
1.5V Minimum, Precise Control of Peak Current Limit
LT1317
2-Cell Micropower DC/DC with Low-Battery Detector
3.3V at 200mA from Two Cells, 600kHz Fixed Frequency
LT1610
Single Cell Micropower DC/DC Converter
3V at 30mA from 1V, 1.7MHz Fixed Frequency
LT1611
Inverting 1.4MHz Switching Regulator in 5-Lead ThinSOT
–5V at 150mA from 5V Input. Tiny SOT-23 Package
LT1613
1.4MHz Switching Regulator in 5-Lead ThinSOT
5V at 200mA from 3.3V Input. Tiny SOT-23 Package
LT1615
Micropower Constant Off-Time DC/DC Converter in 5-Lead ThinSOT
20V at 12mA from 2.5V. Tiny SOT-23 Package
LT1617
Micropower Inverting DC/DC Converter in 5-Lead ThinSOT
–15V at 12mA from 2.5V. Tiny SOT-23 Package
LT1930/LT1930A
1.2MHz/2.2MHz, 1A Switching Regulators in 5-Lead ThinSOT
5V at 450mA from 3.3V Input. Tiny SOT-23 Package
Burst Mode operation is a trademark of Linear Technology Corporation.
1931fa
12
Linear Technology Corporation
LT/LT 1005 REV A • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408)432-1900 ● FAX: (408) 434-0507 ● www.linear-tech.com
© LINEAR TECHNOLOGY CORPORATION 2000