LINER LTC3809IMSE-1-TRPBF No rsensetm, low input voltage, synchronous dc/dc controller with output tracking Datasheet

LTC3809-1
No RSENSETM, Low Input
Voltage, Synchronous DC/DC
Controller with Output Tracking
FEATURES
DESCRIPTION
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The LTC®3809-1 is a synchronous step-down switching
regulator controller that drives external complementary
power MOSFETs using few external components. The
constant frequency current mode architecture with MOSFET
VDS sensing eliminates the need for a current sense resistor
and improves efﬁciency.
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Programmable Output Voltage Tracking
No Current Sense Resistor Required
Constant Frequency Current Mode Operation for
Excellent Line and Load Transient Response
Wide VIN Range: 2.75V to 9.8V
Wide VOUT Range: 0.6V to VIN
0.6V ±1.5% Reference
Low Dropout Operation: 100% Duty Cycle
Selectable Burst Mode®/Pulse-Skipping/Forced
Continuous Operation
Auxiliary Winding Regulation
Internal Soft-Start Circuitry
Selectable Maximum Peak Current Sense Threshold
Output Overvoltage Protection
Micropower Shutdown: IQ = 9μA
Tiny Thermally Enhanced Leadless (3mm × 3mm)
DFN and 10-lead MSOP Packages
Optional Burst Mode operation provides high efﬁciency
operation at light loads. 100% duty cycle provides low
dropout operation, extending operating time in batterypowered systems. Burst Mode is inhibited when the MODE
pin is pulled low to reduce noise and RF interference.
The LTC3809-1 allows either coincident or ratiometric
output voltage tracking. Switching frequency is ﬁxed at
550kHz. Fault protection is provided by an overvoltage
comparator and a short-circuit current limit comparator.
The LTC3809-1 is available in tiny footprint thermally
enhanced DFN and 10-lead MSOP packages.
APPLICATIONS
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, LT, LTC and LTM are registered trademarks of Linear Technology Corporation. Burst
is a registered trademark of Linear Technology Corporation. No RSENSE is a trademark of
Linear Technology Corporation. All other trademarks are the property of their respective
owners. Protected by U.S. Patents including 5481178, 5929620, 6580258, 6304066,
5847554, 6611131, 6498466. Other Patents pending.
1- or 2-Cell Lithium-Ion Powered Devices
Notebook and Palmtop Computers, PDAs
Portable Instruments
Distributed DC Power Systems
TYPICAL APPLICATION
Efﬁciency and Power Loss vs Load Current
High Efﬁciency, 550kHz Step-Down Converter
100
10μF
15k
187k
TG
LTC3809-1
470pF
2.2μH
VFB
SW
ITH
BG
VOUT
2.5V
2A
EFFICIENCY (%)
59k
VIN = 5V
1k
VIN = 4.2V
80
100
TYPICAL POWER
LOSS (VIN = 4.2V)
70
10
47μF
RUN
60
1
GND
38091 TA01
POWER LOSS (mW)
MODE
VIN = 3.3V
90
VIN
IPRG
10k
EFFICIENCY
VIN
2.75V TO 9.8V
FIGURE 8 CIRCUIT
VOUT = 2.5V
50
1
10
100
1k
LOAD CURRENT (mA)
0.1
10k
38091 TA02
38091fc
1
LTC3809-1
ABSOLUTE MAXIMUM RATINGS
(Note 1)
Input Supply Voltage (VIN) ........................ –0.3V to 10V
RUN, TRACK/SS, MODE,
IPRG Voltages ............................... –0.3V to (VIN + 0.3V)
VFB, ITH Voltages ...................................... –0.3V to 2.4V
SW Voltage ......................... –2V to VIN + 1V (10V Max)
TG, BG Peak Output Current (<10μs) ......................... 1A
Operating Temperature Range (Note 2)....–40°C to 85°C
Storage Ambient Temperature Range
DFN....................................................–65°C to 125°C
MSOP ................................................–65°C to 150°C
Junction Temperature (Note 3) ............................ 125°C
Lead Temperature (Soldering, 10 sec)
MSOP Package ................................................. 300°C
PIN CONFIGURATION
TOP VIEW
TOP VIEW
MODE
1
10 SW
TRACK/SS
2
9 VIN
VFB
3
ITH
4
7 BG
RUN
5
6 IPRG
11
MODE
TRACK/SS
VFB
ITH
RUN
8 TG
1
2
3
4
5
11
10
9
8
7
6
SW
VIN
TG
BG
IPRG
MSE PACKAGE
10-LEAD PLASTIC MSOP
DD PACKAGE
10-LEAD (3mm s 3mm) PLASTIC DFN
TJMAX = 125°C, θJA = 40°C/W
EXPOSED PAD (PIN 11) IS GND
(MUST BE SOLDERED TO PCB)
TJMAX = 125°C, θJA = 43°C/W
EXPOSED PAD (PIN 11) IS GND
(MUST BE SOLDERED TO PCB)
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3809EDD-1#PBF
LTC3809EDD-1#TRPBF
LBQZ
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 85°C
LTC3809IDD-1#PBF
LTC3809IDD-1#TRPBF
LBQZ
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 85°C
LTC3809EMSE-1#PBF
LTC3809EMSE-1#TRPBF
LTBQV
10-Lead Plastic MSOP
–40°C to 85°C
LTC3809IMSE-1#PBF
LTC3809IMSE-1#TRPBF
LTBQV
10-Lead Plastic MSOP
–40°C to 85°C
LEAD BASED FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3809EDD-1
LTC3809EDD-1#TR
LBQZ
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 85°C
LTC3809IDD-1
LTC3809IDD-1#TR
LBQZ
10-Lead (3mm × 3mm) Plastic DFN
–40°C to 85°C
LTC3809EMSE-1
LTC3809EMSE-1#TR
LTBQV
10-Lead Plastic MSOP
–40°C to 85°C
LTC3809IMSE-1
LTC3809IMSE-1#TR
LTBQV
10-Lead Plastic MSOP
–40°C to 85°C
Consult LTC Marketing for parts speciﬁed with wider operating temperature ranges. *The temperature grade is identiﬁed by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel speciﬁcations, go to: http://www.linear.com/tapeandreel/
38091fc
2
LTC3809-1
ELECTRICAL CHARACTERISTICS
The l indicates speciﬁcations which apply over the full operating
temperature range, otherwise speciﬁcations are at TA = 25°C. VIN = 4.2V unless otherwise noted.
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
350
105
9
9
500
150
20
20
μA
μA
μA
μA
1.95
2.15
2.25
2.45
2.55
2.75
V
V
0.8
1.1
1.4
V
0.65
1
1.35
μA
Main Control Loops
Input DC Supply Current
Normal Operation
Sleep Mode
Shutdown
UVLO
(Note 4)
Undervoltage Lockout Threshold (UVLO)
VIN Falling
VIN Rising
RUN = 0V
VIN = UVLO Threshold –200mV
l
l
Shutdown Threshold of RUN Pin
Start-Up Current Source
TRACK/SS = 0V
Regulated Feedback Voltage
(Note 5)
0.6
0.609
V
Output Voltage Line Regulation
2.75V < VIN < 9.8V (Note 5)
0.01
0.04
%/V
Output Voltage Load Regulation
ITH = 0.9V (Note 5)
ITH = 1.7V
0.1
–0.1
0.5
–0.5
%
%
VFB Input Current
(Note 5)
9
50
nA
Overvoltage Protect Threshold
Measured at VFB
0.68
0.7
V
l
0.591
0.66
Overvoltage Protect Hysteresis
20
Auxiliary Feedback Threshold
0.325
0.4
mV
0.475
V
Top Gate (TG) Drive Rise Time
CL = 3000pF
40
ns
Top Gate (TG) Drive Fall Time
CL = 3000pF
40
ns
Bottom Gate (BG) Drive Rise Time
CL = 3000pF
50
ns
Bottom Gate (BG) Drive Fall Time
CL = 3000pF
40
ns
Maximum Current Sense Voltage (ΔVSENSE(MAX))
(VIN – SW)
IPRG = Floating (Note 6)
IPRG = 0V (Note 6)
IPRG = VIN (Note 6)
Soft-Start Time (Internal)
Time for VFB to Ramp from 0.05V to 0.55V
Oscillator Frequency
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC3809E-1 is guaranteed to meet speciﬁed performance
from 0°C to 85°C. Speciﬁcations over the –40°C to 85°C operating range
are assured by design characterization, and correlation with statistical
process controls. The LTC3809I-1 is guaranteed to meet speciﬁed
performance over the full –40°C to 85°C operating temperature range.
l
l
l
110
70
185
125
85
204
140
100
223
mV
mV
mV
0.5
0.74
0.9
ms
480
550
600
kHz
Note 3: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formula:
TJ = TA + (PD • θJA °C/W)
Note 4: Dynamic supply current is higher due to gate charge being
delivered at the switching frequency.
Note 5: The LTC3809-1 is tested in a feedback loop that servos ITH to
a speciﬁed voltage and measures the resultant VFB voltage.
Note 6: Peak current sense voltage is reduced dependent on duty cycle
to a percentage of value as shown in Figure 1.
38091fc
3
LTC3809-1
TYPICAL PERFORMANCE CHARACTERISTICS
Efﬁciency vs Load Current
FIGURE 8 CIRCUIT
FIGURE 8 CIRCUIT
95 VIN = 5V, VOUT = 2.5V
EFFICIENCY (%)
EFFICIENCY (%)
VOUT = 1.2V
80
VOUT = 1.8V
75
85
BURST MODE
(MODE = VIN)
80
75
FORCED
CONTINUOUS
(MODE = 0V)
70
65
70
60
65
MODE = VIN
VIN = 5V
60
1
1
10k
40
20
–20
50
10
100
1k
LOAD CURRENT (mA)
60
0
PULSE SKIPPING
(MODE = 0.6V)
55
Burst Mode OPERATION
(ITH RISING)
Burst Mode OPERATION
(ITH FALLING)
FORCED CONTINUOUS
MODE
PULSE SKIPPING
MODE
80
90
VOUT = 3.3V
85
100
100
VOUT = 2.5V
95
90
Maximum Current Sense Voltage
vs ITH Pin Voltage
Efﬁciency vs Load Current
CURRENT LIMIT (%)
100
TA = 25°C, unless otherwise noted.
10
100
1k
LOAD CURRENT (mA)
38091 G01
10k
0.5
38091 G02
Load Step
(Burst Mode Operation)
Load Step
(Forced Continuous Mode)
Load Step
(Pulse-Skipping Mode)
VOUT
200mV/DIV
AC COUPLED
VOUT
200mV/DIV
AC COUPLED
IL
2A/DIV
IL
2A/DIV
IL
2A/DIV
38091 G04
100μs/DIV
VIN = 3.3V
VOUT = 1.8V
ILOAD = 300mA TO 3A
MODE = 0V
FIGURE 8 CIRCUIT
Start-Up with Internal Soft-Start
(TRACK/SS = VIN)
200μs/DIV
VIN = 4.2V
RLOAD = 1
FIGURE 8 CIRCUIT
2
38091 G03
VOUT
200mV/DIV
AC COUPLED
100μs/DIV
VIN = 3.3V
VOUT = 1.8V
ILOAD = 300mA TO 3A
MODE = V IN
FIGURE 8 CIRCUIT
1
1.5
ITH VOLTAGE (V)
38091 G05
100μs/DIV
VIN = 3.3V
VOUT = 1.8V
ILOAD = 300mA TO 3A
MODE = VFB
FIGURE 8 CIRCUIT
38091 G06
Start-Up with External Soft-Start
(CSS = 10nF)
VOUT
1.8V
VOUT
1.8V
500mV/DIV
500mV/DIV
38091 G07
1ms/DIV
38091 G08
VIN = 4.2V
RLOAD = 1
FIGURE 8 CIRCUIT
38091fc
4
LTC3809-1
TYPICAL PERFORMANCE CHARACTERISTICS
Start-Up with Coincident Tracking
(VOUT = 0V at 0s)
Start-Up with Coincident Tracking
(VOUT = 0.8V at 0s)
Vx
2.5V
Vx
2.5V
VOUT
1.8V
VOUT
1.8V
VOUT
1.8V
500mV/DIV
500mV/DIV
500mV/DIV
38091 G10
10ms/DIV
VIN = 4.2V
RTA = 590
RTB = 1.18k
FIGURE 8 CIRCUIT
VIN = 4.2V
RTA = 590
RTB = 1.69k
FIGURE 8 CIRCUIT
Undervoltage Lockout Threshold
vs Temperature
Shutdown (RUN) Threshold
vs Temperature
2.55
0.606
1.20
2.50
0.604
VIN RISING
1.15
0.600
0.598
RUN VOLTAGE (V)
INPUT VOLTAGE (V)
2.45
0.602
2.40
2.35
2.30
VIN FALLING
2.20
0
20 40 60
TEMPERATURE (°C)
80
100
2.15
–60 –40 –20
20 40 60
0
TEMPERATURE (°C)
38091 G012
1.04
IPRG = FLOAT
130
125
120
115
–60 –40 –20
20 40 60
0
TEMPERATURE (°C)
100
1.00
–60 –40 –20
20 40 60
0
TEMPERATURE (°C)
80
100
38091 G14
TRACK/SS Start-Up Current
vs Temperature
TRACK/SS START-UP CURRENT (μA)
135
80
38091 G13
Maximum Current Sense
Threshold vs Temperature
MAXIMUM CURRENT SENSE THRESHOLD (mV)
0.594
–60 –40 –20
1.10
1.05
2.25
0.596
38091 G11
10ms/DIV
VIN = 4.2V
RTA = 590
RTB = 1.18k
FIGURE 8 CIRCUIT
Regulated Feedback Voltage
vs Temperature
FEEDBACK VOLTAGE (V)
Start-Up with Ratiometric Tracking
(VOUT = 0V at 0s)
Vx
2.5V
38091 G09
10ms/DIV
TA = 25°C, unless otherwise noted.
80
100
38091 G15
TRACK/SS = 0V
1.02
1.00
0.98
0.96
0.94
–60 –40 –20
0
20 40 60
TEMPERATURE (°C)
80
100
38091 G16
38091fc
5
LTC3809-1
TYPICAL PERFORMANCE CHARACTERISTICS
Oscillator Frequency
vs Input Voltage
Shutdown Quiescent Current
vs Input Voltage
10
5
18
8
4
16
6
4
2
0
–2
–4
–6
–8
–10
–60 –40 –20
0
20 40 60
TEMPERATURE (°C)
3
SHUTDOWN CURRENT (μA)
NORMALIZED FREQUENCY SHIFT (%)
NORMALIZED FREQUENCY (%)
Oscillator Frequency
vs Temperature
TA = 25°C, unless otherwise noted.
2
1
0
–1
–2
–3
100
3
2
4
7
8
5
6
INPUT VOLTAGE (V)
38091 G17
10
8
6
4
0
9
10
2
3
4
8
7
6
5
INPUT VOLTAGE (V)
38091 G18
9
10
38091 G19
TRACK/SS Start-Up Current
vs TRACK/SS Voltage
Sleep Current vs Input Voltage
130
TRACK/SS STARTUP CURRENT (μA)
1.04
120
SLEEP CURRENT (μA)
12
2
–4
–5
80
14
110
100
90
80
70
2
3
4
8
7
6
5
INPUT VOLTAGE (V)
1.00
0.96
0.92
0.88
0.84
9
10
38091 G20
0
0.1
0.2 0.3 0.4 0.5
TRACK/SS VOLTAGE (V)
0.6
0.7
38091 G21
38091fc
6
LTC3809-1
PIN FUNCTIONS
MODE (Pin 1): This pin performs two functions: 1) auxiliary
winding feedback input, and 2) Burst Mode operation,
pulse skipping or forced continuous mode select.
To select Burst Mode operation at light loads, tie this
pin to VIN. Grounding this pin selects forced continuous
operation which allows the inductor current to reverse.
Tying this pin to VFB selects pulse-skipping mode. Do not
leave this pin ﬂoating.
TRACK/SS (Pin 2): Tracking Input for the Controller or
Optional External Soft-Start Input. This pin allows the
start-up of VOUT to “track” the external voltage at this pin
using an external resistor divider. Tying this pin to VIN
allows VOUT to start up with the internal 0.74ms soft-start.
An external soft-start can be programmed by connecting
a capacitor between this pin and ground. Do not leave
this pin ﬂoating.
VFB (Pin 3): Feedback Pin. This pin receives the remotely
sensed feedback voltage for the controller from an external
resistor divider across the output.
ITH (Pin 4): Current Threshold and Error Ampliﬁer
Compensation Point. Nominal operating range on this pin
is from 0.7V to 2V. The voltage on this pin determines the
threshold of the main current comparator.
RUN (Pin 5): Run Control Input. Forcing this pin below
1.1V shuts down the chip. Driving this pin to VIN or
releasing this pin enables the chip to start-up with the
internal soft-start.
IPRG (Pin 6): Three-State Pin to Select Maximum Peak
Sense Voltage Threshold. This pin selects the maximum
allowed voltage drop between the VIN and SW pins
(i.e., the maximum allowed drop across the external
P-channel MOSFET). Tie to VIN, GND or ﬂoat to select
204mV, 85mV or 125mV respectively.
BG (Pin 7): Bottom (NMOS) Gate Drive Output. This pin
drives the gate of the external N-channel MOSFET. This
pin has an output swing from PGND to VIN.
TG (Pin 8): Top (PMOS) Gate Drive Output. This pin drives
the gate of the external P-channel MOSFET. This pin has
an output swing from PGND to VIN.
VIN (Pin 9): Chip Signal Power Supply. This pin powers
the entire chip, the gate drivers and serves as the positive
input to the differential current comparator.
SW (Pin 10): Switch Node Connection to Inductor. This
pin is also the negative input to the differential current
comparator and an input to the reverse current comparator.
Normally this pin is connected to the drain of the external
P-channel MOSFET, the drain of the external N-channel
MOSFET and the inductor.
GND (Exposed Pad, Pin 11): Ground connection for
internal circuits, the gate drivers and the negative input to
the reverse current comparator. The Exposed Pad must
be soldered to the PCB ground.
38091fc
7
LTC3809-1
FUNCTIONAL DIAGRAM
VIN
CIN
9 VIN
VREF
0.6V
VOLTAGE
REFERENCE
6 IPRG
SLOPE
CLK
+
UNDERVOLTAGE
LOCKOUT
S
R
ICMP
SENSE+
8
Q
–
OSC
VIN
ANTI-SHOOTTHROUGH
7
+
0.15V
SLEEP
–
GND
–
MODE
BURST DEFEAT
TRK/SS
BURSTDIS
FCB
0.68V
RB
0.3V
MUX
MN
+
BURSTDIS
1μA
BG
OV
IREV
1
VOUT
FCB
VIN
t = 0.74ms
INTERNAL
SOFT-START
2
L
SW
COUT
UVSD
RUN
TRACK/SS
10
PVIN
0.7μA
11
MP
GND
SWITCHING
LOGIC AND
BLANKING
CIRCUIT
VIN
5
TG
+
0.54V
–
VFB
UV
+
4
ITH
RC
+
EAMP +
–
–
VREF
0.6V
TRK/SS
3
CC
VFB
RA
38091 FD
IREV
+
SW
–
GND
RICMP
38091fc
8
LTC3809-1
OPERATION
(Refer to Functional Diagram)
Main Control Loop
The LTC3809-1 uses a constant frequency, current mode
architecture. During normal operation, the top external
P-channel power MOSFET is turned on when the clock sets
the RS latch, and is turned off when the current comparator
(ICMP) resets the latch. The peak inductor current at which
ICMP resets the RS latch is determined by the voltage on the
ITH pin, which is driven by the output of the error ampliﬁer
(EAMP). The VFB pin receives the output voltage feedback
signal from an external resistor divider. This feedback
signal is compared to the internal 0.6V reference voltage
by the EAMP. When the load current increases, it causes a
slight decrease in VFB relative to the 0.6V reference, which
in turn causes the ITH voltage to increase until the average
inductor current matches the new load current. While the top
P-channel MOSFET is off, the bottom N-channel MOSFET is
turned on until either the inductor current starts to reverse,
as indicated by the current reversal comparator IRCMP, or
the beginning of the next cycle.
Shutdown, Soft-Start and Tracking Start-Up
(RUN and TRACK/SS Pins)
The LTC3809-1 is shut down by pulling the RUN pin low.
In shutdown, all controller functions are disabled and the
chip draws only 9μA. The TG output is held high (off) and
the BG output low (off) in shutdown. Releasing the RUN
pin allows an internal 0.7μA current source to pull up the
RUN pin to VIN. The controller is enabled when the RUN
pin reaches 1.1V.
The start-up of VOUT is based on the three different connections on the TRACK/SS pin. The start-up of VOUT is
controlled by the LTC3809-1’s internal soft-start when
TRACK/SS is connected to VIN. During soft-start, the error
ampliﬁer EAMP compares the feedback signal VFB to the
internal soft-start ramp (instead of the 0.6V reference),
which rises linearly from 0V to 0.6V in about 1ms. This allows the output voltage to rise smoothly from 0V to its ﬁnal
value while maintaining control of the inductor current.
The 1ms soft-start time can be changed by connecting
the optional external soft-start capacitor CSS between the
TRACK/SS and GND pins. When the controller is enabled
by releasing the RUN pin, the TRACK/SS pin is charged up
by an internal 1μA current source and rises linearly from
0V to above 0.6V. The error ampliﬁer EAMP compares the
feedback signal VFB to this ramp instead, and regulates
VFB linearly from 0V to 0.6V.
When the voltage on the TRACK/SS pin is less than the
0.6V internal reference, the LTC3809-1 regulates the VFB
voltage to the TRACK/SS pin instead of the 0.6V reference.
Therefore VOUT of the LTC3809-1 can track an external
voltage VX during start-up. Typically, a resistor divider on
VX is connected to the TRACK/SS pin to allow the start-up
of VOUT to “track” that of VX . For coincident tracking during
start-up, the regulated ﬁnal value of VX should be larger
than that of VOUT, and the resistor divider on VX has the
same ratio as the divider on VOUT that is connected to VFB .
See detailed discussions in the Run and Soft-Start/Tracking
Functions in the Applications Information Section.
Light Load Operation (Burst Mode Operation,
Continuous Conduction or Pulse-Skipping Mode)
(MODE Pin)
The LTC3809-1 can be programmed for either high efﬁciency
Burst Mode operation, forced continuous conduction mode
or pulse-skipping mode at low load currents. To select
Burst Mode operation, tie the MODE pin to VIN. To select
forced continuous operation, tie the MODE pin to a DC
voltage below 0.4V (e.g., GND). Tying the MODE pin to a
DC voltage above 0.4V and below 1.2V (e.g., VFB) enables
pulse-skipping mode. The 0.4V threshold between forced
continuous operation and pulse-skipping mode can be
used in secondary winding regulation as described in the
Auxiliary Winding Control Using the MODE Pin discussion
in the Applications Information section.
When the LTC3809-1 is in Burst Mode operation, the peak
current in the inductor is set to approximately one-fourth
of the maximum sense voltage even though the voltage on
the ITH pin indicates a lower value. If the average inductor current is higher than the load current, the EAMP will
decrease the voltage on the ITH pin. When the ITH voltage
drops below 0.85V, the internal SLEEP signal goes high
and the external MOSFET is turned off.
38091fc
9
LTC3809-1
OPERATION
(Refer to Functional Diagram)
In sleep mode, much of the internal circuitry is turned off,
reducing the quiescent current that the LTC3809-1 draws.
The load current is supplied by the output capacitor. As
the output voltage decreases, the EAMP increases the
ITH voltage. When the ITH voltage reaches 0.925V, the
SLEEP signal goes low and the controller resumes normal
operation by turning on the external P-channel MOSFET
on the next cycle of the internal oscillator.
When the controller is enabled for Burst Mode or pulseskipping operation, the inductor current is not allowed to
reverse. Hence, the controller operates discontinuously.
The reverse current comparator RICMP senses the
drain-to-source voltage of the bottom external N-channel
MOSFET. This MOSFET is turned off just before the inductor
current reaches zero, preventing it from going negative.
In forced continuous operation, the inductor current is
allowed to reverse at light loads or under large transient
conditions. The peak inductor current is determined by the
voltage on the ITH pin. The P-channel MOSFET is turned
on every cycle (constant frequency) regardless of the ITH
pin voltage. In this mode, the efﬁciency at light loads is
lower than in Burst Mode operation. However, continuous
mode has the advantages of lower output ripple and no
noise at audio frequencies.
When the MODE pin is set to the VFB Pin, the LTC3809-1
operates in PWM pulse-skipping mode at light loads. In
this mode, the current comparator ICMP may remain
tripped for several cycles and force the external P-channel
MOSFET to stay off for the same number of cycles. The
inductor current is not allowed to reverse (discontinuous
operation). This mode, like forced continuous operation,
exhibits low output ripple as well as low audible noise
and reduced RF interference as compared to Burst Mode
operation. However, it provides low current efﬁciency
higher than forced continuous mode, but not nearly as
high as Burst Mode operation. During start-up or an
undervoltage condition (VFB ≤ 0.54V), the LTC3809-1
operates in pulse-skipping mode (no current reversal
allowed), regardless of the state of the MODE pin.
Short-Circuit and Current Limit Protection
The LTC3809-1 monitors the voltage drop ΔVSC (between
the GND and SW pins) across the external N-channel
MOSFET with the short-circuit current limit comparator.
The allowed voltage is determined by:
ΔVSC(MAX) = A • 90mV
where A is a constant determined by the state of the IPRG
pin. Floating the IPRG pin selects A = 1; tying IPRG to VIN
selects A = 5/3; tying IPRG to GND selects A = 2/3.
The inductor current limit for short-circuit protection is
determined by ΔVSC(MAX) and the on-resistance of the
external N-channel MOSFET:
ISC =
ΔVSC(MAX )
RDS(ON)
Once the inductor current exceeds ISC, the short current
comparator will shut off the external P-channel MOSFET
until the inductor current drops below ISC .
Output Overvoltage Protection
As further protection, the overvoltage comparator (OVP)
guards against transient overshoots, as well as other more
serious conditions that may overvoltage the output. When
the feedback voltage on the VFB pin has risen 13.33%
above the reference voltage of 0.6V, the external P-channel
MOSFET is turned off and the N-channel MOSFET is turned
on until the overvoltage is cleared.
38091fc
10
LTC3809-1
OPERATION
(Refer to Functional Diagram)
Dropout Operation
When the input supply voltage (VIN) approaches the output
voltage, the rate of change of the inductor current while the
external P-channel MOSFET is on (ON cycle) decreases.
This reduction means that the P-channel MOSFET will
remain on for more than one oscillator cycle if the inductor
current has not ramped up to the threshold set by the
EAMP on the ITH pin. Further reduction in the input supply
voltage will eventually cause the P-channel MOSFET to be
turned on 100%; i.e., DC. The output voltage will then be
determined by the input voltage minus the voltage drop
across the P-channel MOSFET and the inductor.
Undervoltage Lockout
To prevent operation of the P-channel MOSFET below
safe input voltage levels, an undervoltage lockout is
incorporated in the LTC3809-1. When the input supply
voltage (VIN) drops below 2.25V, the external P- and
N-channel MOSFETs and all internal circuits are turned
off except for the undervoltage block, which draws only
a few microamperes.
where A is a constant determined by the state of the IPRG
pin. Floating the IPRG pin selects A = 1; tying IPRG to
VIN selects A = 5/3; tying IPRG to GND selects A = 2/3.
The maximum value of VITH is typically about 1.98V, so
the maximum sense voltage allowed across the external
P-channel MOSFET is 125mV, 85mV or 204mV for the
three respective states of the IPRG pin.
However, once the controller’s duty cycle exceeds 20%,
slope compensation begins and effectively reduces the
peak sense voltage by a scale factor (SF) given by the
curve in Figure 1.
The peak inductor current is determined by the peak sense
voltage and the on-resistance of the external P-channel
MOSFET:
IPK =
ΔVSENSE(MAX )
RDS(ON)
110
100
90
Peak Current Sense Voltage Selection
and Slope Compensation (IPRG Pin)
When the LTC3809-1 controller is operating below 20%
duty cycle, the peak current sense voltage (between the
VIN and SW pins) allowed across the external P-channel
MOSFET is determined by:
V – 0.7 V
ΔVSENSE(MAX ) = A • ITH
10
SF = I/IMAX (%)
80
70
60
50
40
30
20
10
0
0
10 20 30 40 50 60 70 80 90 100
DUTY CYCLE (%)
38091 F01
Figure 1. Maximum Peak Current vs Duty Cycle
38091fc
11
LTC3809-1
APPLICATIONS INFORMATION
The typical LTC3809-1 application circuit is shown in Figure
8. External component selection for the controller is driven
by the load requirement and begins with the selection of
the inductor and the power MOSFETs.
Power MOSFET Selection
The LTC3809-1’s controller requires two external power
MOSFETs: a P-channel MOSFET for the topside (main)
switch and a N-channel MOSFET for the bottom (synchronous) switch. The main selection criteria for the power
MOSFETs are the breakdown voltage VBR(DSS), threshold
voltage VGS(TH), on-resistance RDS(ON), reverse transfer
capacitance CRSS, turn-off delay tD(OFF) and the total gate
charge QG.
The gate drive voltage is the input supply voltage. Since
the LTC3809-1 is designed for operation down to low input
voltages, a sublogic level MOSFET (RDS(ON) guaranteed at
VGS = 2.5V) is required for applications that work close to
this voltage. When these MOSFETs are used, make sure that
the input supply to the LTC3809-1 is less than the absolute
maximum MOSFET VGS rating, which is typically 8V.
The P-channel MOSFET’s on-resistance is chosen based
on the required load current. The maximum average load
current IOUT(MAX) is equal to the peak inductor current
minus half the peak-to-peak ripple current IRIPPLE. The
LTC3809-1’s current comparator monitors the drain-tosource voltage VDS of the top P-channel MOSFET, which
is sensed between the VIN and SW pins. The peak inductor
current is limited by the current threshold, set by the voltage
on the ITH pin, of the current comparator. The voltage on
the ITH pin is internally clamped, which limits the maximum
current sense threshold ΔVSENSE(MAX) to approximately
125mV when IPRG is ﬂoating (85mV when IPRG is tied
low; 204mV when IPRG is tied high).
The output current that the LTC3809-1 can provide is
given by:
IOUT(MAX ) =
ΔVSENSE(MAX ) IRIPPLE
–
RDS(ON)
2
where IRIPPLE is the inductor peak-to-peak ripple current
(see Inductor Value Calculation).
A reasonable starting point is setting ripple current IRIPPLE
to be 40% of IOUT(MAX). Rearranging the above equation
yields:
5 ΔVSENSE(MAX )
RDS(ON)MAX = •
for Duty Cycle < 20%
6
IOUT(MAX )
However, for operation above 20% duty cycle, slope
compensation has to be taken into consideration to select
the appropriate value of RDS(ON) to provide the required
amount of load current:
ΔVSENSE(MAX )
5
RDS(ON)MAX = • SF •
6
IOUT(MAX )
where SF is a scale factor whose value is obtained from
the curve in Figure 1.
These must be further derated to take into account the
signiﬁcant variation in on-resistance with temperature. The
following equation is a good guide for determining the required RDS(ON)MAX at 25°C (manufacturer’s speciﬁcation),
allowing some margin for variations in the LTC3809-1 and
external component values:
ΔVSENSE(MAX )
5
RDS(ON)MAX = • 0.9 • SF •
6
IOUT(MAX ) • ρT
The ρT is a normalizing term accounting for the temperature
variation in on-resistance, which is typically about 0.4%/°C,
as shown in Figure 2. Junction-to-case temperature TJC is
about 10°C in most applications. For a maximum ambient temperature of 70°C, using ρ80°C ~ 1.3 in the above
equation is a reasonable choice.
The N-channel MOSFET’s on resistance is chosen based
on the short-circuit current limit (ISC). The LTC38091’s short-circuit current limit comparator monitors the
drain-to-source voltage VDS of the bottom N-channel
MOSFET, which is sensed between the GND and SW pins.
38091fc
12
LTC3809-1
APPLICATIONS INFORMATION
VOUT
VIN
V −V
Bottom N-Channel Duty Cycle = IN OUT
VIN
2.0
Top P-Channel Duty Cycle =
1.5
1.0
The MOSFET power dissipations at maximum output
current are:
0.5
0
–50
PTOP =
50
100
0
JUNCTION TEMPERATURE (°C)
150
VOUT
2
2
• IOUT (MAX) • ρT • RDS(ON) + 2 • VIN
VIN
• IOUT (MAX) • CRSS • f
38091 F02
Figure 2. RDS(ON) vs Temperature
The short-circuit current sense threshold ΔVSC is set
approximately 90mV when IPRG is ﬂoating (60mV when
IPRG is tied low; 150mV when IPRG is tied high). The
on-resistance of N-channel MOSFET is determined by:
RDS(ON)MAX =
ΔVSC
ISC(PEAK )
The short-circuit current limit (ISC(PEAK)) should be larger
than the IOUT(MAX) with some margin to avoid interfering
with the peak current sensing loop. On the other hand,
in order to prevent the MOSFETs from excessive heating
and the inductor from saturation, ISC(PEAK) should be
smaller than the minimum value of their current ratings.
A reasonable range is:
IOUT(MAX) < ISC(PEAK) < IRATING(MIN)
Therefore, the on-resistance of N-channel MOSFET should
be chosen within the following range:
ΔVSC
IRATING(MIN)
< RDS(ON) <
ΔVSC
IOUT(MAX )
where ΔVSC is 90mV, 60mV or 150mV with IPRG being
ﬂoated, tied to GND or VIN respectively.
The power dissipated in the MOSFET strongly depends
on its respective duty cycles and load current. When the
LTC3809-1 is operating in continuous mode, the duty
cycles for the MOSFETs are:
PBOT =
VIN – VOUT
2
• IOUT (MAX) • ρT • RDS(ON)
VIN
Both MOSFETs have I2R losses and the PTOP equation
includes an additional term for transition losses, which are
largest at high input voltages. The bottom MOSFET losses
are greatest at high input voltage or during a short-circuit
when the bottom duty cycle is 100%.
The LTC3809-1 utilizes a non-overlapping, anti-shootthrough gate drive control scheme to ensure that the
P- and N-channel MOSFETs are not turned on at the same
time. To function properly, the control scheme requires
that the MOSFETs used are intended for DC/DC switching
applications. Many power MOSFETs, particularly P-channel
MOSFETs, are intended to be used as static switches and
therefore are slow to turn on or off.
Reasonable starting criteria for selecting the P-channel
MOSFET are that it must typically have a gate charge (QG)
less than 25nC to 30nC (at 4.5VGS) and a turn-off delay
(tD(OFF)) of less than approximately 140ns. However, due
to differences in test and speciﬁcation methods of various
MOSFET manufacturers, and in the variations in QG and
tD(OFF) with gate drive (VIN) voltage, the P-channel MOSFET
ultimately should be evaluated in the actual LTC3809-1
application circuit to ensure proper operation.
Shoot-through between the P-channel and N-channel
MOSFETs can most easily be spotted by monitoring the
input supply current. As the input supply voltage increases,
if the input supply current increases dramatically, then the
likely cause is shoot-through. Note that some MOSFETs
38091fc
13
LTC3809-1
APPLICATIONS INFORMATION
that do not work well at high input voltages (e.g., VIN >
5V) may work ﬁne at lower voltages (e.g., 3.3V).
Selecting the N-channel MOSFET is typically easier, since
for a given RDS(ON), the gate charge and turn-on and turn-off
delays are much smaller than for a P-channel MOSFET.
Inductor Value Calculation
Given the desired input and output voltages, the inductor
value and operating frequency, fOSC , directly determine
the inductor’s peak-to-peak ripple current:
IRIPPLE =
VOUT VIN – VOUT
•
VIN
fOSC • L
Lower ripple current reduces core losses in the inductor,
ESR losses in the output capacitors and output voltage
ripple. Thus, highest efﬁciency operation is obtained at
low frequency with a small ripple current. Achieving this,
however, requires a large inductor.
A reasonable starting point is to choose a ripple current
that is about 40% of IOUT(MAX). Note that the largest ripple
current occurs at the highest input voltage. To guarantee
that ripple current does not exceed a speciﬁed maximum,
the inductor should be chosen according to:
L≥
VIN – VOUT VOUT
•
fOSC • IRIPPLE VIN
Burst Mode Operation Considerations
The choice of RDS(ON) and inductor value also determines
the load current at which the LTC3809-1 enters Burst Mode
operation. When bursting, the controller clamps the peak
inductor current to approximately:
1 ΔVSENSE(MAX )
IBURST(PEAK ) = •
4
RDS(ON)
The corresponding average current depends on the
amount of ripple current. Lower inductor values (higher
IRIPPLE) will reduce the load current at which Burst Mode
operation begins.
The ripple current is normally set so that the inductor current
is continuous during the burst periods. Therefore,
IRIPPLE ≤ IBURST(PEAK)
This implies a minimum inductance of:
L MIN ≤
VIN – VOUT
V
• OUT
fOSC • IBURST(PEAK ) VIN
A smaller value than LMIN could be used in the circuit,
although the inductor current will not be continuous
during burst periods, which will result in slightly lower
efﬁciency. In general, though, it is a good idea to keep
IRIPPLE comparable to IBURST(PEAK).
Inductor Core Selection
Once the value of L is known, the type of inductor must be
selected. Actual core loss is independent of core size for a
ﬁxed inductor value, but is very dependent on the inductance selected. As inductance increases, core losses go
down. Unfortunately, increased inductance requires more
turns of wire and therefore copper losses will increase.
Ferrite designs have very low core losses and are preferred at high switching frequencies, so design goals can
concentrate on copper loss and preventing saturation.
Ferrite core material saturates “hard”, which means that
inductance collapses abruptly when the peak design current
is exceeded. Core saturation results in an abrupt increase
in inductor ripple current and consequent output voltage
ripple. Do not allow the core to saturate!
38091fc
14
LTC3809-1
APPLICATIONS INFORMATION
Different core materials and shapes will change the size/
current and price/current relationship of an inductor. Toroid
or shielded pot cores in ferrite or permalloy materials are
small and don’t radiate much energy, but generally cost
more than powdered iron core inductors with similar
characteristics. The choice of which style inductor to use
mainly depends on the price vs size requirements and any
radiated ﬁeld/EMI requirements. New designs for surface
mount inductors are available from Coiltronics, Coilcraft,
Toko and Sumida.
Schottky Diode Selection (Optional)
The schottky diode D in Figure 9 conducts current during the dead time between the conduction of the power
MOSFETs. This prevents the body diode of the bottom
N-channel MOSFET from turning on and storing charge
during the dead time, which could cost as much as 1%
in efﬁciency. A 1A Schottky diode is generally a good
size for most LTC3809-1 applications, since it conducts
a relatively small average current. Larger diode results
in additional transition losses due to its larger junction
capacitance. This diode may be omitted if the efﬁciency
loss can be tolerated.
CIN and COUT Selection
In continuous mode, the source current of the P-channel
MOSFET is a square wave of duty cycle (VOUT /VIN). To
prevent large voltage transients, a low ESR input capacitor
sized for the maximum RMS current must be used. The
maximum RMS capacitor current is given by:
VOUT • ( VIN – VOUT )
1/ 2
CIN Re quiredIRMS ≈ IMAX •
VIN
This formula has a maximum value at VIN = 2VOUT, where
IRMS = IOUT /2. This simple worst-case condition is commonly used for design because even signiﬁcant deviations
do not offer much relief. Note that capacitor manufacturer’s
ripple current ratings are often based on 2000 hours of life.
This makes it advisable to further derate the capacitor or
to choose a capacitor rated at a higher temperature than
required. Several capacitors may be paralleled to meet the
size or height requirements in the design. Due to the high
operating frequency of the LTC3809-1, ceramic capacitors
can also be used for CIN. Always consult the manufacturer
if there is any question.
The selection of COUT is driven by the effective series
resistance (ESR). Typically, once the ESR requirement
is satisﬁed, the capacitance is adequate for ﬁltering. The
output ripple (ΔVOUT) is approximated by:
⎛
⎞
1
ΔVOUT ≈ IRIPPLE • ⎜ ESR +
⎟
8 • f • COUT ⎠
⎝
where f is the operating frequency, COUT is the output
capacitance and IRIPPLE is the ripple current in the inductor. The output ripple is highest at maximum input voltage
since IRIPPLE increase with input voltage.
Setting Output Voltage
The LTC3809-1 output voltage is set by an external
feedback resistor divider carefully placed across the
output, as shown in Figure 3. The regulated output voltage
is determined by:
⎛ R ⎞
VOUT = 0.6 V • ⎜ 1 + B ⎟
⎝ RA ⎠
38091fc
15
LTC3809-1
APPLICATIONS INFORMATION
For most applications, a 59k resistor is suggested for RA.
In applications where minimizing the quiescent current is
critical, RA should be made bigger to limit the feedback
divider current. If RB then results in very high impedance,
it may be beneﬁcial to bypass RB with a 50pF to 100pF
capacitor CFF.
VOUT
LTC3809-1
RB
CFF
Once the controller is enabled, the start-up of VOUT is controlled by the state of the TRACK/SS pin. If the TRACK/SS
pin is connected to VIN, the start-up of VOUT is controlled
by internal soft-start, which slowly ramps the positive
reference to the error ampliﬁer from 0V to 0.6V, allowing
VOUT to rise smoothly from 0V to its ﬁnal value. The default
internal soft-start time is around 0.74ms. The soft-start
time can be changed by placing a capacitor between the
TRACK/SS pin and GND. In this case, the soft-start time
will be approximately:
VFB
tSS = CSS •
RA
38091 F03
600mV
1μA
where 1μA is an internal current source which is always on.
Figure 3. Setting Output Voltage
Run and Soft-Start/Tracking Functions
The LTC3809-1 has a low power shutdown mode which is
controlled by the RUN pin. Pulling the RUN pin below 1.1V
puts the LTC3809-1 into a low quiescent current shutdown
mode (IQ = 9μA). Releasing the RUN pin, an internal 0.7μA
(at VIN = 4.2V) current source will pull the RUN pin up
to VIN, which enables the controller. The RUN pin can be
driven directly from logic as showed in Figure 4.
When the voltage on the TRACK/SS pin is less than the
internal 0.6V reference, the LTC3809-1 regulates the VFB
voltage to the TRACK/SS pin voltage instead of 0.6V.
Therefore the start-up of VOUT can ratiometrically track
an external voltage VX, according to a ratio set by a resistor divider at TRACK/SS pin (Figure 5a). The ratiometric
relation between VOUT and VX is (Figure 5c):
VOUT R TA R A + RB
=
•
VX
R A R TA + R TB
VOUT
VX
3.3V OR 5V
LTC3809-1
LTC3809-1
RUN
LTC3809-1
RUN
RTB
TRACK/SS
38091 F04
RB
VFB
RA
RTA
38091 F5a
Figure 4. RUN Pin Interfacing
Figure 5a. Using the TRACK/SS Pin to Track VX
38091fc
16
LTC3809-1
APPLICATIONS INFORMATION
VOUT
VX
OUTPUT VOLTAGE
OUTPUT VOLTAGE
VX
VOUT
38091 F05b,c
TIME
TIME
(5b) Coincident Tracking
(5c) Ratiometric Tracking
Figure 5b and 5c. Two Different Modes of Output Voltage Tracking
For coincident tracking (VOUT = VX during start-up),
RTA = RA, RTB = RB
VX should always be greater than VOUT when using the
tracking function of TRACK/SS pin.
The internal current source (1μA), which is for external
soft-start, will cause a tracking error at VOUT. For example,
if a 59k resistor is chosen for RTA, the RTA current will be
about 10μA (600mV/59k). In this case, the 1μA internal
current source will cause about 10% (1μA/10μA • 100%)
tracking error, which is about 60mV (600mV • 10%)
referred to VFB. This is acceptable for most applications.
If a better tracking accuracy is required, the value of RTA
should be reduced.
Table 1 summarizes the different states in which the
TRACK/SS can be used.
Table 1. The States of the TRACK/SS Pin
TRACK/SS Pin
FREQUENCY
Capacitor CSS
External Soft-Start
VIN
Internal Soft-Start
Resistor Divider
VOUT Tracking an External Voltage VX
Auxiliary Winding Control Using the MODE Pin
The MODE pin can be used as an auxiliary feedback to
provide a means of regulating a ﬂyback winding output.
When this pin drops below its ground-referenced 0.4V
threshold, continuous mode operation is forced.
During continuous mode, current ﬂows continuously in
the transformer primary side. The auxiliary winding draws
current only when the bottom synchronous N-channel
MOSFET is on. When primary load currents are low and/
or the VIN /VOUT ratio is close to unity, the synchronous
MOSFET may not be on for a sufﬁcient amount of time to
transfer power from the output capacitor to the auxiliary
load. Forced continuous operation will support an auxiliary
winding as long as there is a sufﬁcient synchronous
MOSFET duty factor. The MODE input pin removes
the requirement that power must be drawn from the
transformer primary side in order to extract power from
the auxiliary winding. With the loop in continuous mode,
the auxiliary output may nominally be loaded without
regard to the primary output load.
38091fc
17
LTC3809-1
APPLICATIONS INFORMATION
The auxiliary output voltage VAUX is normally set, as shown
in Figure 6, by the turns ratio N of the transformer:
VAUX = (N + 1) • VOUT
LTC3809-1
R6
TG
MODE
L1
1:N
VAUX
+
1μF
VOUT
SW
R5
+
BG
COUT
38091 F06
Figure 6. Auxiliary Output Loop Connection
However, if the controller goes into pulse-skipping operation
and halts switching due to a light primary load current, then
VAUX will droop. An external resistor divider from VAUX to
the MODE sets a minimum voltage VAUX(MIN):
⎛ R6 ⎞
VAUX(MIN) = 0.4 V • ⎜ 1 + ⎟
⎝ R5 ⎠
If VAUX drops below this value, the MODE voltage forces
temporary continuous switching operation until VAUX is
again above its minimum.
Fault Condition: Short-Circuit and Current Limit
If the LTC3809-1’s load current exceeds the short-circuit
current limit (ISC), which is set by the short-circuit sense
threshold (ΔVSC) and the on resistance (RDS(ON)) of
bottom N-channel MOSFET, the top P-channel MOSFET
is turned off and will not be turned on at the next clock
cycle unless the load current decreases below ISC. In this
case, the controller’s switching frequency is decreased
and the output is regulated by short-circuit (current limit)
protection.
105
NORMALIZED VOLTAGE OR CURRENT (%)
VIN
In a hard short (VOUT = 0V), the top P-channel MOSFET
is turned off and kept off until the short-circuit condition
is cleared. In this case, there is no current path from
input supply (VIN) to either VOUT or GND, which prevents
excessive MOSFET and inductor heating.
100
VREF
95
MAXIMUM
SENSE VOLTAGE
90
85
80
75
2.0 2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8 2.9 3.0
INPUT VOLTAGE (V)
38091 F07
Figure 7. Line Regulation of VREF and Maximum Sense Voltage
Low Supply Voltage
Although the LTC3809-1 can function down to below 2.4V,
the maximum allowable output current is reduced as VIN
decreases below 3V. Figure 7 shows the amount of change
as the supply is reduced down to 2.4V. Also shown is the
effect on VREF.
Minimum On-Time Considerations
Minimum on-time, tON(MIN) is the smallest amount of time
that the LTC3809-1 is capable of turning the top P-channel
MOSFET on. It is determined by internal timing delays and
the gate charge required to turn on the top MOSFET. Low
duty cycle and high frequency applications may approach
the minimum on-time limit and care should be taken to
ensure that:
tON(MIN) <
VOUT
fOSC • VIN
38091fc
18
LTC3809-1
APPLICATIONS INFORMATION
If the duty cycle falls below what can be accommodated
by the minimum on-time, the LTC3809-1 will begin to skip
cycles (unless forced continuous mode is selected). The
output voltage will continue to be regulated, but the ripple
current and ripple voltage will increase. The minimum ontime for the LTC3809-1 is typically about 210ns. However,
as the peak sense voltage (IL(PEAK) • RDS(ON)) decreases,
the minimum on-time gradually increases up to about
260ns. This is of particular concern in forced continuous
applications with low ripple current at light loads. If forced
continuous mode is selected and the duty cycle falls below
the minimum on time requirement, the output will be
regulated by overvoltage protection.
Efﬁciency Considerations
The efﬁciency of a switching regulator is equal to the output
power divided by the input power times 100%. It is often
useful to analyze individual losses to determine what is
limiting efﬁciency and which change would produce the
most improvement. Efﬁciency can be expressed as:
Efﬁciency = 100% – (L1 + L2 + L3 + …)
where L1, L2, etc. are the individual losses as a percentage
of input power.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of
the losses in LTC3809-1 circuits: 1) LTC3809-1 DC bias
current, 2) MOSFET gate-charge current, 3) I2R losses
and 4) transition losses.
1) The VIN (pin) current is the DC supply current, given
in the Electrical Characteristics, which excludes MOSFET
driver currents. VIN current results in a small loss that
increases with VIN.
2) MOSFET gate-charge current results from switching
the gate capacitance of the power MOSFET. Each time a
MOSFET gate is switched from low to high to low again,
a packet of charge dQ moves from VIN to ground. The
resulting dQ/dt is a current out of VIN, which is typically
much larger than the DC supply current. In continuous
mode, IGATECHG = f • QP.
3) I2R losses are calculated from the DC resistances of the
MOSFETs, inductor and/or sense resistor. In continuous
mode, the average output current ﬂows through L but
is “chopped” between the top P-channel MOSFET and
the bottom N-channel MOSFET. The MOSFET RDS(ON)
multiplied by duty cycle can be summed with the resistance
of L to obtain I2R losses.
4) Transition losses apply to the external MOSFET and
increase with higher operating frequencies and input
voltages. Transition losses can be estimated from:
Transition Loss = 2 • VIN2 • IO(MAX) • CRSS • f
Other losses, including CIN and COUT ESR dissipative losses
and inductor core losses, generally account for less than
2% total additional loss.
Checking Transient Response
The regulator loop response can be checked by looking
at the load transient response. Switching regulators take
several cycles to respond to a step in load current. When
a load step occurs, VOUT immediately shifts by an amount
equal to (ΔILOAD) • (ESR), where ESR is the effective series resistance of COUT. ΔILOAD also begins to charge or
discharge COUT generating a feedback error signal used
by the regulator to return VOUT to its steady-state value.
During this recovery time, VOUT can be monitored for
overshoot or ringing that would indicate a stability problem.
OPTI-LOOP compensation allows the transient response
to be optimized over a wide range of output capacitance
and ESR values.
The ITH series RC-CC ﬁlter (see Functional Diagram) sets
the dominant pole-zero loop compensation.
The ITH external components showed in the ﬁgure on the
ﬁrst page of this data sheet will provide adequate compensation for most applications. The values can be modiﬁed
slightly (from 0.2 to 5 times their suggested values) to
optimize transient response once the ﬁnal PC layout is done
and the particular output capacitor type and value have
been determined. The output capacitor needs to be decided
upon because the various types and values determine the
loop feedback factor gain and phase. An output current
38091fc
19
LTC3809-1
APPLICATIONS INFORMATION
pulse of 20% to 100% of full load current having a rise
time of 1μs to 10μs will produce output voltage and ITH
pin waveforms that will give a sense of the overall loop
stability. The gain of the loop will be increased by increasing RC and the bandwidth of the loop will be increased
by decreasing CC. The output voltage settling behavior is
related to the stability of the closed-loop system and will
demonstrate the actual overall supply performance. For
a detailed explanation of optimizing the compensation
components, including a review of control loop theory,
refer to Application Note 76.
A second, more severe transient is caused by switching
in loads with large (>1μF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel
with COUT, causing a rapid drop in VOUT. No regulator can
deliver enough current to prevent this problem if the load
switch resistance is low and it is driven quickly. The only
solution is to limit the rise time of the switch drive so that
the load rise time is limited to approximately (25) • (CLOAD).
Thus a 10μF capacitor would be require a 250μs rise time,
limiting the charging current to about 200mA.
Design Example
As a design example, assume VIN will be operating from a
maximum of 4.2V down to a minimum of 2.75V (powered
by a single lithium-ion battery). Load current requirement
is a maximum of 2A, but most of the time it will be in a
standby mode requiring only 2mA. Efﬁciency at both low
and high load currents is important. Burst Mode operation
at light loads is desired. Output voltage is 1.8V. The IPRG
pin will be left ﬂoating, so the maximum current sense
threshold ΔVSENSE(MAX) is approximately 125mV.
Maximum Duty Cycle =
VOUT
= 65.5%
VIN(MIN)
From Figure 1, SF = 82%.
ΔVSENSE(MAX )
5
RDS(ON)MAX = • 0.9 • SF •
= 0.032Ω
6
IOUT(MAX ) • ρT
A 0.032Ω P-channel MOSFET in Si7540DP is close to
this value.
The N-channel MOSFET in Si7540DP has 0.017Ω RDS(ON).
The short-circuit current is:
ISC =
90mV
= 5.3A
0.017Ω
So the inductor current rating should be higher than 5.3A.
The LTC3809-1 operates at a frequency of 550kHz. For
continuous Burst Mode operation with 600mA IRIPPLE,
the required minimum inductor value is:
LMIN =
⎛
1.8 V
1.8 V ⎞
• ⎜ 1−
⎟ = 1.88μH
550kHz • 600mA ⎝ 2.75V ⎠
A 6A 2.2μH inductor works well for this application.
CIN will require an RMS current rating of at least 1A
at temperature. A COUT with 0.1Ω ESR will cause
approximately 60mV output ripple.
PC Board Layout Checklist
When laying out the printed circuit board, use the following
checklist to ensure proper operation of the LTC3809-1.
• The power loop (input capacitor, MOSFET, inductor,
output capacitor) should be as small as possible and
isolated as much as possible from LTC3809-1.
• Put the feedback resistors close to the VFB pins. The ITH
compensation components should also be very close
to the LTC3809-1.
• The current sense traces should be Kelvin connections
right at the P-channel MOSFET source and drain.
• Keeping the switch node (SW) and the gate driver nodes
(TG, BG) away from the small-signal components,
especially the feedback resistors, and ITH compensation
components.
38091fc
20
LTC3809-1
TYPICAL APPLICATIONS
VIN
2.75V TO 8V
1
10μF
MODE
VIN
6
CITH
220pF RITH
15k
4
2
187k
IPRG
TG
LTC3809EDD-1
ITH
SW
TRACK/SS
BG
9
8
MP
Si7540DP
L
1.5μH
10
7
VOUT
2.5V
(5A AT 5VIN)
MN
Si7540DP
3
VFB
GND
RUN
5
COUT
150μF
+
11
59k
100pF
38091 F08
L: VISHAY IHLP-2525CZ-01
COUT: SANYO 4TPB150MC
Figure 8. 550kHz, Synchronous DC/DC Converter with Internal Soft-Start
VIN
2.75V TO 8V
10μF
1
MODE
VIN
6
470pF
15k
4
IPRG
TG
LTC3809EDD-1
ITH
SW
TRACK/SS
BG
9
8
MP
Si3447BDV
L
1.5μH
10
VOUT
1.8V
2A
10nF
2
118k
3
59k
VFB
GND
11
RUN
7
MN
Si3460DV
5
D
(OPT)
COUT
22μF
x2
100pF
L: VISHAY IHLP-2525CZ-01
D: ON SEMI MBRM120LT3 (OPTIONAL)
38091 F09
Figure 9. 550kHz, Synchronous DC/DC Converter with External Soft-Start, Ceramic Output Capacitor
38091fc
21
LTC3809-1
TYPICAL APPLICATIONS
Synchronous DC/DC Converter with Output Tracking
1
VIN
2.75V TO 8V
10μF
MODE
VIN
6
220pF
15k
4
1.18k
2
Vx
IPRG
TG
LTC3809EDD-1
ITH
SW
TRACK/SS
BG
9
8
MP
Si7540DP
L
1.5μH
10
7
MN
Si7540DP
590Ω
118k
3
VFB
GND
RUN
5
COUT
150μF
VOUT
1.8V
(5A AT 5VIN)
+
11
59k
100pF
38091 TA03
L: VISHAY IHLP-2525CZ-01
COUT: SANYO 4TPB150MC
VOUT < Vx
PACKAGE DESCRIPTION
DD Package
10-Lead Plastic DFN (3mm × 3mm)
(Reference LTC DWG # 05-08-1698)
R = 0.115
TYP
6
0.38 p 0.10
10
0.675 p 0.05
3.50 p 0.05
1.65 p 0.05
2.15 p 0.05 (2 SIDES)
3.00 p 0.10
(4 SIDES)
PACKAGE
OUTLINE
1.65 p 0.10
(2 SIDES)
PIN 1
TOP MARK
(SEE NOTE 6)
(DD10) DFN 1103
5
0.200 REF
0.25 p 0.05
0.50
BSC
2.38 p 0.05
(2 SIDES)
1
0.25 p 0.05
0.50 BSC
0.75 p 0.05
0.00 – 0.05
2.38 p 0.10
(2 SIDES)
BOTTOM VIEW—EXPOSED PAD
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
NOTE:
1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION
OF (WEED-2). CHECK THE LTC WEBSITE DATA SHEET FOR CURRENT STATUS
OF VARIATION ASSIGNMENT
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON
ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE
TOP AND BOTTOM OF PACKAGE
38091fc
22
LTC3809-1
PACKAGE DESCRIPTION
MSE Package
10-Lead Plastic MSOP, Exposed Die Pad
(Reference LTC DWG # 05-08-1664 Rev C)
BOTTOM VIEW OF
EXPOSED PAD OPTION
2.794 p 0.102
(.110 p .004)
5.23
(.206)
MIN
0.889 p 0.127
(.035 p .005)
1
0.05 REF
10
DETAIL “B”
CORNER TAIL IS PART OF
DETAIL “B” THE LEADFRAME FEATURE.
FOR REFERENCE ONLY
NO MEASUREMENT PURPOSE
3.00 p 0.102
(.118 p .004)
(NOTE 3)
10 9 8 7 6
DETAIL “A”
0o – 6o TYP
1 2 3 4 5
GAUGE PLANE
0.53 p 0.152
(.021 p .006)
DETAIL “A”
0.18
(.007)
0.497 p 0.076
(.0196 p .003)
REF
3.00 p 0.102
(.118 p .004)
(NOTE 4)
4.90 p 0.152
(.193 p .006)
0.254
(.010)
0.29
REF
1.83 p 0.102
(.072 p .004)
2.083 p 0.102 3.20 – 3.45
(.082 p .004) (.126 – .136)
0.50
0.305 p 0.038
(.0197)
(.0120 p .0015)
BSC
TYP
RECOMMENDED SOLDER PAD LAYOUT
2.06 p 0.102
(.081 p .004)
SEATING
PLANE
0.86
(.034)
REF
1.10
(.043)
MAX
0.17 – 0.27
(.007 – .011)
TYP
0.50
(.0197)
BSC
0.1016 p 0.0508
(.004 p .002)
MSOP (MSE) 0908 REV C
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
38091fc
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
23
LTC3809-1
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC1628/LTC3728
Dual High Efﬁciency, 2-Phase Synchronous Step Down Controllers
Constant Frequency, Standby, 5V and 3.3V LDOs, VIN to 36V,
LTC1735
High Efﬁciency Synchronous Step-Down Controller
Burst Mode Operation, 16-Pin Narrow SSOP, Fault Protection,
3.5V ≤ VIN ≤ 36V
LTC1773
Synchronous Step-Down Controller
2.65V ≤ VIN ≤ 8.5V, IOUT Up to 4A, 10-Lead MSOP
LTC1778
No RSENSE , Synchronous Step-Down Controller
Current Mode Operation Without Sense Resistor,
Fast Transient Response, 4V ≤ VIN ≤ 36V
LTC1872
Constant Frequency Current Mode Step-Up Controller
2.5V ≤ VIN ≤ 9.8V, SOT-23 Package, 550kHz
LTC3411
1.25A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter
95% Efﬁciency, VIN: 2.5V to 5.5V, VOUT = 0.8V, IQ = 60μA,
ISD = <1μA, MS Package
LTC3412
2.5A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter
95% Efﬁciency, VIN: 2.5V to 5.5V, VOUT = 0.8V, IQ = 60μA,
ISD = <1μA, TSSOP-16E Package
LTC3416
4A, 4MHz, Monolithic Synchronous Step-Down Regulator
Tracking Input to Provide Easy Supply Sequencing,
2.25V ≤ VIN ≤ 5.5V, 20-Lead TSSOP Package
LTC3418
8A, 4MHz, Monolithic Synchronous Regulator
Tracking Input to Provide Easy Supply Sequencing,
2.25V ≤ VIN ≤ 5.5V, QFN Package
LTC3701
2-Phase, Low Input Voltage Dual Step-Down DC/DC Controller
2.5V ≤ VIN ≤ 9.8V, 550kHz, PGOOD, PLL, 16-Lead SSOP
LTC3708
2-Phase, No RSENSE , Dual Synchronous Controller with
Output Tracking
Constant On-Time Dual Controller, VIN Up to 36V, Very Low
Duty Cycle Operation, 5mm × 5mm QFN Package
LTC3736/LTC3736-2 2-Phase, No RSENSE , Dual Synchronous Controller with
Output Tracking
2.75V ≤ VIN ≤ 9.8V, 0.6V ≤ VOUT ≤ VIN , 4mm × 4mm QFN
LTC3736-1
Low EMI, 2-Phase, No RSENSE , Dual Synchronous Controller with
Output Tracking
Integrated Spread Spectrum for 20dB Lower “Noise,”
2.75V ≤ VIN ≤ 9.8V
LTC3737
2-Phase, No RSENSE , Dual DC/DC Controller with Output Tracking
2.75V ≤ VIN ≤ 9.8V, 0.6V ≤ VOUT ≤ VIN , 4mm × 4mm QFN
LTC3772
Micropower, No RSENSE , Constant Frequency Step-Down Controller 40μA No-Load IQ, Non-Synchronous, 2.75V ≤ VIN ≤ 9.8V,
550kHz, 3mm × 2mm DFN or 8-Lead TSOT-23 Packages.
LTC3776
Dual, 2-Phase, No RSENSE , Synchronous Controller for DDR/QDR
Memory Termination
LTC3808
No RSENSE , Low EMI, Synchronous Controller with Output Tracking 2.75V ≤ VIN ≤ 9.8V, 4mm × 3mm DFN, Spread Spectrum for
20dB Lower Peak Noise
LTC3809
No RSENSE , Low EMI, Synchronous DC/DC Controller
Provides VDDQ and VTT with One IC, 2.75V ≤ VIN ≤ 9.8V,
Adjustable Constant Frequency with PLL Up to 850kHz,
Spread Spectrum Operation, 4mm × 4mm QFN and 24-Lead
SSOP Packages
2.75V ≤ VIN ≤ 9.8V, 3mm × 3mm DFN and 10-Lead MSOPE
Packages, Spread Spectrum for 20dB Lower Peak Noise
PolyPhase is a trademark of Linear Technology Corporation.
38091fc
24 Linear Technology Corporation
LT 1108 REV C • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2005

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