LINER LTC3563 500ma, synchronous step-down dc/dc converter with selectable output voltage Datasheet

LTC3563
500mA, Synchronous
Step-Down DC/DC Converter
with Selectable Output Voltage
DESCRIPTIO
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FEATURES
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High Efficiency: Up to 96%
Pin Selectable Output Voltage: 1.28V/1.87V
Low Ripple (<20mVP-P) Burst Mode® Operation:
IQ = 26µA
Very Low Quiescent Current: Only 26µA
2.5V to 5.5V Input Voltage Range
2.25MHz Constant Frequency Operation
Low Dropout Operation: 100% Duty Cycle
No Schottky Diode Required
Internal Soft-Start Limits Inrush Current
Shutdown Mode Draws <1µA Supply Current
±2% Output Voltage Accuracy
Current Mode Operation for Excellent Line and Load
Transient Response
Overtemperature Protected
Available in 2mm × 2mm 6-Lead DFN
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APPLICATIO S
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The switching frequency is internally set at 2.25MHz,
allowing the use of small surface mount inductors and
capacitors. The LTC3563 is specifically designed to work
well with ceramic output capacitors, achieving very low
output voltage ripple and a small PCB footprint.
The LTC3563 is configured for the power saving Burst
Mode Operation.
Cellular Telephones
Wireless and DSL Modems
Digital Cameras
MP3 Players
PDAs and Other Handheld Devices
, LTC, LT, LTM and Burst Mode are registered trademarks of Linear Technology
Corporation. All other trademarks are the property of their respective owners.
Protected by U.S. Patents, including 5481178, 6580258, 6304066, 6127815, 6498466,
6611131, 5994885.
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The LTC®3563 is a high efficiency monolithic synchronous buck converter using a constant frequency, current
mode architecture. A voltage select input allows the user
to program the output voltage to 1.28V or 1.87V. Supply
current during operation is only 26µA, dropping to <1µA in
shutdown. The 2.5V to 5.5V input voltage range makes the
LTC3563 ideally suited for single Li-Ion battery-powered
applications. 100% duty cycle provides low dropout operation, extending battery life in portable systems. Internal
power switches are optimized to provide high efficiency
and eliminate the need for an external Schottky diode.
TYPICAL APPLICATIO
Efficiency and Power Loss vs Output Current
100
1000
90
1.28V 1.87V
VIN
SW
VOUT
1.28V/1.87V
500mA
LTC3563
RUN
VOUT
VSEL
COUT
10µF
CER
3563 TA01a
GND
80
100
70
EFFICIENCY (%)
2.2µH
CIN
10µF
CER
60
50
10
40
30
1
20
10
0
0.1
POWER LOSS (mW)
VIN
2.7V TO 5.5V
VIN = 3.6V
VOUT = 1.87V
1
10
100
OUTPUT CURRENT (mA)
0.1
1000
3563 TA01b
3563f
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LTC3563
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AXI U
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ABSOLUTE
RATI GS
PACKAGE/ORDER INFORMATION
(Note 1)
Input Supply Voltage (VIN) ........................... –0.3V to 6V
VOUT, RUN Voltages .....................................–0.3V to VIN
VSEL Voltage...................................–0.3V to (VIN + 0.3V)
SW Voltage ....................................–0.3V to (VIN + 0.3V)
Operating Ambient Temperature Range
(Note 2).................................................... –40°C to 85°C
Junction Temperature (Note 7) ............................. 125°C
Storage Temperature Range................... –65°C to 125°C
Reflow Peak Body Temperature (DFN) .................. 260°C
TOP VIEW
6 RUN
VOUT 1
VIN 2
7
5 VSEL
4 SW
GND 3
DC PACKAGE
6-LEAD (2mm × 2mm) PLASTIC DFN
TJMAX = 125°C, θJA = 102°C/W, θJC = 20°C/W (SOLDERED TO A 4-LAYER BOARD, NOTE 3)
EXPOSED PAD (PIN 7) IS PGND, MUST BE SOLDERED TO PCB
ORDER PART NUMBER
DC PART MARKING
LTC3563EDC
LCSZ
Order Options Tape and Reel: Add #TR
Lead Free: Add #PBF Lead Free Tape and Reel: Add #TRPBF
Lead Free Part Marking: http://www.linear.com/leadfree/
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 3.6V unless otherwise noted.
SYMBOL
PARAMETER
VIN
Operating Voltage Range
VOUT
Output Voltage (Note 4)
CONDITIONS
VSEL = 0V
VSEL = High
ΔVLINE_REG
Reference Voltage Line Regulation (Note 4) VIN = 2.5V to 5.5V
ΔVLOAD_REG
Output Voltage Load Regulation (Note 4)
ILOAD = 100mA to 500mA
IS
Input DC Supply Current (Note 5)
Active Mode
Sleep Mode
Shutdown
VOUT = 1.1V, VSEL = 0V
VOUT = 1.4V, VSEL = 0V
RUN = 0V
MIN
TYP
MAX
5.5
V
1.28
1.87
1.306
1.908
V
V
0.04
0.2
%/V
0.2
%
26
0.1
500
35
1
µA
µA
µA
1.8
2.25
2.7
MHz
650
1000
1750
mA
●
2.5
●
●
1.254
1.832
●
UNITS
fOSC
Oscillator Frequency
ILIM
Peak Switch Current
VIN = 3V, VFB = 0.5V, Duty Cycle < 35%
RDS(ON)
P-Channel On Resistance (Note 6)
N-Channel On Resistance (Note 6)
ISW = 100mA
ISW = 100mA
0.5
0.35
0.65
0.55
Ω
Ω
ISW(LKG)
Switch Leakage Current
VIN = 5V, VRUN = 0V, VSW = 0V or 5V
±0.01
±1
µA
3563f
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LTC3563
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 3.6V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
VUVLO
Undervoltage Lockout Threshold
VIN Rising
VIN Falling
VRUN
RUN Threshold
●
IRUN
RUN Leakage Current
●
VSEL
VSEL Threshold
●
RSEL
VSEL Pull-Up Resistance
Resistance Between VSEL and VIN
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime. No pin should exceed 6V.
Note 2: The LTC3563 is guaranteed to meet performance specifications
from 0°C to 85°C. Specifications over the –40°C to 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls.
Note 3: Failure to solder the Exposed Pad of the package to the PC board
will result in a thermal resistance much higher than 40°C/W.
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VOUT
50mV/DIV
AC COUPLED
IL
100mA/DIV
VIN = 3.6V
VOUT = 1.87V
ILOAD = 30mA
2µs/DIV
3563 G01
2.3
1.8
0.3
UNITS
V
V
1.2
V
±0.01
±1
µA
0.3
1
1.2
V
1.3
2.2
3
MΩ
Start-Up from Shutdown
RUN
2V/DIV
RUN
2V/DIV
VOUT
1V/DIV
VOUT
1V/DIV
IL
200mA/DIV
IL
500mA/DIV
VIN = 3.6V
VOUT = 1.87V
ILOAD = 0A
MAX
2.0
1.9
TA = 25°C unless otherwise specified.
Start-Up from Shutdown
SW
2V/DIV
TYP
Note 4: The converter is tested in a proprietary test mode that connects
the output of the error amplifier to the SW pin, which is connected to an
external servo loop.
Note 5: Dynamic supply current is higher due to the internal gate charge
being delivered at the switching frequency.
Note 6: The DFN switch on resistance is guaranteed by correlation to wafer
level measurements.
Note 7: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formula:
TJ = TA + (PD) • (θJA).
TYPICAL PERFOR A CE CHARACTERISTICS
Burst Mode Operation
MIN
400µs/DIV
3542 G02
VIN = 3.6V
VOUT = 1.87V
ILOAD = 500mA
400µs/DIV
3563 G03
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LTC3563
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TYPICAL PERFOR A CE CHARACTERISTICS
TA = 25°C unless otherwise specified.
Load Step
Load Step
Load Step
VOUT
100mV/DIV
AC COUPLED
VOUT
100mV/DIV
AC COUPLED
VOUT
100mV/DIV
AC COUPLED
IL
500mA/DIV
IL
500mA/DIV
IL
500mA/DIV
ILOAD
500mA/DIV
ILOAD
500mA/DIV
ILOAD
500mA/DIV
3563 G04
VIN = 3.6V
20µs/DIV
VOUT = 1.87V
ILOAD = 0A TO 500mA
3563 G05
VIN = 3.6V
20µs/DIV
VOUT = 1.87V
ILOAD = 30mA TO 500mA
Regulated Output Voltage
vs Temperature
Load Step
1.31
VOUT
100mV/DIV
AC COUPLED
Oscillator Frequency
vs Temperature
2.5
VSEL = 0V
1.30
2.4
VOUT (V)
ILOAD
500mA/DIV
FREQUENCY (MHz)
1.29
IL
500mA/DIV
1.28
1.27
1.26
2.3
2.2
2.1
3563 G07
VIN = 3.6V
20µs/DIV
VOUT = 1.28V
ILOAD = 30mA TO 500mA
3563 G06
20µs/DIV
VIN = 3.6V
VOUT = 1.28V
ILOAD = 0mA TO 500mA
1.25
1.24
–50 –25
50
25
75
0
TEMPERATURE (°C)
100
2.0
–50
125
–25
50
25
0
75
TEMPERATURE (°C)
2.6
0.4
2.5
0.3
2.3
2.2
2.1
–0.1
–0.2
–0.4
5
4
SUPPLY VOLTAGE (V)
6
3563 G10
1.0
0
1.9
VIN = 3.6V
1.5
0.1
–0.3
3
IOUT = 200mA
0.2
2.0
2
2.0
VOUT ERROR (%)
2.4
VOUT ERROR (%)
FREQUENCY (MHz)
0.5
1.8
Output Voltage vs Load Current
Output Voltage vs Supply Voltage
2.7
125
3563 G09
3563 G08
Oscillator Frequency
vs Supply Voltage
100
0.5
Burst Mode
OPERATION
0
–0.5
–1.0
–1.5
VOUT = 1.28V
VOUT = 1.87V
–0.5
2
2.5
3
3.5 4
4.5 5
INPUT VOLTAGE (V)
5.5
6
3563 G11
VOUT = 1.28V
VOUT = 1.87V
–2.0
1
10
100
LOAD CURRENT (mA)
1000
3563 G12
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LTC3563
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TYPICAL PERFOR A CE CHARACTERISTICS
TA = 25°C unless otherwise specified.
RDS(ON) vs Temperature
RDS(ON) vs Input Voltage
0.9
0.9
0.8
0.8
0.7
0.7
Switch Leakage vs Input Voltage
1000
900
MAIN SWITCH
LEAKAGE CURRENT (pA)
0.6
RDS(ON) (Ω)
MAIN SWITCH
0.5
0.4
0.3
0.2
0.5
0.4
0.3
SYNCHRONOUS
SWITCH
SYNCHRONOUS
SWITCH
0.2
0.1
0
3
4
VIN (V)
5
6
7
–25
0
25
50
75
TEMPERATURE (°C)
100
3563 G13
600
500
MAIN SWITCH
400
300
SYNCHRONOUS
SWITCH
100
0
125
0
1
2
3
VIN (V)
3563 G14
Switch Leakage vs Temperature
4
5
6
3542 G15
Efficiency vs Input Voltage
100
300
VOUT = 1.87V
90
250
80
EFFICIENCY (%)
200
150
100
70
60
50
MAIN SWITCH
SYNCHRONOUS SWITCH
50
0
–50 –25
40
50
25
75
0
TEMPERATURE (°C)
100
30
2.5
125
IOUT = 500mA
IOUT = 100mA
IOUT = 10mA
IOUT = 1mA
IOUT = 0.1mA
3
4.5
4
3.5
INPUT VOLTAGE (V)
5
Efficiency vs Load Current
Efficiency vs Load Current
100
100
90
90
80
80
70
70
60
50
40
30
60
50
40
30
20
10
5.5
3563 G17
3542 G16
EFFICIENCY (%)
2
SWITCH LEAKAGE (nA)
1
0
–50
700
200
VIN = 2.7V
VIN = 3.6V
VIN = 4.2V
0.1
EFFICIENCY (%)
RDS(ON) (Ω)
0.6
800
VOUT = 1.87V
FIGURE 3a CIRCUIT
0
0.1
VIN = 2.7V
VIN = 3.6V
VIN = 4.2V
1
10
100
OUTPUT CURRENT (mA)
1000
3563 G17
20
10
VOUT = 1.28V
FIGURE 3a CIRCUIT
0
0.1
VIN = 2.7V
VIN = 3.6V
VIN = 4.2V
1
10
100
OUTPUT CURRENT (mA)
1000
3563 G18
3563f
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LTC3563
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PI FU CTIO S
VOUT (Pin 1): Output Voltage Feedback. An internal resistive
divider divides the output voltage down for comparison
to the internal 0.6V reference voltage.
VIN (Pin 2): Power Supply Pin. Must be closely decoupled
to GND.
GND (Pin 3): Ground Pin.
SW (Pin 4): Switch Node Connection to Inductor. This pin
connects to the drains of the internal main and synchronous
power MOSFET switches.
VSEL (Pin 5): Output Voltage Selection Pin. This pin controls
the regulated output voltage. When tied to GND, VOUT is
1.28V. When floating or connecting this pin to VIN, VOUT
becomes 1.87V.
RUN (Pin 6): Converter Enable Pin. Forcing this pin above
1.5V enables this part, while forcing it below 0.3V causes
the device to shut down. In shutdown, all functions are
disabled drawing <1µA supply current. This pin must be
driven; do not float.
Exposed Pad (Pin 7): GND. The Exposed Pad is ground. It
must be soldered to PCB ground to provide both electrical
contact and optimum thermal performance.
3563f
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LTC3563
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BLOCK DIAGRA
1
VOUT
SLOPE COMPENSATION
+
OSC
VIN
2
ICOMP
–
VIN
0.6V
5
+
–
–
EA
+
VSEL
VB
+
BURST
LOGIC
SW
ANTISHOOT
THROUGH
4
VIN
+
6
RUN
0.6V REF
SHUTDOWN
IRCMP
–
GND
3
3563 BD
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LTC3563
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OPERATIO
The LTC3563 uses a constant frequency, current mode,
step-down architecture. The operating frequency is set
at 2.25MHz.
The output voltage is set by an internal divider. An error
amplifier compares the divided output voltage with a
reference voltage of 0.6V and adjusts the peak inductor
current accordingly.
Main Control Loop
During normal operation, the top power switch (P-channel
MOSFET) is turned on at the beginning of a clock cycle
when the VOUT voltage is below the regulated voltage.
The current flows into the inductor and the load increases
until the current limit is reached. The switch turns off and
energy stored in the inductor flows through the bottom
switch (N-channel MOSFET) into the load until the next
clock cycle. The peak inductor current is controlled by
the internally compensated output of the error amplifier.
When the load current increases, the feedback voltage
decreases slightly below the reference. This decrease
causes the error amplifier to increase its output voltage
until the average inductor current matches the new load
current. The main control loop is shut down by pulling
the RUN pin to ground.
Burst Mode Operation
During light load currents, the LTC3563 operates in Burst
Mode operation in which the internal power MOSFETs
operate intermittently based on load demand.
In Burst Mode operation, the peak current of the inductor
is set to approximately 60mA regardless of the output load.
Each burst event can last from a few cycles at light loads
to almost continuously cycling with short sleep intervals
at moderate loads. In between these burst events, the
power MOSFETs and any unneeded circuitry are turned off,
reducing the quiescent current to 26µA. In this sleep state,
the load current is being supplied solely from the output
capacitor. As the output voltage drops, the EA amplifier’s
output rises above the sleep threshold and turns the top
MOSFET on. This process repeats at a rate that is dependent
on the load demand. By running cycles periodically, the
switching losses which are dominated by the gate charge
losses of the power MOSFETs are minimized.
Low Supply Operation
To prevent unstable operation, the LTC3563 incorporates
an undervoltage lockout circuit which shuts down the part
when the input voltage drops below 2V.
Internal Soft-Start
At start-up when the RUN pin is brought high, the internal
reference is linearly ramped from 0V to 0.6V in about
1ms. The regulated output voltage follows this ramp from
0% to 100% in 1ms. The current in the inductor during
soft-start is defined by the combination of the current
needed to charge the output capacitance and the current
provided to the load as the output voltage ramps up. The
start-up waveform, shown in the Typical Performance
Characteristics, shows the output voltage start-up from
0V to 1.87V with a 500mA load and VIN = 3.6V.
3563f
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LTC3563
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APPLICATIO S I FOR ATIO
A general LTC3563 application circuit is shown in Figure1.
External component selection is driven by the load requirement and begins with the selection of the inductor L. Once
the inductor is chosen, CIN and COUT can be selected.
L
VIN
2.7V TO 5.5V
VIN
CIN
SW
LTC3563
RUN
1.28V 1.87V
VOUT
VSEL
COUT
VOUT
GND
3563 F01
Figure 1. LTC3563 General Schematic
Inductor Selection
The inductor value has a direct effect on ripple current ΔIL,
which decreases with higher inductance and increases with
higher VIN or VOUT, as shown in following equation:
∆IL =
VOUT ⎛ VOUT ⎞
1–
ƒ O • L ⎜⎝
VIN ⎟⎠
where fO is the switching frequency. A reasonable starting
point for setting ripple current is ΔIL = 0.4 • IOUT(MAX),
where IOUT(MAX) is 500mA. The largest ripple current ΔIL
occurs at the maximum input voltage. To guarantee that
the ripple current stays below a specified maximum, the
inductor value should be chosen according to the following equation:
L=
VOUT ⎛
VOUT ⎞
⎜ 1–
⎟
ƒ O • ∆IL ⎝ VIN(MAX ) ⎠
The DC current rating of the inductor should be at least
equal to the maximum load current plus half the ripple
current to prevent core saturation. Thus, a 600mA rated
inductor should be enough for most applications (500mA
+ 100mA). For better efficiency, choose a low DC-resistance inductor.
The inductor value will also have an effect on Burst Mode
operation. The transition to low current operation begins
when the inductor’s peak current falls below a level set by
the burst clamp. Lower inductor values result in higher
ripple current which causes the transition to occur at lower
load currents. This causes a dip in efficiency in the upper
range of low current operation. In Burst Mode operation,
lower inductance values cause the burst frequency to
increase.
Inductor Core Selection
Different core materials and shapes change the size/current
and price/current relationships of an inductor. Toroid or
shielded pot cores in ferrite or permalloy materials are small
and don’t radiate much energy, but generally cost more
than powdered iron core inductors with similar electrical
characteristics. The choice of which style inductor to use
often depends more on the price vs size requirements
and any radiated field/EMI requirements than on what the
LTC3563 requires to operate. Table 1 shows some typical surface mount inductors that work well in LTC3563
applications.
Input Capacitor (CIN) Selection
In continuous mode, the input current of the converter is a
square wave with a duty cycle of approximately VOUT/VIN.
To prevent large voltage transients, a low equivalent series
resistance (ESR) input capacitor sized for the maximum
RMS current must be used. The maximum RMS capacitor
current is given by:
IRMS ≈ IMAX
VOUT ( VIN – VOUT )
VIN
where the maximum average output current IMAX equals
the peak current minus half the peak-to-peak ripple current, IMAX = ILIM – ΔIL/2. This formula has a maximum at
VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case
is commonly used to design because even significant
deviations do not offer much relief. Note that capacitor
manufacturer’s ripple current ratings are often based on
only 2000 hours life time. This makes it advisable to further
derate the capacitor, or choose a capacitor rated at a higher
temperature than required. Several capacitors may also be
paralleled to meet the size or height requirements of the
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LTC3563
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APPLICATIO S I FOR ATIO
Table 1. Representative Surface Mount Inductors
PART NUMBER
VALUE
(µH)
MAX DC
CURRENT
(A)
DCR
(Ω)
SIZE (mm3)
CDRH2D11
2.2
0.780
0.098
3.2 × 3.2 × 1.2
CDRH3D16
2.2
1.2
0.075
3.8 × 3.8 × 1.8
MANUFACTURER
Sumida
Murata
TDK
CMD4D11
2.2
0.95
0.116
4.4 × 5.8 × 1.2
CDH2D09B
3.3
0.85
0.15
2.8 × 3 × 1
4.9 × 4.9 × 1
CLS4D09
4.7
0.75
0.15
LQH32CN
2.2
0.79
0.097 2.5 × 3.2 × 1.55
LQH43CN
4.7
0.75
0.15
4.5 × 3.2 × 2.6
IVLC453232
2.2
0.85
0.18
4.8 × 3.4 × 3.4
VLF3010AT2R2M1R0
2.2
1.0
0.12
2.8 × 2.6 × 1
design. An additional 0.1µF to 1µF ceramic capacitor is
also recommended on VIN for high frequency decoupling,
when not using an all ceramic capacitor solution.
T495 series, and Sprague 593D and 595D series. Consult
the manufacturer for other specific recommendations.
Ceramic Input and Output Capacitors
Output Capacitor (COUT) Selection
The selection of COUT is driven by the required ESR to
minimize voltage ripple and load step transients. Typically,
once the ESR requirement is satisfied, the RMS current
rating generally far exceeds the IRIPPLE(P-P) requirement,
except for an all ceramic solution. The output ripple (ΔVOUT)
is determined by:
⎛
⎞
1
∆VOUT ≈ ∆IL ⎜ ESR +
8 • ƒO • COUT ⎟⎠
⎝
where fO is the switching frequency, COUT is the output
capacitance and ΔIL is the inductor ripple current. For a fixed
output voltage, the output ripple is highest at maximum
input voltage since ΔIL increases with input voltage.
If tantalum capacitors are used, it is critical that the capacitors are surge tested for use in switching power supplies.
An excellent choice is the AVX TPS series of surface mount
tantalums, available in case heights ranging from 2mm to
4mm. These are specially constructed and tested for low
ESR so they give the lowest ESR for a given volume. Other
capacitor types include Sanyo POSCAP, Kemet T510 and
Higher value, lower cost ceramic capacitors are now
becoming available in smaller case sizes. Their high
ripple current rating, high voltage rating and low ESR
are tempting for switching regulator use. However, the
ESR is so low that it can cause loop stability problems.
Since the LTC3563’s control loop does not depend on
the output capacitor’s ESR for stable operation, ceramic
capacitors can be used to achieve very low output ripple
and small circuit size. X5R or X7R ceramic capacitors are
recommended because these dielectrics have the best
temperature and voltage characteristics of all the ceramics
for a given value and size.
Great care must be taken when using only ceramic input
and output capacitors. When a ceramic capacitor is used
at the input and the power is being supplied through long
wires, such as from a wall adapter, a load step at the output
can induce ringing at the VIN pin. At best, this ringing can
couple to the output and be mistaken as loop instability.
At worst, the ringing at the input can be large enough to
damage the part. For more information, see Application
Note 88. The recommended capacitance value to use is
10µF for both input and output capacitors.
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LTC3563
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APPLICATIO S I FOR ATIO
Efficiency Considerations
The efficiency of a switching regulator is equal to the output
power divided by the input power times 100%. It is often
useful to analyze individual losses to determine what is
limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage of input power.
Although all dissipative elements in the circuit produce
losses, three main sources usually account for most of
the losses in LTC3563 circuits: 1) VIN quiescent current,
2) I2R loss and 3) switching loss. VIN quiescent current
loss dominates the power loss at very low load currents,
whereas the other two dominate at medium to high load
currents. In a typical efficiency plot, the efficiency curve
at very low load currents can be misleading since the
actual power loss is of no consequence as illustrated in
Figure 2.
1000
VIN = 3.6V
POWER LOSS (mW)
100
10
1
0.1
0.1
VOUT = 1.87V
VOUT = 1.28V
1
10
100
OUTPUT CURRENT (mA)
1000
3563 F02
Figure 2. Power Loss vs Load Current
1) The VIN quiescent current is the DC supply current given
in the Electrical Characteristics which excludes MOSFET
charging current. VIN current results in a small (<0.1%)
loss that increases with VIN, even at no load.
2) I2R losses are calculated from the DC resistances of
the internal switches, RSW, and external inductor, RL. In
continuous mode, the average output current flows through
inductor L, but is “chopped” between the internal top and
bottom switches. Thus, the series resistance looking into
the SW pin is a function of both top and bottom MOSFET
RDS(ON) and the duty cycle (D) as follows:
RSW = (RDS(ON)TOP)(D) + (RDS(ON)BOT)(1 – D)
The RDS(ON) for both the top and bottom MOSFETs can
be obtained from the Typical Performance Characteristics
curves. Thus, to obtain I2R losses:
I2R losses = IOUT2(RSW + RL)
3) The switching current is MOSFET gate charging current,
that results from switching the gate capacitance of the
power MOSFETs. Each time a MOSFET gate is switched
from low to high to low again, a packet of charge dQ moves
from VIN to ground. The resulting dQ/dt is a current out of
VIN that is typically much larger than the DC bias current.
In continuous mode, IGATECHG = fO(QT + QB), where QT
and QB are the gate charges of the internal top and bottom
MOSFET switches. The gate charge losses are proportional
to VIN and thus their effects will be more pronounced at
higher supply voltages.
Other “hidden” losses such as copper trace and internal
battery resistances can account for additional efficiency
degradations in portable systems. The internal battery
and fuse resistance losses can be minimized by making
sure that CIN has adequate charge storage and very low
ESR at the switching frequency. Other losses include
diode conduction losses during dead-time and inductor
core losses generally account for less than 2% total additional loss.
3563f
11
LTC3563
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APPLICATIO S I FOR ATIO
Thermal Considerations
In most applications the LTC3563 does not dissipate much
heat due to its high efficiency. But in applications where the
LTC3563 is running at high ambient temperature with low
supply voltage and high duty cycles, such as in dropout,
the heat dissipated may exceed the maximum junction
temperature of the part. If the junction temperature reaches
approximately 150°C, both power switches will be turned
off and the SW node will become high impedance.
To avoid the LTC3563 from exceeding the maximum
junction temperature, the user needs to do some thermal
analysis. The goal of the thermal analysis is to determine
whether the power dissipated exceeds the maximum
junction temperature of the part. The temperature rise is
given by:
TR = (PD)(θJA)
where PD is the power dissipated by the regulator and
θJA is the thermal resistance from the junction of the die
to the ambient.
The junction temperature, TJ, is given by:
TJ = TA + TR
where TA is the ambient temperature.
As an example, consider the LTC3563 with an output
voltage of 1.87V, an input voltage of 2.7V, a load current
of 500mA and an ambient temperature of 70°C. From
the typical performance graph of switch resistance, the
RDS(ON) of the P-channel switch at 70°C is approximately
0.7Ω and the RDS(ON) of the N-channel synchronous
switch is approximately 0.4Ω. The duty cycle in this case
is approximately 70%.
The series resistance looking into the SW pin is:
RSW = 0.7Ω (0.7) + 0.4Ω (0.3) = 0.61Ω
Therefore, for the power dissipated by the part is:
PD = ILOAD2 • RSW = 152.5mW
For the DFN package, the θJA is 40°C/W. Thus, the junction
temperature of the regulator is:
TJ = 70°C + (0.1525)(40) = 76.1°C
which is below the maximum junction temperature of
125°C.
Note that at higher supply voltages, the junction temperature
is lower due to reduced switch resistance (RDS(ON)).
Checking Transient Response
The regulator loop response can be checked by looking
at the load transient response. Switching regulators take
several cycles to respond to a step in load current. When
a load step occurs, VOUT immediately shifts by an amount
equal to ΔILOAD • ESR, where ESR is the effective series
resistance of COUT. ΔILOAD also begins to charge or discharge COUT, generating a feedback error signal used by the
regulator to return VOUT to its steady-state value. During
this recovery time, VOUT can be monitored for overshoot
or ringing that would indicate a stability problem.
The output voltage settling behavior is related to the stability
of the closed-loop system and will demonstrate the actual
overall supply performance. For a detailed explanation of
optimizing the compensation components, including a review of control loop theory, refer to Application Note 76.
In some applications, a more severe transient can be caused
by switching loads with large (>1µF) bypass capacitors.
The discharged bypass capacitors are effectively put in
parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem, if
the switch connecting the load has low resistance and is
driven quickly. The solution is to limit the turn-on speed of
the load switch driver. A Hot SwapTM controller is designed
specifically for this purpose and usually incorporates current limit, short circuit protection and soft-start.
Design Example
As a design example, assume the LTC3563 is used in
a single lithium-ion battery-powered cellular phone application. The VIN will be operating from a maximum of
4.2V down to about 2.7V. The load current requirement
is a maximum of 0.5A, but most of the time it will be in
standby mode, requiring only 2mA. Efficiency at both
low and high load currents is important. Output voltage
is either 1.87V or 1.28V.
3563f
12
LTC3563
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APPLICATIO S I FOR ATIO
With this information we can calculate L using:
L=
CIN will require an RMS current rating of at least
0.25A ≅ ILOAD(MAX)/2 at temperature and COUT will require
ESR of less than 0.2Ω. In most cases, ceramic capacitors
will satisfy these requirements. Select COUT = 10µF and
CIN = 10µF.
⎛ V ⎞
1
• VOUT • ⎜ 1– OUT ⎟
f • ∆IL
VIN ⎠
⎝
Substituting VOUT = 1.87V, VIN = 4.2V, ΔIL = 200mA and
f = 2.25MHz gives:
Figure 3 shows the complete circuit along with its efficiency
curve, load step response and recommended layout.
1.87 V
⎛ 1.87 V ⎞
• ⎜ 1–
= 2.31µH
2.25MHz • 200mA ⎝
4.2V ⎟⎠
With VOUT = 1.28 V
L=
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of
the LTC3563. These items are also illustrated graphically
in Figure 3b. Check the following in your layout:
1.28 V
⎛ 1.28 V ⎞
• ⎜ 1–
= 1.98µH
L=
2.25MHz • 200mA ⎝
4.2V ⎟⎠
Choosing a vendor’s closest inductor value of 2.2µH results
in a maximum ripple current of:
1. The power traces, consisting of the GND trace, the SW
trace and the VIN trace should be kept short, direct and
wide.
For VOUT = 1.87 V
∆IL =
2. Does the (+) plate of CIN connect to VIN as closely as
possible? This capacitor provides the AC current to the
internal power MOSFETs.
1.87 V
⎛ 1.87 V ⎞
• ⎜1 –
= 209.6mA
2.25MHz • 2.2µH ⎝
4.2V ⎟⎠
For VOUT = 1.28 V
3. Keep the (–) plates of CIN and COUT as close as possible.
1.28 V
⎛ 1.28 V ⎞
• ⎜1 –
= 179.8mA
∆IL =
2.25MHz • 2.2µH ⎝
4.2V ⎟⎠
Hot Swap is a trademark of Linear Technology Corporation.
L
VIN
2.7V TO 5.5V
VIN
CIN
SW
LTC3563
RUN
1.28V 1.87V
VOUT
VSEL
COUT
VOUT
3563 F03
GND
Figure 3a. Typical Application
3563f
13
LTC3563
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APPLICATIO S I FOR ATIO
VIA TO VOUT
GND
VIN
6 RUN
VOUT 1
VIN 2
CIN
GND
5 VSEL
4 SW
GND 3
L
COUT
GND
VOUT
3563 F03b
Figure 3b. Layout Diagram
100
90
VOUT
100mV/DIV
AC COUPLED
80
EFFICIENCY (%)
70
IL
500mA/DIV
60
50
40
ILOAD
500mA/DIV
30
20
10
VOUT = 1.87V
FIGURE 3a CIRCUIT
0
0.1
VIN = 2.7V
VIN = 3.6V
VIN = 4.2V
10
100
1
OUTPUT CURRENT (mA)
20µs/DIV
VIN = 3.6V
VOUT = 1.87V
ILOAD = 0A TO 500mA
3563 G04
1000
3563 G17
Figure 3d. Load Step
Figure 3c. Efficiency Curve
3563f
14
LTC3563
U
PACKAGE DESCRIPTIO
DC Package
6-Lead Plastic DFN (2mm × 2mm)
(Reference LTC DWG # 05-08-1703)
0.675 ±0.05
2.50 ±0.05
1.15 ±0.05 0.61 ±0.05
(2 SIDES)
PACKAGE
OUTLINE
0.25 ± 0.05
0.50 BSC
1.42 ±0.05
(2 SIDES)
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
R = 0.115
TYP
0.56 ± 0.05
(2 SIDES)
0.38 ± 0.05
4
6
2.00 ±0.10
(4 SIDES)
PIN 1 BAR
TOP MARK
(SEE NOTE 6)
PIN 1
CHAMFER OF
EXPOSED PAD
3
0.200 REF
0.75 ±0.05
1
(DC6) DFN 1103
0.25 ± 0.05
0.50 BSC
1.37 ±0.05
(2 SIDES)
0.00 – 0.05
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WCCD-2)
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE
TOP AND BOTTOM OF PACKAGE
3563f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
15
LTC3563
U
TYPICAL APPLICATIO
Using Low Profile Components, <1mm Height
2.2µH**
VIN
2.7V TO 5.5V
VIN
CIN*
10µF
CER
VOUT
1.28V/1.87V
500mA
SW
LTC3563
RUN
VOUT
COUT*
10µF
CER
VSEL
1.28V 1.87V
3563 TA02a
GND
*MURATA GRM219R60J106KE19
**TDK VLF3010AT-2R2M1R0
Efficiency vs Output Current
100
90
VIN = 3.6V
80
EFFICIENCY (%)
70
VOUT = 1.87V
VOUT = 1.28V
60
50
40
30
20
10
0
0.1
1
10
100
OUTPUT CURRENT (mA)
1000
3563 TA02b
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC3405/LTC3405B
300mA IOUT, 1.5MHz, Synchronous Step-Down DC/DC
Converter
95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 20µA,
ISD < 1µA, ThinSOT Package
LTC3406/LTC3406B
600mA IOUT, 1.5MHz, Synchronous Step-Down DC/DC
Converter
96% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 20µA,
ISD < 1µA, ThinSOT Package
LTC3407/LTC3407-2
Dual 600mA/800mA IOUT, 1.5MHz/2.25MHz, Synchronous 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 40µA,
Step-Down DC/DC Converter
ISD < 1µA, MS10E, DFN Packages
LTC3409
600mA IOUT, 1.7MHz/2.6MHZ, Synchronous Step-Down
DC/DC Converter
96% Efficiency, VIN: 1.6V to 5.5V, VOUT(MIN) = 0.6V, IQ = 65µA,
ISD < 1µA, DFN Package
LTC3410/LTC3410B
300mA IOUT, 2.25MHz, Synchronous Step-Down DC/DC
Converter
95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 26µA,
ISD < 1µA, SC70 Package
LTC3411
1.25A IOUT, 4MHz, Synchronous Step-Down DC/DC
Converter
95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 60µA,
ISD < 1µA, MS10, DFN Packages
LTC3542
500mA IOUT, 2.25MHz, Synchronous Step-Down DC/DC
Converter
96% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 26µA,
ISD < 1µA, DFN Packages
LTC3548
Dual 400mA/800mA IOUT, 2.25MHz, Synchronous
Step-Down DC/DC Converter
95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 40µA,
ISD < 1µA, MS10, DFN Packages
LTC3561
1A IOUT, 4MHz Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 2.6V to 5.5V, VOUT(MIN) = 0.8V, IQ = 240µA,
ISD < 1µA, 3mm × 3mm DFN Package
3563f
16 Linear Technology Corporation
LT 0107 • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
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