CA5160 CT O DU ODUC T R P PR TE O LE TI TU T E S B 0 O S SUB 0, CA313 LE Data B 4 I 1 Sheet S 3 CA P OS ® 4MHz, BiMOS Microprocessor Operational Amplifier with MOSFET Input/CMOS Output CA5160 is an integrated circuit operational amplifier that combines the advantage of both CMOS and bipolar transistors on a monolithic chip. The CA5160 is a frequency compensated version of the popular CA5130 series. It is designed and guaranteed to operate in microprocessor or logic systems that use +5V supplies. Gate-protected P-Channel MOSFET (PMOS) transistors are used in the input circuit to provide very high input impedance, very low input current, and exceptional speed performance. The use of PMOS field effect transistors in the input stage results in common-mode input voltage capability down to 0.5V below the negative supply terminal, an important attribute in single supply applications. A complementary symmetry MOS (CMOS) transistor pair, capable of swinging the output voltage to within 10mV of either supply voltage terminal (at very high values of load impedance), is employed as the output circuit. The CA5160 operates at supply voltages ranging from +5V to +16V, or ±2.5V to ±8V when using split supplies, and have terminals for adjustment of offset voltage for applications requiring offset-null capability. Terminal provisions are also made to permit strobing of the output stage. It has guaranteed specifications for 5V operation over the full military temperature range of -55oC to 125oC. CA5160M96 (5160) TEMP. RANGE (oC) -55 to 125 FN1924.7 Features • MOSFET Input Stage - Very High ZI; 1.5TΩ (1.5 x 1012Ω) (Typ) - Very Low II; at 15V Operation. . . . . . . . . . . . . 5pA (Typ) at 5V Operation . . . . . . . . . . . . . 2pA (Typ) • Common-Mode Input Voltage Range Includes Negative Supply Rail; Input Terminals Can be Swung 0.5V Below Negative Supply Rail • CMOS Output Stage Permits Signal Swing to Either (or Both) Supply Rails • CA5160 Has Full Military Temperature Range Guaranteed Specifications for V+ = 5V • CA5160 is Guaranteed to Operate Down to 4.5V for AOL • CA5160 is Guaranteed Up to ±7.5V Applications • Ground Referenced Single Supply Amplifiers • Fast Sample-Hold Amplifiers • Long Duration Timers/Monostables • Ideal Interface With Digital CMOS • High Input Impedance Wideband Amplifiers • Voltage Followers (e.g., Follower for Single Supply D/A Converter) • Wien-Bridge Oscillators • Voltage Controlled Oscillators Part Number Information PART NUMBER (BRAND) May 2003 PACKAGE 8 Ld SOIC Tape and Reel PKG. NO. M8.15 • Photo Diode Sensor Amplifiers • 5V Logic Systems • Microprocessor Interface Pinout CA5160 (SOIC) TOP VIEW OFFSET NULL 1 8 STROBE INV. INPUT 2 7 V+ NON INV. INPUT 3 6 OUTPUT V- 4 5 OFFSET NULL + NOTE: CA5160 devices have an on-chip frequency compensation network. Supplementary phase-compensation or frequency roll-off (if desired) can be connected externally between terminals 1 and 8. 1 CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 321-724-7143 | Intersil (and design) is a registered trademark of Intersil Americas Inc. Copyright © Intersil Americas Inc. 2003. All Rights Reserved All other trademarks mentioned are the property of their respective owners. CA5160 Absolute Maximum Ratings Thermal Information Supply Voltage (V+ to V-). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .16V Differential Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .8V DC Input Voltage . . . . . . . . . . . . . . . . . . . . . . (V+ +8V) to (V- -0.5V) Input Current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1mA Output Short Circuit Duration (Note 2). . . . . . . . . . . . . . . . Indefinite Thermal Resistance (Typical, Note 1) θJA (oC/W)θJC (oC/W) SOIC Package . . . . . . . . . . . . . . . . . . . 165 N/A Maximum Junction Temperature (Plastic Package) . . . . . . . .150oC Maximum Storage Temperature Range . . . . . . . . . . -65oC to 150oC Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . .300oC (SOIC - Lead Tips Only) Operating Conditions Temperature Range . . . . . . . . . . . . . . . . . . . . . . . . . -55oC to 125oC CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied. NOTES: 1. θJA is measured with the component mounted on a low effective thermal conductivity test board in free air. See Tech Brief TB379 for details. 2. Short circuit may be applied to ground or to either supply. TA = 25oC, V+ = 5V, V- = 0V, Unless Otherwise Specified Electrical Specifications PARAMETER TEST CONDITIONS SYMBOL MIN TYP MAX UNITS Input Offset Voltage VIO VO = 2.5V - 2 10 mV Input Offset Current IIO VO = 2.5V - 0.1 10 pA II VO = 2.5V - 2 15 pA VCM = 0 to 1V 70 80 - dB VCM = 0 to 2.5V 60 69 - dB VlCR+ 2.5 2.8 - V VlCR- - -0.5 0 V ∆V+ = 1V; ∆V- = 1V 55 67 - dB RL = ∞ 95 117 - dB RL = 10kΩ 85 102 - dB ISOURCE VO = 0V 1.0 3.4 4.0 mA ISINK VO = 5V 1.0 2.2 4.8 mA VOUT RL = ∞ 4.99 5 - V - 0 0.01 V 4.4 4.7 - V - 0 0.01 V 2.5 3.3 - V - 0 0.01 V Input Current Common Mode Rejection Ratio CMRR Common Mode Input Voltage Range Power Supply Rejection Ratio Large Signal Voltage Gain (Note 3) PSRR VO = 0.1 to 4.1V VO = 0.1 to 3.6V Source Current Sink Current Maximum Output Voltage AOL VOM+ VOMVOM+ RL = 10kΩ VOMVOM+ RL = 2kΩ VOMSupply Current ISUPPLY VO = 0V - 50 100 µA ISUPPLY VO = 2.5V - 320 400 µA NOTE: 3. For V+ = 4.5V and V- = GND; VOUT = 0.5V to 3.2V at RL = 10kΩ. 2 CA5160 TA = -55oC to 125oC, V+ = 5V, V- = 0V, Unless Otherwise Specified Electrical Specifications PARAMETER SYMBOL TEST CONDITIONS MIN TYP MAX UNITS Input Offset Voltage VIO VO = 2.5V - 3 15 mV Input Offset Current IIO VO = 2.5 V - 0.1 10 nA II VO = 2.5V - 2 15 nA VCM = 0 to 1V 60 80 - dB VCM = 0 to 2.5V 50 75 - dB VlCR+ 2.5 2.8 - V VlCR- - -0.5 0 V ∆V+ = 2V 40 60 - dB RL = ∞ 90 110 - dB RL = 10kΩ 75 100 - dB ISOURCE VO = 0V 0.6 - 5.0 mA ISINK VO = 5V 0.6 - 6.2 mA VOUT RL= ∞ 4.99 5 - V - 0 0.01 V 4.0 4.3 - V - 0 0.01 V 2.0 2.5 - V - 0 0.01 V Input Current Common Mode Rejection Ratio CMRR Common Mode Input Voltage Range Power Supply Rejection Ratio PSRR Large Signal Voltage Gain (Note 4) VO = 0.1 to 4.1V AOL VO = 0.1 to 3.6V Source Current Sink Current Maximum Output Voltage VOM+ VOMVOM+ RL = 10kΩ VOMVOM+ RL = 2kΩ VOMSupply Current VO = 0V ISUPPLY - 170 220 µA VO = 2.5V ISUPPLY - 410 500 µA MIN TYP MAX UNITS NOTE: 4. For V+ = 4.5V and V- = GND; VOUT = 0.5V to 3.2V at RL = 10kΩ. TA = 25oC, V+ = 15V, V- = 0V, Unless Otherwise Specified Electrical Specifications PARAMETER SYMBOL TEST CONDITIONS Input Offset Voltage VIO VS = ±7.5V - 6 15 mV Input Offset Current IIO VS = ±7.5V - 0.5 30 pA II VS = ±7.5V - 5 50 pA 50 320 - kV/V 94 110 - dB CMRR 70 90 - dB VlCR 10 -0.5 to 12 0 V - 32 320 µV/V 12 13.3 - V - 0.002 0.01 V 14.99 15 - V - 0 0.1 V VO = 0V 12 22 45 mA VO = 15V 12 20 45 mA Input Current Large Signal Voltage Gain AOL Common Mode Rejection Ratio Common Mode Input Voltage Range Power Supply Rejection Ratio Maximum Output Voltage VOM+ VO = 10VP-P RL = 2kΩ PSRR ∆V+ = 1V; ∆V- = 1V VS = ±7.5V VOUT RL = 2kΩ VOMRL = ∞ VOM+ VOMMaximum Output Current IOM+ (Source) IOM- (Sink) 3 IO CA5160 TA = 25oC, V+ = 15V, V- = 0V, Unless Otherwise Specified (Continued) Electrical Specifications PARAMETER SYMBOL Supply Current I+ Input Offset Voltage Temperature Drift TEST CONDITIONS MIN TYP MAX UNITS RL = ∞, VO = 7.5V - 10 15 mA RL = ∞, VO = 0V - 2 3 mA ∆VIO/∆T - 8 - µV/oC For Design Guidance, At TA = 25oC, VSUPPLY = ±7.5V, Unless Otherwise Specified Electrical Specifications PARAMETER SYMBOL TEST CONDITIONS Input Offset Voltage Adjustment Range 10kΩ Across Terminals 4 and 5 or 4 and 1 TYP UNITS ±22 mV 1.5 TΩ Input Resistance RI Input Capacitance CI f = 1MHz 4.3 pF Equivalent Input Noise Voltage eN BW = 0.2MHz, RS = 1MΩ 40 µV BW = 0.2MHz, RS = 10MΩ 50 µV RS = 100Ω, 1kHz 72 nV/√Hz RS = 100Ω, 10kHz 30 nV/√Hz Equivalent Input Noise Voltage eN Unity Gain Crossover Frequency fT 4 MHz Slew Rate SR 10 V/µs 0.09 µs 10 % 1.8 µs Transient Response Rise Time tR Overshoot OS Settling Time (To <0.1%, VIN = 4VP-P) CC = 25pF, RL = 2kΩ (Voltage Follower) tS CC = 25pF, RL = 2kΩ, (Voltage Follower) Block Diagram 7 200µA 1.35mA 8mA (NOTE 5) 0mA (NOTE 6) 200µA V+ NOTES: 5. Total supply voltage (for indicated voltage gains) = 15V with input terminals biased so that Terminal 6 potential is +7.5V above Terminal 4. BIAS CKT. 6. Total supply voltage (for indicated voltage gains) = 15V with output terminal driven to either supply rail. + 3 OUTPUT AV ≈ 6000X AV ≈ 5X INPUT AV ≈ 30X 6 2 - 4 CC 5 1 OFFSET NULL 8 COMPENSATION (WHEN DESIRED) 4 STROBE V- CA5160 Schematic Diagram CURRENT SOURCE FOR Q6 AND Q7 BIAS CIRCUIT Q1 D1 D2 Z1 8.3V R1 40kΩ 7 “CURRENT SOURCE LOAD” FOR Q11 Q2 Q3 Q4 Q5 V+ D3 D4 R2 5kΩ INPUT STAGE D5 NON-INV. INPUT 3 D7 D6 SECOND STAGE OUTPUT STAGE + Q6 Q7 R3 1kΩ R5 1kΩ 6 30 pF R4 1kΩ Q9 OUTPUT 2kΩ 2 INV. INPUT Q8 Q10 Q12 Q11 R6 1kΩ 5 1 OFFSET NULL SUPPLEMENTARY COMP IF DESIRED 8 4 STROBING NOTE: Diodes D5 through D7 provide gate oxide protection for MOSFET Input Stage. Application Information Circuit Description Refer to the block diagram of the CA5160 CMOS Operational Amplifier. The input terminals may be operated down to 0.5V below the negative supply rail, and the output can be swung very close to either supply rail in many applications. Consequently, the CA5160 circuit is ideal for single supply operation. Three class A amplifier stages, having the individual gain capability and current consumption shown in the block diagram, provide the total gain of the CA5160. A biasing circuit provides two potentials for common use in the first and second stages. Terminals 8 and 1 can be used to supplement the internal phase compensation network if additional phase compensation or frequency roll-off is desired. Terminals 8 and 4 can also be used to strobe the output stage into a low quiescent current state. When Terminal 8 is tied to the negative supply rail (Terminal 4) by mechanical or electrical means, the output potential at Terminal 6 essentially rises to the positive supply rail potential at Terminal 7. This condition of essentially zero current drain in the output stage under the strobed “OFF” 5 condition can only be achieved when the ohmic load resistance presented to the amplifier is very high (e.g., when the amplifier output is used to drive CMOS digital circuits in comparator applications). Input Stages The circuit of the CA5160 is shown in the schematic diagram. It consists of a differential input stage using PMOS field effect transistors (Q6, Q7) working into a mirror pair of bipolar transistors (Q9, Q10) functioning as load resistors together with resistors R3 through R6. The mirror pair transistors also function as a differential-to-single-ended converter to provide base drive to the second-stage bipolar transistor (Q11). Offset nulling, when desired, can be effected by connecting a 100,000Ω potentiometer across Terminals 1 and 5 and the potentiometer slider arm to Terminal 4. Cascode-connected PMOS transistors Q2, Q4, are the constant current source for the input stage. The biasing circuit for the constant current source is subsequently described. The small diodes D5 through D7 provide gateoxide protection against high voltage transients, including static electricity during handling for Q6 and Q7. CA5160 Second Stage Offset Nulling Most of the voltage gain in the CA5160 is provided by the second amplifier stage, consisting of bipolar transistor Q11 and its cascode-connected load resistance provided by PMOS transistors Q3 and Q5. The source of bias potentials for these PMOS transistors is described later. Miller Effect compensation (roll off) is accomplished by means of the 30pF capacitor and 2kΩ resistor connected between the base and collector of transistor Q11. These internal components provide sufficient compensation for unity gain operation in most applications. However, additional compensation, if desired, may be used between Terminals 1 and 8. Offset voltage nulling is usually accomplished with a 100,000Ω potentiometer connected across Terminals 1 and 5 and with the potentiometer slider arm connected to Terminal 4. A fine offset null adjustment usually can be affected with the slider arm positioned in the mid point of the potentiometer’s total range. At total supply voltages, somewhat above 8.3V, resistor R2 and zener diode Z1 serve to establish a voltage of 8.3V across the series connected circuit, consisting of resistor R1, diodes D1 through D4, and PMOS transistor Q1. A tap at the junction of resistor R1 and diode D4 provides a gate bias potential of about 4.5V for PMOS transistors Q4 and Q5 with respect to Terminal 7. A potential of about 2.2V is developed across diode connected PMOS transistor Q1 with respect to Terminal 7 to provide gate bias for PMOS transistors Q2 and Q3. It should be noted that Q1 is “mirror connected” to both Q2 and Q3. Since transistors Q1, Q2 and Q3 are designed to be identical, the approximately 200µA current in Q1 establishes a similar current in Q2 and Q3 as constant current sources for both the first and second amplifier stages, respectively. At total supply voltages somewhat less than 8.3V, zener diode Z1 becomes non-conductive and the potential, developed across series connected R1, D1-D4, and Q1 varies directly with variations in supply voltage. Consequently, the gate bias for Q4, Q5 and Q2, Q3 varies in accordance with supply voltage variations. This variation results in deterioration of the power supply rejection ration (PSRR) at total supply voltages below 8.3V. Operation at total supply voltages below about 4.5V results in seriously degraded performance. Output Stage The output stage consists of a drain loaded inverting amplifier using CMOS transistors operating in the Class A mode. When operating into very high resistance loads, the output can be swung within millivolts of either supply rail. Because the output stage is a drain loaded amplifier, its gain is dependent upon the load impedance. The transfer characteristics of the output stage for a load returned to the negative supply rail are shown in Figure 20. Typical op-amp loads are readily driven by the output stage. Because large signal excursions are nonlinear, requiring feedback for good waveform reproduction, transient delays may be encountered. As a voltage follower, the amplifier can achieve 0.01% accuracy levels, including the negative supply rail. 6 As shown in the Table of Electrical Specifications, the input current for the CA5160 Series Op Amps is typically 5pA at TA = 25oC when Terminals 2 and 3 are at a common-mode potential of +7.5V with respect to negative supply Terminal 4. Figure 1 contains data showing the variation of input current as a function of common-mode input voltage at TA = 25oC. These data show that circuit designers can advantageously exploit these characteristics to design circuits which typically require an input current of less than 1pA, provided the common-mode input voltage does not exceed 2V. As previously noted, the input current is essentially the result of the leakage current through the gate-protection diodes in the input circuit and, therefore, a function of the applied voltage. Although the finite resistance of the glass terminal-to-case insulator of the metal can package also contributes an increment of leakage current, there are useful compensating factors. Because the gate-protection network functions as if it is connected to Terminal 4 potential, and the metal can case of the CA5160 is also internally tied to Terminal 4, input terminal 3 is essentially “guarded” from spurious leakage currents. 10 TA = 25oC 7.5 INPUT VOLTAGE (V) Bias-Source Circuit Input Current Variation with Common Mode Input Voltage 5 V+ 15V TO 5V 7 2 CA5160 PA 6 3 2.5 4 VIN 8 0V TO V- -10V 0 -1 0 1 2 3 4 5 6 7 INPUT CURRENT (pA) FIGURE 1. CA5160 INPUT CURRENT vs COMMON MODE VOLTAGE Input Current Variation with Temperature The input current of the CA5160 series circuits is typically 5pA at 25oC. The major portion of this input current is due to leakage current through the gate protective diodes in the input circuit. As with any semiconductor-junction device, including op amps with a junction-FET input stage, the CA5160 In applications requiring the lowest practical input current and incremental increases in current because of “warm-up” effects, it is suggested that an appropriate heat sink be used with the CA5160. In addition, when “sinking” or “sourcing” significant output current the chip temperature increases, causing an increase in the input current. In such cases, heatsinking can also very markedly reduce and stabilize input current variations. 4000 TA = 125oC FOR METAL CAN PACKAGES 6 DIFFERENTIAL DC VOLTAGE 5 (ACROSS TERMINALS 2 AND 3) = 2V OUTPUT STAGE TOGGLED 4 3 2 DIFFERENTIAL DC VOLTAGE (ACROSS TERMINALS 2 AND 3) = 0V OUTPUT VOLTAGE = V+/2 1 0 0 VS = ±7.5V 500 1000 1500 2000 2500 3000 3500 4000 TIME (HOURS) 1000 INPUT CURRENT (pA) 7 OFFSET VOLTAGE SHIFT (mV) leakage current approximately doubles for every 10oC increase in temperature. Figure 2 provides data on the typical variation of input bias current as a function of temperature in the CA5160. FIGURE 3. TYPICAL INCREMENTAL OFFSET VOLTAGE SHIFT vs OPERATING LIFE 100 Power Supply Considerations 10 1 -80 -60 -40 -20 0 20 40 60 80 TEMPERATURE (oC) 100 120 140 FIGURE 2. INPUT CURRENT vs TEMPERATURE Input Offset Voltage (VIO) Variation with DC Bias vs Device Operating Life It is well known that the characteristics of a MOSFET device can change slightly when a DC gate-source bias potential is applied to the device for extended time periods. The magnitude of the change is increased at high temperatures. Users of the CA5160 should be alert to the possible impacts of this effect if the application of the device involves extended operation at high temperatures with a significant differential DC bias voltage applied across Terminals 2 and 3. Figure 3 shows typical data pertinent to shifts in offset voltage encountered with CA5160 devices in metal can packages during life testing. At lower temperatures (metal can and plastic) for example at 85oC, this change in voltage is considerably less. In typical linear applications where the differential voltage is small and symmetrical, these incremental changes are of about the same magnitude as those encountered in an operational amplifier employing a bipolar transistor input stage. The 2VDC differential voltage example represents conditions when the amplifier output state is “toggled”, e.g., as in comparator applications. 7 Because the CA5160 is very useful in single-supply applications, it is pertinent to review some considerations relating to power-supply current consumption under both single-and dual-supply service. Figures 4A and 4B show the CA5160 connected for both dual and single-supply operation. Dual-Supply Operation - When the output voltage at Terminal 6 is 0V, the currents supplied by the two power supplies are equal. When the gate terminals of Q8 and Q12 are driven increasingly positive with respect to ground, current flow through Q12 (from the negative supply) to the load is increased and current flow through Q8 (from the positive supply) decreases correspondingly. When the gate terminals of Q8 and Q12 are driven increasingly negative with respect to ground, current flow through Q8 is increased and current flow through Q12 is decreased accordingly. Single Supply Operation - Initially, let it be assumed that the value of RL is very high (or disconnected), and that the inputterminal bias (Terminals 2 and 3) is such that the output terminal (Number 6) voltage is at V+/2, i.e., the voltage-drops across Q8 and Q12 are of equal magnitude. Figure 21 shows typical quiescent supply-current vs supply-voltage for the CA5160 operated under these conditions. Since the output stage is operating as a Class A amplifier, the supply-current will remain constant under dynamic operating conditions as long as the transistors are operated in the linear portion of their voltage transfer characteristics (see Figure 20). If either Q8 or Q12 are swung out of their linear regions toward cutoff (a nonlinear region), there will be a corresponding reduction in supply-current. In the extreme case, e.g., with Terminal 8 swung down to ground potential (or tied to ground), NMOS transistor Q12 is completely cut off and the supply-current to series-connected transistors Q8, Q12 goes essentially to zero. The two preceding stages in the CA5160, however, continue CA5160 to draw modest supply-current (see the lower curve in Figure 21) even through the output stage is strobed off. Figure 4A shows a dual-supply arrangement for the output stage that can also be strobed off, assuming RL = ∞, by pulling the potential of Terminal 8 down to that of Terminal 4. Let it now be assumed that a load-resistance of nominal value (e.g., 2kΩ) is connected between Terminal 6 and ground in the circuit of Figure 4B. Let it further be assumed again that the input terminal bias (Terminals 2 and 3) is such that the output terminal (Number 6) voltage is V+/2. Since PMOS transistor Q8 must now supply quiescent current to both RL and transistor Q12, it should be apparent that under these conditions the supply current must increase as an inverse function of the RL magnitude. Figure 27 shows the voltage drop across PMOS transistor Q8 as a function of load current at several supply voltages. Figure 20 shows the voltage transfer characteristics of the output stage for several values of load resistance. test-circuit amplifier of Figure 5 is operated at a total supply voltage of 15V. This value of total input-referred noise remains essentially constant, even though the value of source resistance is raised by an order of magnitude. This characteristic is due to the fact that reactance of the input capacitance becomes a significant factor in shunting the source resistance. It should be noted, however, that for values of source resistance very much greater than 1MΩ, the total noise voltage generated can be dominated by the thermal noise contributions of both the feedback and source resistors. +7.5V 0.01µF RS 3 7 + 1MΩ 2 - 4 30.1kΩ V+ 0.01 µF -7.5V 7 3 2 + Q8 OUTPUT Q12 - NOISE VOLTAGE OUTPUT 6 STAGE BW (-3dB) = 200kHz TOTAL NOISE VOLTAGE (INPUT REFERRED) = 40µV (TYP) 6 RL 1kΩ FIGURE 5. TEST-CIRCUIT AMPLIFIER (30dB GAIN) USED FOR WIDEBAND NOISE MEASUREMENTS 4 8 Typical Applications V- FIGURE 4A. DUAL POWER-SUPPLY OPERATION Operational amplifiers with very high input resistances, like the CA5160, are particularly suited to service as voltage followers. Figure 6 shows the circuit of a classical voltage follower, together with pertinent waveforms using the CA5160 in a split supply-configuration. V+ 7 3 + Q8 OUTPUT STAGE 2 6 RL Q12 - Voltage Followers 4 8 FIGURE 4B. SINGLE POWER-SUPPLY OPERATION FIGURE 4. CA5160 OUTPUT STAGE IN DUAL AND SINGLE POWER SUPPLY OPERATION Wideband Noise From the standpoint of low-noise performance considerations, the use of the CA5160 is most advantageous in applications where in the source resistance of the input signal is on the order of 1MΩ or more. In this case, the total input-referred noise voltage is typically only 40µV when the 8 A voltage follower, operated from a single-supply, is shown in Figure 7 together with related waveforms. This follower circuit is linear over a wide dynamic range, as illustrated by the reproduction of the output waveform in Figure 7B with input signal ramping. The waveforms in Figure 7C show that the follower does not lose its input-to-output phase-sense, even though the input is being swung 7.5V below ground potential. This unique characteristic is an important attribute in both operational amplifier and comparator applications. Figure 7C also shows the manner in which the CMOS output stage permits the output signal to swing down to the negative supply rail potential (i.e., ground in the case shown). The digital-toanalog converter (DAC) circuit, described in the following section, illustrates the practical use of the CA5160 in a singlesupply voltage follower application. CA5160 9 Bit CMOS DAC +7.5V 0.01µF 3 7 + 10kΩ 6 2 - 4 2kΩ 0.01 µF -7.5V 2kΩ BW (-3dB) = 4MHz SR = 10V/µs 25pF SIMULATED LOAD CAPACITANCE 0.1µF FIGURE 6A. DUAL SUPPLY FOLLOWER Top Trace: Output Bottom Trace: Input FIGURE 6B. SMALL SIGNAL RESPONSE A typical circuit of a 9 bit Digital-to-Analog Converter (DAC) (see Note) is shown in Figure 8. This system combines the concepts of multiple-switch CMOS ICs, a low cost ladder network of discrete metal-oxide-film resistors, a CA5160 op amp connected as a follower, and an inexpensive monolithic regulator in a simple single power supply arrangement. An additional feature of the DAC is that it is readily interfaced with CMOS input logic, e.g., 10V logic levels are used in the circuit of Figure 8. The circuit uses an R/2R voltage-ladder network, with the output-potential obtained directly by terminating the ladder arms at either the positive or the negative power-supply terminal. Each CD4007A contains three “inverters”, each “inverter” functioning as a single-pole double-throw switch to terminate an arm of the R/2R network at either the positive or negative power-supply terminal. The resistor ladder is an assembly of 1% tolerance metal-oxide film resistors. The five arms requiring the highest accuracy are assembled with series and parallel combinations of 806,000Ω resistors from the same manufacturing lot. A single 15V supply provides a positive bus for the CA5160 follower amplifier and feeds the CA3085 voltage regulator. A “scale-adjust” function is provided by the regulator output control, set to a nominal 10V level in this system. The linevoltage regulation (approximately 0.2%) permits a 9 bit accuracy to be maintained with variations of several volts in the supply. The flexibility afforded by the CMOS building blocks simplifies the design of DAC systems tailored to particular needs. Error Amplifier in Regulated Power Supplies The CA5160 is an ideal choice for error-amplifier service in regulated power supplies since it can function as an erroramplifier when the regulated output voltage is required to approach 0V. The circuit shown in Figure 9 uses a CA5160 as an error amplifier in a continuously adjustable 1A power supply. One of the key features of this circuit is its ability to regulate down to the vicinity of zero with only one DC power supply input. An RC network, connected between the base of the output drive transistor and the input voltage, prevents “turn-on overshoot”, a condition typical of many operational-amplifier regulator circuits. As the amplifier becomes operational, this RC network ceases to have influence on the regulator performance. Top Trace: Output Signal Center Trace: Difference Signal 5mV/Div. Bottom Trace: Input Signal FIGURE 6C. INPUT-OUTPUT DIFFERENCE SIGNAL SHOWING SETTLING TIME FIGURE 6. SPLIT SUPPLY VOLTAGE FOLLOWER WITH ASSOCIATED WAVEFORMS 9 NOTE: “Digital-to-Analog Conversion Using the Intersil CD4007A CMOS IC”, Application Note AN6080. CA5160 Precision Voltage-Controlled Oscillator +15V 3 + 2 - 10kΩ The circuit diagram of a precision voltage-controlled oscillator is shown in Figure 10. The oscillator operates with a tracking error on the order of 0.02% and a temperature coefficient of 0.01%/oC. A multivibrator (A1) generates pulses of constant amplitude (V) and width (T2). Since the output (Terminal 6) of A1 (a CA5130) can swing within about 10mV of either supply-rail, the output pulse amplitude (V) is essentially equal to V+. The average output voltage (EAVG = V T2/T1) is applied to the non-inverting input terminal of comparator A2 (a CA5160) via an integrating network R3, C2. Comparator A2 operates to establish circuit conditions such that EAVG = V1. This circuit condition is accomplished by feeding an output signal from Terminal 6 of A2 through R4, D4 to the inverting terminal (Terminal 2) of A1, thereby adjusting the multivibrator interval, T3. 0.01µF 7 6 4 5 100kΩ 1 OFFSET ADJUST 2kΩ 0.1µF FIGURE 7A. SINGLE SUPPLY FOLLOWER Voltmeter With High Input Resistance 0 FIGURE 7B. OUTPUT SIGNAL WITH INPUT SIGNAL RAMPING 0 0 Top Trace: Output Bottom Trace: Input FIGURE 7C. OUTPUT-WAVEFORM WITH GROUNDREFERENCE SINE-WAVE INPUT FIGURE 7. SINGLE SUPPLY VOLTAGE FOLLOWER WITH ASSOCIATED WAVEFORMS (e.g., FOR USE IN SINGLE-SUPPLY D/A CONVERTER; SEE FIGURE 9 IN AN6080) 10 The voltmeter circuit shown in Figure 11 illustrates an application in which a number of the CA5160 characteristics are exploited. Range-switch SW1 is ganged between input and output circuitry to permit selection of the proper output voltage for feedback to Terminal 2 via 10kΩ current-limiting resistor. The circuit is powered by a single 8.4V mercury battery. With zero input signal, the circuit consumes somewhat less than 500µA plus the meter current required to indicate a given voltage. Thus, at full-scale input, the total supply current rises to slightly more than 1500µA. CA5160 10V LOGIC INPUTS +10.010V 14 LSB 9 8 7 6 3 10 MSB 11 6 5 4 3 2 1 6 3 10 6 3 10 2 CD4007A “SWITCHES” 9 13 7 8 CD4007A “SWITCHES” 1 12 13 5 806K 1% 4 8 402K 1% 200K 1% 806K 1% 1 +10.010V 8 22.1K 1% 6 7 + - REGULATED VOLTAGE ADJUST 3.83K 1% 4 2µF 25V 1 8 12 5 (2) 806K 1% 806K 1% (4) 806K 1% 750K 1% (8) 806K 1% +15V PARALLELED RESISTORS 1K 0.001µF BIT REQUIRED RATIOMATCH 1 Standard 2 ±0.1% 3 ±0.2% 4 ±0.4% 5 ±0.8% 6-9 ±1% ABS. 10K 7 3 + OUTPUT CA3085 3 13 806K 1% 62 2 12 5 100K 1% 806K 1% VOLTAGE +15V REGULATOR 1 CD4007A “SWITCHES” VOLTAGE FOLLOWER CA5160 6 - 4 2 5 LOAD 1 100K OFFSET NULL 2K 0.1µF FIGURE 8. 9 BIT DAC USING CMOS DIGITAL SWITCHES AND CA5160 2N6385 POWER DARLINGTON INPUT 40V + SHORT-CIRCUIT CURRENT LIMIT ADJUSTMENT 1Ω 3 2 OUTPUT 0V 35V AT 1A 10kΩ 0.2µF TURN ON DELAY 2.4kΩ 1W 1kΩ 1.5kΩ 1W 100kΩ 2N2102 1kΩ 1 1N914 56pF + 100µF - 43kΩ 8 2.2kΩ 100µF 25V + 7 + 5µF - CA3086 10 11 1 5 8 9 7 3 5 6 4 10kΩ 12 62kΩ 3 - 2 1 8.2 kΩ 4 14 4.7kΩ 13 1kΩ + 6 - 2 10kΩ 2kΩ 50kΩ 100kΩ 0.01µF - - Hum and Noise Output <250µVRMS; Regulation (No Load to Full Load) <0.005%; Input Regulation <0.01%/V FIGURE 9. CA5160 VOLTAGE REGULATOR CIRCUIT (0.1 TO 35V AT 1A) 11 CA5160 T2 T3 VCO CONTROL VOLTAGE (VI) (0V - 10V) V +15V fo (SENSITIVITY = 1kHz/V) T1 D1 1M 10K 0.01µF R5 100K +15V +15V D2 100K 0.1 µF 7 3 R6 100K C1 500pF 2 2 + A1 MULTIVIBRATOR CA5130 - EAVG = V T2/T1 6 R3 1M 3 4 R1 182K D4 R2 10K A2 COMPARATOR CA5160 + 6 4 5 C2 0.01µF D3 7 - 0.01µF 1 R7 100K D5 D1 - D5 = 1N914 R4 3K FIGURE 10. VOLTAGE CONTROLLED OSCILLATOR 300V 300V 100MΩ 100V 100V 30V 30V 10V BATTERY TEST OFF ON 1.02 MΩ 3 POSITION SLIDE SWITCH 9.9 kΩ 10V + - SW1A 3V INPUT 3V SW1B 1V 1V 300mV 300mV 100mV 100mV 30mV 30mV 10mV 10mV BATTERY +9V BATTERY 3 22MΩ 0.001 µF 7 + 2.7kΩ 2 - 4 5 1 300V 100V ZERO ADJUST 10kΩ 1V CAL. 10V 3V SW1C 3V 1V 300mV 9kΩ 300mV 100mV 900Ω 100mV 30mV 10mV 30mV 10mV 100Ω FIGURE 11. CA5160A HIGH INPUT RESISTANCE DC VOLTMETER 12 30V 10V 1V 9.1kΩ 300V 100V 820Ω 200Ω 30V 100kΩ M 0-1mA 3V CAL. 500Ω 6 CA5160 500 µF SW1D CA5160 8.2kΩ BUFFER VOLTAGE FOLLOWER 20pF +7.5V 0.9 - 7pF C1 VOLTAGE-CONTROLLED CURRENT SOURCE 7 3 + 6 CA3080A 1kΩ 2 - 4 1kΩ 2MΩ SYMMETRY -7.5V 100kΩ +7.5V 6.8MΩ 5 6.2kΩ 3 + 10kΩ 4 - 60pF CA5160 C3 2 - 6 4 -7.5V 10kΩ 6.2kΩ 500Ω FREQ ADJUST 500Ω CA3080 + 6 4 -7.5V EXTERNAL SWEEPING INPUT MIN. FREQ. SET 7.5V 7 2 3 0.1 µF 10kΩ -7.5V MAX FREQ SET +7.5V 30kΩ 0.1µF 7 -7.5V 4.7kΩ +7.5V 430pF 10-80pF C2 5 -7.5V +7.5V HIGH FREQ. SHAPE THRESHOLD DETECTOR CENTERING 100kΩ C4 4 - 60pF 2kΩ HIGH FREQ LEVEL ADJUST 50kΩ 2-1N914 C5 15 - 115pF FIGURE 12A. FUNCTION GENERATOR CIRCUIT NOTE: A square wave signal modulates the external sweeping input to produce 1Hz and 1MHz, showing the 1,000,000/1 frequency range of the function generator. NOTE: The bottom trace is the sweeping signal and the top trace is the actual generator output. The center trace displays the 1MHz signal via delayed oscilloscope triggering of the upper swept output signal. FIGURE 12B. TWO-TONE OUTPUT SIGNAL FROM THE FUNCTION GENERATOR FIGURE 12C. TRIPLE-TRACE OF THE FUNCTION GENERATOR SWEEPING TO 1MHz FIGURE 12. CA5160 1,000,000/1 SINGLE CONTROL FUNCTION GENERATOR - 1MHz TO 1Hz 13 CA5160 +15V 5.1kΩ +15V 100 kΩ 100 kΩ 1MΩ 3 100 kΩ STEP HEIGHT ADJUST 4 - 60pF 8.2kΩ 7 + CA5130 2 15 - 115pF FREQ ADJUST - 1N914 470pF STAIRCASE OUTPUT +15V 6 2 7 10kΩ CA5160 1N914 8 6 CA5130 2kΩ 4 CHARGE COMMUTATING NETWORK + 3 6 +15V + 3 4 MULTIVIBRATOR +15V 7 2 1.5 MΩ - 8 4 INTEGRATOR HYSTERESIS SWITCH MULTIVIBRATOR RETRACE INHIBIT +15mV TO +10V 51kΩ 100kΩ FIGURE 13A. STAIRCASE GENERATOR CIRCUIT Function Generator A function generator having a wide tuning range is shown in Figure 12. The adjustment range, in excess of 1,000,000/1, is accomplished by a single potentiometer. Three operational amplifiers are utilized: a CA5160 as a voltage follower, a CA3080 as a high-speed comparator, and a second CA3080A as a programmable current source. Three variable capacitors C1, C2, and C3 shape the triangular signal between 500kHz and 1MHz. Capacitors C4, C5, and the trimmer potentiometer in series with C5 maintain essentially constant (±10%) amplitude up to 1MHz. STAIRCASE OUTPUT 2V STEPS COMPARATOR OSCILLATOR Top Trace: Staircase Output 2V Steps Center Trace: Comparator Bottom Trace: Oscillator Staircase Generator Figure 13 shows a staircase generator circuit utilizing three CMOS operational amplifiers. Two CA5130s are used; one as a multivibrator, the other as a hysteresis switch. The third amplifier, a CA5160, is used as a linear staircase generator. FIGURE 13B. STAIRCASE GENERATOR WAVEFORM FIGURE 13. STAIRCASE GENERATOR CIRCUIT UTILIZING THREE CMOS OPERATIONAL AMPLIFIERS 10GΩ +15V 1MΩ 0.1µF 10pF +15V 7 10MΩ 3 7 + 6 CA5160 2 2 10kΩ - 4 5 CA3140 3 1 100kΩ 6 + 5.6kΩ 9.9kΩ 560kΩ 4 0.1µF 9.1kΩ 500Ω 100Ω -15V M 500-0-500µA -15V FIGURE 14. CURRENT-TO-VOLTAGE CONVERTER TO PROVIDE A PICOAMMETER WITH ±3pA FULL SCALE DEFLECTION 14 CA5160 100kΩ +15V +15V 2200pF 30pF 1MΩ 3 39kΩ + 6 CA5160 2 8.2Ω 7 2 7 - - 8 1N914 4 5 6 CA3080A 3 1 0.1µF 8.2kΩ 27kΩ 9.1kΩ 2 + 5 DROOP ZERO ADJUST CA3140 1MΩ 100kΩ 4 100kΩ OFFSET VOLTAGE ADJUST +15V 0.1 µF 0.1µF 7 3 6 + 0.1 µF 4 39kΩ 500µA 2kΩ STROBE INPUT SAMPLE - 15V HOLD - 0V FIGURE 15A. SAMPLE AND HOLD CIRCUIT SAMPLED OUTPUT SAMPLED OUTPUT 0VINPUT 0V- INPUT SIGNAL SAMPLING PULSES SAMPLING PULSE Top Trace: Sampled Output Center Trace: Input Signal Bottom Trace: Sampling Pulses FIGURE 15B. SAMPLE AND HOLD WAVEFORM Top Trace: Sampled Output Center Trace: Input Bottom Trace: Sampling Pulses FIGURE 15C. SAMPLE AND HOLD WAVEFORM FIGURE 15. SINGLE SUPPLY SAMPLE AND HOLD SYSTEM, INPUT 0V TO 10V Picoammeter Circuit Figure 14 is a current-to-voltage converter configuration utilizing a CA5160 and CA3140 to provide a picoampere meter for ±3pA full-scale meter deflection. By placing Terminals 2 and 4 of the CA5160 at ground potential, the CA5160 input is operated in the “guarded mode”. Under this operating condition, even slight leakage resistance present between Terminals 3 and 2 or between Terminals 3 and 4 would result in 0V across this leakage resistance, thus substantially reducing the leakage current. If the CA5160 is operated with the same voltage on input Terminals 3 and 2 as on Terminal 4, a further reduction in the input current to the less than 1pA level can be achieved as shown in Figure 1. 15 To further enhance the stability of this circuit, the CA5160 can be operated with its output (Terminal 6) near ground, thus markedly reducing the dissipation by reducing the supply current to the device. The CA3140 stage serves as a X100 gain stage to provide the required plus and minus output swing for the meter and feedback network. A 100-to-1 voltage divider network consisting of a 9.9kΩ resistor in series with a 100Ω resistor sets the voltage at the 10GΩ resistor (in series with Terminal 3) to ±30mV full-scale deflection. This 30mV signal results from ±3V appearing at the top of the voltage divider network which also drives the meter circuitry. By utilizing a switching technique in the meter circuit and in the 9.9kΩ and 100Ω network similar to that used in the voltmeter circuit shown in Figure 11, a current range of 3pA CA5160 to 1nA full scale can be handled with the single 10GΩ resistor. +15V R1 100kΩ Single Supply Sample-and-Hold System Figure 15 shows a single-supply sample-and-hold system using a CA5160 to provide a high input impedance and an input-voltage range of 0V to 10V. The output from the input buffer integrator network is coupled to a CA3080A. The CA3080A functions as a strobeable current source for the CA3140 output integrator and storage capacitor. The CA3140 was chosen because of its low output impedance and constant gain-bandwidth product. Pulse “droop” during the hold interval can be reduced to zero by adjusting the 100kΩ bias-voltage potentiometer on the positive input of the CA3140. This zero adjustment sets the CA3080A output voltage at its zero current position. In this sample-and-hold circuit it is essential that the amplifier bias current be reduced to zero to minimize output signal current during the hold mode. Even with 320mV at the amplifier bias circuit (Terminal 5) at least ±100pA of output current will be available. Wien Bridge Oscillator A simple, single-supply Wien Bridge oscillator using a CA5160 is shown in Figure 16. A pair of parallel-connected 1N914 diodes comprise the gain-setting network which standardizes the output voltage at approximately 1.1V. The 500Ω potentiometer is adjusted so that the oscillator will always start and the oscillation will be maintained. Increasing the amplitude of the voltage may lower the threshold level for starting and for sustaining the oscillation, but will introduce more distortion. +15V R3 51kΩ C2 51pF 7 3 OUTPUT f = 100kHz 2% THD AT 1.1VP-P + 6 CA5160 2 R2 100kΩ 0.1 µF 4 C1 10-80pF 2kΩ 2-1N914 0.01µF 680Ω f= 1 2 π √(R1 || R2) C1 R3 C2 500Ω FIGURE 16. SINGLE-SUPPLY WEIN-BRIDGE OSCILLATOR The current sourcing and sinking capability of the CA5160 output stage is easily supplemented to provide power-boost capability. In the circuit of Figure 17, three CMOS transistorpairs in a single CA3600 lC array are shown parallel-connected with the output stage in the CA5160. In the Class A mode of CA3600E shown, a typical device consumes 20mA of supply current at 15V operation. This arrangement boosts the currenthandling capability of the CA5160 output stage by about 2.5X. The amplifier circuit in Figure 17 employs feedback to establish a closed-loop gain of 20dB. The typical largesignal-bandwidth (-3dB) is 190kHz. Operation with Output-Stage Power-Booster +15V 14 0.01µF 1µF - 1MΩ + 3 680kΩ 2 QP2 11 QP3 + 6 CA5160 INPUT 1µF CA3600 (NOTE) QP1 7 2 - 8 2kΩ 13 1 3 10 500µF 6 12 4 8 QN1 A = 20dB LARGE SIGNAL BW (-3dB) = 190kHz 7 NOTE: See File Number 619. 50Ω 100mW AT 10% THD 5 QN2 4 QN3 9 20kΩ FIGURE 17. CMOS TRANSISTOR ARRAY (CA3600E) CONNECTED AS POWER BOOSTER IN THE OUTPUT STAGE OF THE CA5160. 16 CA5160 VS = ±7.5V TA = 25oC 100 0 50 100 φ OL 80 150 200 60 40 CL = 30pF RL = 2kΩ 20 150 RL = 2kΩ OPEN-LOOP VOLTAGE GAIN (dB) OPEN-LOOP VOLTAGE GAIN (dB) 120 OPEN-LOOP PHASE (DEGREES) Typical Performance Curves 102 103 104 105 106 FREQUENCY (Hz) 107 108 FIGURE 18. OPEN-LOOP VOLTAGE GAIN AND PHASE SHIFT vs FREQUENCY 120 110 100 90 15 17.5 SUPPLY VOLTAGE: V+ = 15V, V- = 0V TA = 25oC 15 LOAD RESISTANCE = 5kΩ 12.5 2kΩ 1kΩ 10 500Ω 7.5 5 2.5 0 0 2.5 5 7.5 10 12.5 15 17.5 20 -50 LOAD RESISTANCE = ∞ TA = 25oC OUTPUT VOLTAGE BALANCED = V+/2 V- = 0 12.5 10 7.5 5 OUTPUT VOLTAGE HIGH = V+ OR LOW = V2.5 6 8 GATE VOLTAGE (TERMINALS 4 AND 8) (V) 18 600 OUTPUT VOLTAGE = V+/2 V- = 0 V+ = 5V, V- = 0V TA = -55oC 10 25oC 8 125oC 6 4 2 525 SUPPLY CURRENT (µA) QUIESCENT SUPPLY CURRENT (mA) 10 12 14 16 POSITIVE SUPPLY VOLTAGE (V) FIGURE 21. QUIESCENT SUPPLY CURRENT vs SUPPLY VOLTAGE 14 12 100 0 22.5 FIGURE 20. VOLTAGE TRANSFER CHARACTERISTICS OF CMOS OUTPUT STAGE 0 50 TEMPERATURE (oC) FIGURE 19. OPEN-LOOP GAIN vs TEMPERATURE QUIESCENT SUPPLY CURRENT (mA) OUTPUT VOLTAGE [TERMS 4 AND 6] (V) 130 80 -100 0 101 140 450 125oC 375 25oC 300 -55oC 225 150 75 0 0 2 4 6 8 10 12 14 16 POSITIVE SUPPLY VOLTAGE (V) FIGURE 22. QUIESCENT SUPPLY CURRENT vs SUPPLY VOLTAGE 17 0 0 0.5 1 1.5 2 2.5 3 3.5 OUTPUT VOLTAGE (V) 4 4.5 FIGURE 23. SUPPLY CURRENT vs OUTPUT VOLTAGE 5 CA5160 Typical Performance Curves (Continued) 8 9 V+ = 5V, V- = 0V 8 7 OUTPUT VOLTAGE SWING (V) OUTPUT VOLTAGE SWING (V) V+ = 5V, V- = 0V 6 5 4 -55oC 3 25oC 125oC 2 7 6 5 4 3 2 1 1 0 1 2 3 4 5 6 7 8 LOAD RESISTANCE (kΩ) 9 10 0 0.1 0.2 11 FIGURE 24. OUTPUT VOLTAGE SWING vs LOAD RESISTANCE VOLTAGE DROP ACROSS PMOS OUTPUT STAGE TRANSISTOR (Q8) (V) OUTPUT CURRENT (mA) 7 6 5 4 SINK 3 2 0 -60 50 V- = 0V TA = 25oC 10 800 10V 15V V+ = 5V 1 0.1 -40 -20 0 20 40 60 80 0.001 0.001 100 120 140 TEMPERATURE (oC) 0.01 0.1 1 1000 INPUT NOISE VOLTAGE (nV/ √Hz) V+ = 15V 10V 5V 1 0.1 0.01 0.001 0.001 0.01 0.1 1 10 100 MAGNITUDE OF LOAD CURRENT (mA) FIGURE 28. VOLTAGE ACROSS NMOS OUTPUT TRANSISTOR (Q12) vs LOAD CURRENT 18 100 FIGURE 27. VOLTAGE ACROSS PMOS OUTPUT TRANSISTOR (Q8) vs LOAD CURRENT 50 V- = 0V TA = 25oC 10 MAGNITUDE OF LOAD CURRENT (mA) FIGURE 26. OUTPUT CURRENT vs TEMPERATURE VOLTAGE DROP ACROSS NMOS OUTPUT - STAGE TRANSISTOR (Q12) (V) 200 0.01 SOURCE 1 10 2 4 6 8 20 40 80 LOAD RESISTANCE (kΩ) FIGURE 25. OUTPUT SWING vs LOAD RESISTANCE 8 V+ = 5V, V- = 0V 0.6 1 TA = 25oC VS = ±7.5V 100 10 1 1 101 102 103 FREQUENCY (Hz) 104 105 FIGURE 29. EQUIVALENT NOISE VOLTAGE vs FREQUENCY CA5160 Small Outline Plastic Packages (SOIC) M8.15 (JEDEC MS-012-AA ISSUE C) 8 LEAD NARROW BODY SMALL OUTLINE PLASTIC PACKAGE N INDEX AREA 0.25(0.010) M H B M INCHES E SYMBOL -B1 2 3 L SEATING PLANE -A- h x 45o A D -C- e A1 B 0.25(0.010) M C 0.10(0.004) C A M B S MILLIMETERS MIN MAX NOTES A 0.0532 0.0688 1.35 1.75 - 0.0040 0.0098 0.10 0.25 - B 0.013 0.020 0.33 0.51 9 C 0.0075 0.0098 0.19 0.25 - D 0.1890 0.1968 4.80 5.00 3 E 0.1497 0.1574 3.80 4.00 4 0.050 BSC 1.27 BSC - H 0.2284 0.2440 5.80 6.20 - h 0.0099 0.0196 0.25 0.50 5 L 0.016 0.050 0.40 1.27 6 8o 0o N α NOTES: MAX A1 e µα MIN 8 0o 8 7 8o 1. Symbols are defined in the “MO Series Symbol List” in Section 2.2 of Publication Number 95. Rev. 0 12/93 2. Dimensioning and tolerancing per ANSI Y14.5M-1982. 3. Dimension “D” does not include mold flash, protrusions or gate burrs. Mold flash, protrusion and gate burrs shall not exceed 0.15mm (0.006 inch) per side. 4. Dimension “E” does not include interlead flash or protrusions. Interlead flash and protrusions shall not exceed 0.25mm (0.010 inch) per side. 5. The chamfer on the body is optional. If it is not present, a visual index feature must be located within the crosshatched area. 6. “L” is the length of terminal for soldering to a substrate. 7. “N” is the number of terminal positions. 8. Terminal numbers are shown for reference only. 9. The lead width “B”, as measured 0.36mm (0.014 inch) or greater above the seating plane, shall not exceed a maximum value of 0.61mm (0.024 inch). 10. Controlling dimension: MILLIMETER. Converted inch dimensions are not necessarily exact. All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems. Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries. For information regarding Intersil Corporation and its products, see www.intersil.com 19